CN110266615B - 低isi比低功率芯片间通信方法和装置 - Google Patents
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Abstract
本发明公开低ISI比低功率芯片间通信方法和装置,所述方法包括:接收向量信令码的码字的符号集合,每个符号接收自多线路总线的一条相应的线路;生成一组模拟输出,每个模拟输出形成为所述码字的所述符号集合中的相应的符号子集的线性结合,所述线性结合由接收相应的所述符号子集的多输入比较器形成,所述线性结合与所述向量信令码的多个相互正交的子信道代码向量中相应的子信道代码向量相关联,其中,所述符号子集中的符号数目对应于相应的所述子信道代码向量中的非零元素数目;以及通过所述一组模拟输出生成一组信息比特。
Description
本申请是申请号为201580006043.3,申请日为2015年2月2日,发明名称为“低ISI比低功率芯片间通信方法和装置”的专利申请的分案申请。
相关申请的交叉引用
本申请要求申请号为61/934,803,申请日为2014年2月2日,发明人为Ali Hormati和Amin Shokrollahi,名称为《利用ISI比评价代码的方法》的美国临时专利申请的优先权,并通过引用将其内容整体并入本文。
参考文献
以下参考文献通过引用整体并入本文,以供所有目的之用:
公开号为2011/0268225,申请号为12/784,414,申请日为2010年5月20日,发明人为Harm Cronie和Amin Shokrollahi,名称为《正交差分向量信令》的美国专利申请(下称“《Cronie I》”);
申请号为13/030,027,申请日为2011年2月17日,发明人为Harm Cronie、AminShokrollahi和Armin Tajalli,名称为《利用稀疏信令码进行抗噪声干扰、高引脚利用率、低功耗通讯的方法和***》的美国专利申请,下称“《Cronie II》”;
申请号为14/158,452,申请日为2014年1月17日,发明人为John Fox、BrianHolden、Peter Hunt、John D Keay、Amin Shokrollahi、Richard Simpson、Anant Singh、Andrew Kevin John Stewart和Giuseppe Surace,名称为《低SSO噪声芯片间通信方法和***》的美国专利申请,下称“《Fox I》”;
申请号为14/178,051,申请日为2014年2月11日,发明人为John Fox、BrianHolden、Peter Hunt、John D Keay、Amin Shokrollahi、Richard Simpson、Andrew KevinJohn Stewart、Giuseppe Surace和Roger Ulrich,名称为《高带宽芯片间通信接口方法和***》的美国专利申请,下称“《Fox II》”;
申请号为13/842,740,申请日为2013年3月15日,发明人为Brian Holden、AminShokrollahi和Anant Singh,名称为《芯片间通信用向量信令码中偏斜耐受方法以及用于芯片间通信用向量信令码的高级检测器》的美国专利申请,下称“《Holden I》”;
申请号为61/839,360,申请日为2013年6月23日,发明人为Amin Shokrollahi,名称为《低接收器复杂度向量信令》的美国临时专利申请,下称“《Shokrollahi I》”;
申请号为61/839,360,申请日为2013年6月23日,发明人为Amin Shokrollahi,名称为《低接收器复杂度向量信令码》的美国临时专利申请,下称“《Shokrollahi II》”;
申请号为61/946,574,申请日为2014年2月28日,发明人为Amin Shokrollahi、Brian Holden和Richard Simpson,名称为《时钟嵌入式向量信令码》的美国临时专利申请,下称“《Shokrollahi III》”;
申请号为62/015,172,申请日为2014年7月10日,发明人为Amin Shokrollahi和Roger Ulrich,名称为《信噪比特性改进型向量信令码》的美国临时专利申请,下称“《Shokrollahi IV》”;
申请号为13/895,206,申请日为2013年5月15日,发明人为Roger Ulrich和PeterHunt,名称为《利用差和高效检测芯片间通信用向量信令码的电路》的美国专利申请,下称“《Ulrich I》”;
申请号为62/026,860,申请日为2014年7月21日,发明人为Roger Ulrich和AminShokrollahi,名称为《总线可逆型正交差分向量信令码》的美国临时专利申请,下称“《Ulrich II》”;
申请号为14/315,306,申请日为2014年6月25日,发明人为Roger Ulrich,名称为《高速芯片间通信用多级驱动器》的美国专利申请,下称“《Ulrich III》”。
此外,本申请中还引用了以下现有技术参考文献:
专利号为7053802,申请日为2004年4月22日,授权日为2006年5月30日,发明人为William Cornelius,名称为《带内嵌定时功能的单端平衡编码型接口》的美国专利,下称“《Cornelius》”;
专利号为8649460,申请日为2010年3月11日,授权日为2014年2月11日,发明人为Frederick Ware和Jade Kizer,名称为《利用嵌入式时钟进行多线路编码的技术》的美国专利,下称“《Ware》”。
技术领域
本发明总体涉及通信领域,尤其涉及可在器件内部或器件之间传送信息的信号的发送和接收。
背景技术
通信***中的一个目的在于将信息从一个物理位置传输至另一物理位置。一般而言,此类信息传输的目标在于可靠、快速且消耗最少的资源。一种常见的信息传输媒介为串行通信链路,此种链路可基于将地面或其他常用基准作为比较对象的单个有线电路或将地面或其他常用基准作为比较对象的多个此类有线电路。常见的一例为使用单端信令(Singled-ended Signaling,SES)。SES的工作原理为,在一条线路中发送信号,然后在接收器端以固定基准值为比较对象测定所述信号。串行通信链路也可基于相互间作为比较对象的多个电路。此方面的常见的一例为使用差分信令(Differential Signaling,DS)。差分信令的工作原理在于,在一条线路中发送信号,并在配对线路中发送所述信号的相反信号。所述信号的信息由上述两线路之间的差值,而非其相对于地面或其他固定基准值的绝对值表示。
与DS相比,有多种信令方法可在保持相同有益特性的同时,增加引脚利用率。向量信令为一种信令方法。通过向量信令,多条线路中的多个信号在保持每个信号的独立性的同时可视为一个整体。该信号整体中的每个信号均称为向量分量,而所述多条线路的数目称为向量“维数”。此外,在一些实施方式中,与差分信令对的情况相同,一条线路中的信号完全取决于另一线路中的信号。因此,在某些情况下,向量维数可指多条线路内的信号的自由度数,而非该多条线路的数目。
向量信令码的任何合适子集均为该码的“子码”。此类子码可本身为一种向量信令码。在二元向量信令中,每个向量分量(或称“符号”)的取值为两个可能取值当中的一值。在非二元向量信令中,每个符号的取值为从由两个以上可能取值所组成的集合中选出的一值。由能表示所有符号的所有取值组成的集合称为代码的“符集”。因此,举例而言,二元向量信令码至少需要一个由两个取值组成的符集,而三元向量信令码至少需要一个由三个取值组成的符集。当作为物理信号在通信介质中传输时,符号可由适合于该介质的具体物理值表示。例如,在一种实施方式中,可由150mV的电压表示符号“+1”,50mV的电压表示符号“-1”;而在另一实施方式中,“+1”可由800mV表示,“-1”可由-800mV表示。
在本文中,向量信令码为由具有相同长度N的向量(称作码字)组成的集合C。集合C的大小的二进制对数与长度N之间的比值称为该向量信令码的引脚利用率。向量信令码示例见《Cronie I》、《Cronie II》、《Fox I》、《Shokrollahi I》、《Shokrollahi II》及《Shokrollahi III》中的正交差分向量信令(Orthogonal Differential VectorSignaling,ODVS)码,该码在本文中用于描述目的。
符号间干扰(Inter-symbol Interference,ISI)是指***之前所发符号的余效所导致的接收器待解码符号的失真。此现象主要因底层通信信道的特性所致,而且经常成为妨碍通信速度提升或通信差错降低的限制因素。众所周知,易受ISI影响的信道例如包括受多径干扰影响的无线通信,以及有线***中的有限带宽信道。由于ISI一定会导致性能退化,因此在试图解码当前符号以获得其内嵌信息时,最好消除其影响。在某些情况下,一系列之前所发符号可发生增益性结合,从而最大程度地影响当前符号的检测余量。由于这些最差状况下符号模式的发生概率通常高于***的目标差错率,因此其所产生的影响以及如何量化和最小化其负面行为成为通信***设计中的主要关注点。在这种具体问题具体解决式的问题模式识别方式之外,没有一种可靠的指标能对ISI对通信***性能的影响进行评估,或能就减轻此类影响作出如何修改信道或编码的指示。
解决ISI问题的一种方式为使用可使等效信道免受ISI影响的均衡器。均衡器为功能处理模块或电路,用于将信道反转,从而使得每个符号间隔内所发送的数据与经***所发的其他符号(完全)无关。在串行器/解串器(Serializer-Deserializer,SerDes)设计中,有限脉冲响应滤波(Finite Impulse Response filtering,FIR)和连续时间线性均衡(Continuous Time Linear Equalization,CTLE)为两种知名的线性均衡方法,分别用于***的发射器端和接收器端,而判定反馈均衡(Decision Feedback Equalization,DFE)为用于接收器端的非线性均衡方法。此外,汤姆林森-原岛预编码(Tomlinson-HarshimaPrecoding)等其他均衡方法也同样为本领域从业者所知。此类预编码通常相当于在发射器端实施均衡。另一方面,均衡器在实现复杂度、功耗以及校准要求层面上,尤其对于数千兆比特/秒的通信***而言,较为昂贵。因此,既需要能准确反映ISI对通信***性能影响的指标,也需要能以高效率、高性能方式降低ISI效应的信道处理方案。
发明内容
此处描述一种高效的通信装置,该装置用于向量信令码,以在集成电路器件之间传输数据以及可选传输时序信号。此外,还描述了根据本文中称为“ISI比”的新指标设计上述装置及其关联代码的方法,该方法可提高通信速度,减小***功耗,降低实现复杂度。
附图说明
图1所示为本发明通信***的一种实施方式。
图2所示为在2条线路上以每符号发送1个比特的差分信令的情形。
图3所示为在2条线路上以每符号发送2个比特的PAM-4信令的情形。
图4所示为本申请中模拟所用例示信道的脉冲响应。
图5为P3接收器的第一情形下多输入比较器实施方式的眼图。
图6为P3接收器的第二种情形下多输入比较器实施方式的眼图。
图7A~7E为5b6w码接收器的5个子信道的眼图。
图8A~8C为ENRZ码接收器的3个子信道的眼图。
图9A和图9B为《Shokrollahi I》所述4.5b5w码的眼图。
图10A和图10B为8b8w码的眼图。
图11为NRZ信令、PAM-4信令和ENRZ信令在相同信道模型下的性能比较图。
图12为作为上述信道常用信令方法的一例的NRZ眼图。
图13为作为上述信道常用信令方法的一例的PAM-4眼图。
图14为Glasswing接收器的框图。
图15为嵌入式时钟恢复型Glasswing接收器的框图。
图16为带式3矩阵第6行所定义输入权重的多输入比较器的一种实施方式的带均衡功能和不带均衡功能的两版本电路图。
图17为带式3矩阵第6行所定义输入权重的多输入比较器的另一实施方式的带均衡功能和不带均衡功能的两版本电路图。
图18为带式3矩阵第3行所定义输入权重的多输入比较器的一种实施方式的带均衡功能和不带均衡功能的两版本电路图。在适当替换线路输入端后,其还可用于表现式3矩阵第5行的比较器。
图19为带式3矩阵第3行所定义输入权重的多输入比较器的另一实施方式的带均衡功能和不带均衡功能的两版本电路图。在将适当替换线路输入端后,其还可用于表现式3矩阵第5行的比较器。
图20为带式3矩阵第2行所定义输入权重的多输入比较器的一种实施方式的带均衡功能和不带均衡功能的两版本电路图。在将适当替换线路输入端后,其还可用于表现式3矩阵第4行的比较器。
图21为Glasswing5b6w发送驱动器的一种实施方式的框图。
图22为Glasswing5b6w_10_5发送驱动器的一种实施方式的框图。
图23为发送方法流程图。
图24为接收方法流程图。
具体实施方式
图1所示为本发明通信***的一种使用向量信令码的实施方式。输入至发射器110的源数据图示为S0,S1,S2,S3,S4,并以源数据字100的形式进入编码器112。源数据字的大小可变,且取决于所述向量信令码的参数。编码器112生成所述向量信令码的码字,所述***为针对该向量信令码设计的***。运行过程中,编码器112所生成的所述码字用于控制驱动器118内的PMOS和NMOS晶体管,从而在通信信道120的N条通信线路125当中的每条线路中生成2个、3个或更多个不同电压或电流,以代表所述码字的N个符号。在图1实施方式中,所述源数据字的大小示为5个比特,而且所述码字大小为6个符号。因此,通信信道110示为由6条信号线路125组成,每条信号线路传输一个码字符号。熟悉编码领域的人员还可将此代码描述为:分组码长为6(即生成六符号输出字),代码大小为32(即具有32个不同码字,足以编码5个二进制比特的数据)。
在通***130内,检测器132读取线路125中的电压或电流,此过程有可能涉及放大、频率补偿和共模信号消除。在本例中,接收结果140图示为R0,R1,R2,R3,R4,且直接由检测器132提供,无需涉及可选解码器138。
容易理解的是,不同代码可具有不同的关联分组码长及不同的关联码字大小。为了描述方便,且不构成任何限制,图1例示为一种使用正交差分向量信令码的***,该代码可编码在6条线路中传输的5个二进制比特值,即所谓的5b6w码。
根据所使用向量信令码的不同,可不设置解码器,或者不设置编码器,或者既不设置解码器也不设置编码器。举例而言,对于《Cronie II》中所公开的8b8w码,既设置有编码器112,也同时设置有解码器138。另一方面,对于本例中的5b6w码,由于上述***可设置为由检测器132直接生成所述接收结果140,因此无需明确设置解码器。
为了保证通信***能正常工作,必须使得通信发射器110和通***130的运行完全同步化。在一些实施方式中,此同步化功能由发射器和接收器共享的外部时钟完成。在其他实施方式中,可像众所周知的串行通信用双相编码或本文所述的其他方法中的那样,将所述时钟功能与上述数据信道中的一条或多条信道相结合。
通信***中的信令
信令为一种利用通信信道发送信息的方法。任何将信息(通常表示为比特)映射至通信信道所载物理量的形式均称为信令,在优选实施方式中,通过成形脉冲对待经通信信道传输的信息进行调制,其中,所述成形脉冲具有有限支撑集的实值函数。传输于线路中的信号形式如下:
其中,Ci为长度为N(与线路数目相等)的码字向量,T为代表一个传输值的单位脉冲间隔的时间长度,P为在间隔[0,T]上定义了调制用脉冲形状的函数。码字Ci决定了任何时点所发送的信息。对Ci所属集合的选择以及对信息比特与Ci间映射方式的选择决定了信令方法。
脉冲振幅调制(Pulse Amplitude Modulation,PAM)为一种信令方法,在该方法中,Ci可取值为表示相应信号范围内等长振幅区间的值[-1,-1+2/(X-1),-1+4/(X-1),....,1-4/(X-1),1-2/(X-1),1]中的一个。此类型信令称为PAM-X信令。通常(但不绝对),X为2的乘方,每个Ci承载log2(X)(X的二进制对数)个信息比特。在许多高速SerDes应用中,X=2,尽管各种标准机构还提出了X=4、X=8或X=16等值。
对于本领域技术人员而言众所周知的是,虽然PAM-X信令可实现更高的传输比特/单位间隔比,但比PAM-2更易受ISI的影响。这一方面的原因通常被错误地归咎于如下事实:PAM-X信令较可能具有更多过渡区间(即信号振幅变化),而遍历这些区间需要不同长度的时间,从而导致更大的ISI。基于这一错误前提,在此之前的建议通常为,任何使用符集大小大于2的信令方法将遭受比PAM-2更大的ISI。与此相反,在下文中,我们证明了符集大小并不直接影响ISI。
在本文中,向量信令码既为由被称为码字的具有相同维数N的向量组成的集合C;也为由被称为多输入比较器(Multi-Input Comparator,MIC)的具有相同维数N的向量组成的第二集合Λ;还为一个“弃用对”集合,每一弃用对均为一个(Ci,λ)对,其中,Ci为集合C的一个元素,λ为集合Λ的一个元素。运行过程中,集合C各元素的坐标为有界值,而且我们选择由-1和1之间的实数对其进行表示。
运行过程中,一个码字由该码字与具有MIC的所有(Ci,λ)非弃用对的λ的标积所组成的向量唯一确定。如果一个(Ci,λ)对为非弃用对,则称其中的Ci对于MIC的该λ而言可用。
以下给出一例:码字集合C由对向量(1,0,0,-1)进行排列组合后获得的12个向量组成,MIC为以下的6个向量:
(1,-1,0,0),(1,0,-1,0),(1,0,0,-1),(0,1,-1,0),(0,1,0,-1),(0,0,1,-1),
而弃用对集合为各码字及多输入比较器对当中多输入比较器与码字的标积为零的对。换句话说,弃用对为以下各对:
((1,0,0,-1),(0,1,-1,0)),((1,0,-1,0),(0,1,0,-1)),((1,-1,0,0),(0,0,1,-1)),((0,1,-1,0),(1,0,0,-1)),
((0,1,0,-1),(1,0,-1,0)),((0,0,1,-1),(1,-1,0,0)),((0,0,-1,1),(1,-1,0,0)),((0,-1,1,0),(1,0,0,-1)),
((0,-1,0,1),(1,0,-1,0)),((-1,0,0,1),(0,1,-1,0)),((-1,0,1,0),(0,1,0,-1)),((-1,1,0,0),(0,0,1,-1))。
在下文中,我们将MIC互换性地理解定义为:一个向量;以及与该向量正交的所有点的集合所给出的超平面。
本文以信令视角对ISI问题进行检视,以图确定如何能以对***内的未校正残留ISI较不敏感的方式传送数据。在通过将信号值与固定基准值相比较,或者将信号值相互之间比较的方式测定信号的情形中,已证明对ISI的敏感性主要由从上述比较器的输出端所观察到的信号电平,而并非由从线路本身所观察到的信号电平确定。此处,我们引入ISI比的概念,该比为一种可帮助我们将ISI噪声的影响量化为所使用信令方法和检测方法的函数的指标。以下,我们将使用此概念设计抗ISI噪声的向量信令方法,并通过模拟结果对这一做法进行证实。
使用多输入比较器的接收器
如《Holden I》所述,系数为a0,a1,...,am-1的多输入比较器(MIC)的一种实际实施方式为一种求和电路,该电路将从多条信号导线中接收的向量(x0,x1,...,xm-1)作为其输入,并输出:
结果=(a0*x0+...+am-1*xm-1) (式2)
其中,(x0,...,xm-1)为所述求和电路的信号权重向量。在许多实施方式中,所需的输出为二进制值,因此需利用模拟比较器或其他此类信号分割电路对上述结果值进行分割,以生成二进制判定输出。由于该情形为一种常见用途,因此上述电路的俗语名称中包含了“比较器”一词。但是,在其他实施方式中,所述求和结果也可施加至PAM-3或PAM-4分割器,以获得三进制或四进制输出;也可实际保留式2的模拟输出,以用于进一步计算。
以矩阵表示法描述的接收器
在数学上,可利用矩阵表示法简洁地描述上述包含代码接收器的一组多输入比较器,其中,矩阵各列对应于输入向量(x0,x1,...,xm-1)的各相继元素,即承载向量信令码的多条信号导线或线路输入;矩阵的每一行对应于定义了一个特定多输入比较器及其输出的向量。在此表示法中,矩阵元素的值对应于由相应行的多输入比较器施加至相应列的输入值上的一个权重向量或一组比例因子。
式3矩阵描述了此类包含代码接收器的一组多输入比较器。
在此实施方式中,矩阵的6个列所代表的6条输入线路由矩阵的行2~6所代表的5个多输入比较器处理。为了后文中所述目的,第一矩阵行全部由值“1”组成,从而生成6×6的方阵。
在本文中,如式3所示,当一个矩阵M符合MT M=D时,称为“正交”矩阵,即该矩阵与其转置矩阵的积为一个仅对角线上具有非零值的对角矩阵。这一定义比常用定义较弱,常用定义要求上述结果为单位矩阵,即对角线上的值等于1。虽然矩阵M也可归一化为符合较强的常用正交性要求的形式,但正如下文所述,在实际应用中,这一归一化过程既不必要也不是期望的。
在功能方面,正交性要求为:每一代表多输入比较器的权重向量行与所有其他行正交,而且每一代表多输入比较器的行的和为零。由于这表示各比较器的输入也相互正交(因而相互独立),因此其可代表不同的通信模,本文称为向量信令码通信***的“子信道”。
基于以上关于模的解释,上述矩阵的首行可视为代表传输介质中的共模通信信道。由于实际***中希望接收器具有共模抑制作用,因此上述第一行的值均设为“1”,从而使得每条线路输出对此矩阵行的共模贡献最大化。由于所有矩阵行根据定义为相互正交,因此其他矩阵行(即接收器输出)可均不受共模信号的影响。具有此共模抑制作用的实施方式无需实施与其描述矩阵第一行相对应的物理比较器。
为避免混淆,需要注意的是,在正交差分向量信令***内,包括表示子信道内所载信号的状态转换在内的数据通信均为实施于整个信道内的码字通信。如本文、《Holden I》及《Ulrich I》所述,在实施方式中,可将特定的输入值与码字间的映射相互关联,或者将这些映射与具体检测器结果相关联,但是不应将此类关联做法与物理通信介质本身的分区、分割或子信道相混淆。同样,正交差分向量信令子信道的概念并不因例示实施方式而局限于特定的正交差分向量信令码、发射器实施方式或接收器实施方式。此外,用于保持内部状态的编码器和/或解码器也可作为本发明实施方式的部件。子信道既可由各单独信号表示,也可由多个信号所传达的状态来表示。
《Shokrollahi II》中描述了可按本文所述方式使用的正交矩阵的构建方法。
生成与接收器矩阵对应的ODVS码
如《Cronie I》和《Cronie II》中所述,正交差分向量信令码可通过将(0,a1,a2,...,an)形式的输入调制向量与矩阵M相乘的方式构建自生成矩阵。在最简单的情况下,此向量的每个ai均为表示一比特传输信息的单个值的正数或负数,如±1。
通过将矩阵M理解为描述***的各种通信模,可容易地看出,在将该矩阵与此类输入向量相乘的过程中,各模均受到所述向量的ai的扰动,而与共模传输对应的第零模根本不受影响。对于熟悉本领域的技术人员而言容易理解的是,在大多数实施方式中,共模传输中消耗的能量是一种不必要的浪费。然而,在本发明的至少一种实施方式中,共模项的非零振幅用于在整个通信信道内提供非零偏移或基线值。
还可以看出的是,利用此方法所生成的代码的各个码字表示各正交通信模的线性结合。在不施加额外约束的情况下(例如为了实施方便的目的),此方法实现了可在N条线路上以N-1个不同子信道通信的***,该***通常实施为N-1比特/N线***。表示编码值所需的各码字值的集合称为代码的符集,而各符集值的数目称为符集大小。
作为另一例,表1所示为通过上述方法自式3矩阵生成的代码。
表1
容易看出,此码的符集由值+1,+1/3,-1/3,-1组成,因此该码为四元码(即符集大小为四)。下文中将此码描述为5b6w码或“Glasswing”码,并将其相应的式3所示接收矩阵描述为“Glasswing接收器”。
子信道内的定时信息
由于ODVS通信***必须以编码传输的形式传输每一数据输入组合,而且该编码传输的速度必须受通信介质容量的制约,因此待传输数据的变化率必须处于奈奎斯特(Nyquist)限之内,其中,码字的传输速率表示采样间隔。作为一例,当二进制时钟信号或选通信号在每次码字传输中具有不多于一个时钟边沿时,其可通过ODVS子信道传输。
在一种实施方式中,当数据输入发生任何变化时,ODVS编码器及其相应线路驱动器可以非同步方式运行。在其他实施方式中,可例如采用内部定时时钟合并多个数据处理阶段,以生成单个高速输出流。在这些实施方式中,所***字元素以本质同步的方式输出(不存在任何逻辑延迟或其他实施制约),因此,在接收器端看来,代码子信道上传输的时钟信号或选通信号将成为数据对齐时钟(例如,其过渡边沿与同一代码的其他子信道上传输的数据边沿同步)。通过本领域公知的方法,可将此类数据对齐时钟转化为适于通过结合本发明而启动数据子信道采样的延迟时钟或眼中心对齐时钟。此类方法可涉及引入固定延时、可调延时、延迟锁定环等。
ISI比
ISI比为信令方案对通信***内符号间干扰的敏感度的度量值。在某种意义上,ISI比也为信令方案因残余未校正ISI而发生的恶化的程度的度量值。
为了规范化上述定义,假设我们现有N条通信线路。在每个单位间隔(UnitInterval,UI)内,编码器从K个长度为N的可能码字中拣取一个码字(根据待发送比特),而且驱动器生成与所述码字的坐标值成正比的电压/电流,并利用这些值对线路进行驱动。在此例中,虽然并不是为了施加任何限制,但是我们假设每条线路上的值使用矩形脉冲形状Pt。之后,所述码字沿信道向接收器传输。在接收器端,线路上的值可经过均衡器,并可能经过接收器前端的增益级,所得新值在比较器网络内(可能)以多种线性方式相结合,从而生成一组判定值。如上所述,各比较器可先求得线路值的线性结合值,然后将这些值与固定基准值比较,或与线路值的其他线性结合值比较。所述接收器的比较器网络具有M个MIC。这些MIC可简捷地表述为上述线路值的“仿射线性形式”。换句话说,每个MIC均与一组系数a0,a1,...,an相关联。运行过程中,MIC的输出为表达式a0+a1x1+...+anxn的符号,其中,x1,...,xn分别为线路1,...,N的值,-a0为相应基准值。如果MIC的基准值为零,则称该MIC为“中心”MIC。否则,该MIC称为非中心MIC。通常,中心MIC也记为上述系数向量的线性形式,即[a0,a1,...,an]。中心MIC可分为两种类型:无参照MIC;及有参照MIC。无参照MIC为系数之和为零的中心MIC。
举例而言,在标准差分信令中,MIC系数为简单的[+1,-1]对,且其结果与零相比较。因此,该MIC为中心无参照MIC。在PAM-4信令中,虽使用相同的MIC,但是在所使用的三个MIC当中,两个为有参照MIC。换句话说,如果最大信号值为1,最小信号值为-1,则所述各MIC由以下三个仿射线性形式表示:
一般而言,MIC的某些系数可以为零,在此情况下,将其对应的输入舍弃即可。
有时,在几何学意义上描述MIC比较有利,其中MIC被视为N维欧几里得空间中的超平面。如此,可使用针对该MIC求值得零的所有向量所组成的向量组确定该MIC。例如,对于基准值为0的MIC[1,-1],上述超平面恰为二维欧几里得平面内的直线x=y。下文中,我们经常互换性地利用一个MIC的相应向量或其关联超平面确定该MIC。在几何学意义上,当MIC的超平面经过原点时,则该MIC为中心MIC;否则,该MIC为非中心MIC。
上述代码的K个码字中的每个码字均可视作N维欧几里得空间内的点。所述MIC的目的在于区分这些点。当两个码字的对应点位于与某个MIC相对应的超平面的相对两侧,且当该MIC对于所述两个码字而言为非弃用对时,则该MIC可将所述两个码字相互区分。
图2和图3所示为在两条线路上分别以每符号发送1个比特和2个比特的差分信令和PAM-4信令的情形。在差分信令的情形中,如图2所示,超平面为一条经过原点的45度角直线,而码字为(1,-1)和(-1,1)坐标点。此两码字由所述超平面相互分离,而且距该超平面的距离相等。对于图3所示PAM-4信令,超平面为:一条经过原点的45度角直线;以及由向量(2/3,-2/3)和(-2/3,2/3)将此线平移后形成的两条平移线。此信令的码字为点(1,-1),(1/3,-1/3),(-1/3,1/3)和(-1,1),这些点均位于垂直于各MIC的直线上。可以看出,各MIC可将所述各码字相互区分。
以下,我们将对基于上述ISI比概念设计的新型信令方法进行展示。所述ISI比的定义对象为MICλ以及码字集合C。MIC m的可用码字索引集合(即对于该多输入比较器而言,其并非为弃用对)表示为Δm。为了精确定义ISI比,我们需要分别区分中心MIC(相应超平面通过原点)和非中心MIC(与上述PAM-4信令中的两个MIC相同,其相应超平面不通过原点)。
中心MIC的ISI比的计算
令d(Ck,MICm)表示码字Ck与MICm的对应超平面之间的距离,则MICm的ISI比定义为:
在几何学意义上,ISI比可视作任何可用码字到MIC超平面的最大距离与任何可用码字到MIC超平面的最小距离的比值。当计算上述各信令实施例的ISI比时,可得如下各数:
差分信令:
MIC的ISI比为1。
PAM-4信令:
中间MIC(即中心MIC)的ISI比为3。
通过考察图2中的差分信令情形可对上述值进行验证,其中,两个码字点与MIC平面距离相等,表明最大距离/最小距离之比为1。在图3所示PAM-4信令中,可看出最远码字点与最近码字点距中心MIC平面的距离为3:1。
以下,将描述根据ISI比解决信令设计问题的其他实施例。
非中心MIC的ISI比的计算
令d(Ck,MICm)表示码字Ck与有参照MICm的对应超平面之间的距离,d0(Ck,MICm)表示码字Ck与过原点的MICm偏移后的形式之间的距离,则非中心MICm的ISI比定义为:
根据此式可知,PAM-4信令的两个非中心MIC的ISI比也等于3。由此可知,PAM-4通信***内的所有MIC的ISI比均为3。
ISI比和水平眼开度
上述k个码字Ck k=1,...,K中的每个码字均在MICm的输出端产生其自身的脉冲响应,表示为:
当假设***内串扰可忽略且信道hi,i(t)均相等(=h(t)),则码字K在MICm输出端产生的脉冲响应表示为:
其中,Ph(t)=P(t)*h(t)为***(包括所有线性均衡器)的脉冲响应,MICm(i)为第m个MIC的第i个坐标,Ck(i)为码字Ck的第i个坐标。假设所述MIC为中心MIC且其系数归一化至平方和等于1,则表示码字K至MICm所对应的中心超平面之间的距离。需要注意的是,在MIC与任意非零实数做标量乘法时,ISI比为不变量。因此,可以认为,代表所述MIC的向量的范数为1(即其系数的平方和为1)。
MICm的输出信号为时移量为UI长度(=T)的整数倍的任选码字的脉冲响应Pm,k(t)的叠加。假设t0为码字C0的参考采样时间,则在时间t0处可得:
对于中心MIC,接收器在该MIC输出端于采样时间t内眼闭,在该时间内,rm(t)相对于其在t0时的符号发生符号转变。这决定了所述MIC输出端的水平眼开度。由于许多通信***的目标在于实现接收器的极低检测误差率(如小于10-12),因此其眼开度由最差状况下的码字模式决定。最差状况下的码字模式对应于离MICm超平面最近的待检测码字(我们称之为Cmin(m)),而离该超平面距离最远的码字为前达和后达码字(我们称之为Cmax(m))。当选择这些码字时,我们可得:
通过改变采样阶段t0,可使得一旦下式变更符号,即发生眼闭:
Ph(t0)-ISIRatio*∑kPh(t0-kT) (式11)
信令方案的ISI比越大,我们从相邻码字中观察到的残余ISI就越大,且眼闭发生得越快。需注意的是,对于给定脉冲响应Ph(t)(由信道、均衡器、发射器脉冲形状和***波特率决定),水平眼开度仅取决于底层信令方案的ISI比。ISI比越大,水平眼开度越小。
对于非中心信令的情形,水平眼开度由距ISI基准值的距离ref(m)决定。相应地,式8改为:
该式同样表明,水平眼开度仅由底层MIC的ISI比决定。
基于ISI比指标的信令设计
基于上述讨论,我们可根据ISI比概念设计信令方案。为了描述目的,但不构成任何限制,下文以有线传输***领域的实施例进行描述。
作为第一实施例,我们希望在信道中传输b个比特的信息,这要求我们的信令方案至少具有2b个码字。在我们的设计中,还涉及以下考虑因素:
·维数N等于信道中的线路数(即各通信元素或子信道相互独立)
·由于发射器中电压摆动的限制,所***字坐标处于[-1,1]范围内。
此设计的目标在于找出码字、MIC以及弃用对,从而使得所述MIC可区分各个码字,并使得所述MIC的最大ISI比最小化。如上节所述,ISI最大的MIC为决定性MIC,而且其相应的水平眼开度决定了整个通信***的误码率。
在以下所述实施例中,使用四线传输线路,而且信道特性以TE(www.te.com)提供的基准通道为基础,该信道为针对IEEE 802.3bj标准(在线网址:www-dot-ieee802.org/3/100GCU/public/channel.html)测量的两线背板信道。所述信道的响应扩展至消除了线路间串扰的四线信道。波特率固定于8G波特每秒。此外,还使用均衡范围为0dB~12dB的连续时间线性均衡器(CTLE),以及设于发射器端的3阶滤波器,其中均衡器的设置已优化至可获得最佳水平眼开度。所得信道的脉冲响应如图4所示。
作为常用信令方法在上述信道的应用示例,图12所示为非归零(NRZ)信令的眼图(吞吐量=4Gbps/线,水平眼开度=106.2皮秒),图13所示为PAM-4信令的眼图(吞吐量=8Gbps/线,水平眼开度=40.0皮秒)。
第一向量信令码设计实施方式称为P3信令,且使用(1,0,-1),(-1,0,1),(0,1,-1)和(0,-1,1)四个码字在上述示例信道的3条线路上发送2个比特。我们对以下两种情形进行考量:
情形1:我们将MIC选为(1,-1,0)和(0,1,-1)。这两个MIC在上述四个码字上的输出分别为值1,-1,-1,1和1,-1,2,-2。通过检验便可容易地发现,这两个MIC可将所述四个码字相互区分。第一MIC的ISI比为1(水平眼开度=106.2皮秒),第二MIC的ISI比为2(水平眼开度=60皮秒)。吞吐量为5.33Gbps/线。相应眼图如图5所示。
情形2:我们将MIC选为(1,-1,0)和(1/2,1/2,-1)。第二MIC的输出变为3/2,-3/2,3/2,-3/2。虽然两个MIC仍能区分所述码字,但是两MIC的ISI比均为1(两MIC的水平眼开度=106.2皮秒)。吞吐量为5.33Gbps/线。相应眼图如图6所示。
此实施例证明,在相同时钟频率下在相同信道中具有相同ISI比的MIC产生近乎相同的水平眼开度,而且选择正确的MIC对于传输***的信号完整性而言至关重要。
第二信令设计实施方式使用线性编码,也就是说,该信令方案由矩阵向量积定义。所述向量包含待传输的比特,而所述矩阵定义了将所述比特转化为码字的转化方式。在此方案中,原则上可在N条线路中发送N-1个比特。代表所述各比特的向量形式为(0,±1,±1,...,±1)。编码矩阵的第一行全部为1,而其他N-1行全与第一行正交,并跨越剩余的(N-1)维子空间。由此可见,此矩阵为本文中上述的正交矩阵。各MIC系数即为所述编码矩阵的逆矩阵的相应N-1个列。
基于正交矩阵的线性编码方案的一个重要特性在于,所有MIC的输出仅为±1形式,因此其ISI比一定为1。可以看出,这一特性对于使用此类代码的通信***的性能具有深远的影响。
在另一实施方式中,我们可使用上述P3码,并构建以下Glasswing编码矩阵,以在6条线路中发送5个比特:
S·A=w
S=信息比特[0,S0,S1,S2,S3,S4]的行向量,所述信息比特代表正负权重(如±1表示逻辑0,1,或采用±1/3表示),w为待传输的码字向量[w0,w1,w2,w3,w4,w5]。如果使用正负权重±1,则可利用系数1/3将传输码字归一化,以生成具有四元符集{±1/3,±1}的码字。此外,由于选择式13的子信道码字,因此形成最终传输码字的子信道代码的线性结合将产生码字元素的缩集。也就是说,所述子信道代码向量中的至少一个向量的大小未归一化,且缩小后的符集的行方向的线性结合产生一个缩集。可以看出,在具有三个非零元素的子信道代码向量中,第一非零元素的大小为第二非零元素的两倍,也为第三非零元素的两倍,而且第一非零元素的位置位于在该位置上仅有一个其他子信道码字向量具有非零元素的位置。
如上式3所述,正确归一化的式13矩阵A的转置矩阵即为各列定义了MIC系数的逆矩阵。此方案的吞吐量为6.66Gbps/线,其眼图如图7A~图7E所示。所有MIC的水平眼开度均为106.2皮秒。本文中,以6条线路传输5个比特的代码(即式13的5b6w码),式3的MIC矩阵,以及表1码字称为“Glasswing”码。
第三信令设计实施方式也使用线性编码,并称为集成NRZ(Ensemble-NRZ,ENRZ)。《Cronie I》中对ENRZ进行了完整描述。ENRZ具有±perm(l,-1/3,-1/3,-1/3)形式的八个码字(其中,“perm”表示所列各系数的所有排列组合形式)以及(1/2,-1/2,1/2,-1/2),(1/2,1/2,-1/2,-1/2)和(1/2,-1/2,-1/2,1/2)形式的三个MIC,并且可在每个单位间隔内在4条线路上发送3个比特。相应编码矩阵为如下乘以比例系数后的4×4阿达玛(Hadamard)矩阵:
选择比例系数1/3的目的在于将最终值限制于-1和1之间。ENRZ信令的吞吐量为6Gbps/线,其眼图见图8A~图8C。三个MIC的水平眼开度为106.2皮秒。
第四实施方式为向量信令码设计,该设计的特征为含有具有不同ISI比的不同MIC,以及含有所述MIC的非可用码字。此类代码允许设计人员在相同吞吐量和相同ISI比的前提下使用更少的MIC。此外,如以下实施例所示,与ISI比较高的MIC相比,ISI比较小的MIC通常为在更多线路上实施线性结合的MIC。由于这些MIC具有更大的水平眼开度,因此当线性结合中涉及较多线路时,上述特征使得此类MIC可容忍更高的偏斜度。此设计概念的一例见《Shokrollahi I》,该例描述了一种具有±(perm(1,1,0,-1)|-1)形式的24个码字的代码,其中,“perm”表示系数的所有排列组合形式。此码可在每个单位间隔内在5条线路中发送4.5个比特。此方案具有7个MIC,前6个MIC均用于在前4条线路之间实施成对比较,而且ISI比均为2。最后一个MIC的系数为(1/4,1/4,1/4,1/4,-1),且ISI比为1。此方案的吞吐量为7.2Gbps/线,眼图见图9A和图9B。从眼图中可看出,与ISI比较高的成对比较MIC的水平眼开度(图9A,水平眼开度=62.5皮秒)相比,最后一个MIC的水平眼开度较大(图9B,水平眼开度=106.2皮秒)。
最后一种实施方式为一种利用13个MIC在8条线路上发送8个比特的代码。相应码字的形式为±(perm(1,1,0,-1)|perm(-1,-1,0,1)),其中,“perm”表示系数的所有排列组合形式。所述各MIC为:作用于前4条线路的6个成对比较MIC;作用于后4条线路的6个成对比较MIC;以及一个MIC(1,1,1,1,-1,-1,-1,-1)/4。最后的输入MIC的ISI比为1,而每个成对比较MIC的ISI比为2。此方案的吞吐量为8Gbps/线,眼图见图10A和图10B。ISI比为1的MIC的水平眼开度为106.2皮秒(图10B),而其余的ISI比为2的MIC的水平眼开度为62.5皮秒(图10A)。
为了对上述各实施方式进行直接比较,我们先计算出上述ENRZ、NRZ及PAM-4信令方案的水平眼开度,然后将水平眼开度关系绘制为吞吐量的函数。由于这些信令方案中获得相同吞吐量所需的波特率各不相同,因此为了比较目的,我们还对吞吐量值实施了归一化。
上述结果示于图11。比较NRZ和ENRZ可知,两种方案的ISI比均等于1,但ENRZ的波特率为NRZ波特率的2/3。比较PAM-4和ENRZ可发现,虽然ENRZ的波特率为PAM-4的4/3,但是其ISI比仅为后者的1/3。因此,ISI比最小的实施方式,即ISI比为1的ENRZ,在整个吞吐量范围内表现最佳,这表明小的ISI比为好的通信***性能的一个指标。
子信道增益归一化
通过对表1代码及式3矩阵所定义的接收器实施数值分析可看出,由矩阵行1,3,5定义的比较器所产生的输出值为±2/3,而由矩阵行2,4定义的比较器所产生的输出值为±1。因此,与差分信令相比,垂直眼开度的损失为20*log10(3)=~9.5dB。此输出电平的变化源于我们对接收矩阵正交性的较为宽松的定义,这是因为对角矩阵MTM=D上非统一的值使得相应子信道具有非统一的增益。对于熟悉本领域的技术人员而言容易理解的是,对上述矩阵实施归一化(即给其元素乘以系数,从而使得上述对角线值均为1)后,可形成所有子信道的增益为恒定统一值的***。然而,这类已知的归一化方法将导致非最优的实施方式,这是因为在实际***中实现大量的不同归一化系数值,况且在许多情况中,这些值中还包括不合理的值。
至少一种实施方式保留了式3所例示说明的带易实施系数值的非归一化矩阵,并同时通过修饰调制各种子信道所用的输入信号振幅的方式,对子信道振幅的变化进行补偿。例如,假设一个***具有8个相同增益子信道以及一个增益为前述增益的0.8倍的子信道,则该***的信噪比(SNR)最终由后一子信道限制。因此,将后一子信道的传输输入增加至{+1.2,-1.2},而非使其保持{+1,-1},可提高相应信道的输出。或者,也可将所有其他子信道输入减小至{+0.8,-0.8},从而降低相应信道的输出,使得所有信道具有以较小发射功率便可发送的相同输出电平。
然而,上述补偿技术并非没有代价。如《Shokrollahi IV》中所述,以上述方式修饰输入向量将导致在信道中传输码字时需要更大的符集大小(并从而使得发射器必须产生更多的不同信号电平)。《Shokrollahi IV》中描述了一些数值方法,用于指导如何选择合适的调制振幅,从而在最小程度扩增所需符集大小的同时,得到匹配程度最高的子信道输出。
通过应用《Shokrollahi IV》中所述方法,Glasswing方案计算所得的最佳初始代码集为(0,±3/8,±1/4,±3/8,±1/4,±3/8),而相应代码的符集为大小为10的(1,7/8,1/2,1/4,1/8,-1/8,-1/4,-1/2,-7/8,-1)。所得码字如表2所示,此间所得的新代码称为5b6w_10_5码。图22所示为可生成上述信号电平的驱动器的一种实施方式。每个驱动器片1010生成一个线路输出信号,而每个驱动器片中的多个驱动器元素可共同生成所需的10个不同输出电平。
表2
在此代码中,式3中的所有比较器产生的输出值均为±3/4。与未修饰的5b6w码相比,垂直眼开度的增加量为20*log10((3/4)/(2/3))=~1dB。5b6w_10_5码的端接功率约为未修饰5b6w码的88%,由此可见,即使端接功率较小,5b6w_10_5也能使得垂直眼开度获得稍许改善。然而,为获得此项改善所需付出的实施代价在于,发射器需提高其复杂度,以将数据编码为能为每条线路在10个而非4个符号值之间进行选择的内部表现形式,并需设置能生成10个而非4个不同输出电平的线路驱动器。这一发射器实施方式可与式3矩阵所定义的任何Glasswing接收器完全兼容,而且所需的线路驱动器功率将比未修饰的5b6w发送驱动器更小。迫于对此通常而言不利的代价及益处之间的权衡,大多数Glasswing发射器实施方式倾向于使用未修饰的5b6w信号电平,其中,当在成本效益方面有益时,采用5b6w_10_5修饰内容的替代实施方式显然可与未修饰码互换和/或互通运行。
基本型Glasswing接收器
图14所示为式3矩阵所定义的Glasswing接收器的一种实施方式。其6条输入线路为w0~w5,5个子信道输出为S0~S4。在本文采用的附图惯例中,多输入比较器210~250的每个输入端由权重值代表,以表示该输入端对定义了每个MIC的式3矩阵行所确定的最终结果输出的相对贡献。因此,210和230可视为现有的双输入差分比较器,每个该差分比较器具有一正一负两个等值相反权重值。比较器220和240各自具有对总正值各贡献一半的两个正输入,以及一个贡献所有负值的输入。比较器250具有三个对总正值各贡献1/3的输入,三个对总负值各贡献1/3的输入。
式3矩阵2~6行所对应的5个多输入比较器的实施方式可采用《Holden I》中的扩展差分对设计,或采用《Ulrich I》的可替代设计。在一些实施方式中,上述多线路比较器级还设置连续时间线性均衡器(CTLE)。《Shokrollahi IV》中给出了适于与Glasswing码联用的此类实施方式的晶体管级电路图,而且这些电路图也作为示例纳为本文的图16~20。每种设计均以含集成CTLE和不含集成CTLE两个版本进行了图示,而且在合适之处还示出了替代实施方式。
图16所示实施方式或图17所示其替代实施方式适于用作图14中的比较器250。图18所示实施方式或图19所示其替代实施方式适于用作比较器220,以及在将输入端w0,w1和w2简单换作w3,w4和w5后,用作比较器240。图20所示实施方式适于用作比较器210,以及在作出相同的输入端替换后,用作比较器230。
一般而言,典型Glasswing码实施方式的信道通常较短且信号传播特性较好,因此无需使用判定反馈均衡(Decision Feedback Equalization,DFE)技术。DFE和其他已知本领域技术可与上述本发明直接结合,但是这不对本发明构成任何限制。在至少某些此类实施方式中,DFE并不针对接收线路信号,而是针对子信道实施。在这种设计中,可针对二进制信号值实施DFE历史和校正计算,这与在直接针对线路信号实施DFE时需在四元线路信号(5b6w)或十元线路信号(5b6w_10_5)实施DFE历史和校正计算的做法不同。
Glasswing发射器
根据通信信道的具体特性以及半导体器件制造方法的不同,Glasswing发射器可采用各种本领域已知的解决方案。极短Glasswing信道和/或中等数据速率Glasswing信道可使用高阻抗“类CMOS”点对点互连,这些点对点互连最好由现有电压模式驱动器驱动。匹配阻抗端接的更高速度和/或更长信道可优选使用电流模式驱动器驱动。
对于本领域技术人员而言容易理解的是,上述所需的多级输出驱动器为数模(D/A)转换的一种特殊实例,而D/A转换为本领域的一个熟知领域。因此,已知D/A转换方法(包括基于电阻链、电阻梯、已调节电压或电流求和或已调节电压或电流选择的D/A转换方法)的术语可与本发明联合。所述调节既可以为调节至预设电压或电流等绝对值,也可以为相对于或正比于集成电路电源电压等给定电平的调节。
图21所示为用于采用《Ulrich III》启示内容的5b6w发射器的电阻性源极端接驱动器的一种实施方式。每个驱动器片910生成一个线路输出信号,而每个驱动器片的多个驱动器元素可共同生成必须由发射器产生的四个不同输出电平,以代表由符号+1,+1/3,-1/3和-1组成的码字符集。因此,熟悉本领域的技术人员容易理解的是,在图21中编码器的输出中,每驱动器片910至少有两个二进制选择信号,以实现对至少代表每条线路的所述四符符集的输出值进行选择。此外,如上所述,为了优化Glasswing接收信道增益,还可通过使用修饰的调制值而使用更大的符集,这一做法的代价在于需增加发射器复杂度,例如,需扩大编码器与线路驱动器间选择信号集合,以实现对至少表示更大符集集合的输出值的选择。图22所示为用于生成多个信号电平的驱动器的一种实施方式。
在实施方式的具体信道特性和***设计目标要求下,Glasswing方案还可与有限脉冲响应滤波等已知传输驱动器均衡技术联用。
在一种实施方式中,参考图23,描述了一种方法2300。在框体2302中,接收一组信息比特。在框体2304中,由编码器生成缩集码字向量。所述编码器形成子信道代码向量的加权和,其中,部分基于由所接收的一组比特中的相应信息比特所确定的相应正负权重对每个子信道代码向量进行加权。在各种实施方式中,所述编码器为正交编码逻辑电路,而且其通过将所接收的一组信息比特映射至相应缩集码字向量并输出缩集码字向量选择信号的方式,生成缩集码字向量。此后,可将所述选择信号提供至所述多个线路驱动器。所述多个线路驱动器中的每个线路驱动器对所述缩集码字向量选择信号的一部分进行使用,以输出表示该缩集码字向量元素的相应电流或电压。如前所述,所述子信道代码向量形成缩集权重矩阵,而且还相互正交并正交于共模向量。
在框体2306中,利用多个线路驱动器,发送所述缩集码字向量。具体而言,所述缩集码字向量包括多个缩集码字向量元素,每个缩集码字向量元素由所述多个线路驱动器中的一个相应线路驱动器经多线路通信总线的一条线路发送。在一些实施方式中,所述缩集码字向量元素选自归一化元素集合{+1,+1/3,-1/3,-1}。
方法2300可与具有5个子信道向量的***联用,其中,所述5个子信道向量用于与六线多线路通信总线联用。上述正负权重选自归一化元素集合{+1,-1}。在其他方法中,至少一个子信道的正负权重可根据时钟信号调制。此外,在一些实施方式中,电压偏移可被建模为一种包括作用于共模子信道向量上的恒定权重的形式。
以下参考图24,描述方法2400。在框体2402中,从多线路通信总线中接收一组信号。所述一组信号代表由子信道代码向量的正负加权和形成的缩集码字向量。在框体2404中,多个子信道多输入比较器生成多个子信道输出信号。所述多个多输入比较器中的每个多输入比较器实施为对应于一个子信道代码向量的输入权重向量,并且输出正负输出信号。在框体2406中,根据相应正负输出信号,确定一组信息比特。例如,通过相对于基准值分割所述正负输出信号的方式,确定该一组信息比特。在一种实施方式中,所述缩集码字向量元素选自归一化元素集合{+1,+1/3,-1/3,-1},所述子信道多输入比较器的数目为5,所述多线路通信总线包括6条线路。
带嵌入式时钟的Glasswing
图15实施方式除了使用图14中基本型Glasswing接收器之外,还包括由一条子信道承载的嵌入式时钟信号。如《Shokrollahi III》中所述,当在子信道内嵌入时钟时,通常选择最大振幅信道承载该时钟。在一般设计实践中,上述选择通常可使得所述时钟信道具有最佳SNR,从而使得其具有最干净的输出结果。
然而,当实际实施方式中各种Glasswing信道的增益变化较为温和时,实际操作中,根据上述原则为时钟选择特定子通道的动机便变得无足轻重。在图15所示实施方式中,式3矩阵底行所定义的子信道被指定为用于承载时钟的子信道,这是因为其在实际实施方式中的对称和延迟特性被证明有利于与现有时钟数据恢复电路的延迟采样行为相结合。此种电路的一例图示为380,且包括:时钟边沿检测器382;固定式、可调式或DLL控制式延时器385;以及采样保持式或等效数据采样器388。
实施方式
在一种实施方式中,一种方法包括:接收一组信息比特;由编码器通过形成子信道代码向量的加权和而生成缩集码字向量,其中,部分基于由所接收的该组信息比特中的相应信息比特所确定的相应的正负权重对每个子信道代码向量进行加权,所述子信道代码向量形成缩集权重矩阵,所述子信道代码向量相互正交且正交于一共模向量;以及利用多个线路驱动器发送所述缩集码字向量,其中,该缩集码字向量包括多个缩集码字向量元素,每个缩集码字向量元素由所述多个线路驱动器当中的一个相应线路驱动器经多线路通信总线的一条线路发送。
在一种实施方式中,所述编码器为正交编码逻辑电路;所述生成缩集码字向量包括:将所接收的该组信息比特映射至相应的缩集码字向量;以及输出一缩集码字向量选择信号。
在一种实施方式中,所述缩集码字向量选择信号被提供至所述多个线路驱动器;所述多个线路驱动器中的每个线路驱动器使用所述缩集码字向量选择信号的一部分来输出表示对应的缩集码字向量元素的相应的电流或电压。
在一种实施方式中,所述子信道代码向量中的至少一个子信道代码向量的大小未归一化,其中,所述缩集权重矩阵的行方向线性结合生成一个缩集。
在一种实施方式中,所述缩集码字向量元素选自归一化元素集合{+1,+1/3,-1/3,-1}。所述子信道向量的数目为5,且所述多线路通信总线包括6条线路。
在一种实施方式中,所述正负权重选自归一化元素集合{+1,-1}。在一种实施方式中,至少一个子信道的正负权重根据时钟信号调制。
在一种实施方式中,提供非调制共模子信道向量,用于实现电压偏移。
在一种实施方式中,具有三个非零元素的子信道代码向量包括第一非零元素,所述第一非零元素的大小为第二非零元素和第三非零元素中每一个的两倍,所述第一非零元素的位置位于仅一个其他子信道码字向量具有非零元素的位置。
在一种实施方式中,一种方法包括:从多线路通信总线中接收一组信号,该组信号代表由子信道代码向量的正负加权和形成的缩集码字向量;利用多个子信道多输入比较器生成多个子信道输出信号,所述多个多输入比较器中的每个多输入比较器实施为对应于一个子信道代码向量的输入权重向量并输出正负输出信号;根据相应正负输出信号,确定一组信息比特。
在一种实施方式中,所述方法可利用多个信号分割器确定所述一组信息比特。在一种实施方式中,所述缩集码字向量元素选自归一化元素集合{+1,+1/3,-1/3,-1},所述子信道多输入比较器的数目为5,所述多线路通信总线包括6条线路。
在一种实施方式中,一种装置包括:多条信号导线,用于接收一组信息比特;编码器,连接至所述信号导线,所述编码器用于通过形成子信道代码向量的和而生成缩集码字向量,其中,部分基于由所接收的该组信息比特中的相应信息比特所确定的相应的正负权重对每个子信道代码向量进行加权,所述子信道代码向量形成缩集权重矩阵,所述子信道代码向量相互正交且正交于一共模向量;以及多个线路驱动器,用于发送所述缩集码字向量,其中,所述缩集码字向量包括多个缩集码字向量元素,每个缩集码字向量元素由所述多个线路驱动器中的一个相应的线路驱动器经多线路通信总线的一条线路发送。
在一种实施方式中,所述编码器还包括逻辑电路,所述逻辑电路将所接收的该组信息比特映射至相应的缩集码字向量,并输出一缩集码字向量选择信号。
在一种实施方式中,所述装置包括用于将所述缩集码字向量选择信号提供至所述多个线路驱动器的线路连接,其中,所述多个线路驱动器中的每个线路驱动器使用所述缩集码字向量选择信号的一部分来输出表示对应的缩集码字向量元素的相应的电流或电压。
在一种实施方式中,所述编码器设置为生成缩集,该缩集包括元素集合{+1,+1/3,-1/3,-1}。所述多线路通信总线包括6条线路,用于发送一组5个子信道代码向量。在一种实施方式中,所述编码器设置为从元素集合{+1,-1}中选择所述正负权重。
所述编码器设置为根据时钟信号对至少一个子信道代码向量进行调制。所述编码器设置为通过提供非调制共模子信道向量而实现电压偏移。
在一种实施方式中,一种装置包括:多线路通信总线,该多线路通信总线设置为接收一组信号,该组信号代表由子信道代码向量的正负加权和形成的缩集码字向量;多个子信道多输入比较器,所述多个多输入比较器中的每个多输入比较器实施为对应于一个子信道代码向量的输入权重向量并输出正负输出信号;多个信号分割器,所述分割器设置为根据相应正负输出信号确定一组信息比特。
在一种实施方式中,所述装置可接收缩集码字向量元素,该元素选自归一化元素集合{+1,+1/3,-1/3,-1};所述子信道多输入比较器的数目为5,所述多线路通信总线包括6条线路。
在另一实施方式中,提供一种诊断工具,该工具使用包括如下步骤的方法:接收多个候选码字;针对多个多输入比较器中的每个多输入比较器,计算可用码字到过原点多输入比较器超平面的最大距离与可用码字到该过原点多输入比较器超平面的最小距离的比值;导出每个中心多输入比较器的所述比值,用于分析。此外,所述超平面可为含有偏移量的偏移超平面,而且上述比值为可用码字到该偏移超平面的最大距离与可用码字到该偏移超平面的最小距离的比值。
本文所呈实施例描述了将向量信令码用于点对点有线通信。为了说明目的,以上将第一发送器件与第二接收器件之间的互连描述为单向信令网络。然而,不应以任何方式将此视为对本发明范围构成了限制。本申请中所公开的方法同样适用于可使用交替信令方向的网络(即半双工网络),以及可在分立的发射器和接收器之间同时双向通信的网络(即全双工网络)。同样,上述本发明的一个以上实例可用于以基本并行的方式传输更大的数据字,以及/或者提供更高的总通信带宽,其中,不同实例既可具有各自的嵌入式时钟,也可两个或更多实例共享一个共同的时钟。除了上述有线互连之外,也可以类似方式使用包括光通信和无线通信在内的其他通信介质。因此,“电压”和“信号电平”等描述性词语应视为包括其在其他度量***中的同等概念,如“光强”、“射频调制”等。本文所使用的“物理信号”一词包括可传送信息的物理现象的任何适用形态和属性。此外,物理信号可以为有形的非暂时性信号。
Claims (15)
1.一种通过多线路总线的多条线路传送信息的方法,其特征在于,包括:
在所述多线路总线的所述多条线路上并行接收向量信令码的码字的符号集合,每个符号接收自所述多线路总线的一条相应的线路;
生成一组模拟输出,每个模拟输出形成为所述码字的所述符号集合中的相应的符号子集的线性结合,所述线性结合由接收相应的所述符号子集的多输入比较器形成,所述线性结合与所述向量信令码的多个相互正交的子信道代码向量中相应的子信道代码向量相关联,其中,所述符号子集中的符号数目对应于相应的所述子信道代码向量中的非零元素数目,并且至少两个子信道代码向量中的非零元素数目不相等;以及
通过所述一组模拟输出生成一组信息比特。
2.如权利要求1所述方法,其特征在于,生成所述一组信息比特包括使用相应的信号分割电路对每个模拟输出进行分割。
3.如权利要求2所述方法,其特征在于,至少一个模拟输出使用PAM-3分割器分割。
4.如权利要求2所述方法,其特征在于,至少一个模拟输出使用PAM-4分割器分割。
5.如权利要求1所述方法,其特征在于,所述一组模拟输出中的至少两个模拟输出由具有不同数目符号的符号子集的线性结合生成。
6.如权利要求1所述方法,其特征在于,每个所述子信道代码向量正交于一共模向量。
7.如权利要求1所述方法,其特征在于,所述一组模拟输出具有相同的输出电平。
8.如权利要求1所述方法,其特征在于,所述向量信令码为四元码。
9.一种通过多线路总线的多条线路传送信息的装置,其特征在于,包括:
多个多输入比较器,用于在所述多线路总线的所述多条线路上并行接收向量信令码的码字的符号集合,每个符号接收自所述多线路总线的一条相应的线路,所述多个多输入比较器中的每个多输入比较器用于接收所述码字的所述符号集合中的相应的符号子集并生成一组模拟输出中的相应一个模拟输出,所述相应一个模拟输出形成为所述相应的符号子集的线性结合,所述线性结合与所述向量信令码的多个相互正交的子信道代码向量中相应的子信道代码向量相关联,其中,所述符号子集中的符号数目对应于相应的所述子信道代码向量中的非零元素数目,并且至少两个子信道代码向量中的非零元素数目不相等;以及
用于通过所述一组模拟输出生成一组信息比特的比较器。
10.如权利要求9所述装置,其特征在于,所述比较器包括使用相应的信号对每个模拟输出进行分割的分割电路。
11.如权利要求10所述装置,其特征在于,所述分割电路包括用于分割至少一个模拟输出的PAM-3分割器。
12.如权利要求10所述装置,其特征在于,所述分割电路包括用于分割至少一个模拟输出的PAM-4分割器。
13.如权利要求9所述装置,其特征在于,所述一组模拟输出中的至少两个模拟输出由具有不同数目符号的符号子集的线性结合生成。
14.如权利要求9所述装置,其特征在于,每个所述子信道代码向量正交于一共模向量。
15.如权利要求9所述装置,其特征在于,所述一组模拟输出具有相同的输出电平。
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