WO2011118709A1 - スイッチング電源回路およびスイッチング電源回路の制御方法 - Google Patents
スイッチング電源回路およびスイッチング電源回路の制御方法 Download PDFInfo
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- WO2011118709A1 WO2011118709A1 PCT/JP2011/057185 JP2011057185W WO2011118709A1 WO 2011118709 A1 WO2011118709 A1 WO 2011118709A1 JP 2011057185 W JP2011057185 W JP 2011057185W WO 2011118709 A1 WO2011118709 A1 WO 2011118709A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0003—Details of control, feedback or regulation circuits
- H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
- H02M3/1586—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel switched with a phase shift, i.e. interleaved
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Definitions
- the present invention relates to a switching power supply circuit and a switching power supply circuit control method, and more particularly to a power factor correction circuit.
- a circuit composed of a reactor, a diode, and a switch element has been proposed as a power factor correction circuit for improving the power factor on the input side. More specifically, a reactor and a switch element are connected in series between two input terminals, and a diode and a switch element are connected in series between two output terminals. The diode is provided with its anode facing the switch element side. A smoothing capacitor is provided between the two output terminals.
- a power factor correction circuit having 2 circuits has been proposed.
- the conduction timings of the switch elements belonging to the two circuits are made different from each other.
- Such a power factor correction circuit is called a so-called interleaved power factor correction circuit.
- Patent Document 1 is disclosed as a technique related to the present invention.
- an object of the present invention is to provide a switching power supply circuit that contributes to the realization of electrical characteristics corresponding to load fluctuations.
- a first aspect of the switching power supply circuit according to the present invention includes first and second input terminals (P1, P2), first and second output terminals (P3, P4), and the second input terminal. And a second power supply line (LL) connecting the second output terminal and a plurality of circuits (3, 3a, 3b), each of the plurality of circuits including the first input terminal and the second input terminal.
- P1, P2 first and second input terminals
- P3, P4 first and second output terminals
- LL second power supply line
- a first power supply line (LH1, LH2) connecting the first output terminal, a reactor (L1, L2) provided on the first power supply line, and the reactor connected in series on the first power supply line; Diodes (D11, D21) disposed with the anode facing the reactor, switch elements (S1, S2) provided between the reactor and the diode, and the second power supply line
- Diodes (D11, D21) disposed with the anode facing the reactor, switch elements (S1, S2) provided between the reactor and the diode, and the second power supply line
- a second aspect of the switching power supply circuit according to the present invention is the switching power supply circuit according to the first aspect, wherein the switch element (S1) belonging to the plurality of circuits (3a) is an insulated gate bipolar transistor.
- the switch element (S1) belonging to the other circuit (3b) is a MOS field effect transistor.
- a third aspect of the switching power supply circuit according to the present invention is the switching power supply circuit according to the first aspect, wherein the switching element (S1) belonging to the one of the plurality of circuits (3a) is a silicon carbide semiconductor or The switching element (S2) formed of a gallium nitride semiconductor and belonging to the other plurality of circuits (3b) is formed of a silicon semiconductor.
- a fourth aspect of the switching power supply circuit according to the present invention is the switching power supply circuit according to any one of the first to third aspects, wherein the reactor (L1) belongs to the plurality of circuits (3a). Is smaller than the impedance of the reactor belonging to the other one of the plurality of circuits (3b).
- a first aspect of the switching power supply circuit control method is the switching power supply circuit control method according to any one of the first to fourth aspects, wherein the plurality of circuits (3a, 3b) A first step of maintaining all the switching elements (S1, S2) belonging to non-conduction, and a current flowing through the first and second input terminals (P1, P2) exceeds a first predetermined value (Iref1) The second step of repeatedly switching between conduction / non-conduction of the switching element (S1) belonging to the one of the plurality of circuits (3a) is performed.
- a second aspect of the switching power supply circuit control method is the switching power supply circuit control method according to the first aspect, wherein in the second step, the first and the first steps in the first step are performed.
- the switching element (S1) belonging to the one of the plurality of circuits (3a) based on a first DC voltage command value higher than the voltage between the two output terminals (P3, P4). Switch repeatedly.
- a third aspect of the switching power supply circuit control method according to the present invention is the switching power supply circuit control method according to the first or second aspect, wherein the first and second input terminals (P1, P2). )
- the first and second input terminals (P1, P2) When the current flowing through the first circuit exceeds a second predetermined value (Iref2) that is larger than the first predetermined value (Iref1), the one circuit (3a) and the other circuit
- Iref2 a second predetermined value that is larger than the first predetermined value (Iref1)
- a fourth aspect of the switching power supply circuit control method according to the present invention is the switching power supply circuit control method according to the third aspect, and is larger than the first DC voltage command value in the third step. Based on the second DC voltage command value, the switching elements (S1, S2) belonging to the one plurality of circuits (3a) and the other one of the plurality of circuits (3b) are turned on / off. Switch repeatedly.
- a fifth aspect of the switching power supply circuit control method is a switching power supply circuit control method according to any one of the first to fourth aspects, wherein the first step and the second step are the same.
- the switching elements (S1, S2) belonging to the plurality of circuits (3, 3a, 3b) Prior to the first relationship between the current and the efficiency of the switching power supply circuit, all of the switching elements (S1, S2) belonging to the plurality of circuits (3, 3a, 3b) are kept nonconductive. And switching between conduction / non-conduction of the switching element (S1) of the plurality of circuits (3a) based on the first DC voltage command value, and the current and the switching power supply
- a fifth step for determining a second relationship between the efficiency of the circuit and the first predetermined value (Iref1) is the same in both the first relationship and the second relationship. Wherein the current is employed to provide the rate.
- the case where the plurality of circuits are two first and second circuits will be described below as an example.
- At least one characteristic of the reactor, the switch element, and the diode of the first circuit is different from at least one characteristic of the reactor, the switch element, and the diode of the second circuit. Therefore, the electrical characteristics of the switching power supply circuit are different between the first state in which only the first circuit is operated and the second state in which only the second circuit is operated. The electrical characteristics of the switching power supply circuit in the third state where both the first and second circuits are operated are different from the electrical characteristics in the first state and the second state.
- the three electric characteristics can be exhibited as the switching power supply circuit by the two first and second circuits.
- the electrical characteristics in each of the first state to the third state affect the power factor, efficiency, and harmonics, respectively. Therefore, the power factor, efficiency, and harmonics can be appropriately adjusted by appropriately selecting the first state and the third state.
- the first state to the third state can be appropriately selected according to the fluctuation of the load connected to the first and second output terminals. Therefore, it is possible to contribute to the realization of electrical characteristics according to the load variation.
- a circuit having a MOS field effect transistor has a smaller current capacity than an insulated gate bipolar transistor, and is widely used in television receivers and the like. Therefore, general-purpose parts can be used, and costs can be reduced.
- the switching element formed of the silicon carbide semiconductor or the gallium nitride semiconductor has lower conduction loss and higher breakdown voltage than the switching element formed of the silicon semiconductor.
- the reactors belonging to a plurality of circuits can be reduced in size.
- the switching power supply circuit when the current is lower than the first predetermined value, all the switching elements are non-conductive. When the current is low, the ratio of the switching loss to the loss generated in the circuit is large. Therefore, the efficiency in the state where the current is lower than the first predetermined value can be increased. On the other hand, when the current is larger than the first predetermined value, the efficiency in this state can be improved by repeatedly switching the conduction / non-conduction of the switching elements belonging to a plurality of circuits.
- the first DC voltage command value in the range in which the current is large as compared with the control method in which the first DC voltage command value is equal to the DC voltage in the first step A stable operation of the circuit in two steps can be realized.
- the efficiency of the circuit in a range where the current is small can be increased.
- the third aspect of the control method of the switching power supply circuit since the ratio of the switching loss accounts for a large part of the loss generated in the circuit when the current is low, the current is lower than the second predetermined value. Efficiency can be increased. On the other hand, when the current is larger than the second predetermined value, the efficiency in this state can be improved by repeatedly switching the conduction / non-conduction of the switching elements belonging to one and the other plurality of circuits.
- the efficiency in the third step is higher than that in the control method in which the second DC voltage command value is made equal to the first DC voltage command value. Can be improved.
- the first step and the second step can be selected so as to obtain the highest efficiency.
- the switching power supply circuit includes a plurality of circuits 3, input terminals P1 and P2, and output terminals P3 and P4.
- a DC voltage is applied between the input terminals P1 and P2.
- a diode rectifier circuit 2 is connected to the input terminals P1 and P2.
- the diode rectifier circuit 2 rectifies the AC voltage from the AC power source 1 and applies the rectified DC voltage between the input terminals P1 and P2.
- the potential applied to the input terminal P2 is lower than the potential applied to the input terminal P1. It is not essential that the diode rectifier circuit 2 is connected to the input terminals P1 and P2. Any configuration that applies a DC voltage between the input terminals P1 and P2 only needs to be connected to the input terminals P1 and P2.
- the plurality of circuits 3 are all connected to the input terminals P1 and P2 and the output terminals P3 and P4. As will be described later, each circuit 3 functions as a booster circuit that boosts the DC voltage applied to the input terminals P1 and P2 and functions as a power factor correction circuit that improves the power factor on the input side.
- a smoothing capacitor C1 is provided between the output terminals P3 and P4.
- the smoothing capacitor C1 smoothes the DC voltage boosted by each circuit 3.
- a capacitor C2 may be provided between the input terminals P1 and P2.
- the capacitor C ⁇ b> 2 can reduce noise of the current input to each circuit 3.
- FIG. 2 shows a conceptual example of a specific configuration of the plurality of circuits 3.
- two circuits 3a and 3b are shown as an example.
- the outputs of the circuits 3a and 3b are input to the inverter 4 via the smoothing capacitor C1. That is, the output terminals P3 and P4 are connected to the inverter 4 on the input side of the inverter 4.
- the input terminal P2 and the output terminal P4 are connected to each other by a power line LL.
- the circuit 3a includes a power line LH1, a reactor L1, a diode D11, and a switch element S1.
- the power supply line LH1 connects the input terminal P1 and the output terminal P3.
- Reactor L1 is provided on power supply line LH1.
- the diode D11 is connected in series with the reactor L1 on the output terminal P3 side with respect to the reactor L1.
- the diode D11 is provided with its anode facing the reactor L1.
- the switch element S1 is provided between the point between the reactor L1 and the diode D11 and the power line LL.
- the control unit 5 controls conduction / non-conduction of the switch element S1.
- the switch element S1 includes a transistor T1 and a diode D12.
- the transistor T1 is an insulated gate bipolar transistor, for example, and is provided with its emitter electrode facing the power supply line LL.
- the switch element S1 does not necessarily have to include the transistor T1 and the diode D12.
- the diode D12 may not be provided.
- a MOS Metal-Oxide-Semiconductor field effect transistor may be employed as the switch element S1.
- the circuit 3b includes a power line LH2, a reactor L2, a diode D21, and a switch element S2.
- the connection relationship among the power supply line LH2, the reactor L2, the diode D21, and the switch element S2 is the same as the connection relationship between the power supply line LH1, the reactor L1, the diode D11, and the switch element S1.
- the switch element S2 includes a transistor T2 and a diode D22.
- the connection relationship between the transistor T2 and the diode D22 is the same as the connection relationship between the transistor T1 and the diode D12.
- the diode D22 is not an essential requirement, and the switch element S1 may be a MOS field effect transistor, for example.
- the control unit 5 controls conduction / non-conduction of the switch element S2.
- control of the switch elements S1 and S2 described below is the control unit 5 unless otherwise specified.
- control unit 5 includes a microcomputer and a storage device.
- the microcomputer executes each processing step (in other words, a procedure) described in the program.
- the storage device is, for example, a ROM (Read-Only-Memory), a RAM (Random-Access-Memory), a rewritable nonvolatile memory (EPROM (Erasable-Programmable-ROM), etc.), and various storage devices such as a hard disk device. One or more can be configured.
- the storage device stores various information, data, and the like, stores a program executed by the microcomputer, and provides a work area for executing the program.
- control unit 5 is not limited to this, and various procedures executed by the control unit 5 or various means or various functions implemented may be realized by hardware.
- the switch element S2 can be made non-conductive and the circuit 3a can be operated alone.
- the switch element S1 can be made non-conductive and the circuit 3b can be operated alone.
- a current flows from the input terminal P1 to the input terminal P2 via the reactor L1 and the switch element S1.
- Such a current increases in accordance with an inclination determined by the inductance of the reactor L1 and the DC voltage between the input terminals P1 and P2 (see the current IL1 in FIG. 3). Electromagnetic energy is accumulated in the reactor L1 by such current.
- the current is reduced by a gradient based on the inductance of the reactor L1 and the capacitance of the smoothing capacitor C1 (see current IL1 in FIG. 3).
- the switch element S1 is turned on again. Thereafter, the above-described operation is repeated. With this operation, the current IL1 changes along a sawtooth shape.
- the mode in which the switch element S1 is turned on immediately after the current IL1 flowing through the reactor L1 reaches zero is called a so-called current critical mode.
- the circuit 3a can function as a switching power supply circuit that boosts the voltage between the input terminals P1 and P2 and applies it between the output terminals P3 and P4. Even in a period in which no current flows to the smoothing capacitor C1 (a period in which the switch element S1 is conductive), a current flows in the diode rectifier circuit 2 via the switch element S1. Therefore, the conduction angle of the current flowing through the diode rectifier circuit 2 can be expanded. In other words, the circuit 3a can function as a power factor correction circuit.
- the current IL1 flowing through the reactor L1 is detected, and the detected current IL1 is input to the control unit 5.
- the control unit 5 detects a zero cross of the current IL1, and outputs a switch signal to the switch element S1 from the time when the zero cross is detected.
- the control part 5 stops the output of the said switch signal when the period determined based on arbitrary DC voltage command values (command value about the voltage between the output terminals P3 and P4) passed. .
- the single operation of the circuit 3b is the same as that of the circuit 3a. Therefore, the circuit 3b functions as a booster circuit and also functions as a power factor correction circuit.
- the current IL2 flowing through the reactor L2 is detected and the detected current IL2 is input to the control unit 5 in order to realize switching for the circuit 3b.
- the control unit 5 detects a zero cross of the current IL2, and outputs a switch signal to the switch element S2 from the time when the zero cross is detected. And the control part 5 stops the output of the said switch signal when the period determined based on arbitrary DC voltage command values passed.
- the current critical mode is adopted, but not limited thereto.
- the switch element S1 or the switch element S2 may be turned on when the current IL1 becomes a predetermined value larger than zero. Such a mode is called a so-called continuous current mode. Further, for example, the switch element S1 or the switch element S2 may be turned on when a predetermined time has elapsed since the current IL1 became zero. Such a mode is called a so-called current discontinuous mode.
- the current critical mode will be described as a representative example below.
- circuits 3a and 3b can be operated in cooperation. Such driving is also called interleaving. The details will be described below.
- the switch element S2 becomes conductive when a predetermined period (the predetermined period may be zero) has elapsed since the switch element S1 became conductive.
- the predetermined period is a period shorter than a period (hereinafter also referred to as a cycle) T from when the switch element S1 is turned on to when it is turned on again.
- a cycle a period
- switch element S1 can be grasped
- switch element S2 can be grasped
- the current I flowing through the diode rectifier circuit 2 is equal to the current IL1 flowing through the reactor L1.
- the current I flowing through the diode rectifier circuit 2 is equal to the sum of the currents IL1 and IL2.
- a portion where the current IL1 is low is filled with a portion where the current IL2 is high (so-called mountain).
- the valley of the current IL2 is filled with a peak of the current IL1. Therefore, the fluctuation component (so-called harmonic component) of the current I can be lowered (see the current I in FIG. 3).
- the shift of the cycle of the currents IL1 and IL2 is not limited to a half cycle, but the harmonic component can be reduced most if the cycle is a half cycle.
- the peak of the current IL2 fills the valley of the current IL1
- the average value of the current IL1 can be increased.
- the maximum value of the current I can be reduced in order to achieve the same average value as when the circuit 3a is operated alone.
- the control unit 5 detects, for example, a zero cross of the current IL1, and outputs a switch signal to the switch element S1 from the time when the zero cross is detected. And the control part 5 stops the output of the said switch signal when the period determined based on arbitrary DC voltage command values passed. In parallel with this, the control unit 5 outputs a switch signal to the switch element S2 after a predetermined period (for example, a half cycle) has elapsed from the time when the zero crossing of the current IL1 is detected, and based on an arbitrary DC voltage command value. The output of the switch signal is stopped when the period determined in this manner elapses.
- a predetermined period for example, a half cycle
- the circuits 3a and 3b can be operated independently, and the circuits 3a and 3b can be operated in cooperation.
- the controller 5 also has a function of switching between the single operation of the circuit 3a, the single operation of the circuit 3b, and the cooperative operation of the circuits 3a and 3b.
- ⁇ Characteristics of circuits 3a and 3b> at least one of the characteristics of the switching elements S1 and S2 belonging to the circuits 3a and 3b, the characteristics of the diodes D11 and D21, and the characteristics of the reactors L1 and L2 is different from each other. Thereby, the electrical characteristics of the circuit 3a and the electrical characteristics of the circuit 3b are different.
- the electrical characteristics of the switching power supply circuit in the first state where the circuit 3a is operated alone are different from the electrical characteristics of the switching power supply circuit in the second state where the circuit 3b is operated independently. Furthermore, the electrical characteristics of the switching power supply circuit in the third state in which the circuits 3a and 3b are operated in cooperation are different from the electrical characteristics of the switching power supply circuit in the first state and the second state.
- the three circuits 3a and 3b can exhibit three electrical characteristics. Further, a 0th state in which both of the two circuits 3a and 3b are not operated, that is, a state in which both of the switch elements S1 and S2 are made non-conductive may be employed. If this is adopted, four electric characteristics can be exhibited by the two circuits 3a and 3b.
- device characteristics that are different in each of the circuits 3a and 3b are listed as examples.
- inductances of reactors L1 and L2 reverse recovery characteristics and forward voltages of diodes D11 and D21, conduction characteristics and gate constants of switch elements S1 and S2 can be given. All of these device characteristics affect any of the power factor, efficiency, and harmonics included in the current as a switching power supply circuit. Moreover, the influence of these device characteristics on the power factor, efficiency, and harmonics depends on the load of the inverter 4 (for example, output current, output frequency, etc.).
- the first state, the second state, and the third state are appropriately selected.
- the switching power supply circuit may be operated by adopting the state. Or you may employ
- the state of the load of the inverter 4 may be classified into four ranges and the 0th state to the 3rd state may be adopted respectively.
- the switching power supply circuit can be operated in the most appropriate operating state according to the load of the inverter 4. It can be operated.
- the present switching power supply circuit can contribute to the realization of electrical characteristics corresponding to load fluctuations.
- circuits 3a and 3b are exemplified and described as the plurality of circuits 3.
- the combination S is represented by the following equation.
- a C b is expressed by the following equation.
- the S switching power supply circuit electrical characteristics can be realized.
- the present switching power supply circuit can contribute to the realization of electrical characteristics corresponding to a finer load state.
- Some of the plurality of circuits 3 may have the same device characteristics.
- FIG. 4 is a diagram showing an example of a switching power supply circuit.
- the switching power supply circuit illustrated in FIG. 4 is different from the switching power supply circuit illustrated in FIG. 2 in that the two switching elements S1 and S2 belong to the circuits 3a and 3b, respectively.
- the switch element S1 is an insulated gate bipolar transistor.
- the switch element S2 is a MOS field effect transistor.
- MOS has been used in the past for metal / oxide / semiconductor stacked structures, and is taken from the acronym Metal-Oxide-Semiconductor.
- the material of the gate insulating film and the gate electrode has been improved from the viewpoint of recent integration and improvement of the manufacturing process.
- polycrystalline silicon has been adopted instead of metal as a material for a gate electrode mainly from the viewpoint of forming a source / drain in a self-aligned manner.
- a material having a high dielectric constant is adopted as a material for the gate insulating film, but the material is not necessarily limited to an oxide.
- MOS is not necessarily limited to the metal / oxide / semiconductor laminated structure, and this specification does not assume such limitation. That is, in view of technical common sense, here, “MOS” has not only an abbreviation derived from the word source but also a broad meaning including a laminated structure of a conductor / insulator / semiconductor.
- the insulated gate bipolar transistor has a large conduction loss as a device characteristic as compared with a MOS field effect transistor, and has a large current capacity as a device characteristic as compared with a MOS field effect transistor.
- the circuit 3b is operated alone. Since the switch element S1 does not conduct in a region where the current I is small, the occurrence of conduction loss can be suppressed, and thus the efficiency can be improved. Further, for example, only the circuit 3a is operated independently in a region where the current is medium (for example, the load of the inverter 4 is a middle region). Therefore, even if the current capacity of the switch element S2 is insufficient, since the switch element S1 having a large current capacity is used, damage to the switch element S2 can be prevented.
- the circuits 3a and 3b are operated in cooperation.
- the current I is large (for example, the load of the inverter 4 is high)
- the circuits 3a and 3b are operated in cooperation.
- the current can be distributed to the switch elements S1 and S2, and the current flowing through the switch elements S1 and S2 can be reduced.
- the maximum value of the current flowing through the switch element S2 may be set lower than the maximum value of the current flowing through the switch element S1. This can be realized in consideration of the following two points.
- the slope of the current flowing through the switching element S2 (that is, the slope when the current IL2 increases) can be lowered.
- the conduction period of the switch element S2 is short, the maximum value of the current flowing through the switch element S2 decreases. Therefore, the maximum value of the current flowing through the switch element S2 can be made lower than the maximum value of the current flowing through the switch element S1 by appropriately setting the inductance of the reactor L2 and the conduction period of the switch element S2.
- the circuit 3b that employs a MOS field effect transistor as the switch element S2 is widely used in, for example, a television receiver. Therefore, the switch element S2 which is a general-purpose component can be employed, and the manufacturing cost can be reduced.
- the switch element S1 may be formed of a SiC (silicon carbide) semiconductor or GaN (gallium nitride semiconductor), and the switch element S2 may be formed of a Si (silicon) semiconductor.
- the switch element S1 formed of a SiC semiconductor or a GaN semiconductor has a smaller conduction loss than the switch element S2 formed of, for example, a Si semiconductor. Therefore, for example, the efficiency in the first state in which the circuit 3a is independently operated can be further increased.
- the manufacturing cost of the switch element S2 formed of Si semiconductor is lower than the manufacturing cost of the switch element S1 formed of SiC semiconductor or GaN semiconductor. Therefore, the manufacturing cost can be reduced as compared with the case where both of the switch elements S1 and S2 are formed of SiC semiconductor or GaN semiconductor.
- a reactor having a small current capacity may be adopted as the reactor L2. That is, the wire diameter of the coil included in the reactor L2 may be reduced. Thereby, reactor L2 can be reduced in size and manufacturing cost can be reduced.
- a circuit 3b employing such a reactor L2 is also widely used in television receivers and the like. Therefore, the reactor L2 which is a general-purpose component can be employed, and the manufacturing cost can be reduced.
- Air conditioner with switching power supply circuit Next, consider the case where the present switching power supply circuit is provided in an air conditioner. At this time, the switching power supply circuit is provided on the input side of the inverter 4 as in the example of FIG. 2, and the inverter 4 applies an AC voltage to a motor (not shown) to control the rotational speed of the motor.
- the motor drives a compressor or a fan included in the air conditioner.
- FIG. 5 shows the occurrence time of each outside air temperature in Japan and the relationship between the outside air temperature and the air conditioning load.
- the air conditioning load in FIG. 5 is recognized as a heating load when the outside air temperature is less than 20 degrees Celsius (hereinafter, omitted), and as a cooling load when the outside air temperature is greater than 20 degrees.
- the period when the outside air temperature is, for example, 5 degrees or less is relatively short throughout the year.
- the period requiring a large value for the heating capacity is short.
- the capability in this case, the heating capability
- the heating intermediate capability P2 that is half the rated capability is the outside temperature. This is consistent with the ability when is 7-8 degrees (T2).
- T2 the heating intermediate capability
- the inverter 4 is frequently operated in a state where the air conditioning load is equal to or less than the heating intermediate capacity P2 throughout the year.
- the cooling intermediate capacity P3 is around 29 degrees (T3). It matches the ability when There are relatively many periods during which the cooling intermediate capacity P3 or less is operated.
- the electrical characteristics of the switching power supply circuit with respect to a moderate load or less are compared with the case where the switching power supply circuit is employed in other fields. Improvement is particularly desired.
- the switching power supply circuit is controlled as follows.
- the 0th state, the first state, and the third state are switched in accordance with the load state.
- the load status is distinguished by the current I, for example. Therefore, here, for example, the 0th state, the first state, and the third state are switched according to the magnitude of the current I.
- FIG. 6 shows an example of the relationship between the magnitude of the current I and the efficiency in each state.
- the relationship between the current I and the efficiency in the 0th state, the first state, and the third state is indicated by a solid line, a broken line, and a one-point difference line, respectively.
- Such a relationship can be obtained in advance by experiments or simulations.
- the relationship in the 0th state can be obtained by calculating the efficiency of the switching power supply circuit while changing the current I while maintaining all the switch elements S1 and S2 non-conductive. The same applies to the first state and the third state.
- the efficiency in each state has a generally convex shape.
- the current value I1 that takes the efficiency peak in the 0th state is smaller than the current value I2 that takes the efficiency peak in the first state, and the current value I2 is smaller than the current value I3 that takes the efficiency peak in the third state.
- the efficiency in the 0th state and the efficiency in the first state take the same value at a current value Iref1 that is larger than the current value I1 and smaller than the current value I2.
- the efficiency in the first state and the efficiency in the third state have the same value at a current value Iref2 that is larger than the current value I2 and smaller than the current value I3.
- the 0th state, the 1st state, and the 3rd state are switched as follows in order to increase the efficiency in a wide range of current. That is, when the current I is smaller than the current value Iref1, the circuits 3a and 3b are not operated, that is, the switch elements S1 and S2 are kept nonconductive and the 0th state is adopted.
- the circuit 3a is operated alone, that is, switching between conduction / non-conduction of the switch element S1 is repeated to adopt the first state.
- the circuits 3a and 3b are operated cooperatively, that is, the switching of the switching elements S1 and S2 is repeatedly switched to adopt the third state.
- the efficiency can be increased in a wide range of the current I (in other words, a wide range of the load).
- the relationship between the current and the efficiency in the switching power supply circuit having only one circuit 3a is shown by a two-dot dashed curve.
- the circuit constants of the switching power supply circuit are set so that the efficiency peaks in the vicinity of the center (near the current value I2) in order to increase the efficiency in the entire region of the current I.
- the efficiency can be improved over a wide range of the current I by the above-described control method.
- the efficiency can be improved over a wide range of loads.
- the efficiency in a region where the load is small can be improved. Therefore, it is particularly effective for an air conditioner having a high frequency of operation below the intermediate capacity.
- the current matches the current value Iref1
- either the 0th state or the first state may be adopted, and when the current matches the current value Iref2, any of the first state or the third state may be adopted.
- the reference for switching the state is not necessarily limited to the current values Iref1 and Iref2, and there may be a slight deviation. Further, such a control method is not limited to the air conditioner, and may be employed in a switching power supply circuit mounted in another device.
- the efficiency in the first state can be further improved as illustrated in FIG.
- the curve indicated by the thin broken line represents the efficiency when the switch element S1 formed by the Si semiconductor is employed, and the curve indicated by the thick broken line is formed by the SiC semiconductor or the GaN semiconductor.
- the efficiency when the switch element S1 is employed is shown.
- the efficiency of the third state of the switch element S1 is increased by the SiC semiconductor or the GaN semiconductor, the illustration is omitted in FIG.
- control unit 5 controls the switch elements S1 and S2 in the first state to the third state based on the DC voltage command value as the command value for the DC voltage between the output terminals P3 and P4. More specifically, for example, the conduction period of the switch elements S1 and S2 may be determined based on a DC voltage command value.
- the control unit 5 employs the first state when the current I is larger than the current value Iref1, for example.
- the control unit 5 may adopt the first DC voltage command value A that is larger than the DC voltage between the output terminals P3 and P4 in the 0th state.
- the larger the current I that is, the larger the load
- the circuit 3 may be unstable. More specifically, for example, the DC voltage may be lower than the value required by the inverter. According to this control method, when the current I is larger than the current value Iref, a larger first DC voltage command value A is adopted.
- the inverter 4 drives the motor.
- the efficiency of the motor decreases as the current I increases. This is because the copper loss increases as the current I increases.
- copper loss can be reduced by increasing the DC voltage. If the DC voltage is increased, the efficiency of the circuit 3 is decreased. However, the decrease in the efficiency of the motor due to this is less than the increase in the efficiency of the motor by reducing the copper loss. Can be improved. Therefore, if the inverter 4 drives a motor, by using this control method, it is possible to suppress the reduction of the motor efficiency in the range where the current I is large or to increase the motor efficiency. Further, according to the present control method, as described above, the efficiency of the circuit can be improved in the range where the current I is small, and as a result, the motor efficiency in this range can be improved.
- the switch element S1 is a master-side switch element
- the switch element S2 is a slave-side switch element.
- the impedances of the reactors L1 and L2 are different from each other.
- the inductance of the reactor L1 is twice the inductance of the reactor L2.
- the number of turns of the coil which reactor L2 has can be made into the half of the number of turns of the coil which reactor L2 has. Therefore, the reactor L2 can be reduced in size and the manufacturing cost can be reduced.
- FIG. 9 shows an example of currents IL1, IL2, and I in such a switching power supply circuit.
- a switch signal is output to the switch element S1, and the switch element S1 becomes conductive. Therefore, the current IL1 increases at a predetermined slope between time t1 and time t3. Then, the output of the switch signal to the switch element S1 is stopped between time t3 and time t5. As a result, the switch element S1 becomes non-conductive, and the current IL1 decreases with a predetermined slope. Since the current IL1 reaches 0 at time t5, a switch signal is output to the switch element S1 again, and the switch element S1 becomes conductive. Thereafter, the above-described operation is repeated in the circuit 3a.
- the slope when the current IL1 increases and the slope when the current IL1 decreases are the same except for the positive and negative.
- the current IL1 when the conduction period and the non-conduction period of the switch element S1 are the same is illustrated. Therefore, in the illustration of FIG. 9, the current IL1 exhibits an isosceles triangle in one cycle T that is a period between time t1 and time t5.
- the current IL2 reaches 0 when half of one cycle T has elapsed (that is, at time t3).
- the switch signal is output again to the switch element S2, and the above-described operation is repeated.
- the current I which is the sum of them has a valley at the time.
- the minimum value of the current I takes the same 0 as the minimum value of the currents IL1 and IL2 at the time.
- the maximum value of current IL1 takes a value larger than the maximum value of each of currents IL1 and IL2 (here, 1.5 times that of currents IL1 and IL2) at times t2 and t4. Therefore, the difference between the maximum value and the minimum value of the current I that is the sum of the currents IL1 and IL2 is large (here, 1.5 times that of the currents IL1 and IL2), and harmonics are easily generated.
- the switch signal may be output to the switch element S2 only during the period from the time t4 to the time t5 in the one cycle T from the time t1 to the time t5.
- the current IL2 becomes a peak during the period when the current IL1 becomes a trough, and the current IL2 becomes zero during the period when the current IL1 becomes a peak. Therefore, the difference between the maximum value and the minimum value of the current I, which is the sum of the currents IL1 and IL2, can be reduced.
- the maximum value and the minimum value of the current I are 1 time and 0.5 times the maximum value of the current IL1 (or current IL2), respectively, and the difference is 0. 0 of the maximum value of the current IL1 (or current IL2). Reduce to 5 times.
- the switch element S2 when the inductance ratio of the reactors L1 and L2 is 2, as illustrated in FIG. 10, the switch element S2 is turned on only in the last quarter of one cycle T. If the ratio of the inductance of the reactor L1 to the reactor L2 is N (N is a number greater than 1), the switching element S2 only needs to be conducted only in the last 2N cycles of one cycle T. In other words, from the first time point (t1) when the switch element S1 is turned on to the second time point (t5) before the second time point (t5) when the switch element S1 is turned on, the third time point (t4) before the 1 / N period and the second time point The switch element S2 may be made conductive only between (t5).
- the inductances and current capacities of reactors L1 and L2 may be different from each other.
- the inductance of reactor L1 is smaller than the inductance of reactor L2, and the current capacity of reactor L1 is larger than the current capacity of reactor L2.
- the wire diameter of the coil which reactor L2 has can be made smaller than the wire diameter of the coil which reactor L2 has. Therefore, even if the inductance of the reactor L2 is large, the reactor L2 can be prevented from being enlarged, and the manufacturing cost can be reduced.
- FIG. 11 shows an example of currents IL1, IL2, and I in such a switching power supply circuit.
- the switching of the switch element S1 in the circuit 3a is the same as the switching according to the description with reference to FIG.
- a switch signal is output to the switch element S2 in a period from time t3 when the half cycle has elapsed from time t1 when the switch element S1 is turned on to time t5 (that is, a half cycle). Conducted. Therefore, the current IL2 increases during the period from time t3 to time t5.
- the inductance of reactor L2 is larger than the inductance of reactor L1, and the conduction periods of switching elements S1 and S2 are the same, so the slope of current IL2 is lower than the slope of current IL1. Therefore, the maximum value of current IL2 is lower than the maximum value of current IL1.
- the circuits 3a and 3b can be operated in cooperation without causing a malfunction of the reactor L2.
- the current capacity of the reactor L2 is set larger than the current IL2.
- the inductance of the reactor L2 may be close to the inductance of the reactor L1.
- the slope of the current IL2 is lowered, and consequently the maximum value of the current IL2 is lowered.
- Such currents IL1, IL2, and I are illustrated in FIG.
- the maximum value of the current IL2 decreases, the maximum value of the current I also decreases. Therefore, the difference between the maximum value and the minimum value of current I can be reduced, and the current capacity of reactor L2 can be reduced.
- the conduction time of the switch elements S1 and S2 may be shifted. In other words, the phase difference between the currents IL1 and IL2 may be changed.
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Abstract
Description
図1に例示するように、スイッチング電源回路は複数の回路3と入力端P1,P2と出力端P3,P4とを備えている。
本スイッチング電源回路においてスイッチ素子S2を非導通とし、回路3aを単独で動作させることができる。同様にスイッチ素子S1を非導通とし、回路3bを単独で動作させることができる。まず回路3aの単独運転について説明する。
本スイッチング電源回路においては回路3a,3bを協働して運転させることができる。かかる運転はインターリーブとも呼ばれる。以下の詳細に説明する。
本実施の形態では、それぞれ回路3a,3bに属するスイッチ素子S1,S2の特性、ダイオードD11,D21の特性、及びリアクトルL1,L2の特性の少なくともいずれかは、互いに相違している。これにより、回路3aの電気特性と回路3bの電気特性とが相違する。
次に本スイッチング電源回路が空気調和機に設けられる場合について考慮する。このとき図2の例示と同様に本スイッチング電源回路はインバータ4の入力側に設けられ、インバータ4は不図示のモータへと交流電圧を印加してモータの回転速度を制御する。モータは空気調和機が有する圧縮機又はファンを駆動する。
制御部5は上述したように第1状態~第3状態において、出力端P3,P4の間の直流電圧についての指令値たる直流電圧指令値に基づいて、スイッチ素子S1,S2を制御する。より詳細には例えば直流電圧指令値に基づいて、スイッチ素子S1,S2の導通期間を決定してもよい。
次に、デバイス特性の異なる回路3a,3bを協働して動作させる際のスイッチングの例について述べる。以下では、スイッチ素子S1をマスター側のスイッチ素子とし、スイッチ素子S2をスレーブ側のスイッチ素子とする。
L1,L2 リアクトル
LH1,LH2,LL 電源線
P1,P2 入力端
P3,P4 出力端
S1,S2 スイッチ素子
Claims (18)
- 第1および第2の入力端(P1,P2)と、
第1および第2の出力端(P3,P4)と、
前記第2の入力端及び前記第2の出力端を接続する第2の電源線(LL)と、
複数の回路(3,3a,3b)と
を備え、
前記複数の回路の各々は、
前記第1の入力端と前記第1の出力端とを接続する第1電源線(LH1,LH2)と、
前記第1電源線上に設けられるリアクトル(L1,L2)と、
前記第1電源線上で前記リアクトルと直列に接続され、アノードを前記リアクトル側に向けて配置されるダイオード(D11,D21)と、
前記リアクトルと前記ダイオードとの間の点と、前記第2電源線との間に設けられるスイッチ素子(S1,S2)と
を有し、
一の前記複数の回路と他の一の前記複数の回路とにそれぞれ属する前記リアクトル、前記スイッチ素子及び前記ダイオードの少なくともいずれか一つの特性が相互に異なる、スイッチング電源回路。 - 前記一の前記複数の回路(3a)に属する前記スイッチ素子(S1)は絶縁ゲートバイポーラトランジスタであって、前記他の一の前記複数の回路(3b)に属する前記スイッチ素子(S1)はMOS電界効果トランジスタである、請求項1に記載のスイッチング電源回路。
- 前記一の前記複数の回路(3a)に属する前記スイッチング素子(S1)は炭化珪素半導体或いは窒化ガリウム半導体によって形成され、前記他の一の前記複数の回路(3b)に属する前記スイッチング素子(S2)は珪素半導体によって形成される、請求項1に記載のスイッチング電源回路。
- 前記一の複数の回路(3a)に属する前記リアクトル(L1)のインピーダンスは、前記他の一の前記複数の回路(3b)に属する前記リアクトルのインピーダンスよりも小さい、請求項1に記載のスイッチング電源回路。
- 前記一の複数の回路(3a)に属する前記リアクトル(L1)のインピーダンスは、前記他の一の前記複数の回路(3b)に属する前記リアクトルのインピーダンスよりも小さい、請求項2に記載のスイッチング電源回路。
- 前記一の複数の回路(3a)に属する前記リアクトル(L1)のインピーダンスは、前記他の一の前記複数の回路(3b)に属する前記リアクトルのインピーダンスよりも小さい、請求項3に記載のスイッチング電源回路。
- 請求項1から6のいずれか一つに記載のスイッチング電源回路の制御方法であって、
前記複数の回路(3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持する第1工程と、
前記第1及び前記第2の入力端(P1,P2)を流れる電流が第1所定値(Iref1)を超えたときに、前記一の前記複数の回路(3a)に属する前記スイッチング素子(S1)の導通/非導通を繰り返し切り替える第2工程と
を実行する、スイッチング電源回路の制御方法。 - 前記第2工程において、前記第1工程における前記第1及び前記第2の出力端(P3,P4)の間の電圧よりも高い第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)に属する前記スイッチング素子(S1)の導通/非導通を繰り返し切り替える、請求項7に記載のスイッチング電源回路の制御方法。
- 前記第1及び前記第2の入力端(P1,P2)を流れる電流が、前記第1所定値(Iref1)よりも大きい第2所定値(Iref2)を超えたときに、前記一の前記複数の回路(3a)および前記他の一の前記複数の回路(3b)にそれぞれ属する前記スイッチング素子(S1,S2)の導通/非導通を繰り返し切り替える第3工程
を更に実行する、請求項7に記載のスイッチング電源回路の制御方法。 - 前記第1及び前記第2の入力端(P1,P2)を流れる電流が、前記第1所定値(Iref1)よりも大きい第2所定値(Iref2)を超えたときに、前記一の前記複数の回路(3a)および前記他の一の前記複数の回路(3b)にそれぞれ属する前記スイッチング素子(S1,S2)の導通/非導通を繰り返し切り替える第3工程
を更に実行する、請求項8に記載のスイッチング電源回路の制御方法。 - 前記第3工程において、前記第1の直流電圧指令値よりも大きい第2の直流電圧指令値に基づいて前記一の前記複数の回路(3a)および前記他の一の前記複数の回路(3b)にそれぞれ属する前記スイッチング素子(S1,S2)の導通/非導通を繰り返し切り替える、請求項9に記載のスイッチング電源回路の制御方法。
- 前記第3工程において、前記第1の直流電圧指令値よりも大きい第2の直流電圧指令値に基づいて前記一の前記複数の回路(3a)および前記他の一の前記複数の回路(3b)にそれぞれ属する前記スイッチング素子(S1,S2)の導通/非導通を繰り返し切り替える、請求項10に記載のスイッチング電源回路の制御方法。
- 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項7に記載のスイッチング電源回路の制御方法。 - 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項8に記載のスイッチング電源回路の制御方法。 - 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項9に記載のスイッチング電源回路の制御方法。 - 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項10に記載のスイッチング電源回路の制御方法。 - 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項11に記載のスイッチング電源回路の制御方法。 - 前記第1工程及び前記第2工程に先立って、
前記複数の回路(3,3a,3b)に属する前記スイッチング素子(S1,S2)の全てを非導通に維持し、前記電流と前記スイッチング電源回路の効率との間の第1の関係を求める第4工程と、
前記第1の直流電圧指令値に基づいて前記一の前記複数の回路(3a)の前記スイッチング素子(S1)の導通/非導通を繰り返し切り替え、前記電流と前記スイッチング電源回路の効率との間の第2の関係を求める第5工程と
が更に実行され、
前記第1所定値(Iref1)として、前記第1関係と前記第2関係のいずれにおいても同じ前記効率を与える前記電流が採用される、請求項12に記載のスイッチング電源回路の制御方法。
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ES11759514T ES2847882T3 (es) | 2010-03-26 | 2011-03-24 | Circuito de alimentación de energía de conmutación y método para controlar el circuito de alimentación de energía de conmutación |
CN201180014255.8A CN102804576B (zh) | 2010-03-26 | 2011-03-24 | 开关电源电路和开关电源电路的控制方法 |
EP11759514.0A EP2555399B1 (en) | 2010-03-26 | 2011-03-24 | Switching power supply circuit, and method for control of switching power supply circuit |
BR112012024228-8A BR112012024228B1 (pt) | 2010-03-26 | 2011-03-24 | método para controle de circuito de fornecimento de energia comutável |
AU2011230323A AU2011230323B2 (en) | 2010-03-26 | 2011-03-24 | Switching power supply circuit, and method for control of switching power supply circuit |
KR1020127021692A KR101346542B1 (ko) | 2010-03-26 | 2011-03-24 | 스위칭 전원 회로 및 스위칭 전원 회로의 제어 방법 |
US13/579,955 US8994343B2 (en) | 2010-03-26 | 2011-03-24 | Switching power supply circuit, and method for control of switching power supply circuit |
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JP7230735B2 (ja) | 2018-08-10 | 2023-03-01 | 株式会社デンソー | 車両用電力変換装置 |
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AU2011230323A1 (en) | 2012-09-06 |
CN102804576B (zh) | 2016-04-27 |
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EP2555399A4 (en) | 2017-12-13 |
US8994343B2 (en) | 2015-03-31 |
KR101346542B1 (ko) | 2013-12-31 |
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CN102804576A (zh) | 2012-11-28 |
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US20120313614A1 (en) | 2012-12-13 |
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