WO2015165191A1 - 一种三相双模式逆变器的稳态控制方法 - Google Patents

一种三相双模式逆变器的稳态控制方法 Download PDF

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WO2015165191A1
WO2015165191A1 PCT/CN2014/086501 CN2014086501W WO2015165191A1 WO 2015165191 A1 WO2015165191 A1 WO 2015165191A1 CN 2014086501 W CN2014086501 W CN 2014086501W WO 2015165191 A1 WO2015165191 A1 WO 2015165191A1
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grid
phase
inverter
circuit
voltage
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PCT/CN2014/086501
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English (en)
French (fr)
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罗安
陈燕东
王明玥
钟庆昌
周乐明
黄媛
陈智勇
周小平
匡慧敏
李鸣慎
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湖南大学
长沙博立电气有限公司
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Priority to US15/307,809 priority Critical patent/US9837931B2/en
Publication of WO2015165191A1 publication Critical patent/WO2015165191A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/46—Controlling of the sharing of output between the generators, converters, or transformers
    • H02J3/48—Controlling the sharing of the in-phase component
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/126—Arrangements for reducing harmonics from ac input or output using passive filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to the field of distributed power generation in a micro-grid, in particular to a steady-state control method for a three-phase dual-mode inverter.
  • the adoption and utilization of distributed power sources (photovoltaics, wind power, fuel cells, etc.) through the microgrid is an effective way to solve the current energy crisis and environmental degradation.
  • the inverter in the microgrid serves as the interface between the distributed power source and the microgrid, and will be distributed energy. Turning into high-quality electrical energy, it is of great significance to study inverters suitable for microgrid operation.
  • the existing research mainly focuses on the control method in which the inverter works alone in the grid-connected mode or off-grid (island) mode, but for the inverter that can work in the grid-connected and off-grid dual mode, and the corresponding smoothing
  • the research on switching control is not perfect.
  • the inverter In a highly flexible microgrid, the inverter should have the ability to simultaneously supply power to the local load and the grid, and in exceptional circumstances, the inverter can disconnect from the grid and directly supply power to the local load.
  • the micro-source and the grid cannot be strictly synchronized, and the control schemes are different, which will generate instantaneous overvoltage or overcurrent.
  • the three phases of the thyristor (SCR) or solid state relay (SSR) cannot be simultaneously Turning off, causing voltage or current imbalance, so that the active or reactive power increases and fluctuates, and finally the DC side voltage cannot be stabilized.
  • the technical problem to be solved by the present invention is to provide a steady-state control method for a three-phase dual-mode inverter according to the deficiencies of the prior art, overcome the shortcomings of the existing dual-mode inverter control, and solve the inverter in the micro-grid.
  • the energy backflow phenomenon due to the phase error ensures the stability of the system operation.
  • the technical solution adopted by the present invention is: a steady-state control method for a three-phase dual-mode inverter, which is suitable for a micro-grid dual-mode inverter parallel system, the micro-grid dual-mode inverter
  • the parallel system includes a plurality of dual mode inverters, a parallel/off-grid switch, a three-phase power grid and an inverter control circuit
  • the dual mode inverter includes a DC storage capacitor, a three-phase inverter circuit, and an inverter control circuit
  • the LC filter circuit, the DC storage capacitor, the three-phase inverter circuit, and the LC filter circuit are sequentially connected, and the LC filter circuit is connected to the line impedance, and the line resistance is
  • the anti-internal switch is connected to the parallel/off-grid switch, and the parallel/off-grid switch is connected to the three-phase power grid
  • the inverter control circuit includes a sampling and conditioning circuit, a phase-locked loop circuit, a controller, and
  • the off-grid mode steady state control method is:
  • the sampling conditioning circuit pairs the three-phase grid voltage u sa , the DC storage capacitor voltage u dc , the LC filter circuit capacitor voltage u oa , u ob , u oc , the line current i oa , i Ob , i oc , incoming current i sa , i sb , and i sc are respectively sampled, and then the sampled data is sent to the controller for processing, the effective value of each sampled value is calculated, and the LC filter circuit capacitor voltage u oa , u ob , u oc and line currents i oa , i ob , i oc are respectively converted into LC filter circuit capacitor voltages u o ⁇ , u o ⁇ and line currents i o ⁇ , i o ⁇ in ⁇ coordinates;
  • K is the control quantity constant, the value range is 0 ⁇ K ⁇ 1, L f is the LC filter circuit inductance value, T c is the PWM carrier period; u dc (k) is the DC side of the dual mode inverter sampled at time k The voltage, u o ⁇ (k), u o ⁇ (k) is the LC filter circuit capacitor voltage at the ⁇ coordinate at time k, i o ⁇ (k), i o ⁇ (k) is the line current at the ⁇ coordinate at time k, a predicted reference value of the line current at the ⁇ coordinate of the k+1 time;
  • the d a , d b , d c and the triangular carrier are bipolarly modulated to obtain the duty cycle signal of the three-phase inverter circuit switch tube, and the driving protection circuit is used to control the opening and closing of the three-phase inverter circuit switch tube.
  • the grid-connected mode steady state control method is:
  • I o1 and I s1 are the output current RMS value and the grid current RMS value of the dual-mode inverter after the power change
  • I o2 and I s2 are the output current RMS and the grid current of the dual-mode inverter before the power change.
  • R', X' are the line impedance and inductive reactance of the dual-mode inverter to the AC bus, respectively, and R" and X" are the line impedance and the inductive reactance of the local load to the AC bus, respectively.
  • the present invention has the beneficial effects that the present invention proposes a steady state control strategy for a dual mode inverter, the steady state control is controlled by the outer loop droop, the voltage quasi-resonant control, and the current inner loop.
  • the beat control is constructed to achieve steady state control in the on/off mode. Since the output voltage of the inverter in the grid-connected mode is always slightly ahead of the grid voltage, the invention introduces phase lead control, which avoids the energy backflow phenomenon caused by the phase error of the inverter in the grid-connected mode, and ensures the micro-inversion.
  • the source can continuously deliver energy to the grid, achieving stable operation in the grid-connected mode.
  • the invention can be widely applied in the microgrid control system, and the effect in the high power system is particularly remarkable.
  • FIG. 1 is a schematic diagram of a parallel structure of a micro-grid dual-mode inverter according to an embodiment of the present invention
  • FIG. 2 is a block diagram of steady state control of a dual mode inverter according to an embodiment of the present invention
  • FIG. 3 is a schematic diagram showing changes in output frequency of an inverter according to an embodiment of the present invention
  • FIG. 3(a) is a change in a droop curve when power is changed
  • FIG. 3(b) is a change in frequency and phase.
  • FIG. 1 is a schematic diagram of a parallel structure of a micro-grid dual-mode inverter according to an embodiment of the present invention, which mainly includes: a full-bridge inverter circuit, a filter, a local load, a parallel/off-grid switch, a power grid, and the like.
  • the distributed power is converted to a constant voltage DC with a voltage of U dc ; the DC is converted to AC by a three-phase PWM inverter circuit; the filter is used to filter out glitch caused by the high frequency switch, u o , i o is a filter
  • the rear inverter outputs voltage and current; the output AC power supplies power to the local load Z load , and the / off-grid switch S is used to connect the micro source to the grid.
  • the voltage of the common connection point (PCC) is u s
  • the grid voltage mentioned in the present invention refers to the voltage at the PCC.
  • FIG. 2 is a block diagram of steady state control of a dual mode inverter according to an embodiment of the present invention.
  • k U is the voltage feedback coefficient
  • k F is the reference voltage feed forward coefficient
  • ⁇ is the inverter leading the grid voltage phase.
  • the dual mode steady state control method is:
  • the sampling conditioning circuit pairs the three-phase grid voltage u sa , the DC storage capacitor voltage u dc , the LC filter circuit capacitor voltage u oa , u ob , u oc , the line current i oa , i Ob , i oc , incoming current i sa , i sb , and i sc are respectively sampled, and then the sampled data is sent to the controller for processing, the effective value of each sampled value is calculated, and the LC filter circuit capacitor voltage u oa , u ob , u oc and line currents i oa , i ob , i oc are respectively converted into LC filter circuit capacitor voltages u o ⁇ , u o ⁇ and line currents i o ⁇ , i o ⁇ in ⁇ coordinates;
  • ⁇ * and U * are the real-time values of the grid voltage angular frequency and amplitude, which vary according to the change of the grid voltage; in the off-grid mode, ⁇ * and U * are set values;
  • Inverter output voltage phase in off-grid mode Zero.
  • the phase of the grid voltage is detected by the phase-locked loop circuit.
  • the leading phase angle is ⁇ , then the output voltage phase equal:
  • K is the control quantity constant, the value range is 0 ⁇ K ⁇ 1, L f is the filter inductance value, T c is the PWM carrier period;
  • u dc (k) is the sampling DC side voltage value at time k, u o ⁇ (k ), u o ⁇ (k) is the value of the grid voltage sampled at time k at the ⁇ coordinate,
  • i o ⁇ (k), i o ⁇ (k) are the values of the output current of the sampled inverter at the time ⁇ at the ⁇ coordinate.
  • i o ⁇ (k+1), i o ⁇ (k+1) are the values of the reference current of the k+1 time at the ⁇ coordinate, respectively;
  • the SPWM modulated wave signals d a , d b , d c are bipolarly modulated with the triangular carrier to obtain the duty cycle signal of the fully controlled power device, and the driving protection circuit is used to control the opening of the fully controlled power device. And shut down.
  • the steady state control of the grid connection mode increases the phase advance control link.
  • the inverter In order to ensure the normal energy flow between the micro-source and the grid, and avoid the energy backflow caused by the phase error of the inverter, the inverter should lead the grid to a small phase ⁇ . When the output power changes, the phase ⁇ also needs to change. Adjusting the phase ⁇ ensures the accuracy of the control and enhances the stability of the inverter in the grid-connected operating mode.
  • the phase ⁇ calculation formula is:
  • the initial value ⁇ 0 ranges from 0.0001 to 0.15, which is related to the rated power of the dual mode inverter.
  • ⁇ is the phase compensation parameter, which is related to the inverter output power.
  • I o1 and I s1 are the output current rms value and the grid-connected current RMS value of the dual-mode inverter after the power change
  • I o2 and I s2 are the output current RMS value and the grid-connected current of the dual-mode inverter before the power change.
  • the effective values, R', X' are the line impedance and inductive reactance of the dual-mode inverter to the AC bus, respectively
  • R" and X" are the line impedance and inductive reactance of the local load to the AC bus, respectively.
  • FIG. 3 is a schematic diagram showing changes in output frequency of an inverter according to an embodiment of the present invention.
  • Fig. 3(a) shows the change in the droop curve when the power changes
  • Fig. 3(b) shows the frequency and phase change.
  • the output voltage of the inverter must be ahead of the grid voltage due to the impedance of the transmission line. Therefore, in order to ensure that the micro-source continuously delivers energy to the grid without causing backflow, it is necessary to ensure that the phase of the micro-source always leads the grid.
  • the phase of the micro-source slightly exceeds the phase of the grid to facilitate the flow of power.
  • the phase-locked loop does not act on the droop control and only provides real-time frequency and phase reference.
  • the increase in output power is taken as an example.
  • the droop characteristic curve needs to change from point a to point d under the condition of constant frequency.
  • the phase difference between the micro-source and the power grid needs to be increased due to the increase in current. If a delay strategy is used to change the phase angle, it takes a delay of nearly one cycle, during which a large number of harmonics are introduced, and the phase angle cannot be changed frequently by delay.
  • the present invention changes the phase angle by changing the frequency.
  • the adjustment process always keeps the droop coefficient constant.
  • the droop characteristic curve shown in Fig. 3(a) when the frequency of the micro source increases, the operating point changes from a to b, and the control only needs to change the rated frequency, and the output power does not change; when the frequency is maintained at the maximum value, Increase the rated power, the working point changes from b to c; the last working point changes from c to d, only changing the rated frequency, the output power of the micro-source can be increased, the phase adjustment is completed, and the frequency before and after the adjustment phase is guaranteed to be constant.
  • This adjustment does not introduce harmonics.
  • the frequency changes it is guaranteed to be in the range of 49.8 ⁇ 50.2Hz.
  • the vector relationship analysis shows that the leading phase ⁇ needs to be adjusted.
  • the phase to be adjusted is set to ⁇ , and in order to avoid system oscillation caused by direct frequency change, the present invention performs phase advance adjustment by means of frequency adjustment.
  • the instantaneous frequency output varies according to the following formula:
  • T a is the adjustment time and f o is the system output frequency before adjustment.

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Abstract

一种三相双模式逆变器的稳态控制方法，离网模式稳态控制由外环功率下垂控制、电压前馈的准谐振控制以及电流内环无差拍控制构成，加快了逆变器的响应速度，抑制了微电网负荷波动的影响；并网模式稳态控制在离网模式稳态控制的基础上，通过引入相位超前控制到功率下垂控制环节中，使得逆变器的输出电压始终略超前于电网电压，避免了逆变器因相位误差造成的能量倒灌现象，实现了并网模式下的稳定可靠运行。

Description

一种三相双模式逆变器的稳态控制方法 技术领域
本发明涉及微电网分布式发电领域，特别是一种三相双模式逆变器的稳态控制方法。
背景技术
通过微电网接纳与利用分布式电源(光伏、风力、燃料电池等)是解决当前能源危机与环境恶化的有效途径，微电网中逆变器作为分布式电源与微电网的接口，将分布式能源转变成高质量的电能，因此研究适合微电网运行的逆变器意义重大。现有研究主要集中在逆变器单独工作在并网模式或离网(孤岛)模式下的控制方法，但对于能够在并网、离网双模式下工作的逆变器，及其相应的平滑切换控制的研究尚不完善。
在一个高度灵活的微电网中，逆变器应该具备同时向本地负载和电网提供电能的能力，且在异常情况下，逆变器能够断开与电网的连接，直接向本地负载供电。并网切换时，由于微源与电网不能严格同步，以及控制方案存在差异，会产生瞬间的过压或过流；离网切换时，由于晶闸管(SCR)或固态继电器(SSR)三相不能同时关断，引起电压或电流不平衡，使得有功或无功功率加大波动，最终导致直流侧电压不能稳定。这些均影响负载正常工作，破坏了供电的可靠性和稳定性。现有的双模式逆变器通常在并网模式采用P/Q控制，在离网模式采用V/f控制。虽然采取了过渡控制，在切换过程中抑制了逆变器输出电流冲击，但其往往只关注逆变器的输出电压、电流以及功率的波动，没有兼顾减小入网电流冲击。而并网和离网模式下的下垂控制，能够实现***的稳态控制，保证负载和功率的均分。但对于其双模式的切换控制，却只能依赖以往已有的平滑切换方式，不能根据下垂控制的特性进行平滑切换，缺乏针对性。
发明内容
本发明所要解决的技术问题是，针对现有技术不足，提供一种三相双模式逆变器的稳态控制方法，克服现有双模式逆变器控制的缺点，解决微电网中逆变器在并网模式下，因相位误差出现的能量倒灌现象，保证***运行的稳定性。
为解决上述技术问题，本发明所采用的技术方案是：一种三相双模式逆变器的稳态控制方法，适用于微电网双模式逆变器并联***，所述微电网双模式逆变器并联***包括多个双模式逆变器、并/离网开关、三相电网和逆变控制电路；所述双模式逆变器包括直流储能电容、三相逆变电路、逆变控制电路、LC滤波电路，所述直流储能电容、三相逆变电路、LC滤波电路依次连接，所述LC滤波电路与线路阻抗连接，所述线路阻 抗通过交流母线与并/离网开关连接，所述并/离网开关接入三相电网；所述逆变控制电路包括采样调理电路、锁相环电路、控制器、驱动保护电路；所述采样调理电路输入端与所述LC滤波电路连接；所述控制器与所述驱动保护电路输入端、采样调理电路输出端、锁相环电路输出端连接；所述锁相环电路输入端与所述交流母线连接；该方法包括离网模式稳态控制方法和并网模式稳态控制方法；
所述离网模式稳态控制方法为：
1)在每个采样周期的起始点，采样调理电路对三相电网电压usa、直流储能电容电压udc、LC滤波电路电容电压uoa、uob、uoc、线路电流ioa、iob、ioc、入网电流isa、isb、isc分别进行采样，然后将采样数据送给控制器进行处理，计算各采样值的有效值，并将LC滤波电路电容电压uoa、uob、uoc和线路电流ioa、iob、ioc分别转换为在αβ坐标下的LC滤波电路电容电压uoα、uoβ和线路电流ioα、ioβ；
2)将LC滤波电路电容电压uoa、uob、uoc与线路电流ioa、iob、ioc分别相乘，得到双模式逆变器的有功功率P和无功功率Q；
3)对双模式逆变器输出电压幅值参考值U*、角频率参考值ω*、有功功率参考值P*、无功功率参考值Q*，以及上述有功功率平P、无功功率Q进行下垂控制运算，得到双模式逆变器的输出电压幅值Uo和角频率ωo；其中，并网模式下，ω*、U*为电网电压角频率和幅值的实时值，根据电网电压的变化而变化；离网模式下，ω*、U*为设定值；
4)由双模式逆变器输出电压有效值Uo、角频率ωo，以及电压输出相位
Figure PCTCN2014086501-appb-000001
合成αβ坐标下的参考电压urefα、urefβ，其中t为采样时间：
Figure PCTCN2014086501-appb-000002
5)将urefα、urefβ分别减去uoα与反馈系数kU的乘积、uoβ与反馈系数kU的乘积， 得到的差值作为准谐振QPR控制器的输入，其中反馈系数kU的取值范围为0.1～2；
6)引入参考电压前馈环节kF·urefα *、kF·urefβ *，将kF·urefα *、kF·urefβ *与准谐振QPR控制器的输出相加，得到线路电流的参考值irefα、irefβ，其中kF为电压前馈系数，取值范围为0.01～5；
7)对线路电流参考值irefα、irefβ，LC滤波电路电容电压uoα、uoβ，直流储能电容电压udc进行电流无差拍控制，得到三相逆变电路开关管控制量dα、dβ：
Figure PCTCN2014086501-appb-000003
其中，K为控制量常数，取值范围0<K<1，Lf为LC滤波电路电感值，Tc为PWM载波周期；udc(k)为k时刻采样的双模式逆变器直流侧电压，uoα(k)、uoβ(k)为k时刻的αβ坐标下的LC滤波电路电容电压，ioα(k)、ioβ(k)为k时刻αβ坐标下线路电流，
Figure PCTCN2014086501-appb-000004
为k+1时刻αβ坐标下的线路电流的预测参考值；
8)对dα、dβ进行坐标变化，得到稳态控制时abc坐标下的三相逆变电路开关管控制量da、db、dc；
9)将da、db、dc与三角载波进行双极性调制，得到三相逆变电路开关管的占空比信号，经驱动保护电路，控制三相逆变电路开关管的开通与关断；
所述并网模式稳态控制方法为：
将上述离网模式稳态控制方法步骤4)中的电压输出相位调整为
Figure PCTCN2014086501-appb-000005
然后根据上述步骤1)～步骤9)的方法控制三相逆变电路开关管的开通与关断；其中，
Figure PCTCN2014086501-appb-000006
表示锁相环电路检测到的三相电网电压相位，δ＝δ0±Δδ，δ0取值范围为0.0001～0.15；Δδ为相位补偿参数，
Figure PCTCN2014086501-appb-000007
其中， Io1、Is1分别为功率变化后双模式逆变器的输出电流有效值和入网电流有效值，Io2、Is2为功率变化前双模式逆变器的输出电流有效值和入网电流有效值，R'、X'分别为双模式逆变器到交流母线的线路阻抗和感抗，R"、X"分别为本地负载到交流母线的线路阻抗和感抗。
与现有技术相比，本发明所具有的有益效果为：本发明提出了一种双模式逆变器稳态控制策略，稳态控制由外环下垂控制、电压准谐振控制以及电流内环无差拍控制构成，实现了并网/离网模式下的稳态控制。由于逆变器在并网模式下的输出电压始终略超前于电网电压，本发明引入了相位超前控制，避免了逆变器在并网模式下，因相位误差出现的能量倒灌现象，保证了微源能够持续不断向电网输送能量，实现了并网模式下的稳定运行。本发明可广泛应用在微电网控制***中，在大功率***中的效果尤为显著。
附图说明
图1为本发明一实施例微电网双模式逆变器并联结构示意图；
图2为本发明一实施例双模式逆变器稳态控制框图；
图3为本发明一实施例逆变器输出频率变化示意图；图3(a)为功率变化时下垂曲线变化；图3(b)为频率及相位变化。
具体实施方式
图1为本发明一实施例微电网双模式逆变器并联结构示意图，主要包括：全桥逆变电路、滤波器、本地负载、并/离网开关、电网等。分布式电源被转换为电压恒定的直流电，电压为Udc；直流电通过三相PWM逆变电路转换为交流电；滤波器用于滤除由高频开关引起的毛刺，uo、io为经过滤波器后的逆变器输出电压和电流；输出交流电向本地负载Zload提供电能，并/离网开关S用于连接微源与电网。其中，公共连接点(PCC)的电压为us，本发明中提到的电网电压均指PCC处电压。
图2为本发明一实施例双模式逆变器稳态控制框图。图中kU为电压反馈系数，kF为参考电压前馈系数；
Figure PCTCN2014086501-appb-000008
为输出电压相位，δ为逆变器超前电网电压相位。所述双模式稳态控制方法为：
1)在每个采样周期的起始点，采样调理电路对三相电网电压usa、直流储能电容电压udc、LC滤波电路电容电压uoa、uob、uoc、线路电流ioa、iob、ioc、入网电流isa、isb、isc分别进行采样，然后将采样数据送给控制器进行处理，计算各采样值的有效值，并将 LC滤波电路电容电压uoa、uob、uoc和线路电流ioa、iob、ioc分别转换为在αβ坐标下的LC滤波电路电容电压uoα、uoβ和线路电流ioα、ioβ；
2)将LC滤波电路电容电压uoa、uob、uoc，与线路电流ioa、iob、ioc分别相乘，得到逆变器的有功功率P、无功功率Q；
3)将逆变器输出电压幅值参考值U*、角频率参考值ω*、有功功率参考值P*、无功功率参考值Q*，以及上述有功功率P、无功功率Q，进行下垂控制运算，得到逆变器输出电压幅值Uo和角频率ωo。其中，并网模式下，ω*、U*为电网电压角频率和幅值的实时值，根据电网电压的变化而变化；离网模式下，ω*、U*为设定值；
4)离网模式下，逆变器输出电压相位
Figure PCTCN2014086501-appb-000009
为零。并网模式下，通过锁相环电路检测到电网电压相位为
Figure PCTCN2014086501-appb-000010
通过相位超前控制取超前相位角为δ，则此时输出电压相位
Figure PCTCN2014086501-appb-000011
等于：
Figure PCTCN2014086501-appb-000012
5)由双模式逆变器输出电压有效值Uo、角频率ωo，以及电压输出相位
Figure PCTCN2014086501-appb-000013
合成αβ坐标下的参考电压urefα、urefβ，其中t为采样时间：
Figure PCTCN2014086501-appb-000014
6)将urefα、urefβ分别减去uoα与反馈系数kU的乘积、uoβ与反馈系数kU的乘积，得到的差值作为准谐振QPR控制器的输入，其中反馈系数kU的取值范围为0.1～2；
7)引入参考电压前馈环节kF·urefα *、kF·urefβ *，将kF·urefα *、kF·urefβ *与准谐振QPR控制器的输出相加，得到线路电流的参考值irefα、irefβ，其中kF为电压前馈系数，取值范围为0.01～5；
8)对线路电流参考值irefα、irefβ，LC滤波电路电容电压uoα、uoβ，直流储能电容电压udc进行电流无差拍控制，得到三相逆变电路开关管控制量dα、dβ：
Figure PCTCN2014086501-appb-000015
其中，K为控制量常数，取值范围0<K<1，Lf为滤波器电感值，Tc为PWM载波周期；udc(k)为k时刻采样直流侧电压值，uoα(k)、uoβ(k)分别为k时刻所采样电网电压在αβ坐标下的值，ioα(k)、ioβ(k)分别为k时刻所采样逆变器输出电流在αβ坐标下的值，ioα(k+1)、ioβ(k+1)分别为k+1时刻参考电流在αβ坐标下的值；
9)将dα、dβ进行坐标变化，得到稳态控制时abc坐标下的开关管控制量da、db、dc；
10)将SPWM调制波信号da、db、dc，与三角载波进行双极性调制，得到全控型功率器件的占空比信号，经驱动保护电路，控制全控型功率器件的开通与关断。
所述的双模式逆变器稳态控制策略，步骤4)中，并网模式稳态控制增加了相位超前控制环节。为保证微源和电网之间正常的能量流动，避免逆变器因相位误差造成能量倒灌现象，逆变器要超前电网一个很小的相位δ。当输出功率变化时，相位δ也需要变化。调节相位δ能够保证控制的精准性，增强并网运行模式下逆变器的稳定性。相位δ计算公式为：
δ＝δ0±Δδ
当输出功率需要增大时，上式取“+”；当输出功率需要减小时，上式取“-”。其中，初始值δ0取值范围为0.0001～0.15，该值与双模式逆变器的额定功率大小有关。Δδ为相位补偿参数，其与逆变器输出功率有关，计算公式为：
Figure PCTCN2014086501-appb-000016
其中，Io1、Is1分别为功率变化后双模式逆变器的输出电流有效值和入网电流有效值，Io2、Is2为功率变化前双模式逆变器的输出电流有效值和入网电流有效值，R'、 X'分别为双模式逆变器到交流母线的线路阻抗和感抗，R"、X"分别为本地负载到交流母线的线路阻抗和感抗。
图3为本发明一实施例逆变器输出频率变化示意图。图3(a)为功率变化时下垂曲线变化，图3(b)为频率及相位变化。无论本地负载呈感性还是容性，由于输电线路阻抗等原因，逆变器的输出电压都要超前于电网电压。因此，为保证微源持续不断地向电网输送能量，而不导致倒灌，就要保证微源的相位始终超前于电网。
在并网情况下，微源相位略超前电网相位有利于功率的流动，锁相环不作用于下垂控制，仅提供实时频率和相位参考。如图3(a)所示，以输出功率增大为例。当输出功率需要增大时，在频率不变的条件下，下垂特性曲线需要从a点变化到d点。由图3(b)可知，输出电压不变时，由于电流增大，微源与电网的相位差也需要增大。如果采用延时策略来改变相角，需要延时近一个周期，期间会引入大量谐波，且不能频繁通过延时来改变相角。本发明通过改变频率来改变相角。
调整过程始终保持下垂系数不变。根据图3(a)所示的下垂特性曲线，当微源频率增大时，工作点由a变为b，控制上只需改变额定频率，输出功率不变；当频率维持在最大值时，增大额定功率，工作点由b变为c；最后工作点由c变为d，只改变额定频率，即可使微源的输出功率增大，完成相位调节，保证调节相位前后频率不变。这样调节不会引入谐波。频率变化时，保证其始终在49.8～50.2Hz范围内。
当逆变器在并网模式下的功率需要变化时，由矢量关系分析得出，超前相位δ需要调节。设待调节相位为Δδ，为避免频率直接变化引起的***振荡，本发明通过频率调节的方式进行相位超前调节。瞬时频率输出根据下式变化：
Figure PCTCN2014086501-appb-000017
其中，Ta为调节时间，fo为调节前***输出频率。当输出功率需要增大时，上式取“+”；当输出功率需要减小时，上式取“-”。

Claims (1)

  1. 一种三相双模式逆变器的稳态控制方法，适用于微电网双模式逆变器并联***，所述微电网双模式逆变器并联***包括多个双模式逆变器、并/离网开关、三相电网和逆变控制电路；所述双模式逆变器包括直流储能电容、三相逆变电路、逆变控制电路、LC滤波电路，所述直流储能电容、三相逆变电路、LC滤波电路依次连接，所述LC滤波电路与线路阻抗连接，所述线路阻抗通过交流母线与并/离网开关连接，所述并/离网开关接入三相电网；所述逆变控制电路包括采样调理电路、锁相环电路、控制器、驱动保护电路；所述采样调理电路输入端与所述LC滤波电路连接；所述控制器与所述驱动保护电路输入端、采样调理电路输出端、锁相环电路输出端连接；所述锁相环电路输入端与所述交流母线连接；其特征在于，该方法包括离网模式稳态控制方法和并网模式稳态控制方法；
    所述离网模式稳态控制方法为：
    1)在每个采样周期的起始点，采样调理电路对三相电网电压usa、直流储能电容电压udc、LC滤波电路电容电压uoa、uob、uoc、线路电流ioa、iob、ioc、入网电流isa、isb、isc分别进行采样，然后将采样数据送给控制器进行处理，计算各采样值的有效值，并将LC滤波电路电容电压uoa、uob、uoc和线路电流ioa、iob、ioc分别转换为在αβ坐标下的LC滤波电路电容电压uoα、uoβ和线路电流ioα、ioβ；
    2)将LC滤波电路电容电压uoa、uob、uoc与线路电流ioa、iob、ioc分别相乘，得到双模式逆变器的有功功率P和无功功率Q；
    3)对双模式逆变器输出电压幅值参考值U*、角频率参考值ω*、有功功率参考值P*、无功功率参考值Q*，以及上述有功功率平P、无功功率Q进行下垂控制运算，得到双模式逆变器的输出电压幅值Uo和角频率ωo；其中，并网模式下，ω*、U*为电网电压角频率和幅值的实时值，根据电网电压的变化而变化；离网模式下，ω*、U*为设定值；
    4)由双模式逆变器输出电压有效值Uo、角频率ωo，以及电压输出相位
    Figure PCTCN2014086501-appb-100001
    合成αβ 坐标下的参考电压urefα、urefβ，其中t为采样时间：
    Figure PCTCN2014086501-appb-100002
    5)将urefα、urefβ分别减去uoα与反馈系数kU的乘积、uoβ与反馈系数kU的乘积，得到的差值作为准谐振QPR控制器的输入；其中反馈系数kU的取值范围为0.1～2；
    6)引入参考电压前馈环节kF·urefα *、kF·urefβ *，将kF·urefα *、kF·urefβ *与准谐振QPR控制器的输出相加，得到线路电流的参考值irefα、irefβ；其中kF为电压前馈系数，取值范围为0.01～5；
    7)对线路电流参考值irefα、irefβ，LC滤波电路电容电压uoα、uoβ，直流储能电容电压udc进行电流无差拍控制，得到三相逆变电路开关管控制量dα、dβ：
    Figure PCTCN2014086501-appb-100003
    其中，K为控制量常数，取值范围0<K<1，Lf为LC滤波电路电感值，Tc为PWM载波周期；udc(k)为k时刻采样的双模式逆变器直流侧电压，uoα(k)、uoβ(k)为k时刻的αβ坐标下的LC滤波电路电容电压，ioα(k)、ioβ(k)为k时刻αβ坐标下线路电流，
    Figure PCTCN2014086501-appb-100004
    为k+1时刻αβ坐标下的线路电流的预测参考值；
    8)对dα、dβ进行坐标变化，得到稳态控制时abc坐标下的三相逆变电路开关管控制量da、db、dc；
    9)将da、db、dc与三角载波进行双极性调制，得到三相逆变电路开关管的占空比信号，经驱动保护电路，控制三相逆变电路开关管的开通与关断；
    所述并网模式稳态控制方法为：
    将上述离网模式稳态控制方法步骤4)中的电压输出相位调整为
    Figure PCTCN2014086501-appb-100005
    然后根据 上述步骤1)～步骤9)的方法控制三相逆变电路开关管的开通与关断；其中，
    Figure PCTCN2014086501-appb-100006
    表示锁相环电路检测到的三相电网电压相位，δ＝δ0±Δδ，δ0取值范围为0.0001～0.15；Δδ为相位补偿参数，
    Figure PCTCN2014086501-appb-100007
    其中，Io1、Is1分别为功率变化后双模式逆变器的输出电流有效值和入网电流有效值，Io2、Is2为功率变化前双模式逆变器的输出电流有效值和入网电流有效值，R'、X'分别为双模式逆变器到交流母线的线路阻抗和感抗，R"、X"分别为本地负载到交流母线的线路阻抗和感抗。
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