CN102045035B - Low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier - Google Patents

Low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier Download PDF

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CN102045035B
CN102045035B CN 201010557194 CN201010557194A CN102045035B CN 102045035 B CN102045035 B CN 102045035B CN 201010557194 CN201010557194 CN 201010557194 CN 201010557194 A CN201010557194 A CN 201010557194A CN 102045035 B CN102045035 B CN 102045035B
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semiconductor
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type metal
current mirror
current
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CN102045035A (en
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吴金
马科
汤欣伟
郑雷
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Southeast University
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Abstract

The invention discloses a low-power consumption broadband high-gain high-slew rate single-stage operational transconductance amplifier, which is formed by successively connecting a constant current bias stage, a differential input stage and a load current mirror transmission output stage in series. The load current mirror transmission output stage comprises eight N type MOS (metal oxide semiconductor) transistors from NM1 to NM8. Through the invention, the inherent limit restraint in a linear operational amplifying circuit is thoroughly solved, and the circuit performance under static AC small signals and dynamic big signals is comprehensively improved and enhanced.

Description

A kind of low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier
Technical field
The present invention relates to a kind of compatible low-power consumption, at a high speed, the linear transconductance single-stage discharge circuit of high accuracy characteristic, belong to the Analog Integrated Operation amplifier technical field, by to load current mirror nonlinear effect and linear transfer than the compatibility of control and integrated, the coordinating and unifying of circuit speed and precision under acquisition efficient current utilization ratio and the power constraints, the General Promotion of realization circuit static, dynamic characteristic.
Background technology
The SoC digital-to-analogue is mixed in the system-on-a-chip, and expanding of system function and performance raising only just have more real meaning under the constraint of low-power consumption.In the prior art, the improvement of circuit speed or dynamic property is that power consumption increases to cost with large driven current density usually, and the requirement that the low-power consumption high-speed, high precision restricts has mutually significantly increased the difficulty of linear discharge circuit Design and implementation.Therefore, low-power consumption high speed discharge circuit designs and realizes, needs to break through the limitation that has linear structure now.
For traditional single-stage OTA linear operation transconductance circuit (seeing Fig. 1), under the low supply voltage restriction, be difficult to take full advantage of the high resistant output characteristic realization high-gain of Cascode structure, the method that improves impedance and gain by reducing output current is difficult to again satisfy the high-speed response requirement under dynamic current adjusting and the driving; On the contrary, after the N value is fixing, by increasing quiescent current I BImprove bandwidth, obtain the high-speed response under the large Slew Rate driving, not only cause the increase of power consumption, cause simultaneously the decline that gains.Therefore, for traditional OTA structure, can only be at quiescent current I B, input difference chooses being optimized between the pipe mutual conductance factor, three kinds of parameters of differential pair load current mirror W/L scale factor N value, with the demand of balance gain, bandwidth, Slew Rate and power consumption, but it is then very difficult to satisfy simultaneously each side's demand.For solving the intrinsic contradictions in the circuit, need to break through by nonlinear effect and the configurable control of parameter restriction and the limitation of original linear circuit structure.
Summary of the invention
The object of the invention is to provide a kind of low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier in order to have solved the intrinsic contradictions between Circuits System speed, precision and the power consumption that is difficult to be in harmonious proportion in the conventional discharge circuit.
A kind of low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier of the present invention, be connected in series successively differential input stage by the constant current biasing, load current mirror transmission output stage three parts consist of, wherein load current mirror transmission output stage is made of eight N-type metal-oxide-semiconductor NM1 to NM8, the drain electrode of N-type metal-oxide-semiconductor NM1 meets respectively output and the N-type metal-oxide-semiconductor NM3 of differential input stage, NM6, the grid of NM7, the drain electrode of N-type metal-oxide-semiconductor NM1 meets respectively another output and the N-type metal-oxide-semiconductor NM4 of differential input stage, NM5, the grid of NM8, the source electrode of N-type metal-oxide-semiconductor NM1 meets respectively N-type metal-oxide-semiconductor NM3, the drain electrode of NM5, the source electrode of N-type metal-oxide-semiconductor NM2 meets respectively N-type metal-oxide-semiconductor NM4, the drain electrode of NM6, N-type metal-oxide-semiconductor NM3, NM4, NM5, NM6, NM7, the source electrode of NM8 connects respectively ground connection, N-type metal-oxide-semiconductor NM7 drain electrode consists of the first output of load current mirror transmission output stage, and N-type metal-oxide-semiconductor NM8 drain electrode consists of the second output of load current mirror transmission output stage.
Described constant current biasing is made of P type metal-oxide-semiconductor PM0, and the drain electrode of P type metal-oxide-semiconductor PM0 meets power supply Vdd, and the source electrode of P type metal-oxide-semiconductor PM0 connects differential input stage.
Described differential input stage is made of two P type metal-oxide-semiconductor PM1, PM2, and the drain electrode of P type metal-oxide-semiconductor PM1, PM2 connects the output of constant current biasing, and the source electrode of P type metal-oxide-semiconductor PM1, PM2 connects the input of load current mirror transmission output stage.
The present invention by playing positive feedback effect under the small-signal the cross-coupled pair tubular construction and wide dynamic range in effectively compatibility and the interaction of load current mirror linearity-nonlinear model adaptive configuration structure, efficiently solve the intrinsic contradictions between Circuits System speed, precision and the power consumption that is difficult to be in harmonious proportion in the conventional discharge circuit.The key of pattern control is under static conditions the load current mirror to be biased under linearity-non-linear critical mode of operation, guarantees low static power consumption; Under ac small signal, utilize the interior cross-coupled pair pipe of the range of linearity to the decline control action of load current mirror linear transfer ratio simultaneously, realize high-gain (A V) double to guarantee high accuracy control, bandwidth (p -3dBAnd GBW) multiplication is to guarantee small-signal at a high speed and the linear process performance; Under large-signal, although cross-coupled pair tube failure, but large input dynamic range can change the differential load current mirror over to degree of depth nonlinear operation pattern automatically, namely realizes the multiplication of slew rate (SR) by the nonlinear effect of current delivery, to improve transient response speed.By configuration and the nonlinear effect of linear transfer breadth length ratio, can thoroughly solve the constraint of the inherent inherent limitation of linear discharge circuit, circuit performance improves comprehensively under realization static state, ac small signal and the dynamic large-signal.
The present invention has under equal low static power consumption, the multiplication of the gain of circuit ac small signal, bandwidth and large-signal voltage Slew Rate.Adopting the CSMC0.18mm standard CMOS process, is under the condition of 29mA and 30pF load capacitance at quiescent current, and low-frequency gain is 71.3dB, and unity gain bandwidth is 6.5MHz, and the forward slew rate under the large-signal is+12.5V/ms that oppositely slew rate is-12.8V/ms.Compare the classical single-stage OTA structure under the equal conditions, increased 24dB, bandwidth has improved 9 times, and the voltage Slew Rate increases 20 times.
Description of drawings
Fig. 1 is the basic circuit diagram of conventional operation trsanscondutance amplifier OTA.
Fig. 2 is the circuit diagram of ordinary lines sex ratio current mirror and Cascode linear current mirror.
Fig. 3 is the circuit diagram of linear/non-linear self-adaptive current mirror.
Fig. 4 is the configurable OTA circuit theory diagrams of nonlinear model.
Fig. 5 is the small-signal Bode diagram of OTA shown in Figure 4.
Fig. 6 is the large-signal voltage Slew Rate of OTA shown in Figure 4.
Embodiment
Be elaborated below in conjunction with the technical scheme of accompanying drawing to invention:
The present invention adopts following technical scheme:
A kind of operation transconductance amplifying circuit (seeing Fig. 4) that under the low-power consumption constraint, still has the high-precision high-speed response, its feature comprises that operation transconductance amplifier belongs to OTA one pass gain circuit structure; Circuits System is made of constant current biasing, differential input stage, load current mirror transmission output stage three parts, wherein the PMOS differential pair tube adopts fixedly tail current biasing, the load current mirror of two pairs of symmetries adopts respectively the dynamic configurable structure of the linearity-nonlinear model that is made of three NMOS pipes, increases simultaneously the pair of cross coupling to the structure of pipe to load current mirror equivalence input W/L modulation control; The circuit output stage adopts the symmetrical Drive Structure of CMOS complementary push-pull.
Two kinds of configurable controls based on circuit structure and mode of operation: utilize its good complementarity that under AD HOC, has, cooperatively interact to solve the inherent intrinsic contradictions restriction of conventional linear circuit, the static state of General Promotion circuit, interchange and mapping; A kind of is the positive feedback control that utilizes under the cross-coupled pair pipe interchange condition, realizes switching or the preparation of the linear carry-over factor of current mirror under the unlike signal pattern, the main improvement that realizes circuit ac small signal characteristic; Another kind is based on the dynamic-configuration of current mirror linearity-nonlinear model, the main lifting that realizes circuit transient state large signal characteristic under the condition that keeps low-power consumption;
The structure of differential input stage and the setting of mode of operation: differential input stage adopts classical three pipe differential configurations, can be three pipe PMOS differential pairs, also can be the nmos differential pair of with it mirror; Differential-pair tail current adopts stable constant biasing, differential pair two input pipes adopt the identical symmetric design of W/L, two differential pair tubes respectively flow through the tail current of half when guaranteeing static state, difference is exported A, B two terminal potentials are identical, variable-current size under the difference AC small-signal drives in two differential pair tubes is identical, polarity is opposite, and simultaneously difference output A, B two ends keep the character of differential signal;
The setting of the configurable structure of current mirror linearity-nonlinear model and mode of operation: the M4 pipe on the output branch road in the four wide amplitude of oscillation Cascode current-mirror structure of pipe (seeing Fig. 2) is removed (or short circuit), can obtain the configurable current-mirror structure of this pattern (seeing Fig. 3).Concrete pattern control is by current mirror input current I 1, input each metal-oxide-semiconductor size W/L of branch road and Cascode bias voltage V BnThe relativeness of three kinds of influencing factors determines.
At bias voltage V BnAnd electric current I 1All under the fixing condition, reduce to have the M1 of same size and the W/L of M3 two pipes, current mirror will be transformed to non-linear nature by linear behavio(u)r, on the contrary, increase W/L current mirror will be transformed to linear behavio(u)r by non-linear nature; W/L design by suitable can be set in this current mirror required linearity or nonlinear operation pattern as required.Differential pair is inputted driving current constant and V to differential load under static conditions BnConstant, utilizing this point characteristic is the pattern that the adjusting of W/L is arranged on load current mirror under the static conditions.
At bias voltage V BnWith input pipe device size W/L all under the fixing condition, increase input current I 1Current mirror will be transformed to non-linear nature by linear behavio(u)r, on the contrary, reduce input current I 1Current mirror will be transformed to linear behavio(u)r by non-linear nature; Therefore, by the variation of input current, can make current mirror realize conversion between the different transport properties.Differential pair changes to differential load input drive current under the little condition of ac small signal, but V BnConstant, W/L no longer changes after setting through static schema, and utilizing this point characteristic is the dynamic self-adapting adjusting of variation realization load current mirror pattern under the ac small signal condition of input current;
The configurable current mirror of differential pair load model, under low static power consumption requires, quiescent biasing is in linearity-non-linear critical working point or slightly be partial to nonlinear mode of operation, the a pair of current mirror of differential load will be respectively to linear and non-linear two mode deflections under the difference condition of small signal, after entering the large-signal input state, one of them load current mirror can be transferred to the nonlinear operation pattern usually;
The cross-coupled pair tubular construction is controlled the adjusting of differential pair load current mirror input pipe equivalence W/L: equivalent W/L is increased, and the current delivery linear coefficient reduces; Equivalent W/L reduces under the ac small signal pattern, and the current delivery linear coefficient increases; Equivalent W/L is constant under the transient state large-signal, and the current delivery linear coefficient is identical with the structure that does not have the cross-coupled pair pipe.
The setting of circuit working pattern: under the low static power consumption constraint, the load current mirror should be biased in linearity-non-linear critical working point pattern, or slightly is partial to non-linear current mirror pattern.This circuit is fit to higher supply voltage and less amplifier tail current, and the dynamic change scope of generation enough makes load current mirror penetration depth nonlinear model under the large-signal, and realizes the Slew Rate maximum lift under high common mode electrical level input.
Fig. 2 and Fig. 3 have provided contrasts linear, linearity-nonlinear adaptive two class current-mirror structure, M 1With M 2The ratio of the W/L of pipe is m:1.For the linear current mirror, it is I that current delivery is closed 2=I 1No matter/m is i.e. I 1How to change, as long as M 2It is constant that pipe is kept saturated constant current district operating state, I 2With I 1Remain that above linear relationship is constant.If under different signal conditions, can realize control or adjusting to the m value, be to make m increase to reduce current delivery under the static direct current, make m reduce to increase current delivery under the ac small signal, be that linear transfer is than the dynamic-configuration of m value under different conditions, just can solve the defective of the linear current mirror of traditional fixed transmission ratio, satisfy the joint demand of circuit DC and AC characteristics.
For the configurable current mirror of adaptive model, M under the static conditions 1Pipe is at specific input current I 1And V BnUnder the voltage bias, be limited to the critical working point in saturated constant current district and linear resistance district, this moment, current delivery kept linear behavio(u)r; Under the dynamic condition along with I 1Increase, M 3Pipe V GS3Increase, because of V thereupon DS1=V Bn-V GS3, then at fixing V BnThe lower V of biasing DS1Descend, force M 1Pipe enters the linear resistance district, the M under the electric current of increase and the linear work district 1Pipe will make its grid voltage significantly improve, if M 2Still keep initial saturated constant current zone state, output current I 2Obviously increase, i.e. original linear current transmission changes non-linear current multiplication transmission into.
Replace linear current mirror that pattern and m value all fix as the differential load of OTA circuit the configurable current mirror of above m value and pattern, utilize input differential signal control inputs drive current I 1, make it that online property-non-linear critical conduction mode is set under static state or small-signal condition, keep low-power consumption and high gain characteristics by linearity or approximately linear current delivery; After entering the moving state of dynamic large-signal, under the input current that increases drives, automatically change degree of depth non-linear current transmission mode over to significantly to improve output current, realize significantly improving of Slew Rate and transient response speed.On this basis, further adopt the cross-coupled pair pipe that effective W/L of load current mirror is regulated, obtain configurable linear transfer Coefficient m under the unlike signal pattern, realize improving of AC characteristic.The coordination of non-linear and the configurable two class controlling mechanisms of m value with cooperate, finally realize comprehensively improving of circuit synthesis performance.
The circuit theory diagrams of the present invention that Fig. 4 provides comprise the impact that two kinds of special constructions interact and effect is controlled current delivery, and the improvement and the raising that bring thus the discharge circuit performance, below divide seven aspects to carry out labor and elaboration.
1, the positive feedback cross coupling structure is to the control of linear current transmission coefficient
The cross coupling structure that NM5, NM6 two pipes consist of only plays the positive feedback regulating action in the linear scope of small-signal, change the effective linear ratio N of current delivery EffNM5 and NM7 and NM6 are K:1 with the ratio of the breadth length ratio of NM8.If I BBe the fixedly tail current biasing of differential pair tube, static lower input difference is I to two branch currents B/ 2, and a, b two node voltages equate, the quiescent current in each load metal-oxide-semiconductor is directly proportional with its W/L.If N T=N+K, N S=N-K, comprehensive each side performance requirement is generally selected N in the Classic couture T=1, static lower N Eff=N T, the output branch current is I o=I B/ (2N T), the total quiescent current of circuit reaches I Q=(1+N T -1) I B2I BThe reduction of quiescent dissipation also is conducive to the raising of output impedance and the increase of low-frequency gain simultaneously.
The input of ac small signal difference drives lower, differential load a, 2 current potentials of b become differential signal, NM5 and NM6 cross-coupled pair pipe satisfy the complementary drive characteristic of ac small signal electric current under this differential signal drives, and make the equivalent linear transfer ratio of load current mirror be reduced to N Eff=N S, transmission current increases.Because the linear behavio(u)r of circuit structure can be kept this linear behavio(u)r simultaneously under the small-signal operation condition, then the direct current transfer ratio is identical with the alternating current transfer ratio, as shown in the formula:
Figure 699744DEST_PATH_IMAGE001
(1)
N under the dc state Eff(DC)=N T, N under the exchange status Eff(AC)=N S, N Eff(DC)〉〉 N Eff(AC).
2, the state factor of the configurable linearity of pattern-nonlinear adaptive current mirror
NM1, NM3, NM7 and NM2, NM4, NM8 consist of respectively the linearity of two groups of symmetries-nonlinear adaptive current mirror, and the linear current transfer ratio that is wherein determined by breadth length ratio is N:1, and two pipes adopt the identical symmetric design of W/L in the input branch road.Difference according to NM3 or NM4 Cascode input pipe operating state can be biased in current mirror under the different mode states.
The configurable current mirror that forms take NM1, NM3, NM7 three pipes is analyzed condition judgement and the Working mode set of current mirror as example.If the input and output electric current of this current mirror is respectively I InWith I o, when three pipes when all being operated in saturated constant-current source service area, being the linear current mirror, its alternating current-direct current signal transport property is identical and satisfy the linear transfer rule; When the NM3 pipe enters the linear resistance district and all the other two pipes still keep saturated constant-current characteristics, then be the non-linear current mirror, its alternating current-direct current signal transmission characteristics separates and satisfies respectively different nonlinear transport and concerns.The operating state of NM3 pipe can be by its Gate voltage and Cascode bias voltage V BnBetween relativeness determine, establish D M3=V GS3-V TN, D Mb=V Bn-V TN, definition a V=D M3/ D Mb, the resistance of NM3 pipe or constant current state and the current mirror transport property that determines thus all can be by a VSize distinguished, therefore be called the state factor of circuit or the nonlinear factor of current mirror.After NM3 enters the linear resistance district, under the prerequisite of ignoring NM1 pipe body bias effect, have:
Figure 650383DEST_PATH_IMAGE002
(2)
If normalized V DS3Be h=V DS3/ D Mb, at k 1=k 3Symmetric design under, obtain h and at a VApproximate under 1 the dark linear resistance district,, namely have:
Figure 408254DEST_PATH_IMAGE003
(3)
Clearly, a VBe approximately anti-phase relation, a with h V=h=1/2 is corresponding to the critical state point in NM3 pipeline resistance area and saturated constant current district, and current mirror still keeps linear transport property under this critical conduction mode; Work as a V1/2 or h<1/2 after, the NM3 pipe enters the nonlinear resistance district, current mirror changes nonlinear transport into by linear transfer character, a VLarger or h is less, nonlinear effect is stronger.Indicator a as state model or nonlinear degree VAnd h, with input current I InThe pass be:
Figure 636104DEST_PATH_IMAGE004
(4)
Work as a V1 o'clock, approximate have a h/ a V0.The linear current model that (2) formula of utilization provides obtains:
Figure 578653DEST_PATH_IMAGE005
(5)
With co-relation explanation, state factor a VTo change with the input current of differential load current mirror a VThe variation that value occurs in 1/2 front and back is corresponding to the variation of load current mirror transport property, and a VNumerically variation after 1/2, the change of nonlinear degree power embodied.Therefore, only have the variation characteristic that utilizes input current, ability is the mode of operation of regulating load current mirror effectively, thereby consists of the basic principle of the type current mirror working mode adaptive dynamic-configuration.
3, the dynamic range of state factor and control thereof
Ignore body bias effect, then fixing D MbD is satisfied in biasing Mb=D M1+ V DS3=D M1+ hD MbConstraint obtains D Mb=D M1/ (1-h) concerning, this formula is all applicable to different input currents, wherein with the initial static electric current I In0Corresponding parameter is D M1,0And h 0, and h 0Then corresponding to a of initial setting up V0If the electric current I of dynamic change In=dI In0, D is then arranged M1=D M1,0d 1/2, according to D under the curent change MbConstant condition obtains d 1/2=(1-h)/(1-h 0), the h=that provides in conjunction with (3) formula again f(a V) functional relation, solve:
Figure 891953DEST_PATH_IMAGE007
(6)
Nonlinear factor a VWith the nonlinear factor a under input current dynamic change scope d, the initial static V0The h of control 0Factor-related.If initial static is selected linearity-non-linear critical conduction mode point a V0=h 0=1/2, under this constraint, solve a V=(1+d/2-d 1/2)/(2-d 1/2).When this current mirror is used for fixedly the tail current differential configuration because of d=2, a then V=1, namely under the dynamic change of 2 times of differential load input currents, the current mirror nonlinear factor is by a under the initial criticality pattern V0=1/2, increase to a under the nonlinear model V=1, the nonlinear effect insufficient strength.
When current mirror is used for fixedly the tail current differential load, because the dynamic range that input current is 2 times is not enough to produce enough large nonlinear factor, therefore adopt dynamic tail current to control to increase the d factor and become the highly effective method of enhancing nonlinear strength, under the above ideal conditions when electric current dynamic change during near 4 times, a V, can realize the non-linear of any intensity.In fact, V GS3, maxLimited by specific circuit architecture and supply voltage, then actual a V, maxThere is upper range.And under fixing tail current condition, be a under improving dynamically VThe factor also can be by reducing h 0And realize.According to h and a VAnti-phase relation, directly increase a V0, be about to the static below-center offset of current mirror in the small nonlinearity district, can increase by the appropriateness of power consumption, i.e. a V0Small size increase exchange a for VSignificantly raising.
4, the configurable linearity of pattern-nonlinear adaptive current mirror transmission characteristic
According to the saturation current model of NM3 and NM7 pipe, output I o=(1/2) k 7D M7 2, D M7=D M3, k 1=k 3, behind NM3 pipe penetration depth linear resistance district, its V DS3Reduce to make this pipe D that overdrives M3Increase, the raising of final decision output current, that is:
(7)
Above relationship description direct current large-signal transmission characteristic, 1/N is the relevant linear carry-over factor of W/L, b DCEmbody the contribution of nonlinear effect, namely nonlinear effect is to the multiplication factor of linear carry-over factor, and has:
Figure 187937DEST_PATH_IMAGE008
(8)
At a VUnder the critical conduction mode of=h=1/2, b DC=1, along with a VThe factor increases, multiplication factor b DCObviously increase.Another aspect that the current mirror nonlinear effect embodies is that the direct current large-signal is separated with the transmission coefficient of ac small signal, and both no longer be not same amount under the linear conditions.But there is the relevance that can not isolate between DC and AC characteristics, utilizes the local derviation of direct current transmission equation is processed, and (5) formula of utilization relation, the interchange transmission equation that obtains is:
(9)
Exchange transmission characteristic and have identical form with the direct current transmission characteristic, wherein the linear transfer coefficient still is 1/N, and the interchange transmission coefficient is b ACEqually, at a VUnder 1 the strong nonlinearity condition, exchange multiplication factor and be approximately:
(10)
After the strong nonlinearity effect occurred, exchanging transmission coefficient obviously increased, and the speed of increase surpasses direct current, and the strong nonlinearity times increment of alternating current-direct current will be above the linear multiplier coefficient simultaneously.
5, amplifier critical performance parameters Changing Pattern
Two kinds of physical effects cause the variation of ac and dc current transmission characteristic, are the roots that causes circuit DC and AC characteristics parameter generating to change, and according to Fig. 4 structure, static lower a, 2 current potentials of b are identical, and the current mirror linear transfer is than being 1/N T, then quiescent dissipation is:
Figure 574553DEST_PATH_IMAGE011
(11)
I in the formula BBe differential-pair tail current, I oBe the output branch current.Under near the ac small signal condition the static point, a, 2 of b are converted into differential signal, because the compensating action of cross-coupled pair pipe, current mirror linear transfer ratio switches to 1/N S, then the amplifier small-signal transconductance is:
Figure 674227DEST_PATH_IMAGE012
(12)
G in the formula M1,2Be differential pair mutual conductance, G M_ref=g M1,2/ N TFor Fig. 1 conventional structure with reference to the mutual conductance of amplifier under identical quiescent dissipation, the linear scale factor a that the raising that then improves the circuit mutual conductance forms from the dimensional configurations effect AC=N T/ N SWith the non-linear b that exchanges the transmission effects generation ACThe acting in conjunction of the factor, namely the product of two factors determines the total multiplication factor of mutual conductance.The unity gain bandwidth of single-stage systems stabilisation is GBW=G m/ C L, so bandwidth GBW and G mHas identical multiplication factor.Under the joint effect that current delivery and mutual conductance change, being changed to of low-frequency gain:
Figure 735724DEST_PATH_IMAGE013
(13)
L is the metal-oxide-semiconductor channel modulation factor in the formula, I InBe the input current of differential load current mirror, b AC, b DCBe respectively the AC and DC multiplication factor that the current mirror nonlinear transport causes near the small signal region of static point.When quiescent point is arranged under the linear model, then the load current mirror is still relatively near linear behavio(u)r under the ac small signal, and not only value is near 1 and b arranged ACb DC, cause the multiplication factor of circuit low frequency DC current gain mainly by a ACThe factor determines.
After input differential signal exceeds dynamic range, be zero the inefficacy because of the electric current in the cross-coupled pair pipe on the one hand, a differential load current mirror is always arranged on the other hand under large-signal drives and inject whole tail currents, usually can make this current mirror enter the strong nonlinearity mode of operation, consequent output slew rate is:
Figure 895441DEST_PATH_IMAGE014
(14)
SR in the formula Ref=I O_ref/ C LBe the output slew rate of reference OTA circuit, it is zero to act on inefficacy because of electric current that current mirror switches to after the large-signal pattern cross-coupled pair pipe by the direct current static point, and the linear multiplier factor that the conversion of its effective W/L under two kinds of patterns produces is a DY=N T/ N, b DYThen enter the direct current transmission multiplication factor b of non-linear current mirror behind the dark linear resistance district for NM3 pipe under the large-signal DC
Clearly, the variation of all kinds of AC-DC parameter indexs of amplifier is all from the contribution of two class multiplication factors, a kind of is the dimension line sex factor that load current mirror W/L comes at different mode incision tape swapping, this factor value under different signal modes changes is different, but irrelevant with concrete size of current, between AD HOC, switch to specific linear constant; Another kind is that current mirror enters the nonlinear factor that brings after the nonlinear transport pattern, and this factor changes with the actual current size, shows with circuit state
Continually varying character.
6, the optimum mode of operation setting of amplifier
Select different bias voltage V Bn(as shown in Figure 4), the OTA discharge circuit can be biased in three kinds of different quiescent operation dot patterns in theory.Wherein, if quiescent point is selected typical linear current delivery pattern, i.e. a V<<1/2, then above all kinds of multiplication factors are 1, and circuit deteriorates to traditional structure, and therefore performance does not select the realistic meaning of this pattern without any improvement; On the contrary, if quiescent point is selected typical linear current delivery pattern, i.e. a V1/2, b DCIncrease quiescent dissipation will obviously be increased, this quasi-mode can't become optimum working mode because not meeting the low power dissipation design requirement.Therefore, for satisfying discharge circuit low-power consumption requirement, obtain the power consumption identical with reference configuration, can only choose a V=1/2, b DC=1 quiescent biasing condition, this is chosen in the critical conduction mode of linearity-nonlinear transport corresponding to load current mirror quiescent point, and this pattern or working point pattern in its vicinity namely correspond to the optimal mode of system works.
Under this pattern, because b is arranged DC b AC1, to compare quiescent dissipation with the reference discharge circuit and remain unchanged, the multiplication factor of mutual conductance, bandwidth and gain is a AC, Slew Rate multiplication factor a little less than 2b DYAbove result shows, under the low-power consumption mode of optimum, and dimension line sex factor a ACThe main multiplication that realizes the small-signal AC characteristic, large-signal direct current degree of depth nonlinear factor b DYThe main multiplication that realizes wide amplitude of oscillation dynamic characteristic has good complementary characteristic between two kinds of effects, can improve the performance of circuit various aspects under its acting in conjunction.
The setting of above critical working point pattern is by static tail current I B, differential pair load current mirror input pipe W/L, Cascode pipe bias voltage V BnThe three determines jointly.For overcoming the impact of NM1 pipe body bias effect, V BnShould be slightly larger than the theoretical prediction value of not considering to serve as a contrast when inclined to one side, the part of increase should compensate cut-in voltage because serving as a contrast the variable quantity DV that partially brings TNZV DS3Under degree of depth nonlinear transport pattern, because of V DS3Very little, the impact of body bias effect is very weak, and body bias effect is to a VAnd the impact of each relevant multiplication factor can be ignored.
Owing to adopting fixedly tail current biasing, dynamically descend 2 times of curent changes, so under above static critical point pattern, a of state factor during by static state V0=1/2 increases to a under the dynamic large-signal V=1, the non-linear multiplication factor of direct current transmission is accordingly by b DC=1 increases to b DY=2, actual output Slew Rate multiplication amplitude is still less than normal, away from the upper range of circuit permission., on the quantitative setting of mode of operation, can slightly adjust for this reason, static lower current mirror is biased in weak non-linear, sacrifice quiescent dissipation as cost take appropriateness, exchange a under the large-signal for VAnd b DYRaising, improve large-signal transient state drive characteristic.On concrete methods of realizing, can suitably reduce on the original basis D MbBias voltage, or at D MbSuitably increase the W/L of NM3, NM4 two pipes under the constant condition of bias voltage.
7, the upper range of multiplication factor
Under optimum working mode, the upper range that need to improve as far as possible all kinds of multiplication factors for obtaining optimum circuit alternating current-direct current and large signal characteristic, but because being subjected to the restriction of actual circuit structure and condition of work, all kinds of multiplication factors all have the specific upper limit, alleviate these restrictions the maximum capacity of bringing into play circuit is played an important role.
Ac small signal parameter multiplication factor a ACThe upper limit by the amplifier bandwidth less than inferior limit p 2Term restriction, output high resistant large electric capacity has determined that the output limit is system's dominant pole, the 1/g of inside circuit mType low-resistance load node construction system time limit.In each site position of Fig. 4, d, e two nodes cause pole frequency higher because of the electric capacity less, and differential pair load current mirror a, b two node places be because of parasitic gate capacitance maximum and the minimum minimum inferior limit of frequency that becomes of equivalent transconductance, and make two pole frequencies equal because of symmetrical configuration.
With a point small-signal equivalent capacity C aBe example, this electric capacity should comprise the drain capacitance of the gate capacitance of NM3, NM7, NM6 three pipes and PM1, NM1 two pipes, if the W/L of above each metal-oxide-semiconductor is larger, considers that from reducing the mismatch aspect L value is larger simultaneously, then the C that forms of large device area aElectric capacity is also larger, pole frequency p A, bLess.Under the constraint of 60 ° of phase margins, circuit bandwidth GBW is less, a of permission ACMultiplication factor upper limit peak value is also less.Under optimal mode, the interchange output impedance of a, b two nodes is approximately r A, b1/g M_Ns, g wherein M_NsBe the stream ac small signal mutual conductance of differential pair load Electronic Speculum under the quiescent point, the parasitic time limit that forms thus is p A, b=g M_Ns/ C a, bandwidth GBW=G m/ C L
Consider the low frequency limit of two coincidences, system stability should satisfy 4GBW<p A, bRestriction.Consider from the differential pair parameter, differential pair tube W/L can make the input mutual conductance and with reference to the Bandwidth Reduction of amplifier, the interchange multiplication factor that then allows increases by reducing; Consider from the differential load current mirror, by increasing N TValue can make the current delivery coefficient and reduce with reference to the bandwidth of amplifier, and the interchange multiplication factor of permission is increased.Regulate according to above bandwidth constraint, in conjunction with differential pair tube and its load pipe gain factor k PM12=k NTSymmetrical or approximately uniform condition, can get:
(15)
At deep submicron process condition, parasitic capacitance C aCan be controlled in the scope of hundreds of pico farad, then output capacitance and C aThe ratio of electric capacity will reach 2 more than the order of magnitude, and the interchange multiplication factor upper limit that allows thus reaches a ACMore than 10.
The various restraining factors that the output Slew Rate is subject to, one of them is the restriction of input Slew Rate, the speed that is increased to circuit when exceeding the inside circuit Slew Rate when the output Slew Rate will determine by the response that discharges and recharges of inner node capacitance, and the speed that discharges and recharges that increases output capacitance is meaningless.Therefore the maximum Slew Rate that discharges and recharges of inner node, the higher limit corresponding to the maximum Slew Rate of output obtains b under the large-signal thus DYBe limited on the factor:
(16)
In addition, differential pair common source end c point common mode electrical level produces material impact to a, the peaked restriction of b two point voltages equally to output current and output Slew Rate.C point current potential is with difference input common mode electrical level V ComAnd change, the highest being increased to makes tail current enter the critical point of linear resistance-saturated constant current, i.e. V C, max=V DD-D PM0, D wherein PM0Be the overdrive voltage of tail current pipe, actual potential is V c=V TP+ D PM1, max, D wherein PM1, maxBe the maximum overdrive voltage of differential pair tube under the dynamic large-signal, the i.e. dynamic range of corresponding difference input.Because c point current potential will be at V cV C, maxChange in the scope, and V A, bV c, then the maximum overdrive voltage of efferent duct is V c-V TN; And the overdrive voltage of efferent duct is a when static VD Mb, and under optimal mode a V1/2.Consider square being directly proportional of output driving current and output saturation pipe overdrive voltage, and the variation of output current switches to by critical linearity from the current mirror mode of operation non-linearly, then be subjected to V cThe b of level restriction DYScope is:
(17)
Found out by following formula, increase supply voltage and adopt low static power consumption to setover to reduce D MbOverdrive voltage can improve the peak level of Slew Rate multiplication factor.At V ComV DD-(V TP+ D PM1, max+ D PM0) in the scope, increase V ComCan improve the maximum output direct current multiplication factor of permission, and the actual multiplication factor that produces, with a in the differential pair load current mirror input current dynamic range VFactor-related, the input current dynamic range is wider, a VChange greatlyr, nonlinear effect is stronger, then the actual non-linear multiplication factor b that produces DYLarger.
Under optimal mode, answer the static tail current I of reasonable disposition BWith the biasing D that overdrives MBRelativeness, by the W/L of each metal-oxide-semiconductor in offset and the differential load current mirror rationally is set, can make the Slew Rate multiplication factor of actual output can reach corresponding theoretical upper limit under its specific common mode incoming level.
8, side circuit design and performance improvement thereof
For Fig. 4 circuit structure, typical work condition is V DD=3.3V, C L=30pF, I B=10mA provides V BnBiasing branch road 10mA, static below-center offset is at a V=1/2 critical working point pattern, the overdrive voltage of the approximate 0.2V of each metal-oxide-semiconductor.
N is chosen in the actual parameter design T=1, N S=0.1, a AC=N T/ N S=10, a DY=N T/ N=1/0.55=1.82.Adopt CSMC 0.18mm standard CMOS process, V TN=0.5V, V TP=0.6V, a, b node parasitic capacitance are approximately C a<0.2pF, C L/ C a100.
Fig. 5 is circuit AC characteristic simulation result, and under 60 ° of phase margin constraints, actual bandwidth is compared with reference to amplifier and improved 9 times, and theoretical prediction is 10 times; Actual low-frequency gain improves 22dB, and theoretical prediction is 20dB.Therefore, for AC characteristic, theoretical model calculates with the side circuit simulation result very identical.
Fig. 6 is for being made of the large-signal dynamic response characteristic of voltage follower discharge circuit, delayed data can be obtained the output Slew Rate that is obtained by emulation testing from figure, equally reference configuration is similarly tested, and both relatively find, the Slew Rate multiplication reaches 20 times, and calculating nonlinear factor by theory is a V=3.2, b DC=10, a DY=1.82, the Slew Rate multiplication reaches 18.6 times, and same and simulation result coincide.
The input signal common mode electrical level is 1.5V in the test, the b of this level restriction DC40, the actual parameter multiplication that produces is all away from its theoretical upper limit, especially under high power supply and high input common mode electrical level.At last, table 1 has provided traditional OTA (Fig. 1) and OTA of the present invention (Fig. 4) performance simulation result's contrast.
Table 1 performance comparison (C L=30pF)
Parameter Tradition OTA structure OTA structure of the present invention The new and old structure contrast
Operating voltage (V) 3.3 3.3 Operating voltage is identical
Quiescent current (mA) 30 29 Quiescent current is identical
Low-frequency gain (dB) 49.4 71.3 Gain improves 22 dB
Unity gain bandwidth (Hz) 714k 6.5M Bandwidth increases by 9 times
Phase margin (deg) 89.4 60.6 Reduce but meet the demands
Positive Slew Rate (V/ms) +0.62 +12.5 Improve 20 times
Negative Slew Rate (V/ms) -0.65 -12.8 Improve 20 times
PSRR(dB) 28.6@10kHz 72@10kHz Increase more than 40 dB
9, sum up
Utilize state and the mode of operation of the size control dynamic self-adapting current mirror of input current, farthest the performance of compatible static state, ac small signal and dynamic large-signal.When static less input current with configurable configuration biases in linear current mirror pattern in the hope of low-power consumption, exchange to change under the large-signal and bring configurable current mirror strong nonlinearity effect because the input current dynamic range increases down.
When taking full advantage of, circuit of the present invention becomes the effect that non-linear current mirror combining structure produces, namely under different working modes, utilize the cross-coupled pair pipe to the time-varying characteristics of W/L Configuration, in conjunction with the nonlinear model control to linearity-nonlinear adaptive current mirror, realized the multiplication of difference current delivery under the low static power consumption condition, overcome the inherence limitation that linear monotype discharge circuit is difficult to overcome, General Promotion the combination property of circuit, novel circuit feasibility and superiority that theoretical model, simulation result and related experiment result all verify.

Claims (3)

1. low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier, it is characterized in that being connected in series successively differential input stage by the constant current biasing, load current mirror transmission output stage three parts consist of, wherein load current mirror transmission output stage is made of eight N-type metal-oxide-semiconductor NM1 to NM8, the drain electrode of N-type metal-oxide-semiconductor NM1 meets respectively output and the N-type metal-oxide-semiconductor NM3 of differential input stage, NM6, the grid of NM7, the drain electrode of N-type metal-oxide-semiconductor NM2 meets respectively another output and the N-type metal-oxide-semiconductor NM4 of differential input stage, NM5, the grid of NM8, the source electrode of N-type metal-oxide-semiconductor NM1 meets respectively N-type metal-oxide-semiconductor NM3, the drain electrode of NM5, the source electrode of N-type metal-oxide-semiconductor NM2 meets respectively N-type metal-oxide-semiconductor NM4, the drain electrode of NM6, N-type metal-oxide-semiconductor NM3, NM4, NM5, NM6, NM7, the source electrode of NM8 connects respectively ground connection, N-type metal-oxide-semiconductor NM7 drain electrode consists of the first output of load current mirror transmission output stage, N-type metal-oxide-semiconductor NM8 drain electrode consists of the second output of load current mirror transmission output stage
Wherein, the N-type metal-oxide-semiconductor NM1, NM3, the NM7 that are operated under linearity-nonlinear model consist of one linearity-nonlinear adaptive current mirror, and the N-type metal-oxide-semiconductor NM2, NM4, the NM8 that are operated under linearity-nonlinear model consist of another linearity-nonlinear adaptive current mirror.
2. a kind of low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier according to claim 1, it is characterized in that described constant current biasing is made of P type metal-oxide-semiconductor PM0, the drain electrode of P type metal-oxide-semiconductor PM0 meets power supply Vdd, and the source electrode of P type metal-oxide-semiconductor PM0 connects differential input stage.
3. a kind of low-power consumption broadband high-gain high-swing rate single-level operation transconductance amplifier according to claim 1, it is characterized in that described differential input stage is made of two P type metal-oxide-semiconductor PM1, PM2, the drain electrode of P type metal-oxide-semiconductor PM1, PM2 connects the output of constant current biasing, and the source electrode of P type metal-oxide-semiconductor PM1, PM2 connects the input of load current mirror transmission output stage.
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