WO2014128995A1 - パッシブレーダ装置 - Google Patents
パッシブレーダ装置 Download PDFInfo
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- WO2014128995A1 WO2014128995A1 PCT/JP2013/072423 JP2013072423W WO2014128995A1 WO 2014128995 A1 WO2014128995 A1 WO 2014128995A1 JP 2013072423 W JP2013072423 W JP 2013072423W WO 2014128995 A1 WO2014128995 A1 WO 2014128995A1
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/06—Systems determining position data of a target
- G01S13/46—Indirect determination of position data
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/003—Bistatic radar systems; Multistatic radar systems
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
- G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
- G01S7/28—Details of pulse systems
- G01S7/285—Receivers
- G01S7/288—Coherent receivers
- G01S7/2883—Coherent receivers using FFT processing
Definitions
- the present invention is a passive radar device that realizes detection and tracking of a target such as an aircraft or a ship by using an existing radio wave source as a radiation source without emitting radio waves by itself, and by detecting a plurality of frequency bands in a coherent manner.
- the present invention relates to a passive radar device that improves tracking performance.
- a passive radar device does not emit radio waves, but uses existing radio wave sources used for communications and broadcasting as a transmission source, and receives and processes radio waves radiated from them and waves reflected by the target. As a result, target tracking of an aircraft, a ship, or the like is realized.
- a radio wave radiated from the transmission source directly reaches the receiving means is called a direct wave, and a radio wave reflected by the target is called a reflected wave. Then, the cross-correlation between the direct wave and the reflected wave is calculated, and the target is detected and tracked by detecting the peak.
- An antenna that receives a direct wave, a receiving unit, and the like are referred to as a reference system, and an antenna that receives a reflected wave, the receiving unit, and the like are referred to as a search system.
- a signal having a plurality of frequency bands is received by a broadband receiver, the Doppler frequency of the received signal of the reference system is compensated, the correlation processing with the received signal of the search system is performed, and the delay / Doppler frequency Is calculated. Then, the target is detected by detecting the peak.
- Patent Document 1 has the following two problems.
- the first problem is that a broadband receiver is required, and the cost is higher than that of a narrowband receiver.
- the amount of calculation of cross-correlation increases.
- the second problem is that the Doppler frequency is different for each frequency band. Therefore, when the Doppler frequency of a signal in multiple frequency bands is compensated simultaneously by one Doppler frequency, accurate compensation cannot be performed and the loss in the cross-correlation processing is lost. Is generated.
- the target bistatic Doppler velocity V (the sum of the target velocity target-transmitter direction component and the target-receiver direction component) and the transmission carrier frequency have the relationship of the following equation (1).
- fd (n) is the Doppler frequency in the nth frequency band
- fc (n) is the carrier frequency of the signal in the nth frequency band
- c is the speed of light.
- a long-time cross-correlation process is generally used to extend the detection distance.
- the frequency resolution ⁇ fd after the cross-correlation process can be expressed by the following equation (2).
- the present invention has been made to solve the above-described problems, and an object of the present invention is to provide a passive radar device that can suppress integration loss due to different Doppler frequencies among a plurality of frequency bands.
- a passive radar device includes a search system receiving antenna that receives a reflected wave transmitted from a radio wave source and reflected by a target, a reference system receiving antenna that receives a direct wave transmitted from the radio wave source, and a search system Search system band dividing means for dividing the received signal of the reflected wave received by the receiving antenna for each frequency band, and reference system band dividing means for dividing the received signal of the direct wave received by the reference system receiving antenna for each frequency band And a search system receiving means for receiving a reflected wave reception signal for each frequency band divided by the search system band dividing means and performing A / D conversion, and a direct for each frequency band divided by the reference system band dividing means.
- Reference system receiving means for receiving a received signal of a wave and performing A / D conversion, a received signal of a reflected wave A / D converted by a search system receiving means, and A / D converted by a reference system receiving means
- Cross-correlation processing means for performing cross-correlation for each frequency band with the received signal of the direct wave
- band synthesis means for performing band synthesis on the cross-correlation result for each frequency band by the cross-correlation processing means, and band synthesis result by the band synthesis means
- target detection means for detecting a target based on the above.
- the passive radar device includes a search system reception antenna that receives a reflected wave transmitted from a radio wave source and reflected by a target, a reference system reception antenna that receives a direct wave transmitted from the radio wave source, Search system broadband receiving means for receiving the reflected wave received by the search system reception antenna in a wide band, performing A / D conversion, and dividing each frequency band; and direct wave received by the reference system reception antenna
- a reference system wideband receiving means for receiving a received signal in a wideband, performing A / D conversion, and dividing it for each frequency band, a reflected wave reception signal divided by the search system wideband receiving means, and a reference system wideband receiving means
- Cross-correlation processing means for performing cross-correlation for each frequency band with the divided direct wave reception signal, and band for combining the cross-correlation results for each frequency band by the cross-correlation processing means.
- forming means in which a target detection means for detecting a target based on the band synthesis result by band synthesizing means
- the present invention since it is configured as described above, it is possible to suppress the integral loss due to the difference in Doppler frequency between a plurality of frequency bands.
- FIG. 1 is a diagram showing a configuration of a passive radar device according to Embodiment 1 of the present invention.
- the passive radar device includes receiving antennas 101 and 102, band dividing means 103 and 104, receiving means 105 and 106, cross-correlation processing means 107, band synthesizing means 108, and peak detecting means 109. Yes.
- the receiving antenna 101, the band dividing means 103, the receiving means 105, the cross correlation processing means 107, the band synthesizing means 108 and the peak detecting means 109 constitute a search system
- the receiving means 106 constitutes a reference system.
- the receiving antenna (search system receiving antenna) 101 receives a radio wave (reflected wave) transmitted from a transmission source 201 which is an existing radio wave source and reflected by a target 202 to be detected such as an aircraft or a ship.
- the reception antenna (reference system reception antenna) 102 receives radio waves (direct waves) transmitted from the transmission source 201.
- the search system that detects the target 202 since direct waves interfere with each other, it is desirable to consider that direct waves are not mixed as much as possible.
- the direct wave received by the reference system is cross-correlated with the received signal of the search system in the cross-correlation processing means 107, it is desirable that the multi-scattered wave generated in a multipath environment or the like is not included. Therefore, the receiving antenna 102 is installed in a place where the transmitting source 201 can be seen as much as possible.
- the band dividing means (search system band dividing means) 103 divides the reception signal (RF signal) of the reflected wave received by the receiving antenna 101 for each frequency band.
- the band dividing means (reference system band dividing means) 104 divides the direct wave reception signal (RF signal) received by the receiving antenna 102 for each frequency band.
- the band dividing means 103 and 104 generally, a component called a demultiplexer, a separator, an antenna duplexer or the like is used, and as shown in FIG. Divide by analog stage. Since the duplexer separates signals by frequency, the loss of SNR (Signal to Noise Ratio) is small. However, since the SNR loss at the time of division may increase depending on the frequency bandwidth, a configuration may be adopted in which division is performed for each of a plurality of frequency bandwidths.
- SNR Signal to Noise Ratio
- the band dividing means 103, 104 is provided with a local signal transmission source, and by changing the frequency from the local signal transmission source in time series and the received signal, the band to be extracted is changed, The received signal may be divided for each frequency band.
- the receiving means (search system receiving means) 105 receives the received signal of the reflected wave for each frequency band divided by the band dividing means 103 and performs A / D conversion.
- the receiving means (reference system receiving means) 106 receives the direct wave reception signal for each frequency band divided by the band dividing means 104 and performs A / D conversion.
- the receiving means 105 and 106 do not need to be wideband receivers, but may be narrowband receivers (low speed A / D converters).
- the cross-correlation processing unit 107 performs cross-correlation for each frequency band between the reflected wave reception signal A / D converted by the reception unit 105 and the direct wave reception signal A / D converted by the reception unit 106. It is. The configuration of the cross correlation processing means 107 will be described later.
- the band synthesizing unit 108 synthesizes the cross correlation result for each frequency band by the cross correlation processing unit 107.
- the peak detection means (target detection means) 109 detects the target 202 based on the band synthesis result by the band synthesis means 108.
- the cross-correlation processing unit 107 includes a Doppler frequency shift unit 1071, FFT units 1072 and 1073, a complex conjugate multiplication unit 1074, and an IFFT unit 1075 for each frequency band.
- the Doppler frequency shift unit 1071 shifts the direct wave reception signal for each frequency band from the receiving unit 106 by the Doppler frequency calculated from the assumed target bistatic Doppler speed and the carrier frequency for each frequency band. It is.
- the FFT unit 1072 performs a fast Fourier transform (FFT) on the received signal of the reflected wave for each frequency band from the receiving unit 105.
- the FFT unit 1073 performs fast Fourier transform (FFT) on the direct wave reception signal for each frequency band frequency-shifted by the Doppler frequency shift unit 1071.
- the complex conjugate multiplier 1074 takes the complex conjugate of the received signal of the direct wave for each frequency band fast Fourier transformed by the FFT unit 1073 and the received signal of the reflected wave for each frequency band fast Fourier transformed by the FFT unit 1073 Multiply.
- the IFFT unit 1075 performs an inverse fast Fourier transform (IFFT: Inverse FFT) on the output signal that is a result of the complex conjugate multiplication by the complex conjugate multiplication unit 1074.
- IFFT inverse fast Fourier transform
- the receiving antenna 101 receives a reflected wave transmitted from the transmission source 201 and reflected by the target 202 (step ST1). Moreover, the receiving antenna 102 receives the direct wave transmitted from the transmission source 201 (step ST2).
- the band dividing means 103 divides the received signal of the reflected wave received by the receiving antenna 101 for each frequency band (step ST3). Further, the band dividing means 104 divides the direct wave reception signal received by the receiving antenna 102 for each frequency band (step ST4).
- the receiving means 105 receives the received signal of the reflected wave for each frequency band divided by the band dividing means 103 and performs A / D conversion (step ST5).
- the receiving means 106 receives the direct wave reception signal for each frequency band divided by the band dividing means 104 and performs A / D conversion (step ST6).
- receiving means 105 and 106 perform amplification of the received signal by LNA (Low Noise Amplifier), mixing with a local signal by a mixer, and the like. Then, the signal is converted into a digital signal by performing A / D conversion after being converted into an IF (intermediate frequency) band signal.
- the mixer is an orthogonal mixer
- the signal after A / D conversion has an I signal and a Q signal, and can be handled as a complex signal.
- the signal after A / D conversion can be converted into a baseband signal by multiplying exp ( ⁇ j ⁇ 2 ⁇ ⁇ fIF ⁇ t). Note that fIF is the IF center frequency.
- the mixer is not a quadrature mixer, the signal after A / D conversion has only an I signal, so it is converted to a complex signal by Hilbert transform or the like, and exp ( ⁇ j ⁇ 2 ⁇ ⁇ fIF ⁇ t) is By multiplication, it is converted into a baseband signal.
- the cross-correlation processing unit 107 calculates the cross-correlation for each frequency band between the reception signal of the reflected wave A / D converted by the reception unit 105 and the reception signal of the direct wave A / D converted by the reception unit 106. Perform (step ST7).
- the received signals (baseband signals) of the search system and the reference system in a certain frequency band #n are s sur (n, t) and s ref (n, t), respectively.
- the baseband signal of the search system is frequency-shifted by the bistatic Doppler frequency fd (n) generated with the movement of the target expressed by the equation (1). ing.
- This fd (n) is usually unknown. Therefore, the Doppler frequency shift unit 1071 prepares a bistatic Doppler velocity V (l) corresponding to the assumed bistatic Doppler frequency range, and shifts the frequency of the reference baseband signal.
- the shifted signal s ref (n, l, t) is given by the following equation (3).
- V (1) is the initial value of the bistatic Doppler velocity to be observed
- ⁇ V is the bistatic Doppler interval, and is desirably 1 / T or less in order to reduce the loss in the cross-correlation processing.
- the FFT unit 1072, 1073, the complex conjugate multiplication unit 1074, and the IFFT unit 1075 use the following equation (5) as a cyclic correlation calculation using a fast Fourier transform (FFT) and an inverse fast Fourier transform (IFFT).
- FFT fast Fourier transform
- IFFT inverse fast Fourier transform
- the band synthesizing unit 108 synthesizes the cross-correlation result CCF (n, l, m) for each frequency band by the cross-correlation processing unit 107 between the bands as shown in the following equation (6), thereby improving the SNR. (Step ST8).
- the peak detection unit 109 calculates the square of the absolute value of CCCF (k, l, m), which is the band synthesis result by the band synthesis unit 108, and detects the target 202 by detecting the peak (step) ST9).
- a peak detection method a method of detecting as a target when the square of the absolute value exceeds a predetermined value set in advance, or a value obtained by multiplying the average value of the cell peripheral range of interest of the square of the absolute value by a coefficient
- CA-CFAR Cell Averaging-Constant False Alarm Rate
- the received signal is divided into signals for each frequency band and A / D converted, and then configured to perform band synthesis by cross-correlation for each frequency band. Since a wideband receiver is not required and cross-correlation with low loss can be realized with a low-speed A / D converter, integration loss due to a difference in Doppler frequency among a plurality of frequency bands can be suppressed.
- Embodiment 2 shows a configuration that solves the above problem by dividing the received signal into blocks, performing cross-correlation in each block, and performing fast Fourier transform between the blocks.
- FIG. 5 is a diagram showing the configuration of the cross-correlation processing means 107 in the second embodiment of the present invention.
- the block division unit 1076 divides the reception signal of the reflected wave for each frequency band from the reception unit 105 for each block.
- the block division unit 1077 divides the direct wave reception signal for each frequency band from the reception unit 106 for each block.
- the FFT unit 1072b performs a fast Fourier transform (FFT) on the received signal of the reflected wave for each frequency band and each block divided by the block dividing unit 1076.
- the FFT unit 1073b performs fast Fourier transform (FFT) on the frequency band divided by the block division unit 1077 and the direct wave reception signal for each block.
- the complex conjugate multiplier 1074 takes the complex conjugate of the received signal of the direct wave for each frequency band and block that has been fast Fourier transformed by the FFT unit 1073b, and for each frequency band and each block that has been fast Fourier transformed by the FFT unit 1072b. Multiply by the received signal of the reflected wave.
- the inter-block FFT unit 1078 performs a fast Fourier transform (FFT) in the block direction on the inverse fast Fourier transform result from the IFFT unit 1075.
- FFT fast Fourier transform
- the correlation calculation using the fast Fourier transform results in a cyclic cross-correlation, but the transmission signal in the passive radar device is generally not a repetitive signal. Therefore, in order to perform the sliding correlation in the correlation calculation using the fast Fourier transform, the received signals of the search system and the reference system are divided as shown in FIG. That is, as shown in FIG. 6A, the block dividing unit 1076 divides the received signal for each time interval Tb, and then adds the received signal of the next block to the received signal of each divided block. A block having a time interval of 2 Tb is generated. On the other hand, as shown in FIG.
- the block dividing unit 1077 divides the received signal every time interval Tb, and then adds a 0 signal corresponding to the time interval Tb to the received signal of each divided block. Thus, a block having a time interval of 2 Tb is generated.
- the target delay time ⁇ is sufficiently smaller than the time interval Tb in order to reduce the amount of correlation calculation, the correlation calculation is performed on the received signal divided for each time interval Tb in both the search system and the reference system. It is also possible to adopt a configuration to perform.
- the fast Fourier transform between the blocks may be configured to perform a Fourier transform on the target bistatic Doppler velocity V (l), as in the equation (3) in the first embodiment.
- the band synthesizing unit 108 synthesizes CCF (n, q, m) in the direction of the frequency band n in the same manner as Expression (6) in the first embodiment as a result of the cross correlation processing unit 107.
- the cross-correlation processing unit 107 divides both the search system and reference system received signals into blocks and performs cross-correlation for each block, and then performs high speed between blocks. Since the configuration is such that the Fourier transform is performed, in addition to the effect in the first embodiment, an increase in the amount of calculation can be avoided even if the cross-correlation time T becomes long.
- the bistatic Doppler velocity V matches or A configuration for synthesizing close cross-correlation functions is described.
- the ratio between the frequency bands of the Doppler frequency is determined by only the ratio of the carrier frequency without depending on the bistatic Doppler velocity V. Therefore, as shown in CCF (n, q * fc (n) / f1 (n), m), a cross-correlation function in which the ratio of the carrier frequency for each frequency band and the ratio of the Doppler frequency are identical or close is synthesized. Also good.
- FIG. 7 is a diagram showing a configuration of a passive radar device according to Embodiment 3 of the present invention.
- the passive radar device according to the third embodiment shown in FIG. 7 deletes the band dividing means 103 and 104 and the receiving means 105 and 106 from the passive radar device according to the first embodiment shown in FIG. 111 is added.
- Other configurations are the same, and the same reference numerals are given and description thereof is omitted.
- the broadband receiving means (search system broadband receiving means) 110 receives a reflected wave reception signal (RF signal) received by the receiving antenna 101 in a wide band, performs A / D conversion, and divides the signal into frequency bands. is there.
- the wideband receiving means (reference system wideband receiving means) 111 receives a direct wave reception signal (RF signal) received by the receiving antenna 102 in a wide band, performs A / D conversion, and divides it into frequency bands. is there.
- signals in a plurality of frequency bands are collectively A / D converted by the broadband receiving means 110 and 111 and then divided into signals for each band by digital processing.
- a broadband receiver and a broadband A / D converter are required as the broadband receiving means 110 and 111.
- the wideband receiving means 110 and 111 are used in place of the band dividing means 103 and 104 and the receiving means 105 and 106, it is possible to use a plurality of frequency bands.
- the integral loss due to the different Doppler frequencies can be suppressed.
- Embodiment 4 FIG.
- the amplitude / phase fluctuations of the transmission signals in a plurality of frequency bands and the amplitude / phase fluctuations due to the frequency characteristics of the receiver are estimated in real time with a simple configuration, so that the signals in the plurality of frequency bands are synthesized without loss.
- the method to do is shown.
- s n (t) is a modulation sequence (broadcast / communication data, etc.) of frequency channel #n
- fc a modulation sequence (broadcast / communication data, etc.) of frequency channel #n
- n is a center frequency (carrier frequency) of frequency channel #n
- ⁇ n is a signal of frequency channel #n. Is the initial phase.
- a receiving antenna / receiver that receives a direct wave of a satellite is called a reference system
- a receiving antenna / receiver that receives a target scattered wave is called a search system. It is assumed that the reference system and search system are synchronized.
- ⁇ n is a complex amplitude at the time of direct wave reception of frequency channel #n
- ⁇ D is a propagation delay time from the satellite to the reference system.
- the received signal of Expression (9) is converted to a baseband signal by multiplying it by a local signal exp (j2 ⁇ fc , nt).
- phi ⁇ n is the phase of a frequency channel #n including the initial phase of the local signal.
- the received signal of Expression (11) is converted to a baseband signal by multiplying the local signal exp (j2 ⁇ fc , nt) as in the reference system.
- T is an integration time
- Pn is the average power of frequency band #n.
- ⁇ t a sampling period and ⁇ f is a Doppler frequency resolution
- the cross-correlation values can be combined coherently by IFFT in the frequency channel direction as follows.
- K having the peak in the absolute value of the equation (16) is as follows. Further, the frequency characteristic of the target of the RCS, if only phase amplitude is constant varies linearly, i.e., the target also by the frequency channel direction of the IFFT when written as exp (j2 ⁇ f c, n ⁇ RCS ) is integral You can see that In this case, however, it should be noted that the peak delay time index is shifted by ⁇ RCS .
- FIG. 8 shows the configuration of the passive radar device according to the fourth embodiment.
- the passive radar device according to Embodiment 4 shown in FIG. 8 is obtained by adding interband compensation means 112 to the passive radar device according to Embodiment 1 shown in FIG.
- Other configurations are the same, and the same reference numerals are given and description thereof is omitted.
- the inter-band compensation unit 112 compensates the amplitude and phase fluctuation for each frequency band with respect to the cross-correlation result for each frequency band by the cross-correlation processing unit 107.
- Band synthesizing means 108 synthesizes the cross-correlation result for each frequency band after compensation by interband compensating means 112.
- the direct wave from the transmission source 201 generally has high power, it is mixed from the side lobe or back lobe of the receiving antenna 101 which is a search system.
- compensation of amplitude / phase characteristics between bands is performed using a direct wave mixed in the receiving antenna 101.
- the search antenna 101 that is the search system and the receive antenna 102 that is the reference system are placed in substantially the same place (within the reciprocal of the signal band ⁇ the speed of light), and the transmission source 201 is stationary when viewed from the receiver.
- the power of the direct wave mixed from the side lobe or back lobe of the receiving antenna 101 is small, it appears as a peak if the integration time T of the cross-correlation function is long.
- the target reflected received power is several tens of times smaller. Therefore, when performing a cross-correlation function with an integration time such that a target reflected signal is detected, a direct wave leakage peak occurs and a high SNR is often obtained.
- the inter-band compensation unit 112 calculates the correction coefficient based on the band # 1 as follows, for example. To do.
- the inter-band compensation unit 112 compensates the cross-correlation function as follows using the correction coefficient calculated as described above, so that the signal power fluctuation between the bands of the transmission signal and the frequency characteristics between the receivers. Can be compensated. Thereafter, the inter-band compensation unit 112 sends the cross-correlation function that has been inter-band compensated as shown in Expression (20) to the band synthesizing unit 108. Thereby, it can synthesize
- the interband compensation unit 112 can obtain the correction coefficient as follows.
- the interband compensation unit 112 compensates for amplitude and phase fluctuations for each frequency band using the amplitude ratio and phase difference of the direct wave leakage component in the cross correlation result for each frequency band by the cross correlation processing unit 107.
- the correction coefficient to be calculated is calculated, and the cross-correlation result is compensated by the correction coefficient.
- a correction coefficient is calculated by temporarily directing the reception antenna 101 toward the transmission source 201, and the reception antenna 101 is set to the target again.
- the inter-band compensation unit 112 uses the amplitude ratio and the phase difference between the reflected wave reception signal and the direct wave reception signal for each frequency band acquired with the receiving antenna 101 directed in the direct wave direction as a correction coefficient, and the correction coefficient.
- the cross-correlation result for each frequency band may be compensated.
- the correction coefficient may be obtained by simply comparing the amplitude and phase between the band of the search system reception signal and the reference system reception signal.
- a correction coefficient between bands is calculated in a series of conventional target detection processes without adding a new device as a compensation means between bands. Therefore, it is possible to compensate for the amplitude / phase fluctuation between the bands in real time while suppressing the expansion of the apparatus scale, and to improve the target SNR in the band synthesizing unit 108.
- Embodiment 5 FIG.
- band synthesizing means 108 in the first embodiment shown in FIG. 1 obtains a high-resolution cross-correlation function by rearranging data in the delay time direction.
- the passive radar device according to the fifth embodiment is the same as the configuration of the passive radar device according to the first embodiment shown in FIG. 1, and only the different parts will be described with the same reference numerals.
- the cross-correlation result for each frequency band by the cross-correlation processing unit 107 is compensated for the phase fluctuation for each frequency band depending on the delay time index of the cross-correlation result,
- the cross-correlation results after inter-band synthesis performed by inverse fast Fourier transform for each frequency band are rearranged in the delay time direction.
- the delay time ambiguity is eliminated by performing band synthesis after compensating for the phase corresponding to the delay using the delay time index of the cross-correlation processing result, so that the accurate delay time is obtained.
- the purpose is to obtain.
- phase compensation is performed using the delay time index l T after cross-correlation processing as follows.
- the intercorrelation function after the phase compensation is subjected to IFFT in the band direction as follows, thereby performing synthesis between bands.
- the delay time range to be estimated in the band synthesizing unit 108 is limited to ⁇ t.
- Equation (26) a cross-correlation function with improved delay time resolution can be obtained. here, It is.
- the band synthesizing unit 108 is configured to perform the inter-band synthesizing process after compensating for the phase depending on the delay time of the cross-correlation function.
- the delay time ambiguity after synthesis is eliminated, and a high-resolution cross-correlation function can be obtained.
- FIG. Embodiments 1 to 5 show the case where the search system is one system, but Embodiment 6 shows a configuration including a plurality of systems.
- FIG. 9 is a diagram showing the configuration of a passive radar device according to Embodiment 6 of the present invention.
- the passive radar apparatus according to the sixth embodiment shown in FIG. 9 includes the receiving antenna 101, the band dividing means 103, the receiving means 105, the cross-correlation processing means 107, and the band synthesis of the passive radar apparatus according to the first embodiment shown in FIG.
- a plurality of means 108 and peak detection means 109 are provided, and a beam forming means 113 is added.
- Other configurations are the same, and the same reference numerals are given and description thereof is omitted.
- the beam forming unit 113 performs beam forming by combining the band combining results obtained by the plurality of band combining units 108 between the plurality of receiving antennas 101.
- the plurality of target detection means 109 detect a target based on the beam forming result by the beam forming means 113.
- the beam forming unit 113 synthesizes the cross-correlation function after band synthesis between the receiving antennas 101 of the search system as follows. That is, the beam forming unit 113 performs beam forming on the band synthesis result by the band synthesizing unit 108 so as to compensate the phase difference depending on the carrier frequency, the beam directing direction, and the positional relationship of the receiving antenna 101.
- the receiving antenna 101 forms an equally spaced array antenna (antenna spacing d, number of antennas p)
- the beam forming process of the cross-correlation function ⁇ com, p [l T , k b, n , m] is as follows: You can write Here, ⁇ q is the beam pointing direction.
- an SNR improvement gain corresponding to the number of antennas is expected. Further, the absolute value (amplitude value) or power value of ⁇ [l T , kb , n , m] may be calculated and then combined (added or averaged) by the number of antennas.
- the beam forming unit 113 may be configured to perform beam forming immediately after the processing by the cross correlation processing unit 107.
- a plurality of search systems are provided, and beam forming is performed by combining the processing results obtained by the cross-correlation processing unit 107 or the band combining unit 108 between the plurality of receiving antennas 101. Since the beam forming means 113 is provided, it is possible to expect SNR improvement gains corresponding to the number of antennas.
- Embodiment 7 shows a configuration in which long-time integration is efficiently performed by integrating the cross-correlation function after band synthesis in the time direction.
- 10 is a diagram showing the configuration of a passive radar device according to Embodiment 7 of the present invention.
- the passive radar device according to the seventh embodiment shown in FIG. 10 is obtained by adding long-time integrating means 114 to the passive radar device according to the first embodiment shown in FIG.
- Other configurations are the same, and the same reference numerals are given and description thereof is omitted.
- the long time integration unit 114 integrates the band synthesis result by the band synthesis unit 108 for a long time.
- the target detection unit 109 detects the target based on the integration result by the long time integration unit 114.
- Non-Patent Document 1 As a long-time integration method by the long-time integration means 114, for example, there is a method disclosed in Non-Patent Document 1.
- a radar reception signal for each pulse hit is divided into several CPIs (Coherent Pulse Intervals). Then, a range Doppler map integrated by FFT between pulse hits in the divided CPI is generated. Then, when integrating it between CPIs, a plurality of hypotheses corresponding to the movement of the target are established and integrated in the time direction. Thereby, the integration loss due to the target range-Doppler frequency shift can be reduced.
- CPIs Coherent Pulse Intervals
- the range Doppler map for each pulse hit can be replaced with the cross-correlation function for each block in the second embodiment. That is, some of the blocks shown in FIG. 5 are combined into one CPI, and a plurality of CPI data is created. In 1 CPI, as shown in the second embodiment, it is considered to perform integration between blocks by FFT to calculate a cross-correlation function, and to make a plurality of hypotheses when integrating them between CPIs.
- the number of delay time (range) indexes after cross-correlation within 1 CPI is L t
- the number of Doppler frequency indexes is K D
- the number of hypotheses accompanying the movement of the target is N HY
- the number of CPI is N CPI
- the computation amount of the band synthesis process may be an N-point IFFT of L t K D N CPI times, and the number of IFFT computations decreases by the number of hypotheses N HY.
- the long-time integration unit 114 that performs long-time integration between the CPIs of the cross-correlation function after the band synthesis processing is provided, the band synthesis processing is performed after the long-time integration. Compared with the configuration, the number of IFFT operations can be reduced by the number of hypotheses.
- the passive radar device can suppress integration loss due to different Doppler frequencies among a plurality of frequency bands, and does not radiate radio waves by itself using an existing radio wave source as a radiation source, such as an aircraft or a ship.
- a passive radar device that realizes target detection and tracking is suitable for use in a passive radar device that improves detection and tracking performance by coherently combining a plurality of frequency bands.
- 101 receiving antenna (search system receiving antenna), 102 receiving antenna (reference system receiving antenna), 103 band dividing means (search system band dividing means), 104 band dividing means (reference system band dividing means), 105 receiving means (search system) Receiving means), 106 receiving means (reference system receiving means), 107 cross-correlation processing means, 108 band synthesizing means, 109 peak detecting means (target detecting means), 110 wide band receiving means (search system wide band receiving means), 111 wide band receiving Means (reference system broadband receiving means), 112 Interband compensation means, 113 Beam forming means, 114 Long-time integration means, 201 Transmission source (radio wave source), 202 Target, 1071, Doppler frequency shift unit, 1072, 1072b, 1073, 1073b FFT unit, 1074 complex conjugate multiplication unit, 10 5 IFFT unit, 1076 block division unit (search-based block division unit), 1077 block dividing unit (see system block division unit), between 1078 block FFT unit.
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Abstract
Description
1つ目の課題は、広帯域受信機が必要であり、狭帯域受信機と比較して高コストとなる点である。また、A/D変換後のサンプルデータ数も増加するため、相互相関の演算量が増加する。
2つ目の課題は、周波数帯域毎にドップラー周波数が異なるので、一つのドップラー周波数により同時に複数周波数帯域の信号のドップラー周波数を補償した場合に、正確な補償ができずに、相互相関処理における損失が発生するという点である。
目標のバイスタティックドップラー速度V(目標速度の目標-送信機方向成分と目標-受信機方向成分との和)と送信搬送波周波数には下式(1)の関係がある。
ここで、fd(n)は、n番目周波数帯域におけるドップラー周波数、fc(n)は、n番目周波数帯域の信号の搬送波周波数、cは光速である。
実施の形態1.
図1はこの発明の実施の形態1に係るパッシブレーダ装置の構成を示す図である。
パッシブレーダ装置は、図1に示すように、受信アンテナ101,102、帯域分割手段103,104、受信手段105,106、相互相関処理手段107、帯域合成手段108及びピーク検出手段109から構成されている。このパッシブレーダ装置のうち、受信アンテナ101、帯域分割手段103、受信手段105、相互相関処理手段107、帯域合成手段108及びピーク検出手段109は捜索系を構成し、受信アンテナ102、帯域分割手段104及び受信手段106は参照系を構成する。
受信アンテナ(参照系受信アンテナ)102は、送信源201から送信された電波(直接波)を受信するものである。
帯域分割手段(参照系帯域分割手段)104は、受信アンテナ102により受信された直接波の受信信号(RF信号)を周波数帯域毎に分割するものである。
受信手段(参照系受信手段)106は、帯域分割手段104により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行うものである。
なお、受信手段105,106は、広帯域受信機である必要はなく、狭帯域受信機(低速なA/D変換機)でよい。
ピーク検出手段(目標探知手段)109は、帯域合成手段108による帯域合成結果に基づいて目標202を探知するものである。
相互相関処理手段107は、図3に示すように、周波数帯域毎に、ドップラー周波数シフト部1071、FFT部1072,1073、複素共役乗算部1074及びIFFT部1075から構成されている。
FFT部1073は、ドップラー周波数シフト部1071により周波数シフトされた周波数帯域毎の直接波の受信信号を高速フーリエ変換(FFT)するものである。
パッシブレーダ装置の動作では、図4に示すように、まず、受信アンテナ101は、送信源201から送信され目標202により反射された反射波を受信する(ステップST1)。また、受信アンテナ102は、送信源201から送信された直接波を受信する(ステップST2)。
なお、上記ミキサが直交ミキサの場合には、A/D変換後の信号はI信号・Q信号を持つので、複素信号として扱うことができる。この場合、A/D変換後の信号に対して、exp(-j・2π・fIF・t)を乗算することにより、ベースバンド信号に変換できる。なお、fIFはIF中心周波数である。また、上記ミキサが直交ミキサでない場合には、A/D変換後の信号はI信号のみしか持たないので、ヒルベルト変換等により複素信号に変換し、exp(-j・2π・fIF・t)を乗算することで、ベースバンド信号に変換する。
ここで、m(m=1,・・・,M)は、相互相関処理後の遅延ビンである。また、遅延時間τは、サンプリング周期Δtを用いて、τ=(m-1)・Δtとして計算できる。
実施の形態1では、相互相関時間Tの信号を高速フーリエ変換し、相互相関を行う場合について示したが、この場合には相互相関時間Tが長くなるにつれて演算量が増加する。そこで、実施の形態2では、受信信号をブロック毎に分割し、各ブロックで相互相関を行った後にブロック間で高速フーリエ変換することで、上記課題を解消する構成について示す。
図5はこの発明の実施の形態2における相互相関処理手段107の構成を示す図である。図5に示す実施の形態2における相互相関処理手段107は、図2に示す実施の形態1における相互相関処理手段107からドップラー周波数シフト部1071を削除し、ブロック分割部1076,1077及びブロック間FFT部1078を追加し、FFT部1072,1073をFFT部1072b,1073bに変更したものである。その他の構成は同様であり、同一の符号を付してその説明を省略する。
ブロック分割部1077は、受信手段106からの周波数帯域毎の直接波の受信信号を、ブロック毎に分割するものである。
FFT部1073bは、ブロック分割部1077により分割された周波数帯域及びブロック毎の直接波の受信信号を高速フーリエ変換(FFT)するものである。
すなわち、図6(a)に示すように、ブロック分割部1076では、時間間隔Tb毎に受信信号を分割した後に、当該分割した各ブロックの受信信号に次のブロックの受信信号を追加することで、時間間隔2Tbのブロックを生成する。一方、図6(b)に示すように、ブロック分割部1077では、時間間隔Tb毎に受信信号を分割した後に、当該分割した各ブロックの受信信号に時間間隔Tb分の0信号を追加することで、時間間隔2Tbのブロックを生成する。ただし、相関演算の演算量を削減するために、目標遅延時間τが時間間隔Tbより十分に小さい場合には、捜索系、参照系ともに時間間隔Tb毎に分割した受信信号に対して相関演算を行う構成とすることも可能である。
図7はこの発明の実施の形態3に係るパッシブレーダ装置の構成を示す図である。図7に示す実施の形態3に係るパッシブレーダ装置は、図1に示す実施の形態1に係るパッシブレーダ装置から帯域分割手段103,104及び受信手段105,106を削除し、広帯域受信手段110,111を追加したものである。その他の構成は同様であり、同一の符号を付してその説明を省略する。
広帯域受信手段(参照系広帯域受信手段)111は、受信アンテナ102により受信された直接波の受信信号(RF信号)を広帯域で受信してA/D変換を行い、周波数帯域毎に分割するものである。
実施の形態4では、複数の周波数帯域の送信信号の振幅・位相変動及び受信機の周波数特性による振幅・位相変動を簡易な構成でリアルタイムに推定することで、複数周波数帯域の信号を損失なく合成する方式について示す。
周波数帯域の送信信号は以下のように書ける。
ここで、sn(t)は周波数チャンネル#nの変調系列(放送/通信データ等)、fc,nは周波数チャンネル#nの中心周波数(搬送波周波数)、φnは周波数チャンネル#nの信号の初期位相である。また、衛星の直接波を受信する受信アンテナ/受信機を参照系、目標散乱波を受信する受信アンテナ/受信機を捜索系と呼ぶ。参照系、捜索系は同期していることを前提とする。
ここで、周波数チャンネル#nの目標散乱波受信時の複素振幅をβn、衛星-目標-捜索系までの伝播遅延時間をτB、目標のバイスタティック速度をvB、光速をcとすると、以下のように書ける。
ここで、fb,n=vB/c*fc,nであり、目標のバイスタティックドップラー周波数である。
また、目標のRCSの周波数特性が、振幅が一定で位相のみが線形に変化する場合、すなわち、exp(j2πfc,nτRCS)と書ける場合においても上記の周波数チャンネル方向のIFFTにより目標が積分されることが分かる。ただし、その場合、ピークとなる遅延時間インデックスがτRCS分だけシフトすることに注意が必要である。
なお、帯域合成手段108は、帯域間補償手段112による補償後の周波数帯域毎の相互相関結果を帯域合成する。
送信源201からの直接波は、一般に電力が大きいため、捜索系である受信アンテナ101のサイドローブあるいはバックローブから混入する。ここでは、受信アンテナ101に混入した直接波を用いて帯域間の振幅・位相特性の補償を行う。捜索系である受信アンテナ101と参照系である受信アンテナ102がほぼ同一の場所に置かれており(信号帯域の逆数×光速の範囲内)、送信源201が受信機から見て静止しているとみなせる場合、捜索系の受信信号と参照系の受信信号の相互相関関数において、送信源201から受信アンテナ101に混入した直接波は、ドップラー周波数インデックスk=0、遅延時間インデックスl=0の箇所にピークsdir,nとして現れる。
その後、帯域間補償手段112は、式(20)のように帯域間の補償を行った相互相関関数を帯域合成手段108に送る。これにより、損失無く合成することができる。
一方、直接波の漏れ込みが極端に低く相互相関関数のピークとして現れない場合には、受信アンテナ101を送信源201方向に一時的に向けて補正係数を算出し、再度、受信アンテナ101を目標方向に指向するという構成も考えられる。即ち、帯域間補償手段112は、受信アンテナ101を直接波方向に向けて取得した周波数帯域毎の反射波の受信信号と直接波の受信信号の振幅比及び位相差を補正係数とし、当該補正係数により当該周波数帯域毎の相互相関結果を補償するように構成してもよい。
実施の形態5では、図1に示す実施の形態1における帯域合成手段108において、遅延時間方向のデータを並び替えることで高分解能な相互相関関数を得る場合について示す。なお、実施の形態5に係るパッシブレーダ装置は、図1に示す実施の形態1に係るパッシブレーダ装置の構成と同様であり、同一の符号を付して異なる部分についてのみ説明を行う。
そこで、実施の形態5では、相互相関処理結果の遅延時間インデックスを用いて、遅延に対応する位相を補償した後に帯域合成を行うことにより、遅延時間のアンビギュイティを解消し、正確な遅延時間を得ることを目的とする。
ここで、m(=0,1,・・・,N-1)は高分解能遅延時間インデックスである。
実施の形態1~5では捜索系が1系統の場合について示したが、実施の形態6では複数系統を備えた構成について示す。図9はこの発明の実施の形態6に係るパッシブレーダ装置の構成を示す図である。図9に示す実施の形態6に係るパッシブレーダ装置は、図1に示す実施の形態1に係るパッシブレーダ装置の受信アンテナ101、帯域分割手段103、受信手段105、相互相関処理手段107、帯域合成手段108、ピーク検出手段109を複数系統設け、ビーム形成手段113を追加したものである。その他の構成は同様であり、同一の符号を付してその説明を省略する。
なお、複数の目標探知手段109は、ビーム形成手段113によるビーム形成結果に基づいて目標を探知する。
ここで、θqはビーム指向方向である。
また、χ[lT,kb,n,m]の絶対値(振幅値)又は電力値を算出した後にアンテナ数分だけ合成(加算又は平均)する構成としても構わない。
また、ビーム形成手段113は、相互相関処理手段107による処理直後にビーム形成を行う構成としてもよい。
実施の形態7では、帯域合成後の相互相関関数を時間方向に積分することで、効率的に長時間の積分を行う構成について示す。図10はこの発明の実施の形態7に係るパッシブレーダ装置の構成を示す図である。図10に示す実施の形態7に係るパッシブレーダ装置は、図1に示す実施の形態1に係るパッシブレーダ装置に長時間積分手段114を追加したものである。その他の構成は同様であり、同一の符号を付してその説明を省略する。
なお、目標探知手段109は、長時間積分手段114による積分結果に基づいて目標を探知する。
Claims (24)
- 電波源から送信されて目標で反射された反射波を受信する捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記捜索系受信アンテナにより受信された反射波の受信信号を周波数帯域毎に分割する捜索系帯域分割手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を周波数帯域毎に分割する参照系帯域分割手段と、
前記捜索系帯域分割手段により分割された周波数帯域毎の反射波の受信信号を受信してA/D変換を行う捜索系受信手段と、
前記参照系帯域分割手段により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行う参照系受信手段と、
前記捜索系受信手段によりA/D変換された反射波の受信信号と、前記参照系受信手段によりA/D変換された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域合成する帯域合成手段と、
前記帯域合成手段による帯域合成結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 前記捜索系帯域分割手段及び前記参照系帯域分割手段は、前記受信信号を周波数帯域毎にアナログ段で分割する
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記捜索系帯域分割手段及び前記参照系帯域分割手段は、ローカル信号発信源を有し、当該ローカル信号発信源からの周波数を時系列に変更した信号と、前記受信信号とのミキシングを行うことで、当該受信信号を周波数帯域毎に分割する
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記相互相関処理手段は、
前記参照系受信手段からの周波数帯域毎の直接波の受信信号を、想定する目標のバイスタティックドップラー速度及び当該周波数帯域毎の搬送波周波数から算出したドップラー周波数の分だけ周波数シフトするドップラー周波数シフト部と、
前記捜索系受信手段からの周波数帯域毎の反射波の受信信号及び前記ドップラー周波数シフト部により周波数シフトされた周波数帯域毎の直接波の受信信号を高速フーリエ変換するFFT部と、
前記FFT部により高速フーリエ変換された周波数帯域毎の直接波の受信信号の複素共役を取り、前記FFT部により高速フーリエ変換された周波数帯域毎の反射波の受信信号と乗算する複素共役乗算部と、
前記複素共役乗算部による複素共役乗算結果を逆高速フーリエ変換するIFFT部とを備えた
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記相互相関処理手段は、
前記捜索系受信手段からの周波数帯域毎の反射波の受信信号及び前記参照系受信手段からの周波数帯域毎の直接波の受信信号を、それぞれブロック毎に分割するブロック分割部と、
前記ブロック分割部により分割された周波数帯域及びブロック毎の反射波の受信信号、並びに周波数帯域及びブロック毎の直接波の受信信号を高速フーリエ変換するFFT部と、
前記FFT部により高速フーリエ変換された周波数帯域及びブロック毎の直接波の受信信号の複素共役を取り、前記FFT部により高速フーリエ変換された周波数帯域及びブロック毎の反射波の受信信号と乗算する複素共役乗算部と、
前記複素共役乗算部による複素共役乗算結果を逆高速フーリエ変換するIFFT部と、
前記IFFT部による逆高速フーリエ変換結果をブロック方向に高速フーリエ変換するブロック間FFT部とを備えた
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記捜索系ブロック分割部は、時間間隔Tb毎に前記受信信号を分割した後に、当該分割した各ブロックの受信信号に次のブロックの受信信号を追加することで、時間間隔2Tbのブロックを生成し、
前記参照系ブロック分割部は、前記時間間隔Tb毎に前記受信信号を分割した後に、当該分割した各ブロックの受信信号に前記時間間隔Tb分の0信号を追加することで、時間間隔2Tbのブロックを生成する
ことを特徴とする請求項5記載のパッシブレーダ装置。 - 前記捜索系ブロック分割部は、時間間隔Tb毎に前記受信信号を分割し、時間間隔Tbのブロックを生成し、
前記参照系ブロック分割部は、時間間隔Tb毎に前記受信信号を分割し、時間間隔Tbのブロックを生成する
ことを特徴とする請求項5記載のパッシブレーダ装置。 - 前記帯域合成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域方向に逆フーリエ変換することで、帯域合成を行う
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記帯域合成手段は、前記各周波数帯域の間隔が一定である場合に、前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域方向に逆高速フーリエ変換することで、帯域合成を行う
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記帯域合成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果のドップラー周波数からドップラー速度を算出し、当該ドップラー速度が一致又は近い相互相関結果を帯域方向に逆高速フーリエ変換することで、帯域合成を行う
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 電波源から送信されて目標で反射された反射波を受信する捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記捜索系受信アンテナにより受信された反射波の受信信号を広帯域で受信してA/D変換を行い、周波数帯域毎に分割する捜索系広帯域受信手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を広帯域で受信してA/D変換を行い、周波数帯域毎に分割する参照系広帯域受信手段と、
前記捜索系広帯域受信手段により分割された反射波の受信信号と、前記参照系広帯域受信手段により分割された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域合成する帯域合成手段と、
前記帯域合成手段による帯域合成結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 前記帯域合成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果のドップラー周波数の比が、当該周波数帯域毎の搬送波周波数の比に一致又は近い相互相関結果を帯域方向に逆フーリエ変換することで、帯域合成を行う
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 前記帯域合成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果に対して、当該相互相関結果の遅延時間インデックスに依存する周波数帯域毎の位相変動を補償し、当該周波数帯域毎の逆高速フーリエ変換を行った帯域間合成後の相互相関結果を遅延時間方向に並べ替える
ことを特徴とする請求項1記載のパッシブレーダ装置。 - 電波源から送信されて目標で反射された反射波を受信する捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記捜索系受信アンテナにより受信された反射波の受信信号を周波数帯域毎に分割する捜索系帯域分割手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を周波数帯域毎に分割する参照系帯域分割手段と、
前記捜索系帯域分割手段により分割された周波数帯域毎の反射波の受信信号を受信してA/D変換を行う捜索系受信手段と、
前記参照系帯域分割手段により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行う参照系受信手段と、
前記捜索系受信手段によりA/D変換された反射波の受信信号と、前記参照系受信手段によりA/D変換された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果に対して、当該周波数帯域毎の振幅及び位相変動を補償する帯域間補償手段と、
前記帯域間補償手段による補償後の周波数帯域毎の相互相関結果を帯域合成する帯域合成手段と、
前記帯域合成手段による帯域合成結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 前記帯域間補償手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果における直接波の漏れ込み成分の振幅比及び位相差を用いて当該周波数帯域毎の振幅及び位相変動を補償する補正係数を算出し、当該補正係数により相互相関結果を補償する
ことを特徴とする請求項14記載のパッシブレーダ装置。 - 前記帯域間補償手段は、前記捜索系受信アンテナを直接波方向に向けて取得した周波数帯域毎の反射波の受信信号と直接波の受信信号の振幅比及び位相差を補正係数とし、当該補正係数により当該周波数帯域毎の相互相関結果を補償する
ことを特徴とする請求項14記載のパッシブレーダ装置。 - 電波源から送信されて目標で反射された反射波を受信する複数の捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記複数の捜索系受信アンテナにより受信された反射波の受信信号を周波数帯域毎に分割する捜索系帯域分割手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を周波数帯域毎に分割する参照系帯域分割手段と、
前記捜索系帯域分割手段により分割された周波数帯域毎の反射波の受信信号を受信してA/D変換を行う捜索系受信手段と、
前記参照系帯域分割手段により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行う参照系受信手段と、
前記捜索系受信手段によりA/D変換された反射波の受信信号と、前記参照系受信手段によりA/D変換された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域合成する帯域合成手段と、
前記帯域合成手段による帯域合成結果を前記複数の捜索系受信アンテナ間で合成することによりビーム形成を行うビーム形成手段と
前記ビーム形成手段によるビーム形成結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 電波源から送信されて目標で反射された反射波を受信する複数の捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記複数の捜索系受信アンテナにより受信された反射波の受信信号を周波数帯域毎に分割する捜索系帯域分割手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を周波数帯域毎に分割する参照系帯域分割手段と、
前記捜索系帯域分割手段により分割された周波数帯域毎の反射波の受信信号を受信してA/D変換を行う捜索系受信手段と、
前記参照系帯域分割手段により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行う参照系受信手段と、
前記捜索系受信手段によりA/D変換された反射波の受信信号と、前記参照系受信手段によりA/D変換された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果を前記複数の捜索系受信アンテナ間で合成することによりビーム形成を行うビーム形成手段と
前記ビーム形成手段によるビーム形成結果を帯域合成する帯域合成手段と、
前記ビーム形成手段による帯域合成結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 前記ビーム形成手段は、前記帯域合成手段による帯域合成結果に対して、搬送波周波数、ビーム指向方向及び前記捜索系受信アンテナの位置関係に依存した位相差を補償するようビーム形成を行う
ことを特徴とする請求項17記載のパッシブレーダ装置。 - 前記ビーム形成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果に対して、搬送波周波数、ビーム指向方向及び前記捜索系受信アンテナの位置関係に依存した位相差を補償するようビーム形成を行う
ことを特徴とする請求項18記載のパッシブレーダ装置。 - 前記ビーム形成手段は、前記帯域合成手段による帯域合成結果に対して、振幅値又は電力値を算出した後に前記捜索系受信アンテナ分だけ加算又は平均する
ことを特徴とする請求項17記載のパッシブレーダ装置。 - 前記ビーム形成手段は、前記相互相関処理手段による周波数帯域毎の相互相関結果に対して、振幅値又は電力値を算出した後に前記捜索系受信アンテナ分だけ加算又は平均する
ことを特徴とする請求項18記載のパッシブレーダ装置。 - 電波源から送信されて目標で反射された反射波を受信する捜索系受信アンテナと、
前記電波源から送信された直接波を受信する参照系受信アンテナと、
前記捜索系受信アンテナにより受信された反射波の受信信号を周波数帯域毎に分割する捜索系帯域分割手段と、
前記参照系受信アンテナにより受信された直接波の受信信号を周波数帯域毎に分割する参照系帯域分割手段と、
前記捜索系帯域分割手段により分割された周波数帯域毎の反射波の受信信号を受信してA/D変換を行う捜索系受信手段と、
前記参照系帯域分割手段により分割された周波数帯域毎の直接波の受信信号を受信してA/D変換を行う参照系受信手段と、
前記捜索系受信手段によりA/D変換された反射波の受信信号と、前記参照系受信手段によりA/D変換された直接波の受信信号との周波数帯域毎の相互相関を行う相互相関処理手段と、
前記相互相関処理手段による周波数帯域毎の相互相関結果を帯域合成する帯域合成手段と、
前記帯域合成手段による帯域合成結果を長時間積分する長時間積分手段と
前記長時間積分手段による積分結果に基づいて前記目標を探知する目標探知手段と
を備えたパッシブレーダ装置。 - 前記長時間積分手段は、前記相互相関処理手段による相互相関結果に対して、前記目標の移動に応じた複数の仮説に基づき時間方向に積分する
ことを特徴とする請求項23記載のパッシブレーダ装置。
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