DK147428B - FILTER EQUIPMENT TO REMOVE UNSUSED ECHO SIGNALS IN AN IMPULS-Doppler RADAR RECEIVER - Google Patents
FILTER EQUIPMENT TO REMOVE UNSUSED ECHO SIGNALS IN AN IMPULS-Doppler RADAR RECEIVER Download PDFInfo
- Publication number
- DK147428B DK147428B DK356178AA DK356178A DK147428B DK 147428 B DK147428 B DK 147428B DK 356178A A DK356178A A DK 356178AA DK 356178 A DK356178 A DK 356178A DK 147428 B DK147428 B DK 147428B
- Authority
- DK
- Denmark
- Prior art keywords
- filter
- frequency
- glitter
- signals
- output
- Prior art date
Links
Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
- G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
- G01S13/50—Systems of measurement based on relative movement of target
- G01S13/52—Discriminating between fixed and moving objects or between objects moving at different speeds
- G01S13/522—Discriminating between fixed and moving objects or between objects moving at different speeds using transmissions of interrupted pulse modulated waves
- G01S13/524—Discriminating between fixed and moving objects or between objects moving at different speeds using transmissions of interrupted pulse modulated waves based upon the phase or frequency shift resulting from movement of objects, with reference to the transmitted signals, e.g. coherent MTi
- G01S13/526—Discriminating between fixed and moving objects or between objects moving at different speeds using transmissions of interrupted pulse modulated waves based upon the phase or frequency shift resulting from movement of objects, with reference to the transmitted signals, e.g. coherent MTi performing filtering on the whole spectrum without loss of range information, e.g. using delay line cancellers or comb filters
Landscapes
- Engineering & Computer Science (AREA)
- Radar, Positioning & Navigation (AREA)
- Remote Sensing (AREA)
- Computer Networks & Wireless Communication (AREA)
- Physics & Mathematics (AREA)
- General Physics & Mathematics (AREA)
- Radar Systems Or Details Thereof (AREA)
- Ultra Sonic Daignosis Equipment (AREA)
- Surgical Instruments (AREA)
- Sheet Holders (AREA)
- Electrotherapy Devices (AREA)
Description
147428147428
Opfindelsen angår en filterindretning, der indgår i en impuls-Doppler-radars modtager med henblik på bortfiltrering af uønskede ekkosignaler, der f.eks. kommer fra jordoverfladen, havoverfladen eller regn, hvilken filterindretning er af den i krav 1's indledning angivne art.BACKGROUND OF THE INVENTION 1. Field of the Invention This invention relates to a filter device included in the receiver of an impulse Doppler radar for filtering out unwanted echo signals, e.g. comes from the soil surface, sea surface or rain, which filter device is of the kind specified in the preamble of claim 1.
En radarmodtager har til formål bl.a. at kunne eliminere sådanne uønskede radar-ekkoer eller "glitter", som skyldes refleksion fra uinteressante mål, som f.eks. jord, hav eller nedbør (regn eller sne), og alene detektere det ønskede bevægelige mål, f.eks. et fly. Til dette formål udnyttes hastighedsforskellen mellem de uønskede mål og det eller de ønskede mål. I en kohærent impuls--Doppler-radar af kendt art udsendes der et impulsmoduleret højfrekvent signal med en bestemt bærefrekvens fo, som efter refleksion fra et bevægeligt mål kommer tilbage med en bestemt ændret frekvens 2 147428 fo - fd, hvor ændringen fd afhænger af Doppler-forskydningen, dvs. det bevægelige måls radialhastighed i forhold til radarstationen.The purpose of a radar receiver is being able to eliminate such unwanted radar echoes or "glitter" that are due to reflection from uninteresting targets such as land, sea or rainfall (rain or snow), and only detect the desired moving target, e.g. a plane. For this purpose, the speed difference between the unwanted targets and the desired target (s) is utilized. In a coherent pulse - Doppler radar of a known kind, a pulse-modulated high-frequency signal with a specific carrier frequency fo is emitted, which, after reflection from a moving target, returns at a certain altered frequency 2 where the change fd depends on Doppler displacement, i.e. the radial velocity of the moving target relative to the radar station.
Det indkommende ekkosignal blandes i modtageren med bærefrekvensen fo, hvorved Doppler-frekvensen fd fremkommer. Dersom det udsendte signal med bærefrekvensen fo ikke var impulsmoduleret, ville der i modtageren fremkomme et rent sinus-signal, hvis frekvens er Doppler-frekvensen fd. Eftersom det udsendte signal er impulsmoduleret med en impulsfrekvens fp = 1/T, hvor T er periodetiden, vil modtageren afgive et impulsmoduleret sinussignal, hvorved den modulerede størrelses frekvens er lig med Doppler-frekvens fd. Det modtagne signal er ikke et rent sinussignal (impulsmoduleret), eftersom det indeholder hastighedskomposanter (sinuskomposanter) også fra uønskede mål (jord, hav, osv.) foruden hastighedskomposan-ten fra det ønskede mål. De modtagne og i modtageren blandede signaler vil således indeholde et antal ønskede og et antal ikke ønskede frekvenskomposanter.The incoming echo signal is mixed in the receiver with the carrier frequency f0, thereby producing the Doppler frequency fd. If the transmitted signal with the carrier frequency fo was not pulse modulated, a pure sine signal would appear in the receiver whose frequency is the Doppler frequency fd. Since the transmitted signal is pulse modulated with a pulse frequency fp = 1 / T, where T is the period of time, the receiver will output a pulse modulated sine signal, the frequency of the modulated size being equal to the Doppler frequency fd. The received signal is not a pure sine wave signal (pulse modulated) since it contains velocity components (sine components) also from unwanted targets (earth, sea, etc.) in addition to the speed component of the desired target. Thus, the received and mixed signals in the receiver will contain a number of desired and a number of undesired frequency components.
Det er tidligere kendt i en impuls-Doppler-radars modtager at anbringe filtre, nemlig de såkaldte Doppler-filtre, hvis funktion det er i størst mulig udstrækning at undertrykke de frekvenskomposan-ter, der skriver sig fra ikke ønskede mål, dvs. de lavfrekvente kom-posanter, der skriver sig fra jord, hav/bølger og nedbør. Doppler-fil-teret kan bestå af et digitalt filter, som bortfiltrerer de komposan-ter, hvis frekvens er mindre end en bestemt værdi svarende til en bestemt målhastighed. Inden for det af radarsenderens periodetid T bestemte frekvensbånd udviser et sådant Doppler-filter en bestemt karakteristik, således som det på tegningens fig. 1 er antydet i afbrudt streg. Det er herved ønskeligt, at filteret for lave frekvenser, f.eks. mindre end 1/8T, udviser spærre-egenskaber, mens det for højere frekvensværdier udviser båndpaskarakteristik, så at eventuelle bevægelige mål, hvis radialhastighed er større end glitterets; kan detekteres. Doppler-filterets anvendelighed afhænger imidlertid af størrelsen af dets spærrebånd. Dersom filter-spærrebåndets øverste grænsefrekvens fmax^l/8T og dersom radarimpulsernes periodetid T er begrænset nedad af den ønskede rækkevidde R, gælder det at T = 2Rmax/c, hvor c = forplantningshastigheden, og den største glitterhastighed inden for filterspærrebåndet bliver vmax = )\c/16Rmax, hvor \ = radarbølgelængden. For eksempelvis \= 0,1m(S-båndet) og 4It is known in the prior art of an impulse-Doppler radar receiver to apply filters, namely the so-called Doppler filters, whose function is to suppress as much as possible the frequency components arising from unwanted targets, ie. the low-frequency compositions that derive from land, sea / waves and precipitation. The Doppler filter may consist of a digital filter which filters out those components whose frequency is less than a certain value corresponding to a certain target speed. Within the frequency band determined by the radar transmitter period T, such a Doppler filter exhibits a certain characteristic, such as that shown in FIG. 1 is indicated in dash. It is hereby desirable that the filter for low frequencies, e.g. less than 1 / 8T, exhibits latching characteristics while for higher frequency values exhibits bandpass characteristics such that any moving targets whose radial velocity is greater than that of the glitter; can be detected. However, the applicability of the Doppler filter depends on the size of its blocking tape. If the upper limit frequency of the filter bar is fmax1 / 8T and if the period time T of the radar pulses is limited downwards by the desired range R, then T = 2Rmax / c, where c = propagation speed, and the largest glitter speed within the filter bar is vmax =) \ c / 16Rmax, where \ = radar wavelength. For example, \ = 0.1m (S-band) and 4
Rmax = 10 m, bliver vmaxa/2m/s, hvad der medfører, at filteret 147428 3 kun kan undertrykke jordglitter, mens tilbageværende glitter med højere frekvenskomposanter forbliver upåvirket.Rmax = 10 m, vmaxa / 2m / s becomes, which means that the filter 147428 3 can only suppress ground glitter while remaining glitter with higher frequency components remains unaffected.
I det tilfælde, hvor radaren arbejder i en lav PRF-modus, dvs. at periodetiden T er således afpasset, at alle radar-ekkoer af interesse reflekteres og modtages inden den næste radarimpuls udsendes, betyder dett^, at målets Doppler-frekvens fd kan være større end impulsgentagelsesfrekvensen 1/T. Dette medfører igen, således som det kan ses af fig. 1, at selv mål-ekkoet kan blive undertrykt af Doppler-filteret ved visse såkaldte blinde hastigheder, nærmere bestemt de hastigheder, der giver en Doppler-frekvens, som er et multiplum af frekvensen 1/T. Det er kendt at undgå undertrykkelse af sådanne mål-ekkoer ved at indføre den såkaldte "staggering", dvs. at periodetiden T bringes til at variere fra én impuls til den næste inden for de udsendte radarimpulser.In the case where the radar operates in a low PRF mode, ie. that the period time T is such that all radar echoes of interest are reflected and received before the next radar pulse is emitted, this means that the target's Doppler frequency fd may be greater than the pulse repetition frequency 1 / T. This again results, as can be seen in FIG. 1, that even the target echo can be suppressed by the Doppler filter at certain so-called blind speeds, in particular those speeds which give a Doppler frequency, which is a multiple of the frequency 1 / T. It is known to avoid suppression of such target echoes by introducing the so-called "staggering", ie. that period time T is caused to vary from one pulse to the next within the radar pulses emitted.
En anden kendt metode til at eliminere det ikke ønskede glitterspektrum består i at udføre en hastighedskompensering inden filtreringen i Doppler-filteret. Dette indebærer, at glitterhastigheden bedømmes eller skønnes, f.eks. ved fasemåling, under på hinanden følgende afsøgninger. Ved f.eks. at styre modtagerens lokaloscillator kan glitterspektret forskydes, så at dets dominerende komposant antager værdien nul og på denne måde kommer til at ligge inden for filterspærrebåndet. Denne metode f orudsætter imidlertid, at glitterspektret har en dominerende komposant, som kan bedømmes korrekt.Another known method of eliminating the unwanted glitter spectrum is to perform a velocity compensation prior to filtering in the Doppler filter. This implies that the glitter speed is judged or estimated, e.g. by phase measurement, successive scans. For example, To control the receiver's local oscillator, the glitter spectrum can be shifted so that its dominant component assumes the value zero and thus lies within the filter barrier band. However, this method assumes that the glitter spectrum has a dominant component that can be judged correctly.
Fra britisk patentskrift nr. 1.301.299 er det kendt at eliminere uønskede ekkosignaler (glitter) i dopplersignaler med stor variation i de uønskede ekkosignalers frekvensspektrum. Det i dette patentskrift omhandlede anlæg arbejder med analoge signaler og analoge filtre, såkaldte aktive filtre, selv om det i slutningen af beskrivelsen- til dette patentskrift er nævnt, at digital teknik kan anvendes.From British Patent Specification No. 1,301,299, it is known to eliminate unwanted echo signals (glitter) in Doppler signals with wide variation in the frequency spectrum of the unwanted echo signals. The system referred to in this patent works with analog signals and analog filters, so-called active filters, although it is mentioned at the end of the description of this patent that digital technology can be used.
US-patentskrift nr. 3.011.166 omhandler et impulsdoppler-radaranlæg, som arbejder med to forskellige impulsrepetitionsfrekvenser (PRF1 og PRF2), dog uden at der anvendes "staggering".U.S. Patent No. 3,011,166 discloses an impulse Doppler radar system which operates at two different pulse repetition frequencies (PRF1 and PRF2), however, without the use of "staggering".
Det problem, der skal løses med dette anlæg, er at eliminere u-ønskede ekkosignaler med en vis frekvens ved anvendelse af disse to forskellige impulsrepetitionsfrekvenser. Modtageren i dette kendte anlæg indeholder til dette formål blandingstrin, til hvilke et modtaget radarsignal altid indeholder den ene impulsrepetitions- 4 147628 frekvens, hvorved det bevægelige måls frekvens (hastighed) kan bestemmes.The problem to be solved with this system is to eliminate unwanted echo signals at a certain frequency using these two different pulse repetition frequencies. For this purpose, the receiver of this known system contains mixing steps to which a received radar signal always contains the one pulse repetition frequency, whereby the frequency (speed) of the moving target can be determined.
Filterindretningen ifølge opfindelsen er baseret på den tidligere i sig selv kendte metode, men udviser den yderligere fordel, at bedømmelsen af det bevægelige glitters hastighed udføres efter bortfiltrering af jordglitteret, hvorved opnås, at dette glitter ikke kommer til at påvirke den efterfølgende bedømmelse.The filter device according to the invention is based on the method known per se, but has the additional advantage that the speed of the moving glitter is evaluated after filtering off the ground glitter, thereby ensuring that this glitter will not affect the subsequent rating.
Opfindelsens formål er at anvise udformningen af en Doppler-filterindretning, der indgår i en impuls-Doppler-radars modtager, hvormed såvel filtreringen af lavfrekvent jord- og havglitter som. filtreringen af tilbageværende glitter med højere Doppler-frekvens kan udføres ved anvendelse af digitale filtre af i sig selv kendt art.The object of the invention is to provide the design of a Doppler filter device which forms part of an impulse Doppler radar receiver, which allows both the filtering of low frequency ground and sea glitter as well. the filtration of residual glitter with higher Doppler frequency can be carried out using digital filters of a kind known per se.
Det angivne formål opnås ifølge opfindelsen ved en filterindretning af den indledningsvis omhandlede art, som udmærker sig ved den r krav l's kendetegnende del angivne udformning.According to the invention, the stated object is achieved by a filter device of the kind described initially, which is characterized by the design of the characterizing part of claim 1.
Den foreliggende opfindelse angår således en filterindretning, som indeholder digitale filtre, der til forskel fra de i eksempelvis britisk patentskrift nr. 1.301.299 omhandlede analoge filtre, har en væsentlig anderledes frekvenskarakteristik, hvorved der opstår særlige problemer ved filtreringen, som ikke er omtalt i dette patentskrift. Ved udsendelse af radarsignaler fra det radarudstyr, i hvilket filterindretningen ifølge opfindelsen indgår, anvendes til forskel fra anlægget i det britiske patentskrift såkaldt "staggering", der som nævnt består i, at periodetiden eller impulsrepetitionsfrekvensen varierer fra en impuls til den næste. Det kendte anlæg angår således ikke doppler-filtrering med digitale filtre, hvori der anvendes "staggering", hvorfor forudsætningerne er helt forskellige fra den foreliggende opfindelses. Det fra det britiske patentskrift nr. 1.301.299 kendte anlæg indeholder et lavpasfilter samt et højpasfilter med variabelt frekvenssvar, som ved hjælp af et styresignal styres i afhængighed af en vis parameter eller komposant i glittersignalet. Der foretages ikke nogen hastighedskompensering af det lavpasfiltrerede dopplersignal i dette kendte anlæg.The present invention thus relates to a filter device which contains digital filters which, unlike, for example, British Patent Specification No. 1,301,299, have a substantially different frequency characteristic, thereby causing particular problems with the filtering which are not disclosed in this patent. When transmitting radar signals from the radar equipment in which the filter device according to the invention is included, unlike the system in the British patent, so-called "staggering", which means that the period time or the pulse repetition frequency varies from one pulse to the next. Thus, the prior art does not relate to Doppler filtering with digital filters using "staggering", so the premises are quite different from the present invention. The system known from British Patent No. 1,301,299 contains a low-pass filter and a high-pass variable frequency response filter which is controlled by means of a control signal depending on a certain parameter or component of the glitter signal. No speed compensation of the low-pass filtered Doppler signal is performed in this known system.
Problemstillingen for det i US-patentskrift nr. 3.011.166 omhandlede impulsdopplerradaranlæg er ikke den samme som for fil- 147428 5 terindretningen ifølge den foreliggende opfindelse, selv om formålet er det samme, nemlig at eliminere uønskede ekkosignaler.The problem of the pulse Doppler radar system disclosed in US Patent No. 3,011,166 is not the same as that of the filter device of the present invention, although the purpose is the same, namely to eliminate unwanted echo signals.
Den foreliggende opfindelse anvendes til et radaranlæg med "staggering" hvilket giver anledning til det problem, at de indgående digitale filtre får uregelmæssig frekvenskarakteristik. Dette problem er ikke omtalt i US-patentskriftet.The present invention is applied to a "staggering" radar system which gives rise to the problem that the incoming digital filters get irregular frequency characteristics. This problem is not disclosed in U.S. Pat.
Opfindelsen skal i det følgende forklares nærmere under henvisning til tegningen, hvor fig. 1 er et frekvensdiagram, som viser dels frekvensspektret for et modtaget radarsignal, dels en bestemt valgt filterkarakteristik, fig. 2 er et blokdiagram over visse dele, der indgår i en radarmodtager, hvilke dele ligger foran Doppler-filterindretningen ifølge opfindelsen, fig. 3 er et blokdiagram, der viser princippet for Doppler--filterindretningen ifølge opfindelsen, fig. 4 viser nærmere udformningen af et i sig selv kendt digitalt filter, der indgår i den i fig. 3 viste filterindretning, og fig. 5 er et blokdiagram over et udførelseseksempel på en Doppler-filterindretning ifølge opfindelsen.The invention will now be explained in more detail with reference to the drawing, in which 1 is a frequency diagram showing partly the frequency spectrum of a received radar signal and partly a particular selected filter characteristic; FIG. Fig. 2 is a block diagram of certain parts forming part of a radar receiver which are in front of the Doppler filter device according to the invention; Figure 3 is a block diagram showing the principle of the Doppler filter device according to the invention; 4 shows in more detail the design of a digital filter known per se which is included in the FIG. 3, and FIG. 5 is a block diagram of an embodiment of a Doppler filter device according to the invention.
Frekvensdiagrammet i fig. 1 viser et glitterspektrum samt det indkommende ekko fra et mål i en vis afstand fra radarudstyret. Filterkarakteristikken for et digitalt filter, der indgår i Doppler-filterindretningen, er antydet med afbrudt streg, og udviser spærrebånd, dels for lave frekvenser, f.eks. frekvenser mindre end 1/8T, dels for frekvenser mellem 7/8T og 9/8T, samt derimellem et pasbånd. Filterkarakteristikken er herved periodisk med en periode på 1/T. Det ønskede måls spektrum er betegnet med s, og det bevægelige glitter har et dominerende spektrum sD, hvis middelfrekvens er betegnet med fD. Doppler-filteret, hvis opbygning skal forklares nærmere under henvisning til fig. 4 og 5, har herved til opgave dels at undertrykke jordglitterspektret sg, dels at undertrykke spektret sD for det dominerende bevægelige glitter, som først og fremmest skyldes nedbør (regn og sne).The frequency diagram of FIG. 1 shows a glitter spectrum as well as the incoming echo from a target at some distance from the radar equipment. The filter characteristic of a digital filter included in the Doppler filter device is indicated by a dashed line, and exhibits blocking bands, partly for low frequencies, e.g. frequencies less than 1 / 8T, partly for frequencies between 7 / 8T and 9 / 8T, and in between a pass band. The filter characteristic is hereby periodic with a period of 1 / T. The spectrum of the desired target is denoted by s, and the moving glitter has a dominant spectrum sD, whose average frequency is denoted by fD. The Doppler filter, the structure of which is explained in more detail with reference to FIG. 4 and 5, are hereby tasked partly to suppress the earth glitter spectrum sg, and partly to suppress the spectrum sD of the dominant moving glitter, which is primarily due to rainfall (rain and snow).
Med henblik på at belyse signalbehandlingen og opbygningen af filterindretningen ifølge opfindelsen, skal først under henvisning til fig. 2 de enheder, som er anbragt foran filterindretningen, skrives nærmere. Over indgangen A optræder et signal A(t) = cos-[2tf (fo+fd)t+ ] fra radarmodtagerens S/M (sende/modtage)-skifter.In order to illustrate the signal processing and structure of the filter device according to the invention, first with reference to FIG. 2, the units located in front of the filter device are written in more detail. Above the input A, a signal A (t) = cos- [2tf (fo + fd) t +] appears from the radar receiver's S / M (send / receive) switch.
147428 6147428 6
Frekvens fd udgør det ønskede måls Doppler-frekvens. Signalet A(t) tilføres to kanaler, I og Q, som hver indeholder en blander Bl henholdsvis B2, samt en analog/digital-omsætter ADI henholdsvis AD2. Til hver blander Bl, B2 tilføres et referencesignal cos 2ΊΤ fot henholdsvis sin fot fra en referenceoscillator OSC i modtageren. Herved fremkommer udgangssignalerne cos (21^-fdt+ $) henholdsvis sin (2/f»'fdt+^), som tilføres hver af de to analog/digital-omsættere ADI og AD2. I disse omsættere eksempleres signalerne ved eksempleringstidspunkterne tn ved hjælp af klokimpulser fra en klok-kreds CL, så at der i de respektive kanaler I og Q fremkommer udgangssignaler X-j. og Xq, såledesFrequency fd is the Doppler frequency of the desired target. The signal A (t) is applied to two channels, I and Q, each containing a mixer B1 and B2 respectively, as well as an analog / digital converter ADI and AD2 respectively. To each mixer B1, B2, a reference signal cos 2ΊΤ feet respectively is fed from a reference oscillator OSC in the receiver. This results in the outputs cos (21 ^ -fdt + $) and sin (2 / f »'fdt + ^, respectively), which are supplied to each of the two analog / digital converters ADI and AD2. In these transducers, the signals at the sampling times tn are exemplified by clock pulses from a clock circuit CL so that output signals X-j appear in the respective channels I and Q. and Xq, thus
Xj = cos (2ir fdtn+ jf)Xj = cos (2ir fdtn + jf)
Xq = sin (21/ fdtn+ <P)Xq = sin (21 / fdtn + <P)
Signalerne X-(tn) og XQ(tn) kan repræsenteres af X(tn) = ej * fdtn (ijT=Q) ^ idet ej2/^ tn = cos(21i» fdtn)+j sin(2W fdtn), således at .2irfdtn[ Γ .21Tfdtnl Χχ = Reje3 I og XQ = linje3 J.The signals X- (tn) and XQ (tn) can be represented by X (tn) = ej * fdtn (ijT = Q) ^ where ej2 / ^ tn = cos (21i »fdtn) + j sin (2W fdtn), so that .2irfdtn [Γ .21Tfdtnl Χχ = Shrimp3 I and XQ = line3 J.
Eksempleringstidspunkterne kan indtræffe på en sådan måde, at der udføres en regelmæssig eksemplering, dvs. at tn = nT (n = 1, 2, 3, ...), eller også på en sådan måde, at tiden mellem på hinanden følgende eksempleringsimpulser varierer inden for et bestemt tidsinterval NT, men det samme eksempleringsmønster optræder igen efter eksempleringstidspunktet tn = NT, dvs. såkaldt "staggering". For det sidstnævnte tilfælde gælder, at eksempleringen sker ved tidspunkterne i) NT+tk, hvor 0=0, 1... og k = 0, 1, 2, ..., N-l.The sampling times may occur in such a manner that a regular sampling is carried out, ie. tn = nT (n = 1, 2, 3, ...), or also in such a way that the time between consecutive sample pulses varies within a specified time interval NT, but the same sample pattern reappears after the sample time tn = NT, ie so-called "staggering". In the latter case, the sampling occurs at the times i) NT + tk, where 0 = 0, 1 ... and k = 0, 1, 2, ..., N-l.
Princippet for Doppler-filterindretningen ifølge opfindelsen fremgår af blokdiagrammet i fig. 3. Filterindretningen indeholder et første digitalt filter DF1 af i sig selv kendt art, hensigtsmæssigt et transversalfilter, der er således dimensioneret, at det eliminerer jord- og havglitter, dvs. glitter med lav hastighed, jfr. den i fig. 1 viste filterkarakteristik. Eftersom filterkarakteristikken for et digitalt filter er periodisk med en periode, der er lig med den reciprokke værdi af eksempleringsfrekvensen, genoptræder spærrebåndet ved frekvensen der svarer til visse højere hastigheder, og de genoptræder periodisk, dersom eksempleringen er regelmæssig. I tilfælde af, at eksempleringsfrekvensen varierer ("staggering"), er karakteristikken 147428 7 for filteret DFl uregelmæssig, og nogen bestemt beliggenhed af dets spærrebånd kan ikke fastlægges, undtagen for meget lave frekvenser svarende til jord- og havglitteret. Det er således i dette sidstnævnte tilfælde ikke generelt muligt at dimensionere filteret DFl således, at der samtidigt sker en udfiltrering af glitter med meget lav hastighed (jord og hav) og glitter med højere hastighed. Filterets indgangssignal er betegnet med x^(tn) og dets udgangssignal med y^ (tn).The principle of the Doppler filter device according to the invention is shown in the block diagram of FIG. 3. The filter device contains a first digital filter DF1 of a kind known per se, suitably a transversal filter which is dimensioned to eliminate land and sea glitter, ie. glitter at low speed, cf. FIG. 1 shows the filter characteristics. Since the filter characteristic of a digital filter is periodic with a period equal to the reciprocal value of the sample rate, the blocking band reappears at the frequency corresponding to certain higher speeds and they reappear periodically if the sample is regular. In case the sample frequency varies ("staggering"), the characteristic of the DF1 filter is irregular, and no particular location of its blocking band can be determined, except for very low frequencies corresponding to the land and sea glitter. Thus, in this latter case, it is not generally possible to size the filter DF1 such that at the same time a very low speed (land and sea) and higher speed glitter is filtered out. The filter input signal is denoted by x ^ (tn) and its output signal by y ^ (tn).
Udgangen af filteret DFl er forbundet med en blok HK, hvis opbygning skal beskrives nærmere under henvisning til fig. 5. Blokken HK foretager en bedømmelse af det glitter, der er tilbage efter filtreringen i det første filter DFl, og foretager en hastighedskompensering af middelfrekvensen fm for det dominerende glitterspektrum.The output of the filter DF1 is connected to a block HK, the structure of which is to be described in more detail with reference to FIG. 5. The block HK assesses the glitter left after filtering in the first filter DF1, and makes a velocity compensation of the mean frequency fm for the dominant glitter spectrum.
Denne kompensering indebære^ at samtlige frekvenskomposanter i indgangssignalet y^(tn) undergår en frekvensforskydning, så at det tilbageblivende bevægelige glitter falder under en bestemt grænsefrekvens, f.eks. under værdien 1/8T i fig. 1. Det til blokken HK tilsluttede, efterfølgende digitale filter DF2 er dimensioneret ifølge samme princip som det første filter DFl, som er således dimensioneret, at dets spærrebånd falder sammen med det glitter, hvis frekvenser har en lav værdi, dvs. jord- og havglitter.This compensation implies that all frequency components of the input signal y ^ (tn) undergo a frequency shift so that the remaining moving glitter falls below a certain limit frequency, e.g. below the value 1 / 8T of FIG. The subsequent digital filter DF2 connected to the block HK is dimensioned according to the same principle as the first filter DF1, which is dimensioned such that its blocking band coincides with the glitter whose frequencies have a low value, ie. earth and sea glitter.
Herved er - takket være udnyttelsen af "staggering" - dimensioneringsproblemet for det andet filter DF2 blevet "transformeret" til det forholdsvis enkle dimensioneringsproblem, der gælder for det første filter DFl. Filteret DF2 eliminerer således resterende bevægeligt glitter (nedbør), og den eneste forudsætning herfor er, at det resterende glitter har et dominerende spektrum, hvis middelfrekvens fm kan bedømmes i blokken HK.Hereby - thanks to the use of "staggering" - the sizing problem for the second filter DF2 has been "transformed" into the relatively simple sizing problem applicable to the first filter DF1. Thus, the DF2 filter eliminates residual moving glitter (precipitation), and the only condition is that the remaining glitter has a dominant spectrum whose average frequency fm can be judged in block HK.
Hvert filter DFl og DF2 udgøres af et digitalt filter af i sig selv kendt udformning, således som vist i fig. 4. Det i fig. 4 viste filter indeholder et antal forsinkelseskredse, f.eks. tre kredse DL1 - DL3, hver med forsinkelsen T, der er lig med radarimpulsernes periodetid. Udgangen i hver forsinkelseskreds er forbundet med en multiplikator MUO - MU3 med koefficienterne LO(n), Ll(n), L2(n) og L3(n) for filteret DFl og med koefficienterne KQ(n), K^(n), K2(n) og K^ (n) for filteret DF2, idet indekset (n) angiver, at koefficienternes værdi kan variere for forskellige eksempleringstidspunkter tn. Samtlige multiplikatorudgange er tilsluttet en additionskreds ADD.Each filter DF1 and DF2 is constituted by a digital filter of a known configuration as shown in FIG. 4. The embodiment of FIG. 4 shows a number of delay circuits, e.g. three circuits DL1 - DL3, each with the delay T equal to the period of the radar pulses. The output of each delay circuit is connected to a multiplier MUO - MU3 with the coefficients LO (n), L1 (n), L2 (n) and L3 (n) of the filter DF1 and with the coefficients KQ (n), K ^ (n), K2 (n) and K ^ (n) for the filter DF2, the index (n) indicating that the value of the coefficients may vary for different sample times tn. All multiplier outputs are connected to an ADD.
147428 8 I det følgende skal kun tilfældet med "staggering" tages i betragtning, nemlig det tilfælde, hvor to på hinanden følgende eksempleringsimpulser varierer i overensstemmelse med det ovenfor omtalte. Tilfældet med regelmæssig eksemplering er et særtilfælde, hvor tn = nT.In the following, only the case of "staggering" must be considered, namely the case where two consecutive sample pulses vary according to the above. The case of regular sampling is a special case where tn = nT.
Den ifølge opfindelsens princip foreslåede hastigskompensering af udgangssignalerne y^(tn) fra filteret DF1 skal først beskrives på grundlag af signalerne, hvorefter en dertil egnet udformning af blokken HK og det efterfølgende filter DF2 (fig. 3) skal beskrives nærmere i forbindelse med fig. 5.The speed compensation proposed according to the principle of the invention for the output signals y1 (tn) from the filter DF1 is first described on the basis of the signals, after which a suitable design of the block HK and the subsequent filter DF2 (fig. 3) will be described in more detail in connection with fig. 5th
Ved "staggering" varierer tiden mellem på hinanden følgende eksempleringsimpulser, men variationen er periodisk med perioden NT, hvad der indebærer, at filterkarakteristikken for filtrene DF1 og DF2 gentages efter tidspunkterne NT, 2NT.....Dersom indgangssignalet til filteret DF1 er (tn) = e32Tr'fd( 0 NT+tn) ^ bliver udgangssignalet fra filteret DF1 yl(tn) L±(n) e^2irfD + Vi* = i=0In "staggering", the time varies between consecutive sample pulses, but the variation is periodic with the period NT, which means that the filter characteristic of filters DF1 and DF2 is repeated after the times NT, 2NT ..... If the input signal to filter DF1 is (tn ) = e32Tr'fd (0 NT + tn) ^, the output of the filter DF1 becomes yl (tn) L ± (n) e ^ 2irfD + Vi * = i = 0
L.(n)e "^^d^W e + n> TL. (n) e "^^ d ^ W e + n> T
i=0 _ _ ... . j27rfn(ON + η) T, hvor ri = antallet af forsinkelses- ' Si 1V e u kredse i filteret DF1.i = 0 _ _ .... j27rfn (ON + η) T, where ri = the number of delay - Si 1V e u circuits in the filter DF1.
Således gælder i dette tilfælde, at amplitude og fase i udgangssignalet y^(tn), der repræsenteres af faktoren Cn(fd), er tidsafhængig.Thus, in this case, the amplitude and phase of the output signal y ^ (tn), represented by the factor Cn (fd), is time dependent.
Hastighedskompenseringen indebærer, at y^ divideres med signalet Cn(fm)e^2lr ^m^ N+n)T, ^vor £^ er resultatet af en måling af det dominerende glitterspektrums middelfrekvens fd efter filteret DF1. Således gælder, atThe velocity compensation implies that y ^ is divided by the signal Cn (fm) e ^ 2lr ^ m ^ N + n) T, ^ our £ ^ is the result of a measurement of the average glitter spectrum fd after the filter DF1. Thus, the
x,(t ) = .3,31 e - fm> (3N + η) Tx, (t) = .3.31 e - fm> (3N + η) T
Cn 9 14742$Cn 9 14742 $
Udgangssignalet fra filteret DF2 fås fraThe output of the filter DF2 is obtained
% Cn-l <fd} “j21T (f,-f ) .l.T j21T(f, - f ) (O N + n) T% Cn-l <fd} “j21T (f, -f) .l.T j21T (f, - f) (O N + n) T
K1 n C η (f.) e -e 1=0 n_1 a hvor r2 = antallet af forsinkelseskredse i filteret DF2.K1 n C η (f.) E -e 1 = 0 n_1 a where r2 = the number of delay circuits in the filter DF2.
Fra udtrykkene for y (tn), x2(tn) og y2(tn) fremgår det, at a) dersom Dopplerfrekvensen fd:s$0, kan signalerne elimineres i det første filter DPI, idet rlFrom the expressions of y (tn), x2 (tn) and y2 (tn), it appears that a) if the Doppler frequency fd: s $ 0, the signals can be eliminated in the first filter DPI, with r1
Cn(fd) L^(n) kan gøres = 0, i=0 b) dersom Dopplerfrekvensen fd f 0, og denne frekvens for det dominerende glitterspektrum er blevet korrekt bedømt i blokken HK, dvs. fmitofd, bliver indgangssignalet til filteret DF2 x2 (tn) = e·^2^" + n^T, hvad der repræsenterer et lavfrek vent signal, og som kan bortfiltreres i filteret DF2 på samme måde som signalet e^2<TTfdtn klev bortfiltreret i filteret DF1 for fd^io.Cn (fd) L ^ (n) can be done = 0, i = 0 b) if the Doppler frequency fd f 0, and this frequency for the dominant glitter spectrum has been correctly evaluated in the block HK, ie. fmitofd, the input signal to the filter DF2 is x2 (tn) = e · ^ 2 ^ "+ n ^ T, which represents a low frequency vent signal and which can be filtered out in filter DF2 in the same way as signal e ^ 2 <TTfdtn in the filter DF1 for fd ^ io.
For beregning af filterkoefficienterne Li(n) og (n) stilles følgende krav til udgangssignalerne y-^tn) og y2 (tn) : y^(tn)«0, når indgangssignalerne udgøres af jordglitter, dvs.For the calculation of the filter coefficients Li (n) and (n), the following requirements are set for the output signals y- ^ tn) and y2 (tn): y ^ (tn) «0 when the input signals are constituted by ground glitter, ie.
fD«0. I særdeleshed kræves, at y^(tn) = O for = /\fK,fD '0th In particular, y = (tn) = 0 for = / \ fK,
K = 1, . ...r.. A£„ vælges inden for jordglitterspektrets frekvens-J- KK = 1,. ... r .. A £ „is selected within the frequency glitter spectrum J-K
område. Dette krav kan opfyldes ved at vælge (/yf|^) = 0, K = 1, ···· r^» ^ = 1/ ..... N, dvs. at filteret 1 eliminerer jordglitteret. Ligningenterritory. This requirement can be met by choosing (/ yf | ^) = 0, K = 1, ···· r ^ »^ = 1 / ..... N, ie. that the filter 1 eliminates the ground glitter. The equation
Cn(/\f ) = o fører til følgende ligningssystem for beregningen af filterkoefficienterne L^in) : f1 L. (n) e' j21rAfK <nT ' ‘n-i’ = 0 i=0 1Cn (/ \ f) = o leads to the following equation system for calculating the filter coefficients L ^ in): f1 L. (n) e 'j21rAfK <nT' 'n-i' = 0 i = 0 1
Dersom AfK vælges symmetrisk omkring frekvensen 0, fører ovenstående sammenhænge til reelle, tidsafhængige koefficienter L^(n).If AfK is chosen symmetrically around the frequency 0, the above relationships result in real time-dependent coefficients L ^ (n).
147428 ίο y2(t )^5-0, når indgangssignalerne udgøres af bevægeligt glitter med frekvensen f^. Glitterets middelfrekvens er målt som værende f »f.. I særdeleshed kræves det, at y. (t ) =0 for FD - Fm = <£fK, K = 1, ^fR vælges således, at f + Λ f K falder inden for det bevægelige glitters frekvensområde. Dette fører til ligningssystemet: £ Cn-l(V Κ1 e -i2*· fK-l-T·· °, K. 1, ... r2 λ κ-|(η) - 1=0 Cn-1 <fm> til bestemmelse af filterkoefficienterne K^(n) .147428 ίο y2 (t) ^ 5-0 when the input signals are constituted by moving glitter with the frequency f ^. The average frequency of the glitter is measured as being f. F. In particular, y. (t) = 0 for FD - Fm = <£ fK, K = 1, ^ fR is chosen such that f + Λ f K falls within the frequency range of the moving glitter. This leads to the equation system: £ Cn-l (V Κ1 e -i2 * · fK-lT ·· °, K. 1, ... r2 λ κ- | (η) - 1 = 0 Cn-1 <fm> to determining the filter coefficients K ^ (n).
Fig. 5 er et blokdiagram over et udførelseseksempel på den i fig. 3 viste blok HK med henblik på at tilvejebringe hastighedskompensering sammen med de to filtre DF1 og DF2. Af udtrykket for x2(tn) ifølge ovenstående fremgår det, at kompenseringen signalmæssigt udføres ved at dividere udgangssignalerne y^(tn) fra filteret DF1 med faktoren Cn (fm) e^Tr N+n) hVor fm repræsente rer en skønnet værdi af middelfrekvensen for det bevægelige glitters dominerende spektrum sD. Fra filteret DF1 fås et signal Re i* 1 (tn)} over I-kanalen og et signal Im j?! (tn) } over Q-kanalen. Til hver kanal I og Q er sluttet en fasemålekreds FK til at måle faseforskellen /\<£) mellem to på hinanden følgende, filtrerede eksempleringer. Dette udføres på kendt måde ved først at opmåle eksempleringsværdiens to komposanter i I- og Q-kanalen, hvorved der opnås en værdi for fasevinkelen $1 i forhold til en bestemt reference. Derpå måles på samme måde fasevinkelen Q2 for den næste eksempleringsværdi, og forskellen /\Q = § 2 -$1 dannes. For hver "staggering"-sekvens tn fås eksempleringsværdier, der giver en sekvens af faseforskelle /\Q n mellem to på hinanden følgende og filtrerede eksempleringer y^(tn). Denne sekvens /\Qn tilføres en akkumulator S for de modtagne faseforskelsværdier /\Q under tidsrummet NT, svarende til en fuldstændig "staggering"-sekvens. Akkumulatoren S kan være af i sig selv kendt art, og kan f.eks. bestå af en tilbagekoblet summator.FIG. 5 is a block diagram of an embodiment of the embodiment shown in FIG. 3 to provide velocity compensation together with the two filters DF1 and DF2. From the expression for x2 (tn) according to the above, it can be seen that the compensation is performed signalally by dividing the output signals y ^ (tn) from the filter DF1 by the factor Cn (fm) e ^ Tr N + n) where fm represents an estimated value of the average frequency for the dominant spectrum of the moving glitter sD. From the filter DF1, a signal Re i * 1 (tn)} is obtained over the I-channel and a signal Im j ?! (tn)} over the Q channel. To each channel I and Q is connected a phase measurement circuit FK to measure the phase difference between two consecutive filtered samples. This is done in known manner by first measuring the two components of the sample value in the I and Q channels, thereby obtaining a value for the phase angle $ 1 relative to a particular reference. Then the phase angle Q2 is then measured for the next sample value and the difference / \ Q = § 2 - $ 1 is formed. For each "staggering" sequence tn, sample values are obtained giving a sequence of phase differences / \ Q n between two consecutive and filtered samples y ^ (tn). This sequence / \ Qn is added to an accumulator S for the received phase difference values / \ Q during the time interval NT, corresponding to a complete "staggering" sequence. The accumulator S may be of a kind known per se, and may e.g. consist of a feedback summator.
Ml og M2 er to lagerenheder, f.eks. i form af programmérbare læselagre ("PROM"). Lageret Ml udgøres af en matrix, hvori værdierne for koefficienterne Cn (fm) er noteret for forskellige værdier af faseforskellen /\Q og for de forskellige værdier af tn i "stagger"--sekvensen. For hvert værdi-par tn, /\Q fås således en bestemt værdi 147428 11 af koefficienterne Cn(fn), eftersom fm beregnes ud fra værdien på /\§) = 2TTT*fm, hvor T er kendt.M1 and M2 are two storage units, e.g. in the form of programmable reading stores ("PROM"). The memory M1 is a matrix in which the values of the coefficients Cn (fm) are noted for different values of the phase difference / \ Q and for the different values of tn in the "stagger" sequence. Thus, for each value pair tn, / \ Q, a specific value is obtained by the coefficients Cn (fn), since fm is calculated from the value of / \ §) = 2TTT * fm, where T is known.
Lagerenheden M2 udgøres af en matrix i form af et programmér-bart læselager, hvori sinus- og cosinus-værdierne for forskellige vinkler $ er listeført, hvilke vinkler fås fra akkumulatoren: S. Lagerenheden Ml har kun en enkelt udgang, hvor værdien l/|cn(fm)| optræder, men to indgange, hvor indgangsværdien /\Q henholdsvis klok-impulserne cl optræder, sidstnævnte ved eksempieringstidspunkterne tn inden for hvert interval C) NT. Lagerenheden M2 har en I- og en Q-udgang, hvori værdierne -sinP henholdsvis cos$ optræder, idet P er den akkumulerede fase. En multiplikator MU er tilsluttet de to udgange i lagerenheden M2 og udgangen i enheden Ml med henblik på at multiplicere faktoren 1/ |Cn(fm)| med sinus- henholdsvis cosinus-værdierne af den akkumulerede fase $. Således fremkommer i I- og Q-udgangene i multiplikatoren MU de to komposanterThe storage unit M2 is a matrix in the form of a programmable readable storage, in which the sine and cosine values of different angles $ are listed, which angles are obtained from the accumulator: S. The storage unit M1 has only a single output where the value l / | cn (sc) | occurs but two inputs where the input value / \ Q and clock pulses cl, respectively, the latter at the sample times tn within each interval C) NT. The storage unit M2 has an I and a Q output in which the values -sinP and cos $, respectively, occur, with P being the accumulated phase. A multiplier MU is connected to the two outputs of the storage unit M2 and the output of the unit M1 in order to multiply the factor 1 / | Cn (fm) | with the sine and cosine values of the accumulated phase $, respectively. Thus, in the I and Q outputs of the multiplier MU, the two components appear
-sin^ cos P-sin ^ cos P
- og - , jCn(fm) | j Cn(fm) j som behøves for at danne den komplekse værdi 1 _ -i 2Tf fm( ^NT+tn)- and -, jCn (fm) | j Cn (fm) j needed to form the complex value 1 _ -i 2Tf fm (^ NT + tn)
i i- Qi i- Q
Cn (fm)Cn (fm)
Ifølge signalbeskrivelsen ovenfor gælder det, at faktorenAccording to the signal description above, the factor
Cn(fd) . e li2,lfd(0NT + tn) skal multipliceres med faktoren 1 _ -j2'ff fm( Ό NT + tn)Cn (fd). e li2, lfd (0NT + tn) must be multiplied by the factor 1 _ -j2'ff fm (Ό NT + tn)
Cn (f m) med henblik på hastighedskompenseringen. Den komplekse multiplikator MK, der er tilsluttet udgangen i filteret DPI, og multiplikatoren MU udfører påi sig selv kendt måde denne komplekse multiplikation, eftersom de komplekse faktorers I- og Q-komposanter er tilgængelige som signalværdier over de respektive kanaler. Over I- og Q-udgangen fra multiplikatoren MK fås således modsvarende signalkomposanter i 147428 12 X (tn) - j2 Tr (fd-fm) ( \) NT+tn) , 2 " Cn(fm> ’ 8 jfr. det ovenstående.Cn (f m) for the speed compensation. The complex multiplier MK connected to the output of the filter DPI and the multiplier MU perform in this known manner in this complex multiplication, since the I and Q components of the complex factors are available as signal values over the respective channels. Thus, over the I and Q outputs of the multiplier MK, corresponding signal components are obtained in 14 X (tn) - j2 Tr (fd-fm) (\) NT + tn), 2 "Cn (fm> '8 cf. the above.
Filteret DF2 udgøres af et digitalt transversalfilter DF4, der er opbygget som vist i fig. 4. For at tilvejebringe en god glitter-undertrykkelse inden for hele hastighedsområdet er det i almindelighed nødvendigt at vælge forskellige filterkoefficienter K^(n) for forskellige opmålte frekvenser fm. Filterkoefficienterne K^(n) bliver derved bestemt af den ovenfor viste relation.The filter DF2 is constituted by a digital transversal filter DF4, which is constructed as shown in FIG. 4. In order to provide a good glitter suppression in the entire velocity range, it is generally necessary to select different filter coefficients K ^ (n) for different measured frequencies fm. The filter coefficients K 1 (n) are thereby determined by the relation shown above.
Til de i filteret DF4 indgående multiplikatorer er der derfor sluttet et lager MF, f.eks. i form af et programmérbart læselager, hvori koefficienterne K-^ (n) for hver værdi af /\φ og for hvert tidspunkt er fast indskrevet i matrix-form. Lageret MF er derfor med begge sine styreindgange forbundet, dels med udgangen i fasemålekredsen FK, hvori værdien af /\Q optræder, dels med den ikke viste klokimpuls-generator, som frembringer klokimpulserne cl i takt med "stagger"--sekvensen tn (inden for hvert interval NT). De af og tn afhængige værdier af koefficienterne (n) afgives til de i filteret DF4 indgående multiplikatorer, og over filterets udgange fremkommer kvadraturkomposanterne af det ønskede filtrerede signal y2(tn).Therefore, a stock MF is connected to the multipliers included in the filter DF4, e.g. in the form of a programmable reading layer in which the coefficients K- ^ (n) for each value of / \ φ and for each time point are fixedly inscribed in matrix form. Therefore, the memory MF is connected to both of its control inputs, partly with the output of the phase measurement circuit FK, in which the value of / \ Q occurs, and partly with the clock pulse generator not shown, which produces the clock pulses cl in step with the "stagger" sequence tn (before for each interval NT). The dependent and tn dependent values of the coefficients (n) are given to the multipliers included in the filter DF4 and above the outputs of the filter the quadrature components of the desired filtered signal y2 (tn) appear.
Filtrene DF1 og DF2 er som nævnt ovenfor opbygget således som vist i fig. 4. Denne figur viser imidlertid kun opbygningen for en enkelt kanal, f.eks. X-kanalen, og multiplikatorerne MU0-MU2 multiplicerer signalkomposanterne af (tn) og *2(tn) i denne kanal med modsvarende komposanter i koefficienterne (n) og (n). Modsvarende filterkredse findes i den anden kanal, nemlig i Q-kanalen, og ved den komplekse multiplikation i multiplikatorerne MU0-MU2 blandes værdierne i I- og Q-kanalerne. Filtrene DF1 og DF2 har for en given orden - der er bestemt af antallet af forsinkelseskredse DL1-DL3 - et bestemt pasbånd,, hvis bredde på kendt måde kan udvides ved at vælge et filter af højere orden.The filters DF1 and DF2, as mentioned above, are constructed as shown in FIG. However, this figure shows only the structure of a single channel, e.g. The X channel, and the multipliers MU0-MU2 multiply the signal components of (tn) and * 2 (tn) in this channel with corresponding components in the coefficients (n) and (n). Corresponding filter circuits exist in the second channel, namely in the Q channel, and for the complex multiplication in the multipliers MU0-MU2, the values in the I and Q channels are mixed. The filters DF1 and DF2 have for a given order - which is determined by the number of delay circuits DL1-DL3 - a specific pass band, the width of which can be expanded in a known way by selecting a higher order filter.
Claims (2)
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
SE7709119 | 1977-08-12 | ||
SE7709119A SE409148B (en) | 1977-08-12 | 1977-08-12 | FILTER DEVICE INCLUDED IN THE RECEIVER OF A PULSE DOPPER RADAR |
Publications (3)
Publication Number | Publication Date |
---|---|
DK356178A DK356178A (en) | 1979-02-13 |
DK147428B true DK147428B (en) | 1984-07-30 |
DK147428C DK147428C (en) | 1985-02-11 |
Family
ID=20331986
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
DK356178A DK147428C (en) | 1977-08-12 | 1978-08-11 | FILTER EQUIPMENT TO REMOVE UNSUSED ECHO SIGNALS IN AN IMPULS-Doppler RADAR RECEIVER |
Country Status (10)
Country | Link |
---|---|
CH (1) | CH632850A5 (en) |
DE (1) | DE2833050C2 (en) |
DK (1) | DK147428C (en) |
FI (1) | FI65863C (en) |
FR (1) | FR2400212A1 (en) |
GB (1) | GB2002617B (en) |
IT (1) | IT1098359B (en) |
NL (1) | NL187872C (en) |
NO (1) | NO145964C (en) |
SE (1) | SE409148B (en) |
Families Citing this family (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
FR2632420B1 (en) * | 1987-10-16 | 1990-10-12 | Thomson Csf | METHOD AND DEVICE FOR COMPENSATING THE SPEED OF CLOUD IN A COHERENT DOPPLER RADAR AT AMBIGUOUS VARIABLE SPEED |
DE19654769A1 (en) * | 1996-12-30 | 1998-07-02 | Teves Gmbh Alfred | Method and device for vehicle control or regulation |
US6049302A (en) * | 1999-05-04 | 2000-04-11 | Boeing North American | Pulsed doppler radar system with small intermediate frequency filters |
GB0301614D0 (en) * | 2003-01-24 | 2013-05-08 | Qinetiq Ltd | Target visibility enhancement system |
Family Cites Families (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3742500A (en) * | 1970-08-24 | 1973-06-26 | Raytheon Co | Mti radar |
DE2212272A1 (en) * | 1971-03-17 | 1972-09-28 | Int Standard Electric Corp | Impulse Doppler radar device |
US3962704A (en) * | 1974-05-31 | 1976-06-08 | Hughes Aircraft Company | Moving target indicator clutter tracker |
US4035799A (en) * | 1975-11-04 | 1977-07-12 | The United States Of America As Represented By The Secretary Of The Navy | Digital mean clutter doppler compensation system |
-
1977
- 1977-08-12 SE SE7709119A patent/SE409148B/en not_active IP Right Cessation
-
1978
- 1978-07-24 GB GB7830900A patent/GB2002617B/en not_active Expired
- 1978-07-26 FR FR7822113A patent/FR2400212A1/en active Granted
- 1978-07-27 DE DE2833050A patent/DE2833050C2/en not_active Expired
- 1978-07-28 FI FI782352A patent/FI65863C/en not_active IP Right Cessation
- 1978-08-02 NL NLAANVRAGE7808152,A patent/NL187872C/en not_active IP Right Cessation
- 1978-08-10 IT IT26665/78A patent/IT1098359B/en active
- 1978-08-11 CH CH855578A patent/CH632850A5/en not_active IP Right Cessation
- 1978-08-11 NO NO782748A patent/NO145964C/en unknown
- 1978-08-11 DK DK356178A patent/DK147428C/en not_active IP Right Cessation
Also Published As
Publication number | Publication date |
---|---|
FR2400212A1 (en) | 1979-03-09 |
CH632850A5 (en) | 1982-10-29 |
NO782748L (en) | 1979-02-13 |
FR2400212B1 (en) | 1984-12-28 |
DK147428C (en) | 1985-02-11 |
NO145964B (en) | 1982-03-22 |
IT1098359B (en) | 1985-09-07 |
DE2833050A1 (en) | 1979-03-01 |
NO145964C (en) | 1982-06-30 |
FI782352A (en) | 1979-02-13 |
NL187872B (en) | 1991-09-02 |
SE7709119L (en) | 1979-02-13 |
SE409148B (en) | 1979-07-30 |
IT7826665A0 (en) | 1978-08-10 |
GB2002617B (en) | 1982-02-10 |
GB2002617A (en) | 1979-02-21 |
NL187872C (en) | 1992-02-03 |
DK356178A (en) | 1979-02-13 |
DE2833050C2 (en) | 1986-08-07 |
FI65863C (en) | 1984-07-10 |
FI65863B (en) | 1984-03-30 |
NL7808152A (en) | 1979-02-14 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
CA2949147C (en) | Radar operation with increased doppler capability | |
DE69507954T2 (en) | METHOD AND DEVICE FOR REDUCING THE DISTANCE DOPPLER AMBIGUITIES IN PULSE DOPPLER RADAR DEVICES | |
JP4988332B2 (en) | Techniques for incoherent integration of targets with vague velocities. | |
US7109916B2 (en) | Device for, in particular bistatic radar applications | |
DE3038961C2 (en) | ||
US5309160A (en) | Radar system and method having variable tracking range | |
DE10104022A1 (en) | Radar device and method for coding a radar device | |
US3707718A (en) | Radar system | |
US4011438A (en) | Simplified digital moving target indicator filter | |
DE112017007143T5 (en) | RADAR DEVICE | |
DE3030515A1 (en) | RADAR DEVICE FOR DETECTING AND LOCATING VEHICLES | |
Kachelmyer | Range-Doppler imaging with laser radar | |
EP0362631B1 (en) | Doppler flow rate meter | |
WO2021013680A2 (en) | Radar device | |
DK147428B (en) | FILTER EQUIPMENT TO REMOVE UNSUSED ECHO SIGNALS IN AN IMPULS-Doppler RADAR RECEIVER | |
RU2354992C2 (en) | Method and system of regenerating signals in form of recurrent pulses with fast tuneable frequency using deconvolution method and their application | |
US4982165A (en) | Set-on oscillator | |
Bandiera et al. | CRLB for I/Q imbalance estimation in FMCW radar receivers | |
Singh et al. | A comparative study of multiple radar waveform design techniques | |
Nhan et al. | A Mathematical Model for Determining the Type of Signal Modulation in a Digital Receiver with Autocorrelation Processing | |
WO2020259903A1 (en) | Ofdm radar sensor system having actively returning repeater | |
RU2262122C1 (en) | System for dection of radar signals | |
Harmuth | Radio Communication with orthogonal time functions | |
Keel | Adaptive clutter rejection filters for airborne Doppler weather radar applied to the detection of low altitude windshear | |
RU2054693C1 (en) | Moving target discrimination method |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
PUP | Patent expired |