WO2019179487A1 - 一种平衡式射频功率放大器、芯片及通信终端 - Google Patents

一种平衡式射频功率放大器、芯片及通信终端 Download PDF

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Publication number
WO2019179487A1
WO2019179487A1 PCT/CN2019/079012 CN2019079012W WO2019179487A1 WO 2019179487 A1 WO2019179487 A1 WO 2019179487A1 CN 2019079012 W CN2019079012 W CN 2019079012W WO 2019179487 A1 WO2019179487 A1 WO 2019179487A1
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Prior art keywords
unit
power
radio frequency
degree
adjustable
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PCT/CN2019/079012
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English (en)
French (fr)
Inventor
陈岗
白云芳
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上海唯捷创芯电子技术有限公司
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Priority to EP19772068.3A priority Critical patent/EP3771093A4/en
Publication of WO2019179487A1 publication Critical patent/WO2019179487A1/zh
Priority to US17/027,708 priority patent/US11863134B2/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H03F3/193—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only with field-effect devices
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/56—Modifications of input or output impedances, not otherwise provided for
    • H03F1/565—Modifications of input or output impedances, not otherwise provided for using inductive elements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/211—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/213—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only in integrated circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H03F3/245—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/60—Amplifiers in which coupling networks have distributed constants, e.g. with waveguide resonators
    • H03F3/602—Combinations of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/18—Indexing scheme relating to amplifiers the bias of the gate of a FET being controlled by a control signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/387—A circuit being added at the output of an amplifier to adapt the output impedance of the amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/391—Indexing scheme relating to amplifiers the output circuit of an amplifying stage comprising an LC-network
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/451—Indexing scheme relating to amplifiers the amplifier being a radio frequency amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2203/00—Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/20—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F2203/21—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F2203/211—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H03F2203/21139—An impedance adaptation circuit being added at the output of a power amplifier stage

Definitions

  • the present invention relates to a radio frequency power amplifier, and more particularly to a balanced radio frequency power amplifier supporting a mobile high power terminal function, and also relates to an integrated circuit chip including the balanced radio frequency power amplifier and a corresponding communication terminal, belonging to a radio frequency integrated circuit Technical field.
  • RF power amplifiers are widely used in wireless communication devices such as mobile phones.
  • various communication terminal devices must improve technical specifications to meet such demands.
  • HPUE High Performance User Equipment
  • the mobile terminal antenna needs to increase the linear power to 26dBm. Due to the reduced design area of the RF antenna of the communication terminal, the RF antenna gain is reduced, and the RF antenna load impedance standing wave ratio (VSWR) becomes larger, which requires the RF power amplifier to output a larger linear power to meet the radio frequency index of the communication terminal.
  • VSWR RF antenna load impedance standing wave ratio
  • the method for increasing the linear power of the radio frequency power amplifier of the communication terminal mainly includes increasing the output voltage and increasing the output current; increasing the output voltage increases the power supply voltage of the output port of the radio frequency power amplifier, so that the radio frequency The peak-to-peak value of the output voltage waveform of the power amplifier is increased to increase the linear power of the RF power amplifier; increasing the output current increases the output current capability of the RF power amplifier at the maximum power.
  • Increasing the power supply voltage of the RF power amplifier is mainly divided into two methods: constant increase and instantaneous increase.
  • constant increase is that the implementation is simple, and the representative technology is APT (Average Power Tracking), but has the following disadvantages: 1) higher requirements for the ruggedness of the power tube; 2) additional increase in communication terminals is required.
  • the power booster boosts the power supply voltage of the RF power amplifier; 3) reduces the power added efficiency at the RF power amplifier's back-off power.
  • the advantage of instantaneous increase is to increase the power added efficiency of the RF power amplifier back-off power.
  • the representative technology is ET (Envelop Tracking), but it also has the following disadvantages: 1) Higher requirements for the ruggedness of the power tube 2) The implementation is complicated, and the baseband chip of the communication terminal needs to work together with the RF front end; 3) It is very technically difficult to process the larger bandwidth RF signal.
  • Increasing the output current of the RF power amplifier is mainly divided into two ways: increasing the output power tube area and connecting multiple power amplifiers in parallel.
  • the advantage of increasing the output power tube area is that the implementation is simple, but the parasitic parameters are too large.
  • the Band41 frequency band currently supporting the HPUE function belongs to a higher frequency band, and the larger parasitic parameters deteriorate the performance of the high-band RF power amplifier. , its linear power boost will also be limited.
  • the multi-channel power amplifier is connected in parallel with respect to increasing the power tube area, avoiding the problem of excessive parasitic parameters, but cannot solve the RF power amplifier output power loss caused by the antenna load impedance variation of the communication terminal.
  • the primary technical problem to be solved by the present invention is to provide a balanced RF power amplifier.
  • Another technical problem to be solved by the present invention is to provide an integrated circuit chip including the balanced radio frequency power amplifier and a corresponding communication terminal.
  • a balanced RF power amplifier including a control unit, a first driver stage unit, a 90 degree power splitter unit, a first power stage unit, a second power level unit, and an adjustable a 90 degree power combiner unit, the control unit being respectively coupled to the first drive stage unit, the first power stage unit, the second power stage unit, and the adjustable 90 degree power combiner unit, the An input end of a driver stage unit is connected to the RF signal input end, an output end of the first driver stage unit is connected to an input end of the 90 degree power splitter unit, and an output end of the 90 degree power splitter unit is respectively Connected to the input ends of the first power stage unit and the second power stage unit, the output ends of the first power stage unit and the second power stage unit are respectively coupled to the adjustable 90 degree power combiner
  • the input end of the unit is connected, and the output end of the adjustable 90-degree power combiner unit is connected to the radio frequency transmitting path;
  • the RF input signal is amplified by the first driving stage unit and input to the 90-degree power divider unit, and the 90-degree power divider unit divides the RF input signal into two equal-amplitude RFs with a phase difference of 90 degrees.
  • the input signal is input to the first power stage unit and the second power stage unit for amplification, and then input to the adjustable 90 degree power combiner unit, and the adjustable 90 is controlled by the control unit.
  • the power combiner unit combines two RF input signals into one RF input signal into the RF transmission path when the phase difference and amplitude difference of the two equal-amplitude RF input signals are different at different frequencies.
  • the second driver stage unit and the third driver stage unit are replaced by the first driver stage unit, and the second driver stage unit and the third driver stage unit are disposed at the 90 degree power division. Between the first power stage unit and the second power stage unit, an input end of the 90 degree power splitter unit is connected to the radio frequency signal input end.
  • a switch module unit is disposed between the adjustable 90-degree power combiner unit and the radio frequency transmission path, and the input end of the switch module unit and the adjustable 90-degree power combiner unit The output ends of the switch module unit are respectively connected to the radio frequency transmission path and the radio frequency receiving path;
  • the control unit is connected to the switch module unit, and is configured to control a switch state of the switch module unit according to a frequency band requirement, and pass the RF input signal output by the adjustable 90-degree power combiner unit to a frequency band corresponding to the frequency band.
  • the RF transmission path is input to the next stage circuit.
  • any one of a Band7 duplexer, a Band38 filter, a Band40 filter, and a Band41 filter is disposed.
  • the adjustable 90 degree power combiner unit comprises a 90 degree phase shifter and a Wilkinson power combiner
  • the 90 degree phase shifter comprises a phase lag impedance transform network and a phase lead impedance transform network
  • An input of the phase-lag impedance conversion network is coupled to an output of the first power stage unit, and an output of the phase-lag impedance conversion network is coupled to an input of the Wilkinson power combiner
  • An input of the lead impedance conversion network is coupled to an output of the second power stage unit, and an output of the phase lead impedance conversion network is coupled to another input of the Wilkinson power combiner.
  • the phase-lag impedance conversion network includes a first inductor, a first variable capacitor, and a second variable capacitor; one end of the first inductor serves as an input end of the phase-lag impedance conversion network, and is used for Connected to an output end of the first power stage unit; the other end of the first inductor is respectively connected to one end of the first variable capacitor and the second variable capacitor, the first variable capacitor The other end is grounded, and the other end of the second variable capacitor serves as an output of the phase-lag impedance conversion network for connection to an input of the Wilkinson power combiner.
  • the phase lead impedance conversion network includes a third variable capacitor and a second inductor; one end of the third variable capacitor is used as an input end of the phase lead impedance conversion network, and is used for An output end of the second power stage unit is connected; the other end of the third variable capacitor is used as an output end of the phase lead impedance conversion network, and is configured to respectively synthesize with one end of the second inductor and the Wilkinson power The other input of the device is connected, and the other end of the second inductor is grounded.
  • the Wilkinson power combiner includes a fourth variable capacitor, a variable resistor, a fifth variable capacitor, a third inductor, a fourth inductor, and a sixth variable capacitor;
  • One end of the capacitor is used as an input end of the Wilkinson power combiner for respectively connecting an output end of the phase-lag impedance conversion network, an output end of the variable resistor, and one end of the third inductor;
  • One end of the variable resistor serves as another input end of the Wilkinson power combiner for respectively outputting the output of the phase lead impedance conversion network, one end of the fifth variable capacitor, and the first One end of the fourth inductor is connected;
  • the other end of the third inductor and the fourth inductor serves as an output end of the adjustable 90-degree power combiner unit for respectively, and one end of the sixth variable capacitor, the switch
  • the input ends of the module unit are connected; the other ends of the fourth variable capacitor, the fifth variable capacitor and the sixth variable capacitor are respectively grounded.
  • each variable capacitor in the adjustable 90-degree power combiner unit is composed of a capacitor and an n-way switched capacitor group in parallel; or each of the adjustable 90-degree power combiner units
  • the variable capacitor is composed of n pairs of the switched capacitor groups connected in parallel, and the n is a positive integer.
  • each set of switched capacitors is composed of a capacitor and a switch connected in series, and switches in each group of the switched capacitors are respectively connected to the control unit, and the control is The unit closes or opens a specific number of switches in the switched capacitor group, and obtains the capacitance value of the corresponding switched capacitor.
  • each variable capacitor in the adjustable 90-degree power combiner unit is composed of a capacitor and an n-way switched capacitor group in series; or each of the adjustable 90-degree power combiner units is variable
  • the capacitor consists of a series of n-switched capacitors connected in series.
  • each set of switched capacitors is composed of a capacitor and a switch in parallel, and switches in each set of the switched capacitors are respectively connected to the control unit, and the control is The unit closes or opens a specific number of switches in the switched capacitor group, and obtains the capacitance value of the corresponding switched capacitor.
  • each variable resistor in the adjustable 90-degree power combiner unit is composed of a resistor and an n-way switch resistor group in parallel; or each of the adjustable 90-degree power combiner units is variable
  • the resistor consists of a parallel combination of n-switch resistors.
  • each group of switch resistors is composed of a resistor and a switch in series, and switches in each group of the switch resistors are respectively connected to the control unit, and the control unit is closed. Or open a specific number of switches in the switch resistor group to obtain the resistance value of the corresponding switch resistor.
  • each variable resistor in the adjustable 90-degree power combiner unit is composed of a resistor and an n-way switch resistor group in series; or each of the adjustable 90-degree power combiner units is variable
  • the resistor consists of a series of n-switch resistors connected in series.
  • each group of switch resistors is composed of a resistor and a switch in parallel, and switches in each group of the switch resistors are respectively connected to the control unit, and the control unit is closed. Or open a specific number of switches in the switch resistor group to obtain the resistance value of the corresponding switch resistor.
  • the first matching network is configured in the 90-degree power splitter unit for impedance matching, and participates in the 90-degree power splitter unit to divide the RF input signal into two phases with a phase difference of 90 degrees.
  • the road is equal to the RF input signal.
  • the adjustable 90-degree power combiner unit is provided with a second matching network for implementing impedance matching, and participates in matching the adjustable 90-degree power combiner unit to make two equal-amplitude RF input signals.
  • the phase difference and amplitude difference at different frequencies are minimal.
  • the 90 degree power combiner unit is replaced by a non-adjustable structure; the 90 degree power combiner of the non-adjustable structure is a matching network built by the inductor, the capacitor and the resistor device. And a phase shifting network; or the 90 degree power combiner of the non-adjustable structure is an impedance and phase transform network built by the metal coupling device; or the 90 degree power combiner of the non-adjustable structure is an impedance and phase transform constructed for the transmission line network The internet.
  • the switch module unit includes n sets of transceiver switches, and the common ends of each set of the transceiver switches are respectively connected to the output ends of the adjustable 90-degree power combiner unit, and each group of the transceivers One output of the switch is respectively connected to a corresponding RF transmission path, and the other output of each group of transceiver switches is respectively connected to a corresponding RF receiving path.
  • an integrated circuit chip including the balanced RF power amplifier described above is provided.
  • a communication terminal including the above-described balanced radio frequency power amplifier.
  • the balanced RF power amplifier provided by the invention divides the RF input signal into two equal-amplitude signals with a phase difference of 90 degrees through a 90-degree splitter unit, and amplifies the two RF input signals and inputs them to the adjustable 90
  • the power combiner controls the adjustable capacitor and the adjustable resistor in the adjustable 90-degree power combiner through the control unit, so that the two RF input signals are combined into one RF when the phase difference and amplitude difference at different frequencies are the smallest.
  • the input signal is input to the next stage circuit through a specific RF transmission path. Therefore, the balanced RF power amplifier not only improves the maximum linear power of the output, but also reduces the sensitivity to changes in the RF antenna load, thereby enabling support for the Mobile High Power Terminal (HPUE) function.
  • HPUE Mobile High Power Terminal
  • FIG. 1 is a schematic structural view 1 of a balanced radio frequency power amplifier according to Embodiment 1 of the present invention.
  • FIG. 2 is a schematic structural view 2 of a balanced radio frequency power amplifier according to Embodiment 1 of the present invention
  • FIG. 3 is a schematic structural view 3 of a balanced radio frequency power amplifier according to Embodiment 1 of the present invention.
  • FIG. 4 is a schematic structural view 1 of a balanced radio frequency power amplifier according to Embodiment 2 of the present invention.
  • FIG. 5 is a schematic structural view 2 of a balanced radio frequency power amplifier according to Embodiment 2 of the present invention.
  • FIG. 6 is a schematic structural view 3 of a balanced radio frequency power amplifier according to Embodiment 2 of the present invention.
  • FIG. 7 is a circuit schematic diagram of a 90-degree adjustable power combiner in a balanced RF power amplifier according to the present invention.
  • FIG. 8 is a schematic structural diagram of a 90-degree power combiner unit with a non-adjustable structure instead of an adjustable 90-degree power combiner unit in the balanced RF power amplifier provided by the present invention
  • 9A is a circuit schematic diagram of each variable capacitor in a tunable 90-degree power combiner in a balanced RF power amplifier according to the present invention.
  • 9B is another circuit schematic diagram of each variable capacitor in the adjustable 90-degree power combiner in the balanced RF power amplifier provided by the present invention.
  • 10A is a circuit schematic diagram of each variable resistor in a tunable 90-degree power combiner in a balanced RF power amplifier according to the present invention
  • 10B is another circuit schematic diagram of each variable resistor in the adjustable 90-degree power combiner in the balanced RF power amplifier provided by the present invention.
  • FIG. 11 is a balanced RF power amplifier according to the present invention having a load impedance standing wave ratio (VSWR) of 3:1, and the phase difference of the two RF input signals in the adjustable 90 degree power combiner is 80 degrees and 90 degrees, respectively.
  • a balanced RF power amplifier according to the present invention having a load impedance standing wave ratio (VSWR) of 3:1, and the amplitude difference of the two RF input signals in the adjustable 90 degree power combiner is -1 dBc, 0 dBc, respectively. , in the case of +1dBc, a plot of the output power (Pout) as a function of load phase;
  • VSWR load impedance standing wave ratio
  • 13 is a design result of an adjustable 90-degree power combiner with a center band of 2.5 GHz in a balanced RF power amplifier according to the present invention
  • FIG. 15 is a schematic diagram of a balanced RF power amplifier according to the present invention, wherein the center frequency band is moved to Band 41 (2.496 GHz to 2.69 GHz) by changing the values of the variable capacitor and the variable resistor in the adjustable 90-degree power combiner. Design result
  • 16 is a schematic structural view of a conventional single-ended structure RF power amplifier
  • FIG. 17 is a schematic diagram showing a comparison of gain (Gain) of a conventional single-ended RF power amplifier and a balanced RF power amplifier provided by the present invention
  • FIG. 18 is a schematic diagram showing a comparison of output power (Pout) in a conventional single-ended structure RF power amplifier and a balanced RF power amplifier provided by the present invention with a load impedance standing wave ratio (VSWR) of 3:1;
  • Pout output power
  • VSWR load impedance standing wave ratio
  • FIG. 19 is a graph showing a comparison of output power (Pout) of a first power stage unit and a second power stage unit in a balanced RF power amplifier with a load impedance standing wave ratio (VSWR) of 3:1. schematic diagram.
  • the present invention provides a balanced RF power amplifier for supporting related indicators of a mobile high power terminal (HPUE), achieving the goal of increasing the maximum output linear power of the RF power amplifier and reducing the load sensitivity of the RF power amplifier.
  • HPUE mobile high power terminal
  • the balanced RF power amplifier provided in this embodiment includes a control unit 100, a first driver stage unit 110, a 90-degree power divider unit 160, a first power stage unit 120, and a second power stage unit 121.
  • the 90 degree power combiner unit 170 and the switch module unit 130 are adjustable.
  • the control unit 100 is respectively connected to the first driving stage unit 110, the first power level unit 120, the second power level unit 121, the adjustable 90-degree power combiner unit 170, and the switch module unit 130, and can be used for controlling the first A quiescent current of a driver stage unit 110, a first power stage unit 120, and a second power stage unit 121.
  • the input end of the first driver stage unit 110 is connected to the RF signal input end, the output end of the first driver stage unit 110 is connected to the input end of the 90 degree power splitter unit 160, and the output ends of the 90 degree power splitter unit 160 are respectively
  • the input ends of the first power stage unit 120 and the second power stage unit 121 are connected, and the outputs of the first power stage unit 120 and the second power stage unit 121 are respectively connected to the input ends of the adjustable 90-degree power combiner unit 170.
  • the output end of the adjustable 90-degree power combiner unit 170 is connected to the input end of the switch module unit 130.
  • the output ends of the switch module unit 130 are respectively connected to the radio frequency transmission path (A1 to An) and the radio frequency receiving path (B1 to Bm). Connection, where m and n are positive integers.
  • the RF input signal When an RF input signal enters the balanced RF power amplifier, the RF input signal is input to the first driver stage unit 110 through the RF signal input terminal for amplification, and the RF input signal amplified by the first driver stage unit 110 is input.
  • the 90-degree splitter unit 160 divides the RF input signal into two equal-amplitude RF input signals with a phase difference of 90 degrees (or close to 90 degrees, the same below), corresponding to the input.
  • the two power level units 120 and the second power stage unit 121 are amplified, and the two RF input signals amplified by the first power stage unit 120 and the second power stage unit 121 are correspondingly input to the adjustable 90 degree power combiner unit 170.
  • the adjustable 90-degree power combiner unit 170 is controlled by the control unit 100 such that the phase difference and the amplitude difference between the two RF input signals at different frequencies are minimum (preferably, the phase difference is 0 degrees or close to 0 degrees, and the amplitude difference is 0dBc or close to 0dBc), the two RF input signals are combined into one RF input signal and input to the switch module unit 130.
  • the control unit 100 controls the switch module according to the frequency band requirement. Switch state 130, the combined RF input signal by a particular radio frequency transmission path is inputted to a next circuit.
  • the RF input signal is divided into phase differences.
  • the design complexity of the RF power amplifier can be set in the 90-degree splitter unit 160, and the first matching network is used to implement the 90-degree splitter unit 160 and the first drive-level unit 110, respectively, and the first power. Impedance matching between the stage unit 120 and the second power stage unit 121.
  • the adjustable 90 degree power combiner unit 170 causes the two RF input signals to have a phase difference of 0 degrees or close to 0 degrees at different frequencies, an amplitude difference of 0 dBc or close to 0 dBc, and combine the two RF input signals into one RF input signal to maximize input.
  • a second matching network can also be set in the adjustable 90-degree power combiner unit 170, and the second matching network can be implemented.
  • the 90 degree power combiner unit 170 is matched with the impedance between the first power stage unit 120, the second power stage unit 121, and the switch module unit 130, respectively.
  • the switch module unit 130 includes n sets of transceiving switches (single pole double throw switches S1 to Sn, n being a positive integer).
  • the common ends of each set of transceiver switches are respectively connected to the output ends of the adjustable 90-degree power combiner unit 170, and one output end of each set of transceiver switches is respectively connected with a corresponding RF transmission path, and the other of each group of transceiver switches The output ends are respectively connected to corresponding RF receiving paths.
  • a Band7 duplexer can be disposed between the switch module unit 130 and the RF transmission path of the embodiment. Any one of a Band38 filter, a Band40 filter, and a Band41 filter.
  • a Band7 duplexer can be disposed between the first group of transceiver switches S1 and the RF transmission path, and a Band38 filter, a Band40 filter, and a Band 40 filter are disposed between the last group of transceiver switches Sn and the RF transmission path. Any of the Band41 filters.
  • the switch module unit 130 of the embodiment may be removed, and the two equal-amplitude RF input signals are combined into one way by the adjustable 90-degree power combiner unit 170.
  • the RF input signal is directly input to a specific RF transmission path and then input to the next stage circuit.
  • the balanced RF power amplifier provided in this embodiment is different from the balanced RF power amplifier provided in Embodiment 1 in that a 90-degree splitter unit 160 is disposed in the first drive stage unit. 110 is front, and the first driver stage unit 110 is changed to the second driver stage unit 111 and the third driver stage unit 112. Therefore, the input end of the 90-degree power splitter unit 160 is connected to the RF signal input end, and the output end of the 90-degree power splitter unit 160 is connected to the input end of the second drive stage unit 111 and the third drive stage unit 112, The outputs of the second driver stage unit 111 and the third driver stage unit 112 are connected to the input ends of the first power stage unit 120 and the second power stage unit 121.
  • the first part can be set in the 90-degree splitter unit.
  • a second matching network can be set in the adjustable 90 degree power combiner unit 170.
  • the RF input signal When an RF input signal enters the balanced RF power amplifier, the RF input signal is input to the 90-degree splitter unit 160 through the RF signal input terminal, and the 90-degree splitter unit 160 divides the RF input signal into phase differences.
  • the two-way equal-amplitude RF input signal of 90 degrees is correspondingly input to the second driving stage unit 111 and the third driving stage unit 112 for amplification, and the two driving steps of the second driving stage unit 111 and the third driving stage unit 112 are amplified.
  • the RF input signal is input to the first power stage unit 120 and the second power stage unit 121 for further amplification, and then input to the adjustable 90-degree power combiner unit 170, and the control unit 100 controls the adjustable 90 degrees.
  • the combiner unit 170 converts the phase difference of the two RF input signals at different frequencies to 0 degrees or close to 0 degrees, and when the amplitude difference is 0 dBc or close to 0 dBc, the two RF input signals are combined into one RF input signal input.
  • the control unit 100 controls the switch state of the switch module unit 130, and passes the combined RF input signal. Fixed RF transmit path input to the next-stage circuit.
  • Band7 duplex can be set between the switch module unit 130 and the RF transmission path of this embodiment. Any one of the device, the Band38 filter, the Band40 filter, and the Band41 filter.
  • a Band7 duplexer can be disposed between the first group of transceiver switches S1 and the radio frequency transmission path, and a Band38 filter, a Band40 filter, and a Band 40 filter are disposed between the last group of transceiver switches Sn and the RF transmission path. Any of the Band41 filters.
  • the switch module unit 130 of the embodiment may be removed, and the two equal-amplitude RF input signals are combined into one way by the adjustable 90-degree power combiner unit 170.
  • the RF input signal is directly input to a specific RF transmission path and then input to the next stage circuit.
  • first driving stage unit 110, the second driving stage unit 111, and the third driving stage unit 112 in Embodiment 1 and Embodiment 2 may be a single-level driving level unit or a two-level driving level unit.
  • the first power stage unit 120 and the second power stage unit 121 may be a single-stage power stage unit or a two-stage power stage unit.
  • the first power stage unit 120 and the second power stage unit 121 may be a heterojunction transistor (HBT) or a high electron mobility transistor fabricated on a gallium arsenide (GaAs) substrate or a germanium-silicon (SiGe) substrate.
  • HEMT high electron mobility transistor
  • p-HEMT high electron mobility transistor
  • BJT bipolar junction transistor
  • CMOS complementary metal-oxide-semiconductor transistor
  • the adjustable 90-degree power combiner unit 170 includes a 90-degree phase shifter 1 and a Wilkinson power combiner 2.
  • the 90-degree phase shifter 1 includes a phase-lag impedance conversion network 3 and a phase lead impedance conversion network 4, and an input end of the phase-lag impedance conversion network 3 is connected to an output end of the first power stage unit 120, and the phase-lag impedance conversion network 3 is connected.
  • the output is connected to an input of the Wilkinson power combiner 2; the input of the phase lead impedance conversion network 4 is connected to the output of the second power stage unit 121, and the output of the phase lead impedance conversion network 4 is connected to Will The other input of the Kinsen power combiner 2 is connected.
  • the two equal-amplitude RF input signals with the phase difference of 90 degrees amplified by the first power stage unit 120 and the second power stage unit 121 enter the phase-lag impedance conversion network 3 according to the phase relationship of the two equal-amplitude RF input signals respectively.
  • the phase leads the impedance conversion network 4, so that the phase difference between the two RF input signals at different frequencies is converted to 0 degrees, and the amplitude difference is close to (equal to or approximately equal to) 0 dBc, and then input to the Wilkinson power synthesizer 2, Will The Jinsen Power Synthesizer 2 synthesizes two RF input signals with a phase difference of 0 degrees and an amplitude difference close to (equal to or approximately equal to) 0 dBc into one RF input signal, and inputs it to the switch module unit 130, or directly inputs to a specific one. In the RF transmission path.
  • the phase-lag impedance conversion network 3 includes a first inductor 301, a first variable capacitor 302, and a second variable capacitor 309; wherein one end of the first inductor 301 serves as an input terminal of the phase-lag impedance conversion network 3
  • the other end of the first inductor 301 is connected to one end of the first variable capacitor 302 and the second variable capacitor 309, and the other end of the first variable capacitor 302 is connected to the output end of the first power level unit 120.
  • the other end of the second variable capacitor 309 serves as an output of the phase-lag impedance conversion network 3 for connection to an input of the Wilkinson power combiner 2.
  • the phase lead impedance conversion network 4 includes a third variable capacitor 303 and a second inductor 304; one end of the third variable capacitor 303 serves as an input of the phase lead impedance conversion network 4 for output with the second power stage unit 121.
  • the other end of the third variable capacitor 303 serves as an output of the phase lead impedance conversion network 4 for respectively connecting one end of the second inductor 304 and the other input of the Wilkinson power combiner 2; the second inductor The other end of 304 is grounded.
  • the Wilkinson power combiner 2 includes a fourth variable capacitor 310, a variable resistor 308, a fifth variable capacitor 311, a third inductor 305, a fourth inductor 306, and a sixth variable capacitor 307; wherein the fourth variable One end of the capacitor 310 is used as an input terminal of the Wilkinson power combiner 2 for respectively connecting the output of the phase-lag impedance conversion network 3, the other end of the variable resistor 308, and one end of the third inductor 305; One end of the 308 is used as the other input terminal of the Wilkinson power combiner 2, and is respectively connected to the output end of the phase lead impedance conversion network 4, one end of the fifth variable capacitor 311, and one end of the fourth inductor 306; The other ends of the inductor 305 and the fourth inductor 306 are used as output ends of the adjustable 90-degree power combiner unit 170 for respectively connecting one end of the sixth variable capacitor 307 and the input end of the switch module unit 130; The other ends of the variable capacitor 310, the fifth variable
  • the adjustable 90-degree power combiner can make two phase differences
  • the phase difference of the 90-degree equal-amplitude RF input signal at different frequencies is maintained at 0 degrees, and the amplitude difference is close to (equal to or approximately equal to) 0 dBc.
  • the variable capacitor in the adjustable 90-degree power combiner 170 can be changed to a fixed capacitor, and the variable resistor can be changed to a fixed resistor, that is, an unadjustable structure (as shown in FIG.
  • the 90 degree power combiner unit 171 replaces the adjustable 90 degree power combiner unit 170); for example, the 90 degree power combiner of the non-adjustable structure may be a matching network and a phase shifting network built by an inductor, a capacitor, a resistor device, or It is an impedance and phase transformation network built by metal coupling devices, and it can also be an impedance and phase transformation network built by a transmission line network. Also, in order to minimize the number of components in the adjustable 90-degree power combiner, the variable capacitance in the adjustable 90-degree power combiner can be combined with nearby variable capacitors, fixed capacitors or inductors.
  • each variable capacitor in the adjustable 90-degree power combiner unit 170 may be composed of a capacitor C0 and an n-way switched capacitor group in parallel, or may be composed of an n-way switched capacitor group independently connected in parallel;
  • each group of switched capacitors is composed of a capacitor and a switch in series, and the switches in each group of switched capacitors are respectively connected to the control unit 100; for example, the switched capacitor group can be connected in series by the capacitor C1 and the capacitor C2 in series.
  • Switch K2 ... capacitor Cn series switch Kn and other n switch capacitors of the same structure are connected in parallel; switch K1, switch K2 ...
  • switch Kn are respectively connected with the control unit 100, through the control unit 100 to close or open the switch K1 to switch Kn specific
  • the number of switches is used to obtain the capacitance value of the corresponding switched capacitor, thereby optimizing the phase difference and amplitude difference of the two RF input signals in the adjustable 90 degree power combiner, so that the phase difference between the two RF input signals at different frequencies Maintaining at 0 degrees, the amplitude difference is close to (equal to or approximately equal to) 0dBc, thereby not only improving the maximum output linear power of the balanced RF power amplifier, but also RF power amplifier output power value of the minimum phase variation with load.
  • each variable capacitor in the adjustable 90-degree power combiner unit 170 may also be composed of a capacitor C0 and an n-way switched capacitor group connected in series, or may be composed of an n-way switched capacitor group independently connected in series;
  • each group of switch capacitors is composed of a capacitor and a switch in parallel, and the switches in each group of switch capacitors are respectively connected to the control unit 100; for example, the switch capacitor group can be connected by a capacitor C1 in parallel with the switch K1 and the capacitor C2.
  • Parallel switch K2 ... capacitor Cn parallel switch Kn and other n switch capacitors of the same structure are connected in series; switch K1, switch K2 ...
  • switch Kn are respectively connected with the control unit 100, through the control unit 100 to close or open the switch K1 to the switch Kn
  • a certain number of switches are used to obtain the capacitance values of the corresponding switched capacitors, thereby optimizing the phase difference and amplitude difference of the two RF input signals in the adjustable 90-degree power combiner, so that the phases of the two RF input signals at different frequencies
  • the difference is maintained at 0 degrees or close to 0 degrees, and the amplitude difference is close to (equal to or approximately equal to) 0 dBc, thereby not only improving the maximum output linear power of the balanced RF power amplifier. Further such that the output power of the RF power amplifier according to the present balanced load with a minimum phase change.
  • each variable resistor in the adjustable 90-degree power combiner unit 170 may be composed of a resistor R0 and an n-way switch resistor group in parallel, or may be composed of an n-way switch resistor group independently connected in parallel;
  • each group of switch resistors is composed of a resistor and a switch in series, and the switches in each group of switch resistors are respectively connected to the control unit 100; for example, the switch resistor group can be connected in series by the resistor R1 and the resistor R2 in series.
  • Switch K2 ... resistor Rn series switch Kn and other n switch resistors of the same structure are connected in parallel; switch K1, switch K2 ...
  • switch Kn are respectively connected with the control unit 100, through the control unit 100 to close or open the switch K1 to switch Kn specific
  • the number of switches to obtain the resistance value of the corresponding switch resistor thereby optimizing the phase difference and amplitude difference of the two RF input signals in the adjustable 90-degree power combiner, so that the phase difference between the two RF input signals at different frequencies Keep at 0 degrees or close to 0 degrees, the amplitude difference is close to (equal to or approximately equal to) 0dBc, which not only improves the maximum output linear power of the balanced RF power amplifier It also minimizes the output power of the balanced RF power amplifier with load phase changes.
  • each variable resistor in the adjustable 90-degree power combiner unit 170 may be composed of a resistor R0 and an n-way switch resistor group connected in series, or may be composed of an n-way switch resistor group independently connected in series;
  • each set of switch resistors is composed of a resistor and a switch in parallel, and the switches in each set of switch resistors are respectively connected to the control unit 100; for example, the switch resistor group can be connected in parallel by the resistor R1 and the resistor R2 in parallel.
  • Switch K2 ... resistor Rn parallel switch Kn and other n switch resistors of the same structure are connected in series; switch K1, switch K2 ...
  • switch Kn are respectively connected with the control unit 100, through the control unit 100 to close or open the switch K1 to the switch Kn specific
  • the number of switches to obtain the resistance value of the corresponding switch resistor thereby optimizing the phase difference and amplitude difference of the two RF input signals in the adjustable 90-degree power combiner, so that the phase difference between the two RF input signals at different frequencies Keep at 0 degrees or close to 0 degrees, the amplitude difference is close to (equal to or approximately equal to) 0dBc, which not only improves the maximum output linear power of the balanced RF power amplifier This also enables balanced RF power amplifier with an output power of the minimum phase variation load.
  • Each of the switches in FIG. 9A to FIG. 10B can be designed on a silicon-on-insulator (Silicon On Insulator, SOI) chip or on a gallium arsenide (GaAs) chip. It can also be designed on a Silicon Germanium (SiGe) chip.
  • SOI silicon-on-insulator
  • GaAs gallium arsenide
  • SiGe Silicon Germanium
  • Each of the resistors and capacitors in Figures 9A through 10B can be designed on an integrated circuit chip or can be implemented in discrete devices.
  • the communication terminal integrates the RF power amplifiers of Band7, Band38, Band40 and Band41 into one integrated circuit chip.
  • the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the phase difference between the two RF input signals input to the adjustable 90-degree power combiner 170 is 80 degrees.
  • the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, the output power varies with the load phase, which is approximately 1.1 dBc; curve 208 is input to an adjustable 90 degrees.
  • the phase difference between the two RF input signals in the power combiner is reduced to 80 degrees.
  • the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the output power varies with the load phase. 2dBc; curve 209 is the phase difference of the two RF input signals input to the adjustable 90 degree power combiner is increased to 100 degrees, and the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1.
  • VSWR load impedance standing wave ratio
  • the output power varies with the phase of the load, which is approximately 2.6dBc. Therefore, when designing the adjustable 90-degree power combiner, the phase difference between the two RF input signals input into the adjustable 90-degree power combiner is maintained at 90 degrees.
  • the balanced RF power amplifier is at the load impedance standing wave ratio (VSWR). In the case of 3:1, the output power varies minimally with load phase.
  • the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the amplitude difference between the two RF input signals input to the adjustable 90-degree power combiner is -1dBc, 0dBc. , +1dBc, the output power (Pout) curve with load phase change; as can be seen from the figure, curve 210 is the amplitude difference between the two RF input signals input to the adjustable 90-degree power combiner is 0dBc
  • the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the output power varies with the load phase, which is about 1.1 dBc; the curve 211 is input to the adjustable 90-degree power combiner.
  • the balanced RF power amplifier When the amplitude difference between the two RF input signals is -1dBc, the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the output power varies with the load phase, which is about 1.4dBc; curve 212
  • the balanced RF power amplifier When the amplitude difference between the two RF input signals input into the adjustable 90-degree power combiner is +1dBc, the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1, and the output power is Load phase change, about 1.6dBc; so design
  • the balanced RF power amplifier When the 90-degree power combiner is adjusted, the balanced RF power amplifier has a load impedance standing wave ratio (VSWR) of 3 when the amplitude difference between the two RF input signals input into the adjustable 90-degree power combiner is reduced to 0 dBc. In the case of 1 case, the output power changes minimally with the load phase.
  • the RF power amplifier has a load impedance standing wave ratio (VSWR) of In the case of 3:1, the output power changes with load phase change. Therefore, in the balanced RF power amplifier, the phase difference of the two RF input signals input into the adjustable 90-degree power combiner is maintained at around 90 degrees in the frequency range of 2.3 GHz to 2.69 GHz, and the amplitude difference is close to 0 dBc.
  • VSWR load impedance standing wave ratio
  • the phase difference between the two RF input signals input into the adjustable 90-degree power combiner is 86 degrees (curve 213) to 90 degrees ( Curve 214), the amplitude difference is +0.5dBc to -0.5dBc, therefore, the balanced RF power amplifier can maintain better performance in the Band7 band.
  • the central frequency band is shifted to the Band40 (2.3 GHz to 2.4 GHz) frequency range by changing the values of the variable capacitor and the variable resistor in the adjustable 90-degree power combiner, and the input is adjusted to 90 degrees.
  • the phase difference between the two RF input signals in the combiner is 88 degrees (curve 213) to 90 degrees (curve 214), and the amplitude difference is +0.18dBc to +0.23dBc. Therefore, the balanced RF power amplifier is in the Band40 frequency band. Can maintain good performance.
  • the center band is shifted to the Band41 (2.496 GHz to 2.69 GHz) frequency range by changing the values of the variable and variable resistors in the adjustable 90-degree power combiner; input to an adjustable 90-degree power
  • the phase difference between the two RF input signals in the combiner is 87 degrees (curve 213) to 89 degrees (curve 214), and the amplitude difference is -0.21dBc to -0.05dBc. Therefore, the balanced RF power amplifier is in Band38 and Band41.
  • the frequency band can maintain good performance.
  • the existing single-ended structure RF power amplifier includes a control unit 100, a driver stage unit 110, a power stage unit 120, an output matching network unit 150, and a switch module unit 130; wherein, the control unit 100 and the drive unit respectively The stage unit 110, the power stage unit 120 and the switch module unit 130 are connected; the input end of the drive stage unit 110 is connected to the RF signal input end, and the output end of the drive stage unit 110 is connected to the input end of the power stage unit 120, the power stage unit The output of 120 is coupled to the input of output matching network unit 150, and the output of output matching network unit 150 is coupled to the input of switch module unit 130.
  • the received RF input signal is amplified by the driver stage unit 110, transmitted to the power stage unit 120 for amplification, and then transmitted to the output matching network unit 150.
  • the output matching network unit 150 participates in impedance conversion and suppresses harmonic energy in the RF input signal. And transmitting the radio frequency input signal to the switch module unit 130, and controlling the switch state of the switch module unit 130 through the control unit 100 according to the frequency band requirement, so that the radio frequency input signal is input to the next stage circuit through a specific radio frequency transmission path.
  • the advantage of the single-ended RF power amplifier is that the structure is simple, but it is difficult to meet the maximum linear power requirement proposed by the high power terminal.
  • the maximum output linear power of the single-ended structure RF power amplifier also has a large change.
  • the performance of the balanced RF power amplifier is compared to the performance of an existing single-ended RF power amplifier through Figures 17 and 18.
  • the situation encountered by the radio frequency antenna of the communication terminal is complicated.
  • the load phase of the radio frequency antenna varies greatly under different holding modes of the mobile phone; the communication protocol has clear requirements on the maximum linear power of the communication terminal, for example, in Band7.
  • the maximum output linear power of the antenna of the communication terminal is not less than 23 dBm; therefore, the communication terminal needs to output the maximum value under the condition that the load phase of the RF antenna changes greatly. Linear power.
  • the balanced RF power amplifier has an existing single-ended RF power amplifier and a Gain curve of the balanced RF power amplifier at a load of 50 Ohm; the existing one shown by the curve 201
  • the single-ended structure RF power amplifier has a single-output power amplifier unit, and the maximum output linear power of the single-ended RF power amplifier is 34dBm (2.51 watts); the balanced RF power amplifier shown in curve 202 has two power levels.
  • Amplifying unit, the balanced output of the RF power amplifier has a maximum output linear power of 37dBm (5.01 watts), which is nearly twice that of the existing single-ended RF power amplifier; therefore, the balanced RF power amplifier is compared to the existing single-ended structure.
  • RF power amplifiers better support the linear power requirements of mobile high power terminal (HPUE) functions.
  • curve 203 is the relationship between the maximum output linear power and the RF antenna load phase when the existing single-ended RF power amplifier has a load impedance standing wave ratio (VSWR) of 3:1.
  • VSWR load impedance standing wave ratio
  • Curve 204 is the relationship between the maximum output linear power and the RF antenna load phase in the case of a balanced RF power amplifier with a load impedance standing wave ratio (VSWR) of 3:1.
  • VSWR load impedance standing wave ratio
  • the balanced RF The maximum output linear power variation of the power amplifier is much smaller than the maximum output linear power variation of the existing single-ended RF power amplifier; the minimum saturation power of the balanced RF power amplifier is much larger than the existing one under different RF antenna load phase changes.
  • the load of the first power stage unit 120 and the second power stage unit 121 of the balanced radio frequency power amplifier also changes, because the 90-degree power splitter divides the radio frequency input signal into The two-channel RF input signal has a phase difference of 90 degrees. Therefore, the load variation trend of the first power stage unit 120 and the second power stage unit 121 is opposite, resulting in the first power stage unit 120 and the second power stage unit 121.
  • the maximum output linear power change trend is reversed.
  • the curve 205 is the maximum linear power curve of the first power stage unit 120 of the balanced RF power amplifier outputted under the load phase of different RF antennas
  • the curve 206 is the second power stage unit 121 of the balanced RF power amplifier at different RF levels.
  • the maximum linear power curve outputted under the load phase of the antenna between 0 and 60 degrees of the load phase, the curve 205 rises and the curve 206 falls; between the load phase is between 60 and 100 degrees, the curve 205 is in the high power segment.
  • the curve 206 is in the low power segment; between the load phase being between 100 and 180 degrees, the curve 205 is decreased and the curve 206 is rising; since the curve 205 and the curve 206 are opposite in the load phase from 0 degrees to 180 degrees, Therefore, the maximum output linear power superposition result of the first power stage unit 120 and the second power stage unit 121 does not change much, and the characteristic that the balanced RF power amplifier is insensitive to the RF antenna load is realized.
  • the balanced RF power amplifier provided by the invention divides the RF input signal into two equal-amplitude signals with a phase difference of 90 degrees through a 90-degree splitter unit, and amplifies the two RF input signals and inputs them to the adjustable 90
  • the power combiner controls the value of the adjustable capacitor and the adjustable resistor in the adjustable 90-degree power combiner through the control unit, so that the phase difference of the two RF input signals at different frequencies is converted to 0 degrees or close to 0 degrees.
  • the amplitude difference is close to 0dBc, it is synthesized into one RF input signal, and is input to the next stage circuit through a specific RF transmission path. Therefore, the balanced RF power amplifier not only improves the maximum linear power of the output, but also reduces the sensitivity to changes in the RF antenna load, thereby enabling support for the Mobile High Power Terminal (HPUE) function.
  • HPUE Mobile High Power Terminal
  • the balanced RF power amplifier provided by the present invention can be applied to a power amplifier circuit module of a plurality of modulated signals, including but not limited to WCDMA, TDSCDMA, CDMA2000, LTE, Wifi, and the like.
  • the balanced RF power amplifier can also be used in different frequency bands. Currently, it is Band7, Band38, Band40, Band41, and can also be applied to 5G frequency bands, such as Band42, Band43, and so on.
  • the balanced RF power amplifier provided by the present invention can also be used in an integrated circuit chip.
  • the specific structure of the balanced RF power amplifier in the integrated circuit chip will not be described in detail here.
  • the balanced RF power amplifier can also be used in communication terminals as an important component of radio frequency integrated circuits.
  • the term "communication terminal” as used herein refers to a computer device that can be used in a mobile environment and supports various communication systems such as GSM, EDGE, TD_SCDMA, TDD_LTE, and FDD_LTE, including mobile phones, notebook computers, tablet computers, and on-board computers.
  • the technical solution provided by the present invention is also applicable to other radio frequency integrated circuit applications, such as a communication base station.

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Abstract

本发明公开了一种平衡式射频功率放大器、芯片及通信终端。该射频功率放大器通过90度功分器单元将射频输入信号分成相位差为90度的两路等幅信号，并将该两路射频输入信号进行放大后输入到可调90度功合器，通过控制单元控制可调90度功合器中的可调电容和可调电阻的数值，使得两路射频输入信号在不同频率处的相位差和幅度差最小时合成为一路射频输入信号，并通过特定的射频发射路径输入到下一级电路。本平衡式射频功率放大器不仅提高了输出的最大线性功率，还降低了对射频天线负载变化的敏感度，从而实现对移动高功率终端功能的支持。

Description

一种平衡式射频功率放大器、芯片及通信终端 技术领域
本发明涉及一种射频功率放大器，尤其涉及一种支持移动高功率终端功能的平衡式射频功率放大器，同时也涉及包括该平衡式射频功率放大器的集成电路芯片及相应的通信终端，属于射频集成电路技术领域。
背景技术
射频功率放大器广泛应用在手机等无线通信设备中，为了适应现代通信事业对通信速度越来越高的要求，各种通信终端设备必须提高技术指标，以满足这种需求。例如，当前业界提出了针对Band 41(2496MHz～2690MHz)频段实现支持移动高功率终端(High Performance User Equipment，简称为HPUE)功能的要求，因此，移动终端天线发射线性功率需要提高到26dBm。由于通信终端射频天线设计面积减小，导致射频天线增益降低，射频天线负载阻抗驻波比(VSWR)变大，这就要求射频功率放大器输出更大的线性功率来满足通信终端射频指标。
在现有技术中，用于提高通信终端射频功率放大器的线性功率的方法主要包括增大输出电压和增大输出电流两种；增大输出电压即增加射频功率放大器输出端口的电源电压，使得射频功率放大器输出电压波形峰峰值得到增加，从而提高射频功率放大器的线性功率；增大输出电流即增加射频功率放大器在最大功率处的输出电流能力。
增大射频功率放大器的电源电压主要分为恒定增大和瞬时增大两种方式。恒定增大的优点是实现方式简单，代表技术为APT(Average Power Tracking)，但有以下缺点：1)对功率管的坚固性(ruggedness)提出了更高的要求；2)需要通信终端额外增加电源升压器来提升射频功率放大器的电源电压；3)降低射频功率放大器回退功率处的功率附加效率。瞬时增大的优点是提升射频功率放大器回退功率处的功率附加效率，代表技术为ET(Envelop Tracking)，但也有以下缺点：1)对功率管的坚固性(ruggedness)提出了更高的要求；2)实现方式复 杂，需要通信终端的基带芯片和射频前端一起配合工作；3)处理更大带宽射频信号有很大的技术难度。
增大射频功率放大器的输出电流主要分为增大输出功率管面积和多路功率放大器并联两种方式。增大输出功率管面积的优点是实现方式简单，但会带来寄生参数过大的问题，当前支持HPUE功能的Band41频段属于较高频段，较大的寄生参数会恶化高频段射频功率放大器的性能，其线性功率提升也会受限。多路功率放大器并联相对于增大功率管面积，避免了寄生参数过大的问题，但不能解决通信终端由于天线负载阻抗变化导致的射频功率放大器输出功率损失。
发明内容
本发明所要解决的首要技术问题在于提供一种平衡式射频功率放大器。
本发明所要解决的另一技术问题在于提供一种包括该平衡式射频功率放大器的集成电路芯片及相应的通信终端。
为了实现上述发明目的，本发明采用下述的技术方案：
根据本发明实施例的第一方面，提供一种平衡式射频功率放大器，包括控制单元、第一驱动级单元、90度功分器单元、第一功率级单元、第二功率级单元、可调90度功合器单元，所述控制单元分别与第一驱动级单元、所述第一功率级单元、所述第二功率级单元和所述可调90度功合器单元连接，所述第一驱动级单元的输入端与射频信号输入端连接，所述第一驱动级单元的输出端与所述90度功分器单元的输入端连接，所述90度功分器单元的输出端分别与所述第一功率级单元、所述第二功率级单元的输入端连接，所述第一功率级单元与所述第二功率级单元的输出端分别与所述可调90度功合器单元的输入端连接，所述可调90度功合器单元的输出端与射频发射路径连接；
所述射频输入信号经所述第一驱动级单元放大后输入到所述90度功分器单元，所述90度功分器单元将射频输入信号分成相位差为90度的两路等幅射频输入信号，对应输入到所述第一功率级单元、所述第二功率级单元中进行放大后，输入到所述可调90度功合器单元，通过所述控制单元控制所述可调90度功合器单元，使得两路等幅射频输入信号在不同频率处的相位差和幅度差最小时，将两路射频输入信号 合成为一路射频输入信号输入到射频发射路径中。
其中较优地，采用第二驱动级单元、第三驱动级单元替代所述第一驱动级单元，并且所述第二驱动级单元与所述第三驱动级单元设置在所述90度功分器单元与所述第一功率级单元和所述第二功率级单元之间，所述90度功分器单元的输入端与所述射频信号输入端连接。
其中较优地，所述可调90度功合器单元与所述射频发射路径之间设置有开关模组单元，所述开关模组单元的输入端与所述可调90度功合器单元的输出端连接，所述开关模组单元的输出端分别与所述射频发射路径和射频接收路径连接；
所述控制单元与所述开关模组单元连接，用于根据频段要求控制所述开关模组单元的开关状态，将所述可调90度功合器单元输出的射频输入信号通过与频段对应的射频发射路径输入到下一级电路。
其中较优地，所述开关模组单元与所述射频发射路径之间设置有Band7双工器、Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。
其中较优地，所述可调90度功合器单元包括90度移相器和威尔金森功率合成器；所述90度移相器包括相位滞后阻抗变换网络和相位超前阻抗变换网络，所述相位滞后阻抗变换网络的输入端与所述第一功率级单元的输出端连接，所述相位滞后阻抗变换网络的输出端与所述威尔金森功率合成器的一个输入端连接；所述相位超前阻抗变换网络的输入端与所述第二功率级单元的输出端连接，所述相位超前阻抗变换网络的输出端与所述威尔金森功率合成器的另一个输入端连接。
其中较优地，所述相位滞后阻抗变换网络包括第一电感、第一可变电容和第二可变电容；所述第一电感的一端作为所述相位滞后阻抗变换网络的输入端，用于与所述第一功率级单元的输出端连接；所述第一电感的另一端分别与所述第一可变电容和所述第二可变电容的一端连接，所述第一可变电容的另一端接地，所述第二可变电容的另一端作为所述相位滞后阻抗变换网络的输出端，用于与所述威尔金森功率合成器的一个输入端连接。
其中较优地，所述相位超前阻抗变换网络包括第三可变电容和第二电感；所述第三可变电容的一端作为所述相位超前阻抗变换网络的 输入端，用于与所述第二功率级单元的输出端连接；所述第三可变电容的另一端作为所述相位超前阻抗变换网络的输出端，用于分别与所述第二电感的一端和所述威尔金森功率合成器的另一个输入端连接，所述第二电感的另一端接地。
其中较优地，所述威尔金森功率合成器包括第四可变电容、可变电阻、第五可变电容、第三电感、第四电感和第六可变电容；所述第四可变电容的一端作为所述威尔金森功率合成器的一个输入端，用于分别与所述相位滞后阻抗变换网络的输出端、所述可变电阻的输出端和所述第三电感的一端连接；所述可变电阻的一端作为所述威尔金森功率合成器的另一个输入端，用于分别与所述相位超前阻抗变换网络的输出端、所述第五可变电容的一端和所述第四电感的一端连接；所述第三电感和第四电感的另一端作为所述可调90度功合器单元的输出端，用于分别与所述第六可变电容的一端、所述开关模组单元的输入端连接；所述第四可变电容、所述第五可变电容和所述第六可变电容的另一端分别接地。
其中较优地，所述可调90度功合器单元中的每个可变电容由电容和n路开关电容组并联组成；或者所述可调90度功合器单元中的每个所述可变电容由n路所述开关电容组并联组成，所述n为正整数。
其中较优地，在n路所述开关电容组中，每组开关电容由一个电容与一个开关串联组成，并且，每组所述开关电容中的开关分别与所述控制单元连接，所述控制单元闭合或打开n路所述开关电容组中特定数量的开关，得到相应的开关电容的电容值。
其中较优地，所述可调90度功合器单元中的每个可变电容由电容和n路开关电容组串联组成；或者所述可调90度功合器单元中的每个可变电容由n路开关电容组串联组成。
其中较优地，在n路所述开关电容组中，每组开关电容由一个电容与一个开关并联组成，并且，每组所述开关电容中的开关分别与所述控制单元连接，所述控制单元闭合或打开n路所述开关电容组中特定数量的开关，得到相应的开关电容的电容值。
其中较优地，所述可调90度功合器单元中的每个可变电阻由电阻和n路开关电阻组并联组成；或者所述可调90度功合器单元中的每个 可变电阻由n路开关电阻组并联组成。
其中较优地，在n路所述开关电阻组中，每组开关电阻由一个电阻与一个开关串联组成，并且，每组所述开关电阻中的开关分别与控制单元连接，所述控制单元闭合或打开n路所述开关电阻组中特定数量的开关，得到相应的开关电阻的电阻值。
其中较优地，所述可调90度功合器单元中的每个可变电阻由电阻和n路开关电阻组串联组成；或者所述可调90度功合器单元中的每个可变电阻由n路开关电阻组串联组成。
其中较优地，在n路所述开关电阻组中，每组开关电阻由一个电阻与一个开关并联组成，并且，每组所述开关电阻中的开关分别与控制单元连接，所述控制单元闭合或打开n路所述开关电阻组中特定数量的开关，得到相应的开关电阻的电阻值。
其中较优地，所述90度功分器单元中设置有第一匹配网络，用于实现阻抗匹配，并参与配合所述90度功分器单元将射频输入信号分成相位差为90度的两路等幅射频输入信号。
其中较优地，所述可调90度功合器单元中设置有第二匹配网络，用于实现阻抗匹配，并参与配合所述可调90度功合器单元使得两路等幅射频输入信号在不同频率处的相位差和幅度差最小。
其中较优地，采用不可调结构的90度功合器单元替代所述可调90度功合器单元；所述不可调结构的90度功合器为电感、电容和电阻器件搭建的匹配网络和移相网络；或者所述不可调结构的90度功合器为金属耦合器件搭建的阻抗和相位变换网络；或者所述不可调结构的90度功合器为传输线网络搭建的阻抗和相位变换网络。
其中较优地，所述开关模组单元包括n组收发开关，每一组所述收发开关的公共端分别与所述可调90度功合器单元的输出端连接，每一组所述收发开关的一个输出端分别与对应的射频发射路径连接，每一组收发开关的另一个输出端分别与对应的射频接收路径连接。
根据本发明实施例的第二方面，提供一种集成电路芯片，所述集成电路芯片中包括上述的平衡式射频功率放大器。
根据本发明实施例的第三方面，提供一种通信终端，所述通信终端中包括上述的平衡式射频功率放大器。
本发明所提供的平衡式射频功率放大器通过90度功分器单元将射频输入信号分成相位差为90度的两路等幅信号，并将该两路射频输入信号进行放大后输入到可调90度功合器，通过控制单元控制可调90度功合器中的可调电容和可调电阻的数值，使得两路射频输入信号在不同频率处的相位差和幅度差最小时合成为一路射频输入信号，并通过特定的射频发射路径输入到下一级电路。因此，本平衡式射频功率放大器不仅提高了输出的最大线性功率，还降低了对射频天线负载变化的敏感度，从而实现对移动高功率终端(HPUE)功能的支持。
附图说明
图1为本发明实施例1所提供的平衡式射频功率放大器的结构示意图1；
图2为本发明实施例1所提供的平衡式射频功率放大器的结构示意图2；
图3为本发明实施例1所提供的平衡式射频功率放大器的结构示意图3；
图4为本发明实施例2所提供的平衡式射频功率放大器的结构示意图1；
图5为本发明实施例2所提供的平衡式射频功率放大器的结构示意图2；
图6为本发明实施例2所提供的平衡式射频功率放大器的结构示意图3；
图7为本发明所提供的平衡式射频功率放大器中，可调90度功合器的电路原理图；
图8为本发明所提供的平衡式射频功率放大器中，采用不可调结构的90度功合器单元代替可调90度功合器单元的结构示意图；
图9A为本发明所提供的平衡式射频功率放大器中，可调90度功合器中的每个可变电容的一种电路原理图；
图9B为本发明所提供的平衡式射频功率放大器中，可调90度功合器中的每个可变电容的另一种电路原理图；
图10A为本发明所提供的平衡式射频功率放大器中，可调90度功合器中的每个可变电阻的一种电路原理图；
图10B为本发明所提供的平衡式射频功率放大器中，可调90度功合器中的每个可变电阻的另一种电路原理图；
图11为本发明所提供的平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1，可调90度功合器中的两路射频输入信号的相位差分别为80度、90度、100度情况下，输出功率(Pout)随负载相位变化的曲线示意图；
图12为本发明所提供的平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1，可调90度功合器中的两路射频输入信号的幅度差分别为－1dBc、0dBc、+1dBc情况下，输出功率(Pout)随负载相位变化的曲线示意图；
图13为本发明所提供的平衡式射频功率放大器中，中心频带为2.5GHz的可调90度功合器的设计结果；
图14为本发明所提供的平衡式射频功率放大器中，通过改变可调90度功合器中的可变电容和可变电阻的数值，将中心频带移到Band40(2.3GHz～2.4GHz)的设计结果；
图15为本发明所提供的平衡式射频功率放大器中，通过改变可调90度功合器中的可变电容和可变电阻的数值，将中心频带移到Band41(2.496GHz～2.69GHz)的设计结果；
图16为现有的单端结构射频功率放大器的结构示意图；
图17为现有的单端结构射频功率放大器与本发明所提供的平衡式射频功率放大器的增益(Gain)对比的曲线示意图；
图18为现有的单端结构射频功率放大器与本发明所提供的平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1的情况下，输出功率(Pout)对比的曲线示意图；
图19为本发明所提供的平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1的情况下，第一功率级单元与第二功率级单元的输出功率(Pout)对比的曲线示意图。
具体实施方式
下面结合附图和具体实施例对本发明的技术内容做进一步的详细说明。
本发明提供了一种平衡式射频功率放大器，用于支持移动高功率终 端(HPUE)的相关指标，实现提高射频功率放大器的最大输出线性功率和降低射频功率放大器对负载敏感度的目的。下面通过不同的实施例对本平衡式射频功率放大器的结构和工作原理进行详细说明。
实施例1
如图1所示，本实施例所提供的平衡式射频功率放大器包括控制单元100、第一驱动级单元110、90度功分器单元160、第一功率级单元120、第二功率级单元121、可调90度功合器单元170和开关模组单元130。其中，控制单元100分别与第一驱动级单元110、第一功率级单元120、第二功率级单元121、可调90度功合器单元170和开关模组单元130连接，可以用于控制第一驱动级单元110、第一功率级单元120、第二功率级单元121的静态电流。第一驱动级单元110的输入端与射频信号输入端连接，第一驱动级单元110的输出端与90度功分器单元160的输入端连接，90度功分器单元160的输出端分别与第一功率级单元120、第二功率级单元121的输入端连接，第一功率级单元120与第二功率级单元121的输出端分别与可调90度功合器单元170的输入端连接，可调90度功合器单元170的输出端与开关模组单元130的输入端连接，开关模组单元130的输出端分别与射频发射路径(A1～An)和射频接收路径(B1～Bm)连接，其中m、n均为正整数。
当有射频输入信号进入本平衡式射频功率放大器时，该射频输入信号通过射频信号输入端输入到第一驱动级单元110中进行放大，经第一驱动级单元110放大后的射频输入信号会输入到90度功分器单元160中，90度功分器单元160将该射频输入信号分成相位差为90度(或接近90度，下同)的两路等幅射频输入信号，对应输入到第一功率级单元120、第二功率级单元121中进行放大，经第一功率级单元120、第二功率级单元121放大后的两路射频输入信号对应输入到可调90度功合器单元170，通过控制单元100控制可调90度功合器单元170，使得两路射频输入信号在不同频率处的相位差和幅度差最小时(优选为相位差为0度或者接近0度，幅度差为0dBc或者接近0dBc)，将两路射频输入信号合成为一路射频输入信号输入到开关模组单元130，此时，根据频段要求控制单元100会控制开关模组单元130的开关状态，将合并后的射频输入信号通过特定射频发射路径输入到下一级电 路。
为了使经第一驱动级单元110放大后的射频输入信号能最大限度的输入到90度功分器单元160中，并且还能参与配合90度功分器单元160将射频输入信号分成相位差为90度的两路等幅射频输入信号，以及将两路等幅射频输入信号最大限度的输入到第一功率级单元120、第二功率级单元121中进行放大，同时尽可能的降低本平衡式射频功率放大器的设计复杂度，可以在90度功分器单元160中设置第一匹配网络，通过该第一匹配网络实现90度功分器单元160分别与第一驱动级单元110、第一功率级单元120和第二功率级单元121之间的阻抗匹配。
同样，为了使经第一功率级单元120和第二功率级单元121放大的射频输入信号能最大限度的输入到可调90度功合器单元170中，并且还能参与配合可调90度功合器单元170使得两路射频输入信号在不同频率处的相位差为0度或者接近0度，幅度差为0dBc或者接近0dBc，以及将两路射频输入信号合成为一路射频输入信号最大限度地输入到开关模组单元130，同时尽可能的降低本平衡式射频功率放大器的设计复杂度，也可以在可调90度功合器单元170中设置第二匹配网络，通过该第二匹配网络实现可调90度功合器单元170分别与第一功率级单元120、第二功率级单元121和开关模组单元130之间的阻抗匹配。
如图1所示，开关模组单元130包括n组收发开关(单刀双掷开关S1～Sn，n为正整数)。每一组收发开关的公共端分别与可调90度功合器单元170的输出端连接，每一组收发开关的一个输出端分别与对应的射频发射路径连接，每一组收发开关的另一个输出端分别与对应的射频接收路径连接。
为了缩小本平衡式射频功率放大器的封装尺寸，优化与集成滤波器件之间的匹配，简化通信终端设计，可以在本实施例的开关模组单元130与射频发射路径之间设置Band7双工器、Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。例如，如图2所示，可以在第一组收发开关S1和射频发射路径之间设置Band7双工器，在最后一组收发开关Sn和射频发射路径之间设置Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。
为了方便通信终端用户自定义应用，如图3所示，还可以将本实施例的开关模组单元130去掉，通过可调90度功合器单元170将两路等幅射频输入信号合成为一路射频输入信号后直接输入到特定的射频发射路径后，然后输入到下一级电路。
实施例2
如图4所示，本实施例所提供的平衡式射频功率放大器与实施例1所提供的平衡式射频功率放大器的不同之处在于，将90度功分器单元160设置在第一驱动级单元110前面，并且将第一驱动级单元110更改为第二驱动级单元111、第三驱动级单元112。因此，90度功分器单元160的输入端与射频信号输入端连接，90度功分器单元160的输出端对应与第二驱动级单元111、第三驱动级单元112的输入端连接，第二驱动级单元111、第三驱动级单元112的输出端对应与第一功率级单元120、第二功率级单元121的输入端连接。本实施例所提供的平衡式射频功率放大器与实施例1所提供的平衡式射频功率放大器的相同部分不再赘述；同实施例1中所述，在90度功分器单元中可以设置第一匹配网络，在可调90度功合器单元170中可以设置第二匹配网络。
当有射频输入信号进入本平衡式射频功率放大器时，该射频输入信号通过射频信号输入端输入到90度功分器单元160中，90度功分器单元160将该射频输入信号分成相位差为90度的两路等幅射频输入信号，对应输入到第二驱动级单元111、第三驱动级单元112中进行放大，经第二驱动级单元111、第三驱动级单元112放大的两路等幅射频输入信号将对应输入到第一功率级单元120、第二功率级单元121中进行进一步放大后，对应输入到可调90度功合器单元170中，通过控制单元100控制可调90度功合器单元170，使得两路射频输入信号在不同频率处的相位差变换为0度或者接近0度，幅度差为0dBc或者接近0dBc时，将两路射频输入信号合成为一路射频输入信号输入到开关模组单元130，此时，根据频段要求控制单元100会控制开关模组单元130的开关状态，将合并后的射频输入信号通过特定射频发射路径输入到下一级电路。
同样，为了缩小本平衡式射频功率放大器的封装尺寸，优化与集成滤波器件之间的匹配，简化通信终端设计，可以在本实施例的开关 模组单元130与射频发射路径之间设置Band7双工器、Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。例如，如图5所示，可以在第一组收发开关S1和射频发射路径之间设置Band7双工器，在最后一组收发开关Sn和射频发射路径之间设置Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。
为了方便通信终端用户自定义应用，如图6所示，还可以将本实施例的开关模组单元130去掉，通过可调90度功合器单元170将两路等幅射频输入信号合成为一路射频输入信号后直接输入到特定的射频发射路径后，然后输入到下一级电路。
另外，实施例1和实施例2中的第一驱动级单元110、第二驱动级单元111和第三驱动级单元112可以为单级驱动级单元，也可以为两级驱动级单元。第一功率级单元120、第二功率级单元121可以为单级功率级单元，也可以为两级功率级单元。并且，第一功率级单元120、第二功率级单元121可以为砷化镓(GaAs)衬底上或锗硅(SiGe)衬底上制作的异质结晶体管(HBT)或高电子迁移率晶体管(HEMT)或赝高电子迁移率晶体管(p－HEMT)，也可以为双极结晶体管(BJT)，或硅基衬底上制作的互补金属氧化物半导体晶体管(CMOS)。
如图7所示，在实施例1和实施例2所提供的平衡式射频功率放大器中，可调90度功合器单元170包括90度移相器1和威尔金森功率合成器2。其中，90度移相器1包括相位滞后阻抗变换网络3和相位超前阻抗变换网络4，相位滞后阻抗变换网络3的输入端与第一功率级单元120的输出端连接，相位滞后阻抗变换网络3的输出端与威尔金森功率合成器2的一个输入端连接；相位超前阻抗变换网络4的输入端与第二功率级单元121的输出端连接，相位超前阻抗变换网络4的输出端与威尔金森功率合成器2的另一个输入端连接。经第一功率级单元120和第二功率级单元121放大后的两路相位差为90度的等幅射频输入信号，根据两路等幅射频输入信号相位关系分别进入相位滞后阻抗变换网络3和相位超前阻抗变换网络4，使得两路射频输入信号在不同频率处的相位差变换为0度、幅度差接近(等于或约等于)0dBc后，输入到威尔金森功率合成器2中，威尔金森功率合成器2将两路相位差为0度、幅度差接近(等于或约等于)0dBc的射频输 入信号合成为一路射频输入信号，并输入到开关模组单元130，或者直接输入到特定的射频发射路径中。
如图7所示，相位滞后阻抗变换网络3包括第一电感301、第一可变电容302和第二可变电容309；其中，第一电感301的一端作为相位滞后阻抗变换网络3的输入端，用于与第一功率级单元120的输出端连接；第一电感301的另一端分别与第一可变电容302和第二可变电容309的一端连接，第一可变电容302的另一端接地，第二可变电容309的另一端作为相位滞后阻抗变换网络3的输出端，用于与威尔金森功率合成器2的一个输入端连接。相位超前阻抗变换网络4包括第三可变电容303和第二电感304；第三可变电容303的一端作为相位超前阻抗变换网络4的输入端，用于与第二功率级单元121的输出端连接；第三可变电容303的另一端作为相位超前阻抗变换网络4的输出端，用于分别与第二电感304的一端和威尔金森功率合成器2的另一个输入端连接；第二电感304的另一端接地。威尔金森功率合成器2包括第四可变电容310、可变电阻308、第五可变电容311、第三电感305、第四电感306和第六可变电容307；其中，第四可变电容310的一端作为威尔金森功率合成器2的一个输入端，用于分别与相位滞后阻抗变换网络3的输出端、可变电阻308的另一端和第三电感305的一端连接；可变电阻308的一端作为威尔金森功率合成器2的另一个输入端，用于分别与相位超前阻抗变换网络4的输出端、第五可变电容311的一端和第四电感306的一端连接；第三电感305和第四电感306的另一端作为可调90度功合器单元170的输出端，用于分别与第六可变电容307的一端、开关模组单元130的输入端连接；第四可变电容310、第五可变电容311和第六可变电容307的另一端分别接地。
因此，根据不同的频率值，通过控制单元100改变可调90度功合器单元170中的多个可变电容和可变电阻的数值，可以使得可调90度功合器将两路相位差为90度的等幅射频输入信号在不同频率处的相位差保持在0度、幅度差接近(等于或约等于)0dBc。根据实际设计需求，可以将可调90度功合器170中的可变电容改为固定电容，可变电阻可以改为固定电阻，即为不可调结构(如图8所示，采用不可调结 构的90度功合器单元171代替可调90度功合器单元170)；例如，不可调结构的90度功合器可以是电感、电容、电阻器件搭建的匹配网络和移相网络，也可以是金属耦合器件搭建的阻抗和相位变换网络，还可以是传输线网络搭建的阻抗和相位变换网络。并且，为了尽可能的减少可调90度功合器中的器件数量，还可以将可调90度功合器中的可变电容与附近的可变电容、固定电容或电感进行合并。
如图9A所示，可调90度功合器单元170中的每个可变电容可以由电容C0和n路开关电容组并联组成，也可以由n路开关电容组独立并联组成；在n路开关电容组中，每组开关电容由一个电容与一个开关串联组成，并且，每组开关电容中的开关分别与控制单元100连接；例如，开关电容组可以由电容C1串联开关K1、电容C2串联开关K2……电容Cn串联开关Kn等n个同样结构的开关电容并联组成；开关K1、开关K2……开关Kn分别与控制单元100连接，通过控制单元100闭合或打开开关K1至开关Kn中特定数量的开关，来得到相应的开关电容的电容值，进而优化可调90度功合器中的两路射频输入信号的相位差和幅度差，使得两路射频输入信号在不同频率处的相位差保持在0度、幅度差接近(等于或约等于)0dBc，从而不仅使本平衡式射频功率放大器的最大输出线性功率得到提高，还使得本平衡式射频功率放大器的输出功率随着负载相位变化最小。
如图9B所示，可调90度功合器单元170中的每个可变电容还可以由电容C0和n路开关电容组串联组成，也可以由n路开关电容组独立串联组成；在n路开关电容组中，每组开关电容由一个电容与一个开关并联组成，并且，每组开关电容中的开关分别与控制单元100连接；例如，开关电容组可以由电容C1并联开关K1、电容C2并联开关K2……电容Cn并联开关Kn等n个同样结构的开关电容串联组成；开关K1、开关K2……开关Kn分别与控制单元100连接，通过控制单元100闭合或打开开关K1至开关Kn中特定数量的开关，来得到相应的开关电容的电容值，进而优化可调90度功合器中的两路射频输入信号的相位差和幅度差，使得两路射频输入信号在不同频率处的相位差保持在0度或者接近0度、幅度差接近(等于或约等于)0dBc，从而不仅使本平衡式射频功率放大器的最大输出线性功率得到提高，还使得本 平衡式射频功率放大器的输出功率随着负载相位变化最小。
如图10A所示，可调90度功合器单元170中的每个可变电阻可以由电阻R0和n路开关电阻组并联组成，也可以由n路开关电阻组独立并联组成；在n路开关电阻组中，每组开关电阻由一个电阻与一个开关串联组成，并且，每组开关电阻中的开关分别与控制单元100连接；例如，开关电阻组可以由电阻R1串联开关K1、电阻R2串联开关K2……电阻Rn串联开关Kn等n个同样结构的开关电阻并联组成；开关K1、开关K2……开关Kn分别与控制单元100连接，通过控制单元100闭合或打开开关K1至开关Kn中特定数量的开关，来得到相应的开关电阻的电阻值，进而优化可调90度功合器中的两路射频输入信号的相位差和幅度差，使得两路射频输入信号在不同频率处的相位差保持在0度或者接近0度、幅度差接近(等于或约等于)0dBc，从而不仅使本平衡式射频功率放大器的最大输出线性功率得到提高，还使得本平衡式射频功率放大器的输出功率随着负载相位变化最小。
如图9B所示，可调90度功合器单元170中的每个可变电阻可以由电阻R0和n路开关电阻组串联组成，也可以由n路开关电阻组独立串联组成；在n路开关电阻组中，每组开关电阻由一个电阻与一个开关并联组成，并且，每组开关电阻中的开关分别与控制单元100连接；例如，开关电阻组可以由电阻R1并联开关K1、电阻R2并联开关K2……电阻Rn并联开关Kn等n个同样结构的开关电阻串联组成；开关K1、开关K2……开关Kn分别与控制单元100连接，通过控制单元100闭合或打开开关K1至开关Kn中特定数量的开关，来得到相应的开关电阻的电阻值，进而优化可调90度功合器中的两路射频输入信号的相位差和幅度差，使得两路射频输入信号在不同频率处的相位差保持在0度或者接近0度、幅度差接近(等于或约等于)0dBc，从而不仅使本平衡式射频功率放大器的最大输出线性功率得到提高，还使得本平衡式射频功率放大器的输出功率随着负载相位变化最小。
图9A～图10B中的每个开关可以在绝缘衬底硅基材料(Silicon On Insulator，简称SOI)芯片上设计，也可以在砷化镓衬底材料(Gallium Arsenide，简称GaAs)芯片上设计，也可以在锗硅衬底材料(Silicon Germanium，简称SiGe)芯片上设计。图9A～图10B中的每个电阻和 电容可以在集成电路芯片上进行设计，也可以用分立器件实现。
由于Band7(2.5GHz～2.57GHz)、Band38(2.57GHz～2.62GHz)、Band40(2.3GHz～2.4GHz)和Band41(2.496GHz～2.69GHz)频段都处在2.3GHz至2.69GHz频率范围内，因此，通信终端会将Band7、Band38、Band40和Band41这几个频段的射频功率放大器都集成到一颗集成电路芯片上。
如图11所示，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1，输入到可调90度功合器170中的两路射频输入信号的相位差为80度、90度、100度情况下，随负载相位变化的输出功率(Pout)曲线；从图中可以看到，曲线207是输入到可调90度功合器170中的两路射频输入信号的相位差保持在90度时，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为1.1dBc；曲线208是输入到可调90度功合器中的两路射频输入信号的相位差减小到80度，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为2dBc；曲线209是输入到可调90度功合器中的两路射频输入信号的相位差增大到100度，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为2.6dBc；所以设计可调90度功合器时，输入到可调90度功合器中的两路射频输入信号的相位差保持在90度，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化最小。
如图12所示，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1，输入到可调90度功合器中的两路射频输入信号的幅度差为－1dBc、0dBc、+1dBc情况下，随负载相位变化的输出功率(Pout)曲线；从图中可以看到，曲线210是输入到可调90度功合器中的两路射频输入信号的幅度差为0dBc时，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为1.1dBc；曲线211是输入到可调90度功合器中的两路射频输入信号的幅度差为－1dBc时，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为1.4dBc；曲线212是输入到可调90度功合器中的两路射频输入信号的幅度差为 +1dBc时，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化，大约为1.6dBc；所以设计可调90度功合器时，输入到可调90度功合器中的两路射频输入信号的幅度差缩小为0dBc时，本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，输出功率随着负载相位变化最小。
综上所述，输入到可调90度功合器170中的两路射频输入信号的相位差偏离90度越多或幅度差越大时，射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，随负载相位变化的输出功率变化越大。因此，本平衡式射频功率放大器在2.3GHz至2.69GHz频率范围内，输入到可调90度功合器中的两路射频输入信号的相位差保持为90度附近，幅度差接近0dBc。
如图13所示，在中心频带为2.4GHz至2.6GHz的频率范围内，输入到可调90度功合器中的两路射频输入信号的相位差为86度(曲线213)至90度(曲线214)，幅度差为+0.5dBc至－0.5dBc，因此，在Band7频段本平衡式射频功率放大器可以保持较好的性能。
如图14所示，通过改变可调90度功合器中可变电容和可变电阻的数值将中心频带移到Band40(2.3GHz～2.4GHz)的频率范围内，输入到可调90度功合器中的两路射频输入信号的相位差为88度(曲线213)至90度(曲线214)，幅度差为+0.18dBc至+0.23dBc，因此，在本平衡式射频功率放大器在Band40频段可以保持较好的性能。
如图15所示，通过改变可调90度功合器中可变电容和可变电阻的数值将中心频带移到Band41(2.496GHz～2.69GHz)的频率范围内；输入到可调90度功合器中的两路射频输入信号的相位差为87度(曲线213)至89度(曲线214)，幅度差为－0.21dBc至－0.05dBc，因此，本平衡式射频功率放大器在Band38和Band41频段可以保持较好的性能。
如图16所示，现有的单端结构射频功率放大器包括控制单元100、驱动级单元110、功率级单元120、输出匹配网络单元150和开关模组单元130；其中，控制单元100分别与驱动级单元110、功率级单元120和开关模组单元130连接；驱动级单元110的输入端与射频信号输入端连接，驱动级单元110的输出端与功率级单元120的输入端连 接，功率级单元120的输出端与输出匹配网络单元150的输入端连接，输出匹配网络单元150的输出端与开关模组单元130的输入端连接。通过驱动级单元110对接收的射频输入信号进行放大，传输到功率级单元120进行放大后，传输到输出匹配网络单元150，输出匹配网络单元150参与阻抗转换和抑制射频输入信号中的谐波能量，并将射频输入信号传输到开关模组单元130，根据频段要求，通过控制单元100控制开关模组单元130的开关状态，使得射频输入信号通过特定的射频发射路径输入到下一级电路。该单端结构射频功率放大器的优点是结构简单，但很难满足高功率终端提出的最大线性功率要求。并且，当通信终端射频天线负载有较大变化时，该单端结构射频功率放大器的最大输出线性功率也会有较大变化。下面通过图17和图18，将本平衡式射频功率放大器与现有的单端结构射频功率放大器的性能进行对比。
众所周知，通信终端的射频天线遇到的情况很复杂，例如，手机在不同握持方式下，射频天线的负载相位变化很大；通信协议对通信终端的最大线性功率有明确要求，例如，在Band7频段，且功率等级为3(PC3)的标准下，通信终端的天线的最大输出线性功率不低于23dBm；因此，通信终端需要在射频天线的负载相位变化较大的条件下输出满足要求的最大线性功率。
如图17所示，本平衡式射频功率放大器在负载为50Ohm情况下，现有的单端结构射频功率放大器与本平衡式射频功率放大器增益(Gain)曲线对比；曲线201所示的现有的单端结构射频功率放大器由于只有单路功率级放大单元，单端结构射频功率放大器的最大输出线性功率为34dBm(2.51瓦)；曲线202所示的本平衡式射频功率放大器由于有两路功率级放大单元，本平衡式射频功率放大器最大输出线性功率为37dBm(5.01瓦)，接近现有的单端结构射频功率放大器的两倍；因此，本平衡式射频功率放大器相对于现有的单端结构射频功率放大器能更好地支持移动高功率终端(HPUE)功能的线性功率要求。
如图18所示，曲线203为现有的单端结构射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，最大输出线性功率与射频天线负 载相位的关系，当射频天线负载相位为80度时，现有的单端结构射频功率放大器输出的最大线性功率为27.2dBm，当射频天线负载相位为160度时，现有的单端结构射频功率放大器输出的最大线性功率为36.4dBm，因此，射频天线负载相位变化引起9.2dBc的最大线性功率变化。当射频天线负载相位为60度时，现有的单端结构射频功率放大器输出的最大线性功率为32.9dBm，当射频天线负载相位为100度时，现有的单端结构射频功率放大器输出的最大线性功率为33.8dBm，射频天线负载阻抗变化引起0.9dBc的最大线性功率变化。曲线204为本平衡式射频功率放大器在负载阻抗驻波比(VSWR)为3：1情况下，最大输出线性功率与射频天线负载相位的关系，在射频天线负载变化的情况下，本平衡式射频功率放大器的最大输出线性功率变化远小于现有的单端结构射频功率放大器的最大输出线性功率变化；在不同射频天线负载相位变化下，本平衡式射频功率放大器的最小饱和功率远大于现有的单端结构射频功率放大器。
如图19所示，当射频天线负载变化时，本平衡式射频功率放大器的第一功率级单元120与第二功率级单元121的负载也会变化，由于90度功分器将射频输入信号分成两路相位差为90度的等幅射频输入信号，因此，第一功率级单元120与第二功率级单元121的负载变化趋势相反，导致第一功率级单元120与第二功率级单元121的最大输出线性功率变化趋势相反。曲线205为本平衡式射频功率放大器的第一功率级单元120在不同射频天线的负载相位下输出的最大线性功率曲线，曲线206为本平衡式射频功率放大器的第二功率级单元121在不同射频天线的负载相位下输出的最大线性功率曲线；在负载相位为0度至60度之间，曲线205上升，曲线206下降；在负载相位为60度至100度之间，曲线205处于高功率段，曲线206处于低功率段；在负载相位为100度至180度之间，曲线205下降，曲线206上升；由于曲线205和曲线206在负载相位为0度至180度之间的变化趋势相反，因此，第一功率级单元120与第二功率级单元121的最大输出线性功率叠加结果变化不大，实现了本平衡式射频功率放大器对射频天线负载不敏感的特性。
本发明所提供的平衡式射频功率放大器通过90度功分器单元将 射频输入信号分成相位差为90度的两路等幅信号，并将该两路射频输入信号进行放大后输入到可调90度功合器，通过控制单元控制可调90度功合器中的可调电容和可调电阻的数值，使得两路射频输入信号在不同频率处的相位差变换为0度或者接近0度，幅度差接近0dBc时合成为一路射频输入信号，并通过特定的射频发射路径输入到下一级电路。因此，本平衡式射频功率放大器不仅提高了输出的最大线性功率，还降低了对射频天线负载变化的敏感度，从而实现对移动高功率终端(HPUE)功能的支持。
本发明所提供的平衡式射频功率放大器可以应用在多种调制信号的功率放大器电路模块中，调制信号包括但不限于WCDMA、TDSCDMA、CDMA2000、LTE、Wifi等。该平衡式射频功率放大器还可以应用在不同制式的频段中，目前为Band7、Band38、Band40、Band41，也可以应用到5G频段，如Band42、Band43等等。
本发明所提供的平衡式射频功率放大器还可以被用在集成电路芯片中。对于该集成电路芯片中的平衡式射频功率放大器的具体结构，在此就不再一一详述了。
另外，该平衡式射频功率放大器还可以被用在通信终端中，作为射频集成电路的重要组成部分。这里所说的通信终端是指可以在移动环境中使用，支持GSM、EDGE、TD_SCDMA、TDD_LTE、FDD_LTE等多种通信制式的计算机设备，包括移动电话、笔记本电脑、平板电脑、车载电脑等。此外，本发明所提供的技术方案也适用于其他射频集成电路应用的场合，例如通信基站等。
以上对本发明所提供的平衡式射频功率放大器、芯片及通信终端进行了详细的说明。对本领域的一般技术人员而言，在不背离本发明实质精神的前提下对它所做的任何显而易见的改动，都将属于本发明专利权的保护范围。

Claims (22)

  1. 一种平衡式射频功率放大器，其特征在于包括控制单元、第一驱动级单元、90度功分器单元、第一功率级单元、第二功率级单元、可调90度功合器单元，所述控制单元分别与第一驱动级单元、所述第一功率级单元、所述第二功率级单元和所述可调90度功合器单元连接，所述第一驱动级单元的输入端与射频信号输入端连接，所述第一驱动级单元的输出端与所述90度功分器单元的输入端连接，所述90度功分器单元的输出端分别与所述第一功率级单元、所述第二功率级单元的输入端连接，所述第一功率级单元与所述第二功率级单元的输出端分别与所述可调90度功合器单元的输入端连接，所述可调90度功合器单元的输出端与射频发射路径连接；
    所述射频输入信号经所述第一驱动级单元放大后输入到所述90度功分器单元，所述90度功分器单元将射频输入信号分成相位差为90度的两路等幅射频输入信号，对应输入到所述第一功率级单元、所述第二功率级单元中进行放大后，输入到所述可调90度功合器单元，通过所述控制单元控制所述可调90度功合器单元，使得两路等幅射频输入信号在不同频率处的相位差和幅度差最小时，将两路射频输入信号合成为一路射频输入信号输入到射频发射路径中。
  2. 如权利要求1所述的平衡式射频功率放大器，其特征在于：
    采用第二驱动级单元、第三驱动级单元替代所述第一驱动级单元，并且所述第二驱动级单元、所述第三驱动级单元设置在所述90度功分器单元与所述第一功率级单元和所述第二功率级单元之间，所述90度功分器单元的输入端与所述射频信号输入端连接。
  3. 如权利要求1或2所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元与所述射频发射路径之间设置有开关模组单元，所述开关模组单元的输入端与所述可调90度功合器单元的输出端连接，所述开关模组单元的输出端分别与所述射频发射路径和射频接收路径连接；
    所述控制单元与所述开关模组单元连接，用于根据频段要求控制所述开关模组单元的开关状态，将所述可调90度功合器单元输出的射 频输入信号通过与频段对应的射频发射路径输入到下一级电路。
  4. 如权利要求3所述的平衡式射频功率放大器，其特征在于：
    所述开关模组单元与所述射频发射路径之间设置有Band7双工器、Band38滤波器、Band40滤波器和Band41滤波器之中的任意一种。
  5. 如权利要求1或2所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元包括90度移相器和威尔金森功率合成器；所述90度移相器包括相位滞后阻抗变换网络和相位超前阻抗变换网络，所述相位滞后阻抗变换网络的输入端与所述第一功率级单元的输出端连接，所述相位滞后阻抗变换网络的输出端与所述威尔金森功率合成器的一个输入端连接；所述相位超前阻抗变换网络的输入端与所述第二功率级单元的输出端连接，所述相位超前阻抗变换网络的输出端与所述威尔金森功率合成器的另一个输入端连接。
  6. 如权利要求5所述的平衡式射频功率放大器，其特征在于：
    所述相位滞后阻抗变换网络包括第一电感、第一可变电容和第二可变电容；所述第一电感的一端作为所述相位滞后阻抗变换网络的输入端，用于与所述第一功率级单元的输出端连接；所述第一电感的另一端分别与所述第一可变电容和所述第二可变电容的一端连接，所述第一可变电容的另一端接地，所述第二可变电容的另一端作为所述相位滞后阻抗变换网络的输出端，用于与所述威尔金森功率合成器的一个输入端连接。
  7. 如权利要求6所述的平衡式射频功率放大器，其特征在于：
    所述相位超前阻抗变换网络包括第三可变电容和第二电感；所述第三可变电容的一端作为所述相位超前阻抗变换网络的输入端，用于与所述第二功率级单元的输出端连接；所述第三可变电容的另一端作为所述相位超前阻抗变换网络的输出端，用于分别与所述第二电感的一端和所述威尔金森功率合成器的另一个输入端连接，所述第二电感的另一端接地。
  8. 如权利要求7所述的平衡式射频功率放大器，其特征在于：
    所述威尔金森功率合成器包括第四可变电容、可变电阻、第五可变电容、第三电感、第四电感和第六可变电容；所述第四可变电容的一端作为所述威尔金森功率合成器的一个输入端，用于分别与所述相 位滞后阻抗变换网络的输出端、所述可变电阻的另一端和所述第三电感的一端连接；所述可变电阻的一端作为所述威尔金森功率合成器的另一个输入端，用于分别与所述相位超前阻抗变换网络的输出端、所述第五可变电容的一端和所述第四电感的一端连接；所述第三电感和第四电感的另一端作为所述可调90度功合器单元的输出端，用于分别与所述第六可变电容的一端、所述开关模组单元的输入端连接；所述第四可变电容、所述第五可变电容和所述第六可变电容的另一端分别接地。
  9. 如权利要求8所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元中的每个可变电容由电容和n路开关电容组并联组成；或者所述可调90度功合器单元中的每个所述可变电容由n路所述开关电容组并联组成，所述n为正整数。
  10. 如权利要求9所述的平衡式射频功率放大器，其特征在于：
    在n路所述开关电容组中，每组开关电容由一个电容与一个开关串联组成，并且，每组所述开关电容中的开关分别与所述控制单元连接，所述控制单元闭合或打开n路所述开关电容组中特定数量的开关，得到相应的开关电容的电容值。
  11. 如权利要求8所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元中的每个可变电容由电容和n路开关电容组串联组成；或者所述可调90度功合器单元中的每个可变电容由n路开关电容组串联组成。
  12. 如权利要求11所述的平衡式射频功率放大器，其特征在于：
    在n路所述开关电容组中，每组开关电容由一个电容与一个开关并联组成，并且，每组所述开关电容中的开关分别与所述控制单元连接，所述控制单元闭合或打开n路所述开关电容组中特定数量的开关，得到相应的开关电容的电容值。
  13. 如权利要求8所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元中的每个可变电阻由电阻和n路开关电阻组并联组成；或者所述可调90度功合器单元中的每个可变电阻由n路开关电阻组并联组成。
  14. 如权利要求13所述的平衡式射频功率放大器，其特征在于：
    在n路所述开关电阻组中，每组开关电阻由一个电阻与一个开关串联组成，并且，每组所述开关电阻中的开关分别与控制单元连接，所述控制单元闭合或打开n路所述开关电阻组中特定数量的开关，得到相应的开关电阻的电阻值。
  15. 如权利要求8所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元中的每个可变电阻由电阻和n路开关电阻组串联组成；或者所述可调90度功合器单元中的每个可变电阻由n路开关电阻组串联组成。
  16. 如权利要求15所述的平衡式射频功率放大器，其特征在于：
    在n路所述开关电阻组中，每组开关电阻由一个电阻与一个开关并联组成，并且，每组所述开关电阻中的开关分别与控制单元连接，所述控制单元闭合或打开n路所述开关电阻组中特定数量的开关，得到相应的开关电阻的电阻值。
  17. 如权利要求1或2所述的平衡式射频功率放大器，其特征在于：
    所述90度功分器单元中设置有第一匹配网络，用于实现阻抗匹配，并参与配合所述90度功分器单元将射频输入信号分成相位差为90度的两路等幅射频输入信号。
  18. 如权利要求1或2所述的平衡式射频功率放大器，其特征在于：
    所述可调90度功合器单元中设置有第二匹配网络，用于实现阻抗匹配，并参与配合所述可调90度功合器单元使得两路等幅射频输入信号在不同频率处的相位差和幅度差最小。
  19. 如权利要求18所述的平衡式射频功率放大器，其特征在于：
    采用不可调结构的90度功合器单元替代所述可调90度功合器单元；所述不可调结构的90度功合器为电感、电容和电阻器件搭建的匹配网络和移相网络；或者所述不可调结构的90度功合器为金属耦合器件搭建的阻抗和相位变换网络；或者所述不可调结构的90度功合器为传输线网络搭建的阻抗和相位变换网络。
  20. 如权利要求3所述的平衡式射频功率放大器，其特征在于：
    所述开关模组单元包括n组收发开关，每一组所述收发开关的公 共端分别与所述可调90度功合器单元的输出端连接，每一组所述收发开关的一个输出端分别与对应的射频发射路径连接，每一组收发开关的另一个输出端分别与对应的射频接收路径连接。
  21. 一种集成电路芯片，其特征在于所述集成电路芯片中包括权利要求1～20中任意一项所述的平衡式射频功率放大器。
  22. 一种通信终端，其特征在于所述通信终端中包括权利要求1～20中任意一项所述的平衡式射频功率放大器。
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