WO2003096534A1 - Procede et dispositif de mise au point d'un dispositif de reglage de la qualite sonore, programme de mise au point d'un dispositif de reglage de la qualite sonore et dispositif de reglage de la qualite sonore - Google Patents

Procede et dispositif de mise au point d'un dispositif de reglage de la qualite sonore, programme de mise au point d'un dispositif de reglage de la qualite sonore et dispositif de reglage de la qualite sonore Download PDF

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Publication number
WO2003096534A1
WO2003096534A1 PCT/JP2003/005263 JP0305263W WO03096534A1 WO 2003096534 A1 WO2003096534 A1 WO 2003096534A1 JP 0305263 W JP0305263 W JP 0305263W WO 03096534 A1 WO03096534 A1 WO 03096534A1
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Prior art keywords
numerical sequence
filter
input
function
sound quality
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PCT/JP2003/005263
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English (en)
Japanese (ja)
Inventor
Yukio Koyanagi
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Neuro Solution Corp.
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Publication date
Application filed by Neuro Solution Corp. filed Critical Neuro Solution Corp.
Priority to JP2004504382A priority Critical patent/JPWO2003096534A1/ja
Publication of WO2003096534A1 publication Critical patent/WO2003096534A1/fr
Priority to US10/979,733 priority patent/US7400676B2/en

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    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
    • G10L19/26—Pre-filtering or post-filtering
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H17/00—Networks using digital techniques
    • H03H17/02—Frequency selective networks
    • H03H17/06—Non-recursive filters

Definitions

  • the present invention relates to a method and a device for designing a sound quality adjusting device, a program for designing a sound quality adjusting device, and a sound quality adjusting device.
  • the present invention relates to emphasizing or de-emphasizing a desired frequency band of an audio signal by digital signal processing. It is suitable for use in a method of designing a device (equalizer) for improving sound quality. Background technology ''
  • the input audio signal is passed through a low-pass filter and a high-pass filter, and the gain of the output signal of each filter and the input audio signal are controlled and all are added.
  • the gain for each filter output and the gain for the input audio signal it is possible to arbitrarily emphasize sound in a desired frequency band.
  • the gain for the output signal of the low-pass filter may be increased.
  • the gain for the output signal of the high-pass filter may be increased.
  • IIR Infinite
  • FIR Finite Impulse Response: finite-length impulse response
  • the FIR filter has the following advantages. First, the circuit is always stable because the pole of the transfer function of the FIR filter is only at the origin of the z-plane. Secondly, these IIR filters and FIR filters that can achieve perfectly accurate linear phase characteristics are based on single-pass filters, such as high-pass filters, band-pass filters, and band-elimination filters. Other filters are derived by performing processing such as frequency conversion from a low-pass filter.
  • a convolution operation using a window function, Chebyshev approximation, or the like is performed based on a ratio between the sampling frequency and the power-off frequency, thereby obtaining a transfer function of the filter. Is calculated, and it is further replaced with frequency components.
  • the above-mentioned conventional filter design method of the sound quality adjustment device requires a high degree of specialized knowledge such as frequency conversion, and has a problem that the sound quality adjustment device cannot be easily designed.
  • the calculation of frequency conversion using a window function or Chebyshev approximation is very complicated. Therefore, if this is realized by software, the processing load becomes heavy, and if it is realized by hardware, the circuit scale becomes large.
  • the present invention has been made to solve such a problem, and an object of the present invention is to enable a sound quality adjustment device using a FIR digital filter to be simply designed. Disclosure of the invention
  • a desired frequency characteristic is Input a numerical sequence or function to represent, perform an inverse Fourier transform of the input numerical sequence or function, extract the resulting real term, and apply the first half and second half to the numerical sequence consisting of the extracted real number term. And a rounding process that multiplies the result by 2 n (where n is a natural number), rounds off the decimal point, and then multiplies the result by 1/2 ”.
  • the numerical sequence obtained as described above is determined as a filter coefficient group of the first filter constituting the sound quality adjustment device.
  • a predetermined operation is performed on the input numerical sequence or function as described above, and the result is subjected to inverse Fourier transform, rearrangement processing, and rounding processing, whereby the first reference is obtained with the gain reference value as an axis.
  • a filter coefficient group of a second filter having frequency characteristics symmetric to those of the first filter is obtained.
  • a numerical sequence or function representing a desired frequency characteristic which has a number of data points larger than the number of taps of the digital filter, is input, and the input numerical sequence or function is input.
  • Inverse Fourier transform of the function to extract the real term of the result, and to rearrange the former half and the latter half of the numerical sequence consisting of the extracted real term, and the numerical value consisting of the above real term
  • a process of multiplying the sequence by a predetermined window function is performed, and the obtained numerical sequence is determined as a filter coefficient group of a first filter constituting the sound quality adjustment device.
  • the first filter is performed with the reference value as an axis.
  • a filter coefficient group of the second filter having frequency characteristics symmetrical to the evening is obtained.
  • FIG. 1 shows a processing procedure of a design method of a sound quality adjustment device according to the present embodiment. It is a flow chart.
  • FIG. 2 is a flowchart showing a processing procedure of a digital file design method according to the present embodiment.
  • FIG. 3 is a diagram showing an example of the frequency characteristics of the low-pass filter to be designed.
  • FIG. 4 is a diagram showing an example of the frequency characteristics of the eight-pass filter to be designed.
  • FIG. 5 is a diagram showing the input in step S 11 of FIG.
  • FIG. 6 is a diagram showing an example of desired frequency characteristics to be obtained.
  • FIG. 6 is a diagram showing the relationship between the input data length m and the maximum frequency error when designing a FIR filter for a speech signal having a sampling frequency of 44.1 kHz.
  • FIG. 7 is a diagram for explaining the rearrangement process in step S13 of FIG.
  • FIG. 8 is a diagram showing a filter coefficient group obtained by applying the design method of the present embodiment from a numerical sequence representing a desired frequency characteristic.
  • FIG. 9 is a diagram showing coefficients used when obtaining a filter coefficient group of the sound quality adjustment device in units of 2 dB as shown in FIGS. 3 and 4.
  • FIG. 10 is a diagram showing the overall configuration of the sound quality adjusting device according to the present embodiment.
  • FIG. 11 is a diagram showing the configuration of the first mouth-pass filter shown in FIG.
  • FIG. 12 is a diagram showing a configuration of the second low-pass filter shown in FIG.
  • FIG. 13 is a diagram showing a configuration of the first high-pass filter shown in FIG.
  • FIG. 14 is a diagram showing a configuration of the second high-pass filter shown in FIG.
  • FIG. 15 is a diagram illustrating a configuration of the signal processing unit illustrated in FIG. 10. BEST MODE FOR CARRYING OUT THE INVENTION
  • FIG. 1 is a flowchart showing a processing procedure of a design method of a sound quality adjustment device according to the present embodiment.
  • FIG. 2 is a flowchart showing a processing procedure of a design method of a digital filter constituting the sound quality adjustment device of the present embodiment.
  • FIG. 4 and FIG. 4 are diagrams showing frequency characteristics of the sound quality adjustment device to be designed. In this frequency characteristic, both the frequency axis (horizontal axis) and the gain axis (vertical axis) are on a logarithmic scale.
  • the sound quality adjustment device designed in the present embodiment is a type that performs a one-pass filter process or a high-pass filter process on an input audio signal, controls the gain of the output signal of each filter and the input audio signal, and adds them all together. belongs to. Therefore, the design of this sound quality adjustment device is achieved by designing a low-pass filter and a high-pass filter.
  • the filter designed here is a type of FIR filter that has a delay line with taps composed of a plurality of delay units, multiplies the signal of each tap by a given filter coefficient group, and then adds and outputs the result.
  • the impulse response represented by the finite time length is used as the filter coefficient as it is. Therefore, designing an FIR filter means determining a group of filter coefficients so as to obtain the desired frequency characteristics.
  • a first low-pass filter (BASS 1) having a basic frequency characteristic of a bass is designed (step S 1). No. As shown in FIG. 3, the low-pass filter 1 has a maximum amplitude (12 dB) in the positive direction from the gain reference value 1 (0 dB).
  • a first low-pass filter is designed according to the procedure shown in FIG. That is, first, a numerical sequence representing a waveform having a desired frequency characteristic is input (step S11). At this time, it is preferable that the numerical sequence to be input has as much data as possible. Originally, to construct an ideal filter, an infinite number of filter coefficients and an infinite number of filter taps were necessary.
  • the number of input data corresponding to the number of filter coefficients it is preferable to increase the number of input data corresponding to the number of filter coefficients to such an extent that the frequency error falls within a required range. At least, enter a numerical sequence so that the number of data is greater than the number of filter coefficients to be obtained (the number of taps in the digital filter).
  • the frequency-gain characteristics of a filter with the logarithmic scale gain standardized by "1" are drawn and converted into numerical data.
  • the input data should be symmetric about the center of the sampling frequency.
  • the input data length (the length of the graph, that is, the number of numerical sequences) m is a value that falls within the required range of the frequency error, and is 2 to simplify the inverse FFT processing in step S12. k .
  • the relationship between the input data length m and the maximum frequency error is as shown in FIG.
  • the maximum frequency error referred to here corresponds to the frequency per graduation of daraf, and is obtained by the calculation of 44. l KHz z Zm.
  • a frequency characteristic corresponding to a low-pass filter having an input data length m of 5 12 is shown.
  • individual numerical values may be directly input, or a desired frequency characteristic may be input on a two-dimensional input coordinate for representing the frequency-gain characteristic.
  • a waveform may be drawn, and the drawn waveform may be replaced with a numerical value sequence corresponding to the drawn waveform.
  • data input can be performed while confirming a desired frequency characteristic as an image, which makes it easy to intuitively input a data representing the desired frequency characteristic.
  • a two-dimensional plane representing frequency-gain characteristics is displayed on a display screen of a combination display, and a waveform of a desired frequency characteristic is drawn on the two-dimensional plane using a GUI (Graphical User Interface) or the like.
  • GUI Graphic User Interface
  • a method of converting it into numerical data there is a method of converting it into numerical data.
  • a pointing device such as a digitizer or a plotter may be used.
  • the method described here is merely an example, and a numerical sequence may be input by other methods.
  • the desired frequency characteristic is input as a numerical sequence here, it may be input as a function representing the waveform of the frequency characteristic.
  • the input frequency characteristic is subjected to an inverse Fourier transform (inverse FFT) as a transfer function, and a real term of the result is extracted (step S12).
  • inverse FFT inverse Fourier transform
  • a waveform having a frequency-gain characteristic corresponding to the numerical sequence is obtained. Therefore, if a numerical sequence or function representing the waveform of the desired frequency-gain characteristic is input and inverse FFT is performed to extract the real term thereof, it is necessary to realize the frequency-gain characteristic.
  • the original numeric sequence is obtained. This numerical sequence corresponds to the filter coefficient group to be obtained.
  • the numerical sequence itself obtained by the inverse FFT is not arranged in the order that can be used as it is as the filter coefficient group. That is, for any type of digital filter, the sequence of filter coefficient values has the highest median value. It has the symmetry that the value gradually decreases while repeating the amplitude as it moves away from the center. In contrast, the numerical sequence obtained by the inverse FFT has the smallest median value and the largest value at both ends.
  • the numerical sequence is divided into a first half and a second half, and the numerical sequence is rearranged so that the median of the numerical sequence obtained by the inverse FFT is located at both ends (step S13). That is, as shown in FIG. 7, the value of the 0th clock is changed to the value of the 256th clock (hereinafter, referred to as 0 ⁇ 256), 1 ⁇ 257, 2 ⁇ 258, , 2 5 5 ⁇ 5 1 1, 2 5 6 ⁇ 0, 2 5 7 ⁇ 1,--By sorting as 5 1 1 ⁇ 2 5 5, the median becomes the maximum value, To be.
  • step S14 a windowing operation is further performed in this embodiment (step S14).
  • the number of input data is increased to such an extent that an error from a desired frequency characteristic falls within a required range.
  • This number of input data corresponds to the number of filter coefficients. Therefore, if a numerical sequence obtained by processing such as inverse FFT from this input data is used as it is as a filter coefficient group, the number of taps in the digital filter becomes very large, and the circuit scale becomes large. . Therefore, the number of taps is reduced to the required number by performing windowing operation.
  • the window functions used at this time include various functions such as a rectangular window, a Hamming window, a Hanning window, and a Haatlet window.
  • any window function may be applied, it is particularly preferable to use a Hanning window.
  • the Hanning window is a function in which the values at both ends of the window are 0, and the values gradually decrease from the median toward both ends. For example, if you use a square window, Although it is forcibly cut off to a finite number, ringing (undulation phenomenon) occurs on the filter characteristics. On the other hand, if the filter coefficient does not stop at a finite value but transitions smoothly to 0, the occurrence of ringing can be suppressed.
  • the width of the window used at this time must be determined in relation to the amount of attenuation of the input data.
  • the width up to the termination of the Hanning window is, for example, 64.
  • step S 14 the central part of the numerical sequence (516 data strings) obtained by the permutation is multiplied by the Hanning window (64 data strings) having a width of 64. At this time, all coefficients outside the Hanning window are calculated as zero.
  • the numerical sequence obtained by such a windowing operation can be used as it is as a filter coefficient group.
  • the filter coefficient group obtained by inverse FFT and windowing operation is a set of complex and random values with an extremely large number of digits below the decimal point. Therefore, if this numerical sequence is used directly as a filter coefficient group, the number of multipliers required for the digital filter becomes enormous, which is not practical.
  • all filter coefficients have an integer multiple of 1 Z 2 ". Therefore, the integer multiple of the signal from each tap of the digital filter is calculated. It is possible to configure a digital filter so that multiplication is performed individually, all multiplication outputs are added, and then multiplied by 12 "at a time.
  • the integer multiple can be represented by binary addition, such as 2 1 + 2 '' + ⁇ ⁇ ⁇ (where i and j are arbitrary integers).
  • the number of multipliers used in the entire digital filter can be greatly reduced, and the configuration can be simplified. Also, since the numerical sequence obtained by the inverse FFT is rounded after multiplying by 2 n , the rounding error can be reduced as compared with the case where the decimal part of the numerical sequence is simply rounded. As a result, the filter coefficient group can be simplified without lowering the accuracy of the filter characteristics.
  • a numerical sequence obtained by such a rounding operation is finally determined as a filter coefficient group.
  • the processes in steps S13 to S15 do not necessarily need to be performed in this order, and it is sufficient if the rounding operation is performed at least after the windowing operation.
  • a windowing operation may be performed before sorting. In this case, multiply the Hanning window so that the coefficient value at both ends of the window is "1" and the coefficient value at the center of the window is "0".
  • the filter coefficient group (64 filter coefficients) obtained in this way realizes almost exactly the frequency characteristics of the input data as shown in FIG.
  • the phase characteristics are linear and stable.
  • the second low-pass filter has a maximum amplitude ( ⁇ 12 dB) in a negative direction from the gain reference value 1 (0 dB).
  • the axis has characteristics that are line-symmetric with the first low-pass filter.
  • the design of the second mouth-pass filter is also performed according to the procedure shown in FIG.
  • step SI1 the input data of the second mouth-to-pass filter is obtained by substituting the numerical sequence input for designing the first mouth-to-pass filter into equation (1). Then, by performing the same processing as in steps S12 to S15 on the input data, a filter coefficient group of the second mouth-pass filter is obtained.
  • the first and second high-pass filters are designed in the same manner as the above-described first and second low-pass filter designing methods (steps S 3 and S 4).
  • the width of the Hanning window used in the windowing operation in step S14 is set to 8.
  • the attenuation of the real term resulting from the inverse FFT is large, so that the window width can be reduced to eight.
  • the first and second high-pass filters are designed after the first and second single-pass filters are designed. However, the order may be reversed.
  • the first low-pass filter or the After designing the first high-pass filter, the second low-pass filter or the second high-pass filter is designed, but this order may be reversed.
  • FIG. 8 is a diagram showing the filter coefficient groups of LPF 1,2 and HPF 1,2 obtained as described above.
  • a filter block including one delay line and four FIR filters for delaying an input audio signal can be formed.
  • the filter coefficient group of the sound quality adjustment device in 2 dB units as shown in FIGS. 3 and 4 is obtained. Can be obtained.
  • FIG. 10 is a diagram showing an overall configuration example of a sound quality adjustment device using the four filter blocks shown in FIG.
  • reference numerals 11 to 14 denote first and second single-pass filters and first and second high-pass filters designed by the procedure shown in FIGS. 1 and 2 described above.
  • the first mouth-pass filter 11 also serves as a delay line for the input audio signal.
  • Reference numeral 15 denotes a signal processing unit, which inputs signals (one delay line output and four filter outputs) output from each of the filters 11 to 14 and outputs the signals by controlling their gain. .
  • FIGS. 11 to 14 are diagrams showing the internal configuration of the above four filters 11 to 14. These filters 11 to 14 include multiple cascaded filters.
  • the input signal is sequentially delayed by one clock CK by the D-type flip-flop. Then, a signal extracted from the output tap of each D-type flip-flop is multiplied by an integer value obtained by multiplying the filter coefficient by 248 by each of a plurality of coefficients, and all of the multiplication results are obtained.
  • the output is added by multiple adders.
  • the first mouth-pass filter 11 is also provided with a delay line that allows the input audio signal to pass through a plurality of D-type flip-flops.
  • FIG. 15 is a diagram showing the internal configuration of the signal processing unit 15.
  • reference numeral 21 denotes a first decoder, which inputs and decodes gain control signals of the first and second single-pass filters 11 1 and 12.
  • Reference numerals 22 to 24 denote a plurality of switches, which perform a switching operation based on the decoding result of the first decoder 21. By this switching operation, one of the output signals of the first and second low-pass filters 11 and 12 is selected, and the gain thereof is controlled.
  • Reference numeral 25 denotes a divider, which divides the signal passed through the switch 22 from the first and second single-pass filters 11 and 12 by 2048. As shown in FIGS. 11 and 12, inside the first and second mouth-pass filters 1 1 and 1 2, an integer value obtained by multiplying the filter coefficient group shown in FIG. Is multiplied by each tap output. Therefore, in order to return the amplitude to a correct value, the filter output is divided by 204 in the divider 25.
  • Reference numeral 26 denotes a plurality of coefficient units, which multiply the signal passed through the divider 25 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the decoding result by the first decoder 21.
  • Reference numeral 31 denotes a second decoder, which inputs and decodes the gain control signals of the first and second high-pass filters 13 and 14.
  • Reference numerals 32 to 34 denote a plurality of switches, which are switched based on the decoding result of the second decoder 31. Switching operation is performed. By this switching operation, one of the output signals of the first and second high-pass filters 13 and 14 is selected, and the gain thereof is controlled.
  • Reference numeral 35 denotes a divider, which divides the signal passed through the switch 32 from the first and second high-pass filters 13 and 1 by 2048. As shown in FIGS. 13 and 14, inside the first and second high-pass filters 13 and 14, the integer values obtained by multiplying the filter coefficient group shown in FIG. Multiplies the tap output. Therefore, in order to return the amplitude to the correct value, the filter output is divided by 204 in the divider 35.
  • Reference numeral 36 denotes a plurality of coefficient units, which multiply the signal passing through the divider 35 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the result of decoding by the second decoder 31.
  • 4 1 is an adder, which adds an audio signal input from the delay line of the first one-pass filter 11 to one of the outputs of the first and second low-pass filters 11 1 and 12.
  • Reference numeral 2 denotes an adder, which adds an audio signal input from the delay line of the first single-pass filter 11 to one of the outputs of the first and second high-pass filters 13 and 14.
  • An adder 43 adds the outputs of the adders 41 and 42 and finally outputs a sound signal whose sound quality has been adjusted.
  • An apparatus for implementing the above-described method for designing a sound quality adjustment apparatus according to the present embodiment can be implemented by any of a hardware configuration, a DSP, and software.
  • the design device of the present embodiment is actually configured by a computer CPU or MPU, RAM, ROM, etc., and a program stored in RAM, ROM, a hard disk, or the like is used. Realized by working Cut.
  • the present invention can be realized by recording a program that causes a computer to perform the functions of the present embodiment on a recording medium such as a CD-R ⁇ M and reading the program into the computer.
  • a recording medium for recording the above program a flexible disk, a hard disk, a magnetic tape, an optical disk, a magneto-optical disk, a DVD, a nonvolatile memory card, and the like can be used in addition to the CD-ROM.
  • it can be realized by downloading the above program in the evening via a network such as the Internet.
  • a numerical value sequence representing a waveform of a desired frequency characteristic is input as an image, and this is subjected to an inverse Fourier transform to thereby obtain a filter coefficient of each filter constituting the sound quality adjustment device. Since the groups are determined, the coefficients of the FIR digital filter that achieve the desired frequency characteristics can be easily determined without special mathematical or electrical engineering knowledge.
  • the filter coefficient group without reducing the accuracy of the filter.
  • Component multiplier (divide ) Can be greatly reduced.
  • the result of the inverse Fourier transform is multiplied by a window function of a required length, so that the input data length is increased to reduce the frequency error, and at the same time, The number of filter coefficients (the number of taps of the digital filter) can be reduced. This simplifies the configuration of the sound quality adjustment device to be designed, and achieves desired frequency characteristics with high accuracy.
  • a filter coefficient group of the first filter is obtained by inputting a waveform of a desired frequency characteristic as a numerical sequence or a function and performing processing such as inverse Fourier transform on the waveform.
  • processing such as inverse Fourier transform on the waveform.
  • a special rounding operation is performed on the numerical sequence obtained by the inverse Fourier transform, so that the filter coefficient group to be obtained is simplified without lowering the accuracy of the filter characteristics.
  • the number of multipliers used in the filter component can be greatly reduced. This makes it possible to simply design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale.
  • the windowing operation is performed on the result of the inverse Fourier transform, so that the first input numerical sequence is lengthened to reduce the frequency error, and at the same time, the filter coefficient is reduced.
  • the number (the number of taps in the digital filter) can be reduced, and the configuration of the digital filter to be designed can be simplified. As a result, it is possible to easily design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale.
  • the present invention is useful for easily designing a sound quality adjustment device using a FIR digital filter.

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • Computational Linguistics (AREA)
  • Signal Processing (AREA)
  • Health & Medical Sciences (AREA)
  • Audiology, Speech & Language Pathology (AREA)
  • Human Computer Interaction (AREA)
  • Acoustics & Sound (AREA)
  • Multimedia (AREA)
  • Computer Hardware Design (AREA)
  • Mathematical Physics (AREA)
  • Tone Control, Compression And Expansion, Limiting Amplitude (AREA)

Abstract

Une forme d'onde ayant la caractéristique de fréquence souhaitée est introduite sous la forme d'une chaîne de valeurs numériques, puis elle est soumise à une transformation de Fourrier rapide (FFT) inverse pour obtenir un groupage de coefficients de filtre. Ainsi, sans posséder le savoir de l'expert, et simplement en introduisant une forme d'onde ayant la caractéristique de fréquence souhaitée, telle qu'une image, on peut facilement mettre au point un premier filtre à réponse impulsionnelle finie (filtre FIR) qui servira de dispositif de réglage de la qualité sonore. Par ailleurs, en soumettant la chaîne de valeurs numériques introduite à un calcul prédéterminé et en soumettant le résultat à une FFT inverse, on peut facilement mettre au point un second filtre FIR qui possède une caractéristique de fréquence symétrique au premier filtre FIR par rapport à la valeur de référence de gain comme axe.
PCT/JP2003/005263 2002-05-09 2003-04-24 Procede et dispositif de mise au point d'un dispositif de reglage de la qualite sonore, programme de mise au point d'un dispositif de reglage de la qualite sonore et dispositif de reglage de la qualite sonore WO2003096534A1 (fr)

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JP2004504382A JPWO2003096534A1 (ja) 2002-05-09 2003-04-24 音質調整装置の設計方法および設計装置、音質調整装置設計用プログラム、音質調整装置
US10/979,733 US7400676B2 (en) 2002-05-09 2004-11-03 Tone quality adjustment device designing method and designing device, tone quality adjustment device designing program, and tone quality adjustment device

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JP2002-134572 2002-05-09
JP2002134572 2002-05-09

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WO2011001589A1 (fr) * 2009-06-29 2011-01-06 三菱電機株式会社 Dispositif de traitement de signal audio

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JPH1079644A (ja) * 1996-09-05 1998-03-24 New Japan Radio Co Ltd デジタルフィルタ
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JPS6196817A (ja) * 1984-10-17 1986-05-15 Sharp Corp フイルタ−
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JPH05259783A (ja) * 1992-03-11 1993-10-08 Sansui Electric Co Ltd トーンコントロール回路
JPH0766685A (ja) * 1993-08-24 1995-03-10 Pioneer Electron Corp ディジタル・グラフィックイコライザ
JP2001273278A (ja) * 1993-12-14 2001-10-05 Masaharu Ishii 適性化装置および適性化方法
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WO2011001589A1 (fr) * 2009-06-29 2011-01-06 三菱電機株式会社 Dispositif de traitement de signal audio
JP5265008B2 (ja) * 2009-06-29 2013-08-14 三菱電機株式会社 オーディオ信号処理装置
US9299362B2 (en) 2009-06-29 2016-03-29 Mitsubishi Electric Corporation Audio signal processing device

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