WO2003096534A1 - Tone quality adjustment device designing method and designing device, tone quality adjustment device designing program, and tone quality adjustment device - Google Patents

Tone quality adjustment device designing method and designing device, tone quality adjustment device designing program, and tone quality adjustment device Download PDF

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Publication number
WO2003096534A1
WO2003096534A1 PCT/JP2003/005263 JP0305263W WO03096534A1 WO 2003096534 A1 WO2003096534 A1 WO 2003096534A1 JP 0305263 W JP0305263 W JP 0305263W WO 03096534 A1 WO03096534 A1 WO 03096534A1
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Prior art keywords
numerical sequence
filter
input
function
sound quality
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PCT/JP2003/005263
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French (fr)
Japanese (ja)
Inventor
Yukio Koyanagi
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Neuro Solution Corp.
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Application filed by Neuro Solution Corp. filed Critical Neuro Solution Corp.
Priority to JP2004504382A priority Critical patent/JPWO2003096534A1/en
Publication of WO2003096534A1 publication Critical patent/WO2003096534A1/en
Priority to US10/979,733 priority patent/US7400676B2/en

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    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
    • G10L19/26—Pre-filtering or post-filtering
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H17/00—Networks using digital techniques
    • H03H17/02—Frequency selective networks
    • H03H17/06—Non-recursive filters

Definitions

  • the present invention relates to a method and a device for designing a sound quality adjusting device, a program for designing a sound quality adjusting device, and a sound quality adjusting device.
  • the present invention relates to emphasizing or de-emphasizing a desired frequency band of an audio signal by digital signal processing. It is suitable for use in a method of designing a device (equalizer) for improving sound quality. Background technology ''
  • the input audio signal is passed through a low-pass filter and a high-pass filter, and the gain of the output signal of each filter and the input audio signal are controlled and all are added.
  • the gain for each filter output and the gain for the input audio signal it is possible to arbitrarily emphasize sound in a desired frequency band.
  • the gain for the output signal of the low-pass filter may be increased.
  • the gain for the output signal of the high-pass filter may be increased.
  • IIR Infinite
  • FIR Finite Impulse Response: finite-length impulse response
  • the FIR filter has the following advantages. First, the circuit is always stable because the pole of the transfer function of the FIR filter is only at the origin of the z-plane. Secondly, these IIR filters and FIR filters that can achieve perfectly accurate linear phase characteristics are based on single-pass filters, such as high-pass filters, band-pass filters, and band-elimination filters. Other filters are derived by performing processing such as frequency conversion from a low-pass filter.
  • a convolution operation using a window function, Chebyshev approximation, or the like is performed based on a ratio between the sampling frequency and the power-off frequency, thereby obtaining a transfer function of the filter. Is calculated, and it is further replaced with frequency components.
  • the above-mentioned conventional filter design method of the sound quality adjustment device requires a high degree of specialized knowledge such as frequency conversion, and has a problem that the sound quality adjustment device cannot be easily designed.
  • the calculation of frequency conversion using a window function or Chebyshev approximation is very complicated. Therefore, if this is realized by software, the processing load becomes heavy, and if it is realized by hardware, the circuit scale becomes large.
  • the present invention has been made to solve such a problem, and an object of the present invention is to enable a sound quality adjustment device using a FIR digital filter to be simply designed. Disclosure of the invention
  • a desired frequency characteristic is Input a numerical sequence or function to represent, perform an inverse Fourier transform of the input numerical sequence or function, extract the resulting real term, and apply the first half and second half to the numerical sequence consisting of the extracted real number term. And a rounding process that multiplies the result by 2 n (where n is a natural number), rounds off the decimal point, and then multiplies the result by 1/2 ”.
  • the numerical sequence obtained as described above is determined as a filter coefficient group of the first filter constituting the sound quality adjustment device.
  • a predetermined operation is performed on the input numerical sequence or function as described above, and the result is subjected to inverse Fourier transform, rearrangement processing, and rounding processing, whereby the first reference is obtained with the gain reference value as an axis.
  • a filter coefficient group of a second filter having frequency characteristics symmetric to those of the first filter is obtained.
  • a numerical sequence or function representing a desired frequency characteristic which has a number of data points larger than the number of taps of the digital filter, is input, and the input numerical sequence or function is input.
  • Inverse Fourier transform of the function to extract the real term of the result, and to rearrange the former half and the latter half of the numerical sequence consisting of the extracted real term, and the numerical value consisting of the above real term
  • a process of multiplying the sequence by a predetermined window function is performed, and the obtained numerical sequence is determined as a filter coefficient group of a first filter constituting the sound quality adjustment device.
  • the first filter is performed with the reference value as an axis.
  • a filter coefficient group of the second filter having frequency characteristics symmetrical to the evening is obtained.
  • FIG. 1 shows a processing procedure of a design method of a sound quality adjustment device according to the present embodiment. It is a flow chart.
  • FIG. 2 is a flowchart showing a processing procedure of a digital file design method according to the present embodiment.
  • FIG. 3 is a diagram showing an example of the frequency characteristics of the low-pass filter to be designed.
  • FIG. 4 is a diagram showing an example of the frequency characteristics of the eight-pass filter to be designed.
  • FIG. 5 is a diagram showing the input in step S 11 of FIG.
  • FIG. 6 is a diagram showing an example of desired frequency characteristics to be obtained.
  • FIG. 6 is a diagram showing the relationship between the input data length m and the maximum frequency error when designing a FIR filter for a speech signal having a sampling frequency of 44.1 kHz.
  • FIG. 7 is a diagram for explaining the rearrangement process in step S13 of FIG.
  • FIG. 8 is a diagram showing a filter coefficient group obtained by applying the design method of the present embodiment from a numerical sequence representing a desired frequency characteristic.
  • FIG. 9 is a diagram showing coefficients used when obtaining a filter coefficient group of the sound quality adjustment device in units of 2 dB as shown in FIGS. 3 and 4.
  • FIG. 10 is a diagram showing the overall configuration of the sound quality adjusting device according to the present embodiment.
  • FIG. 11 is a diagram showing the configuration of the first mouth-pass filter shown in FIG.
  • FIG. 12 is a diagram showing a configuration of the second low-pass filter shown in FIG.
  • FIG. 13 is a diagram showing a configuration of the first high-pass filter shown in FIG.
  • FIG. 14 is a diagram showing a configuration of the second high-pass filter shown in FIG.
  • FIG. 15 is a diagram illustrating a configuration of the signal processing unit illustrated in FIG. 10. BEST MODE FOR CARRYING OUT THE INVENTION
  • FIG. 1 is a flowchart showing a processing procedure of a design method of a sound quality adjustment device according to the present embodiment.
  • FIG. 2 is a flowchart showing a processing procedure of a design method of a digital filter constituting the sound quality adjustment device of the present embodiment.
  • FIG. 4 and FIG. 4 are diagrams showing frequency characteristics of the sound quality adjustment device to be designed. In this frequency characteristic, both the frequency axis (horizontal axis) and the gain axis (vertical axis) are on a logarithmic scale.
  • the sound quality adjustment device designed in the present embodiment is a type that performs a one-pass filter process or a high-pass filter process on an input audio signal, controls the gain of the output signal of each filter and the input audio signal, and adds them all together. belongs to. Therefore, the design of this sound quality adjustment device is achieved by designing a low-pass filter and a high-pass filter.
  • the filter designed here is a type of FIR filter that has a delay line with taps composed of a plurality of delay units, multiplies the signal of each tap by a given filter coefficient group, and then adds and outputs the result.
  • the impulse response represented by the finite time length is used as the filter coefficient as it is. Therefore, designing an FIR filter means determining a group of filter coefficients so as to obtain the desired frequency characteristics.
  • a first low-pass filter (BASS 1) having a basic frequency characteristic of a bass is designed (step S 1). No. As shown in FIG. 3, the low-pass filter 1 has a maximum amplitude (12 dB) in the positive direction from the gain reference value 1 (0 dB).
  • a first low-pass filter is designed according to the procedure shown in FIG. That is, first, a numerical sequence representing a waveform having a desired frequency characteristic is input (step S11). At this time, it is preferable that the numerical sequence to be input has as much data as possible. Originally, to construct an ideal filter, an infinite number of filter coefficients and an infinite number of filter taps were necessary.
  • the number of input data corresponding to the number of filter coefficients it is preferable to increase the number of input data corresponding to the number of filter coefficients to such an extent that the frequency error falls within a required range. At least, enter a numerical sequence so that the number of data is greater than the number of filter coefficients to be obtained (the number of taps in the digital filter).
  • the frequency-gain characteristics of a filter with the logarithmic scale gain standardized by "1" are drawn and converted into numerical data.
  • the input data should be symmetric about the center of the sampling frequency.
  • the input data length (the length of the graph, that is, the number of numerical sequences) m is a value that falls within the required range of the frequency error, and is 2 to simplify the inverse FFT processing in step S12. k .
  • the relationship between the input data length m and the maximum frequency error is as shown in FIG.
  • the maximum frequency error referred to here corresponds to the frequency per graduation of daraf, and is obtained by the calculation of 44. l KHz z Zm.
  • a frequency characteristic corresponding to a low-pass filter having an input data length m of 5 12 is shown.
  • individual numerical values may be directly input, or a desired frequency characteristic may be input on a two-dimensional input coordinate for representing the frequency-gain characteristic.
  • a waveform may be drawn, and the drawn waveform may be replaced with a numerical value sequence corresponding to the drawn waveform.
  • data input can be performed while confirming a desired frequency characteristic as an image, which makes it easy to intuitively input a data representing the desired frequency characteristic.
  • a two-dimensional plane representing frequency-gain characteristics is displayed on a display screen of a combination display, and a waveform of a desired frequency characteristic is drawn on the two-dimensional plane using a GUI (Graphical User Interface) or the like.
  • GUI Graphic User Interface
  • a method of converting it into numerical data there is a method of converting it into numerical data.
  • a pointing device such as a digitizer or a plotter may be used.
  • the method described here is merely an example, and a numerical sequence may be input by other methods.
  • the desired frequency characteristic is input as a numerical sequence here, it may be input as a function representing the waveform of the frequency characteristic.
  • the input frequency characteristic is subjected to an inverse Fourier transform (inverse FFT) as a transfer function, and a real term of the result is extracted (step S12).
  • inverse FFT inverse Fourier transform
  • a waveform having a frequency-gain characteristic corresponding to the numerical sequence is obtained. Therefore, if a numerical sequence or function representing the waveform of the desired frequency-gain characteristic is input and inverse FFT is performed to extract the real term thereof, it is necessary to realize the frequency-gain characteristic.
  • the original numeric sequence is obtained. This numerical sequence corresponds to the filter coefficient group to be obtained.
  • the numerical sequence itself obtained by the inverse FFT is not arranged in the order that can be used as it is as the filter coefficient group. That is, for any type of digital filter, the sequence of filter coefficient values has the highest median value. It has the symmetry that the value gradually decreases while repeating the amplitude as it moves away from the center. In contrast, the numerical sequence obtained by the inverse FFT has the smallest median value and the largest value at both ends.
  • the numerical sequence is divided into a first half and a second half, and the numerical sequence is rearranged so that the median of the numerical sequence obtained by the inverse FFT is located at both ends (step S13). That is, as shown in FIG. 7, the value of the 0th clock is changed to the value of the 256th clock (hereinafter, referred to as 0 ⁇ 256), 1 ⁇ 257, 2 ⁇ 258, , 2 5 5 ⁇ 5 1 1, 2 5 6 ⁇ 0, 2 5 7 ⁇ 1,--By sorting as 5 1 1 ⁇ 2 5 5, the median becomes the maximum value, To be.
  • step S14 a windowing operation is further performed in this embodiment (step S14).
  • the number of input data is increased to such an extent that an error from a desired frequency characteristic falls within a required range.
  • This number of input data corresponds to the number of filter coefficients. Therefore, if a numerical sequence obtained by processing such as inverse FFT from this input data is used as it is as a filter coefficient group, the number of taps in the digital filter becomes very large, and the circuit scale becomes large. . Therefore, the number of taps is reduced to the required number by performing windowing operation.
  • the window functions used at this time include various functions such as a rectangular window, a Hamming window, a Hanning window, and a Haatlet window.
  • any window function may be applied, it is particularly preferable to use a Hanning window.
  • the Hanning window is a function in which the values at both ends of the window are 0, and the values gradually decrease from the median toward both ends. For example, if you use a square window, Although it is forcibly cut off to a finite number, ringing (undulation phenomenon) occurs on the filter characteristics. On the other hand, if the filter coefficient does not stop at a finite value but transitions smoothly to 0, the occurrence of ringing can be suppressed.
  • the width of the window used at this time must be determined in relation to the amount of attenuation of the input data.
  • the width up to the termination of the Hanning window is, for example, 64.
  • step S 14 the central part of the numerical sequence (516 data strings) obtained by the permutation is multiplied by the Hanning window (64 data strings) having a width of 64. At this time, all coefficients outside the Hanning window are calculated as zero.
  • the numerical sequence obtained by such a windowing operation can be used as it is as a filter coefficient group.
  • the filter coefficient group obtained by inverse FFT and windowing operation is a set of complex and random values with an extremely large number of digits below the decimal point. Therefore, if this numerical sequence is used directly as a filter coefficient group, the number of multipliers required for the digital filter becomes enormous, which is not practical.
  • all filter coefficients have an integer multiple of 1 Z 2 ". Therefore, the integer multiple of the signal from each tap of the digital filter is calculated. It is possible to configure a digital filter so that multiplication is performed individually, all multiplication outputs are added, and then multiplied by 12 "at a time.
  • the integer multiple can be represented by binary addition, such as 2 1 + 2 '' + ⁇ ⁇ ⁇ (where i and j are arbitrary integers).
  • the number of multipliers used in the entire digital filter can be greatly reduced, and the configuration can be simplified. Also, since the numerical sequence obtained by the inverse FFT is rounded after multiplying by 2 n , the rounding error can be reduced as compared with the case where the decimal part of the numerical sequence is simply rounded. As a result, the filter coefficient group can be simplified without lowering the accuracy of the filter characteristics.
  • a numerical sequence obtained by such a rounding operation is finally determined as a filter coefficient group.
  • the processes in steps S13 to S15 do not necessarily need to be performed in this order, and it is sufficient if the rounding operation is performed at least after the windowing operation.
  • a windowing operation may be performed before sorting. In this case, multiply the Hanning window so that the coefficient value at both ends of the window is "1" and the coefficient value at the center of the window is "0".
  • the filter coefficient group (64 filter coefficients) obtained in this way realizes almost exactly the frequency characteristics of the input data as shown in FIG.
  • the phase characteristics are linear and stable.
  • the second low-pass filter has a maximum amplitude ( ⁇ 12 dB) in a negative direction from the gain reference value 1 (0 dB).
  • the axis has characteristics that are line-symmetric with the first low-pass filter.
  • the design of the second mouth-pass filter is also performed according to the procedure shown in FIG.
  • step SI1 the input data of the second mouth-to-pass filter is obtained by substituting the numerical sequence input for designing the first mouth-to-pass filter into equation (1). Then, by performing the same processing as in steps S12 to S15 on the input data, a filter coefficient group of the second mouth-pass filter is obtained.
  • the first and second high-pass filters are designed in the same manner as the above-described first and second low-pass filter designing methods (steps S 3 and S 4).
  • the width of the Hanning window used in the windowing operation in step S14 is set to 8.
  • the attenuation of the real term resulting from the inverse FFT is large, so that the window width can be reduced to eight.
  • the first and second high-pass filters are designed after the first and second single-pass filters are designed. However, the order may be reversed.
  • the first low-pass filter or the After designing the first high-pass filter, the second low-pass filter or the second high-pass filter is designed, but this order may be reversed.
  • FIG. 8 is a diagram showing the filter coefficient groups of LPF 1,2 and HPF 1,2 obtained as described above.
  • a filter block including one delay line and four FIR filters for delaying an input audio signal can be formed.
  • the filter coefficient group of the sound quality adjustment device in 2 dB units as shown in FIGS. 3 and 4 is obtained. Can be obtained.
  • FIG. 10 is a diagram showing an overall configuration example of a sound quality adjustment device using the four filter blocks shown in FIG.
  • reference numerals 11 to 14 denote first and second single-pass filters and first and second high-pass filters designed by the procedure shown in FIGS. 1 and 2 described above.
  • the first mouth-pass filter 11 also serves as a delay line for the input audio signal.
  • Reference numeral 15 denotes a signal processing unit, which inputs signals (one delay line output and four filter outputs) output from each of the filters 11 to 14 and outputs the signals by controlling their gain. .
  • FIGS. 11 to 14 are diagrams showing the internal configuration of the above four filters 11 to 14. These filters 11 to 14 include multiple cascaded filters.
  • the input signal is sequentially delayed by one clock CK by the D-type flip-flop. Then, a signal extracted from the output tap of each D-type flip-flop is multiplied by an integer value obtained by multiplying the filter coefficient by 248 by each of a plurality of coefficients, and all of the multiplication results are obtained.
  • the output is added by multiple adders.
  • the first mouth-pass filter 11 is also provided with a delay line that allows the input audio signal to pass through a plurality of D-type flip-flops.
  • FIG. 15 is a diagram showing the internal configuration of the signal processing unit 15.
  • reference numeral 21 denotes a first decoder, which inputs and decodes gain control signals of the first and second single-pass filters 11 1 and 12.
  • Reference numerals 22 to 24 denote a plurality of switches, which perform a switching operation based on the decoding result of the first decoder 21. By this switching operation, one of the output signals of the first and second low-pass filters 11 and 12 is selected, and the gain thereof is controlled.
  • Reference numeral 25 denotes a divider, which divides the signal passed through the switch 22 from the first and second single-pass filters 11 and 12 by 2048. As shown in FIGS. 11 and 12, inside the first and second mouth-pass filters 1 1 and 1 2, an integer value obtained by multiplying the filter coefficient group shown in FIG. Is multiplied by each tap output. Therefore, in order to return the amplitude to a correct value, the filter output is divided by 204 in the divider 25.
  • Reference numeral 26 denotes a plurality of coefficient units, which multiply the signal passed through the divider 25 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the decoding result by the first decoder 21.
  • Reference numeral 31 denotes a second decoder, which inputs and decodes the gain control signals of the first and second high-pass filters 13 and 14.
  • Reference numerals 32 to 34 denote a plurality of switches, which are switched based on the decoding result of the second decoder 31. Switching operation is performed. By this switching operation, one of the output signals of the first and second high-pass filters 13 and 14 is selected, and the gain thereof is controlled.
  • Reference numeral 35 denotes a divider, which divides the signal passed through the switch 32 from the first and second high-pass filters 13 and 1 by 2048. As shown in FIGS. 13 and 14, inside the first and second high-pass filters 13 and 14, the integer values obtained by multiplying the filter coefficient group shown in FIG. Multiplies the tap output. Therefore, in order to return the amplitude to the correct value, the filter output is divided by 204 in the divider 35.
  • Reference numeral 36 denotes a plurality of coefficient units, which multiply the signal passing through the divider 35 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the result of decoding by the second decoder 31.
  • 4 1 is an adder, which adds an audio signal input from the delay line of the first one-pass filter 11 to one of the outputs of the first and second low-pass filters 11 1 and 12.
  • Reference numeral 2 denotes an adder, which adds an audio signal input from the delay line of the first single-pass filter 11 to one of the outputs of the first and second high-pass filters 13 and 14.
  • An adder 43 adds the outputs of the adders 41 and 42 and finally outputs a sound signal whose sound quality has been adjusted.
  • An apparatus for implementing the above-described method for designing a sound quality adjustment apparatus according to the present embodiment can be implemented by any of a hardware configuration, a DSP, and software.
  • the design device of the present embodiment is actually configured by a computer CPU or MPU, RAM, ROM, etc., and a program stored in RAM, ROM, a hard disk, or the like is used. Realized by working Cut.
  • the present invention can be realized by recording a program that causes a computer to perform the functions of the present embodiment on a recording medium such as a CD-R ⁇ M and reading the program into the computer.
  • a recording medium for recording the above program a flexible disk, a hard disk, a magnetic tape, an optical disk, a magneto-optical disk, a DVD, a nonvolatile memory card, and the like can be used in addition to the CD-ROM.
  • it can be realized by downloading the above program in the evening via a network such as the Internet.
  • a numerical value sequence representing a waveform of a desired frequency characteristic is input as an image, and this is subjected to an inverse Fourier transform to thereby obtain a filter coefficient of each filter constituting the sound quality adjustment device. Since the groups are determined, the coefficients of the FIR digital filter that achieve the desired frequency characteristics can be easily determined without special mathematical or electrical engineering knowledge.
  • the filter coefficient group without reducing the accuracy of the filter.
  • Component multiplier (divide ) Can be greatly reduced.
  • the result of the inverse Fourier transform is multiplied by a window function of a required length, so that the input data length is increased to reduce the frequency error, and at the same time, The number of filter coefficients (the number of taps of the digital filter) can be reduced. This simplifies the configuration of the sound quality adjustment device to be designed, and achieves desired frequency characteristics with high accuracy.
  • a filter coefficient group of the first filter is obtained by inputting a waveform of a desired frequency characteristic as a numerical sequence or a function and performing processing such as inverse Fourier transform on the waveform.
  • processing such as inverse Fourier transform on the waveform.
  • a special rounding operation is performed on the numerical sequence obtained by the inverse Fourier transform, so that the filter coefficient group to be obtained is simplified without lowering the accuracy of the filter characteristics.
  • the number of multipliers used in the filter component can be greatly reduced. This makes it possible to simply design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale.
  • the windowing operation is performed on the result of the inverse Fourier transform, so that the first input numerical sequence is lengthened to reduce the frequency error, and at the same time, the filter coefficient is reduced.
  • the number (the number of taps in the digital filter) can be reduced, and the configuration of the digital filter to be designed can be simplified. As a result, it is possible to easily design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale.
  • the present invention is useful for easily designing a sound quality adjustment device using a FIR digital filter.

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Abstract

A waveform of a desired frequency characteristic is input as a numeric value string and is subjected to reverse FFT so as to obtain a filter coefficient group. Thus, without having any expert knowledge, only by inputting a waveform of a desired frequency characteristic as an image, it is possible to easily design a first FIR filter constituting a tone quality adjustment device. Moreover, by performing a predetermined calculation on the numeric value string input and performing a reverse FFT to the result, it is possible to easily design a second FIR filter having a frequency characteristic symmetric to the first FIR filter with respect to the gain reference value as an axis.

Description

明 細 書 音質調整装置の設計方法および設計装置、 音質調整装置設計用プログラ ム、 音質調整装置 技術分野  Description Design method and design apparatus for sound quality adjustment device, program for sound quality adjustment device design, sound quality adjustment device
本発明は、 音質調整装置の設計方法および設計装置、 音質調整装置設 計用プログラム、 音質調整装置に関し、 特に、 デジタル信号処理によつ て音声信号の所望の周波数帯域を強調あるいは非強調して音質を改善す るための装置 （イコライザ） の設計方法に用いて好適なものである。 背景技術 '  The present invention relates to a method and a device for designing a sound quality adjusting device, a program for designing a sound quality adjusting device, and a sound quality adjusting device. In particular, the present invention relates to emphasizing or de-emphasizing a desired frequency band of an audio signal by digital signal processing. It is suitable for use in a method of designing a device (equalizer) for improving sound quality. Background technology ''
従来、 音声信号を出力する装置において、 出力音声の音質を改善する ための方法として種々のものが提案されている。 その中でも比較的簡単 な方法の 1つに、 入力音声信号に対してローパスフィルタ処理やハイパ スフィルタ処理を施すというものがある。  Conventionally, in a device for outputting an audio signal, various methods have been proposed as methods for improving the sound quality of an output audio. One of the relatively simple methods is to apply low-pass or high-pass filtering to the input audio signal.
この種の音質調整装置では、 入力音声信号をローパスフィルタとハイ パスフィルタとに通し、 各フィルタの出力信号と入力音声信号との利得 を制御して全て合算する。 このとき、 各フィルタ出力に対する利得と入 力音声信号に対する利得とを任意に設定することにより、 所望の周波数 帯域の音を任意に強調することが可能となる。  In this type of sound quality adjustment device, the input audio signal is passed through a low-pass filter and a high-pass filter, and the gain of the output signal of each filter and the input audio signal are controlled and all are added. At this time, by arbitrarily setting the gain for each filter output and the gain for the input audio signal, it is possible to arbitrarily emphasize sound in a desired frequency band.
例えば、 低周波領域の音 （いわゆる低音） を強調したいときは、 ロー パスフィルタの出力信号に対する利得を大きくすれば良い。 また、 高周 波領域の音 （いわゆる高音） を強調したいときは、 ハイパスフィルタの 出力信号に対する利得を大きくすれば良い。  For example, when it is desired to emphasize low-frequency sounds (so-called low-pitched sounds), the gain for the output signal of the low-pass filter may be increased. Also, when it is desired to emphasize the sound in the high frequency region (so-called high sound), the gain for the output signal of the high-pass filter may be increased.
この種の音質調整装置に用いるフィルタとしては、 I I R ( I n f i n i t e Impulse Response： 無限長インパルス応答） フィルタや F I R (Finite Impulse Response： 有限長インパルス応答） フィル夕が多く用いられる 。 このうち F I Rフィルタは、 次のような利点を持つ。 第 1 に、 F I R フィル夕の伝達関数の極は z平面の原点のみにあるため、 回路は常に安 定である。 第 2 に、 完全に正確な直線位相特性を実現することができる これらの I I Rフィルタや F I Rフィルタでは、 基本となるのは口一 パスフィルタであり、 ハイパスフィルタ、 帯域通過フィルタ、 帯域消去 フィルタ等のその他のフィルタは、 ローパスフィルタから周波数変換等 の処理を行う ことによって導かれる。 こ こでの周波数変換処理では、 サ ンプリング周波数と力ッ 卜オフ周波数との比率をもとに、 窓関数やチェ ビシェフ近似法などを用いた畳み込み演算等を行う ことにより、 フィル 夕の伝達関数を求め、 それを更に周波数成分に置き換える処理を行って いる。 IIR (Infinite) Impulse Response: Infinite-length impulse response) Filter and FIR (Finite Impulse Response: finite-length impulse response) filter are often used. The FIR filter has the following advantages. First, the circuit is always stable because the pole of the transfer function of the FIR filter is only at the origin of the z-plane. Secondly, these IIR filters and FIR filters that can achieve perfectly accurate linear phase characteristics are based on single-pass filters, such as high-pass filters, band-pass filters, and band-elimination filters. Other filters are derived by performing processing such as frequency conversion from a low-pass filter. In this frequency conversion processing, a convolution operation using a window function, Chebyshev approximation, or the like is performed based on a ratio between the sampling frequency and the power-off frequency, thereby obtaining a transfer function of the filter. Is calculated, and it is further replaced with frequency components.
しかしながら、 上記従来の音質調整装置のフィルタ設計法では、 周波 数変換などの高度な専門知識が必要であり、 音質調整装置を容易には設 計できないという問題があった。 また、 窓関数やチェビシェフ近似法な どを用いた周波数変換は、 その計算が非常に複雑である。 そのため、 こ れをソフ トウエアで実現すると処理負荷が重くなり、 ハードウェアで実 現すると回路規模が大きくなるという問題があった。  However, the above-mentioned conventional filter design method of the sound quality adjustment device requires a high degree of specialized knowledge such as frequency conversion, and has a problem that the sound quality adjustment device cannot be easily designed. In addition, the calculation of frequency conversion using a window function or Chebyshev approximation is very complicated. Therefore, if this is realized by software, the processing load becomes heavy, and if it is realized by hardware, the circuit scale becomes large.
本発明は、 このような問題を解決するために成されたものであり、 F I Rデジタルフィルタを用いた音質調整装置を簡易的に設計できるよう にすることを目的とする。 発明の開示  The present invention has been made to solve such a problem, and an object of the present invention is to enable a sound quality adjustment device using a FIR digital filter to be simply designed. Disclosure of the invention
上記課題を解決するために、 本発明においては、 所望の周波数特性を 表す数値列もしくは関数を入力し、 当該入力した数値列もしくは関数を 逆フーリエ変換してその結果の実数項を抽出し、 当該抽出した実数項か ら成る数値列に対して、 その前半部と後半部とを並べ替える処理と、 上 記実数項から成る数値列を 2 n倍 （ nは自然数） して小数点以下を丸めた 後その結果を 1 / 2 "倍する丸め処理とを行い、 これによつて得られた数 値列を音質調整装置を構成する第 1 のフィルタのフィルタ係数群として 決定する。 In order to solve the above-mentioned problems, in the present invention, a desired frequency characteristic is Input a numerical sequence or function to represent, perform an inverse Fourier transform of the input numerical sequence or function, extract the resulting real term, and apply the first half and second half to the numerical sequence consisting of the extracted real number term. And a rounding process that multiplies the result by 2 n (where n is a natural number), rounds off the decimal point, and then multiplies the result by 1/2 ”. The numerical sequence obtained as described above is determined as a filter coefficient group of the first filter constituting the sound quality adjustment device.
また、 上記のように入力した数値列もしくは関数に所定の演算を行い 、 その結果に対して逆フーリエ変換、 並べ替え処理および丸め処理を行 う ことにより、 ゲインの基準値を軸として上記第 1 のフィルタと対称的 な周波数特性を有する第 2のフィルタのフィル夕係数群を求める。  Further, a predetermined operation is performed on the input numerical sequence or function as described above, and the result is subjected to inverse Fourier transform, rearrangement processing, and rounding processing, whereby the first reference is obtained with the gain reference value as an axis. A filter coefficient group of a second filter having frequency characteristics symmetric to those of the first filter is obtained.
本発明の他の態様では、 所望の周波数特性を表す数値列もしくは関数 であって、 デジタルフィル夕のタップ数より も多いデータ点を有する数 値列もしくは関数を入力し、 当該入力した数値列もしくは関数を逆フ一 リエ変換してその結果の実数項を抽出し、 当該抽出した実数項から成る 数値列に対して、 その前半部と後半部とを並べ替える処理と、 上記実数 項から成る数値列に所定の窓関数を掛ける処理とを行い、 これによつて 得られた数値列を音質調整装置を構成する第 1 のフィルタのフィルタ係 数群として決定する。  In another aspect of the present invention, a numerical sequence or function representing a desired frequency characteristic, which has a number of data points larger than the number of taps of the digital filter, is input, and the input numerical sequence or function is input. Inverse Fourier transform of the function to extract the real term of the result, and to rearrange the former half and the latter half of the numerical sequence consisting of the extracted real term, and the numerical value consisting of the above real term A process of multiplying the sequence by a predetermined window function is performed, and the obtained numerical sequence is determined as a filter coefficient group of a first filter constituting the sound quality adjustment device.
また、 上記のように入力した数値列もしくは関数に所定の演算を行い 、 その結果に対して逆フーリエ変換、 並べ替え処理および窓掛け処理を 行う ことにより、 基準値を軸として上記第 1 のフィル夕と対称的な周波 数特性を有する第 2のフィルタのフィルタ係数群を求める。 図面の簡単な説明  Further, by performing a predetermined operation on the input numerical sequence or function as described above, and performing an inverse Fourier transform, a rearrangement process, and a windowing process on the result, the first filter is performed with the reference value as an axis. A filter coefficient group of the second filter having frequency characteristics symmetrical to the evening is obtained. BRIEF DESCRIPTION OF THE FIGURES
図 1 は、 本実施形態による音質調整装置の設計方法の処理手順を示す フローチヤ一トである。 FIG. 1 shows a processing procedure of a design method of a sound quality adjustment device according to the present embodiment. It is a flow chart.
図 2は、 本実施形態によるデジタルフィル夕の設計方法の処理手順を 示すフローチヤ一 トである。  FIG. 2 is a flowchart showing a processing procedure of a digital file design method according to the present embodiment.
図 3は、 設計するローパスフィルタの周波数特性の例を示す図である 図 4は、 設計する八ィパスフィル夕の周波数特性の例を示す図である 図 5は、 図 2のステップ S 1 1で入力する所望の周波数特性の例を示 す図である。  FIG. 3 is a diagram showing an example of the frequency characteristics of the low-pass filter to be designed. FIG. 4 is a diagram showing an example of the frequency characteristics of the eight-pass filter to be designed. FIG. 5 is a diagram showing the input in step S 11 of FIG. FIG. 6 is a diagram showing an example of desired frequency characteristics to be obtained.
図 6は、 サンプリング周波数が 4 4 . 1 K H z の音声信号を対象どす る F I Rフィルタを設計する場合における、 入力データ長 mと最大周波 数誤差との関係を示す図である。  FIG. 6 is a diagram showing the relationship between the input data length m and the maximum frequency error when designing a FIR filter for a speech signal having a sampling frequency of 44.1 kHz.
図 7は、 図 2のステップ S 1 3 における並べ替え処理を説明するため の図である。  FIG. 7 is a diagram for explaining the rearrangement process in step S13 of FIG.
図 8は、 所望の周波数特性を表す数値列から本実施形態の設計方法を 適用して求められるフィルタ係数群を示す図である。  FIG. 8 is a diagram showing a filter coefficient group obtained by applying the design method of the present embodiment from a numerical sequence representing a desired frequency characteristic.
図 9は、 図 3および図 4に示すような 2 d B単位の音質調整装置のフ ィルタ係数群を得る際に用いられる係数を示す図である。  FIG. 9 is a diagram showing coefficients used when obtaining a filter coefficient group of the sound quality adjustment device in units of 2 dB as shown in FIGS. 3 and 4.
図 1 0は、 本実施形態による音質調整装置の全体構成を示す図である 図 1 1は、 図 1 0 に示した第 1 の口一パスフィルタの構成を示す図で ある。  FIG. 10 is a diagram showing the overall configuration of the sound quality adjusting device according to the present embodiment. FIG. 11 is a diagram showing the configuration of the first mouth-pass filter shown in FIG.
図 1 2は、 図 1 0 に示した第 2のローパスフィルタの構成を示す図で ある。  FIG. 12 is a diagram showing a configuration of the second low-pass filter shown in FIG.
図 1 3は、 図 1 0 に示した第 1 のハイパスフィルタの構成を示す図で ある。 図 1 4は、 図 1 0 に示した第 2のハイパスフィルタの構成を示す図で ある。 FIG. 13 is a diagram showing a configuration of the first high-pass filter shown in FIG. FIG. 14 is a diagram showing a configuration of the second high-pass filter shown in FIG.
図 1 5は、 図 1 0 に示した信号処理部の構成を示す図である。 発明を実施するための最良の形態  FIG. 15 is a diagram illustrating a configuration of the signal processing unit illustrated in FIG. 10. BEST MODE FOR CARRYING OUT THE INVENTION
以下、 本発明の一実施形態を図面に基づいて説明する。  Hereinafter, an embodiment of the present invention will be described with reference to the drawings.
図 1 は、 本実施形態による音質調整装置の設計方法の処理手順を示す フローチャート、 図 2は、 本実施形態の音質調整装置を構成するデジ夕 ルフィルタの設計方法の処理手順を示すフローチャート、 図 3および図 4は、 設計する音質調整装置の周波数特性を示す図である。 なお、 この 周波数特性においては、 周波数軸 （横軸） もゲイン軸 （縦軸） も対数目 盛り としている。  FIG. 1 is a flowchart showing a processing procedure of a design method of a sound quality adjustment device according to the present embodiment. FIG. 2 is a flowchart showing a processing procedure of a design method of a digital filter constituting the sound quality adjustment device of the present embodiment. FIG. 4 and FIG. 4 are diagrams showing frequency characteristics of the sound quality adjustment device to be designed. In this frequency characteristic, both the frequency axis (horizontal axis) and the gain axis (vertical axis) are on a logarithmic scale.
本実施形態において設計する音質調整装置は、 入力音声信号に対して 口一パスフィルタ処理やハイパスフィルタ処理を施し、 各フィルタの出 力信号と入力音声信号との利得を制御して全て合算するタイプのもので ある。 したがって、 この音質調整装置の設計は、 ローパスフィルタとハ ィパスフィルタとを設計することによって成される。  The sound quality adjustment device designed in the present embodiment is a type that performs a one-pass filter process or a high-pass filter process on an input audio signal, controls the gain of the output signal of each filter and the input audio signal, and adds them all together. belongs to. Therefore, the design of this sound quality adjustment device is achieved by designing a low-pass filter and a high-pass filter.
ここで設計するフィルタは、 複数の遅延器から成るタップ付き遅延線 を備え、 各タップの信号を、 与えられるフィルタ係数群によりそれぞれ 数倍した後、 加算して出力するタイプの F I Rフィルタである。 F I R フィルタは、 有限時間長で表されるインパルス応答がそのままフィルタ の係数となっている。 したがって、 F I Rフィルタを設計するという こ とは、 希望の周波数特性が得られるようにフィルタ係数群を決定すると いう ことである。  The filter designed here is a type of FIR filter that has a delay line with taps composed of a plurality of delay units, multiplies the signal of each tap by a given filter coefficient group, and then adds and outputs the result. In the FIR filter, the impulse response represented by the finite time length is used as the filter coefficient as it is. Therefore, designing an FIR filter means determining a group of filter coefficients so as to obtain the desired frequency characteristics.
図 1 に示すように、 まず低音部の基本となる周波数特性を持つ第 1 の ローパスフィルタ （B A S S 1 ) を設計する （ステップ S 1 ) 。 こ.の第 1 のローパスフィルタは、 図 3に示すように、 ゲインの基準値 1 ( 0 d B ) より正の方向で最大の振幅 （ 1 2 d B ) を有するフィルタである。 このステップ S 1では、 図 2 に示す手順に従って第 1 のローパスフィ ルタを設計する。 すなわち、 まず所望の周波数特性の波形を表す数値列 を入力する （ステップ S 1 1 ) 。 このとき入力する数値列は、 できるだ けデータ数が多くなるようにするのが好ましい。 本来、 理想的なフィル 夕を構成するには、 フィルタ係数を無限個必要とし、 フィルタのタップ 数も無限個にする必要がある。 したがって、 所望の周波数特性との誤差 を小さくするためには、 フィルタ係数の数に対応する入力デ一夕の数を 、 周波数誤差が必要な範囲内に入る程度まで多くするのが好ましい。 少 なく とも、 求めるフィルタ係数の数 （デジタルフィル夕のタップ数） よ りもデータ数が多くなるように数値列を入力する。 As shown in FIG. 1, first, a first low-pass filter (BASS 1) having a basic frequency characteristic of a bass is designed (step S 1). No. As shown in FIG. 3, the low-pass filter 1 has a maximum amplitude (12 dB) in the positive direction from the gain reference value 1 (0 dB). In this step S1, a first low-pass filter is designed according to the procedure shown in FIG. That is, first, a numerical sequence representing a waveform having a desired frequency characteristic is input (step S11). At this time, it is preferable that the numerical sequence to be input has as much data as possible. Originally, to construct an ideal filter, an infinite number of filter coefficients and an infinite number of filter taps were necessary. Therefore, in order to reduce the error from the desired frequency characteristic, it is preferable to increase the number of input data corresponding to the number of filter coefficients to such an extent that the frequency error falls within a required range. At least, enter a numerical sequence so that the number of data is greater than the number of filter coefficients to be obtained (the number of taps in the digital filter).
具体的には、 図 5 に示すように、 対数目盛りのゲインを " 1 " で基準 化したフィルタの周波数一ゲイン特性を描いて、 これを数値データ化す る。 入力データは、 サンプリ ング周波数の中央を軸として対称となるよ うにする。 このとき、 入力データ長 （グラフの長さ、 すなわち数値列の 数） mは、 周波数誤差が必要な範囲内に入る値で、 かつ、 ステップ S 1 2における逆 F F T処理の簡易化のために 2 kとなるようにする。 Specifically, as shown in Fig. 5, the frequency-gain characteristics of a filter with the logarithmic scale gain standardized by "1" are drawn and converted into numerical data. The input data should be symmetric about the center of the sampling frequency. At this time, the input data length (the length of the graph, that is, the number of numerical sequences) m is a value that falls within the required range of the frequency error, and is 2 to simplify the inverse FFT processing in step S12. k .
例えば、 サンプリ ング周波数が 4 4. 1 K H z の音声信号を対象とす る F I Rフィルタを設計する場合、 入力データ長 mと最大周波数誤差と の関係は、 図 6 に示すようになる。 こ こで言う最大周波数誤差は、 ダラ フの 1 目盛り当たりの周波数に相当し、 4 4. l KH z Zmの演算によ つて求められる。 図 5 に示すグラフの例では、 入力データ長 mが 5 1 2 のローパスフィルタに相当する周波数特性を示している。  For example, when designing a FIR filter for a speech signal having a sampling frequency of 44.1 kHz, the relationship between the input data length m and the maximum frequency error is as shown in FIG. The maximum frequency error referred to here corresponds to the frequency per graduation of daraf, and is obtained by the calculation of 44. l KHz z Zm. In the example of the graph shown in FIG. 5, a frequency characteristic corresponding to a low-pass filter having an input data length m of 5 12 is shown.
このデータ入力は、 個々の数値を直接入力しても良いし、 周波数ーゲ イン特性を表すための 2次元入力座標上において、 所望の周波数特性の 波形を描画し、 描画された波形をそれに対応する数値列に置換入力する ようにしても良い。 後者の入力手法を用いれば、 所望の周波数特性をィ メ一ジとして確認しながらデータ入力を行う ことができるので、 所望の 周波数特性を表すデ一夕の入力を直感的に行いやすくすることができる 後者の入力手法を実現する手段は幾つか考えられる。 例えば、 コンビ ユー夕のディスプレイ画面上に周波数一ゲイ ン特性を表す 2次元平面を 表示して、 その 2次元平面上に所望の周波数特性の波形を GU I (Graph ical User Interface) 等により描画し、 それを数値データ化するという 方法が考えられる。 また、 コンピュータ画面上の GU I の代わりに、 デ ィ ジタイザやプロッタ等のボインティ ングデバイスを用いても良い。 こ こに挙げている手法は単なる例に過ぎず、 これ以外の手法により数値列 を入力するようにしても良い。 また、 こ こでは所望の周波数特性を数値 列として入力しているが、 当該周波数特性の波形を表す関数として入力 するようにしても良い。 In this data input, individual numerical values may be directly input, or a desired frequency characteristic may be input on a two-dimensional input coordinate for representing the frequency-gain characteristic. A waveform may be drawn, and the drawn waveform may be replaced with a numerical value sequence corresponding to the drawn waveform. If the latter input method is used, data input can be performed while confirming a desired frequency characteristic as an image, which makes it easy to intuitively input a data representing the desired frequency characteristic. There are several ways to implement the latter input method. For example, a two-dimensional plane representing frequency-gain characteristics is displayed on a display screen of a combination display, and a waveform of a desired frequency characteristic is drawn on the two-dimensional plane using a GUI (Graphical User Interface) or the like. However, there is a method of converting it into numerical data. Further, instead of the GUI on the computer screen, a pointing device such as a digitizer or a plotter may be used. The method described here is merely an example, and a numerical sequence may be input by other methods. Although the desired frequency characteristic is input as a numerical sequence here, it may be input as a function representing the waveform of the frequency characteristic.
次に、 このようにして入力された周波数特性を伝達関数として逆フー リエ変換 （逆 F F T) し、 その結果の実数項を抽出する （ステップ S 1 2 ) 。 周知のように、 ある数値列に対してフーリエ変換 （F F T) の処 理を行うと、 その数値列に対応した周波数一ゲイン特性の波形が得られ る。 したがって、 所望の周波数—ゲイン特性の波形を表す数値列もしく は関数を入力してそれを逆 F F Tし、 その実数項を抽出すれば、 当該周 波数一ゲイン特性を実現するのに必要な元の数値列が得られる。 この数 値列が、 求めるフィルタ係数群に相当するものである。  Next, the input frequency characteristic is subjected to an inverse Fourier transform (inverse FFT) as a transfer function, and a real term of the result is extracted (step S12). As is well known, when a Fourier transform (FFT) process is performed on a numerical sequence, a waveform having a frequency-gain characteristic corresponding to the numerical sequence is obtained. Therefore, if a numerical sequence or function representing the waveform of the desired frequency-gain characteristic is input and inverse FFT is performed to extract the real term thereof, it is necessary to realize the frequency-gain characteristic. The original numeric sequence is obtained. This numerical sequence corresponds to the filter coefficient group to be obtained.
ただし、 逆 F F Tにより求められた数値列そのものは、 フィルタ係数 群としてそのまま使える順番には並んでいない。 すなわち、 どのような タイプのデジタルフィルタでも、 フィルタ係数の数値列は、 中央値が最 も大きく、 中央から離れるに従って振幅を繰り返しながら値が徐々に小 さくなるという対称性を持っている。 これに対して、 逆 F F Tにより求 められた数値列は、 中央値が最も小さく、 両端の値が最も大きくなつて いる。 However, the numerical sequence itself obtained by the inverse FFT is not arranged in the order that can be used as it is as the filter coefficient group. That is, for any type of digital filter, the sequence of filter coefficient values has the highest median value. It has the symmetry that the value gradually decreases while repeating the amplitude as it moves away from the center. In contrast, the numerical sequence obtained by the inverse FFT has the smallest median value and the largest value at both ends.
そこで、 逆 F F Tにより求められた数値列の中央値が両端にくるよう に、 当該数値列を前半部と後半部とに分けてそれらを並べ替える （ステ ップ S 1 3 ) 。 すなわち、 図 7 に示すように、 0クロック目の数値を 2 5 6 クロック目の数値に （以下、 0→ 2 5 6 と表記する） 、 1→ 2 5 7 、 2→ 2 5 8、 · · · 、 2 5 5→ 5 1 1、 2 5 6→ 0 , 2 5 7→ 1、 · - · 5 1 1→ 2 5 5のように並べ替えることにより、 中央値が最大値と なり前後対称となるようにする。  Therefore, the numerical sequence is divided into a first half and a second half, and the numerical sequence is rearranged so that the median of the numerical sequence obtained by the inverse FFT is located at both ends (step S13). That is, as shown in FIG. 7, the value of the 0th clock is changed to the value of the 256th clock (hereinafter, referred to as 0 → 256), 1 → 257, 2 → 258, , 2 5 5 → 5 1 1, 2 5 6 → 0, 2 5 7 → 1,--By sorting as 5 1 1 → 2 5 5, the median becomes the maximum value, To be.
このようにして得られた数値列をそのままフィルタ係数群として決定 することも可能であるが、 本実施形態では更に、 窓掛け演算を行う （ス テツプ S 1 4 ) 。 上述のように、 ステップ S 1 1 のデ一タ入力段階にお いては、 所望の周波数特性との誤差が必要な範囲内に入る程度まで入力 データの数を多く している。 この入力データ数はフィルタ係数の数に対 応するものである。 したがって、 この入力データから逆 F F Tなどの処 理によって求められた数値列をそのままフィル夕係数群として用いると 、 デジタルフィル夕のタップ数が非常に多くなり、 回路規模が大きなも のになつてしまう。 そこで、 窓掛け演算を行うことによって、 タップ数 を必要な数に減らすようにしている。  Although the numerical sequence obtained in this way can be directly determined as a filter coefficient group, a windowing operation is further performed in this embodiment (step S14). As described above, in the data input stage of step S11, the number of input data is increased to such an extent that an error from a desired frequency characteristic falls within a required range. This number of input data corresponds to the number of filter coefficients. Therefore, if a numerical sequence obtained by processing such as inverse FFT from this input data is used as it is as a filter coefficient group, the number of taps in the digital filter becomes very large, and the circuit scale becomes large. . Therefore, the number of taps is reduced to the required number by performing windowing operation.
このとき用いる窓関数には、 方形窓、 ハミング窓、 ハニング窓、 ハー トレツ ト窓などの各種の関数が存在する。 何れの窓関数を適用しても良 いが、 特にハニング窓を用いることが好ましい。 ハニング窓は、 窓の両 端の値が 0で、 しかも中央値から両端に向かって値がなだらかに減衰し ていく関数だからである。 例えば方形窓を用いた場合には、 タップ数を 有限個に強制的に打ち切ることになるが、 これではフィルタ特性上にリ ンギング （波打ち現象） が発生してしまう。 これに対し、 フィルタ係数 を有限の値で打ち切るのではなく、 なだらかに 0 に移行するようにすれ ば、 リ ンギングの発生を抑制することができる。 The window functions used at this time include various functions such as a rectangular window, a Hamming window, a Hanning window, and a Haatlet window. Although any window function may be applied, it is particularly preferable to use a Hanning window. This is because the Hanning window is a function in which the values at both ends of the window are 0, and the values gradually decrease from the median toward both ends. For example, if you use a square window, Although it is forcibly cut off to a finite number, ringing (undulation phenomenon) occurs on the filter characteristics. On the other hand, if the filter coefficient does not stop at a finite value but transitions smoothly to 0, the occurrence of ringing can be suppressed.
また、 このとき用いる窓の幅は、 入力データの減衰量の大きさと関連 して決める必要がある。 図 5 に示した入力データの場合は、 減衰が緩や かなので、 ハニング窓の打ち切りまでの幅を、 例えば 6 4とする。 ステ ップ S 1 4では、 この幅 6 4のハニング窓 （ 6 4個のデータ列） を、 並 ぺ替えによって求められた数値列 （ 5 1 6個のデータ列） の中央部分に 乗算する。 このとき、 ハニング窓の範囲外の係数は全て 0 として計算す る。  The width of the window used at this time must be determined in relation to the amount of attenuation of the input data. In the case of the input data shown in FIG. 5, since the attenuation is moderate, the width up to the termination of the Hanning window is, for example, 64. In step S 14, the central part of the numerical sequence (516 data strings) obtained by the permutation is multiplied by the Hanning window (64 data strings) having a width of 64. At this time, all coefficients outside the Hanning window are calculated as zero.
このような窓掛け演算によって得られた数値列をそのままフィルタ係 数群として用いることも可能である。 しかし、 逆 F F Tおよび窓掛け演 算によって求まるフィルタ係数群は、 少数点以下の桁数が非常に多く、 かつ複雑でランダムな値の集合である。 そのため、 この数値列をそのま まフィルタ係数群として用いると、 デジタルフィルタに必要な乗算器の 数が膨大となり、 現実的でない。  The numerical sequence obtained by such a windowing operation can be used as it is as a filter coefficient group. However, the filter coefficient group obtained by inverse FFT and windowing operation is a set of complex and random values with an extremely large number of digits below the decimal point. Therefore, if this numerical sequence is used directly as a filter coefficient group, the number of multipliers required for the digital filter becomes enormous, which is not practical.
そのため、 数値列の少数点数桁以下を切り捨てるなどしてフィルタ係 数を丸める必要がある。 ところが、 単なる切り捨てによる丸め処理では 、 その結果の数値列は桁数が減っているだけで依然として複雑でランダ ムな値であり、 やはり多くの乗算器を必要とする。 また、 単なる切り捨 てでは、 得られるフィルタ係数群の精度が悪く、 所望の周波数特性との 誤差が大きくなつてしまう。  For this reason, it is necessary to round the filter coefficient by truncating the decimal digits of the numerical sequence. However, in rounding by simply truncation, the resulting sequence of numbers is still a complex and random value with only a reduced number of digits, and still requires many multipliers. In addition, mere truncation results in poor accuracy of the obtained filter coefficient group, resulting in a large error from a desired frequency characteristic.
そこで、 本実施形態では、 以下に述べるような丸め演算を行う （ステ ップ S 1 5 ) 。 すなわち、 上記ステップ S 1 4で窓掛けされた後の数値 列を 2 n倍 （ nは自然数で、 例えば n = 2 0 4 8 ) して小数点以下を丸め (整数化する） 、 その結果を 1 Z 2 11倍する処理を行う。 Therefore, in the present embodiment, the rounding operation described below is performed (step S15). That is, the numerical sequence after windowing in step S14 above is multiplied by 2 n (where n is a natural number, for example, n = 2 0 4 8), and the decimal part is rounded. (Integer conversion), and the result is multiplied by 1 Z 2 11 times.
このような丸め演算によれば、 全てのフィルタ係数,は 1 Z 2 "の整数倍 の値を持つようになる。 よって、 デジタルフィル夕の各タップからの信 号に対して整数倍の部分を個別に乗算し、 それぞれの乗算出力を全て加 算した後にまとめて 1 2 "倍するようにデジタルフィルタを構成するこ とが可能となる。 しかも、 整数倍の部分は、 2 1 + 2 '' + · · · ( i , j は 任意の整数） のように 2進数の足し算で表現できる。 According to such a rounding operation, all filter coefficients have an integer multiple of 1 Z 2 ". Therefore, the integer multiple of the signal from each tap of the digital filter is calculated. It is possible to configure a digital filter so that multiplication is performed individually, all multiplication outputs are added, and then multiplied by 12 "at a time. In addition, the integer multiple can be represented by binary addition, such as 2 1 + 2 '' + · · · (where i and j are arbitrary integers).
これにより、 デジタルフィルタ全体として乗算器の使用数を大きく削 減し、 構成を簡素化することができる。 また、 逆 F F Tにより得られた 数値列を 2 n倍してから丸めているので、 数値列の小数点数桁以下を単に 丸める場合に比べて丸め誤差を小さくすることができる。 これにより、 フィルタ特性の精度を落とすことなくフィルタ係数群を簡素化すること ができる。 As a result, the number of multipliers used in the entire digital filter can be greatly reduced, and the configuration can be simplified. Also, since the numerical sequence obtained by the inverse FFT is rounded after multiplying by 2 n , the rounding error can be reduced as compared with the case where the decimal part of the numerical sequence is simply rounded. As a result, the filter coefficient group can be simplified without lowering the accuracy of the filter characteristics.
本実施形態においては、 このような丸め演算によって求められた数値 列を最終的にフィルタ係数群として決定する。 なお、 上述のステップ S 1 3 〜 S 1 5の処理は、 必ずしもこの順番で行う必要はなく、 少なく と も窓掛け演算より後に丸め演算を行うのであれば良い。 例えば、 窓掛け 演算を並べ替えの前に行っても良い。 この場合は、 窓の両端の係数値が " 1 " で、 窓の中央部の係数値が " 0 " となるようなハニング窓を乗算 する。 このように窓掛け演算を一連の手順の中の早い段階で行うことに より、 以降の演算に使用するデータ数を減らすことができ、 演算にかか る処理負荷を軽減することができる。  In the present embodiment, a numerical sequence obtained by such a rounding operation is finally determined as a filter coefficient group. Note that the processes in steps S13 to S15 do not necessarily need to be performed in this order, and it is sufficient if the rounding operation is performed at least after the windowing operation. For example, a windowing operation may be performed before sorting. In this case, multiply the Hanning window so that the coefficient value at both ends of the window is "1" and the coefficient value at the center of the window is "0". By performing the windowing operation at an early stage in the series of procedures, the number of data used for subsequent operations can be reduced, and the processing load on the operation can be reduced.
このようにして求められるフィルタ係数群 （ 6 4個のフィルタ係数） は、 図 5のような入力デ一夕の周波数特性をほぼ正確に実現している。 しかも、 位相特性も直線で安定な特性を実現できている。  The filter coefficient group (64 filter coefficients) obtained in this way realizes almost exactly the frequency characteristics of the input data as shown in FIG. In addition, the phase characteristics are linear and stable.
以上のようにして第 1 の口一パスフィル夕を設計したら、 次に第 2の ローパスフィルタ （B A S S 2 ) を設計する （ステップ S 2 ) 。 この第 2のローパスフィルタは、 図 3に示すように、 ゲインの基準値 1 ( 0 d B ) より負の方向で最大の振幅 （― 1 2 d B) を有するフィルタであり 、 基準値 1 を軸として第 1 のローパスフィルタと線対称な特性を有する ものである。 この第 2の口一パスフィルタの設計も、 図 2のフ口一チヤ 一卜に示す手順に従って行う。 After designing the first mouth-to-pass fill as described above, Design a low-pass filter (BASS 2) (step S 2). As shown in FIG. 3, the second low-pass filter has a maximum amplitude (−12 dB) in a negative direction from the gain reference value 1 (0 dB). The axis has characteristics that are line-symmetric with the first low-pass filter. The design of the second mouth-pass filter is also performed according to the procedure shown in FIG.
縦横の両軸を対数目盛り とした L O G— L O G平面において、 基準値 1 より上側の曲線 ί ( X ) を 1 + g ( X ) で表すとすると、 この曲線 f ( X ) に対して基準値 1 を軸として線対称な曲線 f ( ) ' は、 次の式（ 1)のようになる。  LOG with both the vertical and horizontal axes on a logarithmic scale—In the LOG plane, if the curve ί (X) above the reference value 1 is represented by 1 + g (X), the reference value 1 for this curve f (X) The curve f () ′, which is symmetrical with respect to, is given by the following equation (1).
f ( ) ' = 1 - 1 κ ( l + g ( χ ) ) · · · (l)  f () '= 1-1 κ (l + g (χ))
そこで、 ステップ S I 1では、 第 1 の口一パスフィルタを設計するため に入力された数値列を式（1)に代入することにより、 第 2の口一パスフィ ル夕の入力データを得る。 そして、 この入力データに対してステップ S 1 2〜 S 1 5 と同様の処理を行う ことにより、 第 2の口一パスフィルタ のフィルタ係数群を求める。 Therefore, in step SI1, the input data of the second mouth-to-pass filter is obtained by substituting the numerical sequence input for designing the first mouth-to-pass filter into equation (1). Then, by performing the same processing as in steps S12 to S15 on the input data, a filter coefficient group of the second mouth-pass filter is obtained.
さらに、 以上のような第 1および第 2のローパスフィルタの設計法と 同様にして、 第 1および第 2のハイパスフィルタを設計する （ステップ S 3 , S 4 ) 。 ただし、 ハイパスフィルタの設計では、 ステップ S 1 4 の窓掛け演算で用いるハニング窓の幅を 8 とする。 ハイパスフィルタの 場合、 逆 F F Tした結果の実数項の減衰量が大きいので、 窓の幅を 8 ま で短くすることが可能である。 窓の幅を小さくすることにより、 フィル 夕のタップ数も少なくすることができる。  Further, the first and second high-pass filters are designed in the same manner as the above-described first and second low-pass filter designing methods (steps S 3 and S 4). However, in the design of the high-pass filter, the width of the Hanning window used in the windowing operation in step S14 is set to 8. In the case of a high-pass filter, the attenuation of the real term resulting from the inverse FFT is large, so that the window width can be reduced to eight. By reducing the width of the window, the number of taps in the evening can be reduced.
なお、 図 1 の例では、 第 1および第 2の口一パスフィルタを設計した 後に第 1および第 2のハイパスフィル夕を設計しているが、 この順番は 逆でも良い。 また、 図 1 の例では、 第 1 のローパスフィルタあるいは第 1のハイパスフィルタを設計した後に第 2のローパスフィルタあるいは 第 2のハイパスフィルタを設計しているが、 この順番も逆で良い。 In the example of FIG. 1, the first and second high-pass filters are designed after the first and second single-pass filters are designed. However, the order may be reversed. In the example of Fig. 1, the first low-pass filter or the After designing the first high-pass filter, the second low-pass filter or the second high-pass filter is designed, but this order may be reversed.
図 8は、 以上のようにして求められた L P F 1 , 2 、 H P F 1 , 2の フィルタ係数群を示す図である。 これらのフィル夕係数群を用いて、 入 力音声信号を遅延させる 1つのディ レイラインと 4つの F I Rフィルタ とから成るフィルタプロックを形成することができる。 また、 このフィ ルタ係数群に対して図 9 に示すような係数を乗じて " 1 " に加えること により、 図 3および図 4に示すような 2 d B単位の音質調整装置のフィ ルタ係数群を得ることができる。  FIG. 8 is a diagram showing the filter coefficient groups of LPF 1,2 and HPF 1,2 obtained as described above. By using these filter coefficient groups, a filter block including one delay line and four FIR filters for delaying an input audio signal can be formed. Also, by multiplying this filter coefficient group by a coefficient as shown in FIG. 9 and adding it to “1”, the filter coefficient group of the sound quality adjustment device in 2 dB units as shown in FIGS. 3 and 4 is obtained. Can be obtained.
このようにして設計した音質調整装置を実際に構成する場合、 囪 8 に 示すフィルタ係数群を 2 0 4 8倍すると全ての値が整数となる。 このよ うにすると、 複数の遅延器から成るタップ付き遅延線の各タップの信号 にフィルタ係数を乗算する計算で乗算器を用いる必要がなく、 ビッ トシ フタと加算器だけの構成で済む。 また、 各タップの信号とフィルタ係数 との積和演算後の数値を 2 0 4 8で割る割算器も、 下位 1 1 ビッ トを処 理すれば良く、 構成を簡素化することができる。  When actually configuring the sound quality adjustment device designed in this way, if the filter coefficient group shown in 囪 8 is multiplied by 2048, all values become integers. With this configuration, it is not necessary to use a multiplier in the calculation of multiplying the signal of each tap of the tapped delay line including a plurality of delay elements by the filter coefficient, and only the configuration of the bit shifter and the adder is sufficient. Also, a divider that divides the numerical value of the product of each tap signal and the filter coefficient after the product-sum operation by 24048 may process only the lower 11 bits, and the configuration can be simplified.
図 1 0は、 図 8 に示す 4つのフィルタブロックを用いた音質調整装置 の全体構成例を示す図である。 図 8 において、 1 1 〜 1 4は上記図 1ぉ よび図 2に示す手順によって設計した第 1および第 2の口一パスフィル タ、 第 1およぴ第 2のハイパスフィルタである。 このうち第 1 の口一パ スフィルタ 1 1 は、 入力音声信号のディ レイラインも兼ねている。 1 5 は信号処理部であり、 各フィルタ 1 1 〜 1 4から出力される信号 （ 1つ のディ レイ ライン出力と 4つのフィルタ出力） とを入力し、 それらの利 得を制御して出力する。  FIG. 10 is a diagram showing an overall configuration example of a sound quality adjustment device using the four filter blocks shown in FIG. In FIG. 8, reference numerals 11 to 14 denote first and second single-pass filters and first and second high-pass filters designed by the procedure shown in FIGS. 1 and 2 described above. The first mouth-pass filter 11 also serves as a delay line for the input audio signal. Reference numeral 15 denotes a signal processing unit, which inputs signals (one delay line output and four filter outputs) output from each of the filters 11 to 14 and outputs the signals by controlling their gain. .
図 1 1〜図 1 4は、 上記 4つのフィルタ 1 1 〜 1 4の内部構成を示す 図である。 これらのフィルタ 1 1 〜 1 4では、 縦続接続された複数個の D型フリ ップフロップによって入力信号を 1 クロック C Kずつ順次遅延 させる。 そして、 各 D型フリ ップフロップの出力タップから取り出した 信号に対し、 フィルタ係数を 2 0 4 8倍した結果の整数値を複数個の係 数器によってそれぞれ乗算し、 それらの乗算結果をすベて複数個の加算 器で加算して出力する。 なお、 第 1 の口一パスフィルタ 1 1では、 入力 音声信号が複数の D型フリ ップフロップを通過するだけのディ レイライ ンも設けられている。 FIGS. 11 to 14 are diagrams showing the internal configuration of the above four filters 11 to 14. These filters 11 to 14 include multiple cascaded filters. The input signal is sequentially delayed by one clock CK by the D-type flip-flop. Then, a signal extracted from the output tap of each D-type flip-flop is multiplied by an integer value obtained by multiplying the filter coefficient by 248 by each of a plurality of coefficients, and all of the multiplication results are obtained. The output is added by multiple adders. Note that the first mouth-pass filter 11 is also provided with a delay line that allows the input audio signal to pass through a plurality of D-type flip-flops.
図 1 5は、 信号処理部 1 5の内部構成を示す図である。 図 1 5 におい て、 2 1は第 1 のデコーダであり、 第 1および第 2の口一パスフィルタ 1 1 , 1 2の利得制御信号を入力してデコードする。 2 2〜 2 4は複数 のスィ ッチであり、 第 1 のデコーダ 2 1 によるデコード結果に基づいて スイッチング動作する。 このスイッチング動作により、 第 1およぴ第 2 のローパスフィルタ 1 1 , 1 2の何れかの出力信号を選択し、 その利得 を制御するようになっている。  FIG. 15 is a diagram showing the internal configuration of the signal processing unit 15. In FIG. 15, reference numeral 21 denotes a first decoder, which inputs and decodes gain control signals of the first and second single-pass filters 11 1 and 12. Reference numerals 22 to 24 denote a plurality of switches, which perform a switching operation based on the decoding result of the first decoder 21. By this switching operation, one of the output signals of the first and second low-pass filters 11 and 12 is selected, and the gain thereof is controlled.
2 5は割算器であり、 第 1および第 2の口一パスフィルタ 1 1， 1 2 よりスィッチ 2 2 を通過した信号を 2 0 4 8で除算する。 図 1 1および 図 1 2 に示したように、 第 1および第 2の口一パスフィルタ 1 1 , 1 2 の内部では、 図 8 に示すフィルタ係数群を 2 0 4 8倍した結果の整数値 を各タップ出力に乗算している。 よって、 振幅を正しい値に戻すために 、 割算器 2 5 においてフィルタ出力を 2 0 4 8で除算している。 2 6 は 複数の係数器であり、 割算器 2 5 を通過した信号に対して、 図 9 に示す 係数値の何れかを乗算する。 どの係数を乗算するかは、 第 1のデコーダ 2 1 によるデコー ド結果に応じて決められる。  Reference numeral 25 denotes a divider, which divides the signal passed through the switch 22 from the first and second single-pass filters 11 and 12 by 2048. As shown in FIGS. 11 and 12, inside the first and second mouth-pass filters 1 1 and 1 2, an integer value obtained by multiplying the filter coefficient group shown in FIG. Is multiplied by each tap output. Therefore, in order to return the amplitude to a correct value, the filter output is divided by 204 in the divider 25. Reference numeral 26 denotes a plurality of coefficient units, which multiply the signal passed through the divider 25 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the decoding result by the first decoder 21.
3 1 は第 2のデコーダであり、 第 1および第 2のハイパスフィルタ 1 3 , 1 4の利得制御信号を入力してデコードする。 3 2 ~ 3 4は複数の スィ ッチであり、 第 2のデコーダ 3 1 によるデコード結果に基づいてス イ ッチング動作する。 このスイ ッチング動作により、 第 1および第 2の ハイパスフィルタ 1 3 , 1 4の何れかの出力信号を選択し、 その利得を 制御するようになっている。 Reference numeral 31 denotes a second decoder, which inputs and decodes the gain control signals of the first and second high-pass filters 13 and 14. Reference numerals 32 to 34 denote a plurality of switches, which are switched based on the decoding result of the second decoder 31. Switching operation is performed. By this switching operation, one of the output signals of the first and second high-pass filters 13 and 14 is selected, and the gain thereof is controlled.
3 5は割算器であり、 第 1および第 2のハイパスフィルタ 1 3， 1 よりスィッチ 3 2 を通過した信号を 2 0 4 8で除算する。 図 1 3および 図 1 4に示したように、 第 1および第 2のハイパスフィルタ 1 3， 1 4 の内部では、 図 8 に示すフィルタ係数群を 2 0 4 8倍した結果の整数値 を各タップ出力に乗算している。 よって、 振幅を正しい値に戻すために 、 割算器 3 5 においてフィルタ出力を 2 0 4 8で除算している。 3 6は 複数の係数器であり、 割算器 3 5 を通過した信号に対して、 図 9 に示す 係数値の何れかを乗算する。 どの係数を乗算するかは、 第 2のデコーダ 3 1 によるデコード結果に応じて決められる。  Reference numeral 35 denotes a divider, which divides the signal passed through the switch 32 from the first and second high-pass filters 13 and 1 by 2048. As shown in FIGS. 13 and 14, inside the first and second high-pass filters 13 and 14, the integer values obtained by multiplying the filter coefficient group shown in FIG. Multiplies the tap output. Therefore, in order to return the amplitude to the correct value, the filter output is divided by 204 in the divider 35. Reference numeral 36 denotes a plurality of coefficient units, which multiply the signal passing through the divider 35 by any one of the coefficient values shown in FIG. Which coefficient is to be multiplied is determined according to the result of decoding by the second decoder 31.
4 1 は加算器であり、 第 1 の口一パスフィルタ 1 1 のディ レイライン から入力された音声信号と、 第 1および第 2のローパスフィルタ 1 1 , 1 2の何れかの出力に対して利得制御をした後の音声信号とを加算する 。 4 2は加算器であり、 第 1 の口一パスフィルタ 1 1のディ レイライン から入力された音声信号と、 第 1および第 2のハイパスフィルタ 1 3 , 1 4の何れかの出力に対して利得制御をした後の音声信号とを加算する 。 4 3は加算器であり、 各加算器 4 1， 4 2の出力どうしを加算して、 最終的に音質調整のされた音声信号を出力する。  4 1 is an adder, which adds an audio signal input from the delay line of the first one-pass filter 11 to one of the outputs of the first and second low-pass filters 11 1 and 12. The audio signal after the gain control is added. Reference numeral 2 denotes an adder, which adds an audio signal input from the delay line of the first single-pass filter 11 to one of the outputs of the first and second high-pass filters 13 and 14. The audio signal after the gain control is added. An adder 43 adds the outputs of the adders 41 and 42 and finally outputs a sound signal whose sound quality has been adjusted.
以上に説明した本実施形態による音質調整装置の設計方法を実現する ための装置は、 ハードウェア構成、 D S P、 ソフ トウェアの何れによつ ても実現することが可能である。 例えばソフ トウエアによって実現する 場合、 本実施形態の設計装置は、 実際にはコンピュータの C P Uあるい は M P U、 R A M、 R O Mなどで構成され、 R A Mや R O Mあるいはハ ー ドディスク等に記憶されたプログラムが動作することによって実現で ぎる。 An apparatus for implementing the above-described method for designing a sound quality adjustment apparatus according to the present embodiment can be implemented by any of a hardware configuration, a DSP, and software. For example, when realized by software, the design device of the present embodiment is actually configured by a computer CPU or MPU, RAM, ROM, etc., and a program stored in RAM, ROM, a hard disk, or the like is used. Realized by working Cut.
したがって、 コンピュータが上記本実施形態の機能を果たすように動 作させるプログラムを例えば C D— R〇 Mのような記録媒体に記録し、 コンピュータに読み込ませることによって実現できるものである。 上記 プログラムを記録する記録媒体としては、 C D— R O M以外に、 フレキ シブルディスク、 ハードディスク、 磁気テープ、 光ディスク、 光磁気デ イスク、 D V D、 不揮発性メモリカード等を用いることができる。 また 、 上記プログラムをインターネッ ト等のネッ トワークを介してコンビュ —夕にダウンロードすることによつても実現できる。  Therefore, the present invention can be realized by recording a program that causes a computer to perform the functions of the present embodiment on a recording medium such as a CD-R〇M and reading the program into the computer. As a recording medium for recording the above program, a flexible disk, a hard disk, a magnetic tape, an optical disk, a magneto-optical disk, a DVD, a nonvolatile memory card, and the like can be used in addition to the CD-ROM. In addition, it can be realized by downloading the above program in the evening via a network such as the Internet.
また、 コンピュータが供給されたプログラムを実行することにより上 述の実施形態の機能が実現されるだけでなく、 そのプログラムがコンビ ユー夕において稼働している〇 S (ォペレ一ティ ングシステム） あるい は他のアプリケ一ショ ンソフ 卜等と共同して上述の実施形態の機能が実 現される場合や、 供給されたプログラムの処理の全てあるいは一部がコ ンピュ一夕の機能拡張ポ一 ドゃ機能拡張ュニッ トにより行われて上述の 実施形態の機能が実現される場合も、 かかるプログラムは本発明の実施 形態に含まれる。  In addition, not only the functions of the above-described embodiments are realized by the computer executing the supplied program, but also the program is running at the convenience of the computer. S (operating system) or In the case where the functions of the above-described embodiment are realized in cooperation with another application software or the like, or when all or a part of the processing of the supplied program is executed by a computer expansion function port. Such a program is also included in the embodiment of the present invention when the function of the above-described embodiment is implemented by the function extension unit.
以上詳しく説明したように、 本実施形態では、 所望の周波数特性の波 形を表す数値列をイメージとして入力し、 これを逆フーリエ変換するこ とによって、 音質調整装置を構成する各フィルタのフィルタ係数群を求 めるようにしたので、 特別な数学知識や電気工学知識がなくても、 所望 の周波数特性を実現する F I Rデジタルフィルタの係数を簡単に決定す ることができる。  As described above in detail, in the present embodiment, a numerical value sequence representing a waveform of a desired frequency characteristic is input as an image, and this is subjected to an inverse Fourier transform to thereby obtain a filter coefficient of each filter constituting the sound quality adjustment device. Since the groups are determined, the coefficients of the FIR digital filter that achieve the desired frequency characteristics can be easily determined without special mathematical or electrical engineering knowledge.
また、 本実施形態では、 逆フーリエ変換により求められた数値列に対 して特殊な丸め演算を行う ことにより、 フィルタの精度を落とさずにフ ィルタ係数群を簡素化することができ、 フィル夕構成要素の乗算器 （割 算器） の使用数を大幅に削減することができる。 さ らに、 本実施形態で は、 逆フーリエ変換の結果に対して必要な長さの窓関数を乗ずるように したので、 入力デ一夕長を長く して周波数誤差を小さく抑制すると同時 に、 フィルタ係数の数 （デジタルフィルタのタップ数） を少なく抑える ことができる。 これにより、 設計する音質調整装置の構成を簡素化する とともに、 希望する周波数特性を高精度に実現することができる。 Further, in the present embodiment, by performing a special rounding operation on the numerical sequence obtained by the inverse Fourier transform, it is possible to simplify the filter coefficient group without reducing the accuracy of the filter. Component multiplier (divide ) Can be greatly reduced. Further, in the present embodiment, the result of the inverse Fourier transform is multiplied by a window function of a required length, so that the input data length is increased to reduce the frequency error, and at the same time, The number of filter coefficients (the number of taps of the digital filter) can be reduced. This simplifies the configuration of the sound quality adjustment device to be designed, and achieves desired frequency characteristics with high accuracy.
なお、 上記実施形態は、 何れも本発明を実施するにあたっての具体化 の一例を示したものに過ぎず、 これによつて本発明の技術的範囲が限定 的に解釈されてはならないものである。 すなわち、 本発明はその精神、 またはその主要な特徴から逸脱することなく、 様々な形で実施すること ができる。  It should be noted that each of the above-described embodiments is merely an example of a specific embodiment for carrying out the present invention, and the technical scope of the present invention should not be interpreted in a limited manner. . That is, the present invention can be implemented in various forms without departing from the spirit or the main features.
以上説明したように本発明によれば、 所望の周波数特性の波形を数値 列もしくは関数として入力し、 これに逆フーリエ変換等の処理を行うこ とによって第 1 のフィルタのフィルタ係数群を求めるとともに、 上記の ように入力した数値列もしくは関数に所定の演算を行い、 その結果に対 して逆フーリエ変換等の処理を行う ことにより、 ゲインの基準値を軸と して上記第 1 のフィルタと対称的な周波数特性を有する第 2のフィルタ のフィル夕係数群を求めるようにしたので、 専門知識がなくても、 音質 調整装置を構成する F I Rデジタルフィル夕を簡易的に設計することが できる。  As described above, according to the present invention, a filter coefficient group of the first filter is obtained by inputting a waveform of a desired frequency characteristic as a numerical sequence or a function and performing processing such as inverse Fourier transform on the waveform. By performing a predetermined operation on the input numerical sequence or function as described above and performing a process such as an inverse Fourier transform on the result, the first filter is processed with the gain reference value as an axis. Since the filter coefficient group of the second filter having symmetrical frequency characteristics is determined, the FIR digital filter constituting the sound quality adjustment device can be simply designed without any specialized knowledge.
また、 本発明によれば、 逆フーリエ変換により求められた数値列に対 して特殊な丸め演算を行うようにしたので、 フィルタ特性の精度を落と すことなく、 求めるフィルタ係数群を簡素化することができ、 フィルタ 構成要素の乗算器の使用数を大幅に削減することができる。 これにより 、 希望する周波数特性を小さな回路規模で高精度に実現することが可能 な音質調整装置を簡易的に設計することができる。 また、 本発明によれば、 逆フ一リエ変換の結果に対して窓掛け演算を 行うようにしたので、 最初に入力する数値列を長く して周波数誤差を小 さく抑制すると同時に、 フィルタ係数の数 （デジタルフィル夕のタップ 数） を少なく抑え、 設計するデジタルフィルタの構成を簡素化すること ができる。 これにより、 希望する周波数特性を小さな回路規模で高精度 に実現することが可能な音質調整装置を簡易的に設計することができる Further, according to the present invention, a special rounding operation is performed on the numerical sequence obtained by the inverse Fourier transform, so that the filter coefficient group to be obtained is simplified without lowering the accuracy of the filter characteristics. The number of multipliers used in the filter component can be greatly reduced. This makes it possible to simply design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale. In addition, according to the present invention, the windowing operation is performed on the result of the inverse Fourier transform, so that the first input numerical sequence is lengthened to reduce the frequency error, and at the same time, the filter coefficient is reduced. The number (the number of taps in the digital filter) can be reduced, and the configuration of the digital filter to be designed can be simplified. As a result, it is possible to easily design a sound quality adjustment device capable of achieving a desired frequency characteristic with high accuracy on a small circuit scale.
産業上の利用可能性 Industrial applicability
本発明は、 F I Rデジタルフィルタを用いた音質調整装置を簡易的に 設計できるようにするのに有用である。  INDUSTRIAL APPLICABILITY The present invention is useful for easily designing a sound quality adjustment device using a FIR digital filter.

Claims

請 求 の 範 囲 The scope of the claims
1 . 複数の遅延器から成るタツプ付き遅延線における各タップの信号を フィルタ係数群によりそれぞれ数倍した後加算して出力するタイプのデ ジタルフィルタを用いた音質調整装置の設計方法であって、 1. A design method of a sound quality adjusting device using a digital filter of a type in which a signal of each tap in a tap-added delay line composed of a plurality of delay units is multiplied by a filter coefficient group and added and output,
所望の周波数特性を表す.数値列もしくは関数を入力し、 当該入力した 数値列もしくは関数を逆フーリエ変換してその結果の実数項を抽出し、 当該抽出した実数項から成る数値列に対して、 その前半部と後半部とを 並べ替える処理と、 上記実数項から成る数値列を 2 "倍 （nは自然数） し て小数点以下を丸めた後その結果を 1 2 n倍する丸め処理とを行い、 こ れによって得られた数値列を第 1のフィルタのフィル夕係数群として求 めるとともに、 Represents a desired frequency characteristic.Input a numerical sequence or function, inverse Fourier transform the input numerical sequence or function, extract the resulting real term, its front half portion and the latter half portion and the sort process, a numerical sequence consisting of the real term 2 "times (n is a natural number) is performed and the result 1 2 n multiplied rounding after rounding the fractional and , And the numerical sequence obtained as a result is obtained as a filter coefficient group of the first filter.
上記入力した数値列もしくは関数に所定の演算を行い、 その結果に対 して上記逆フ一リエ変換、 並べ替え処理および丸め処理を行う ことによ つて、 ゲインの基準値を軸として上記第 1 のフィルタと対称的な周波数 特性を有する第 2のフィルタのフィルタ係数群を求めるようにしたこと を特徴とする音質調整装置の設計方法。  A predetermined operation is performed on the input numerical sequence or function, and the result is subjected to the above inverse Fourier transform, rearrangement processing, and rounding processing. A filter coefficient group of a second filter having a frequency characteristic symmetric to that of the second filter is obtained.
2 . 周波数軸およびゲイン軸の両軸が対数目盛りの 2次元平面において 上記入力した数値列もしくは関数を f ( ) = 1 + g ( x ) で表した場 合に、 上記入力した数値列もしくは関数に対して行う上記所定の演算は f ( X ) ' = 1 - 1 / ( 1 + g ( x ) )  2. When the above input numerical sequence or function is represented by f () = 1 + g (x) on a two-dimensional plane where both the frequency axis and the gain axis are logarithmic scales, the above input numerical sequence or function The above predetermined operation to be performed is f (X) '= 1-1 / (1 + g (x))
であることを特徴とする請求の範囲第 1項に記載の音質調整装置の設計 方法。 2. The method for designing a sound quality adjusting device according to claim 1, wherein:
3 . 複数の遅延器から成るタツプ付き遅延線における各夕ップの信号を フィルタ係数群によりそれぞれ数倍した後加算して出力するタイプのデ ジタルフィルタを用いた音質調整装置の設計方法であって、 所望の周波数特性を表す数値列もしくは関数であって、 上記デジ夕ル フィルタのタップ数より も多いデータ点を有する数値列もしくは関数を 入力し、 当該入力した数値列もしくは関数を逆フーリエ変換してその結 果の実数項を抽出し、 当該抽出した実数項から成る数値列に対して、 そ の前半部と後半部とを並べ替える処理と、 上記実数項から成る数値列に 所定の窓関数を掛ける処理とを行い、 これによつて得られた数値列を第 1のフィル夕のフィルタ係数群として求めるとともに、 3. A type of data in which each evening signal in a tapped delay line composed of a plurality of delay units is multiplied by a filter coefficient group and added and output. A method for designing a sound quality adjustment device using a digital filter, wherein a numerical sequence or function representing a desired frequency characteristic and having a number of data points greater than the number of taps of the digital filter is input. A process in which the input numerical sequence or function is subjected to inverse Fourier transform to extract a real term of the result, and the former half and the latter half are rearranged in the numerical sequence composed of the extracted real number term. And performing a process of multiplying the numerical sequence composed of the real number terms by a predetermined window function, thereby obtaining the numerical sequence obtained as a filter coefficient group of the first filter,
上記入力した数値列もしくは関数に所定の演算を行い、 その結果に対 して上記逆フーリエ変換、 並べ替え処理および窓掛け処理を行う ことに よって、 ゲインの基準値を軸として上記第 1 のフィルタと対称的な周波 数特性を有する第 2のフィル夕のフィルタ係数群を求めるようにしたこ とを特徴とする音質調整装置の設計方法。  A predetermined operation is performed on the input numerical sequence or function, and the result is subjected to the inverse Fourier transform, the rearrangement process, and the windowing process, whereby the first filter is set with the gain reference value as an axis. A filter coefficient group of a second filter having a frequency characteristic symmetrical to that of the first embodiment.
4 . 上記逆フ一リエ変換の結果の実数項から成る数値列が並べ替えられ る前もしくは並べ替えられた後の数値列、 あるいは、 上記窓関数が掛け られた後の数値列を 2 n倍 （ nは自然数） して小数点以下を丸め、 その結 果を 1 / 2 "倍する丸め処理を更に行うように成し、 4. The number sequence before or after the numerical sequence consisting of the real terms resulting from the inverse Fourier transform is rearranged, or the numerical sequence after the window function is multiplied by 2 n times (Where n is a natural number), round the decimal point, and perform further rounding to multiply the result by 1/2 ".
上記所望の周波数特性を表す数値列もしくは関数を入力して逆フーリ ェ変換、 並べ替え処理、 窓掛け処理および丸め処理を行う ことによって 上記第 1 のフィルタのフィルタ係数群を求めるとともに、  By inputting a numerical sequence or a function representing the desired frequency characteristic and performing an inverse Fourier transform, a rearrangement process, a windowing process, and a rounding process, a filter coefficient group of the first filter is obtained.
上記入力した数値列もしくは関数に所定の演算を行い、 その結果に対 して上記逆フーリエ変換、 並べ替え処理、 窓掛け処理および丸め処理を 行う ことによって上記第 2のフィルタのフィルタ係数群を求めるように したことを特徴とする請求の範囲第 3項に記載の音質調整装置の設計方 法。  A predetermined operation is performed on the input numerical sequence or function, and the result is subjected to the inverse Fourier transform, the rearrangement process, the windowing process, and the rounding process, thereby obtaining a filter coefficient group of the second filter. 4. The method for designing a sound quality adjusting device according to claim 3, wherein the sound quality adjusting device is designed as follows.
5 . 周波数軸およびゲイン軸の両軸が対数目盛りの 2次元平面において 上記入力した数値列もしくは関数を f ( X ) = 1 + g ( x ) で表した場 合に、 上記入力した数値列もしくは関数に対して行う上記所定の演算は f ( X ) ' = 1 - 1 / ( 1 + g ( x ) ) 5. In the two-dimensional plane where both the frequency axis and the gain axis are logarithmic When the input numerical sequence or function is represented by f (X) = 1 + g (x), the predetermined operation to be performed on the input numerical sequence or function is f (X) '= 1- 1 / (1 + g (x))
であることを特徴とする請求の範囲第 3項に記載の音質調整装置の設計 方法。 4. The method for designing a sound quality adjusting device according to claim 3, wherein:
6 . 複数の遅延器から成るタツプ付き遅延線における各夕ップの信号を フィルタ係数群によりそれぞれ数倍した後加算して出力するタイプのデ ジタルフィルタを用いた音質調整装置の設計装置であって、  6. A design apparatus for a sound quality adjustment device using a digital filter of a type that multiplies each signal of each evening by a filter coefficient group in a delay line with taps composed of a plurality of delay units and then adds and outputs the result. hand,
所望の周波数特性の波形を表す数値列もしくは関数を入力する入力手 段と、  An input means for inputting a numerical sequence or a function representing a waveform having a desired frequency characteristic;
上記入力手段により入力された数値列もしくは関数を逆フーリエ変換 し、 その結果の実数項を抽出する逆フーリエ変換手段と、  Inverse Fourier transform means for performing an inverse Fourier transform on the numerical sequence or function input by the input means, and extracting a real term of the result;
上記逆フーリエ変換により求められた数値列の前半部と後半部とを並 ベ替える並べ替え手段と、  Reordering means for rearranging the first half and the second half of the numerical sequence obtained by the inverse Fourier transform;
上記並べ替え手段により並べ替えられる前もしくは並べ替えられた後 の上記実数項の数値列を 2 n倍 （ nは自然数） して小数点以下を丸め、 そ の結果を 1 / 2 -倍する処理を行う丸め手段とを備え、 The process of multiplying the number sequence of the real term before or after sorting by 2 n (where n is a natural number) by 2 n (where n is a natural number), rounding the decimal part, and multiplying the result by 1/2- And a rounding means for performing
上記入力手段により入力された所望の周波数特性を表す数値列もしく は関数に対して上記逆フーリエ変換手段、 上記並べ替え手段および上記 丸め手段の処理を行うことによって第 1 のフィルタのフィルタ係数群を 求めるとともに、 上記入力手段により入力された数値列もしくは関数に 所定の演算を行い、 その結果に対して上記逆フーリエ変換手段、 上記並 ベ替え手段および上記丸め手段の処理を行う ことによって、 ゲインの基 準値を軸として上記第 1のフィル夕と対称的な周波数特性を有する第 2 のフィル夕のフィルタ係数群を求めるようにしたことを特徴とする音質 調整装置の設計装置。 By performing the processing of the inverse Fourier transform means, the rearranging means, and the rounding means on a numerical sequence or a function representing a desired frequency characteristic inputted by the input means, a filter coefficient group of a first filter is obtained. And a predetermined operation is performed on the numerical sequence or function input by the input means, and the result is processed by the inverse Fourier transform means, the rearrangement means, and the rounding means. A filter coefficient group of a second filter having frequency characteristics symmetrical to the first filter with respect to the reference value of the second filter. Adjustment device design equipment.
7 . 上記入力手段は、 周波数—ゲイン特性を表すための 2次元入力座標 上において上記所望の周波数特性の波形を描画するための手段と、 描画 された波形を上記数値列もしくは関数として入力するための手段とを含 むことを特徴とする請求の範囲第 6項に記載の音質調整装置の設計装置  7. The input means is means for drawing a waveform of the desired frequency characteristic on two-dimensional input coordinates for representing frequency-gain characteristics, and for inputting the drawn waveform as the numerical sequence or function. 7. A sound quality adjusting device designing apparatus according to claim 6, comprising:
8 . 周波数軸およびゲイン軸の両軸が対数目盛りの 2次元平面において 上記入力手段により入力した数値列もしくは関数を f ( ) = 1 + g ( x ) で表した場合に、 上記入力した数値列もしくは関数に対して行う上 記所定の演算は、 8. When the numerical sequence or function input by the above input means is represented by f () = 1 + g (x) on a two-dimensional plane in which both the frequency axis and the gain axis are logarithmic scales, the numerical sequence input above Alternatively, the predetermined operation to be performed on the function is
f ( X ) ' = 1 - 1 / ( 1 + g ( x ) )  f (X) '= 1-1 / (1 + g (x))
であることを特徴とする請求の範囲第 6項に記載の音質調整装置の設計 7. The design of the sound quality adjusting device according to claim 6, wherein
9 . 複数の遅延器から成るタツプ付き遅延線における各夕ップの信号を フィルタ係数群によりそれぞれ数倍した後加算して出力するタイプのデ ジタルフィルタを用いた音質調整装置の設計装置であって、 9. A sound quality adjustment device using a digital filter of a type that multiplies each evening signal by a filter coefficient group and adds and outputs each signal in a tap delay line including a plurality of delay devices. hand,
所望の周波数特性の波形を表す数値列もしくは関数であって、 上記デ ジタルフィルタのタップ数よりも多いデータ点を有する数値列もしくは 関数を入力する入力手段と、  Input means for inputting a numerical sequence or a function representing a waveform of a desired frequency characteristic, the numerical sequence or the function having more data points than the number of taps of the digital filter;
上記入力手段により入力された数値列もしくは関数を逆フ一リエ変換 し、 その結果の実数項を抽出する逆フーリエ変換手段と、  Inverse Fourier transform means for performing an inverse Fourier transform on the numerical sequence or function input by the input means and extracting a real term of the result;
上記逆フーリエ変換により求められた数値列の前半部と後半部とを並 ベ替える並べ替え手段と、  Reordering means for rearranging the first half and the second half of the numerical sequence obtained by the inverse Fourier transform;
上記並べ替え手段により並べ替えられる前もしくは並べ替えられた後 の数値列に対して所定の窓関数を掛ける窓処理手段とを備え、  Window processing means for applying a predetermined window function to the numerical sequence before or after the sorting by the sorting means,
上記入力手段により入力された所望の周波数特性を表す数値列もしく は関数に対して上記逆フーリエ変換手段、 上記並べ替え手段および上記 窓処理手段の処理を行うことによって第 1 のフィルタのフィルタ係数群 を求めるとともに、 上記入力手段により入力された数値列もしくは関数 に所定の演算を行い、 その結果に対して上記逆フーリエ変換手段、 上記 並べ替え手段および上記窓処理手段の処理を行う ことによって、 ゲイン の基準値を軸として上記第 1 のフィルタと対称的な周波数特性を有する 第 2のフィル夕のフィルタ係数群を求めるようにしたことを特徵とする 音質調整装置の設計装置。 A numerical sequence or a numerical sequence representing the desired frequency characteristic input by the input means Obtains a filter coefficient group of the first filter by performing the inverse Fourier transform means, the rearranging means, and the window processing means on the function, and obtains a numerical value sequence or a function inputted by the input means. A predetermined operation is performed, and the result is processed by the inverse Fourier transform means, the rearranging means, and the window processing means, whereby a frequency symmetrical to the first filter with respect to a gain reference value as an axis is obtained. A sound quality adjustment device design apparatus characterized in that a filter coefficient group of a second filter having characteristics is obtained.
1 0 . 上記並べ替え手段により並べ替えられる前もしくは並べ替えられ た後の数値列、 あるいは、 上記窓処理手段により窓掛けが行われた後の 数値列を 2 π倍 （ ηは自然数） して小数点以下を丸め、 その結果を 1 / 2 η倍する処理を行う丸め手段を備え、 1 0. The rearrangement means by sorting is before or sorted numerical sequence after or,, 2 [pi times (eta is a natural number) the numerical sequence after the windowing is performed by the window processing means to A rounding means is provided for rounding the decimal portion and multiplying the result by 1/2 η ,
上記入力手段により入力された所望の周波数特性を表す数値列もしく は関数に対して上記逆フーリエ変換手段、 上記並べ替え手段、 上記窓処 理手段および上記丸め手段の処理を行う ことによって上記第 1 のフィル 夕のフィルタ係数群を求めるとともに、 上記入力手段により入力された 数値列もしくは関数に所定の演算を行い、 その結果に対して上記逆フー リエ変換手段、 上記並べ替え手段、 上記窓処理手段および上記丸め手段 の処理を行う ことによって上記第 2のフィルタのフィルタ係数群を求め るようにしたことを特徴とする請求の範囲第 9項に記載の音質調整装置 の設計装置。  By performing the processing of the inverse Fourier transform means, the rearranging means, the window processing means and the rounding means on a numerical sequence or a function representing a desired frequency characteristic inputted by the input means, In addition to obtaining a filter coefficient group for the filter of No. 1 and performing a predetermined operation on the numerical sequence or function input by the input means, the inverse Fourier transform means, the rearrangement means, and the window processing are performed on the result. 10. The sound quality adjustment device designing apparatus according to claim 9, wherein a filter coefficient group of the second filter is obtained by performing processing of the rounding means and the rounding means.
1 1 . 上記入力手段は、 周波数一ゲイン特性を表すための 2次元入力座 標上において上記所望の周波数特性の波形を描画するための手段と、 描 画された波形を上記数値列もしくは関数として入力するための手段とを 含むことを特徴とする請求の範囲第 9項に記載の音質調整装置の設計装 11. The input means includes means for drawing a waveform of the desired frequency characteristic on a two-dimensional input coordinate for representing a frequency-gain characteristic, and the drawn waveform as the numerical sequence or function. Means for inputting the sound quality of the sound quality adjusting device according to claim 9.
1 2 . 周波数軸お'よびゲイン軸の両軸が対数目盛りの 2次元平面におい て上記入力手段により入力した数値列もしくは関数を f ( X ) = 1 + g ( ) で表した場合に、 上記入力した数値列もしくは関数に対して行う 上記所定の演算は、 1 2. When the numerical sequence or function input by the above input means is represented by f (X) = 1 + g () on a two-dimensional plane in which both the frequency axis and the gain axis are logarithmic scales, The above predetermined operation to be performed on the input numerical sequence or function is
f ( ) ， = 1 - 1 / ( 1 + g ( X ) )  f (), = 1-1 / (1 + g (X))
であることを特徴とする請求の範囲第 9項に記載の音質調整装置の設計 装置。 10. The sound quality adjusting device designing apparatus according to claim 9, wherein:
1 3 . 請求の範囲第 6項に記載の各手段としてコンピュータを機能させ るための音質調整装置設計用プログラム。  13. A sound quality adjustment device designing program for causing a computer to function as each means described in claim 6.
1 4 . 請求の範囲第 9項に記載の各手段としてコンピュータを機能させ るための音質調整装置設計用プログラム。 '  14. A sound quality adjusting device design program for causing a computer to function as each means described in claim 9. '
1 5 . 請求の範囲第 1項に記載の設計方法を用いて設計された音質調整 装置。  15. A sound quality adjustment device designed using the design method according to claim 1.
1 6 . 請求の範囲第 3項に記載の設計方法を用いて設計された音質調整 装置。  16. A sound quality adjustment device designed using the design method according to claim 3.
1 7 . 請求の範囲第 6項に記載の設計装置を用いて設計された音質調整  17. Sound quality adjustment designed using the design device according to claim 6.
1 8 . 請求の範囲第 9項に記載の設計装置を用いて設計された音質調整 装置。 18. A sound quality adjustment device designed using the design device according to claim 9.
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