US8130854B2 - Nonlinear precoding method for a digital broadcast channel - Google Patents

Nonlinear precoding method for a digital broadcast channel Download PDF

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US8130854B2
US8130854B2 US10/564,832 US56483204A US8130854B2 US 8130854 B2 US8130854 B2 US 8130854B2 US 56483204 A US56483204 A US 56483204A US 8130854 B2 US8130854 B2 US 8130854B2
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matrix
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transmit
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US20060198459A1 (en
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Robert Fischer
Christoph Windpassinger
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Siemens AG
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/01—Equalisers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H04L25/497—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems by correlative coding, e.g. partial response coding or echo modulation coding transmitters and receivers for partial response systems
    • H04L25/4975—Correlative coding using Tomlinson precoding, Harashima precoding, Trellis precoding or GPRS
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/02—Arrangements for detecting or preventing errors in the information received by diversity reception
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/02—Arrangements for detecting or preventing errors in the information received by diversity reception
    • H04L1/06—Arrangements for detecting or preventing errors in the information received by diversity reception using space diversity
    • H04L1/0618—Space-time coding
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03343—Arrangements at the transmitter end
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/0342—QAM

Definitions

  • the invention relates to a nonlinear precoding method based on a modulo arithmetic for the transmit-side preequalization of K user signals to be transmitted at the same time and frequency in a digital broadcast channel with known transmission behavior set up between a central transmitting station and K decentralized, non-interconnected receiving stations, user signals consisting of data symbols a k with k from 1 to K from an M k -level signal constellation having a signal point spacing A k with a periodic multiple representation of the undisturbedly transmitted data symbols a k in data symbol intervals congruent for K receive-side modulo decision devices, a transmit-power-minimizing selection of representatives v k from the range of values a k +A k ⁇ M k ⁇ z kk where z kk is from the set of integers, and linear preequalization of the selected representatives v k to form transmit signals x k to be transmitted.
  • a plurality of user signals present at a common (i.e. central) transmitter are digitally transmitted to a plurality of decentralized (i.e. scattered over a service area) receivers (e.g. mobile stations).
  • Signal transmission user signal ⁇ receive signal is unidirectional in the downlink.
  • the particular feature of signal transmission in a broadcast channel is the lack of cooperability between the individual receivers. At no receiver are the signals of the other receivers known, and communication between the individual receivers is not possible. Consequently there can be no joint data processing of the receive signals in a central receiver. Transmission-improving signal conditioning can therefore only take place at the transmit side in the common transmitter.
  • Signal transmission can be wireline, but tends to be non-wireline.
  • CDMA Code Division Multiple Access
  • SDMA Space Division Multiple Access
  • MIMO system Multiple Input Multiple Output
  • multi-antenna systems are being increasingly used in which the signals are transmitted via a large number of transmitting antennas to a large number of receiving antennas, the numbers of antennas possibly being the same or different and having an impact on signal processing.
  • time and space diversity can be advantageously utilized in a MIMO system.
  • the problem arising from a plurality of receivers being supplied from a common transmitter is that the individual users are supplied not only with their own wanted signals, but that other users' signals are superimposed thereon, resulting in interference signals.
  • the occurrence of crosstalk interferences is synonymous with loss of the orthogonality which would be present in the case of ideal transmission behavior with decoupled subchannels.
  • On the transmit side it must therefore be attempted, knowing the user signals and the transmission conditions currently obtaining in the broadcast channel, i.e. the individual crosstalk factors between the individual users, to generate a suitable common transmit signal in such a way that each user receives his desired signal but without interference from the other signals.
  • the complex-valued elements h kI of the channel matrix H describe the couplings between the transmission paths I ⁇ k, i.e. the crosstalk of the user I onto the user k.
  • the ideal channel matrix H without couplings is a diagonal matrix, preferably an identity matrix (value 1 on the main diagonal).
  • the channel matrix H can be estimated by various known methods with backchannel or, in the case of duplexing with time division multiplex, also without backchannel and is assumed to be known at the central transmitter (presence of the so-called Channel State Information CSI).
  • H ⁇ 1 represents the inverse matrix to H, which can only be formed, however, if the transmission matrix is non-singular (determinant of the matrix is non-zero). It is achieved, however, that no interference signals are produced at the receivers and the data symbols a k appear directly (with only additive noise superimposed). There is therefore complete decoupling of the individual direct transmission paths k ⁇ k (orthogonality). However, the disadvantage of this procedure is the associated, in some cases very substantial, increase in the average transmit power required. This effect is greater the more strongly the matrix H ⁇ 1 tends to a singular matrix.
  • Precoding methods can be developed from the twin problem to this situation, i.e. the multiple access scenario (multiple access problem e.g. in the uplink transmit direction in which a plurality of distributed users access a common receiver).
  • nonlinear equalization can be performed by successive elimination of the interference signals which is implemented, for example, in the known V-BLAST method and can be termed Zero Forcing Decision Feedback Equalization (ZF-DFE) completely eliminating (Zero Forcing ZF) the interference signals.
  • ZF-DFE Zero Forcing Decision Feedback Equalization
  • An established preceding method is known according to Tomlinson and Harashima (THP—Tomlinson-Harashima Precoding) and is based on the use of modulo arithmetic. This procedure is described for the first time by M. Tomlinson in publication I “New Automatic Equaliser Employing Modulo Arithmetic” (Electronics Letters, vol. 7, Nos. 5/6, pp. 138-139, March 1971) and by H. Harashima and H.
  • This preequalization can be used in the central transmitter instead of receive-side feedback equalization which is only possible in the case of a central receiver.
  • THP operates on a nonlinear basis.
  • the output signal is constantly held between predefined modulo limits by a simple addition rule, by which the transmit power can be significantly reduced compared to linear methods.
  • This limiting is performed symbol-by-symbol without memory and is equivalently representable as the addition of a correction symbol which may assume an integral multiple of A k ⁇ M k .
  • the now apparently linear preequalization in this approach completely nullifies the channel distortion.
  • the multiple representation and selection of a suitable representative v k one more degree of freedom is therefore provided for signal processing.
  • the binary symbol “0” can, for example, be represented by .
  • the transmit-side nonlinear preprocessing can be derived from DFE and has, in mathematical terms, a unitary matrix F operated in the forward direction whose function is to transform the channel matrix into triangular form, and a matrix B present in the nonlinearly operating feedback loop in the form of a lower triangular matrix with unit main diagonal. If the overall channel matrix for the transmission behavior is of triangular form, the interference signals occurring can be precompensated bit by bit in the feedback branch of the central transmitter using modulo arithmetic. At the individual receivers, the data then appears as if the other users (with parallel transmission paths to the other receivers) did not exist.
  • each receiver the transmitted data symbols a k can be recovered or estimated values for them can be formed by threshold decision-making which takes account of the periodic continuation of the amplitude values or signal point spacings (modulo decision device).
  • threshold decision-making which takes account of the periodic continuation of the amplitude values or signal point spacings (modulo decision device).
  • the disadvantage of these preceding methods is that no “diversity gain” can be achieved because of the complete prevention of mutual interference signals.
  • Each transmission subsystem one user signal to the associated receiver
  • a diversity gain can in principle be achieved. If in the case of two transmission paths one of them has poor transmission conditions, it is highly probable that the other transmission path is quite usable.
  • the precoding method according to the invention for transmit-side, joint preprocessing of the user signals is based on the above described THP and returns to the idea of periodic continuation of the possible representatives for the individual data symbols a k , the mutually superimposed interference signals no longer being zero forced but being included by the joint preprocessing by assigning values from a precisely defined set of values to these interference values also, the values permitted for characterizing the interference signals being selected such that the receive-side modulo decision devices can still decide reliably on the transmitted data symbol a k even with the interference signals present.
  • the interference signals may now assume values which mean that a different representative v k from the possible set of representatives, but which represents the same data symbol a k , appears at the receiver, the range of values (a k +A k ⁇ M k z kk , where z kk is a positive or negative integer including zero) for the data symbols transmitted undisturbedly from user k to receiver k differing by just the original data symbol a k from the range of values (A k ⁇ M k ⁇ M k ⁇ z Ik , where z Ik is a positive or negative integer including zero) for the superimposed interference signals from user I to receiver k, the subscriber k being excluded.
  • interference signals present by producing a permissible shifting of the modulo-coded user signals into decision intervals for identical data symbols, are therefore taken into account and co-processed.
  • the periodic shifting means that the decision intervals are different, the result of the decision and its reliability are identical.
  • the mutual interference signals may therefore assume the values . . . ⁇ 4; ⁇ 2; 0; +2; +4; . . . (even numbers) and therefore be even-numbered.
  • the mapping of the interference signals to multiples of whole numbers also applies to any other selection of M k -level signal constellations.
  • the precoding according to the invention can also be applied to the quadrature amplitude modulation schemes (e.g. 4QAM or 16QAM) using complex number space.
  • the data vector a consists of data symbols a k (also known as signal points) from a complex-valued QAM alphabet.
  • the real parts of the symbols are transmitted with a cosine wave modulation and the imaginary parts with a sine wave modulation (quadrature mixing).
  • complex-valued channel descriptions complex-valued matrix entries
  • the channel matrix H r therefore attains twice the dimension (2K).
  • demodulators are provided which detect the voltage values phase-correctly and re-assign the real components real and imaginary part. Then finally transformation back to the complex-valued space again takes place.
  • the broadcast channel present is notionally subdivided into two sections.
  • the first part is completely equalized by preceding, the user signals are therefore decoupled, but a periodic continuation of the data symbols takes place.
  • the current representative v k for the data symbols a k is selected on an ongoing basis from the possible values which differ by integral multiples of the original level number M k , so that after linear preequalization of the selected representatives v k the required transmit power is minimal.
  • the second part of the channel is not equalized and therefore produces residual interference between the user signals.
  • the residual interference is so constituted that it does not adversely affect decision making in the receivers and, on the other, that equalization of the first part of the channel is possible with lower receive-side gain and therefore lower noise amplification or the diversity of the channel can be at least partially utilized.
  • the residual interference may assume values coinciding with the spacing of the possible representatives, the interference being reflected only in the (virtual) selection of a different representative, and its effect being completely eliminated in the modulo decision device already present.
  • the crucial advantage of the invention is the greatly increased power efficiency of signal transmission.
  • a lower bit error rate i.e. reliable reception, can be achieved at the same average transmit power as with the known precoding methods.
  • a diversity gain can be obtained which makes itself positively felt in a more rapid reduction in the bit error rate as the transmission quality of the broadcast channel improves.
  • the residual interference matrix R is only dependent on the current transmission behavior of the channel. As long as the channel matrix H (or H r ) does not change (burst transmission), the residual interference matrix R does not change either. On the main diagonal the residual interference matrix R is occupied by ones (direct signal paths), all the other elements assume row-wise only integral (positive or negative) multiples of the level number M k . In the case of binary transmission per component, these elements are only even (positive or negative) numbers:
  • Optimum decomposition methods must be selected for their complexity, which should be as low as possible, and for required receive-side gain that is as small as possible (e.g. by Automatic Gain Control AGC).
  • the known method specifically considering the case of two transmitting and two receiving antennas differs fundamentally from the invention in that there, in a multi-antenna system, all the user receive signals are known on the receive side and joint signal processing is possible.
  • the precoding method according to the invention relates to the problem in a multiuser system with exclusively downlink direction.
  • a common transmitter is present at which all the user signals are known and can be processed.
  • the receivers scattered over a service area cannot cooperate, i.e. each receiver sees only its own receive signal (no joint processing possible).
  • the known (partial) equalization takes place exclusively on the receive side on an exclusively linear basis, i.e. the reduced portion of the channel is equalized using the inverse channel matrix.
  • the invention operates exclusively nonlinearly on the transmit side on the basis of THP.
  • the preequalization method of the present invention is, on the other hand, designed for decentralized receivers that cannot cooperate, wherein lies a particular difficulty for signal processing, because it can only take place on the transmit side.
  • this reduced matrix is further factorized into suitable matrices.
  • F is a matrix with orthogonal columns
  • B is the lower triangular matrix
  • P a permutation matrix (each row and each column contains a single 1)
  • g the receive-side gain factor (automatic gain control). All three matrices and the scalar can be unambiguously determined from H red according to a predefined criterion (preferably minimum g).
  • FIG. 1 shows a broadcast channel
  • FIG. 2 shows the decoupling of the broadcast channel by a related art precoding method
  • FIG. 3 shows the broadcast channel with the preceding method according to the invention superimposed on it
  • FIG. 4 shows a block diagram of the preceding method according to the invention
  • FIG. 5 shows bit error curves for various equalization methods
  • FIG. 6 shows the gain factors for various equalization methods.
  • FIG. 1 schematically illustrates the structure of a broadcast channel BC for digital communication of K user signals ST k from a common, central transmitter CT (e.g. a base station) to K decentralized receivers DR k (e.g. mobile stations) which shall in each case only receive their own receive signal SR k and have no contact with the adjacent receivers DR k .
  • Transmission takes place exclusively in the downlink direction, non-wireline radio transmission being used in the case illustrated.
  • the broadcast channel BC considered in its entirety has a large number of inputs and a large number of outputs and can therefore be interpreted as a MIMO channel (Multiple Input Multiple Output).
  • a multiuser system is present here which must be differentiated from a multi-antenna system which likewise defines a MIMO channel.
  • THP Tomlinson-Harashima-Precoding
  • each data symbol a k (assigned to the user signals ST k ) a special value of an integral multiple of the product of the level number M k and the signal point spacing A k of the signal constellation (A k ⁇ M k ⁇ z where z is a positive or negative integer including zero) and the best value in respect of minimum transmit power is selected and the signal representative thus obtained is linearly preequalized.
  • THP is used on the transmit side to produce in the central transmitter CT a common transmit signal such that each decentralized receiver DR k receives its required receive signal SR k . Interference signals present are completely eliminated with this preceding method THP so that channel diversity cannot be used.
  • FIG. 3 illustrates the application of the nonlinear precoding method according to the invention, taking interference signals into account.
  • EIIP Extended-Integer Interference Precoding
  • FIG. 3 the basic principle of partial channel equalization on which EIIP is based can be clearly seen, whereby the broadcast channel BC is virtually converted into a reduced channel without coupling (first addition positions) which undergoes nonlinear precoding (shown in linearized form), and a superimposition of the suitably formed interference signals (second addition positions) is discriminated.
  • FIG. 4 shows the entire transmission system as it is provided in the proposed partially equalizing precoding method EIIP.
  • the channel matrix H denotes the actual transmission channel with K users.
  • all the transmit signals can be jointly accessed, which is indicated by a wide vector arrow.
  • the user signals y k with k from 1 . . . K are only processed singly, here indicated by individual scalar arrows.
  • On the receive side further noise n k is superimposed.
  • the receivers each consist only of a scaling device (Automatic Gain Control) and a threshold decision device (indicated in FIG.
  • the transmitter consists of the first three functional blocks. This involves a permutation matrix P T depending on the existing channel matrix H (or H red ), a feedback loop with a nonlinear modulo operation MOD, the identity matrix I and a lower triangular matrix B as well as a matrix F with orthogonal columns.
  • the data symbols to be transmitted (taken from a QAM alphabet) are combined in the K-dimensional vector a. Each receiver wishes to receive its data symbol a k (and that alone).
  • This vector with complex entries is first converted into a real vector (separation of the complex components into real and imaginary part as already described above), symbolized by the notation a/a r .
  • the further processing in the transmitter takes place on a real-value basis.
  • the transmitter produces transmit symbols, combined in the vector x r . These are then translated to a complex-valued representation (combination of real and imaginary part to form a complex number; reverse process as above), as the channel processes complex-valued input symbols.
  • the first stage of the transmitter is a permutation (re-sorting) of the components of the vector a r .
  • the next functional block is the nonlinearly operating feedback loop known in precoding methods.
  • the interference signals occurring during transmission over the channel are already pre-compensated.
  • Another unitary matrix F is applied which converts the general channel matrix into a lower triangular matrix without increasing the transmit power. Only thus can successive processing, as required, take place in the transmitter.
  • the matrices P, B and F are computed uniquely from the reduced form of the channel matrix as described above.
  • the precoding therefore equalizes only this reduced portion; the interference signals due to the residual interference matrix R (see above) remain.
  • the mode of operation of transmission is illustrated in the middle and bottom row in FIG. 4 .
  • First the preceding loop is replaced by its linearized representation.
  • the modulo operation is replaced by the addition of a correction term d.
  • the remaining, linear feedback loop (forward transmission One; feedback B-I) is then realized precisely by the matrix B ⁇ 1 (inverse matrix of B).
  • the channel matrix is represented, as described in the exemplary embodiments above, as a cascade of the reduced channel matrix H red and the residual interference matrix R. Because of the specific construction of the matrices B and F from H red , the cascade of B 1 , F and H red produces precisely the matrix P/g (again above equation), thereby producing the structure shown in the bottom row.
  • the permutation matrices P T and P cancel each other out; as transmission matrix, there therefore remains only the residual interference matrix R.
  • the main diagonal is one, the wanted signals are transmitted ideally.
  • the secondary diagonal elements which describe the crosstalk between the users, are even-numbered in the case of binary transmission; only even-numbered interferences therefore occur. However, this does not impair the existing modulo decision.
  • FIG. 5 shows the average bit error curves of the users for various signal processing methods.
  • the average bit error rate BER in each case is plotted against the ratio (expressed in dB) of the average transmit energy E b per information bit to the spectral power density N 0 of the additive noise.
  • the decomposition of the reduced channel matrix H red into g, F, B and P the same bit error characteristic is produced for both users.
  • FIG. 6 plots the gain factors g EIIP-PREC using nonlinear preceding according to the invention against the gain factors g PREC which arise with a real-valued preceding method not taking account of the interference (the representation is in dB as the inverse of the square, as the signal-to-noise ratio SNR is proportional to this term and this term directly describes the capability of the method).
  • the corresponding gain factors g EIIP-PREC are shown as the upper end of the bar. The length of the bar then indicates the achievable gain. Large gains are apparent particularly in situations in which the known methods produce very poor results.

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  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Physics & Mathematics (AREA)
  • Spectroscopy & Molecular Physics (AREA)
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DE10333514 2003-07-17
DE10333514A DE10333514B4 (de) 2003-07-17 2003-07-17 Nichtlineares Vorcodierungsverfahren für einen digitalen Broadcastkanal
DE10333514.5 2003-07-17
PCT/DE2004/001455 WO2005011219A1 (de) 2003-07-17 2004-07-07 Nichtlineares vorcodierungsverfahren für einen digitalen broadcastkanal

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