TWM547787U - Measuring device - Google Patents
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Description
本新型係關於一種量測裝置，特別是一種量測壓控振盪器相位雜訊用的量測裝置。The present invention relates to a measuring device, and more particularly to a measuring device for measuring phase noise of a voltage controlled oscillator.
壓控振盪器(voltage-controlled oscillator, VCO)被廣泛的利用於鎖相迴路等通訊系統中的電路內。對於壓控振盪器來說，其自由振盪(free-running)狀態下的相位雜訊是在應用壓控振盪器時，一個重要且必須依實際需求而被考量的特性。然而，傳統的相位雜訊分析儀無法精準的量到頻率飄移的相位雜訊。且在量測時，頻譜分析儀的頻率量測窗(span)不能設定太小。A voltage-controlled oscillator (VCO) is widely used in circuits in communication systems such as phase-locked loops. For a voltage-controlled oscillator, the phase noise in its free-running state is an important factor that must be considered in actual application when applying a voltage-controlled oscillator. However, conventional phase noise analyzers cannot accurately measure the phase noise of frequency drift. And when measuring, the frequency analyzer's frequency measurement window (span) cannot be set too small.
為了解決上述問題，在本領域中，安捷倫 (Agilent/Keysight)研發出型號為E5052B的分析儀。然而在應用E5052B分析儀時，需要待測的振盪器的頻率漂移低於中心頻率的0.4%才能實際量測。因此，於某些實施的環境中仍然無法將E5052B分析儀用於量測壓控振盪器在自由振盪狀態下的相位雜訊。In order to solve the above problems, in the field, Agilent/Keysight developed an analyzer model E5052B. However, when applying the E5052B analyzer, the frequency drift of the oscillator to be tested needs to be lower than 0.4% of the center frequency to actually measure. Therefore, the E5052B analyzer cannot be used to measure the phase noise of the voltage controlled oscillator in the free-running state in some implementation environments.
鑑於上述問題，本新型旨在提供一種量測裝置，以量測在自由振盪狀態下待測的壓控振盪器之相位雜訊。In view of the above problems, the present invention aims to provide a measuring device for measuring phase noise of a voltage controlled oscillator to be tested in a free-running state.
依據本新型一實施例的量測裝置，適於量測壓控振盪器(voltage-controlled oscillator, VCO)在第一頻率偏差值內的相位雜訊(phase noise, PN)，所述量測裝置相位偵測器與迴路濾波器。相位偵測器用以依據參考時脈信號與來自壓控振盪器的振盪信號，以產生相位差異信號，其中相位差異信號相較於參考時脈信號與振盪信號的差值具有第一倍率。迴路濾波器(loop-filter)分別電性連接相位偵測器與壓控振盪器，除了提供系統穩定度之外，並以第一低通頻寬將相位差異信號濾波以對壓控振盪器提供穩定的控制電壓。其中量測裝置與壓控振盪器的系統頻寬小於第一頻率偏差值，且系統頻寬係依據第一低通頻寬、第一倍率與壓控振盪器的頻率增益值的乘積所定義。The measuring device according to an embodiment of the present invention is adapted to measure phase noise (PN) of a voltage-controlled oscillator (VCO) within a first frequency deviation value, the measuring device Phase detector and loop filter. The phase detector is configured to generate a phase difference signal according to the reference clock signal and the oscillation signal from the voltage controlled oscillator, wherein the phase difference signal has a first magnification compared to the difference between the reference clock signal and the oscillation signal. The loop filter is electrically connected to the phase detector and the voltage controlled oscillator respectively, in addition to providing system stability, and filtering the phase difference signal with the first low pass bandwidth to provide the voltage controlled oscillator. Stable control voltage. The system bandwidth of the measuring device and the voltage controlled oscillator is smaller than the first frequency deviation value, and the system bandwidth is defined by a product of the first low pass bandwidth, the first power ratio and the frequency gain value of the voltage controlled oscillator.
本新型所揭示的量測裝置，藉由上述迴路濾波器的選擇，使得當壓控振盪器與量測裝置所構成的鎖相迴路鎖定時，其相位雜訊大致等於壓控振盪器在自由振盪狀態下的相位雜訊。The measuring device disclosed by the present invention, by the selection of the loop filter described above, makes the phase noise of the voltage controlled oscillator and the measuring device locked to be substantially equal to that of the voltage controlled oscillator when the phase locked loop formed by the voltage controlled oscillator and the measuring device is locked. Phase noise in the state.
以上之關於本揭露內容之說明及以下之實施方式之說明係用以示範與解釋本新型之精神與原理，並且提供本新型之專利申請範圍更進一步之解釋。The above description of the disclosure and the following description of the embodiments are intended to illustrate and explain the spirit and principles of the present invention, and to provide further explanation of the scope of the patent application of the present invention.
以下在實施方式中詳細敘述本新型之詳細特徵以及優點，其內容足以使任何熟習相關技藝者了解本新型之技術內容並據以實施，且根據本說明書所揭露之內容、申請專利範圍及圖式，任何熟習相關技藝者可輕易地理解本新型相關之目的及優點。以下之實施例係進一步詳細說明本新型之觀點，但非以任何觀點限制本新型之範疇。The detailed features and advantages of the present invention are described in detail in the following detailed description of the embodiments of the present invention. Any related art and related art can easily understand the related purposes and advantages of the present invention. The following examples are intended to describe the present invention in further detail, but do not limit the scope of the present invention in any way.
請參照圖1，其係依據本新型的量測裝置功能方塊圖。如圖1所示，量測裝置1000用來電性連接至待測物壓控振盪器2000上，並且量測裝置1000依據壓控振盪器2000的振盪信號V OSC來對壓控振盪器2000提供控制電壓V C。於一實施例中，量測裝置1000具有相位偵測器1100與迴路濾波器(loop-filter)1200。 Please refer to FIG. 1 , which is a functional block diagram of the measuring device according to the present invention. As shown in FIG. 1, the measuring device 1000 is electrically connected to the object voltage controlled oscillator 2000, and the measuring device 1000 provides control of the voltage controlled oscillator 2000 according to the oscillation signal V OSC of the voltage controlled oscillator 2000. Voltage V C . In one embodiment, the measurement device 1000 has a phase detector 1100 and a loop-filter 1200.
相位偵測器1100用以依據參考時脈信號CLK與振盪信號V OSC，以產生相位差異信號V PD。於一實施例中，當參考時脈信號CLK與振盪信號V OSC的邏輯值相同時，相位差異信號V PD的邏輯值為假(0)，當參考時脈信號CLK與振盪信號V OSC的邏輯值相異時，相位差異信號V PD的邏輯值為真(1)。於一實施例中，定義相位偵測器1100的增益(K PD)為相位差異信號V PD除以參考時脈信號CLK與振盪信號V OSC的差值，以下稱此增益為第一倍率。於一實施例中，相位差異信號V PD為一個脈衝信號(pulse signal)，而第一倍率定義為相位差異信號V PD的脈衝積(電壓*電流)除以參考時脈信號CLK與振盪信號V OSC的相位差值所得到的比值。 The phase detector 1100 is configured to generate the phase difference signal V PD according to the reference clock signal CLK and the oscillation signal V OSC . In one embodiment, when the reference clock signal CLK is the same as the logic value of the oscillation signal V OSC , the logic value of the phase difference signal V PD is false (0), when referring to the logic of the clock signal CLK and the oscillation signal V OSC When the values are different, the logical value of the phase difference signal V PD is true (1). In one embodiment, the gain (K PD ) of the phase detector 1100 is defined as the phase difference signal V PD divided by the difference between the reference clock signal CLK and the oscillating signal V OSC , which is hereinafter referred to as the first magnification. In one embodiment, the phase difference signal V PD is a pulse signal, and the first magnification is defined as a pulse product (voltage * current) of the phase difference signal V PD divided by the reference clock signal CLK and the oscillation signal V. The ratio obtained by the phase difference of the OSC .
迴路濾波器1200分別電性連接相位偵測器1100與壓控振盪器2000，以第一低通頻寬LBW1(Loop Bandwidth)將相位差異信號V PD濾波以對壓控振盪器2000提供控制電壓V C。其中，第一低通頻寬LBW1的調整需要使得系統頻寬小於壓控振盪器2000的輸出頻率偏差值(frequency offset)，舉例來說，若欲量測待測壓控振盪器2000在其以2.5吉赫(GHz)的頻率振盪時，在頻率偏差值為100兆赫(MHz)的範圍內(2.4GHz~2.6GHz)的相位雜訊，則調整第一低通頻寬LBW1使得系統頻寬小於100兆赫。於一實施例中，調整後的系統頻寬小於等於壓控振盪器2000的頻率偏差值的十分之一。其中，於本實施例中，系統頻寬係依據第一低通頻寬LBW1、壓控振盪器2000的頻率增益值(K VCO)與第一倍率所決定。 The loop filter 1200 is electrically connected to the phase detector 1100 and the voltage controlled oscillator 2000, respectively, and filters the phase difference signal V PD with a first low pass bandwidth LBW1 (Loop Bandwidth) to provide a control voltage V to the voltage controlled oscillator 2000. C. The adjustment of the first low-pass bandwidth LBW1 needs to make the system bandwidth smaller than the output frequency offset of the voltage controlled oscillator 2000. For example, if the voltage-controlled oscillator 2000 to be measured is to be measured, At a frequency oscillation of 2.5 GHz, the phase noise of the frequency deviation value of 100 megahertz (MHz) (2.4 GHz to 2.6 GHz) is adjusted, and the first low-pass bandwidth LBW1 is adjusted so that the system bandwidth is smaller than 100 MHz. In one embodiment, the adjusted system bandwidth is less than or equal to one tenth of the frequency deviation of the voltage controlled oscillator 2000. In this embodiment, the system bandwidth is determined according to the first low pass bandwidth LBW1, the frequency gain value (K VCO ) of the voltage controlled oscillator 2000, and the first multiple.
具體來說，於一實施例中，請參照圖2，其係依據本新型一實施例的迴路濾波器的功能方塊圖。如圖2所示，迴路濾波器1200具有低通濾波器LPF1、低通濾波器LPF2與低通濾波器LPF3。其中低通濾波器LPF1具有第一濾波頻寬例如為10千赫茲，低通濾波器LPF2具有第二濾波頻寬例如為100千赫茲，而低通濾波器LPF3具有第三濾波頻寬例如為1兆赫(MHz)。迴路濾波器1200並且具有選擇電路SEL。選擇電路SEL用以依據選擇信號，使低通濾波器LPF1至LPF3其中之一電性連接至相位偵測器1100與壓控振盪器2000以提供濾波功能。舉例來說，當使用者想要量測壓控振盪器2000在2.5吉赫(GHz)的振盪頻率下，於頻率偏差值為100兆赫(MHz)的範圍內的相位雜訊，則選擇信號例如由使用者所提供，指定選擇電路SEL將低通濾波器LPF2電性連接到相位偵測器1100與壓控振盪器2000。使得系統頻寬小於100兆赫，則如此，當壓控振盪器2000達到鎖定的狀態時，其相位雜訊相當於壓控振盪器2000在自由振盪(free-runnig)狀態下的相位雜訊。Specifically, in an embodiment, please refer to FIG. 2 , which is a functional block diagram of a loop filter according to an embodiment of the present invention. As shown in FIG. 2, the loop filter 1200 has a low pass filter LPF1, a low pass filter LPF2, and a low pass filter LPF3. The low pass filter LPF1 has a first filter bandwidth of, for example, 10 kHz, the low pass filter LPF2 has a second filter bandwidth of, for example, 100 kHz, and the low pass filter LPF3 has a third filter bandwidth of, for example, 1 Megahertz (MHz). The loop filter 1200 also has a selection circuit SEL. The selection circuit SEL is configured to electrically connect one of the low pass filters LPF1 to LPF3 to the phase detector 1100 and the voltage controlled oscillator 2000 according to the selection signal to provide a filtering function. For example, when the user wants to measure the phase noise of the voltage controlled oscillator 2000 at an oscillation frequency of 2.5 gigahertz (GHz) at a frequency offset value of 100 megahertz (MHz), the selection signal is, for example, Provided by the user, the designation selection circuit SEL electrically connects the low pass filter LPF2 to the phase detector 1100 and the voltage controlled oscillator 2000. When the system bandwidth is less than 100 MHz, then when the voltage controlled oscillator 2000 reaches the locked state, its phase noise is equivalent to the phase noise of the voltage controlled oscillator 2000 in the free-runn state.
於另一實施例中，迴路濾波器1200具有主動濾波器，依據相位差異信號V PD調整第一低通頻寬LBW1。具體來說，前述圖2中的選擇電路SEL更具有依據相位差異信號V PD來判斷壓控制盪器2000是否鎖定的能力。而在實作中，選擇電路SEL先選擇低通濾波器LPF3來對相位差異信號VPD進行濾波。由於低通濾波器LPF3所具有的頻寬為三個低通濾波器中最大的，因此壓控振盪器2000較容易、也較快鎖定。然後當判斷壓控振盪器2000的振盪信號已經被鎖定了，選擇電路SEL自動地切換到低通濾波器LPF2。若是在使用低通濾波器LPF2時能達到鎖定，則選擇電路SEL進一步切換使用低通濾波器LPF1。若是在使用低通濾波器LPF2時不能達到鎖定，則選擇電路SEL切換回使用低通濾波器LPF3。如此，量測者可以快速地得到初步地量測結果，並且進一步地可以得到精確地量測結果。 In another embodiment, the loop filter 1200 has an active filter that adjusts the first low pass bandwidth LBW1 according to the phase difference signal V PD . Specifically, the selection circuit SEL in FIG. 2 described above further has the ability to determine whether the voltage control device 2000 is locked according to the phase difference signal V PD . In practice, the selection circuit SEL first selects the low pass filter LPF3 to filter the phase difference signal VPD. Since the low pass filter LPF3 has the largest bandwidth among the three low pass filters, the voltage controlled oscillator 2000 is easier and faster to lock. Then, when it is judged that the oscillation signal of the voltage controlled oscillator 2000 has been locked, the selection circuit SEL is automatically switched to the low pass filter LPF2. If the lock can be achieved when the low pass filter LPF2 is used, the selection circuit SEL further switches the use of the low pass filter LPF1. If the lock cannot be achieved when the low pass filter LPF2 is used, the selection circuit SEL switches back to use the low pass filter LPF3. In this way, the measurer can quickly obtain the preliminary measurement result, and further can accurately obtain the measurement result.
於前述實施例中，迴路濾波器1200中的低通濾波器可以是由混合信號電路所構成，也可以是由類比電路所構成。舉例來說，以混合信號電路(mixed-signal circuit)所構成的低通濾波器可以由一個數位信號處理器(digital signal processor, DSP)與一個數位類比轉換器(digital-to-analog converter, DAC)來實現。於這樣的實施例中，選擇電路SEL係直接下達指令給數位信號處理器來調整其信號轉移函數(也就是頻寬)。In the foregoing embodiment, the low pass filter in the loop filter 1200 may be composed of a mixed signal circuit or an analog circuit. For example, a low-pass filter composed of a mixed-signal circuit can be composed of a digital signal processor (DSP) and a digital-to-analog converter (DAC). )to realise. In such an embodiment, the selection circuit SEL directly issues an instruction to the digital signal processor to adjust its signal transfer function (i.e., bandwidth).
而以類比電路構成的低通濾波器例如為由電阻與電容等被動電子元件所構成的低通濾波器，其構成與設計當為所屬領域舉有通常知識者所熟知，於此不再贅述。而選擇電路SEL切換各低通濾波器的方式係藉由實際上選擇性地將被選擇的低通濾波器電性連接至相位偵測器1100與壓控振盪器2000。The low-pass filter formed by the analog circuit is, for example, a low-pass filter composed of passive electronic components such as a resistor and a capacitor, and its configuration and design are well known to those skilled in the art, and will not be described herein. The selection circuit SEL switches the low pass filters by electrically connecting the selected low pass filter to the phase detector 1100 and the voltage controlled oscillator 2000.
於一實施例中，請參照圖3，其係依據本新型另一實施例的量測裝置功能方塊圖。如圖3所示，量測裝置1000A相較於量測裝置1000，更具有電荷泵浦(charge pump)1300，分別電性連接相位偵測器1100與迴路濾波器1200，電荷泵浦1300依據相位差異信號V PD對該迴路濾波器提供充放電電流信號I CP，並且迴路濾波器將充放電電流信號I CP濾波以產生控制電壓Vc。電荷泵浦1300用來依據相位差異信號V PD，來提供準確的充放電電流信號I CP。其架構為所屬領域具有通常知識者所熟析，於此不再另外描述。於此一實施例中，系統頻寬係依據第一低通頻寬LBW1、壓控振盪器2000的頻率增益值(K VCO)、第一倍率與信號增益所決定。其中，信號增益定義為充放電電流信號I CP與相位差益信號V PD的比值。具體來說，信號增益為充放電電流信號I CP的脈衝積與相位差益信號V PD的脈衝積比值。 In an embodiment, please refer to FIG. 3 , which is a functional block diagram of a measuring device according to another embodiment of the present invention. As shown in FIG. 3, the measuring device 1000A has a charge pump 1300 compared to the measuring device 1000, and is electrically connected to the phase detector 1100 and the loop filter 1200, respectively. The charge pump 1300 is phase-dependent. The difference signal V PD provides a charge and discharge current signal I CP to the loop filter, and the loop filter filters the charge and discharge current signal I CP to generate a control voltage Vc. The charge pump 1300 is used to provide an accurate charge and discharge current signal I CP based on the phase difference signal V PD . The architecture is well known to those of ordinary skill in the art and will not be described separately herein. In this embodiment, the system bandwidth is determined according to the first low pass bandwidth LBW1, the frequency gain value (K VCO ) of the voltage controlled oscillator 2000, the first power factor, and the signal gain. The signal gain is defined as the ratio of the charge and discharge current signal I CP to the phase difference signal V PD . Specifically, the signal gain is a pulse product ratio of the pulse product of the charge and discharge current signal I CP and the phase difference signal V PD .
於再一實施例中，請參照圖4，其係依據本新型再一實施例的量測裝置功能方塊圖。如圖4所示，量測裝置1000B相較於量測裝置1000更具有除頻器1400，分別電性連接壓控振盪器2000與相位偵測器1100，除頻器1400用以將振盪信號V OSC除頻，且相位偵測器1100係比較參考時脈信號CLK與被除頻的振盪信號，也就是信號V DIV，以產生相位差異信號VPD。具體來說，除頻器1400係將具有第一頻率的振盪信號V OSC除頻而產生具有第二頻率的振盪信號V DIV。通常第二頻率實質上等於參考時脈信號CLK的頻率。而除頻器1400例如整數除頻器、分數除頻器。其實現方式例如以正反器(D-flipflop, DFF)來實現，所屬領域具有通常知識者當能理解除頻器的實作方式，於此不再贅述。於此一實施例中，系統頻寬係依據第一低通頻寬LBW1、壓控振盪器2000的頻率增益值(K VCO)、第一倍率與除頻倍率所決定。其中，除頻倍率定義為信號V DIV與振盪信號V OSC的比值。具體來說，除頻倍率為信號V DIV的頻率與振盪信號V OSC的頻率的比值。 In still another embodiment, please refer to FIG. 4 , which is a functional block diagram of a measuring device according to still another embodiment of the present invention. As shown in FIG. 4, the measuring device 1000B has a frequency divider 1400 compared to the measuring device 1000, and is electrically connected to the voltage controlled oscillator 2000 and the phase detector 1100, respectively, and the frequency divider 1400 is used to oscillate the signal V. The OSC divides the frequency, and the phase detector 1100 compares the reference clock signal CLK with the frequency-divided oscillating signal, that is, the signal V DIV to generate a phase difference signal VPD. Specifically, the frequency divider 1400 divides the oscillating signal V OSC having the first frequency to generate an oscillating signal V DIV having a second frequency. Typically the second frequency is substantially equal to the frequency of the reference clock signal CLK. The frequency divider 1400 is, for example, an integer frequency divider and a fractional frequency divider. The implementation is implemented, for example, by a D-flipflop (DFF). Those skilled in the art can understand the implementation of the frequency divider, and details are not described herein. In this embodiment, the system bandwidth is determined according to the first low pass bandwidth LBW1, the frequency gain value (K VCO ) of the voltage controlled oscillator 2000, the first power ratio, and the frequency division ratio. Wherein, the frequency division ratio is defined as the ratio of the signal V DIV to the oscillation signal V OSC . Specifically, the frequency division ratio is a ratio of the frequency of the signal V DIV to the frequency of the oscillation signal V OSC .
於又一實施例中，請參照圖5，其係依據本新型又一實施例的量測裝置功能方塊圖。如圖5所示，量測裝置1000C相較於量測裝置1000更具有混頻器1500與低通濾波器1600。其中混頻器1500用以將振盪信號V OSC與載波信號V CON混頻來產生混頻信號V MIX。具體來說，混頻器1500例如為吉爾伯特乘法器(Gilbert Multiplier)。而載波信號V CON係依據參考時脈信號CLK所產生。 In still another embodiment, please refer to FIG. 5 , which is a functional block diagram of a measuring device according to still another embodiment of the present invention. As shown in FIG. 5, the measuring device 1000C has a mixer 1500 and a low pass filter 1600 as compared to the measuring device 1000. The mixer 1500 is configured to mix the oscillating signal V OSC with the carrier signal V CON to generate a mixed signal V MIX . Specifically, the mixer 1500 is, for example, a Gilbert Multiplier. The carrier signal V CON is generated according to the reference clock signal CLK.
低通濾波器1600用以對混頻信號V MIX濾波以產生轉換後的振盪信號V OSC’。由於混頻信號V MIX實際上具有兩個主要頻率，也就是載波信號的頻率與振盪信號的頻率和，以及載波信號的頻率與振盪信號的頻率差。而經過低通濾波器1600後，僅留下載波信號的頻率與振盪信號的頻率差的成分。也就是利用混頻與低通濾波的手段來快速的達到降頻的效果。如此，振盪信號V OSC’再經過放大器AMP放大後，或可直接被送入相位偵測器1100，或是更易於被除頻器1400所處理。於一實施例中，請參照圖6，其係依據本新型一實施例的量測裝置功能方塊圖。如圖6所示，量測裝置1000D可以具有前述圖1、圖3至圖5所提及的所有元件。其架構的連接方式如圖所示，所屬領域具有通常知識者於詳閱前述實施例並參酌圖6後當能理解，於此不再贅述。 The low pass filter 1600 is configured to filter the mixed signal V MIX to generate the converted oscillating signal V OSC '. Since the mixing signal V MIX actually has two main frequencies, that is, the frequency of the carrier signal and the frequency of the oscillating signal, and the frequency difference between the frequency of the carrier signal and the oscillating signal. After passing through the low pass filter 1600, only the component of the frequency difference between the frequency of the carrier signal and the oscillating signal is left. That is to use the means of mixing and low-pass filtering to quickly achieve the effect of frequency reduction. Thus, the oscillating signal V OSC ' may be amplified by the amplifier AMP or may be directly sent to the phase detector 1100 or may be more easily processed by the frequency divider 1400. In an embodiment, please refer to FIG. 6 , which is a functional block diagram of a measuring device according to an embodiment of the present invention. As shown in FIG. 6, the measuring device 1000D may have all of the elements mentioned in the foregoing FIGS. 1, 3 to 5. The connection manner of the architecture is as shown in the figure, and those skilled in the art can understand the foregoing embodiments and refer to FIG. 6 as will be understood, and details are not described herein.
於一實施例中，請回到圖1，量測裝置1000更具有頻譜分析儀1700，用以依據參考時脈信號CLK與振盪信號V OSC，分析壓控振盪器2000的相位雜訊。 In an embodiment, returning to FIG. 1 , the measuring device 1000 further has a spectrum analyzer 1700 for analyzing phase noise of the voltage controlled oscillator 2000 according to the reference clock signal CLK and the oscillation signal V OSC .
於另一實施例中，前述頻譜分析儀1700可以不屬於量測裝置1000。而是由量測裝置1000外接於頻譜分析儀。由量測裝置1000對頻譜分析儀提供參考時脈信號CLK，並由壓控振盪器2000對頻譜分析儀提供振盪信號。於另一實施例中，參考時脈信號CLK實際上是由頻譜分析儀1700(或量測裝置1000所外接的頻譜分析儀)所提供。In another embodiment, the aforementioned spectrum analyzer 1700 may not belong to the measurement device 1000. Instead, the measuring device 1000 is externally connected to the spectrum analyzer. The reference analyzer provides a reference clock signal CLK to the spectrum analyzer and provides an oscillating signal to the spectrum analyzer by the voltage controlled oscillator 2000. In another embodiment, the reference clock signal CLK is actually provided by the spectrum analyzer 1700 (or a spectrum analyzer external to the measuring device 1000).
為了理解本新型的理論基礎，請參照圖7，其係依據本新型一實施例中(例如圖6的架構)的鎖相迴路的線性模型。其中，前述的量測裝置1000與壓控振盪器2000結合起來即為鎖相迴路。而鎖相迴路的線性模型如圖7所示，其中，S CLK為時脈信號的雜訊，S PD為相位偵測器1100的雜訊，S LF為迴路濾波器1200的雜訊，S VCO為待測的壓控振盪器2000的雜訊，S CON為載波的雜訊，S MIX為混頻器的雜訊，S AMP為放大器的雜訊，S DIV為除頻器的雜訊，K P定義為相位偵測器1100的放大倍率，Z(s)定義為迴路濾波器1200的轉移函數。而推導出鎖相迴路的相位雜訊函數S OSC如下： In order to understand the theoretical basis of the present invention, reference is made to Figure 7, which is a linear model of a phase locked loop in accordance with an embodiment of the present invention (e.g., the architecture of Figure 6). Wherein, the foregoing measuring device 1000 and the voltage controlled oscillator 2000 are combined as a phase locked loop. The linear model of the phase-locked loop is shown in Figure 7, where S CLK is the noise of the clock signal, S PD is the noise of the phase detector 1100, S LF is the noise of the loop filter 1200, S VCO For the noise of the voltage controlled oscillator 2000 to be tested, S CON is the carrier noise, S MIX is the noise of the mixer, S AMP is the noise of the amplifier, S DIV is the noise of the frequency divider, K P is defined as the magnification of the phase detector 1100, and Z(s) is defined as the transfer function of the loop filter 1200. The phase noise function S OSC of the phase-locked loop is derived as follows:
其中系統增益G(s)定義為K PZ(s)(K VCO/s)/N，其中N為除頻器的倍率。則若選擇頻率偏差值遠大於系統增益G(s)的頻寬(系統頻寬)，則轉移函數Z(s)可以視為零，因此系統增益G(s)也可以視為零。於此，鎖相迴路的雜訊函數S OSC(f offset)可以大致等於壓控振盪器1200的相位雜訊S VCO。換句話說，於此狀況下，僅需量測由壓控振盪器2000與量測裝置1000構成的鎖相迴路的相位雜訊，即能得到壓控振盪器2000的相位雜訊。於實務上，前述所謂的遠大於，於本領域具有通常知識者當能理解其係依據實際需求而選擇，例如選擇頻率偏差值為系統頻寬的3倍、5倍、10倍乃至20倍。倍率越小，則所量測到的振盪信號V OSC的相位雜訊與實際上壓控振盪器2000在自由振盪狀態下的的相位雜訊的差異越大。而倍率越大，則鎖定所需要花費的時間越久。因此所屬領域具有通常知識者當能依照所需要的精確度與所能花費的量測時間，來適當地選擇其倍率。 The system gain G(s) is defined as K P Z(s)(K VCO /s)/N, where N is the override of the frequency divider. If the frequency deviation value is selected to be much larger than the bandwidth of the system gain G(s) (system bandwidth), the transfer function Z(s) can be regarded as zero, so the system gain G(s) can also be regarded as zero. Here, the noise function S OSC (f offset ) of the phase locked loop may be substantially equal to the phase noise S VCO of the voltage controlled oscillator 1200. In other words, in this case, only the phase noise of the phase-locked loop formed by the voltage controlled oscillator 2000 and the measuring device 1000 needs to be measured, that is, the phase noise of the voltage controlled oscillator 2000 can be obtained. In practice, the so-called so-called far greater than the ordinary knowledge in the field can understand that it is selected according to actual needs, for example, the frequency deviation value is selected to be 3 times, 5 times, 10 times or even 20 times the system bandwidth. The smaller the magnification, the greater the difference between the phase noise of the measured oscillating signal V OSC and the phase noise of the voltage controlled oscillator 2000 in the free oscillation state. The larger the magnification, the longer it takes to lock. Therefore, those skilled in the art can appropriately select the magnification according to the required accuracy and the measurement time that can be spent.
請參照圖8A與圖8B，其中圖8A係為本新型一實施例中的量測結果示意圖，圖8B為以本新型所提供之量測裝置進行頻譜量測的結果。其中待測的壓控振盪器型號為型號為Hittite HMC-C030 8-12.5GHz。如圖8A所示，其中曲線C1係由型號為SSA的頻譜分析儀量測的結果，曲線C2為依據本新型實施例中的量測裝置量測待測的壓控振盪器的結果，曲線C3係由型號為E4448A的頻譜分析儀量測的結果，曲線C4與曲線C5分別為理論上待測的壓控振盪器於不同溫度下的相位雜訊特性曲線。由圖8A可以看到，本新型的量測結果(曲線C2)實際至少在鎖定的頻率以上(右側)，很接近待測物的特性曲線(曲線C4與C5)、以及由安捷倫的量測裝置量測的結果曲線C1與C3。而由圖8B可以看到，於頻譜量測時，本新型所提供的量測裝置可以將頻率量測窗設定至100Hz，而毋需受限於振盪頻率。Please refer to FIG. 8A and FIG. 8B , wherein FIG. 8A is a schematic diagram of measurement results in an embodiment of the present invention, and FIG. 8B is a result of performing spectrum measurement by the measurement device provided by the present invention. The model of the voltage controlled oscillator to be tested is the Hittite HMC-C030 8-12.5 GHz. As shown in FIG. 8A, the curve C1 is the result measured by the spectrum analyzer of the type SSA, and the curve C2 is the result of measuring the voltage controlled oscillator to be tested according to the measuring device in the new embodiment, the curve C3 It is measured by the spectrum analyzer of model E4448A. Curve C4 and curve C5 are the phase noise characteristic curves of the voltage controlled oscillator theoretically tested at different temperatures. As can be seen from Fig. 8A, the measurement result (curve C2) of the present invention is actually at least above the locked frequency (right side), very close to the characteristic curve of the object to be tested (curves C4 and C5), and the measuring device by Agilent. The measured results are curves C1 and C3. As can be seen from FIG. 8B, in the spectrum measurement, the measuring device provided by the present invention can set the frequency measuring window to 100 Hz without being limited by the oscillation frequency.
綜上所述，所屬領域具有通常知識者當能明白，預量測一個壓控振盪器時，可以依據前述原則來選擇適當的迴路濾波器的頻寬，則可以量到壓控振盪器在自由振盪狀態下的相位雜訊。In summary, those skilled in the art can understand that when pre-measuring a voltage-controlled oscillator, the bandwidth of the appropriate loop filter can be selected according to the foregoing principles, and the voltage-controlled oscillator can be measured freely. Phase noise in an oscillating state.
雖然本新型以前述之實施例揭露如上，然其並非用以限定本新型。在不脫離本新型之精神和範圍內，所為之更動與潤飾，均屬本新型之專利保護範圍。關於本新型所界定之保護範圍請參考所附之申請專利範圍。Although the present invention has been disclosed above in the foregoing embodiments, it is not intended to limit the present invention. The changes and refinements of the present invention are within the scope of the patent protection of the present invention without departing from the spirit and scope of the present invention. Please refer to the attached patent application for the scope of protection defined by this new model.
1000、1000A~1000D‧‧‧量測裝置
1100‧‧‧相位偵測器
1200‧‧‧迴路濾波器
1300‧‧‧電荷泵浦
1400‧‧‧除頻器
1500‧‧‧混頻器
1600‧‧‧低通濾波器
1700‧‧‧頻譜分析儀
2000‧‧‧壓控振盪器
AMP‧‧‧放大器
VC‧‧‧控制電壓
VPD‧‧‧相位差異信號
KP‧‧‧相位偵測器的放大倍率
G(s)‧‧‧系統增益
ICP‧‧‧充放電電流信號
VOSC、VOSC’‧‧‧振盪信號
CLK‧‧‧參考時脈信號
VDIV‧‧‧信號
VMIX‧‧‧混頻信號
VCON‧‧‧載波信號
C1~C5‧‧‧曲線
SCLK、SPD、SLF、SVCO、SCON、SMIX、SAMP、SDIV‧‧‧雜訊1000, 1000A~1000D‧‧‧Measurement device
1100‧‧‧ phase detector
1200‧‧‧ Loop Filter
1300‧‧‧Charge pumping
1400‧‧‧Delephone
1500‧‧‧ Mixer
1600‧‧‧ low pass filter
1700‧‧‧ spectrum analyzer
2000‧‧‧Variable Control Oscillator
AMP‧‧Amplifier
V C ‧‧‧Control voltage
V PD ‧‧‧ phase difference signal
K P ‧‧‧ phase detector magnification
G(s)‧‧‧ system gain
I CP ‧‧‧Charge and discharge current signal
V OSC , V OSC '‧‧‧ oscillating signal
CLK‧‧‧ reference clock signal
V DIV ‧‧‧ signal
V MIX ‧‧‧mixing signal
V CON ‧‧‧Carrier signal
C1~C5‧‧‧ Curve
S CLK , S PD , S LF , S VCO , S CON , S MIX , S AMP , S DIV ‧‧‧ Noise
圖1係依據本新型的一量測裝置功能方塊圖。 圖2係依據本新型一實施例的迴路濾波器的功能方塊圖。 圖3係依據本新型另一實施例的量測裝置功能方塊圖。 圖4係依據本新型再一實施例的量測裝置功能方塊圖。 圖5係依據本新型又一實施例的量測裝置功能方塊圖。 圖6係依據本新型一實施例的量測裝置功能方塊圖。 圖7係依據本新型一實施例中的鎖相迴路的線性模型。 圖8A係為本新型一實施例中的量測結果示意圖。 圖8B為以本新型所提供之量測裝置進行頻譜量測的結果。1 is a functional block diagram of a measuring device according to the present invention. 2 is a functional block diagram of a loop filter in accordance with an embodiment of the present invention. 3 is a functional block diagram of a measuring device in accordance with another embodiment of the present invention. 4 is a functional block diagram of a measuring device in accordance with still another embodiment of the present invention. FIG. 5 is a functional block diagram of a measuring device according to still another embodiment of the present invention. 6 is a functional block diagram of a measuring device in accordance with an embodiment of the present invention. Figure 7 is a linear model of a phase locked loop in accordance with an embodiment of the present invention. FIG. 8A is a schematic diagram of measurement results in an embodiment of the present invention. Figure 8B is a result of spectral measurement by the measuring device provided by the present invention.
1000‧‧‧量測裝置 1000‧‧‧Measuring device
1100‧‧‧相位偵測器 1100‧‧‧ phase detector
1200‧‧‧迴路濾波器 1200‧‧‧ Loop Filter
1700‧‧‧頻譜分析儀 1700‧‧‧ spectrum analyzer
2000‧‧‧壓控振盪器 2000‧‧‧Variable Control Oscillator
VC‧‧‧控制電壓 V C ‧‧‧Control voltage
VPD‧‧‧相位差異信號 V PD ‧‧‧ phase difference signal
VOSC‧‧‧振盪信號 V OSC ‧‧‧Oscillation signal
CLK‧‧‧參考時脈信號 CLK‧‧‧ reference clock signal
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| CN108362941A (en) * | 2018-03-29 | 2018-08-03 | 珠海迈科智能科技股份有限公司 | A kind of device and method of test Tuner module crystal oscillator frequency deviations |
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| CN108362941A (en) * | 2018-03-29 | 2018-08-03 | 珠海迈科智能科技股份有限公司 | A kind of device and method of test Tuner module crystal oscillator frequency deviations |
| CN108362941B (en) * | 2018-03-29 | 2023-07-18 | 珠海迈科智能科技股份有限公司 | Equipment and method for testing frequency deviation of Tuner module crystal oscillator |
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