TW202308279A - High voltage conversion ratio dc converter - Google Patents
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本發明係有關於一種高電壓轉換比直流轉換器,尤其是指一種具有高升壓特性,但是不必操作在極大導通比;高輸出電壓時,功率開關具有遠低於輸出電壓的低電壓應力,可以降低功率開關的導通損失;高輸入電流應用時,具有低輸入電流漣波;而且漏電感能改善二極體的反向恢復損失,使得其適合高升壓、高效率和高功率之應用,而在其整體施行使用上更增實用功效特性者。The present invention relates to a DC converter with a high voltage conversion ratio, in particular to a DC converter with high boost characteristics, but does not have to operate at a large conduction ratio; when the output voltage is high, the power switch has a low voltage stress far below the output voltage, It can reduce the conduction loss of the power switch; when high input current is applied, it has low input current ripple; and the leakage inductance can improve the reverse recovery loss of the diode, making it suitable for high boost, high efficiency and high power applications. And in its overall implementation and use, it has more practical and functional characteristics.
按,《巴黎協定》希望各國透過再生能源,用更經濟、有效的方式達成減排目標,追求經濟的「綠色成長」。爰此,再生能源的利用必定是各國產業發展的重點方向,包含太陽能、風力能、水力能、地熱能、潮汐能、生質能及燃料電池等。例如在歐洲、日本與美國裝設於屋頂的住宅型太陽能併網電力系統,最近成為成長快速的市場。在再生能源電力系統應用中,太陽能發電系統及燃料電池發電系統的技術發展越來越成熟,常常在分散式發電系統[distributed generation system]扮演重要的角色。According to the Paris Agreement, countries are expected to use renewable energy to achieve emission reduction targets in a more economical and effective way, and to pursue economic "green growth". Therefore, the utilization of renewable energy must be the key direction of industrial development in various countries, including solar energy, wind energy, hydraulic energy, geothermal energy, tidal energy, biomass energy and fuel cells. For example, residential solar grid-connected power systems installed on rooftops in Europe, Japan, and the United States have recently become a fast-growing market. In the application of renewable energy power system, the technological development of solar power generation system and fuel cell power generation system is becoming more and more mature, and they often play an important role in distributed generation system [distributed generation system].
由於住宅型應用[residential applications]的安全性與可靠性的問題,太陽能電池模組與燃料電池所產生的輸出電壓是屬於低電壓,一般不超過 ,為了達到併網發電系統或直流微電網的需求,必須先將此低電壓利用高升壓DC-DC轉換器,升壓至一個高直流排電壓。例如:對於一個單相 的電網系統而言,此高直流排電壓常為 ,以利全橋換流器[inverter]的DC-AC轉換。理論上,操作在極高導通比的傳統升壓型[boost]轉換器能夠得到高電壓增益,但是實務上受到寄生元件的影響,電壓轉換比受限在約5倍以下,因此當電壓增益超過5倍的需求時,研發嶄新的高升壓轉換器拓樸是必要的。因此近幾年高升壓DC-DC轉換器是電力電子工程領域中常見的研究主題之一。 Due to the safety and reliability of residential applications, the output voltage generated by solar cell modules and fuel cells is low voltage, generally not exceeding , in order to meet the needs of grid-connected power generation systems or DC microgrids, this low voltage must first be boosted to a high DC row voltage by using a high-boost DC-DC converter. Example: For a single phase For the power grid system, this high DC row voltage is often , to facilitate the DC-AC conversion of the full-bridge converter [inverter]. Theoretically, a traditional step-up [boost] converter operating at a very high conduction ratio can obtain a high voltage gain, but in practice, due to the influence of parasitic elements, the voltage conversion ratio is limited to less than about 5 times, so when the voltage gain exceeds 5 times the demand, it is necessary to develop a new high-boost converter topology. Therefore, high-boost DC-DC converters are one of the common research topics in the field of power electronics engineering in recent years.
請參閱第二十三圖現有之傳統升壓型轉換器電路圖所示,該升壓型轉換器(2)電路中 為電感的等效串聯電阻,當考慮理想元件[ ]且操作在連續導通模式[CCM]模式時,其輸出電壓增益 Please refer to the existing traditional boost converter circuit diagram shown in Figure 23, in the boost converter (2) circuit is the equivalent series resistance of the inductor, when considering ideal components[ ] and operating in the continuous conduction mode [CCM] mode, its output voltage gain
電壓增益完全決定於開關導通比[duty ratio] 。理論上要得到高電壓增益,轉換器必須操作在極大導通比;但是實務上,由於寄生元件的存在,例如 ,則電壓增益 與效率 對導通比的表示式分別為 The voltage gain is completely determined by the switch conduction ratio [duty ratio] . Theoretically, to obtain a high voltage gain, the converter must operate at a very large conduction ratio; but in practice, due to the existence of parasitic elements, such as , then the voltage gain and efficiency The expressions for the conduction ratio are
請再參閱第二十四圖現有之傳統升壓型轉換器的輸出電壓增益對開關導通比的關係曲線圖及第二十五圖現有之傳統升壓型轉換器的效率對開關導通比的關係曲線圖所示,可知操作在極大導通比的轉換器電壓增益是有所限制,而且轉換效率不佳,另外操作在極大導通比的升壓型轉換器衍生了以下問題:容易產生很大的輸入電流漣波,使得太陽能電池模組輸出端的電解電容數量必須增加,減少燃料電池的使用壽命;此外,輸出二極體的反向恢復損失相當大。Please refer to Figure 24 for the relationship between the output voltage gain of the existing conventional boost converter and the switch conduction ratio and Figure 25 for the relationship between the efficiency of the conventional boost converter and the switch conduction ratio As shown in the graph, it can be seen that the voltage gain of the converter operating at a very large conduction ratio is limited, and the conversion efficiency is not good. In addition, the boost converter operating at a very large conduction ratio has the following problems: it is easy to generate a large input Due to the current ripple, the number of electrolytic capacitors at the output of the solar cell module must be increased, reducing the service life of the fuel cell; in addition, the reverse recovery loss of the output diode is quite large.
使得為了適合高功率應用及降低輸入電流漣波的特性,請參閱第二十六圖現有之交錯式升壓型轉換器電路圖所示,即有業者發展出交錯式升壓型轉換器(3),然而該交錯式升壓型轉換器(3)之功率開關仍需承受高電壓應力,其導通損失會隨開關導通比增大而增加;因此研發交錯式DC-DC轉換器拓樸具有高升壓特性,但是不必操作在極大開關導通比,改善二極體的反向恢復損失問題,是重要的考量。In order to be suitable for high-power applications and reduce the characteristics of input current ripple, please refer to the existing interleaved boost converter circuit diagram in Figure 26, that is, the industry has developed an interleaved boost converter (3) , however, the power switch of the interleaved boost converter (3) still needs to withstand high voltage stress, and its conduction loss will increase with the increase of the switch conduction ratio; therefore, the research and development of the interleaved DC-DC converter topology has high Voltage characteristics, but it is not necessary to operate at a maximum switch conduction ratio, and it is an important consideration to improve the reverse recovery loss of the diode.
再者,典型交錯式升壓型轉換器之開關電壓應力為高壓的輸出電壓,由於高耐壓的MOSFET,一般都具有高導通電阻 的特性,導致較高的導通損失;因此在開關成本、導通電阻、耐壓限制與轉換效率的考量之下,研發高升壓DC-DC轉換,而功率開關具有低電壓應力,是另一個重要的考量。 Furthermore, the switching voltage stress of a typical interleaved boost converter is a high-voltage output voltage. Due to the high withstand voltage MOSFET, it generally has a high on-resistance Therefore, under the consideration of switching cost, on-resistance, withstand voltage limit and conversion efficiency, it is another important to develop high-boost DC-DC conversion, and the power switch has low voltage stress. considerations.
緣是,發明人有鑑於此,秉持多年該相關行業之豐富設計開發及實際製作經驗,針對現有之結構及缺失再予以研究改良,提供一種高電壓轉換比直流轉換器,以期達到更佳實用價值性之目的者。The reason is that, in view of this, the inventor has been adhering to years of rich experience in design, development and actual production in this related industry, and then researched and improved the existing structure and defects to provide a high-voltage conversion ratio DC converter in order to achieve better practical value. sexual purpose.
本發明之主要目的在於提供一種高電壓轉換比直流轉換器,主要係具有高升壓特性,但是不必操作在極大導通比;高輸出電壓時,功率開關具有遠低於輸出電壓的低電壓應力,可以降低功率開關的導通損失;高輸入電流應用時,具有低輸入電流漣波;而且漏電感能改善二極體的反向恢復損失,使得其適合高升壓、高效率和高功率之應用,而在其整體施行使用上更增實用功效特性者。The main purpose of the present invention is to provide a high voltage conversion ratio DC converter, which mainly has high boost characteristics, but does not need to operate at a large conduction ratio; when the output voltage is high, the power switch has a low voltage stress far below the output voltage, It can reduce the conduction loss of the power switch; when high input current is applied, it has low input current ripple; and the leakage inductance can improve the reverse recovery loss of the diode, making it suitable for high boost, high efficiency and high power applications. And in its overall implementation and use, it has more practical and functional characteristics.
為令本發明所運用之技術內容、發明目的及其達成之功效有更完整且清楚的揭露,茲於下詳細說明之,並請一併參閱所揭之圖式及圖號:In order to have a more complete and clear disclosure of the technical content used in the present invention, the purpose of the invention and the effects achieved, it will be described in detail below, and please also refer to the disclosed drawings and drawing numbers:
首先,請參閱第一圖本發明之電路圖及第二圖本發明之等效電路圖所示,本發明之轉換器(1)主要係於輸入電壓 之正極分別連接第一耦合電感一次側 之第一端及第二耦合電感一次側 之第一端,該第一耦合電感一次側 形成有第一磁化電感 ,該第二耦合電感一次側 形成有第二磁化電感 ,於該第一耦合電感一次側 之第二端分別連接有第一功率開關 之第一端、第一輸出電容 之第一端及第二輸出電容 之第二端,且於該第一耦合電感一次側 之第二端與該第一功率開關 之第一端、該第一輸出電容 之第一端及該第二輸出電容 之第二端之間形成有第一漏電感 ,而該第二耦合電感一次側 之第二端分別連接有第二功率開關 之第一端及第二輸出二極體 之正極,並於該第二耦合電感一次側 之第二端與該第二功率開關 之第一端及第二輸出二極體 之正極之間形成有第二漏電感 ,該輸入電壓 之負極分別連接該第二功率開關 之第二端、該第一功率開關 之第二端及第一輸出二極體 之負極,該第二輸出二極體 之負極分別連接該第二輸出電容 之第一端、第一舉升電容 之第一端及第二耦合電感二次側 之第一端,該第二耦合電感二次側 之第二端連接第一耦合電感二次側 之第二端,該第一耦合電感二次側 之第一端分別連接第二舉升電容 之第一端及第一舉升二極體 之正極,該第一舉升電容 之第二端分別連接該第一舉升二極體 之負極及第二舉升二極體 之正極,該第二舉升電容 之第二端分別連接該第二舉升二極體 之負極及輸出二極體 之正極,該輸出二極體 之負極分別連接輸出電容 之第一端及負載 之第一端,該第一輸出二極體 之正極則分別連接該第一輸出電容 之第二端、該輸出電容 之第二端及該負載 之第二端。 First of all, please refer to the circuit diagram of the present invention shown in the first figure and the equivalent circuit diagram of the present invention in the second figure, the converter (1) of the present invention mainly depends on the input voltage The positive poles are respectively connected to the primary side of the first coupled inductor The first end and the primary side of the second coupled inductor The first end, the primary side of the first coupled inductor formed with a first magnetizing inductance , the second coupled inductor primary side formed with a second magnetizing inductance , on the primary side of the first coupled inductor The second end is respectively connected with the first power switch The first terminal, the first output capacitor The first terminal and the second output capacitor The second terminal, and on the primary side of the first coupled inductor The second terminal of the first power switch The first end of the first output capacitor The first terminal and the second output capacitor A first leakage inductance is formed between the second terminal of , and the primary side of the second coupled inductor The second end is respectively connected with the second power switch The first terminal and the second output diode positive pole, and on the primary side of the second coupled inductor The second terminal and the second power switch The first terminal and the second output diode A second leakage inductance is formed between the positive poles , the input voltage The negative poles of are respectively connected to the second power switch The second end of the first power switch The second terminal and the first output diode the negative pole of the second output diode The negative poles of the two are respectively connected to the second output capacitor The first terminal, the first lift capacitor The first terminal and the secondary side of the second coupled inductor The first end of the second coupled inductor secondary side The second terminal is connected to the secondary side of the first coupled inductor The second terminal, the secondary side of the first coupled inductor The first end of each is connected to the second lift capacitor The first terminal and the first lifting diode positive terminal, the first lift capacitor The second terminal is connected to the first lifting diode The negative electrode and the second lifting diode positive terminal, the second lift capacitor The second terminal is connected to the second lifting diode Negative pole and output diode the positive terminal, the output diode The negative poles are respectively connected to the output capacitor The first terminal and the load The first terminal, the first output diode The positive poles are respectively connected to the first output capacitor The second terminal, the output capacitor The second terminal and the load the second end.
而該轉換器(1)在連續導通模式[CCM]中,為了達到高升壓性能,導通比大於0.5,而且該第一功率開關 和該第二功率開關 以工作相位相差半切換週期的交錯式操作。穩態分析時,根據該轉換器(1)各功率開關及各二極體的ON/OFF狀態,該轉換器(1)在一個切換週期內可分成9個線性操作階段,假設: In the continuous conduction mode [CCM] of the converter (1), in order to achieve high boost performance, the conduction ratio is greater than 0.5, and the first power switch and the second power switch Interleaved operation with working phase difference of half switching period. In the steady-state analysis, according to the ON/OFF state of each power switch and each diode of the converter (1), the converter (1) can be divided into 9 linear operation stages within one switching cycle, assuming:
1.所有功率半導體元件[各開關及各二極體]均為理想,即導通壓降為零。1. All power semiconductor components [switches and diodes] are ideal, that is, the conduction voltage drop is zero.
2.各電容 、 、 與 與 夠大,各電容電壓 、 、 、 和 可視為定電壓,因此輸出電壓 可視為常數。 2. Capacitors , , and and large enough that each capacitor voltage , , , and can be regarded as a constant voltage, so the output voltage can be regarded as a constant.
3.兩個耦合電感的匝數比相等[ ]且磁化電感值相等[ ],漏電感值相等[ ],磁化電感遠大於漏電感,耦合電感的耦合係數 。 3. The turns ratio of the two coupled inductors is equal [ ] and the magnetizing inductance value is equal to [ ], the leakage inductance value is equal to [ ], the magnetizing inductance is much larger than the leakage inductance, the coupling coefficient of the coupled inductance .
4.耦合電感的磁化電感電流操作在連續導通模式[Continuous Conduction Mode,CCM]。4. The magnetizing inductance current of the coupled inductor operates in continuous conduction mode [Continuous Conduction Mode, CCM].
其各線性階段線性等效電路以及主要元件波形如下所示,請再一併參閱第三圖本發明之主要元件時序波形圖所示:The linear equivalent circuit of each linear stage and the waveform of the main components are as follows, please refer to the third figure together with the timing waveform diagram of the main components of the present invention:
第一階段[ ]:[第一功率開關 :OFF→ON、第二功率開關 :ON、第一輸出二極體 :OFF、第二輸出二極體 :OFF、第一舉升二極體 :OFF、第二舉升二極體 :ON、輸出二極體 :OFF]:請再一併參閱第四圖本發明之第一操作階段等效電路圖所示,第一階段開始於 ,第一功率開關 切換成ON,且第二功率開關 仍保持ON,第一輸出二極體 、第二輸出二極體 、第一舉升二極體 、輸出二極體 均為逆向偏壓,由於第一漏電感 的存在,第一功率開關 具有零電流切換[ZCS]的柔切性能,降低切換損失。第一漏電感電流 上升,當 時,第一磁化電感 所儲存的能量仍然藉由耦合電感傳送至二次側,第二舉升二極體 保持導通狀態,第二舉升二極體電流 下降。第一漏電感 控制了第二舉升二極體電流 的下降速率,緩和了第二舉升二極體 反向恢復問題。當 ,第二舉升二極體電流 下降至0,第二舉升二極體 轉態成OFF時,本階段結束。 The first stage[ ]: [First power switch : OFF→ON, second power switch : ON, the first output diode : OFF, second output diode : OFF, the first lift diode : OFF, second lift diode : ON, output diode : OFF]: Please refer to Figure 4 again, as shown in the equivalent circuit diagram of the first operation stage of the present invention, the first stage begins at , the first power switch switches ON, and the second power switch remains ON, the first output diode , the second output diode , the first lifting diode , output diode are reverse biased, due to the first leakage inductance presence of the first power switch It has soft cutting performance of zero current switching [ZCS] and reduces switching loss. The first leakage inductance current rise when , the first magnetizing inductance The stored energy is still transmitted to the secondary side through the coupled inductor, the second lift diode remains on, the second lift-up diode current decline. first leakage inductance controls the second lift diode current rate of fall, moderated by the second lift diode Reverse recovery problem. when , the second lift diode current drops to 0, the second lift diode When the state is turned OFF, this phase ends.
第二階段[ ]:[第一功率開關 :ON、第二功率開關 :ON、第一輸出二極體 :OFF、第二輸出二極體 :OFF、第一舉升二極體 :OFF、第二舉升二極體 :ON→OFF、輸出二極體 :OFF]:請再一併參閱第五圖本發明之第二操作階段等效電路圖所示,第二階段開始於 ,第二舉升二極體 轉態成OFF,所有二極體均為逆向偏壓而OFF,第一功率開關 、第二功率開關 皆為ON。輸入電壓 跨於兩個耦合電感的一次側,即跨於第一磁化電感 和第一漏電感 以及第二磁化電感 和第二漏電感 ,第一漏電感電流 和第二漏電感電流 呈線性上升,斜率均為 ,從能量觀點而言,兩個耦合電感的一次側在本階段作儲存能量的動作。當 ,第二功率開關 切換成OFF時,本階段結束。 second stage[ ]: [First power switch : ON, second power switch : ON, the first output diode : OFF, second output diode : OFF, the first lift diode : OFF, second lift diode : ON→OFF, output diode : OFF]: Please refer to Figure 5 again, as shown in the equivalent circuit diagram of the second operation stage of the present invention, the second stage begins at , the second lift diode Turning to OFF, all diodes are reverse biased and OFF, the first power switch , the second power switch All are ON. Input voltage Across the primary sides of the two coupled inductors, that is, across the first magnetizing inductance and the first leakage inductance and the second magnetizing inductance and the second leakage inductance , the first leakage inductance current and the second leakage inductance current rises linearly, with a slope of , from an energy point of view, the primary side of the two coupled inductors stores energy at this stage. when , the second power switch When switched to OFF, this phase ends.
第三階段[ ]:[第一功率開關 :ON、第二功率開關 :ON→OFF、第一輸出二極體 :OFF、第二輸出二極體 :ON、第一舉升二極體 :ON、第二舉升二極體 :ON、輸出二極體 :ON]:請再一併參閱第六圖本發明之第三操作階段等效電路圖所示,第三階段開始於 ,第二功率開關 切換為OFF,第一功率開關 保持為ON,第一輸出二極體 和第二舉升二極體 為逆向偏壓。第二漏電感電流 的連續性使得第二輸出二極體 轉態為ON,第二漏電感電流 流經第二輸出二極體 、第二輸出電容 和第一功率開關 ,對第二輸出電容 充電。耦合電感之第二磁化電感 以返馳式模式傳送能量至二次側,使得第一舉升二極體 轉態為ON。第一舉升二極體電流 對第一舉升電容 充電,此階段輸出二極體 導通,第二舉升電容 對輸出電容 對負載 放電,此時第二漏電感電流 呈線性下降。當 ,第二漏電感電流 線性下降,當 ,第二輸出電容 轉為向上流動,本階段結束。 The third phase[ ]: [First power switch : ON, second power switch : ON→OFF, the first output diode : OFF, second output diode : ON, the first lifting diode : ON, the second lifting diode : ON, output diode : ON]: Please refer to Figure 6 again, as shown in the equivalent circuit diagram of the third operation stage of the present invention, the third stage begins at , the second power switch switch to OFF, the first power switch remains ON, the first output diode and a second boost diode for reverse bias. Second leakage inductance current The continuity makes the second output diode transition to ON, the second leakage inductance current flows through the second output diode , the second output capacitor and the first power switch , for the second output capacitor Charge. The second magnetizing inductance of the coupled inductor transfers energy to the secondary in flyback mode such that the first lift diode Turn the state to ON. 1st lift diode current to the first lift capacitor charging, the output diode at this stage turn on, the second boost capacitor to the output capacitor on load discharge, at this time the second leakage inductance current Decreases linearly. when , the second leakage inductance current declines linearly, when , the second output capacitor Turn to upward flow and this phase ends.
第四階段[ ]:[第一功率開關 :ON、第二功率開關 :OFF、第一輸出二極體 :OFF、第二輸出二極體 :ON、第一舉升二極體 :ON、第二舉升二極體 :ON、輸出二極體 :ON]:請再一併參閱第七圖本發明之第四操作階段等效電路圖所示,第四階段開始於 ,第二功率開關 保持OFF,第一功率開關 保持ON,第一輸出二極體 和第二舉升二極體 逆向偏壓。與第三階段比較,除了第二輸出電容 的流向不同,其餘的電路操作相同。第二漏電感電流 持續線性下降,此階段輸出電容 和第一舉升電容 為儲存能量,第二舉升電容 為釋放能量。當 ,第二漏電感 儲存的能量完全釋放完畢,即 ,第二輸出二極體 自然轉態成OFF,本階段結束。 The fourth stage [ ]: [First power switch : ON, second power switch : OFF, the first output diode : OFF, second output diode : ON, the first lifting diode : ON, the second lifting diode : ON, output diode : ON]: Please refer to Figure 7 again, as shown in the equivalent circuit diagram of the fourth operation stage of the present invention, the fourth stage begins at , the second power switch kept OFF, the first power switch remains ON, the first output diode and a second boost diode reverse bias. compared with the third stage, except that the second output capacitor The flow direction is different, and the rest of the circuit operates the same. Second leakage inductance current continuous linear decline, at this stage the output capacitance and the first lift capacitor To store energy, the second boost capacitor to release energy. when , the second leakage inductance The stored energy is completely released, that is, , the second output diode Natural transition to OFF, this phase ends.
第五階段[ ]:[第一功率開關 :ON、第二功率開關 :OFF、第一輸出二極體 :OFF、第二輸出二極體 :ON→OFF、第一舉升二極體 :ON、第二舉升二極體 :OFF、輸出二極體 :ON]:請再一併參閱第八圖本發明之第五操作階段等效電路圖所示,第五階段開始於 ,此時第二漏電感 的能量釋放完畢,第二輸出二極體 轉態成OFF。第二磁化電感電流 完全由耦合電感之一次側反射到二次側,第一舉升二極體電流 對第一舉升電容 充電,第二舉升電容 對輸出電容 和負載 釋放能量,此時第一功率開關 的電流等於第一磁化電感 和第二磁化電感 的電流總和。當 ,第二功率開關 切換為ON時,本階段結束。 fifth stage [ ]: [First power switch : ON, second power switch : OFF, the first output diode : OFF, second output diode : ON→OFF, the first lifting diode : ON, the second lifting diode : OFF, output diode : ON]: Please refer to the eighth figure and the equivalent circuit diagram of the fifth operation stage of the present invention, the fifth stage starts at , at this time the second leakage inductance After the energy is released, the second output diode Turn the state to OFF. second magnetizing inductance current completely reflected from the primary side of the coupled inductor to the secondary side, the first lift diode current to the first lift capacitor charging, the second lift capacitor to the output capacitor and load release energy, at this time the first power switch A current equal to the first magnetizing inductance and the second magnetizing inductance the sum of the currents. when , the second power switch When toggled ON, this phase ends.
第六階段[ ]:[第一功率開關 :ON、第二功率開關 :OFF→ON、第一輸出二極體 :OFF、第二輸出二極體 :OFF、第一舉升二極體 :ON、第二舉升二極體 :OFF、輸出二極體 :ON]:請再一併參閱第九圖本發明之第六操作階段等效電路圖所示,第六階段開始於 ,第二功率開關 切換成ON,且第一功率開關 保持ON,第一輸出二極體 、第二輸出二極體 和第二舉升二極體 逆向偏壓。由於第二漏電感 的存在,第二功率開關 具有零電流切換[ZCS]的柔切性能,降低切換損失。第二漏電感電流 上升,當 時,第二磁化電感 的儲能仍然藉由耦合電感傳送二次側。第一舉升二極體 和輸出二極體 仍保持如前一階段的導通狀態,第一舉升二極體電流 和輸出二極體電流 下降。第一漏電感 和第二漏電感 控制了第一舉升二極體 和輸出二極體 電流下降速率,因此可緩和第一舉升二極體 和輸出二極體 反向恢復問題。當 ,第一舉升二極體電流 和輸出二極體電流 下降至0,第一舉升二極體 和輸出二極體 轉態成OFF時,本階段結束。 Sixth stage [ ]: [First power switch : ON, second power switch : OFF→ON, the first output diode : OFF, second output diode : OFF, the first lift diode : ON, the second lifting diode : OFF, output diode : ON]: Please refer to Figure 9 again, as shown in the equivalent circuit diagram of the sixth operation stage of the present invention, the sixth stage begins at , the second power switch switched ON, and the first power switch remains ON, the first output diode , the second output diode and a second boost diode reverse bias. Due to the second leakage inductance presence of a second power switch It has soft cutting performance of zero current switching [ZCS] and reduces switching loss. Second leakage inductance current rise when , the second magnetizing inductance The stored energy is still transferred to the secondary side through the coupled inductor. first lift diode and output diode remains on as in the previous stage, the first boost diode current and output diode current decline. first leakage inductance and the second leakage inductance controls the first lift diode and output diode current drop rate, thus moderating the first lift diode and output diode Reverse recovery problem. when , the first lift diode current and output diode current down to 0, the first lift diode and output diode When the state is turned OFF, this phase ends.
第七階段[ ]:[第一功率開關 :ON、第二功率開關 :ON、第一輸出二極體 :OFF、第二輸出二極體 :OFF、第一舉升二極體 :ON→OFF、第二舉升二極體 :OFF、輸出二極體 :ON→OFF]:請再一併參閱第十圖本發明之第七操作階段等效電路圖所示,第七階段開始於 ,第一舉升二極體 和輸出二極體 轉態成OFF,第一輸出二極體 、第二輸出二極體 和第二舉升二極體 均為逆向偏壓,第一功率開關 和第二功率開關 皆為ON。輸入電壓 跨於兩個耦合電感的一次側,即跨於第一磁化電感 和第一漏電感 以及第二磁化電感 和第二漏電感 ,第一漏電感電流 和第二漏電感電流 呈線性上升,斜率均為 ,從能量觀點而言,兩個耦合電感的一次側在本階段作儲存能量的動作,輸出電容 對負載 釋放能量。當 ,第一功率開關 切換成OFF時,本階段結束。 The seventh stage [ ]: [First power switch : ON, second power switch : ON, the first output diode : OFF, second output diode : OFF, the first lift diode : ON→OFF, the second lifting diode : OFF, output diode : ON→OFF]: Please refer to Figure 10 again, as shown in the equivalent circuit diagram of the seventh operation stage of the present invention, the seventh stage begins at , the first lift diode and output diode transition to OFF, the first output diode , the second output diode and a second boost diode are reverse biased, the first power switch and a second power switch All are ON. Input voltage Across the primary sides of the two coupled inductors, that is, across the first magnetizing inductance and the first leakage inductance and the second magnetizing inductance and the second leakage inductance , the first leakage inductance current and the second leakage inductance current rises linearly, with a slope of , from the perspective of energy, the primary sides of the two coupled inductors store energy at this stage, and the output capacitor on load emit energy. when , the first power switch When switched to OFF, this phase ends.
第八階段[ ]:[第一功率開關 :ON→OFF、第二功率開關 :ON、第一輸出二極體 :ON、第二輸出二極體 :OFF、第一舉升二極體 :OFF、第二舉升二極體 :ON、輸出二極體 :OFF]:請再一併參閱第十一圖本發明之第八操作階段等效電路圖所示,第八階段開始於 ,第一功率開關 切換為OFF,第二功率開關 保持為ON。第一漏電感電流 的連續性使得第一輸出二極體 轉態為ON,第一漏電感電流 流經第一輸出電容 和第一輸出二極體 ,對第一輸出電容 充電,耦合電感之第一磁化電感 的儲能以返馳式模式傳送至二次側,使得第二舉升二極體 轉態為ON,本階段第一舉升電容 經由第二舉升二極體 對第二舉升電容 充電,第二舉升二極體電流 對第一舉升電容 、第二舉升電容 充電此時第一漏電感電流 呈線性下降。當 ,第一漏電感 儲存的能量完全釋放完畢,即 ,第一輸出二極體 自然轉態成OFF時,本階段結束。 Eighth stage [ ]: [First power switch : ON→OFF, second power switch : ON, the first output diode : ON, second output diode : OFF, the first lift diode : OFF, second lift diode : ON, output diode : OFF]: Please refer to Figure 11 and the equivalent circuit diagram of the eighth operation stage of the present invention, the eighth stage starts at , the first power switch switch to OFF, the second power switch Keep it ON. The first leakage inductance current The continuity makes the first output diode transition to ON, the first leakage inductance current flows through the first output capacitor and the first output diode , for the first output capacitor Charging, the first magnetizing inductance of the coupled inductance The stored energy is transferred to the secondary side in flyback mode, making the second lift diode Turning to ON, the first lift capacitor in this stage via the second lift diode to the second boost capacitor charging, the second lift diode current to the first lift capacitor , the second lift capacitor charging at this time the first leakage inductance current Decreases linearly. when , the first leakage inductance The stored energy is completely released, that is, , the first output diode This phase ends when the natural transition is turned OFF.
第九階段[ ]:[第一功率開關 :OFF、第二功率開關 :ON、第一輸出二極體 :ON→OFF、第二輸出二極體 :OFF、第一舉升二極體 :OFF、第二舉升二極體 :ON、輸出二極體 :OFF]:請再一併參閱第十二圖本發明之第九操作階段等效電路圖所示,第九階段開始於 ,此時第一漏電感 的能量完全釋放到第一輸出電容 ,第一輸出二極體 自然轉態成OFF。第一磁化電感電流 由耦合電感之一次側完全反射到二次側,第一舉升電容 經由第二舉升二極體 對第二舉升電容 充電,此時第二功率開關 的電流等於第一磁化電感 和第二磁化電感 的電流總和。當 ,第一功率開關 切換為ON時,本階段結束,進入下一個切換週期。 Ninth stage [ ]: [First power switch : OFF, second power switch : ON, the first output diode : ON→OFF, second output diode : OFF, the first lift diode : OFF, second lift diode : ON, output diode : OFF]: Please refer to Figure 12 and the equivalent circuit diagram of the ninth operation stage of the present invention, the ninth stage starts at , at this time the first leakage inductance The energy is fully released to the first output capacitor , the first output diode Natural transition to OFF. first magnetizing inductance current Completely reflected from the primary side of the coupled inductor to the secondary side, the first lift capacitor via the second lift diode to the second boost capacitor charging, at this time the second power switch A current equal to the first magnetizing inductance and the second magnetizing inductance the sum of the currents. when , the first power switch When switching to ON, this stage ends and enters the next switching cycle.
由以上的該轉換器(1)電路動作分析可知,轉換器有以下優點:第一功率開關 和第二功率開關 具有零電流切換[ZCS]性能,可減少切換損失;第一輸出二極體 和第二輸出二極體 沒有反向恢復損失;因漏電感的存在,能夠緩和第一舉升二極體 和第二舉升二極體 的反向恢復問題。漏電感能量可回收再利用,不但可改善效率,也可避免造成突波電壓。 From the above analysis of the circuit action of the converter (1), it can be seen that the converter has the following advantages: the first power switch and a second power switch It has zero-current switching [ZCS] performance, which can reduce switching loss; the first output diode and a second output diode No reverse recovery loss; due to the presence of leakage inductance, the first lift diode can be moderated and a second boost diode reverse recovery problem. Leakage inductance energy can be recycled and reused, which not only improves efficiency, but also avoids surge voltage.
以下進行該轉換器(1)穩態特性分析,為了簡化分析,假設各開關及各二極體導通壓降為零,並且忽略時間極短的暫態階段,只考慮第二、三、四、五、七及八階段。各電容夠大,忽略電壓漣波,使得電容電壓在一個切換週期內視為常數。The steady-state characteristic analysis of the converter (1) is carried out below. In order to simplify the analysis, it is assumed that the conduction voltage drop of each switch and each diode is zero, and the extremely short transient phase is ignored, and only the second, third, fourth, and second phases are considered. Five, seven and eight stages. Each capacitor is large enough to ignore the voltage ripple so that the capacitor voltage is considered constant within one switching cycle.
電壓增益分析:Voltage Gain Analysis:
由於第一輸出電容 和第二輸出電容 的電壓可視為傳統升壓型轉換器的輸出電壓,因此根據第一磁化電感 和第二磁化電感 滿足伏秒平衡定理,可推導得到電壓 和 為 Due to the first output capacitor and a second output capacitor The voltage of can be regarded as the output voltage of a conventional boost converter, so according to the first magnetizing inductance and the second magnetizing inductance Satisfying the volt-second balance theorem, the voltage can be derived and for
(1) (1)
耦合電感二次側的輸出電容電壓 和 ,可藉由耦合電感一次側電壓反射至二次測電壓推導而得到。在第三階段,第一功率開關 :ON、第二功率開關 :OFF,而且第一舉升二極體 導通,電壓 為 The output capacitor voltage on the secondary side of the coupled inductor and , which can be obtained by reflecting the primary side voltage of the coupled inductor to the secondary measured voltage. In the third stage, the first power switch : ON, second power switch : OFF, and the first lift diode conduction, voltage for
(2) (2)
在第八階段,第一功率開關 :OFF、第二功率開關 :ON,而且第二舉升二極體 導通,電壓 為 In the eighth stage, the first power switch : OFF, second power switch : ON, and the second lift diode conduction, voltage for
(3) (3)
在第三階段電路中,根據KVL定理,可知In the third stage circuit, according to the KVL theorem, we know
(4) (4)
將(1)、(2)、(3)式的結果代入(4)式,可得總輸出電壓 為 Substituting the results of (1), (2), and (3) into (4), the total output voltage can be obtained for
(5) (5)
因此本轉換器的電壓增益 為 Therefore the voltage gain of this converter for
(6) (6)
當 時,電壓增益 與不同耦合電感的耦合係數 的關係曲線,即如第十三圖本發明之不同耦合係數和電壓增益的關係曲線圖所示,可知耦合係數 對電壓增益的影響非常小。若耦合係數 ,則理想的電壓增益M為 when When the voltage gain Coupling coefficient with different coupled inductance The relationship curve, that is, as shown in the relationship curve diagram of different coupling coefficients and voltage gains of the present invention in the thirteenth figure, it can be known that the coupling coefficient The effect on voltage gain is very small. If the coupling coefficient , then the ideal voltage gain M is
(7) (7)
從上式可知本轉換器的電壓增益具有耦合電感匝數比 和導通比 兩個設計自由度。該轉換器(1)可藉由適當設計耦合電感的匝數比,達到高升壓比,且不必操作在極大的導通比。對應於耦合電感匝數比 及導通比 的電壓增益曲線,請參閱第十四圖本發明之電壓增益與導通比及不同耦合電感匝數比之曲線圖所示,可知當導通比 , 時,電壓增益為20倍。 It can be seen from the above formula that the voltage gain of the converter has the coupling inductor turns ratio and conduction ratio Two degrees of design freedom. The converter (1) can achieve a high step-up ratio by properly designing the turns ratio of the coupling inductor, and does not need to operate at a very large conduction ratio. Corresponding to the coupled inductor turns ratio and conduction ratio For the voltage gain curve, please refer to the fourteenth graph of the voltage gain and conduction ratio of the present invention and the curves of different coupling inductor turns ratios, it can be seen that when the conduction ratio , , the voltage gain is 20 times.
功率開關和二極體的電壓應力分析:Voltage Stress Analysis of Power Switches and Diodes:
在功率開關和二極體的電壓應力分析方面,為了簡化分析,忽略電容電壓漣波、二極體導通壓降及耦合電感的漏電感,及假設耦和係數k=1。In terms of the voltage stress analysis of the power switch and the diode, in order to simplify the analysis, the capacitor voltage ripple, the diode conduction voltage drop and the leakage inductance of the coupled inductor are ignored, and the coupling coefficient k=1 is assumed.
由該轉換器(1)操作的第三階段可求得第二功率開關 的電壓應力 The third stage of operation of the converter (1) results in the second power switch voltage stress
(8) (8)
由第八階段可求得第一功率開關 的電壓應力 From the eighth stage, the first power switch can be obtained voltage stress
(9) (9)
另一方面,由第三和第八階段也可求得二極體的電壓應力On the other hand, the voltage stress of the diode can also be obtained from the third and eighth stages
(10) (10)
(11) (11)
(12) (12)
(13) (13)
由於傳統交錯式升壓型轉換器的功率開關電壓應力為輸出電壓 ,而該轉換器(1)的開關電壓應力僅為輸出電壓 的 倍,因此可使用低額定耐壓具有較低導通電阻的MOSFET,可降低開關導通損失。另一方面,較低電壓應力的二極體可採用順向壓降較低的功率二極體,可降低導通損失。 Since the power switch voltage stress of a conventional interleaved boost converter is the output voltage , while the switching voltage stress of this converter (1) is only the output voltage of times, so MOSFETs with low rated withstand voltage and low on-resistance can be used, which can reduce switch conduction loss. On the other hand, diodes with lower voltage stress can use power diodes with lower forward voltage drop, which can reduce conduction loss.
依據上述電路動作分析結果,利用Is-Spice軟體作先期的模擬,轉換器規格:輸入電壓32V、輸出電壓400V、最大輸出功率1000W、切換頻率50kHz,耦合電感匝數比
,驗證該轉換器(1)的特點,以下以模擬波形驗證與說明該轉換器(1)的特點[請再一併參閱第十五圖本發明之模擬電路示意圖所示]。
Based on the analysis results of the above circuit, Is-Spice software was used for preliminary simulation. The converter specifications:
A.驗證穩態特性:A. Verify steady-state characteristics:
首先驗證該轉換器(1)之穩態特性,滿載1000W時,請參閱第十六圖本發明之開關驅動信號、輸入電壓與輸出電壓波形圖所示,當輸入電壓32V、輸出電壓400V、耦合電感匝數比 ,則導通比的理論值大約 ,模擬結果符合該轉換器(1)電壓增益的公式。 First verify the steady-state characteristics of the converter (1). When the full load is 1000W, please refer to the sixteenth figure of the present invention for the switch drive signal, input voltage and output voltage waveform diagram. Inductance turns ratio , the theoretical value of the conduction ratio is about , the simulation results conform to the formula of the converter (1) voltage gain.
B.驗證開關電壓應力:B. Verify switch voltage stress:
請參閱第十七圖本發明之開關驅動信號與開關跨壓信號波形圖所示,可知當第一功率開關 、第二功率開關 為OFF時,其跨壓 或 都約為80V,僅為輸出電壓400V的五分之一,符合分析結果,比較傳統的升壓型轉換器,開關電壓應力為輸出電壓,該轉換器(1)的開關具有低電壓應力的優點。 Please refer to the seventeenth figure shown in the waveform diagram of the switch drive signal and the switch cross-voltage signal of the present invention, it can be known that when the first power switch , the second power switch is OFF, its voltage across the or Both are about 80V, which is only one-fifth of the output voltage of 400V, which is consistent with the analysis results. Compared with the traditional boost converter, the switch voltage stress is the output voltage. The switch of the converter (1) has the advantage of low voltage stress .
C.驗證具有低輸入漣波電流性能與CCM操作:C. Verify low input ripple current performance with CCM operation:
請參閱第十八圖本發明之滿載1000W時,耦合電感的漏電感電流及總輸入電流波形圖所示,可知 、 的漣波電流大小大約33A,而輸入電流的漣波電流大小僅為約1.6A,很明顯地,交錯式操作具有降低輸入漣波電流效用。請參閱第十九圖本發明之耦合電感的磁化電感電流波形圖所示,驗證該轉換器(1)操作在連續導通模式[CCM]。 Please refer to the 18th figure of the present invention when the full load is 1000W, the leakage inductance current of the coupled inductor and the total input current waveform diagram, we can know , The ripple current of the input current is about 33A, while the ripple current of the input current is only about 1.6A. Obviously, the interleaved operation has the effect of reducing the input ripple current. Please refer to Figure 19, which shows the magnetizing inductance current waveform of the coupled inductor of the present invention, and verify that the converter (1) operates in continuous conduction mode [CCM].
D.驗證二極體反向恢復電流問題:D. Verify the diode reverse recovery current problem:
請參閱第二十圖本發明之輸出二極體的電流及電壓波形圖所示,可知 和 都沒有反向恢復問題,因此沒有反向恢復損失。另一方面可看出,第一輸出二極體 電壓應力約為80V,只有輸出電壓的五之一,第二輸出二極體 電壓應力大約為160V,只有輸出電壓的五分之二,符合分析結果。 Please refer to the current and voltage waveforms of the output diode of the present invention in Figure 20, it can be known and Neither has reverse recovery issues, so there is no reverse recovery penalty. On the other hand it can be seen that the first output diode The voltage stress is about 80V, only one-fifth of the output voltage, the second output diode The voltage stress is about 160V, only two-fifths of the output voltage, which is consistent with the analysis results.
請參閱第二十一圖本發明之舉升二極體和輸出二極體的電流及電壓波形圖所示,第一舉升二極體 和第二舉升二極體 的電壓應力均為160V,符合分析結果。第一舉升二極體 、第二舉升二極體 和輸出二極體 的電流幾乎沒有反向恢復電流,因為耦合電感中第一漏電感 和第二漏電感 的存在緩和了反向恢復問題。 Please refer to the current and voltage waveforms of the lifting diode and the output diode in Figure 21 of the present invention, the first lifting diode and a second boost diode The voltage stresses are all 160V, which is in line with the analysis results. first lift diode , the second lifting diode and output diode The current has almost no reverse recovery current, because the first leakage inductance in the coupled inductor and the second leakage inductance The existence of mit alleviates the reverse recovery problem.
E.驗證輸出電容電壓:E. Verify output capacitor voltage:
請參閱第二十二圖本發明之電容的電壓波形圖所示,電容電壓 、 和 大約都等於80V,電容電壓 大約等於160V,電容電壓 約等於400V,符合分析結果。 Please refer to Figure 22 shown in the voltage waveform diagram of the capacitor of the present invention, the capacitor voltage , and Approximately equal to 80V, the capacitor voltage Approximately equal to 160V, the capacitor voltage It is approximately equal to 400V, which is in line with the analysis results.
根據以上的模擬波形驗證,該轉換器(1)的特性與優點歸納如下:According to the above simulation waveform verification, the characteristics and advantages of the converter (1) are summarized as follows:
1.電壓增益公式、各開關電壓應力、各二極體電壓應力及每個輸出電容電壓值都與穩態特性分析的推導結果都十分符合。1. The voltage gain formula, the voltage stress of each switch, the voltage stress of each diode, and the voltage value of each output capacitor are all in good agreement with the derivation results of the steady-state characteristic analysis.
2.高電壓增益的達成,確實不必操作在極大的導通比。2. To achieve high voltage gain, it is really not necessary to operate at a very large conduction ratio.
3.轉換器兩個功率開關的電壓應力只有輸出電壓的五分之一,可以使用導通電阻較小的低額定耐壓MOSFET,以降低導通損失。3. The voltage stress of the two power switches of the converter is only one-fifth of the output voltage, and a low-rated withstand voltage MOSFET with a small on-resistance can be used to reduce the conduction loss.
4.由於輸出二極體在轉態成OFF之前,其流經的電流先降為零,所以輸出二極體沒有反向恢復功率損失問題。4. Since the current flowing through the output diode drops to zero before it turns OFF, the output diode does not have the problem of reverse recovery power loss.
5.耦合電感的漏電感能量,能夠回收再利用,避免了造成功率開關的電壓突波問題。5. The leakage inductance energy of the coupled inductor can be recycled and reused, avoiding the voltage surge problem caused by the power switch.
6.由於交錯式操作,使得耦合電感一次側繞組的電流漣波能相消,降低輸入電流漣波大小。6. Due to the interleaved operation, the current ripple of the primary side winding of the coupled inductor can be eliminated, reducing the size of the input current ripple.
然而前述之實施例或圖式並非限定本發明之產品結構或使用方式,任何所屬技術領域中具有通常知識者之適當變化或修飾,皆應視為不脫離本發明之專利範疇。However, the aforementioned embodiments or drawings do not limit the product structure or usage of the present invention, and any appropriate changes or modifications by those with ordinary knowledge in the technical field shall be considered as not departing from the patent scope of the present invention.
綜上所述,本發明實施例確能達到所預期之使用功效,又其所揭露之具體構造,不僅未曾見諸於同類產品中,亦未曾公開於申請前,誠已完全符合專利法之規定與要求,爰依法提出發明專利之申請,懇請惠予審查,並賜准專利,則實感德便。To sum up, the embodiment of the present invention can indeed achieve the expected use effect, and the specific structure disclosed by it has not only never been seen in similar products, nor has it been disclosed before the application, and it has fully complied with the provisions of the Patent Law In accordance with the requirements, it is very convenient to file an application for a patent for invention in accordance with the law, and sincerely ask for the review and approval of the patent.
1:轉換器1: Converter
:輸入電壓 :Input voltage
:第一耦合電感一次側 : Primary side of the first coupled inductor
:第一耦合電感二次側 : Secondary side of the first coupled inductor
:第一磁化電感 : first magnetizing inductance
:第一漏電感 : The first leakage inductance
:第二耦合電感一次側 : Primary side of the second coupled inductor
:第二耦合電感二次側 : Secondary side of the second coupled inductor
:第二磁化電感 : second magnetizing inductance
:第二漏電感 : Second leakage inductance
:第一功率開關 : first power switch
:第二功率開關 : Second power switch
:第一輸出電容 : first output capacitance
:第二輸出電容 : Second output capacitor
:第一舉升電容 : The first lift capacitor
:第二舉升電容 : Second lift capacitor
:輸出電容 : output capacitance
:第一輸出二極體 : The first output diode
:第二輸出二極體 : Second output diode
:第一舉升二極體 : 1st lift diode
:第二舉升二極體 : Second lift diode
:輸出二極體 : output diode
:負載 :load
2:升壓型轉換器2: Boost converter
3:交錯式升壓型轉換器3: Interleaved Boost Converter
第一圖:本發明之電路圖The first figure: the circuit diagram of the present invention
第二圖:本發明之等效電路圖The second figure: the equivalent circuit diagram of the present invention
第三圖:本發明之主要元件時序波形圖Figure 3: Timing waveform diagram of the main components of the present invention
第四圖:本發明之第一操作階段等效電路圖Figure 4: Equivalent circuit diagram of the first operating stage of the present invention
第五圖:本發明之第二操作階段等效電路圖The fifth figure: the equivalent circuit diagram of the second operation stage of the present invention
第六圖:本發明之第三操作階段等效電路圖Figure 6: Equivalent circuit diagram of the third operating stage of the present invention
第七圖:本發明之第四操作階段等效電路圖The seventh figure: the equivalent circuit diagram of the fourth operation stage of the present invention
第八圖:本發明之第五操作階段等效電路圖Figure 8: Equivalent circuit diagram of the fifth operating stage of the present invention
第九圖:本發明之第六操作階段等效電路圖Figure 9: Equivalent circuit diagram of the sixth operating stage of the present invention
第十圖:本發明之第七操作階段等效電路圖Figure 10: Equivalent circuit diagram of the seventh operation stage of the present invention
第十一圖:本發明之第八操作階段等效電路圖Figure 11: Equivalent circuit diagram of the eighth operation stage of the present invention
第十二圖:本發明之第九操作階段等效電路圖Figure 12: Equivalent circuit diagram of the ninth operating stage of the present invention
第十三圖:本發明之不同耦合係數和電壓增益的關係曲線圖The thirteenth figure: the relational graph of different coupling coefficients and voltage gain of the present invention
第十四圖:本發明之電壓增益與導通比及不同耦合電感匝數比之曲線圖Figure 14: The graph of voltage gain, conduction ratio and turns ratio of different coupling inductors according to the present invention
第十五圖:本發明之模擬電路示意圖Figure 15: Schematic diagram of the analog circuit of the present invention
第十六圖:本發明之開關驅動信號、輸入電壓與輸出電壓波形圖Figure 16: Switch drive signal, input voltage and output voltage waveform diagram of the present invention
第十七圖:本發明之開關驅動信號與開關跨壓信號波形圖Figure 17: Waveform diagram of switch drive signal and switch cross-voltage signal of the present invention
第十八圖:本發明之滿載1000W時,耦合電感的漏電感電流及總輸入電流波形圖Figure 18: When the present invention is fully loaded with 1000W, the leakage inductance current of the coupled inductor and the waveform diagram of the total input current
第十九圖:本發明之耦合電感的磁化電感電流波形圖Figure 19: The magnetizing inductance current waveform diagram of the coupled inductance of the present invention
第二十圖:本發明之輸出二極體的電流及電壓波形圖Figure 20: Current and voltage waveforms of the output diode of the present invention
第二十一圖:本發明之舉升二極體和輸出二極體的電流及電壓波形圖Figure 21: Current and voltage waveforms of the lifting diode and the output diode of the present invention
第二十二圖:本發明之電容的電壓波形圖Figure 22: the voltage waveform diagram of the capacitor of the present invention
第二十三圖:現有之傳統升壓型轉換器電路圖Figure 23: Circuit Diagram of Existing Traditional Boost Converter
第二十四圖:現有之傳統升壓型轉換器的輸出電壓增益對開關導通比的關係曲線圖Figure 24: The relationship between the output voltage gain and the switch conduction ratio of the existing conventional boost converter
第二十五圖:現有之傳統升壓型轉換器的效率對開關導通比的關係曲線圖Figure 25: The relationship between the efficiency of the existing traditional boost converter and the switch conduction ratio
第二十六圖:現有之交錯式升壓型轉換器電路圖Figure 26: Circuit diagram of the existing interleaved boost converter
1:轉換器 1: Converter
V in:輸入電壓 V in : input voltage
N p1:第一耦合電感一次側 N p 1 : Primary side of the first coupled inductor
N s1:第一耦合電感二次側 N s 1 : Secondary side of the first coupled inductor
L m1:第一磁化電感 L m 1 : first magnetizing inductance
L k1:第一漏電感 L k 1 : the first leakage inductance
N p2:第二耦合電感一次側 N p 2 : Primary side of the second coupled inductor
N s2:第二耦合電感二次側 N s 2 : Secondary side of the second coupled inductor
L m2:第二磁化電感 L m 2 : second magnetizing inductance
L k2:第二漏電感 L k 2 : Second leakage inductance
S 1:第一功率開關 S 1 : the first power switch
S 2:第二功率開關 S 2 : Second power switch
C 1:第一輸出電容 C 1 : the first output capacitor
C 2:第二輸出電容 C 2 : Second output capacitor
C 3:第一舉升電容 C 3 : the first lift capacitor
C 4:第二舉升電容 C 4 : Second lift capacitor
C o:輸出電容 C o : output capacitance
D 1:第一輸出二極體 D 1 : The first output diode
D 2:第二輸出二極體 D 2 : Second output diode
D 3:第一舉升二極體 D 3 : The first lift diode
D 4:第二舉升二極體 D 4 : Second lift diode
D o :輸出二極體 D o : output diode
R o :負載 R o : load
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