TW202308279A - High voltage conversion ratio dc converter - Google Patents

High voltage conversion ratio dc converter Download PDF

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TW202308279A
TW202308279A TW110128276A TW110128276A TW202308279A TW 202308279 A TW202308279 A TW 202308279A TW 110128276 A TW110128276 A TW 110128276A TW 110128276 A TW110128276 A TW 110128276A TW 202308279 A TW202308279 A TW 202308279A
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diode
output
voltage
terminal
power switch
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TW110128276A
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TWI762396B (en
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陳信助
楊松霈
黃昭明
買柏豪
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崑山科技大學
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Abstract

This invention relates to a high voltage conversion ratio DC-DC converter, which consists of two sets of step-up converters connected in series with parallel input and output, and the voltage lift module is connected in series with the output side to enhance the voltage gain; the voltage lift module consists of two coupling inductors connected in series with two lift capacitors and lift diodes on the secondary side, and the two power switches are operated in a staggered manner with half switching period difference, so that the current ripple of the primary winding of the coupling inductors can be partially eliminated to reduce the input current ripple size. Thus, with high boost characteristics, but without operating at very large on-ratios, the power switch has a low voltage stress far below the output voltage at high output voltages, which reduces the conduction loss of the power switch. When high input current is applied, it has low input current ripple, and the leakage inductance can improve the reverse recovery loss of the diode, making it suitable for high boost, high efficiency and high power applications, and increasing the practical efficiency characteristics in its overall implementation.

Description

高電壓轉換比直流轉換器High Voltage Conversion Ratio DC Converter

本發明係有關於一種高電壓轉換比直流轉換器,尤其是指一種具有高升壓特性,但是不必操作在極大導通比;高輸出電壓時,功率開關具有遠低於輸出電壓的低電壓應力,可以降低功率開關的導通損失;高輸入電流應用時,具有低輸入電流漣波;而且漏電感能改善二極體的反向恢復損失,使得其適合高升壓、高效率和高功率之應用,而在其整體施行使用上更增實用功效特性者。The present invention relates to a DC converter with a high voltage conversion ratio, in particular to a DC converter with high boost characteristics, but does not have to operate at a large conduction ratio; when the output voltage is high, the power switch has a low voltage stress far below the output voltage, It can reduce the conduction loss of the power switch; when high input current is applied, it has low input current ripple; and the leakage inductance can improve the reverse recovery loss of the diode, making it suitable for high boost, high efficiency and high power applications. And in its overall implementation and use, it has more practical and functional characteristics.

按,《巴黎協定》希望各國透過再生能源,用更經濟、有效的方式達成減排目標,追求經濟的「綠色成長」。爰此,再生能源的利用必定是各國產業發展的重點方向,包含太陽能、風力能、水力能、地熱能、潮汐能、生質能及燃料電池等。例如在歐洲、日本與美國裝設於屋頂的住宅型太陽能併網電力系統,最近成為成長快速的市場。在再生能源電力系統應用中,太陽能發電系統及燃料電池發電系統的技術發展越來越成熟,常常在分散式發電系統[distributed generation system]扮演重要的角色。According to the Paris Agreement, countries are expected to use renewable energy to achieve emission reduction targets in a more economical and effective way, and to pursue economic "green growth". Therefore, the utilization of renewable energy must be the key direction of industrial development in various countries, including solar energy, wind energy, hydraulic energy, geothermal energy, tidal energy, biomass energy and fuel cells. For example, residential solar grid-connected power systems installed on rooftops in Europe, Japan, and the United States have recently become a fast-growing market. In the application of renewable energy power system, the technological development of solar power generation system and fuel cell power generation system is becoming more and more mature, and they often play an important role in distributed generation system [distributed generation system].

由於住宅型應用[residential applications]的安全性與可靠性的問題,太陽能電池模組與燃料電池所產生的輸出電壓是屬於低電壓,一般不超過

Figure 02_image045
,為了達到併網發電系統或直流微電網的需求,必須先將此低電壓利用高升壓DC-DC轉換器,升壓至一個高直流排電壓。例如:對於一個單相
Figure 02_image047
的電網系統而言,此高直流排電壓常為
Figure 02_image049
,以利全橋換流器[inverter]的DC-AC轉換。理論上,操作在極高導通比的傳統升壓型[boost]轉換器能夠得到高電壓增益,但是實務上受到寄生元件的影響,電壓轉換比受限在約5倍以下,因此當電壓增益超過5倍的需求時,研發嶄新的高升壓轉換器拓樸是必要的。因此近幾年高升壓DC-DC轉換器是電力電子工程領域中常見的研究主題之一。 Due to the safety and reliability of residential applications, the output voltage generated by solar cell modules and fuel cells is low voltage, generally not exceeding
Figure 02_image045
, in order to meet the needs of grid-connected power generation systems or DC microgrids, this low voltage must first be boosted to a high DC row voltage by using a high-boost DC-DC converter. Example: For a single phase
Figure 02_image047
For the power grid system, this high DC row voltage is often
Figure 02_image049
, to facilitate the DC-AC conversion of the full-bridge converter [inverter]. Theoretically, a traditional step-up [boost] converter operating at a very high conduction ratio can obtain a high voltage gain, but in practice, due to the influence of parasitic elements, the voltage conversion ratio is limited to less than about 5 times, so when the voltage gain exceeds 5 times the demand, it is necessary to develop a new high-boost converter topology. Therefore, high-boost DC-DC converters are one of the common research topics in the field of power electronics engineering in recent years.

請參閱第二十三圖現有之傳統升壓型轉換器電路圖所示,該升壓型轉換器(2)電路中

Figure 02_image051
為電感的等效串聯電阻,當考慮理想元件[
Figure 02_image053
]且操作在連續導通模式[CCM]模式時,其輸出電壓增益
Figure 02_image055
Please refer to the existing traditional boost converter circuit diagram shown in Figure 23, in the boost converter (2) circuit
Figure 02_image051
is the equivalent series resistance of the inductor, when considering ideal components[
Figure 02_image053
] and operating in the continuous conduction mode [CCM] mode, its output voltage gain
Figure 02_image055

Figure 02_image057
Figure 02_image057

電壓增益完全決定於開關導通比[duty ratio]

Figure 02_image059
。理論上要得到高電壓增益,轉換器必須操作在極大導通比;但是實務上,由於寄生元件的存在,例如
Figure 02_image061
,則電壓增益
Figure 02_image063
與效率
Figure 02_image064
對導通比的表示式分別為 The voltage gain is completely determined by the switch conduction ratio [duty ratio]
Figure 02_image059
. Theoretically, to obtain a high voltage gain, the converter must operate at a very large conduction ratio; but in practice, due to the existence of parasitic elements, such as
Figure 02_image061
, then the voltage gain
Figure 02_image063
and efficiency
Figure 02_image064
The expressions for the conduction ratio are

Figure 02_image066
Figure 02_image066

Figure 02_image068
Figure 02_image068

請再參閱第二十四圖現有之傳統升壓型轉換器的輸出電壓增益對開關導通比的關係曲線圖及第二十五圖現有之傳統升壓型轉換器的效率對開關導通比的關係曲線圖所示,可知操作在極大導通比的轉換器電壓增益是有所限制,而且轉換效率不佳,另外操作在極大導通比的升壓型轉換器衍生了以下問題:容易產生很大的輸入電流漣波,使得太陽能電池模組輸出端的電解電容數量必須增加,減少燃料電池的使用壽命;此外,輸出二極體的反向恢復損失相當大。Please refer to Figure 24 for the relationship between the output voltage gain of the existing conventional boost converter and the switch conduction ratio and Figure 25 for the relationship between the efficiency of the conventional boost converter and the switch conduction ratio As shown in the graph, it can be seen that the voltage gain of the converter operating at a very large conduction ratio is limited, and the conversion efficiency is not good. In addition, the boost converter operating at a very large conduction ratio has the following problems: it is easy to generate a large input Due to the current ripple, the number of electrolytic capacitors at the output of the solar cell module must be increased, reducing the service life of the fuel cell; in addition, the reverse recovery loss of the output diode is quite large.

使得為了適合高功率應用及降低輸入電流漣波的特性,請參閱第二十六圖現有之交錯式升壓型轉換器電路圖所示,即有業者發展出交錯式升壓型轉換器(3),然而該交錯式升壓型轉換器(3)之功率開關仍需承受高電壓應力,其導通損失會隨開關導通比增大而增加;因此研發交錯式DC-DC轉換器拓樸具有高升壓特性,但是不必操作在極大開關導通比,改善二極體的反向恢復損失問題,是重要的考量。In order to be suitable for high-power applications and reduce the characteristics of input current ripple, please refer to the existing interleaved boost converter circuit diagram in Figure 26, that is, the industry has developed an interleaved boost converter (3) , however, the power switch of the interleaved boost converter (3) still needs to withstand high voltage stress, and its conduction loss will increase with the increase of the switch conduction ratio; therefore, the research and development of the interleaved DC-DC converter topology has high Voltage characteristics, but it is not necessary to operate at a maximum switch conduction ratio, and it is an important consideration to improve the reverse recovery loss of the diode.

再者,典型交錯式升壓型轉換器之開關電壓應力為高壓的輸出電壓,由於高耐壓的MOSFET,一般都具有高導通電阻

Figure 02_image070
的特性,導致較高的導通損失;因此在開關成本、導通電阻、耐壓限制與轉換效率的考量之下,研發高升壓DC-DC轉換,而功率開關具有低電壓應力,是另一個重要的考量。 Furthermore, the switching voltage stress of a typical interleaved boost converter is a high-voltage output voltage. Due to the high withstand voltage MOSFET, it generally has a high on-resistance
Figure 02_image070
Therefore, under the consideration of switching cost, on-resistance, withstand voltage limit and conversion efficiency, it is another important to develop high-boost DC-DC conversion, and the power switch has low voltage stress. considerations.

緣是,發明人有鑑於此,秉持多年該相關行業之豐富設計開發及實際製作經驗,針對現有之結構及缺失再予以研究改良,提供一種高電壓轉換比直流轉換器,以期達到更佳實用價值性之目的者。The reason is that, in view of this, the inventor has been adhering to years of rich experience in design, development and actual production in this related industry, and then researched and improved the existing structure and defects to provide a high-voltage conversion ratio DC converter in order to achieve better practical value. sexual purpose.

本發明之主要目的在於提供一種高電壓轉換比直流轉換器,主要係具有高升壓特性,但是不必操作在極大導通比;高輸出電壓時,功率開關具有遠低於輸出電壓的低電壓應力,可以降低功率開關的導通損失;高輸入電流應用時,具有低輸入電流漣波;而且漏電感能改善二極體的反向恢復損失,使得其適合高升壓、高效率和高功率之應用,而在其整體施行使用上更增實用功效特性者。The main purpose of the present invention is to provide a high voltage conversion ratio DC converter, which mainly has high boost characteristics, but does not need to operate at a large conduction ratio; when the output voltage is high, the power switch has a low voltage stress far below the output voltage, It can reduce the conduction loss of the power switch; when high input current is applied, it has low input current ripple; and the leakage inductance can improve the reverse recovery loss of the diode, making it suitable for high boost, high efficiency and high power applications. And in its overall implementation and use, it has more practical and functional characteristics.

為令本發明所運用之技術內容、發明目的及其達成之功效有更完整且清楚的揭露,茲於下詳細說明之,並請一併參閱所揭之圖式及圖號:In order to have a more complete and clear disclosure of the technical content used in the present invention, the purpose of the invention and the effects achieved, it will be described in detail below, and please also refer to the disclosed drawings and drawing numbers:

首先,請參閱第一圖本發明之電路圖及第二圖本發明之等效電路圖所示,本發明之轉換器(1)主要係於輸入電壓

Figure 02_image001
之正極分別連接第一耦合電感一次側
Figure 02_image003
之第一端及第二耦合電感一次側
Figure 02_image011
之第一端,該第一耦合電感一次側
Figure 02_image072
形成有第一磁化電感
Figure 02_image007
,該第二耦合電感一次側
Figure 02_image011
形成有第二磁化電感
Figure 02_image015
,於該第一耦合電感一次側
Figure 02_image003
之第二端分別連接有第一功率開關
Figure 02_image019
之第一端、第一輸出電容
Figure 02_image023
之第一端及第二輸出電容
Figure 02_image025
之第二端,且於該第一耦合電感一次側
Figure 02_image003
之第二端與該第一功率開關
Figure 02_image019
之第一端、該第一輸出電容
Figure 02_image023
之第一端及該第二輸出電容
Figure 02_image025
之第二端之間形成有第一漏電感
Figure 02_image009
,而該第二耦合電感一次側
Figure 02_image011
之第二端分別連接有第二功率開關
Figure 02_image021
之第一端及第二輸出二極體
Figure 02_image035
之正極,並於該第二耦合電感一次側
Figure 02_image011
之第二端與該第二功率開關
Figure 02_image021
之第一端及第二輸出二極體
Figure 02_image035
之正極之間形成有第二漏電感
Figure 02_image017
,該輸入電壓
Figure 02_image001
之負極分別連接該第二功率開關
Figure 02_image021
之第二端、該第一功率開關
Figure 02_image019
之第二端及第一輸出二極體
Figure 02_image033
之負極,該第二輸出二極體
Figure 02_image035
之負極分別連接該第二輸出電容
Figure 02_image025
之第一端、第一舉升電容
Figure 02_image027
之第一端及第二耦合電感二次側
Figure 02_image013
之第一端,該第二耦合電感二次側
Figure 02_image013
之第二端連接第一耦合電感二次側
Figure 02_image005
之第二端,該第一耦合電感二次側
Figure 02_image005
之第一端分別連接第二舉升電容
Figure 02_image029
之第一端及第一舉升二極體
Figure 02_image037
之正極,該第一舉升電容
Figure 02_image027
之第二端分別連接該第一舉升二極體
Figure 02_image037
之負極及第二舉升二極體
Figure 02_image039
之正極,該第二舉升電容
Figure 02_image029
之第二端分別連接該第二舉升二極體
Figure 02_image078
之負極及輸出二極體
Figure 02_image041
之正極,該輸出二極體
Figure 02_image079
之負極分別連接輸出電容
Figure 02_image031
之第一端及負載
Figure 02_image043
之第一端,該第一輸出二極體
Figure 02_image033
之正極則分別連接該第一輸出電容
Figure 02_image023
之第二端、該輸出電容
Figure 02_image031
之第二端及該負載
Figure 02_image043
之第二端。 First of all, please refer to the circuit diagram of the present invention shown in the first figure and the equivalent circuit diagram of the present invention in the second figure, the converter (1) of the present invention mainly depends on the input voltage
Figure 02_image001
The positive poles are respectively connected to the primary side of the first coupled inductor
Figure 02_image003
The first end and the primary side of the second coupled inductor
Figure 02_image011
The first end, the primary side of the first coupled inductor
Figure 02_image072
formed with a first magnetizing inductance
Figure 02_image007
, the second coupled inductor primary side
Figure 02_image011
formed with a second magnetizing inductance
Figure 02_image015
, on the primary side of the first coupled inductor
Figure 02_image003
The second end is respectively connected with the first power switch
Figure 02_image019
The first terminal, the first output capacitor
Figure 02_image023
The first terminal and the second output capacitor
Figure 02_image025
The second terminal, and on the primary side of the first coupled inductor
Figure 02_image003
The second terminal of the first power switch
Figure 02_image019
The first end of the first output capacitor
Figure 02_image023
The first terminal and the second output capacitor
Figure 02_image025
A first leakage inductance is formed between the second terminal of
Figure 02_image009
, and the primary side of the second coupled inductor
Figure 02_image011
The second end is respectively connected with the second power switch
Figure 02_image021
The first terminal and the second output diode
Figure 02_image035
positive pole, and on the primary side of the second coupled inductor
Figure 02_image011
The second terminal and the second power switch
Figure 02_image021
The first terminal and the second output diode
Figure 02_image035
A second leakage inductance is formed between the positive poles
Figure 02_image017
, the input voltage
Figure 02_image001
The negative poles of are respectively connected to the second power switch
Figure 02_image021
The second end of the first power switch
Figure 02_image019
The second terminal and the first output diode
Figure 02_image033
the negative pole of the second output diode
Figure 02_image035
The negative poles of the two are respectively connected to the second output capacitor
Figure 02_image025
The first terminal, the first lift capacitor
Figure 02_image027
The first terminal and the secondary side of the second coupled inductor
Figure 02_image013
The first end of the second coupled inductor secondary side
Figure 02_image013
The second terminal is connected to the secondary side of the first coupled inductor
Figure 02_image005
The second terminal, the secondary side of the first coupled inductor
Figure 02_image005
The first end of each is connected to the second lift capacitor
Figure 02_image029
The first terminal and the first lifting diode
Figure 02_image037
positive terminal, the first lift capacitor
Figure 02_image027
The second terminal is connected to the first lifting diode
Figure 02_image037
The negative electrode and the second lifting diode
Figure 02_image039
positive terminal, the second lift capacitor
Figure 02_image029
The second terminal is connected to the second lifting diode
Figure 02_image078
Negative pole and output diode
Figure 02_image041
the positive terminal, the output diode
Figure 02_image079
The negative poles are respectively connected to the output capacitor
Figure 02_image031
The first terminal and the load
Figure 02_image043
The first terminal, the first output diode
Figure 02_image033
The positive poles are respectively connected to the first output capacitor
Figure 02_image023
The second terminal, the output capacitor
Figure 02_image031
The second terminal and the load
Figure 02_image043
the second end.

而該轉換器(1)在連續導通模式[CCM]中,為了達到高升壓性能,導通比大於0.5,而且該第一功率開關

Figure 02_image081
和該第二功率開關
Figure 02_image082
以工作相位相差半切換週期的交錯式操作。穩態分析時,根據該轉換器(1)各功率開關及各二極體的ON/OFF狀態,該轉換器(1)在一個切換週期內可分成9個線性操作階段,假設: In the continuous conduction mode [CCM] of the converter (1), in order to achieve high boost performance, the conduction ratio is greater than 0.5, and the first power switch
Figure 02_image081
and the second power switch
Figure 02_image082
Interleaved operation with working phase difference of half switching period. In the steady-state analysis, according to the ON/OFF state of each power switch and each diode of the converter (1), the converter (1) can be divided into 9 linear operation stages within one switching cycle, assuming:

1.所有功率半導體元件[各開關及各二極體]均為理想,即導通壓降為零。1. All power semiconductor components [switches and diodes] are ideal, that is, the conduction voltage drop is zero.

2.各電容

Figure 02_image084
Figure 02_image086
Figure 02_image088
Figure 02_image090
Figure 02_image092
夠大,各電容電壓
Figure 02_image094
Figure 02_image096
Figure 02_image098
Figure 02_image100
Figure 02_image102
可視為定電壓,因此輸出電壓
Figure 02_image104
可視為常數。 2. Capacitors
Figure 02_image084
,
Figure 02_image086
,
Figure 02_image088
and
Figure 02_image090
and
Figure 02_image092
large enough that each capacitor voltage
Figure 02_image094
,
Figure 02_image096
,
Figure 02_image098
,
Figure 02_image100
and
Figure 02_image102
can be regarded as a constant voltage, so the output voltage
Figure 02_image104
can be regarded as a constant.

3.兩個耦合電感的匝數比相等[

Figure 02_image106
]且磁化電感值相等[
Figure 02_image108
],漏電感值相等[
Figure 02_image110
],磁化電感遠大於漏電感,耦合電感的耦合係數
Figure 02_image112
。 3. The turns ratio of the two coupled inductors is equal [
Figure 02_image106
] and the magnetizing inductance value is equal to [
Figure 02_image108
], the leakage inductance value is equal to [
Figure 02_image110
], the magnetizing inductance is much larger than the leakage inductance, the coupling coefficient of the coupled inductance
Figure 02_image112
.

4.耦合電感的磁化電感電流操作在連續導通模式[Continuous Conduction Mode,CCM]。4. The magnetizing inductance current of the coupled inductor operates in continuous conduction mode [Continuous Conduction Mode, CCM].

其各線性階段線性等效電路以及主要元件波形如下所示,請再一併參閱第三圖本發明之主要元件時序波形圖所示:The linear equivalent circuit of each linear stage and the waveform of the main components are as follows, please refer to the third figure together with the timing waveform diagram of the main components of the present invention:

第一階段[

Figure 02_image114
]:[第一功率開關
Figure 02_image081
:OFF→ON、第二功率開關
Figure 02_image082
:ON、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:OFF、第二舉升二極體
Figure 02_image039
:ON、輸出二極體
Figure 02_image116
:OFF]:請再一併參閱第四圖本發明之第一操作階段等效電路圖所示,第一階段開始於
Figure 02_image118
,第一功率開關
Figure 02_image081
切換成ON,且第二功率開關
Figure 02_image082
仍保持ON,第一輸出二極體
Figure 02_image033
、第二輸出二極體
Figure 02_image035
、第一舉升二極體
Figure 02_image037
、輸出二極體
Figure 02_image116
均為逆向偏壓,由於第一漏電感
Figure 02_image009
的存在,第一功率開關
Figure 02_image081
具有零電流切換[ZCS]的柔切性能,降低切換損失。第一漏電感電流
Figure 02_image120
上升,當
Figure 02_image122
時,第一磁化電感
Figure 02_image007
所儲存的能量仍然藉由耦合電感傳送至二次側,第二舉升二極體
Figure 02_image039
保持導通狀態,第二舉升二極體電流
Figure 02_image124
下降。第一漏電感
Figure 02_image009
控制了第二舉升二極體電流
Figure 02_image126
的下降速率,緩和了第二舉升二極體
Figure 02_image039
反向恢復問題。當
Figure 02_image127
,第二舉升二極體電流
Figure 02_image126
下降至0,第二舉升二極體
Figure 02_image039
轉態成OFF時,本階段結束。 The first stage[
Figure 02_image114
]: [First power switch
Figure 02_image081
: OFF→ON, second power switch
Figure 02_image082
: ON, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: OFF, second lift diode
Figure 02_image039
: ON, output diode
Figure 02_image116
: OFF]: Please refer to Figure 4 again, as shown in the equivalent circuit diagram of the first operation stage of the present invention, the first stage begins at
Figure 02_image118
, the first power switch
Figure 02_image081
switches ON, and the second power switch
Figure 02_image082
remains ON, the first output diode
Figure 02_image033
, the second output diode
Figure 02_image035
, the first lifting diode
Figure 02_image037
, output diode
Figure 02_image116
are reverse biased, due to the first leakage inductance
Figure 02_image009
presence of the first power switch
Figure 02_image081
It has soft cutting performance of zero current switching [ZCS] and reduces switching loss. The first leakage inductance current
Figure 02_image120
rise when
Figure 02_image122
, the first magnetizing inductance
Figure 02_image007
The stored energy is still transmitted to the secondary side through the coupled inductor, the second lift diode
Figure 02_image039
remains on, the second lift-up diode current
Figure 02_image124
decline. first leakage inductance
Figure 02_image009
controls the second lift diode current
Figure 02_image126
rate of fall, moderated by the second lift diode
Figure 02_image039
Reverse recovery problem. when
Figure 02_image127
, the second lift diode current
Figure 02_image126
drops to 0, the second lift diode
Figure 02_image039
When the state is turned OFF, this phase ends.

第二階段[

Figure 02_image129
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:ON、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:OFF、第二舉升二極體
Figure 02_image039
:ON→OFF、輸出二極體
Figure 02_image116
:OFF]:請再一併參閱第五圖本發明之第二操作階段等效電路圖所示,第二階段開始於
Figure 02_image127
,第二舉升二極體
Figure 02_image039
轉態成OFF,所有二極體均為逆向偏壓而OFF,第一功率開關
Figure 02_image081
、第二功率開關
Figure 02_image082
皆為ON。輸入電壓
Figure 02_image001
跨於兩個耦合電感的一次側,即跨於第一磁化電感
Figure 02_image007
和第一漏電感
Figure 02_image009
以及第二磁化電感
Figure 02_image015
和第二漏電感
Figure 02_image017
,第一漏電感電流
Figure 02_image120
和第二漏電感電流
Figure 02_image131
呈線性上升,斜率均為
Figure 02_image133
,從能量觀點而言,兩個耦合電感的一次側在本階段作儲存能量的動作。當
Figure 02_image135
,第二功率開關
Figure 02_image082
切換成OFF時,本階段結束。 second stage[
Figure 02_image129
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: ON, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: OFF, second lift diode
Figure 02_image039
: ON→OFF, output diode
Figure 02_image116
: OFF]: Please refer to Figure 5 again, as shown in the equivalent circuit diagram of the second operation stage of the present invention, the second stage begins at
Figure 02_image127
, the second lift diode
Figure 02_image039
Turning to OFF, all diodes are reverse biased and OFF, the first power switch
Figure 02_image081
, the second power switch
Figure 02_image082
All are ON. Input voltage
Figure 02_image001
Across the primary sides of the two coupled inductors, that is, across the first magnetizing inductance
Figure 02_image007
and the first leakage inductance
Figure 02_image009
and the second magnetizing inductance
Figure 02_image015
and the second leakage inductance
Figure 02_image017
, the first leakage inductance current
Figure 02_image120
and the second leakage inductance current
Figure 02_image131
rises linearly, with a slope of
Figure 02_image133
, from an energy point of view, the primary side of the two coupled inductors stores energy at this stage. when
Figure 02_image135
, the second power switch
Figure 02_image082
When switched to OFF, this phase ends.

第三階段[

Figure 02_image137
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:ON→OFF、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:ON、第一舉升二極體
Figure 02_image037
:ON、第二舉升二極體
Figure 02_image039
:ON、輸出二極體
Figure 02_image116
:ON]:請再一併參閱第六圖本發明之第三操作階段等效電路圖所示,第三階段開始於
Figure 02_image139
,第二功率開關
Figure 02_image082
切換為OFF,第一功率開關
Figure 02_image081
保持為ON,第一輸出二極體
Figure 02_image033
和第二舉升二極體
Figure 02_image039
為逆向偏壓。第二漏電感電流
Figure 02_image131
的連續性使得第二輸出二極體
Figure 02_image035
轉態為ON,第二漏電感電流
Figure 02_image131
流經第二輸出二極體
Figure 02_image035
、第二輸出電容
Figure 02_image025
和第一功率開關
Figure 02_image081
,對第二輸出電容
Figure 02_image025
充電。耦合電感之第二磁化電感
Figure 02_image015
以返馳式模式傳送能量至二次側,使得第一舉升二極體
Figure 02_image037
轉態為ON。第一舉升二極體電流
Figure 02_image140
對第一舉升電容
Figure 02_image027
充電,此階段輸出二極體
Figure 02_image116
導通,第二舉升電容
Figure 02_image029
對輸出電容
Figure 02_image031
對負載
Figure 02_image043
放電,此時第二漏電感電流
Figure 02_image131
呈線性下降。當
Figure 02_image142
,第二漏電感電流
Figure 02_image144
線性下降,當
Figure 02_image145
,第二輸出電容
Figure 02_image147
轉為向上流動,本階段結束。 The third phase[
Figure 02_image137
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: ON→OFF, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: ON, the first lifting diode
Figure 02_image037
: ON, the second lifting diode
Figure 02_image039
: ON, output diode
Figure 02_image116
: ON]: Please refer to Figure 6 again, as shown in the equivalent circuit diagram of the third operation stage of the present invention, the third stage begins at
Figure 02_image139
, the second power switch
Figure 02_image082
switch to OFF, the first power switch
Figure 02_image081
remains ON, the first output diode
Figure 02_image033
and a second boost diode
Figure 02_image039
for reverse bias. Second leakage inductance current
Figure 02_image131
The continuity makes the second output diode
Figure 02_image035
transition to ON, the second leakage inductance current
Figure 02_image131
flows through the second output diode
Figure 02_image035
, the second output capacitor
Figure 02_image025
and the first power switch
Figure 02_image081
, for the second output capacitor
Figure 02_image025
Charge. The second magnetizing inductance of the coupled inductor
Figure 02_image015
transfers energy to the secondary in flyback mode such that the first lift diode
Figure 02_image037
Turn the state to ON. 1st lift diode current
Figure 02_image140
to the first lift capacitor
Figure 02_image027
charging, the output diode at this stage
Figure 02_image116
turn on, the second boost capacitor
Figure 02_image029
to the output capacitor
Figure 02_image031
on load
Figure 02_image043
discharge, at this time the second leakage inductance current
Figure 02_image131
Decreases linearly. when
Figure 02_image142
, the second leakage inductance current
Figure 02_image144
declines linearly, when
Figure 02_image145
, the second output capacitor
Figure 02_image147
Turn to upward flow and this phase ends.

第四階段[

Figure 02_image149
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:OFF、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:ON、第一舉升二極體
Figure 02_image037
:ON、第二舉升二極體
Figure 02_image039
:ON、輸出二極體
Figure 02_image116
:ON]:請再一併參閱第七圖本發明之第四操作階段等效電路圖所示,第四階段開始於
Figure 02_image151
,第二功率開關
Figure 02_image082
保持OFF,第一功率開關
Figure 02_image081
保持ON,第一輸出二極體
Figure 02_image033
和第二舉升二極體
Figure 02_image039
逆向偏壓。與第三階段比較,除了第二輸出電容
Figure 02_image147
的流向不同,其餘的電路操作相同。第二漏電感電流
Figure 02_image131
持續線性下降,此階段輸出電容
Figure 02_image031
和第一舉升電容
Figure 02_image027
為儲存能量,第二舉升電容
Figure 02_image029
為釋放能量。當
Figure 02_image152
,第二漏電感
Figure 02_image017
儲存的能量完全釋放完畢,即
Figure 02_image154
,第二輸出二極體
Figure 02_image035
自然轉態成OFF,本階段結束。 The fourth stage [
Figure 02_image149
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: OFF, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: ON, the first lifting diode
Figure 02_image037
: ON, the second lifting diode
Figure 02_image039
: ON, output diode
Figure 02_image116
: ON]: Please refer to Figure 7 again, as shown in the equivalent circuit diagram of the fourth operation stage of the present invention, the fourth stage begins at
Figure 02_image151
, the second power switch
Figure 02_image082
kept OFF, the first power switch
Figure 02_image081
remains ON, the first output diode
Figure 02_image033
and a second boost diode
Figure 02_image039
reverse bias. compared with the third stage, except that the second output capacitor
Figure 02_image147
The flow direction is different, and the rest of the circuit operates the same. Second leakage inductance current
Figure 02_image131
continuous linear decline, at this stage the output capacitance
Figure 02_image031
and the first lift capacitor
Figure 02_image027
To store energy, the second boost capacitor
Figure 02_image029
to release energy. when
Figure 02_image152
, the second leakage inductance
Figure 02_image017
The stored energy is completely released, that is,
Figure 02_image154
, the second output diode
Figure 02_image035
Natural transition to OFF, this phase ends.

第五階段[

Figure 02_image156
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:OFF、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:ON→OFF、第一舉升二極體
Figure 02_image037
:ON、第二舉升二極體
Figure 02_image039
:OFF、輸出二極體
Figure 02_image116
:ON]:請再一併參閱第八圖本發明之第五操作階段等效電路圖所示,第五階段開始於
Figure 02_image158
,此時第二漏電感
Figure 02_image017
的能量釋放完畢,第二輸出二極體
Figure 02_image035
轉態成OFF。第二磁化電感電流
Figure 02_image159
完全由耦合電感之一次側反射到二次側,第一舉升二極體電流
Figure 02_image161
對第一舉升電容
Figure 02_image027
充電,第二舉升電容
Figure 02_image029
對輸出電容
Figure 02_image031
和負載
Figure 02_image043
釋放能量,此時第一功率開關
Figure 02_image081
的電流等於第一磁化電感
Figure 02_image007
和第二磁化電感
Figure 02_image015
的電流總和。當
Figure 02_image162
,第二功率開關
Figure 02_image082
切換為ON時,本階段結束。 fifth stage [
Figure 02_image156
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: OFF, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: ON→OFF, the first lifting diode
Figure 02_image037
: ON, the second lifting diode
Figure 02_image039
: OFF, output diode
Figure 02_image116
: ON]: Please refer to the eighth figure and the equivalent circuit diagram of the fifth operation stage of the present invention, the fifth stage starts at
Figure 02_image158
, at this time the second leakage inductance
Figure 02_image017
After the energy is released, the second output diode
Figure 02_image035
Turn the state to OFF. second magnetizing inductance current
Figure 02_image159
completely reflected from the primary side of the coupled inductor to the secondary side, the first lift diode current
Figure 02_image161
to the first lift capacitor
Figure 02_image027
charging, the second lift capacitor
Figure 02_image029
to the output capacitor
Figure 02_image031
and load
Figure 02_image043
release energy, at this time the first power switch
Figure 02_image081
A current equal to the first magnetizing inductance
Figure 02_image007
and the second magnetizing inductance
Figure 02_image015
the sum of the currents. when
Figure 02_image162
, the second power switch
Figure 02_image082
When toggled ON, this phase ends.

第六階段[

Figure 02_image164
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:OFF→ON、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:ON、第二舉升二極體
Figure 02_image039
:OFF、輸出二極體
Figure 02_image116
:ON]:請再一併參閱第九圖本發明之第六操作階段等效電路圖所示,第六階段開始於
Figure 02_image166
,第二功率開關
Figure 02_image082
切換成ON,且第一功率開關
Figure 02_image081
保持ON,第一輸出二極體
Figure 02_image033
、第二輸出二極體
Figure 02_image035
和第二舉升二極體
Figure 02_image039
逆向偏壓。由於第二漏電感
Figure 02_image017
的存在,第二功率開關
Figure 02_image082
具有零電流切換[ZCS]的柔切性能,降低切換損失。第二漏電感電流
Figure 02_image131
上升,當
Figure 02_image167
時,第二磁化電感
Figure 02_image015
的儲能仍然藉由耦合電感傳送二次側。第一舉升二極體
Figure 02_image037
和輸出二極體
Figure 02_image116
仍保持如前一階段的導通狀態,第一舉升二極體電流
Figure 02_image169
和輸出二極體電流
Figure 02_image170
下降。第一漏電感
Figure 02_image009
和第二漏電感
Figure 02_image017
控制了第一舉升二極體
Figure 02_image037
和輸出二極體
Figure 02_image116
電流下降速率,因此可緩和第一舉升二極體
Figure 02_image037
和輸出二極體
Figure 02_image116
反向恢復問題。當
Figure 02_image172
,第一舉升二極體電流
Figure 02_image169
和輸出二極體電流
Figure 02_image170
下降至0,第一舉升二極體
Figure 02_image037
和輸出二極體
Figure 02_image116
轉態成OFF時,本階段結束。 Sixth stage [
Figure 02_image164
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: OFF→ON, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: ON, the second lifting diode
Figure 02_image039
: OFF, output diode
Figure 02_image116
: ON]: Please refer to Figure 9 again, as shown in the equivalent circuit diagram of the sixth operation stage of the present invention, the sixth stage begins at
Figure 02_image166
, the second power switch
Figure 02_image082
switched ON, and the first power switch
Figure 02_image081
remains ON, the first output diode
Figure 02_image033
, the second output diode
Figure 02_image035
and a second boost diode
Figure 02_image039
reverse bias. Due to the second leakage inductance
Figure 02_image017
presence of a second power switch
Figure 02_image082
It has soft cutting performance of zero current switching [ZCS] and reduces switching loss. Second leakage inductance current
Figure 02_image131
rise when
Figure 02_image167
, the second magnetizing inductance
Figure 02_image015
The stored energy is still transferred to the secondary side through the coupled inductor. first lift diode
Figure 02_image037
and output diode
Figure 02_image116
remains on as in the previous stage, the first boost diode current
Figure 02_image169
and output diode current
Figure 02_image170
decline. first leakage inductance
Figure 02_image009
and the second leakage inductance
Figure 02_image017
controls the first lift diode
Figure 02_image037
and output diode
Figure 02_image116
current drop rate, thus moderating the first lift diode
Figure 02_image037
and output diode
Figure 02_image116
Reverse recovery problem. when
Figure 02_image172
, the first lift diode current
Figure 02_image169
and output diode current
Figure 02_image170
down to 0, the first lift diode
Figure 02_image037
and output diode
Figure 02_image116
When the state is turned OFF, this phase ends.

第七階段[

Figure 02_image174
]:[第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:ON、第一輸出二極體
Figure 02_image033
:OFF、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:ON→OFF、第二舉升二極體
Figure 02_image039
:OFF、輸出二極體
Figure 02_image116
:ON→OFF]:請再一併參閱第十圖本發明之第七操作階段等效電路圖所示,第七階段開始於
Figure 02_image176
,第一舉升二極體
Figure 02_image037
和輸出二極體
Figure 02_image116
轉態成OFF,第一輸出二極體
Figure 02_image033
、第二輸出二極體
Figure 02_image035
和第二舉升二極體
Figure 02_image039
均為逆向偏壓,第一功率開關
Figure 02_image081
和第二功率開關
Figure 02_image082
皆為ON。輸入電壓
Figure 02_image001
跨於兩個耦合電感的一次側,即跨於第一磁化電感
Figure 02_image007
和第一漏電感
Figure 02_image009
以及第二磁化電感
Figure 02_image015
和第二漏電感
Figure 02_image017
,第一漏電感電流
Figure 02_image120
和第二漏電感電流
Figure 02_image131
呈線性上升,斜率均為
Figure 02_image133
,從能量觀點而言,兩個耦合電感的一次側在本階段作儲存能量的動作,輸出電容
Figure 02_image031
對負載
Figure 02_image043
釋放能量。當
Figure 02_image177
,第一功率開關
Figure 02_image081
切換成OFF時,本階段結束。 The seventh stage [
Figure 02_image174
]: [First power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: ON, the first output diode
Figure 02_image033
: OFF, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: ON→OFF, the second lifting diode
Figure 02_image039
: OFF, output diode
Figure 02_image116
: ON→OFF]: Please refer to Figure 10 again, as shown in the equivalent circuit diagram of the seventh operation stage of the present invention, the seventh stage begins at
Figure 02_image176
, the first lift diode
Figure 02_image037
and output diode
Figure 02_image116
transition to OFF, the first output diode
Figure 02_image033
, the second output diode
Figure 02_image035
and a second boost diode
Figure 02_image039
are reverse biased, the first power switch
Figure 02_image081
and a second power switch
Figure 02_image082
All are ON. Input voltage
Figure 02_image001
Across the primary sides of the two coupled inductors, that is, across the first magnetizing inductance
Figure 02_image007
and the first leakage inductance
Figure 02_image009
and the second magnetizing inductance
Figure 02_image015
and the second leakage inductance
Figure 02_image017
, the first leakage inductance current
Figure 02_image120
and the second leakage inductance current
Figure 02_image131
rises linearly, with a slope of
Figure 02_image133
, from the perspective of energy, the primary sides of the two coupled inductors store energy at this stage, and the output capacitor
Figure 02_image031
on load
Figure 02_image043
emit energy. when
Figure 02_image177
, the first power switch
Figure 02_image081
When switched to OFF, this phase ends.

第八階段[

Figure 02_image179
]:[第一功率開關
Figure 02_image081
:ON→OFF、第二功率開關
Figure 02_image082
:ON、第一輸出二極體
Figure 02_image033
:ON、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:OFF、第二舉升二極體
Figure 02_image039
:ON、輸出二極體
Figure 02_image116
:OFF]:請再一併參閱第十一圖本發明之第八操作階段等效電路圖所示,第八階段開始於
Figure 02_image181
,第一功率開關
Figure 02_image081
切換為OFF,第二功率開關
Figure 02_image082
保持為ON。第一漏電感電流
Figure 02_image120
的連續性使得第一輸出二極體
Figure 02_image033
轉態為ON,第一漏電感電流
Figure 02_image120
流經第一輸出電容
Figure 02_image023
和第一輸出二極體
Figure 02_image033
,對第一輸出電容
Figure 02_image023
充電,耦合電感之第一磁化電感
Figure 02_image007
的儲能以返馳式模式傳送至二次側,使得第二舉升二極體
Figure 02_image039
轉態為ON,本階段第一舉升電容
Figure 02_image027
經由第二舉升二極體
Figure 02_image039
對第二舉升電容
Figure 02_image029
充電,第二舉升二極體電流
Figure 02_image124
對第一舉升電容
Figure 02_image027
、第二舉升電容
Figure 02_image029
充電此時第一漏電感電流
Figure 02_image120
呈線性下降。當
Figure 02_image182
,第一漏電感
Figure 02_image009
儲存的能量完全釋放完畢,即
Figure 02_image184
,第一輸出二極體
Figure 02_image033
自然轉態成OFF時,本階段結束。 Eighth stage [
Figure 02_image179
]: [First power switch
Figure 02_image081
: ON→OFF, second power switch
Figure 02_image082
: ON, the first output diode
Figure 02_image033
: ON, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: OFF, second lift diode
Figure 02_image039
: ON, output diode
Figure 02_image116
: OFF]: Please refer to Figure 11 and the equivalent circuit diagram of the eighth operation stage of the present invention, the eighth stage starts at
Figure 02_image181
, the first power switch
Figure 02_image081
switch to OFF, the second power switch
Figure 02_image082
Keep it ON. The first leakage inductance current
Figure 02_image120
The continuity makes the first output diode
Figure 02_image033
transition to ON, the first leakage inductance current
Figure 02_image120
flows through the first output capacitor
Figure 02_image023
and the first output diode
Figure 02_image033
, for the first output capacitor
Figure 02_image023
Charging, the first magnetizing inductance of the coupled inductance
Figure 02_image007
The stored energy is transferred to the secondary side in flyback mode, making the second lift diode
Figure 02_image039
Turning to ON, the first lift capacitor in this stage
Figure 02_image027
via the second lift diode
Figure 02_image039
to the second boost capacitor
Figure 02_image029
charging, the second lift diode current
Figure 02_image124
to the first lift capacitor
Figure 02_image027
, the second lift capacitor
Figure 02_image029
charging at this time the first leakage inductance current
Figure 02_image120
Decreases linearly. when
Figure 02_image182
, the first leakage inductance
Figure 02_image009
The stored energy is completely released, that is,
Figure 02_image184
, the first output diode
Figure 02_image033
This phase ends when the natural transition is turned OFF.

第九階段[

Figure 02_image186
]:[第一功率開關
Figure 02_image081
:OFF、第二功率開關
Figure 02_image082
:ON、第一輸出二極體
Figure 02_image033
:ON→OFF、第二輸出二極體
Figure 02_image035
:OFF、第一舉升二極體
Figure 02_image037
:OFF、第二舉升二極體
Figure 02_image039
:ON、輸出二極體
Figure 02_image116
:OFF]:請再一併參閱第十二圖本發明之第九操作階段等效電路圖所示,第九階段開始於
Figure 02_image188
,此時第一漏電感
Figure 02_image009
的能量完全釋放到第一輸出電容
Figure 02_image023
,第一輸出二極體
Figure 02_image033
自然轉態成OFF。第一磁化電感電流
Figure 02_image189
由耦合電感之一次側完全反射到二次側,第一舉升電容
Figure 02_image027
經由第二舉升二極體
Figure 02_image039
對第二舉升電容
Figure 02_image029
充電,此時第二功率開關
Figure 02_image082
的電流等於第一磁化電感
Figure 02_image007
和第二磁化電感
Figure 02_image015
的電流總和。當
Figure 02_image191
,第一功率開關
Figure 02_image081
切換為ON時,本階段結束,進入下一個切換週期。 Ninth stage [
Figure 02_image186
]: [First power switch
Figure 02_image081
: OFF, second power switch
Figure 02_image082
: ON, the first output diode
Figure 02_image033
: ON→OFF, second output diode
Figure 02_image035
: OFF, the first lift diode
Figure 02_image037
: OFF, second lift diode
Figure 02_image039
: ON, output diode
Figure 02_image116
: OFF]: Please refer to Figure 12 and the equivalent circuit diagram of the ninth operation stage of the present invention, the ninth stage starts at
Figure 02_image188
, at this time the first leakage inductance
Figure 02_image009
The energy is fully released to the first output capacitor
Figure 02_image023
, the first output diode
Figure 02_image033
Natural transition to OFF. first magnetizing inductance current
Figure 02_image189
Completely reflected from the primary side of the coupled inductor to the secondary side, the first lift capacitor
Figure 02_image027
via the second lift diode
Figure 02_image039
to the second boost capacitor
Figure 02_image029
charging, at this time the second power switch
Figure 02_image082
A current equal to the first magnetizing inductance
Figure 02_image007
and the second magnetizing inductance
Figure 02_image015
the sum of the currents. when
Figure 02_image191
, the first power switch
Figure 02_image081
When switching to ON, this stage ends and enters the next switching cycle.

由以上的該轉換器(1)電路動作分析可知,轉換器有以下優點:第一功率開關

Figure 02_image081
和第二功率開關
Figure 02_image082
具有零電流切換[ZCS]性能,可減少切換損失;第一輸出二極體
Figure 02_image033
和第二輸出二極體
Figure 02_image035
沒有反向恢復損失;因漏電感的存在,能夠緩和第一舉升二極體
Figure 02_image037
和第二舉升二極體
Figure 02_image039
的反向恢復問題。漏電感能量可回收再利用,不但可改善效率,也可避免造成突波電壓。 From the above analysis of the circuit action of the converter (1), it can be seen that the converter has the following advantages: the first power switch
Figure 02_image081
and a second power switch
Figure 02_image082
It has zero-current switching [ZCS] performance, which can reduce switching loss; the first output diode
Figure 02_image033
and a second output diode
Figure 02_image035
No reverse recovery loss; due to the presence of leakage inductance, the first lift diode can be moderated
Figure 02_image037
and a second boost diode
Figure 02_image039
reverse recovery problem. Leakage inductance energy can be recycled and reused, which not only improves efficiency, but also avoids surge voltage.

以下進行該轉換器(1)穩態特性分析,為了簡化分析,假設各開關及各二極體導通壓降為零,並且忽略時間極短的暫態階段,只考慮第二、三、四、五、七及八階段。各電容夠大,忽略電壓漣波,使得電容電壓在一個切換週期內視為常數。The steady-state characteristic analysis of the converter (1) is carried out below. In order to simplify the analysis, it is assumed that the conduction voltage drop of each switch and each diode is zero, and the extremely short transient phase is ignored, and only the second, third, fourth, and second phases are considered. Five, seven and eight stages. Each capacitor is large enough to ignore the voltage ripple so that the capacitor voltage is considered constant within one switching cycle.

電壓增益分析:Voltage Gain Analysis:

由於第一輸出電容

Figure 02_image023
和第二輸出電容
Figure 02_image025
的電壓可視為傳統升壓型轉換器的輸出電壓,因此根據第一磁化電感
Figure 02_image007
和第二磁化電感
Figure 02_image015
滿足伏秒平衡定理,可推導得到電壓
Figure 02_image193
Figure 02_image195
為 Due to the first output capacitor
Figure 02_image023
and a second output capacitor
Figure 02_image025
The voltage of can be regarded as the output voltage of a conventional boost converter, so according to the first magnetizing inductance
Figure 02_image007
and the second magnetizing inductance
Figure 02_image015
Satisfying the volt-second balance theorem, the voltage can be derived
Figure 02_image193
and
Figure 02_image195
for

Figure 02_image197
(1)
Figure 02_image197
(1)

耦合電感二次側的輸出電容電壓

Figure 02_image199
Figure 02_image201
,可藉由耦合電感一次側電壓反射至二次測電壓推導而得到。在第三階段,第一功率開關
Figure 02_image081
:ON、第二功率開關
Figure 02_image082
:OFF,而且第一舉升二極體
Figure 02_image037
導通,電壓
Figure 02_image199
為 The output capacitor voltage on the secondary side of the coupled inductor
Figure 02_image199
and
Figure 02_image201
, which can be obtained by reflecting the primary side voltage of the coupled inductor to the secondary measured voltage. In the third stage, the first power switch
Figure 02_image081
: ON, second power switch
Figure 02_image082
: OFF, and the first lift diode
Figure 02_image037
conduction, voltage
Figure 02_image199
for

Figure 02_image203
(2)
Figure 02_image203
(2)

在第八階段,第一功率開關

Figure 02_image081
:OFF、第二功率開關
Figure 02_image082
:ON,而且第二舉升二極體
Figure 02_image039
導通,電壓
Figure 02_image205
為 In the eighth stage, the first power switch
Figure 02_image081
: OFF, second power switch
Figure 02_image082
: ON, and the second lift diode
Figure 02_image039
conduction, voltage
Figure 02_image205
for

Figure 02_image206
(3)
Figure 02_image206
(3)

在第三階段電路中,根據KVL定理,可知In the third stage circuit, according to the KVL theorem, we know

Figure 02_image208
(4)
Figure 02_image208
(4)

將(1)、(2)、(3)式的結果代入(4)式,可得總輸出電壓

Figure 02_image210
為 Substituting the results of (1), (2), and (3) into (4), the total output voltage can be obtained
Figure 02_image210
for

Figure 02_image212
(5)
Figure 02_image212
(5)

因此本轉換器的電壓增益

Figure 02_image214
為 Therefore the voltage gain of this converter
Figure 02_image214
for

Figure 02_image216
(6)
Figure 02_image216
(6)

Figure 02_image218
時,電壓增益
Figure 02_image214
與不同耦合電感的耦合係數
Figure 02_image220
的關係曲線,即如第十三圖本發明之不同耦合係數和電壓增益的關係曲線圖所示,可知耦合係數
Figure 02_image222
對電壓增益的影響非常小。若耦合係數
Figure 02_image223
,則理想的電壓增益M為 when
Figure 02_image218
When the voltage gain
Figure 02_image214
Coupling coefficient with different coupled inductance
Figure 02_image220
The relationship curve, that is, as shown in the relationship curve diagram of different coupling coefficients and voltage gains of the present invention in the thirteenth figure, it can be known that the coupling coefficient
Figure 02_image222
The effect on voltage gain is very small. If the coupling coefficient
Figure 02_image223
, then the ideal voltage gain M is

Figure 02_image225
(7)
Figure 02_image225
(7)

從上式可知本轉換器的電壓增益具有耦合電感匝數比

Figure 02_image227
和導通比
Figure 02_image229
兩個設計自由度。該轉換器(1)可藉由適當設計耦合電感的匝數比,達到高升壓比,且不必操作在極大的導通比。對應於耦合電感匝數比
Figure 02_image231
及導通比
Figure 02_image232
的電壓增益曲線,請參閱第十四圖本發明之電壓增益與導通比及不同耦合電感匝數比之曲線圖所示,可知當導通比
Figure 02_image233
Figure 02_image235
時,電壓增益為20倍。 It can be seen from the above formula that the voltage gain of the converter has the coupling inductor turns ratio
Figure 02_image227
and conduction ratio
Figure 02_image229
Two degrees of design freedom. The converter (1) can achieve a high step-up ratio by properly designing the turns ratio of the coupling inductor, and does not need to operate at a very large conduction ratio. Corresponding to the coupled inductor turns ratio
Figure 02_image231
and conduction ratio
Figure 02_image232
For the voltage gain curve, please refer to the fourteenth graph of the voltage gain and conduction ratio of the present invention and the curves of different coupling inductor turns ratios, it can be seen that when the conduction ratio
Figure 02_image233
,
Figure 02_image235
, the voltage gain is 20 times.

功率開關和二極體的電壓應力分析:Voltage Stress Analysis of Power Switches and Diodes:

在功率開關和二極體的電壓應力分析方面,為了簡化分析,忽略電容電壓漣波、二極體導通壓降及耦合電感的漏電感,及假設耦和係數k=1。In terms of the voltage stress analysis of the power switch and the diode, in order to simplify the analysis, the capacitor voltage ripple, the diode conduction voltage drop and the leakage inductance of the coupled inductor are ignored, and the coupling coefficient k=1 is assumed.

由該轉換器(1)操作的第三階段可求得第二功率開關

Figure 02_image021
的電壓應力 The third stage of operation of the converter (1) results in the second power switch
Figure 02_image021
voltage stress

Figure 02_image237
(8)
Figure 02_image237
(8)

由第八階段可求得第一功率開關

Figure 02_image019
的電壓應力 From the eighth stage, the first power switch can be obtained
Figure 02_image019
voltage stress

Figure 02_image239
(9)
Figure 02_image239
(9)

另一方面,由第三和第八階段也可求得二極體的電壓應力On the other hand, the voltage stress of the diode can also be obtained from the third and eighth stages

Figure 02_image241
(10)
Figure 02_image241
(10)

Figure 02_image243
(11)
Figure 02_image243
(11)

Figure 02_image245
(12)
Figure 02_image245
(12)

Figure 02_image247
(13)
Figure 02_image247
(13)

由於傳統交錯式升壓型轉換器的功率開關電壓應力為輸出電壓

Figure 02_image249
,而該轉換器(1)的開關電壓應力僅為輸出電壓
Figure 02_image250
Figure 02_image251
倍,因此可使用低額定耐壓具有較低導通電阻的MOSFET,可降低開關導通損失。另一方面,較低電壓應力的二極體可採用順向壓降較低的功率二極體,可降低導通損失。 Since the power switch voltage stress of a conventional interleaved boost converter is the output voltage
Figure 02_image249
, while the switching voltage stress of this converter (1) is only the output voltage
Figure 02_image250
of
Figure 02_image251
times, so MOSFETs with low rated withstand voltage and low on-resistance can be used, which can reduce switch conduction loss. On the other hand, diodes with lower voltage stress can use power diodes with lower forward voltage drop, which can reduce conduction loss.

依據上述電路動作分析結果,利用Is-Spice軟體作先期的模擬,轉換器規格:輸入電壓32V、輸出電壓400V、最大輸出功率1000W、切換頻率50kHz,耦合電感匝數比

Figure 02_image253
,驗證該轉換器(1)的特點,以下以模擬波形驗證與說明該轉換器(1)的特點[請再一併參閱第十五圖本發明之模擬電路示意圖所示]。 Based on the analysis results of the above circuit, Is-Spice software was used for preliminary simulation. The converter specifications: input voltage 32V, output voltage 400V, maximum output power 1000W, switching frequency 50kHz, coupling inductor turns ratio
Figure 02_image253
, to verify the characteristics of the converter (1), the following verification and description of the characteristics of the converter (1) with analog waveforms [please refer to the fifteenth figure shown in the analog circuit schematic diagram of the present invention].

A.驗證穩態特性:A. Verify steady-state characteristics:

首先驗證該轉換器(1)之穩態特性,滿載1000W時,請參閱第十六圖本發明之開關驅動信號、輸入電壓與輸出電壓波形圖所示,當輸入電壓32V、輸出電壓400V、耦合電感匝數比

Figure 02_image253
,則導通比的理論值大約
Figure 02_image255
,模擬結果符合該轉換器(1)電壓增益的公式。 First verify the steady-state characteristics of the converter (1). When the full load is 1000W, please refer to the sixteenth figure of the present invention for the switch drive signal, input voltage and output voltage waveform diagram. Inductance turns ratio
Figure 02_image253
, the theoretical value of the conduction ratio is about
Figure 02_image255
, the simulation results conform to the formula of the converter (1) voltage gain.

B.驗證開關電壓應力:B. Verify switch voltage stress:

請參閱第十七圖本發明之開關驅動信號與開關跨壓信號波形圖所示,可知當第一功率開關

Figure 02_image081
、第二功率開關
Figure 02_image257
為OFF時,其跨壓
Figure 02_image259
Figure 02_image261
都約為80V,僅為輸出電壓400V的五分之一,符合分析結果,比較傳統的升壓型轉換器,開關電壓應力為輸出電壓,該轉換器(1)的開關具有低電壓應力的優點。 Please refer to the seventeenth figure shown in the waveform diagram of the switch drive signal and the switch cross-voltage signal of the present invention, it can be known that when the first power switch
Figure 02_image081
, the second power switch
Figure 02_image257
is OFF, its voltage across the
Figure 02_image259
or
Figure 02_image261
Both are about 80V, which is only one-fifth of the output voltage of 400V, which is consistent with the analysis results. Compared with the traditional boost converter, the switch voltage stress is the output voltage. The switch of the converter (1) has the advantage of low voltage stress .

C.驗證具有低輸入漣波電流性能與CCM操作:C. Verify low input ripple current performance with CCM operation:

請參閱第十八圖本發明之滿載1000W時,耦合電感的漏電感電流及總輸入電流波形圖所示,可知

Figure 02_image263
Figure 02_image264
的漣波電流大小大約33A,而輸入電流的漣波電流大小僅為約1.6A,很明顯地,交錯式操作具有降低輸入漣波電流效用。請參閱第十九圖本發明之耦合電感的磁化電感電流波形圖所示,驗證該轉換器(1)操作在連續導通模式[CCM]。 Please refer to the 18th figure of the present invention when the full load is 1000W, the leakage inductance current of the coupled inductor and the total input current waveform diagram, we can know
Figure 02_image263
,
Figure 02_image264
The ripple current of the input current is about 33A, while the ripple current of the input current is only about 1.6A. Obviously, the interleaved operation has the effect of reducing the input ripple current. Please refer to Figure 19, which shows the magnetizing inductance current waveform of the coupled inductor of the present invention, and verify that the converter (1) operates in continuous conduction mode [CCM].

D.驗證二極體反向恢復電流問題:D. Verify the diode reverse recovery current problem:

請參閱第二十圖本發明之輸出二極體的電流及電壓波形圖所示,可知

Figure 02_image266
Figure 02_image268
都沒有反向恢復問題,因此沒有反向恢復損失。另一方面可看出,第一輸出二極體
Figure 02_image033
電壓應力約為80V,只有輸出電壓的五之一,第二輸出二極體
Figure 02_image035
電壓應力大約為160V,只有輸出電壓的五分之二,符合分析結果。 Please refer to the current and voltage waveforms of the output diode of the present invention in Figure 20, it can be known
Figure 02_image266
and
Figure 02_image268
Neither has reverse recovery issues, so there is no reverse recovery penalty. On the other hand it can be seen that the first output diode
Figure 02_image033
The voltage stress is about 80V, only one-fifth of the output voltage, the second output diode
Figure 02_image035
The voltage stress is about 160V, only two-fifths of the output voltage, which is consistent with the analysis results.

請參閱第二十一圖本發明之舉升二極體和輸出二極體的電流及電壓波形圖所示,第一舉升二極體

Figure 02_image037
和第二舉升二極體
Figure 02_image039
的電壓應力均為160V,符合分析結果。第一舉升二極體
Figure 02_image037
、第二舉升二極體
Figure 02_image039
和輸出二極體
Figure 02_image041
的電流幾乎沒有反向恢復電流,因為耦合電感中第一漏電感
Figure 02_image009
和第二漏電感
Figure 02_image017
的存在緩和了反向恢復問題。 Please refer to the current and voltage waveforms of the lifting diode and the output diode in Figure 21 of the present invention, the first lifting diode
Figure 02_image037
and a second boost diode
Figure 02_image039
The voltage stresses are all 160V, which is in line with the analysis results. first lift diode
Figure 02_image037
, the second lifting diode
Figure 02_image039
and output diode
Figure 02_image041
The current has almost no reverse recovery current, because the first leakage inductance in the coupled inductor
Figure 02_image009
and the second leakage inductance
Figure 02_image017
The existence of mit alleviates the reverse recovery problem.

E.驗證輸出電容電壓:E. Verify output capacitor voltage:

請參閱第二十二圖本發明之電容的電壓波形圖所示,電容電壓

Figure 02_image270
Figure 02_image271
Figure 02_image272
大約都等於80V,電容電壓
Figure 02_image273
大約等於160V,電容電壓
Figure 02_image275
約等於400V,符合分析結果。 Please refer to Figure 22 shown in the voltage waveform diagram of the capacitor of the present invention, the capacitor voltage
Figure 02_image270
,
Figure 02_image271
and
Figure 02_image272
Approximately equal to 80V, the capacitor voltage
Figure 02_image273
Approximately equal to 160V, the capacitor voltage
Figure 02_image275
It is approximately equal to 400V, which is in line with the analysis results.

根據以上的模擬波形驗證,該轉換器(1)的特性與優點歸納如下:According to the above simulation waveform verification, the characteristics and advantages of the converter (1) are summarized as follows:

1.電壓增益公式、各開關電壓應力、各二極體電壓應力及每個輸出電容電壓值都與穩態特性分析的推導結果都十分符合。1. The voltage gain formula, the voltage stress of each switch, the voltage stress of each diode, and the voltage value of each output capacitor are all in good agreement with the derivation results of the steady-state characteristic analysis.

2.高電壓增益的達成,確實不必操作在極大的導通比。2. To achieve high voltage gain, it is really not necessary to operate at a very large conduction ratio.

3.轉換器兩個功率開關的電壓應力只有輸出電壓的五分之一,可以使用導通電阻較小的低額定耐壓MOSFET,以降低導通損失。3. The voltage stress of the two power switches of the converter is only one-fifth of the output voltage, and a low-rated withstand voltage MOSFET with a small on-resistance can be used to reduce the conduction loss.

4.由於輸出二極體在轉態成OFF之前,其流經的電流先降為零,所以輸出二極體沒有反向恢復功率損失問題。4. Since the current flowing through the output diode drops to zero before it turns OFF, the output diode does not have the problem of reverse recovery power loss.

5.耦合電感的漏電感能量,能夠回收再利用,避免了造成功率開關的電壓突波問題。5. The leakage inductance energy of the coupled inductor can be recycled and reused, avoiding the voltage surge problem caused by the power switch.

6.由於交錯式操作,使得耦合電感一次側繞組的電流漣波能相消,降低輸入電流漣波大小。6. Due to the interleaved operation, the current ripple of the primary side winding of the coupled inductor can be eliminated, reducing the size of the input current ripple.

然而前述之實施例或圖式並非限定本發明之產品結構或使用方式,任何所屬技術領域中具有通常知識者之適當變化或修飾,皆應視為不脫離本發明之專利範疇。However, the aforementioned embodiments or drawings do not limit the product structure or usage of the present invention, and any appropriate changes or modifications by those with ordinary knowledge in the technical field shall be considered as not departing from the patent scope of the present invention.

綜上所述,本發明實施例確能達到所預期之使用功效,又其所揭露之具體構造,不僅未曾見諸於同類產品中,亦未曾公開於申請前,誠已完全符合專利法之規定與要求,爰依法提出發明專利之申請,懇請惠予審查,並賜准專利,則實感德便。To sum up, the embodiment of the present invention can indeed achieve the expected use effect, and the specific structure disclosed by it has not only never been seen in similar products, nor has it been disclosed before the application, and it has fully complied with the provisions of the Patent Law In accordance with the requirements, it is very convenient to file an application for a patent for invention in accordance with the law, and sincerely ask for the review and approval of the patent.

1:轉換器1: Converter

Figure 02_image001
:輸入電壓
Figure 02_image001
:Input voltage

Figure 02_image003
:第一耦合電感一次側
Figure 02_image003
: Primary side of the first coupled inductor

Figure 02_image005
:第一耦合電感二次側
Figure 02_image005
: Secondary side of the first coupled inductor

Figure 02_image007
:第一磁化電感
Figure 02_image007
: first magnetizing inductance

Figure 02_image009
:第一漏電感
Figure 02_image009
: The first leakage inductance

Figure 02_image011
:第二耦合電感一次側
Figure 02_image011
: Primary side of the second coupled inductor

Figure 02_image013
:第二耦合電感二次側
Figure 02_image013
: Secondary side of the second coupled inductor

Figure 02_image015
:第二磁化電感
Figure 02_image015
: second magnetizing inductance

Figure 02_image017
:第二漏電感
Figure 02_image017
: Second leakage inductance

Figure 02_image019
:第一功率開關
Figure 02_image019
: first power switch

Figure 02_image021
:第二功率開關
Figure 02_image021
: Second power switch

Figure 02_image023
:第一輸出電容
Figure 02_image023
: first output capacitance

Figure 02_image025
:第二輸出電容
Figure 02_image025
: Second output capacitor

Figure 02_image027
:第一舉升電容
Figure 02_image027
: The first lift capacitor

Figure 02_image029
:第二舉升電容
Figure 02_image029
: Second lift capacitor

Figure 02_image031
:輸出電容
Figure 02_image031
: output capacitance

Figure 02_image033
:第一輸出二極體
Figure 02_image033
: The first output diode

Figure 02_image035
:第二輸出二極體
Figure 02_image035
: Second output diode

Figure 02_image037
:第一舉升二極體
Figure 02_image037
: 1st lift diode

Figure 02_image039
:第二舉升二極體
Figure 02_image039
: Second lift diode

Figure 02_image041
:輸出二極體
Figure 02_image041
: output diode

Figure 02_image043
:負載
Figure 02_image043
:load

2:升壓型轉換器2: Boost converter

3:交錯式升壓型轉換器3: Interleaved Boost Converter

第一圖:本發明之電路圖The first figure: the circuit diagram of the present invention

第二圖:本發明之等效電路圖The second figure: the equivalent circuit diagram of the present invention

第三圖:本發明之主要元件時序波形圖Figure 3: Timing waveform diagram of the main components of the present invention

第四圖:本發明之第一操作階段等效電路圖Figure 4: Equivalent circuit diagram of the first operating stage of the present invention

第五圖:本發明之第二操作階段等效電路圖The fifth figure: the equivalent circuit diagram of the second operation stage of the present invention

第六圖:本發明之第三操作階段等效電路圖Figure 6: Equivalent circuit diagram of the third operating stage of the present invention

第七圖:本發明之第四操作階段等效電路圖The seventh figure: the equivalent circuit diagram of the fourth operation stage of the present invention

第八圖:本發明之第五操作階段等效電路圖Figure 8: Equivalent circuit diagram of the fifth operating stage of the present invention

第九圖:本發明之第六操作階段等效電路圖Figure 9: Equivalent circuit diagram of the sixth operating stage of the present invention

第十圖:本發明之第七操作階段等效電路圖Figure 10: Equivalent circuit diagram of the seventh operation stage of the present invention

第十一圖:本發明之第八操作階段等效電路圖Figure 11: Equivalent circuit diagram of the eighth operation stage of the present invention

第十二圖:本發明之第九操作階段等效電路圖Figure 12: Equivalent circuit diagram of the ninth operating stage of the present invention

第十三圖:本發明之不同耦合係數和電壓增益的關係曲線圖The thirteenth figure: the relational graph of different coupling coefficients and voltage gain of the present invention

第十四圖:本發明之電壓增益與導通比及不同耦合電感匝數比之曲線圖Figure 14: The graph of voltage gain, conduction ratio and turns ratio of different coupling inductors according to the present invention

第十五圖:本發明之模擬電路示意圖Figure 15: Schematic diagram of the analog circuit of the present invention

第十六圖:本發明之開關驅動信號、輸入電壓與輸出電壓波形圖Figure 16: Switch drive signal, input voltage and output voltage waveform diagram of the present invention

第十七圖:本發明之開關驅動信號與開關跨壓信號波形圖Figure 17: Waveform diagram of switch drive signal and switch cross-voltage signal of the present invention

第十八圖:本發明之滿載1000W時,耦合電感的漏電感電流及總輸入電流波形圖Figure 18: When the present invention is fully loaded with 1000W, the leakage inductance current of the coupled inductor and the waveform diagram of the total input current

第十九圖:本發明之耦合電感的磁化電感電流波形圖Figure 19: The magnetizing inductance current waveform diagram of the coupled inductance of the present invention

第二十圖:本發明之輸出二極體的電流及電壓波形圖Figure 20: Current and voltage waveforms of the output diode of the present invention

第二十一圖:本發明之舉升二極體和輸出二極體的電流及電壓波形圖Figure 21: Current and voltage waveforms of the lifting diode and the output diode of the present invention

第二十二圖:本發明之電容的電壓波形圖Figure 22: the voltage waveform diagram of the capacitor of the present invention

第二十三圖:現有之傳統升壓型轉換器電路圖Figure 23: Circuit Diagram of Existing Traditional Boost Converter

第二十四圖:現有之傳統升壓型轉換器的輸出電壓增益對開關導通比的關係曲線圖Figure 24: The relationship between the output voltage gain and the switch conduction ratio of the existing conventional boost converter

第二十五圖:現有之傳統升壓型轉換器的效率對開關導通比的關係曲線圖Figure 25: The relationship between the efficiency of the existing traditional boost converter and the switch conduction ratio

第二十六圖:現有之交錯式升壓型轉換器電路圖Figure 26: Circuit diagram of the existing interleaved boost converter

1:轉換器 1: Converter

V in:輸入電壓 V in : input voltage

N p1:第一耦合電感一次側 N p 1 : Primary side of the first coupled inductor

N s1:第一耦合電感二次側 N s 1 : Secondary side of the first coupled inductor

L m1:第一磁化電感 L m 1 : first magnetizing inductance

L k1:第一漏電感 L k 1 : the first leakage inductance

N p2:第二耦合電感一次側 N p 2 : Primary side of the second coupled inductor

N s2:第二耦合電感二次側 N s 2 : Secondary side of the second coupled inductor

L m2:第二磁化電感 L m 2 : second magnetizing inductance

L k2:第二漏電感 L k 2 : Second leakage inductance

S 1:第一功率開關 S 1 : the first power switch

S 2:第二功率開關 S 2 : Second power switch

C 1:第一輸出電容 C 1 : the first output capacitor

C 2:第二輸出電容 C 2 : Second output capacitor

C 3:第一舉升電容 C 3 : the first lift capacitor

C 4:第二舉升電容 C 4 : Second lift capacitor

C o:輸出電容 C o : output capacitance

D 1:第一輸出二極體 D 1 : The first output diode

D 2:第二輸出二極體 D 2 : Second output diode

D 3:第一舉升二極體 D 3 : The first lift diode

D 4:第二舉升二極體 D 4 : Second lift diode

D o :輸出二極體 D o : output diode

R o :負載 R o : load

Claims (5)

一種高電壓轉換比直流轉換器,其主要係令轉換器於輸入電壓之正極分別連接第一耦合電感一次側之第一端及第二耦合電感一次側之第一端,於該第一耦合電感一次側之第二端分別連接有第一功率開關之第一端、第一輸出電容之第一端及第二輸出電容之第二端,而該第二耦合電感一次側之第二端分別連接有第二功率開關之第一端及第二輸出二極體之正極,該輸入電壓之負極分別連接該第二功率開關之第二端、該第一功率開關之第二端及第一輸出二極體之負極,該第二輸出二極體之負極分別連接該第二輸出電容之第一端、第一舉升電容之第一端及第二耦合電感二次側之第一端,該第二耦合電感二次側之第二端連接第一耦合電感二次側之第二端,該第一耦合電感二次側之第一端分別連接第二舉升電容之第一端及第一舉升二極體之正極,該第一舉升電容之第二端分別連接該第一舉升二極體之負極及第二舉升二極體之正極,該第二舉升電容之第二端分別連接該第二舉升二極體之負極及輸出二極體之正極,該輸出二極體之負極分別連接輸出電容之第一端及負載之第一端,該第一輸出二極體之正極則分別連接該第一輸出電容之第二端、該輸出電容之第二端及該負載之第二端。A high voltage conversion ratio DC converter, which mainly connects the positive pole of the input voltage to the first terminal of the primary side of the first coupled inductor and the first terminal of the primary side of the second coupled inductor, and connects the first terminal of the primary side of the coupled inductor to the positive pole of the input voltage. The second terminal of the primary side is respectively connected to the first terminal of the first power switch, the first terminal of the first output capacitor and the second terminal of the second output capacitor, and the second terminal of the primary side of the second coupling inductor is respectively connected to There is a first terminal of the second power switch and a positive pole of the second output diode, and the negative pole of the input voltage is connected to the second terminal of the second power switch, the second terminal of the first power switch and the first output diode respectively. The negative pole of the polar body, the negative pole of the second output diode is respectively connected to the first end of the second output capacitor, the first end of the first lifting capacitor and the first end of the second side of the second coupling inductor, the first end of the second output capacitor The second terminal of the secondary side of the second coupled inductor is connected to the second terminal of the secondary side of the first coupled inductor, and the first terminal of the secondary side of the first coupled inductor is respectively connected to the first terminal of the second lifting capacitor and the first lifting capacitor. The positive pole of the lifting diode, the second terminal of the first lifting capacitor is respectively connected to the negative pole of the first lifting diode and the positive pole of the second lifting diode, the second terminal of the second lifting capacitor Connect the negative pole of the second lifting diode and the positive pole of the output diode respectively, the negative pole of the output diode is respectively connected to the first end of the output capacitor and the first end of the load, and the first end of the first output diode The positive electrode is respectively connected to the second end of the first output capacitor, the second end of the output capacitor and the second end of the load. 如請求項1所述高電壓轉換比直流轉換器,其中,該轉換器於該第一耦合電感一次側形成有第一磁化電感。The high voltage conversion ratio DC converter according to claim 1, wherein the converter has a first magnetizing inductance formed on the primary side of the first coupled inductance. 如請求項1所述高電壓轉換比直流轉換器,其中,該轉換器於該第二耦合電感一次側形成有第二磁化電感。The high voltage conversion ratio DC converter as claimed in claim 1, wherein the converter has a second magnetizing inductance formed on the primary side of the second coupled inductance. 如請求項1所述高電壓轉換比直流轉換器,其中,該轉換器於該第一耦合電感一次側之第二端與該第一功率開關之第一端、該第一輸出電容之第一端及該第二輸出電容之第二端之間形成有第一漏電感。The high voltage conversion ratio DC converter according to claim 1, wherein the second terminal of the converter on the primary side of the first coupled inductor is connected to the first terminal of the first power switch and the first terminal of the first output capacitor. A first leakage inductance is formed between the terminal and the second terminal of the second output capacitor. 如請求項1所述高電壓轉換比直流轉換器,其中,該轉換器於該第二耦合電感一次側之第二端與該第二功率開關之第一端及第二輸出二極體之正極之間形成有第二漏電感。The high voltage conversion ratio DC converter as described in claim 1, wherein the second end of the converter on the primary side of the second coupled inductor is connected to the first end of the second power switch and the positive pole of the second output diode A second leakage inductance is formed between them.
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