JP2017046407A - Rotational position detection device, air conditioner, and rotational position detection method - Google Patents

Rotational position detection device, air conditioner, and rotational position detection method Download PDF

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JP2017046407A
JP2017046407A JP2015165756A JP2015165756A JP2017046407A JP 2017046407 A JP2017046407 A JP 2017046407A JP 2015165756 A JP2015165756 A JP 2015165756A JP 2015165756 A JP2015165756 A JP 2015165756A JP 2017046407 A JP2017046407 A JP 2017046407A
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axis
motor
rotational position
position detection
induced voltage
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JP6490540B2 (en
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佐理 前川
Sari Maekawa
佐理 前川
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Toshiba Corp
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F24—HEATING; RANGES; VENTILATING
    • F24F—AIR-CONDITIONING; AIR-HUMIDIFICATION; VENTILATION; USE OF AIR CURRENTS FOR SCREENING
    • F24F11/00—Control or safety arrangements
    • F24F11/89—Arrangement or mounting of control or safety devices

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  • Power Engineering (AREA)
  • Chemical & Material Sciences (AREA)
  • Combustion & Propulsion (AREA)
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  • Control Of Motors That Do Not Use Commutators (AREA)
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Abstract

PROBLEM TO BE SOLVED: To provide a rotational position detection device capable of detecting a rotational position without using a position sensor even in a driving range near a limit in field weakening control.SOLUTION: A rotational position detection device includes a position estimation calculation section for performing estimation calculation of a rotational position of a motor so that a differential value calculated by weighting an induction voltage estimated value of a dc axis estimated by calculating by using a motor rotational speed, a motor current command value, a motor application voltage command value, and a motor equivalent circuit constant and an induction voltage estimated value of a qc axis becomes zero, where the estimation axis in the magnetic flux direction of a permanent magnet disposed in the rotor of the permanent magnet motor is defined as the dc axis while the direction orthogonal to the dc axis as the qc axis.SELECTED DRAWING: Figure 1

Description

本発明の実施形態は、永久磁石モータの回転位置を検出する装置及び方法並びに空気調和機に関する。   Embodiments described herein relate generally to an apparatus and method for detecting a rotational position of a permanent magnet motor, and an air conditioner.

従来、永久磁石同期モータの回転位置を推定する方法としては、例えばモータの速度に比例する誘起電圧，特にｄ軸誘起電圧をモータへの入力電圧と電流より演算し、ｄ軸誘起電圧に基づいて推定する方法が広く用いられている。このとき、実際のｄ軸誘起電圧は求めることができないため、図９に示すように、磁石の磁束方向の推定軸をｄｃ軸とし、ｄｃ軸誘起電圧Ｅdcを用いて推定する。   Conventionally, as a method for estimating the rotational position of a permanent magnet synchronous motor, for example, an induced voltage proportional to the speed of the motor, in particular, a d-axis induced voltage is calculated from an input voltage and a current to the motor, and based on the d-axis induced voltage. The estimation method is widely used. At this time, since the actual d-axis induced voltage cannot be obtained, as shown in FIG. 9, the estimated axis in the magnetic flux direction of the magnet is set to the dc axis, and is estimated using the dc-axis induced voltage Edc.

具体的には、永久磁石の磁束方向をｄ軸方向と定義し、それと直交する方向をｑ軸とした時モータが回転することで発生する逆起電圧Ｅｑはｑ軸方向にのみ発生する。従って、回転子位置推定直交座標をｄｃ−ｑｃ軸と定義し、ｄｃ軸方向の誘起電圧Ｅdcがゼロとなるように回転子位置推定角度を逐次補正する。すると、ｄ軸とｄｃ軸、ｑ軸とｑｃ軸とが一致するようになり、真の回転子位置を推定できる（例えば特許文献１参照）。この場合、ｄｃ軸方向の誘起電圧Ｅdcは直接検出できないため、モータ定数と回転子速度，ｄｃ−ｑｃ軸の電圧・電流値を用いて演算で求める。   Specifically, when the magnetic flux direction of the permanent magnet is defined as the d-axis direction and the direction perpendicular to the permanent magnet is defined as the q-axis, the counter electromotive voltage Eq generated when the motor rotates is generated only in the q-axis direction. Therefore, the rotor position estimation orthogonal coordinate is defined as the dc-qc axis, and the rotor position estimation angle is sequentially corrected so that the induced voltage Edc in the dc axis direction becomes zero. Then, the d-axis and dc-axis and the q-axis and qc-axis coincide with each other, and the true rotor position can be estimated (see, for example, Patent Document 1). In this case, since the induced voltage Edc in the dc axis direction cannot be directly detected, it is obtained by calculation using the motor constant, the rotor speed, and the voltage / current value of the dc-qc axis.

また、ノイズによる影響を回避するために、演算に電流検出値でなく電流指令値を用いる手法も提案されており、モータを高速領域で駆動する際には、ｄ軸電流を通電して磁石磁束を弱める弱め界磁制御が広く用いられている。   In order to avoid the influence of noise, a method of using a current command value instead of a current detection value for calculation has been proposed. When driving a motor in a high speed region, a d-axis current is applied to magnet magnetic flux. Field-weakening control is weakly used.

特開２００３−２５０２９３号公報JP 2003-250293 A

ここで上述のように、モータの真の回転位置方向をｄ軸、そこから９０度遅れた方向をｑ軸とし、ｄ軸の位置をθとする。そして、回転位置の推定方向をｄｃ軸、そこから９０度遅れた方向をｑｃ軸とし、ｄｃ軸の位置をθｃとし、ｄ軸とｄｃ軸の軸誤差を回転位置検出誤差Δθとして（１）式で表す。
Δθ＝θ−θｃ …（１）
誤差Δθがゼロになれば推定軸＝d軸となり、推定軸に基づいた制御が機能する。誤差Δθをゼロに制御するため、様々な位置推定手法が存在する。 Here, as described above, the true rotational position direction of the motor is d-axis, the direction delayed by 90 degrees is q-axis, and the d-axis position is θ. The estimated direction of the rotational position is the dc axis, the direction delayed by 90 degrees is the qc axis, the position of the dc axis is θc, and the axis error between the d axis and the dc axis is the rotational position detection error Δθ. Represented by
Δθ = θ−θc (1)
When the error Δθ becomes zero, the estimated axis becomes d-axis, and the control based on the estimated axis functions. There are various position estimation methods for controlling the error Δθ to zero.

また、弱め界磁制御は、モータの出力電圧Ｖdqが直流電圧を超えないように、d軸電流を通電する制御方法であり、モータ出力電圧を抑制できる。（２）式はｄｑ軸の定常状態における電圧方程式であるが、ｄ軸電流Ｉｄを負方向に増加させ続けることで、ｑ軸電圧Ｖｑが減少し、（３）式で示すｄ軸電圧Ｖｄとｑ軸電圧Ｖｑとの２乗根である出力電圧Ｖｄｑの抑制効果も増加していく。

Figure 2017046407
The field weakening control is a control method in which the d-axis current is applied so that the motor output voltage Vdq does not exceed the DC voltage, and the motor output voltage can be suppressed. The equation (2) is a voltage equation in the steady state of the dq axis, but the q-axis voltage Vq is decreased by continuing to increase the d-axis current Id in the negative direction, and the d-axis voltage Vd represented by the equation (3) The effect of suppressing the output voltage Vdq, which is the square root of the q-axis voltage Vq, also increases.
Figure 2017046407

しかしながら、ｄ軸電流Ｉｄについては（４）式で示すＩｄ＿Limitが限界となり、ｄ軸電流Ｉｄを増やそうとしても出力電圧が抑制できなくなる。
Ｉｄ＿Limit＝−φｆ／Ｌｄ …（４）
一方、ｄ軸電流ＩｄがＩｄ＿Limitよりも小さくなると出力電圧が増加してしまうため、電流制御するための出力電圧が直流電圧Ｖdc以上となって不足する。そのため、この限界点付近では、ｄｑ軸の電流指令値に対して実際のｄｑ軸電流の追従性が悪くなる場合がある。すなわち、指令値≠実電流という状態である。 However, with respect to the d-axis current Id, Id_Limit expressed by the equation (4) becomes a limit, and the output voltage cannot be suppressed even if the d-axis current Id is increased.
Id_Limit = −φ f / Ld (4)
On the other hand, when the d-axis current Id becomes smaller than Id_Limit, the output voltage increases, so that the output voltage for current control becomes not less than the DC voltage Vdc. For this reason, in the vicinity of this limit point, the actual dq-axis current followability may deteriorate with respect to the dq-axis current command value. That is, the command value is not equal to the actual current.

このような状態において、電流指令値を用いたｄｃ軸誘起電圧の演算に基づく回転位置推定では、電流指令値と検出電流に差異が発生し、ｄｃ軸誘起電圧と回転位置の間の近似的な比例関係が崩れる場合がある。すると、正常に回転位置推定できず脱調停止を引き起こすことになる。   In such a state, in the rotational position estimation based on the calculation of the dc axis induced voltage using the current command value, a difference occurs between the current command value and the detected current, and an approximate difference between the dc axis induced voltage and the rotational position is generated. The proportional relationship may break. As a result, the rotational position cannot be estimated normally, and a step-out stop is caused.

そこで、弱め開示制御における限界付近の駆動範囲においても、位置センサを用いることなく回転位置を検出できる回転位置検出装置，及び前記装置を備えた空気調和機並びに回転位置検出方法を提供する。   Therefore, there are provided a rotational position detecting device capable of detecting a rotational position without using a position sensor, an air conditioner equipped with the device, and a rotational position detecting method even in a drive range near the limit in the weak disclosure control.

実施形態の回転位置検出装置は、永久磁石モータの回転子に配置されている永久磁石磁束方向の推定軸をｄｃ軸，このｄｃ軸に直交する方向をｑｃ軸とし、モータ回転速度，モータ電流指令値，モータ印加電圧指令値及びモータ等価回路定数を用いて推定演算したｄｃ軸の誘起電圧推定値と、ｑｃ軸の誘起電圧誤差推定値とをそれぞれ重み付けして演算した差分値がゼロとなるように、モータの回転位置を推定演算する位置推定演算部を備える。   The rotational position detection device of the embodiment uses a dc axis as an estimated axis of a permanent magnet magnetic flux direction arranged on a rotor of a permanent magnet motor, and a qc axis as a direction orthogonal to the dc axis, and a motor rotation speed and a motor current command. The difference value calculated by weighting the induced voltage estimated value of the dc axis estimated using the value, the motor applied voltage command value, and the motor equivalent circuit constant, and the induced voltage error estimated value of the qc axis is zero. In addition, a position estimation calculation unit for estimating and calculating the rotational position of the motor is provided.

一実施形態であり、ベクトル制御部の構成を示す機能ブロック図Functional block diagram showing the configuration of the vector control unit according to one embodiment モータ制御装置の構成を示す機能ブロック図Functional block diagram showing the configuration of the motor controller 空気調和機の構成を示す図The figure which shows the composition of the air conditioner ｄｃ軸誘起電圧Ｅdc，誘起電圧誤差ΔＥqc及び重み付き差分値（ｗｄ・Ｅdc−ｗｑ・ΔＥqc）の波形を示す図The figure which shows the waveform of dc axis | shaft induced voltage Edc, induced voltage error (DELTA) Eqc, and a weighted difference value (wd * Edc-wq * (DELTA) Eqc). 従来の制御において、モータが脱調により停止する直前のｄｃ軸誘起電圧Ｅdc，位置推定誤差Δθ及び推定速度ωcの波形を示す図The figure which shows the waveform of dc axis induced voltage Edc, position estimation error (DELTA) (theta), and estimated speed (omega) c just before a motor stops by a step-out in the conventional control. 回転位置推定部の構成を示す機能ブロック図Functional block diagram showing the configuration of the rotational position estimation unit 回転速度ωに応じて制御ゲインＫｐ,Ｋｉを変化させる状態を示す図The figure which shows the state which changes control gain Kp and Ki according to rotational speed (omega). 本実施形態を適用した場合の図５相当図FIG. 5 equivalent diagram when this embodiment is applied ｄ−ｑ軸及びｄｃ−ｑｃ軸の関係を示すベクトル図Vector diagram showing the relationship between dq axis and dc-qc axis

以下、回転位置検出装置を、空気調和機の圧縮機モータを駆動するモータ制御装置に適用した一実施形態について図１から図８を参照して説明する。図３において、ヒートポンプシステム１を構成する圧縮機（負荷）２は、圧縮部３とモータ４を同一の鉄製密閉容器５内に収容して構成され、モータ４のロータシャフトが圧縮部３に連結されている。そして、圧縮機２、四方弁６、室内側熱交換器７、減圧装置８、室外側熱交換器９は、熱伝達媒体流路たるパイプにより閉ループを構成するように接続されている。尚、圧縮機２は、例えばロータリ型の圧縮機であり、モータ４は、例えば３相ＩＰＭ（Interior Permanent Magnet）モータ（ブラシレスＤＣモータ，永久磁石同期モータ）である。   Hereinafter, an embodiment in which a rotational position detection device is applied to a motor control device that drives a compressor motor of an air conditioner will be described with reference to FIGS. 1 to 8. In FIG. 3, the compressor (load) 2 constituting the heat pump system 1 is configured by accommodating the compression unit 3 and the motor 4 in the same iron hermetic container 5, and the rotor shaft of the motor 4 is connected to the compression unit 3. Has been. The compressor 2, the four-way valve 6, the indoor heat exchanger 7, the pressure reducing device 8, and the outdoor heat exchanger 9 are connected by a pipe serving as a heat transfer medium flow path so as to form a closed loop. The compressor 2 is, for example, a rotary compressor, and the motor 4 is, for example, a three-phase IPM (Interior Permanent Magnet) motor (brushless DC motor, permanent magnet synchronous motor).

空気調和機Ｅは、上記のヒートポンプシステム１を有して構成されている。暖房時には、四方弁６は実線で示す状態にあり、圧縮機２の圧縮部３で圧縮された高温冷媒は、四方弁６から室内側熱交換器７に供給されて凝縮し、その後、減圧装置８で減圧され、低温となって室外側熱交換器９に流れ、ここで蒸発して圧縮機２へと戻る。一方、冷房時には、四方弁６は破線で示す状態に切り替えられる。   The air conditioner E includes the heat pump system 1 described above. At the time of heating, the four-way valve 6 is in a state indicated by a solid line, and the high-temperature refrigerant compressed by the compression unit 3 of the compressor 2 is supplied from the four-way valve 6 to the indoor heat exchanger 7 to be condensed, and then the decompression device. The pressure is reduced at 8 and the temperature becomes low and flows to the outdoor heat exchanger 9 where it evaporates and returns to the compressor 2. On the other hand, at the time of cooling, the four-way valve 6 is switched to a state indicated by a broken line.

このため、圧縮機２の圧縮部３で圧縮された高温冷媒は、四方弁６から室外側熱交換器９に供給されて凝縮し、その後、減圧装置８で減圧され、低温となって室内側熱交換器７に流れ、ここで蒸発して圧縮機２へと戻る。そして、室内側、室外側の各熱交換器７，９には、それぞれファン１０，１１により送風が行われ、その送風によって各熱交換器７，９と室内空気、室外空気の熱交換が効率良く行われるように構成されている。   For this reason, the high-temperature refrigerant | coolant compressed with the compression part 3 of the compressor 2 is supplied to the outdoor side heat exchanger 9 from the four-way valve 6, is condensed, and is decompressed by the decompression device 8, and becomes low temperature indoor side It flows into the heat exchanger 7, where it evaporates and returns to the compressor 2. The indoor and outdoor heat exchangers 7 and 9 are blown by the fans 10 and 11, respectively, and the heat exchange between the heat exchangers 7 and 9 and the indoor air and outdoor air is efficient. It is structured to be performed well.

図２は、モータ制御装置の構成を示す機能ブロック図である。直流電源部２１は、直流電源のシンボルで示しているが、商用交流電源から直流電源を生成している場合には、整流回路や平滑コンデンサ等を含んでいる。直流電源部２１には、正側母線２２ａ，負側母線２２ｂを介してインバータ回路２３が接続されている。インバータ回路２３は、スイッチング素子として例えばＮチャネル型のパワーＭＯＳＦＥＴ２４（Ｕ＋，Ｖ＋，Ｗ＋，Ｕ−，Ｖ−，Ｗ−）を３相ブリッジ接続して構成されており、各相の出力端子はモータ４の各相巻線にそれぞれ接続されている。   FIG. 2 is a functional block diagram showing the configuration of the motor control device. The DC power supply unit 21 is indicated by a DC power supply symbol, but includes a rectifier circuit, a smoothing capacitor, and the like when a DC power supply is generated from a commercial AC power supply. An inverter circuit 23 is connected to the DC power supply unit 21 through a positive bus 22a and a negative bus 22b. The inverter circuit 23 is configured by connecting, for example, N-channel type power MOSFETs 24 (U +, V +, W +, U−, V−, W−) as switching elements in a three-phase bridge, and the output terminals of each phase are motors. Each of the four phase windings is connected.

下側のＦＥＴ２４Ｕ−，２４Ｖ−，２４Ｗ−のソースと負側母線２２ｂとの間には、シャント抵抗（電流検出素子）２５Ｕ，２５Ｖ，２５Ｗが接続されており、シャント抵抗２５の端子電圧は電流検出部２６により検出される。電流検出部２６は、前記端子電圧をＡ／Ｄ変換して読み込み、Ｕ，Ｖ，Ｗ各相の電流Ｉｕ，Ｉｖ，Ｉｗを検出する。電流検出部２６が検出した各相電流は、ベクトル演算部３０に入力される。   Shunt resistors (current detection elements) 25U, 25V, and 25W are connected between the sources of the lower FETs 24U−, 24V−, and 24W− and the negative bus 22b, and the terminal voltage of the shunt resistor 25 is a current. It is detected by the detection unit 26. The current detector 26 reads the terminal voltage after A / D conversion, and detects the currents Iu, Iv, and Iw of the U, V, and W phases. Each phase current detected by the current detection unit 26 is input to the vector calculation unit 30.

図１に示すように、ベクトル制御部３０では、入力される各相電流Ｉｕ，Ｉｖ，Ｉｗが３相／２相変換部４１においてｄｃ軸電流Ｉdc，ｑｃ軸電流Ｉqcに変換される。速度制御部４２には、制御条件を設定するマイクロコンピュータ等の機能部分よりモータ４の回転速度指令ωrefが入力されると共に、回転位置推定部４３により推定されたモータ４の回転速度ωｃが入力されている。速度制御部４２は、回転速度指令ωRefと回転速度ωｃとの差分に基づいてｑｃ軸電流指令Ｉqc＿Refを生成し、電流制御部４４に出力する。   As shown in FIG. 1, in the vector control unit 30, input phase currents Iu, Iv, and Iw are converted into a dc-axis current Idc and a qc-axis current Iqc in a three-phase / two-phase conversion unit 41. The speed controller 42 receives the rotational speed command ωref of the motor 4 from a functional part such as a microcomputer that sets control conditions, and also receives the rotational speed ωc of the motor 4 estimated by the rotational position estimator 43. ing. The speed control unit 42 generates a qc-axis current command Iqc_Ref based on the difference between the rotation speed command ωRef and the rotation speed ωc, and outputs it to the current control unit 44.

電流制御部４４には、上述のマイコンよりｄｃ軸電流指令Ｉdc＿Refが入力されており、３相／２相変換部４１よりｄ軸電流Ｉdc，ｑｃ軸電流Ｉqcが入力されている。また、回転速度指令ωRef，ｑｃ軸電流指令Ｉqc＿Ref及びｄ軸電流指令Ｉdc＿Refは、回転位置推定部４３（位置推定演算部）にも入力されている。電流制御部４４は、ｑｃ軸電流指令Ｉqc＿Refとｑｃ軸電流Ｉqcとの差分値に基づいてｑｃ軸電圧Ｖqcを求め、ｄｃ軸電流指令Ｉdc＿Refとｄｃ軸電流Ｉdcとの差分値に基づいてｄｃ軸電圧Ｖdcを求める。これらは、回転位置推定部４３及び２相／３相変換部４５に入力される。   The current control unit 44 receives the dc-axis current command Idc_Ref from the above-described microcomputer, and receives the d-axis current Idc and the qc-axis current Iqc from the three-phase / two-phase conversion unit 41. The rotational speed command ωRef, the qc-axis current command Iqc_Ref, and the d-axis current command Idc_Ref are also input to the rotational position estimation unit 43 (position estimation calculation unit). The current control unit 44 obtains the qc-axis voltage Vqc based on the difference value between the qc-axis current command Iqc_Ref and the qc-axis current Iqc, and determines the dc-axis voltage based on the difference value between the dc-axis current command Idc_Ref and the dc-axis current Idc. Obtain Vdc. These are input to the rotational position estimation unit 43 and the 2-phase / 3-phase conversion unit 45.

回転位置推定部４３により推定された回転位置θｃは、３相／２相変換部４１及び２相／３相変換部４５に入力されている。２相／３相変換部４５は、入力されるｑｃ軸電圧Ｖqc，ｄｃ軸電圧Ｖdcを３相電圧Ｖｕ，Ｖｖ，Ｖｗに変換して、図２に示すＤＵＴＹ生成部３１に出力する。   The rotational position θc estimated by the rotational position estimation unit 43 is input to the three-phase / 2-phase conversion unit 41 and the two-phase / 3-phase conversion unit 45. The two-phase / three-phase conversion unit 45 converts the input qc-axis voltage Vqc and dc-axis voltage Vdc into three-phase voltages Vu, Vv, and Vw, and outputs them to the DUTY generation unit 31 shown in FIG.

ＤＵＴＹ生成部３１は、各相のＰＷＭ信号を生成するためのデューティＵ＿ＤＵＴＹ，Ｖ＿ＤＵＴＹ，Ｗ＿ＤＵＴＹを決定し、各相デューティＵ，Ｖ，Ｗ＿ＤＵＴＹは、ＰＷＭ信号生成部３２に与えられ、キャリアとのレベルが比較されることで３相ＰＷＭ信号が生成される。また、３相ＰＷＭ信号を反転させた下アーム側の信号も生成されて、必要に応じてデッドタイムが付加された後、それらが駆動回路３３に出力される。駆動回路３３は、与えられたＰＷＭ信号に従い、インバータ回路２３を構成する６つのパワーＭＯＳＦＥＴ２４（Ｕ＋，Ｖ＋，Ｗ＋，Ｕ−，Ｖ−，Ｗ−）の各ゲートに、ゲート信号を出力する（上アーム側については、必要なレベルだけ昇圧した電位で出力する）。尚、以上において、構成２７〜３２，３４の機能は、ＣＰＵを含むマイクロコンピュータのハードウェア及びソフトウェアにより実現される機能である。   The DUTY generation unit 31 determines the duties U_DUTY, V_DUTY, and W_DUTY for generating the PWM signals of the respective phases, and the phase duties U, V, and W_DUTY are given to the PWM signal generation unit 32, and the level with the carrier is determined. A three-phase PWM signal is generated by comparison. Further, a signal on the lower arm side obtained by inverting the three-phase PWM signal is also generated, and after a dead time is added as necessary, they are output to the drive circuit 33. The drive circuit 33 outputs a gate signal to each gate of the six power MOSFETs 24 (U +, V +, W +, U−, V−, W−) constituting the inverter circuit 23 in accordance with the given PWM signal (upper). On the arm side, the voltage is output at a potential boosted by a necessary level). In the above, the functions of the configurations 27 to 32 and 34 are functions realized by hardware and software of a microcomputer including a CPU.

ここで、本実施形態における回転位置検出方法の概要を説明する。位置センサレス制御では、（５）式で検出したｄｃ軸誘起電圧Ｅdcが（６）式及び図４に示すように位置推定誤差Δθに近似的な比例関係を持つ。したがって、ｄｃ軸誘起電圧Ｅdｃを比例積分器で制御することで推定位置θcを実位置θに一致させることができる。ｄｃ軸誘起電圧Ｅdcは、図９で示したように位置推定誤差Δθがある場合に発生し、その大きさは（６）式となる。推定誤差が無い場合、ｄｃ軸誘起電圧Ｅdcはゼロとなり、誘起電圧はｑｃ軸方向のみに発生する。
Ｅdc＝Ｖdc−Ｒ・Ｉdc＿Ref＋ωｃ・Ｌq・Ｉqc＿Ref …（５）
Ｅdc＝ωφｆ（−sinΔθ） …（６） Here, an outline of the rotational position detection method in the present embodiment will be described. In the position sensorless control, the dc axis induced voltage Edc detected by the equation (5) has an approximate proportional relationship to the position estimation error Δθ as shown in the equation (6) and FIG. Therefore, the estimated position θc can be matched with the actual position θ by controlling the dc-axis induced voltage Edc with a proportional integrator. The dc-axis induced voltage Edc occurs when there is a position estimation error Δθ as shown in FIG. 9, and the magnitude thereof is given by equation (6). When there is no estimation error, the dc axis induced voltage Edc is zero, and the induced voltage is generated only in the qc axis direction.
Edc = Vdc−R · Idc_Ref + ωc · Lq · Iqc_Ref (5)
Edc = ωφ f (−sin Δθ) (6)

しかし、高速領域での弱め界磁制御における限界点付近の負荷点では、前述したように一時的に電流指令値に実電流が追従しない場合が存在するため、図４の関係が成立しない場合がある。図５は、モータが脱調により停止する直前のｄｃ軸誘起電圧Ｅdcと、位置推定誤差Δθ及び推定速度ωcを示している。位置推定誤差Δθは、モータに位置センサを取り付け測定した。ある点から位置推定誤差Δθが増加していく一方で、誘起電圧Ｅdcはゼロから変化せず、推定速度ωcが低下して脱調に至っている。誘起電圧Ｅdcが変化しないため、センサレス制御のループが位置推定誤差を減らすように働かず、正確な位置が推定できなくなったことが脱調の原因である。   However, at the load point near the limit point in field-weakening control in the high-speed region, there is a case where the actual current does not temporarily follow the current command value as described above, so the relationship of FIG. FIG. 5 shows the dc-axis induced voltage Edc, the position estimation error Δθ, and the estimated speed ωc immediately before the motor stops due to step-out. The position estimation error Δθ was measured by attaching a position sensor to the motor. While the position estimation error Δθ increases from a certain point, the induced voltage Edc does not change from zero, and the estimated speed ωc decreases, leading to step-out. Since the induced voltage Edc does not change, the sensorless control loop does not work to reduce the position estimation error, and the accurate position cannot be estimated.

これに対し、同図でΔＥqcとして表示しているのはｑｃ軸の誘起電圧誤差であり（７）式で演算している。尚、ΔＥqcの推定誤差に対する特性は（８）式で表され、位置推定誤差Δθに応じて変化するが、その方向は誤差の符合に対し対称となる。
ΔＥqc＝Ｖqc−Ｒ・Ｉqc＿Ref−ωｃ・Ｌｄ・Ｉdc＿Ref …（７）
ΔＥqc＝ωφｆ（１−cosΔθ） …（８） On the other hand, what is indicated as ΔEqc in the figure is an induced voltage error on the qc axis, which is calculated by equation (7). The characteristic of ΔEqc with respect to the estimation error is expressed by equation (8) and changes according to the position estimation error Δθ, but the direction is symmetric with respect to the sign of the error.
ΔEqc = Vqc−R · Iqc_Ref−ωc · Ld · Idc_Ref (7)
ΔEqc = ωφ f (1−cos Δθ) (8)

図５に示すように、誘起電圧誤差ΔＥqcは弱め界磁制御の限界付近においても、位置推定誤差Δθの増加に伴い変化して行くことが分かる。つまり、誘起電圧誤差ΔＥqcをセンサレス制御のアルゴリズムに組み込むことで、トルク限界付近のセンサレス制御性能を改善できる。   As shown in FIG. 5, it can be seen that the induced voltage error ΔEqc changes as the position estimation error Δθ increases even near the limit of field weakening control. That is, the sensorless control performance near the torque limit can be improved by incorporating the induced voltage error ΔEqc into the sensorless control algorithm.

図６は，誘起電圧誤差ΔＥqcとｄｃ軸誘起電圧Ｅdcとを用いたセンサレス制御部，すなわち回転位置推定部４３の構成である。回転位置推定部４３の外部にある減算器５１において、ｄ軸の位置θと推定位置θｃとの差分がとられ、（６）式が演算されてｄ軸誘起電圧Ｅdcが得られ、（８）式が演算されて誘起電圧誤差ΔＥqcが得られる。   FIG. 6 shows a configuration of a sensorless control unit using the induced voltage error ΔEqc and the dc-axis induced voltage Edc, that is, the rotational position estimating unit 43. In the subtractor 51 outside the rotational position estimation unit 43, the difference between the d-axis position θ and the estimated position θc is calculated, and the equation (6) is calculated to obtain the d-axis induced voltage Edc, (8) The induced voltage error ΔEqc is obtained by calculating the equation.

ｄ軸誘起電圧Ｅdcには、増幅器５２により重み値（ゲイン）ｗｄが付与され、誘起電圧誤差ΔＥqcには、増幅器５３により重み値ｗｑが付与される。そして、減算器５４により両者の差である重み付き差分値（ｗｄ・Ｅdc−ｗｑ・ΔＥqc）が演算される（図４参照）。重み付き差分値には、増幅器５５，５６により比例制御ゲインＫｐ，積分制御ゲインＫｉが付与される。増幅器５６の出力は積分器５７により積分され、加算器５８により増幅器５５の出力と加算される。加算器５８の出力は、速度推定誤差Δωとなる。   The d-axis induced voltage Edc is given a weight value (gain) wd by the amplifier 52, and the induced voltage error ΔEqc is given a weight value wq by the amplifier 53. Then, a subtractor 54 calculates a weighted difference value (wd · Edc−wq · ΔEqc) that is the difference between the two (see FIG. 4). A proportional control gain Kp and an integral control gain Ki are given to the weighted difference values by the amplifiers 55 and 56. The output of the amplifier 56 is integrated by the integrator 57 and added by the adder 58 with the output of the amplifier 55. The output of the adder 58 is a speed estimation error Δω.

次段の減算器５９により、回転速度指令値ωRefと速度推定誤差Δωとの差分がとられて推定速度ωｃが得られ、推定速度ωｃを積分器６０により積分して推定位置θｃが得られる。すなわち、センサレス制御のＰＩフィードバックループに入力する値を、ｄｃ軸の誘起電圧Ｅdcとｑｃ軸の誘起電圧誤差ΔＥqcの重み付き差分値としている点が特徴である。ｑｃ軸側の重み値ｗｑ，ｄｃ軸側の重み値ｗｄは、動作が安定するように任意の値にして良い。また、（ｗｄ＞ｗｑ）に設定する（例えばｗｄ＝１，ｗｑ＝０．５など）。これは主としてｄｃ軸側を用いて推定し、ｑｃ軸を補助的に用いるためである。さらに、制御ゲインＫｐ,Ｋｉは、回転速度ωに応じて図７のように可変する構成とする。   The subtractor 59 in the next stage takes the difference between the rotational speed command value ωRef and the speed estimation error Δω to obtain the estimated speed ωc, and the estimated speed ωc is integrated by the integrator 60 to obtain the estimated position θc. In other words, the value input to the PI feedback loop of sensorless control is a weighted difference value between the induced voltage Edc on the dc axis and the induced voltage error ΔEqc on the qc axis. The weight value wq on the qc axis side and the weight value wd on the dc axis side may be set to arbitrary values so that the operation is stabilized. Further, (wd> wq) is set (for example, wd = 1, wq = 0.5, etc.). This is mainly because the estimation is performed using the dc axis side and the qc axis is used as an auxiliary. Further, the control gains Kp and Ki are configured to vary as shown in FIG. 7 according to the rotational speed ω.

誘起電圧誤差ΔＥqcを組み入れた効果により、電流指令値に対する実電流の追従性が悪化する弱め界磁限界付近で誘起電圧Ｅdcに基づくセンサレス制御が機能しない場合においても、安定したセンサレス制御が可能となる。図８は実際の空気調和機において、弱め界磁電流Ｉｄを（４）式で示す限界値まで通電して駆動した場合の波形である。電流Ｉｄは限界値に達しているがセンサレス制御が維持できており、その後回転速度指令値が低下していくまで安定したモータの駆動状態が継続している。回転速度の表示は、低下した後また上から表示されているが、これは計器の表示にアンダーフローが起きているためで、実際は低下し続けている。   Due to the effect of incorporating the induced voltage error ΔEqc, stable sensorless control is possible even when sensorless control based on the induced voltage Edc does not function near the field-weakening limit where the followability of the actual current with respect to the current command value deteriorates. . FIG. 8 shows waveforms when the field weakening current Id is energized to the limit value shown by the equation (4) in an actual air conditioner. Although the current Id has reached the limit value, the sensorless control can be maintained, and then the stable motor driving state continues until the rotational speed command value decreases. The display of the rotation speed is displayed from the top again after the decrease, but this is due to the underflow occurring in the display of the instrument, and the actual decrease continues.

以上のように本実施形態によれば、回転位置推定部４３は、モータ４の回転子に配置されている永久磁石磁束方向の推定軸をｄｃ軸，ｄｃ軸に直交する方向をｑｃ軸とし、モータ回転速度ω，モータ電流指令値ＩRef，モータ印加電圧指令値Ｖqc，Ｖdc及びモータ等価回路定数Ｒ，Ｌを用いて推定演算したｄｃ軸の誘起電圧推定値Ｅdcと、ｑｃ軸の誘起電圧推定値ΔＥqcとをそれぞれ重み付けして演算した差分値（ｗｄ・Ｅdc−ｗｑ・ΔＥqc）がゼロとなるように、モータ４の回転位置を推定演算する。   As described above, according to the present embodiment, the rotational position estimating unit 43 sets the estimated axis of the permanent magnet magnetic flux direction arranged in the rotor of the motor 4 as the dc axis and the direction orthogonal to the dc axis as the qc axis, Estimated dc-axis estimated voltage Edc and qc-axis estimated voltage estimated using motor rotation speed ω, motor current command value IRef, motor applied voltage command values Vqc and Vdc, and motor equivalent circuit constants R and L The rotational position of the motor 4 is estimated and calculated so that the difference value (wd · Edc−wq · ΔEqc) calculated by weighting ΔEqc becomes zero.

これにより、弱め界磁制御における限界付近の駆動範囲においても、回転位置を推定できる。したがって、モータ制御装置１５は、正弦波状の電流を通電しながら、モータ４の駆動制御を安定して行うことができる。また、回転位置推定部４３は、ｄｃ軸の誘起電圧推定値Ｅdcの重みｗｄを、ｑｃ軸の誘起電圧推定値ΔＥqcの重みよりも大きく設定する。これにより、主としてｄｃ軸側を用いて推定を行う制御形態に適合した重み値を付与できる。   Thus, the rotational position can be estimated even in the driving range near the limit in field weakening control. Therefore, the motor control device 15 can stably perform drive control of the motor 4 while energizing a sine wave current. Further, the rotational position estimating unit 43 sets the weight wd of the induced voltage estimated value Edc on the dc axis to be larger than the weight of the estimated voltage estimated value ΔEqc on the qc axis. Thereby, the weight value suitable for the control form which estimates mainly using the dc axis | shaft side can be provided.

更に、回転位置推定部４３は、前記差分値に基づくモータ４の回転位置の推定演算を、比例・積分演算により行うので、回転位置の変動に対して適切に追従するように位置推定を行うことができる。加えて、比例・積分演算における制御ゲインＫｐ,Ｋｉを回転速度ωに基づき変化させるので、回転速度ωに適した制御ゲインＫｐ,Ｋｉを付与して制御を安定させることができる。   Furthermore, since the rotational position estimation unit 43 performs the estimation calculation of the rotational position of the motor 4 based on the difference value by proportional / integral calculation, the rotational position estimation unit 43 performs position estimation so as to appropriately follow the fluctuation of the rotational position. Can do. In addition, since the control gains Kp and Ki in the proportional / integral calculation are changed based on the rotational speed ω, the control gains Kp and Ki suitable for the rotational speed ω can be applied to stabilize the control.

（その他の実施形態）
制御ゲインＫｐ,Ｋｉを固定値としても良い。
重み値については、必ずしも（ｗｄ＞ｗｑ）に設定する必要はなく、個別の制御形態に応じて適宜変更すれば良い。
スイッチング素子はＭＯＳＦＥＴ以外にＩＧＢＴやパワートランジスタさらにはＳｉＣ,ＧａＮ等のワイドギャップ半導体等を使用しても良い。
空気調和機以外のモータ制御装置についても適用可能である。 (Other embodiments)
The control gains Kp and Ki may be fixed values.
The weight value is not necessarily set to (wd> wq), and may be appropriately changed according to the individual control mode.
As the switching element, an IGBT, a power transistor, or a wide gap semiconductor such as SiC or GaN may be used in addition to the MOSFET.
The present invention is also applicable to motor control devices other than air conditioners.

本発明のいくつかの実施形態を説明したが、これらの実施形態は例として提示したものであり、発明の範囲を限定することは意図していない。これら新規な実施形態は、その他の様々な形態で実施されることが可能であり、発明の要旨を逸脱しない範囲で種々の省略、置き換え、変更を行うことができる。これらの実施形態やその変形は、発明の範囲や要旨に含まれると共に、特許請求の範囲に記載された発明とその均等の範囲に含まれる。   Although several embodiments of the present invention have been described, these embodiments have been presented by way of example and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.

図面中、４はモータ（永久磁石モータ）、２３はインバータ回路、３０はベクトル演算部、４３は回転位置推定部（位置推定演算部）、Ｅは空気調和機を示す。   In the drawings, 4 is a motor (permanent magnet motor), 23 is an inverter circuit, 30 is a vector calculation unit, 43 is a rotational position estimation unit (position estimation calculation unit), and E is an air conditioner.

Claims (9)

永久磁石モータの回転子に配置されている永久磁石磁束方向の推定軸をｄｃ軸，このｄｃ軸に直交する方向をｑｃ軸とし、
モータ回転速度，モータ電流指令値，モータ印加電圧指令値及びモータ等価回路定数を用いて推定演算したｄｃ軸の誘起電圧推定値と、ｑｃ軸の誘起電圧誤差推定値とをそれぞれ重み付けして演算した差分値がゼロとなるように、モータの回転位置を推定演算する位置推定演算部を備える回転位置検出装置。 The estimated axis of the direction of the permanent magnet magnetic flux arranged in the rotor of the permanent magnet motor is defined as the dc axis, and the direction orthogonal to the dc axis is defined as the qc axis.
The dc-axis induced voltage estimated value and the qc-axis induced voltage error estimated value estimated and calculated using the motor rotation speed, motor current command value, motor applied voltage command value, and motor equivalent circuit constant were respectively weighted and calculated. A rotational position detection device including a position estimation calculation unit that estimates and calculates a rotational position of a motor so that a difference value becomes zero. 前記位置推定演算部は、前記ｄｃ軸の誘起電圧推定値の重みを、前記ｑｃ軸の誘起電圧誤差推定値の重みよりも大きく設定する請求項１記載の回転位置検出装置。   The rotational position detection device according to claim 1, wherein the position estimation calculation unit sets a weight of the induced voltage estimated value of the dc axis to be larger than a weight of the induced voltage error estimated value of the qc axis. 前記位置推定演算部は、前記差分値に基づくモータの回転位置の推定演算を、比例・積分演算により行う請求項１又は２記載の回転位置検出装置。   The rotational position detection device according to claim 1, wherein the position estimation calculation unit performs estimation calculation of the rotational position of the motor based on the difference value by proportional / integral calculation. 前記位置推定演算部は、前記比例・積分演算における制御ゲインを、モータ回転速度に基づき変化させる請求項３記載の回転位置検出装置。   The rotational position detection device according to claim 3, wherein the position estimation calculation unit changes a control gain in the proportional / integral calculation based on a motor rotation speed. 請求項１から４の何れか一項に記載の回転位置検出装置を備え、前記回転位置に基づき永久磁石モータの電流指令値を生成して運転する空気調和機。   An air conditioner comprising the rotational position detection device according to any one of claims 1 to 4 and operating by generating a current command value of a permanent magnet motor based on the rotational position. 永久磁石モータの回転子に配置されている永久磁石磁束方向の推定軸をｄｃ軸，このｄｃ軸に直交する方向をｑｃ軸とし、
モータ回転速度，モータ電流指令値，モータ印加電圧指令値及びモータ等価回路定数を用いて推定演算したｄｃ軸の誘起電圧推定値と、ｑｃ軸の誘起電圧誤差推定値とをそれぞれ重み付けして演算した差分値がゼロとなるように、モータの回転位置を推定演算する回転位置検出方法。 The estimated axis of the direction of the permanent magnet magnetic flux arranged in the rotor of the permanent magnet motor is defined as the dc axis, and the direction orthogonal to the dc axis is defined as the qc axis.
The dc-axis induced voltage estimated value and the qc-axis induced voltage error estimated value estimated and calculated using the motor rotation speed, motor current command value, motor applied voltage command value, and motor equivalent circuit constant were respectively weighted and calculated. A rotational position detection method for estimating and calculating a rotational position of a motor so that a difference value becomes zero. 前記ｄｃ軸の誘起電圧推定値の重みを、前記ｑｃ軸の誘起電圧誤差推定値の重みよりも大きく設定する請求項６記載の回転位置検出方法。   The rotational position detection method according to claim 6, wherein a weight of the induced voltage estimated value of the dc axis is set larger than a weight of the induced voltage error estimated value of the qc axis. 前記差分値に基づくモータの回転位置の推定演算を、比例・積分演算により行う請求項６又は７記載の回転位置検出方法。   The rotational position detection method according to claim 6 or 7, wherein the estimation calculation of the rotational position of the motor based on the difference value is performed by proportional / integral calculation. 前記比例・積分演算における制御ゲインを、モータ回転速度に基づき変化させる請求項８記載の回転位置検出方法。   The rotational position detection method according to claim 8, wherein a control gain in the proportional / integral calculation is changed based on a motor rotational speed.
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