WO2023098826A1 - 谐振型双有源桥式变换电路的控制方法、控制器及变换器 - Google Patents

谐振型双有源桥式变换电路的控制方法、控制器及变换器 Download PDF

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WO2023098826A1
WO2023098826A1 PCT/CN2022/135959 CN2022135959W WO2023098826A1 WO 2023098826 A1 WO2023098826 A1 WO 2023098826A1 CN 2022135959 W CN2022135959 W CN 2022135959W WO 2023098826 A1 WO2023098826 A1 WO 2023098826A1
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Prior art keywords
resonant
bridge
voltage
circuit
phase shift
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PCT/CN2022/135959
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English (en)
French (fr)
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翁炳文
赵一
吴俊雄
廖亚锋
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杭州禾迈电力电子股份有限公司
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Priority to AU2022399746A priority Critical patent/AU2022399746A1/en
Publication of WO2023098826A1 publication Critical patent/WO2023098826A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/38—Means for preventing simultaneous conduction of switches
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention belongs to the technical field of power electronics, and in particular relates to a control method, a controller and a converter of a resonant double active bridge conversion circuit.
  • DC/DC converters have attracted more and more attention.
  • dual active bridge DC/DC converter which can realize the electrical isolation of the primary and secondary sides through the transformer, is a research hotspot in the DC/DC converter.
  • the current DAB (Dual Active Bridge, dual active bridge) conversion circuit has some defects, such as the small voltage gain range of the circuit, and the soft switching failure of the switching device under certain working conditions.
  • an existing improvement scheme is to add an LC resonant circuit to the DAB conversion circuit, and at the same time, for the DAB conversion circuit with a resonant network, either frequency control, or phase shift control, or according to the input and output voltage
  • the size and power are divided into different working intervals, and each interval preferably uses phase shift control or frequency control.
  • the DAB conversion circuit using the above control can obtain a wider working range and/or higher efficiency, but the calculation amount is relatively large and the control is complicated.
  • a control method, device, and converter for a resonant dual active bridge conversion circuit are provided to solve the problem of using frequency conversion and phase shifting to adjust the resonant dual active bridge conversion circuit in the prior art Computationally heavy and complex problems to control.
  • the embodiment of the present application proposes a control method for a resonant dual active bridge conversion circuit.
  • the resonant dual active bridge conversion circuit includes a primary side circuit, at least one secondary side circuit, a resonant cavity, and a transformer.
  • the AC side of the primary side circuit is electrically connected to the primary side of the transformer through a resonant cavity
  • the resonant cavity includes a resonant capacitor and a resonant inductance connected in series
  • the AC side of the secondary side circuit is electrically connected to the secondary side of the transformer, so Said method comprises the following steps:
  • a control signal is generated according to the switching frequency, the phase shift angle in the bridge at the resonant frequency, and the phase angle difference to control the switching on and off of the switching tubes in the primary side circuit and the secondary side circuit.
  • phase angle difference is used to calculate the phase shift angle in the bridge of the primary circuit at the resonant frequency, and the phase shift angle between the primary circuit and the secondary circuit and the phase shift angle in the bridge are at the resonant frequency.
  • the voltage and current in the relational expression of the resonant inductor current and the resonant capacitor voltage are standardized to obtain the relationship between the resonant inductor current and the resonant capacitor voltage Mode;
  • the relationship between the resonant inductor current and the resonant capacitor voltage and the functional relationship between the resonant inductor current, the resonant capacitor voltage and the voltage gain are geometrically graphed, and the phase shift angle and phase shift angle in the bridge at the resonant frequency are obtained according to the geometric relationship. Angle difference.
  • phase shift angle ⁇ 1 and the phase angle difference ⁇ 2 in the bridge under the resonant frequency are:
  • the value of H3 is determined by the dead time, which is a known quantity, M is the voltage gain, and Mc is the per unit value of the peak voltage of the resonant capacitor with the positive value of the midpoint voltage of the primary side bridge arm as the reference value.
  • phase shift angle ⁇ 1 and the phase angle difference ⁇ 2 in the bridge under the resonant frequency are:
  • the value of H3 is determined by the dead time, which is a known quantity, M is the voltage gain, and Mc is the per unit value of the peak voltage of the resonant capacitor with the positive value of the midpoint voltage of the primary side bridge arm as the reference value.
  • the step of generating a control signal according to the switching frequency, the phase angle in the bridge under the resonant frequency and the phase angle difference includes: according to the switching frequency and the phase angle ⁇ 1 and the phase angle difference in the bridge under the resonant frequency ⁇ 2 calculates the phase shift angle ⁇ p in the bridge and the phase shift angle ⁇ ps between the bridges at the switching frequency, and the phase shift angle ⁇ p in the bridge and the phase shift angle ⁇ ps between the bridges at the switching frequency are respectively :
  • f r is the resonant frequency of the resonant cavity
  • f s is the switching frequency of the switching tube.
  • step of calculating the switching frequency of the resonant dual active bridge conversion circuit according to the secondary current includes:
  • the secondary current is compared with the reference current to obtain an error amplification value, and the switching frequency is calculated according to the error amplification value.
  • the primary side circuit includes a first bridge arm and a second bridge arm, the upper switch in the first bridge arm is different from the conduction signal of the lower switch in the second bridge arm by an in-bridge phase shift angle, and the second The conduction signals of the lower switching transistor in the first bridge arm and the upper switching transistor in the second bridge arm are different by an in-bridge phase shift angle.
  • the secondary side circuit includes a third bridge arm and a fourth bridge arm, the upper switch tube in the third bridge arm and the lower switch tube in the fourth bridge arm switch simultaneously, and the switch tube in the third bridge arm
  • the lower switching tube of the first bridge arm and the upper switching tube of the fourth bridge arm switch simultaneously, and the conduction signal of the upper switching tube of the first bridge arm is different from that of the upper switching tube of the third bridge arm by an inter-bridge phase shift angle.
  • the embodiment of the present application also provides a controller, which is used to control the resonant dual active bridge conversion circuit, and the resonant dual active bridge conversion circuit includes a primary circuit, at least one secondary circuit, A resonant cavity and a transformer, the AC side of the primary circuit is electrically connected to the primary side of the transformer through the resonant cavity, the resonant cavity includes a series resonant capacitor and a resonant inductance, the AC side of the secondary circuit is connected to the secondary circuit of the transformer connection, the controller includes:
  • the sampling unit samples the voltage on the DC side of the primary circuit, the voltage on the DC side of the secondary circuit, and the current on the secondary side;
  • the voltage gain calculation unit calculates the voltage gain according to the voltage on the DC side of the primary circuit and the voltage on the DC side of the secondary circuit;
  • the phase shift angle calculation unit uses the voltage gain to calculate the phase shift angle in the bridge of the primary circuit at the resonance frequency, and the phase angle difference between the phase shift angle between the bridges and the phase shift angle in the bridge at the resonance frequency at the switching frequency;
  • a switching frequency calculation unit which calculates the switching frequency of the resonant dual active bridge conversion circuit according to the secondary side current
  • the PWM generation unit generates a control signal according to the phase shift angle within the bridge and the phase shift angle between the bridges at the switching frequency and the resonant frequency, and the control signal is used to control the switching on and off of the switching tubes in the primary circuit and the secondary circuit.
  • the embodiment of the present application further provides a converter, the converter comprising:
  • a resonant type dual active bridge conversion circuit includes a primary circuit, at least one secondary circuit, a resonant cavity, and a transformer with at least one winding on the primary and secondary sides respectively, the primary side
  • the AC side of the circuit is electrically connected to the primary winding of the transformer through a resonant cavity.
  • the resonant cavity includes a series resonant capacitor and a resonant inductance.
  • the secondary circuit corresponds to the secondary winding of the transformer one by one.
  • the AC side of each of the secondary circuits is electrically connected to the corresponding secondary winding;
  • a controller the controller executes the control method of the resonant dual active bridge conversion circuit provided in the present application.
  • the resonant double active bridge conversion circuit of the present invention adopts the phase-shift control and frequency control combined control mode, adopts the time domain analysis method to obtain the relational expression of the resonant capacitor voltage and the resonant inductor current, and converts the obtained
  • the relationship between the resonant capacitor voltage and the resonant inductor current is geometrically graphed, and the phase angle between the phase shift angle in the bridge and the phase shift angle between the bridge phase angle and the phase shift angle in the bridge is obtained at the resonant frequency that is only related to the voltage gain difference, according to the phase shift angle and phase angle difference in the bridge at the resonant frequency, calculate the phase shift angle in the bridge and the phase shift angle between the bridges at the switching frequency, according to the phase shift angle in the bridge, the phase shift angle between the bridge and the calculated
  • the switching frequency controls the conversion circuit, so that each switch in the circuit realizes ZVS, improves work efficiency, and at the same time has a wide range of voltage gain and a high degree of
  • a converter including multiple secondary circuits and the DC sides of the secondary circuits are connected in parallel, which can prevent the problem of excessive current stress of the switching device on the low-voltage side (secondary side) in high power situations.
  • Fig. 1 is the converter schematic diagram that the application provides
  • Fig. 2 is the schematic diagram of the controller provided by the present application.
  • Fig. 3 is the equivalent circuit schematic diagram of the resonant dual active bridge conversion circuit provided by the present application.
  • Fig. 4 is a schematic diagram of the control signal of the switching tube and the voltage and current waveform of the converter in one cycle when the voltage gain M ⁇ 1 provided by the present application;
  • FIG. 5 is a schematic diagram of the relationship between the resonant capacitor voltage u CN (t) and the resonant inductor current i LN (t) in different time periods when the voltage gain M ⁇ 1 provided by the present application;
  • Fig. 6 is a schematic structural diagram of a resonant dual active bridge conversion circuit with two windings on the primary and secondary sides of the transformer;
  • Fig. 7 is a schematic structural diagram of a resonant dual active bridge conversion circuit with multiple windings on the primary and secondary sides of the transformer.
  • FIG. 1 it is a schematic diagram of the converter provided in this application.
  • the converter includes a resonant double active bridge conversion circuit and a controller.
  • the resonant double active bridge conversion circuit includes a primary side circuit, a secondary side circuit, a transformer and a resonant cavity. The working state of the resonant double active bridge conversion circuit is controlled by the controller.
  • the DC side of the primary circuit is electrically connected to the input/output source U 1
  • the AC side of the primary circuit is electrically connected to the primary winding of the transformer through the resonant cavity.
  • the AC side of the secondary circuit is electrically connected to the secondary winding of the transformer, and the DC side of the secondary circuit is electrically connected to the input/output source U 2 .
  • the primary side circuit includes a first full bridge and a voltage stabilizing capacitor C 1
  • the first full bridge includes a first bridge arm and a second bridge arm.
  • the first bridge arm includes the second switch tube Q2 and the fourth switch tube Q4
  • the connection point of the second switch tube Q2 and the fourth switch tube Q4 is taken as the midpoint of the first bridge arm
  • the first bridge arm The midpoint of is used as the first end of the AC side of the primary circuit.
  • the second switching tube Q2 is used as the upper switching tube of the first bridge arm
  • the fourth switching tube Q4 is used as the lower switching tube of the first bridge arm.
  • the second bridge arm includes the first switching tube Q1 and the third switching tube Q3 connected in series, the connection point of the first switching tube Q1 and the third switching tube Q3 is taken as the midpoint of the second bridge arm, and the second bridge arm
  • the midpoint of is used as the second end of the AC side of the primary circuit.
  • the first switching tube Q1 is used as the upper switching tube of the second bridge arm
  • the third switching tube Q3 is used as the lower switching tube of the second bridge arm
  • the midpoint of the second bridge arm is provided between the midpoint of the first bridge arm.
  • the midpoint voltage of the primary bridge arm is Up.
  • the first switching tube Q1 to the fourth switching tube Q4 can be MOS tubes or IGBTs. Specifically, the MOS tubes in this application are described as examples.
  • the drain of the first switching tube Q1 and the second switching tube Q2 The drain is connected and used as the positive terminal of the DC side of the primary circuit, and is electrically connected to the positive terminal of the input/output source U 1 .
  • the source of the first switching tube Q1 is connected to the drain of the third switching tube Q3 , the connection point is the midpoint of the second bridge arm, the source of the second switching tube Q2 is connected to the drain of the fourth switching tube Q4 The poles are connected, and the connection point is used as the midpoint of the first bridge arm.
  • the source of the third switching tube Q3 and the source of the fourth switching tube Q4 serve as the negative terminal of the DC side of the primary circuit, and are electrically connected to the negative terminal of the input/output source U1 .
  • the gates of the first switching transistor Q1 to the fourth switching transistor Q4 are respectively connected to the controller, and are turned on and off according to a control signal provided by the controller.
  • the voltage stabilizing capacitor C1 is connected in parallel with the first bridge arm, and is used for stabilizing the voltage of the DC side of the primary circuit.
  • the resonant cavity includes a resonant capacitor Cr and a resonant inductance Lr connected in series.
  • One end of the resonant capacitor Cr is connected to the midpoint of the second bridge arm, the other end of the resonant capacitor Cr is connected to one end of the resonant inductance Lr, and the other end of the resonant inductance Lr is electrically connected to one end of the primary winding of the transformer.
  • the other end of the side winding is electrically connected to the midpoint of the first bridge arm.
  • the secondary side circuit includes a second full bridge and a voltage stabilizing capacitor C 2
  • the second full bridge includes a third bridge arm and a fourth bridge arm connected in parallel
  • the third bridge arm includes a fifth switching tube S 1 and the sixth switching tube S 3
  • the fourth bridge arm includes the seventh switching tube S 2 and the eighth switching tube S 4 connected in series
  • the connection point between the fifth switching tube S 1 and the sixth switching tube S 3 serves as the third bridge
  • the midpoint of the arm, the connection point of the seventh switching tube S2 and the eighth switching tube S4 serves as the midpoint of the fourth bridge arm
  • the midpoint of the third bridge arm serves as the first end of the AC side of the secondary circuit
  • the fourth The midpoint of the bridge arm serves as the second end of the AC side of the secondary circuit.
  • the fifth switching tube S1 is used as the upper switching tube of the third bridge arm
  • the sixth switching tube S3 is used as the lower switching tube of the third bridge arm
  • the seventh switching tube S2 is used as the upper switching tube of the fourth bridge arm
  • the eighth switching tube S3 is used as the upper switching tube of the fourth bridge arm.
  • the switching tube S4 is used as the lower switching tube of the fourth bridge arm
  • the midpoint voltage Us of the secondary bridge arm is provided between the midpoint of the third bridge arm and the midpoint of the fourth bridge arm.
  • the fifth switching tube S1 to the eighth switching tube S4 are MOS tubes or IGBTs.
  • the MOS tubes in this application are described as examples.
  • the drain of the fifth switching tube S1 is connected to the drain of the seventh switching tube S2 and serves as a secondary
  • the positive end of the DC side of the side circuit is electrically connected to the positive pole of the input/output source U 2 .
  • the source of the fifth switching transistor S1 is connected to the drain of the sixth switching transistor S3 , and the connection point is used as the midpoint of the third bridge arm.
  • the source of the sixth switching tube S3 is connected to the source of the eighth switching tube S3 and serves as the negative terminal of the DC side of the secondary circuit of the third bridge arm, and is electrically connected to the negative terminal of the input/output source U2 .
  • the gates of the fifth switching transistor S1 to the eighth switching transistor S4 are respectively connected to the controller, and are turned on and off according to a control signal provided by the controller.
  • the voltage stabilizing capacitor C2 is connected in parallel with the third bridge arm, and is used to stabilize the voltage of the DC side of the secondary circuit.
  • the first end of the AC side of the secondary circuit is electrically connected to one end of the secondary winding of the transformer, and the second end of the AC side of the secondary circuit is electrically connected to the other end of the secondary winding of the transformer.
  • a capacitor may be connected in series between the AC side of the secondary circuit and the secondary winding of the transformer.
  • the controller controls the first switching tube Q 1 to the fourth switching tube Q 4 , the fifth switching tube S 1 to the eighth switching tube S 4 according to the voltage on the DC side of the primary side circuit, the voltage on the DC side of the secondary side circuit, and the secondary side current.
  • Perform phase shift and frequency modulation control generate control signals for controlling the first switching tube Q1 to the fourth switching tube Q4 , the fifth switching tube S1 to the eighth switching tube S4 , and control the first switching tube Q1 to the fourth switching tube
  • the controller samples the voltage on the DC side of the primary circuit and the voltage on the DC side of the secondary circuit, calculates the voltage gain according to the voltage on the DC side of the primary circuit and the voltage on the DC side of the secondary circuit, and uses the voltage gain Calculate the phase shift angle ⁇ 1 within the bridge of the primary side circuit at the resonant frequency, and the phase angle difference ⁇ 2 between the phase shift angle ⁇ ps between the bridges at the switching frequency and the phase shift angle ⁇ p within the bridge at the resonant frequency, Sampling the secondary current of the secondary circuit, calculating the switching frequency according to the secondary current, and calculating the inter-bridge shift under the switching frequency according to the switching frequency and the internal phase shift angle ⁇ 1 and the phase angle difference ⁇ 2 Phase angle ⁇ ps and bridge internal phase shift angle ⁇ p , according to the bridge internal phase shift angle ⁇ p , inter-bridge phase shift angle ⁇ ps and switching frequency to generate a control signal to control the first switching tube Q 1 to the fourth switching tube
  • the first switching tube Q1 and the third switching tube Q3 of the first full bridge are complementary conduction, and the second switching tube Q2 and the The four switching transistors Q 4 are turned on in a complementary manner.
  • a dead zone is set between the complementary switching tubes.
  • the conduction signals of the first switching transistor Q 1 and the fourth switching transistor Q 4 differ by an in-bridge phase shift angle ⁇ p .
  • the turn-on signals of the second switching tube Q2 and the third switching tube Q3 also differ by an in-bridge phase shift angle ⁇ p .
  • the fifth switching transistor S1 and the seventh switching transistor S3 are turned on in a complementary manner, and the sixth switching transistor S2 and the eighth switching transistor S4 are turned on in a complementary manner. And the fifth switching tube S1 and the eighth switching tube S4 are turned on and off at the same time, and the sixth switching tube S2 and the seventh switching tube S3 are turned on and off at the same time.
  • a dead zone is set between the complementary conduction switch tubes.
  • the turn-on signal of the first switch tube Q1 and the fourth switch tube Q4 of the first bridge arm differ by an intra-bridge phase shift angle ⁇ p
  • the turn-on signal of the first switch tube Q1 of the first full bridge is different from that of the first switch tube Q4 .
  • the turn-on signal of the fifth switching tube S 1 of the two full bridges differs by an inter-bridge phase shift angle ⁇ ps , so at the switching frequency, the phase difference between the midpoint voltage Up of the primary bridge arm and the midpoint voltage Us of the secondary bridge arm is by the phase angle ⁇ ps - ⁇ p .
  • the power flow direction can be controlled by controlling the inter-bridge phase shift angle ⁇ ps and the intra-bridge phase shift angle ⁇ p .
  • the power can be transferred from the primary circuit to the secondary circuit, or from the secondary circuit to the primary circuit.
  • Fig. 2 shows the schematic block diagram of the controller of the embodiment of the present invention, as shown in Fig. 2, described controller comprises:
  • the sampling unit samples the voltage on the DC side of the primary circuit, the voltage on the DC side of the secondary circuit, and the current on the secondary side;
  • the voltage gain calculation unit calculates the voltage gain according to the voltage on the DC side of the primary circuit and the voltage on the DC side of the secondary circuit;
  • the phase shift angle calculation unit uses the voltage gain to calculate the phase shift angle ⁇ 1 in the bridge of the primary circuit at the resonant frequency, and the phase angle difference ⁇ between the phase shift angle ⁇ ps between the bridges and the phase shift angle ⁇ p in the bridge 2 ;
  • the error amplification module compares the secondary current with the reference current I ref to obtain an error amplification value
  • a switching frequency calculation unit which calculates the switching frequency of the resonant dual active bridge conversion circuit according to the secondary side current
  • the PWM generation unit generates control signals PWM1-PWM8 according to the phase shift angle ⁇ 1 within the bridge and the phase shift angle ⁇ 2 between the bridges at the switching frequency and the resonant frequency, which are used to control the opening of the switching tubes in the primary side circuit and the secondary side circuit respectively with shutdown.
  • the switching frequency calculation unit includes an error amplification module and a PI control module.
  • the error amplification module compares the secondary current with the reference current I ref to obtain an error amplification value.
  • the PI control module calculates the switching frequency according to the error amplification value.
  • the PWM generating unit obtains the intra-bridge phase shift angle ⁇ p at the switching frequency and the inter-bridge phase shift angle ⁇ ps at the switching frequency according to the intra -bridge phase shift angle ⁇ 1 , the phase angle difference ⁇ 2 and the switching frequency at the resonant frequency, Generate control signals PWM1-PWM8 according to the switching frequency and the phase shift angle ⁇ p within the bridge and the phase shift angle ⁇ ps between the bridges at the switching frequency, and the control signals PWM1-PWM4 are used to control the first switching tube Q1 to the fourth switching tube respectively Q 4 , the control signals PWM5-PWM8 are respectively used to control the fifth switching tube S 1 to the fourth switching tube S 4 .
  • FIG. 3 it is an equivalent circuit of the resonant dual active bridge conversion circuit provided by the present application.
  • the relationship between the resonant capacitor voltage u Cr (t) and the resonant inductor current i Lr (t) in the equivalent circuit is as follows:
  • UP (t) is the real-time midpoint voltage of the primary-side bridge arm
  • U S (t) is the real-time value converted from the output voltage of the secondary-side bridge arm to the primary side.
  • U Cr (0) is the initial voltage of the resonant capacitor Cr
  • ⁇ r is 2 ⁇ fr
  • f r is the resonant frequency of the resonant cavity
  • I Lr (0) is the initial current of the resonant inductor Lr
  • Z r is the basic impedance
  • the present application uses the positive value of the midpoint voltage U P of the original side bridge arm as the reference value for unitization. If the midpoint voltage U P of the original side bridge arm is used as the reference value, each of the above quantities will be unitized. :
  • U CN is the per unit value of the initial voltage of the resonant capacitor Cr
  • u CN (t) is the per unit value of the real-time voltage of the resonant capacitor Cr
  • I LN is the per unit value of the initial current of the resonant inductor Lr
  • i LN ( t) is the per unit value of the real-time current of the resonant inductor Lr
  • U SN (t) is the per unit value of the real-time midpoint voltage of the secondary bridge arm converted to the primary side U S (t)
  • U PN (t ) is the real-time per unit value of the midpoint voltage U P (t) of the primary side bridge arm.
  • (u CN (t), i LN (t)) is exactly a point on a circle with (X, 0) as the center and R x as the radius.
  • the relationship between the resonant inductor current i LN (t) and the resonant capacitor voltage u CN (t) is in the form of a circle.
  • the midpoint voltage U P (t) of the primary bridge arm and the midpoint voltage of the secondary bridge arm are converted into The value of U S (t) to the original side is different, the center and radius of the circle are changing, and the trajectory of the circle is different.
  • FIG 4 it is a schematic diagram of the control signals of each switch tube and the voltage and current waveforms of the converter in a sine cycle when the voltage gain M ⁇ 1, where Up is the midpoint voltage of the primary side bridge arm, and N*Us is the secondary The midpoint voltage of the side bridge arm is converted to the value of the primary side, Up-N*Us is the voltage of the resonant inductor Lr, and Ir is the current of the resonant inductor.
  • the second switching tube Q 2 is turned off, and the fourth switching tube Q 4 is turned on when the resonant inductor current Ir changes from negative to 0, and the fourth switching tube Q 4 realizes ZVS.
  • the fourth switching tube Q4 is turned on.
  • the midpoint voltage Up of the primary side bridge arm is positive, and the seventh switching tube S2 and the sixth switching tube S3 of the secondary side circuit are kept conducting.
  • the voltage at both ends of the resonant cavity is Up+N*Us, the resonant inductor current Ir increases rapidly, and the direction changes from negative to positive.
  • the seventh switching tube S 2 and the sixth switching tube S 3 are turned off, but due to the dead time, the fifth switching tube S 1 and the eighth switching tube S 4 are not turned on immediately , at this time, the resonant inductor current Ir will continue to flow through the anti-parallel diodes of the fifth switching tube S1 and the eighth switching tube S4 , the voltage across the fifth switching tube S1 and the eighth switching tube S4 drops to 0, and the second The fifth switching tube S1 and the eighth switching tube S4 are turned on, and the fifth switching tube S1 and the eighth switching tube S4 realize soft switching ZVS.
  • the midpoint voltage Us of the secondary bridge arm changes from negative to positive.
  • the first switching tube Q1 and the fourth switching tube Q4 remain on, the voltage at both ends of the resonant cavity is still positive, and the direction of the resonant inductor current Ir is positive. , and increase slowly.
  • the first switching tube Q 1 is turned off, the third switching tube Q 3 is not turned on due to the dead time, and the resonant inductor current Ir continues through the anti-parallel diode of the third switching tube Q 3 flow, the voltage across the third switching tube Q3 drops to 0, the third switching tube Q3 is turned on, and the third switching tube Q3 realizes soft switching ZVS.
  • the fourth switching tube Q4 remains on, the midpoint voltage Up of the primary side bridge arm drops to 0, the fifth switching tube S1 and the eighth switching tube S4 keep conducting, and the midpoint voltage Us of the secondary side bridge arm is positive, the voltage at both ends of the resonant cavity is negative, and the resonant inductor current Ir decreases gradually from the positive value.
  • the three phases [t 5 , t 6 ], [t 6 , t 7 ] and [t 0 , t 1 ] of the negative half cycle are respectively related to the first phase [t 1 , t 2 ] and the second phase of the positive half cycle [t 2 , t 3 ] and the third stage [t 3 , t 4 ] correspond one-to-one, so details are not repeated here.
  • the relationship between the resonant inductor current i LN (t), the resonant capacitor voltage u CN (t) and the voltage gain M is divided into three sections, corresponding to the first Stage [t 1 , t 2 ], second stage [t 2 , t 3 ] and third stage [t 3 , t 4 ], as shown in Figure 4, the first stage [t 1 , t 2 ] corresponds to the bridge
  • the third stage [t 3 , t 4 ] corresponds to the in-bridge phase shift angle ⁇ p .
  • the second stage [t 1 , t 2 ], the third stage [t 2 , t 3 ] and the fourth stage [t 3 , t 4 ] correspond to three arc trajectories respectively.
  • the sine , the law of cosines to obtain the phase shift angle ⁇ 1 within the bridge and the phase angle difference ⁇ 2 at the resonant frequency correspond to the geometric relationship in the geometric figure.
  • the value of H3 is determined by the dead time, which is a known quantity. It can be seen that the values of phase shift angle ⁇ 1 and phase angle difference ⁇ 2 in the bridge are only related to the voltage gain M, and have nothing to do with the switching frequency. The variable is single, and the calculation is simple, and the control program is greatly simplified.
  • the intra-bridge phase shift angle ⁇ 1 and the phase angle difference ⁇ 2 calculate the intra-bridge phase shift angle ⁇ p and the inter-bridge phase shift angle ⁇ ps of the primary circuit at the switching frequency.
  • f r is the resonant frequency of the resonant cavity
  • f s is the switching frequency of the switching tube
  • the resonant frequency of the resonant cavity is determined by the resonant capacitor Cr and the resonant inductance Lr.
  • Cr is the resonant capacitor
  • Lr is the resonant inductance
  • the resonant dual active bridge conversion circuit of the present invention adopts the phase-shift control and frequency control combined control mode, adopts the time domain analysis method to obtain the relational expression of the resonant capacitor voltage and the resonant inductor current, and converts the resonant
  • the relationship between the capacitor voltage and the resonant inductor current is geometrically graphical, and the phase angle difference between the phase shift angle in the bridge and the phase shift angle between the bridge and the phase shift angle in the bridge is obtained at the resonant frequency that is only related to the voltage gain according to the geometry.
  • phase shift angle and phase angle difference in the bridge at the resonant frequency calculate the phase shift angle in the bridge and the phase shift angle between the bridges at the switching frequency, according to the phase shift angle in the bridge, the phase shift angle between the bridge and the calculated switching frequency
  • the conversion circuit is controlled, so that each switch in the circuit realizes ZVS, and the work efficiency is improved.
  • the range of voltage gain is wide and the degree of freedom of control is high.
  • the invention has the advantages of simple control, small amount of calculation, high efficiency and practicality.
  • the present application provides a control method for a resonant dual active bridge conversion circuit, including the following steps:
  • a multi-winding transformer can be used.
  • Each secondary winding of the transformer passes through a rectifier bridge and is connected in parallel to the low-voltage bus (secondary side circuit DC on the busbar on the side).
  • the primary and secondary sides of the transformer respectively include two windings.
  • the resonant dual active bridge conversion circuit includes a primary side circuit, two secondary side circuits, a resonant cavity, and a transformer with two windings on the primary side and the secondary side respectively, and each secondary side winding of the transformer is connected to a secondary side circuit.
  • the DC sides of the two secondary circuits are connected in parallel, the two primary windings of the transformer are connected in series, and the control signals of the two secondary circuits are the same.
  • the primary and secondary sides of the transformer respectively include more than two windings.
  • the resonant dual active bridge conversion circuit includes a primary side circuit, a plurality of secondary side circuits, a resonant cavity, and a transformer in which the primary side and the secondary side respectively include multiple windings.
  • the number of secondary side circuits corresponds to the number of windings, and each of the transformers
  • the secondary winding is connected to one secondary circuit, the DC sides of multiple secondary circuits are connected in parallel, multiple primary windings of the transformer are connected in series, and the control signals of the multiple secondary circuits are the same.
  • the controller provided by the present application is used to control the working state of the circuit according to the control method of the above-mentioned resonant dual active bridge conversion circuit.

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Abstract

本申请公开了一种谐振型双有源桥式变换电路的控制方法，包括以下步骤：对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；利用输入电压和输出电压计算电压增益；利用电压增益计算原边电路在谐振频率下的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；根据副边电流计算谐振型双有源桥式变换电路的开关频率；根据开关频率、谐振频率下的桥内移相角和相角差产生控制信号，控制原边电路和副边电路中开关管的开通与关断。本申请仅用电压增益计算出两个移相角，然后通过频率控制调整电路的输出电流，使电路的控制变得简单，计算量小。本申请还提供一种控制器和一种变换器。

Description

谐振型双有源桥式变换电路的控制方法、控制器及变换器
本申请要求于2021年12月02日提交的申请号为CN202111457868.0，发明名称为“谐振型双有源桥式变换电路的控制方法、控制器及变换器”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本发明属于电力电子技术领域，特别涉及谐振型双有源桥式变换电路的控制方法、控制器及变换器。
背景技术
近年由于电动汽车充电、光伏储能等行业的兴起，DC/DC变换器越来越受到人们的重视。特别是双有源桥式DC/DC变换器，可以通过变压器实现原副边电气隔离，是DC/DC变换器中的研究热点。
不过，当前DAB(Dual Active Bridge，双有源桥式)变换电路存在一些缺陷，比如电路的电压增益范围小、某些工况下开关器件的软开关失效等。
为解决上述问题，现有的一种改进方案为在DAB变换电路中加入LC谐振电路，同时对该带谐振网络的DAB变换电路，或采用频率控制，或采用移相控制，或根据输入输出电压大小和功率大小，分成不同的工作区间，每个区间择优使用移相控制或频率控制。采用以上控制的DAB变换电路可以获得更宽的工作范围和/或更高的效率，但是计算量比较大，控制复杂。
发明内容
根据本申请的各种实施例，提供一种谐振型双有源桥式变换电路的控制方法、装置及变换器，解决现有技术中使用变频移相方式调节谐振型双有源桥式变换电路计算量大，控制复杂的问题。
第一方面，本申请实施例提出一种谐振型双有源桥式变换电路的控制方法，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及变压器，所述原边电路的交流侧通过谐振腔与变压器的原边电性连 接，所述谐振腔包括串联的谐振电容和谐振电感，副边电路的交流侧与变压器的副边电性连接，所述方法包括以下步骤：
对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
利用所述电压增益计算所述原边电路在谐振频率下的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；
根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号，控制原边电路和副边电路中开关管的开通与关断。
进一步的，利用所述电压增益计算谐振频率下所述原边电路的桥内移相角，以及，原边电路和副边电路之间的桥间移相角与桥内移相角在谐振频率下的相角差的步骤包括：
获取谐振电感电流和谐振电容电压在时域内的关系式；
以原边桥臂中点电压的正值为基准值，对谐振电感电流的关系式和谐振电容电压的关系式的中的电压和电流进行标幺化，获得谐振电感电流与谐振电容电压的关系式；
根据谐振电感电流与谐振电容电压的关系式以及每个正弦半波周期中实时的原边桥臂中点电压和副边桥臂中点电压获得不同时刻的谐振电感电流、谐振电容电压与电压增益之间的函数关系；
将所述谐振电感电流与谐振电容电压的关系式以及谐振电感电流、谐振电容电压与电压增益之间的函数关系几何图形化，根据几何关系得到所述谐振频率下的桥内移相角和相角差。
进一步的，当电压增益小于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
Figure PCTCN2022135959-appb-000001
其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
进一步的，当电压增益大于或等于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
Figure PCTCN2022135959-appb-000002
其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
进一步的，根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号的步骤包括根据所述开关频率以及在谐振频率下的桥内移相角Φ 1和相角差Φ 2计算在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps，在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps分别为：
Figure PCTCN2022135959-appb-000003
其中，f r为谐振腔的谐振频率，f s为开关管的开关频率。
进一步的，所述根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率的步骤包括：
将所述副边电流与所述参考电流进行比较，获得误差放大值，根据所述误差放大值计算所述开关频率。
进一步的，所述原边电路包括第一桥臂和第二桥臂，第一桥臂中的上开关管与第二桥臂中的下开关管导通信号相差一个桥内移相角，第一桥臂中的下开关管与第二桥臂中的上开关管导通信号相差一个桥内移相角。
进一步的，所述副边电路包括第三桥臂和第四桥臂，所述第三桥臂中的上开关管和第四桥臂中的下开关管同时开关，所述第三桥臂中的下开关管和第四桥臂中的上开关管同时开关，所述第一桥臂的上开关管与所述第三桥臂的上开关管的导通信号相差一个桥间移相角。
第二方面，本申请实施例还提供一种控制器，用于控制谐振型双有源桥式变换电路，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及变压器，所述原边电路的交流侧通过谐振腔与变压器的原边电性连接，所述谐振腔包括串联的谐振电容和谐振电感，副边电路的交流侧与变压器的副边电性连接，所述控制器包括：
采样单元，分别对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
电压增益计算单元，根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
移相角计算单元，利用电压增益计算谐振频率下的原边电路的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；
开关频率计算单元，根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
PWM生成单元，根据开关频率和谐振频率下的桥内移相角以及桥间移相角产生控制信号，所述控制信号用于控制原边电路和副边电路中开关管的开通与关断。
第三方面，本申请实施例还提供一种变换器，所述变换器包括：
谐振型双有源桥式变换电路，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及原副边分别包括至少一个绕组的变压器，所述原边电路的交流侧通过谐振腔与所述变压器的原边绕组电性连接，所述谐振腔包括串联的谐振电容和谐振电感，所述副边电路与所述变压器的副边绕组一一对应，每个所述副边电路的交流侧与对应的副边绕组电性连接；
控制器，所述控制器执行本申请提供的谐振型双有源桥式变换电路的控制方法。
与现有技术相比，本发明谐振型双有源桥式变换电路采用移相控制和频率控制相结合控制方式，采用时域分析法获得谐振电容电压和谐振电感电流的关系式，并将所述谐振电容电压和谐振电感电流的关系几何图形化，根据几何图形获得仅和电压增益相关的谐振频率下的桥内移相角和桥间移相角与桥内移相角之间的相角差，根据谐振频率下的桥内移相角和相角差，计算开关频率下的桥内移相角和桥间移相角，根据桥内移相角、桥间移相角以及计算出的开关频率对变换电路进行控制，使电路中各开关实现ZVS，提高工作效率，同时电压增益的范围广、控制自由度高。本发明控制简单，计算量小，高效实用。
进一步的，提供了一种包括多个副边电路且副边电路的直流侧并联的变换器，可以防止在大功率场合低压侧(副边)开关器件的电流应力过大的问题。
本申请的一个或多个实施例的细节在以下附图和描述中提出，以使本申请的其他特征、目的和优点更加简明易懂。
附图说明
为了更好地描述和说明这里公开的本申请的实施例和/或示例，可以参考一幅或多幅附图。用于描述附图的附加细节或示例不应当被认为是对所公开的申请、目前描述的实施例和/或示例以及目前理解的这些申请的最佳模式中的任何一者的范围的限制。
图1为本申请提供的变换器示意图；
图2为本申请提供的控制器示意图；
图3为本申请提供的谐振型双有源桥式变换电路的等效电路示意图；
图4为本申请提供的电压增益M＜1时，一个周期内开关管的控制信号及变换器的电压电流波形示意图；
图5为本申请提供的电压增益M<1时，不同时间段内谐振电容电压u CN(t)和谐振电感电流i LN(t)的关系示意图；
图6为变压器原副边分别有两个绕组的谐振型双有源桥式变换电路结构示意图；
图7为变压器原副边分别有多个绕组的谐振型双有源桥式变换电路结构示意图。
具体实施方式
以下将结合附图所示的具体实施方式对本发明进行详细描述，但这些实施方式并不限制本发明，本领域的普通技术人员根据这些实施方式所做出的结构、方法、或功能上的变换均包含在本发明的保护范围内。
如图1所示，为本申请提供的变换器示意图。变换器包括谐振型双有源桥式变换电路以及控制器。其中，谐振型双有源桥式变换电路包括原边电路、副边电路、变压器以及谐振腔。通过控制器对谐振型双有源桥式变换电路的工作状态进行控制。
其中，原边电路的直流侧与输入\输出源U 1电性连接，原边电路的交流侧通过谐振腔与变压器的原边绕组电性连接。副边电路的交流侧与变压器的副边绕组电性连接，副边电路的直流侧与输入\输出源U 2电性连接。
如图1所示，原边电路包括第一全桥和稳压电容C 1，第一全桥包括第一桥臂和第二桥臂。其中，第一桥臂包括第二开关管Q 2和第四开关管Q 4，第二开关管Q 2和第四开关管Q 4的连接点作为第一桥臂的中点，第一桥臂的中点作为原边电路交流侧的第一端。第二开关管Q 2作为第一桥臂的上开关管，第四开关管Q 4作为第一桥臂的下开关管。第二桥臂包括串联的第一开关管Q 1和第三开关管Q 3，第一开关管Q 1和第三开关管Q 3的连接点作为第二桥臂的中点，第二桥臂的中点作为原边电路交流侧的第二端。第一开关管Q 1作为第二桥臂的上开关管，第三开关管Q 3作为第二桥臂的下开关管，第二桥臂的中点和第一桥臂的中点之间提供原边桥臂中点电压Up。
第一开关管Q 1至第四开关管Q 4可以为MOS管或IGBT，具体的，本申请MOS管为例说明，此时，第一开关管Q 1的漏极和第二开关管Q 2的漏极连接并作为原边电路的直流侧正端，与输入\输出源U 1的正极电性连接。第一开关管Q 1的源极和第三开关管Q 3的漏极连接，连接点作为第二桥臂的中点，第二开关管Q 2的源极与第四开关管Q 4的漏极相连，连接点作为第一桥臂的中点。第三开关管Q 3的源极和第四开关管Q 4的源极作为原边电路的直流侧负端，与输入\输出源U 1的负极电性连接。第一开关管Q 1至第四开关管Q 4的栅极分别与控制器连接，根据控制器提供的控制信号导通和关断。
稳压电容C 1与第一桥臂并联，用于稳定原边电路直流侧的电压。谐振腔包括串联的谐振电容Cr和谐振电感Lr。其中谐振电容Cr的一端与第二桥臂的中点相连，谐振电容Cr的另一端与谐振电感Lr的一端相连，谐振电感Lr的另一端与变压器的原边绕组的一端电性连接，变压器原边绕组的另一端与第一桥臂的中点电性连接。
如图1所示，副边电路包括第二全桥和稳压电容C 2，第二全桥包括并联的第三桥臂和第四桥臂，其中，第三桥臂包括第五开关管S 1和第六开关管S 3，第四桥臂包括串联的第七开关管S 2和第八开关管S 4；第五开关管S 1与第六开关管S 3的连接点作为第三桥臂的中点，第七开关管S 2和第八开关管S 4的连接点作为第四桥臂的中点，第三桥臂的中点作为副边电路交流侧的第一端，第四桥臂的中点作为副边电路交流侧的第二端。第五开关管S 1作为第三桥臂的上开关管，第六开关管S 3作为第三桥臂的下开关管，第七开关管S 2作为第四桥臂的上开关管，第八开关管S 4作为第四桥臂的下开关管，第三桥臂的中点和第四桥臂的中点之间提供副边桥臂中点电压Us。
第五开关管S 1至第八开关管S 4为MOS管或IGBT，本申请MOS管为例说明，第五开关管S 1的漏极和第七开关管S 2的漏极连接并作为副边电路的直流侧正端，与输入\输出源U 2的正极电性连接。第五开关管S 1的源极与第六开关管S 3的漏极相连，连接点作为第三桥臂的中点。第六开关管S 3的源极和第八开关管S 3的源极连接并作为第三桥臂的副边电路的直流侧负端，与输入\输出源U 2的负极电性连接。
第五开关管S 1至第八开关管S 4的栅极分别与控制器连接，根据控制器提供的控制信号导通和关断。
稳压电容C 2与第三桥臂并联，用于稳定副边电路直流侧的电压。在副边电路中，副边电路交流侧的第一端与变压器副边绕组的一端电性连接，副边电路交流侧的第二端与变压器副边绕组的另一端电性连接。作为一种可选的实现方式，副边电路交流侧可以与变压器的副边绕组之间串联一个电容。
控制器根据原边电路直流侧的电压、副边电路直流侧的电压以及副边电流对第一开关管Q 1至第四开关管Q 4、第五开关管S 1至第八开关管S 4进行移相和调频控制，产生控制第一开关管Q 1至第四开关管Q 4、第五开关管S 1至第八开关管S 4的控制信号，控制第一开关管Q 1至第四开关管Q 4、第五开关管S 1至第八开关管S 4的导通与关断。
具体的，控制器对原边电路直流侧的电压、副边电路直流侧的电压进行采样，根据原边电路直流侧的电压和副边电路直流侧的电压，计算电压增益，利用所述电压增益计算谐振频率下所述原边电路的桥内移相角Φ 1，以及，开关频率下的桥间移相角Φ ps与桥内移相角Φ p在谐振频率下的相角差Φ 2，采样副边电路的副边电流，根据所述副边电流计算开关频率，根据所述开关频率以及所述桥内移相角Φ 1、相角差Φ 2计算开关频率下的所述桥间移相角Φ ps和桥内移相角Φ p，根据所述桥内移相角Φ p、桥间移相角Φ ps以及开关频率产生控制信号，控制第一开关管Q 1至第四开关管Q 4、第五开关管S 1至第八开关管S 4的导通与关断。
对于谐振型双有源桥式变换电路，作为一种可选的实现方式，第一全桥的第一开关管Q 1和第三开关管Q 3互补导通，第二开关管Q 2和第四开关管Q 4互补导通。为了保证同一桥臂的开关管之间无直通风险，在互补导通的开关管之间设置死区。第一开关管Q 1和第四开关管Q 4的导通信号相差一个桥内移相角Φ p。第二开关管Q 2和第三开关管Q 3的导通信号也相差一个桥内移相角Φ p。
作为一种可选的实现方式，第二全桥中第五开关管S 1和第七开关管S 3互补导通，第六开关管S 2和第八开关管S 4互补导通。且第五开关管S 1和第 八开关管S 4同时导通与关断，第六开关管S 2和第七开关管S 3同时导通与关断。在互补导通的开关管之间设置死区。
由于第一桥臂的第一开关管Q 1和第四开关管Q 4管的导通信号相差一个桥内移相角Φ p，第一全桥的第一开关管Q 1的开通信号与第二全桥的第五开关管S 1开通信号相差一个桥间移相角Φ ps，因此在开关频率下，原边桥臂中点电压Up与副边桥臂中点电压Us相位相差相角Φ ps-Φ p。
通过控制桥间移相角Φ ps、桥内移相角Φ p可以控制功率流向，功率可以从原边电路向副边电路传递，也可以从副边电路向原边电路传递。
图2示出本发明实施例的控制器的示意性框图，如图2所示，所述控制器包括：
采样单元，分别对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
电压增益计算单元，根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
移相角计算单元，利用电压增益计算谐振频率下的原边电路的桥内移相角Φ 1，以及，桥间移相角Φ ps与桥内移相角Φ p之间的相角差Φ 2；
误差放大模块，将副边电流与参考电流I ref相比较，获得误差放大值；
开关频率计算单元，根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
PWM生成单元，根据开关频率和谐振频率下的桥内移相角Φ 1以及桥间移相角Φ 2产生控制信号PWM1-PWM8，分别用于控制原边电路和副边电路中开关管的开通与关断。
具体的，开关频率计算单元包括误差放大模块和PI控制模块，误差放大模块将副边电流与参考电流I ref相比较，获得误差放大值，PI控制模块根据误差放大值计算开关频率。
PWM生成单元根据谐振频率下的桥内移相角Φ 1、相角差Φ 2以及开关频率，获得开关频率下的桥内移相角Φ p以及开关频率下的桥间移相角Φ ps，根据开关频率和开关频率下的桥内移相角Φ p以及桥间移相角Φ ps产生控制信号PWM1-PWM8，控制信号PWM1-PWM4分别用于控制第一开关管Q 1至第四开关管Q 4，控制信号PWM5-PWM8分别用于控制第五开关管S 1至第四开关管S 4。
以下将对桥内移相角Φ p和桥间移相角Φ ps的计算原理说明如下：
具体的，获取谐振电感电流和谐振电容电压在时域内的关系式。
如图3所示，为本申请提供的谐振型双有源桥式变换电路的等效电路。在等效电路中的谐振电容电压u Cr(t)以及谐振电感电流i Lr(t)关系式如下：
Figure PCTCN2022135959-appb-000004
其中，U P(t)为实时的原边桥臂中点电压，U S(t)为实时的副边桥臂输出电压折算到原边的值。U Cr(0)为谐振电容Cr的初始电压，ω r为2πf r，f r为谐振腔的谐振频率，I Lr(0)为谐振电感Lr的初始电流，Z r为基本阻抗，
Figure PCTCN2022135959-appb-000005
进一步的，对谐振电感电流i Lr(t)的关系式和谐振电容电压u Cr(t)的关系式中的电压和电流进行标幺化，获得谐振电感电流i Lr(t)关于谐振电容电压u Cr(t)的关系式。
具体的，本申请以原边桥臂中点电压U P的正值为基准值进行标幺化，若以原边桥臂中点电压U P为基准值，将上面的各个量标幺化处理：
Figure PCTCN2022135959-appb-000006
Figure PCTCN2022135959-appb-000007
其中，U CN为谐振电容Cr的初始电压的标幺值，u CN(t)为谐振电容Cr的实时电压的标幺值，I LN为谐振电感Lr的初始电流的标幺值，i LN(t)为谐振电感 Lr的实时电流的标幺值，U SN(t)为实时的副边桥臂中点电压折算到原边后的值U S(t)的标幺值，U PN(t)为实时的原边桥臂中点电压U P(t)标幺值。
具体的，令X＝U PN(t)-U SN(t)，可得：
Figure PCTCN2022135959-appb-000008
整理后可得：
[u CN(t)-X] 2+[i LN(t)] 2＝R x 2
其中
Figure PCTCN2022135959-appb-000009
由此可知，(u CN(t)，i LN(t))正好是以(X，0)为圆心，以R x为半径的圆上的点。谐振电感电流i LN(t)与谐振电容电压u CN(t)的关系式为圆的形式，在不同时刻，原边桥臂中点电压U P(t)和副边桥臂中点电压折算到原边的值U S(t)的值不同，圆心和半径在变化，圆的轨迹不同。
进一步的，根据谐振电感电流i LN(t)与谐振电容电压u Cr(t)的关系式以及每个正弦半波周期中原边桥臂中点电压U P(t)和副边桥臂中点电压折算到原边的值U S(t)获得不同时刻谐振电感电流i LN(t)、谐振电容电压u Cr(t)与电压增益M之间的函数关系。
如图4所示，为电压增益M＜1时，一个正弦周期内各开关管的控制信号及变换器的电压电流波形示意图，其中，Up为原边桥臂中点电压、N*Us为副边桥臂中点电压折算到原边的值、Up-N*Us为谐振电感Lr的电压、Ir为谐振电感电流。
变换电路在正半周期的工作过程说明如下：
第一阶段[t 1,t 2]，第二开关管Q 2关断，第四开关管Q 4在谐振电感电流Ir由负变为0时开通，第四开关管Q 4实现ZVS。经过死区时间后，第四开关管Q 4导通，此时原边桥臂中点电压Up为正，副边电路第七开关管S 2和第六开关管S 3保持导通，此时谐振腔两端电压为Up+N*Us，谐振电感电流Ir迅速增大，方向由负变为正。
第二阶段[t 2,t 3]，第七开关管S 2和第六开关管S 3关断，由于死区时间的存在，第五开关管S 1和第八开关管S 4没有立刻开通，此时谐振电感电流Ir会经由第五开关管S 1和第八开关管S 4的反并联二极管续流，第五开关管S 1和第八开关管S 4两端电压降到0，第五开关管S 1和第八开关管S 4导通，第五开关管S 1和第八开关管S 4实现软开关ZVS。副边桥臂中点电压Us由负变为正，此时第一开关管Q 1和第四开关管Q 4保持导通，谐振腔两端电压仍是正值，谐振电感电流Ir方向为正，且缓慢增加。
第三阶段[t 3,t 4]，第一开关管Q 1关断，第三开关管Q 3由于死区时间并未开通，谐振电感电流Ir经由第三开关管Q 3的反并联二极管续流，第三开关管Q 3两端电压降到0，第三开关管Q 3开通，第三开关管Q 3实现软开关ZVS。此时第四开关管Q 4保持导通状态，原边桥臂中点电压Up降到0，第五开关管S 1和第八开关管S 4保持导通，副边桥臂中点电压Us为正，谐振腔两端的电压为负值，谐振电感电流Ir由正值逐渐减小。
负半周期的三个阶段[t 5,t 6]、[t 6,t 7]以及[t 0,t 1]分别与正半周期的第一阶段[t 1,t 2]、第二阶段[t 2,t 3]以及第三阶段[t 3,t 4]一一对应，因此不再赘述。本发明实施例的变换电路在M>1的工作模态，与M<1时的差异在于移相角不同，具体工作过程也不详细列出。
进一步的，根据以上分析可知，在一个正弦半波周期，谐振电感电流i LN(t)、谐振电容电压u CN(t)与电压增益M之间的关系式分为三段，分别对应第一阶段[t 1,t 2]、第二阶段[t 2,t 3]以及第三阶段[t 3,t 4]，由图4可知，第一阶段[t 1,t 2]对应所述桥间移相角Φ ps与所述桥内移相角Φ p之间的相角差，第三阶段[t 3,t 4]对应所述桥内移相角Φ p。
将所述谐振电感电流i LN(t)与谐振电容电压u CN(t)的关系式以及谐振电感电流i LN(t)、谐振电容电压u CN(t)与电压增益M之间的关系式几何图形化，根据几何关系得到谐振频率下的所述桥内移相角Φ 1和所述相角差Φ 2。
具体的，在直角坐标系中作出不同时间段内谐振电容电压u CN(t)和谐振电感电流i LN(t)的几何图形，如图5所示，为电压增益M<1时，不同时间段内谐振电容电压u CN(t)和谐振电感电流i LN(t)的示意图。其中，Mc为谐振电容Cr的电压峰值以原边桥臂中点电压的正值U P为基准值的标幺值。第二阶段[t 1,t 2]、 第三阶段[t 2,t 3]以及第四阶段[t 3,t 4]分别对应三段圆弧轨迹，根据几何图形中的几何关系，利用正弦、余弦定理得到所述谐振频率下的桥内移相角Φ 1和所述相角差Φ 2。
最终得到M<1时，桥内移相角Φ 1和相角差Φ 2的表达式为：
Figure PCTCN2022135959-appb-000010
上式中，H 3的值由死区时间决定，为已知量。可以看出，桥内移相角Φ 1和相角差Φ 2的值仅与电压增益M有关，与开关频率并无关系，变量单一，且计算简便，控制程序大大简化。
作为另一种可选的实现方式，当电压增益M大于或等于1时，u CN(t)和i LN(t)的方程，以及对应的图形会有不同，下面仅给出当M大于或等于1时，桥内移相角Φ 1和相角差Φ 2的计算公式如下，具体过程不再详细说明。
Figure PCTCN2022135959-appb-000011
根据桥内移相角Φ 1和相角差Φ 2计算开关频率下的原边电路的桥内移相角Φ p，以及桥间移相角Φ ps。
具体的，
Figure PCTCN2022135959-appb-000012
其中，f r为谐振腔的谐振频率，f s为开关管的开关频率，谐振腔的谐振频率由谐振电容Cr和谐振电感Lr决定。
具体的，
Figure PCTCN2022135959-appb-000013
Cr为谐振电容，Lr为谐振电感。
综上所述，本发明谐振型双有源桥式变换电路采用移相控制和频率控制相结合控制方式，采用时域分析法获得谐振电容电压和谐振电感电流的关系式，并将所述谐振电容电压和谐振电感电流的关系几何图形化，根据几何图形获得仅和电压增益相关的谐振频率下的桥内移相角和桥间移相角与桥内移 相角之间的相角差，根据谐振频率下的桥内移相角和相角差，计算开关频率下的桥内移相角和桥间移相角，根据桥内移相角、桥间移相角以及计算出的开关频率对变换电路进行控制，使电路中各开关实现ZVS，提高工作效率，同时电压增益的范围广、控制自由度高。本发明控制简单，计算量小，高效实用。
作为一种可选的实现方式，本申请提供一种谐振型双有源桥式变换电路的控制方法，包括以下步骤：
S 1、分别对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样。
S 2、根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益。
S 3、利用电压增益计算谐振频率下的原边电路的桥内移相角Φ 1，以及，桥间移相角Φ ps与桥内移相角Φ p在谐振频率下的相角差Φ 2。
S 4、根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率。
S5、根据开关频率和谐振频率下的桥内移相角Φ 1以及桥间移相角Φ 2产生控制信号PWM1-PWM8，分别用于控制原边电路和副边电路中开关管的开通与关断。
在大功率场合，为防止低压侧(副边)开关器件的电流应力过大，可以使用多绕组变压器，变压器的每个副边绕组各经过一个整流桥，再并联至低压母线(副边电路直流侧的母线)上。如图6所示，作为另一种可选的实现方式，本申请提供的变换器中，变压器原副边分别包括两个绕组。具体的，谐振型双有源桥式变换电路包括原边电路、两个副边电路、谐振腔以及原副边分别具有两个绕组的变压器，变压器的每个副边绕组连接一个副边电路，两个副边电路的直流侧并联连接，变压器的两个原边绕组串联连接，两个副边电路的控制信号相同。
如图7所示，作为另一种可选的实现方式，本申请提供的变换器中，变压器原副边分别包括两个以上的绕组。具体的，谐振型双有源桥式变换电路包括原边电路、多个副边电路、谐振腔以及原副边分别包括多个绕组的变压器，副边电路数量与绕组数量对应，变压器的每个副边绕组连接一个副边电 路，多个副边电路的直流侧并联连接，变压器的多个原边绕组串联连接，多个副边电路的控制信号相同。利用本申请提供的控制器根据上述谐振型双有源桥式变换电路的控制方法对电路的工作状态进行控制。
以上所揭露的仅为本发明的较佳实施例而已，然其并非用以限定本发明之权利范围，本领域普通技术人员可以理解：在不脱离本发明及所附的权利要求的精神和范围内，改变、修饰、替代、组合、简化，均应为等效的置换方式，仍属于发明所涵盖的范围。

Claims (17)

  1. 一种谐振型双有源桥式变换电路的控制方法，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及变压器，所述原边电路的交流侧通过谐振腔与变压器的原边电性连接，所述谐振腔包括串联的谐振电容和谐振电感，副边电路的交流侧与变压器的副边电性连接，
    其特征在于，所述方法包括以下步骤：
    对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
    根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
    利用所述电压增益计算所述原边电路在谐振频率下的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；
    根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
    根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号，控制原边电路和副边电路中开关管的开通与关断。
  2. 根据权利要求1所述的控制方法，其特征在于，利用所述电压增益计算谐振频率下所述原边电路的桥内移相角，以及，原边电路和副边电路之间的桥间移相角与桥内移相角在谐振频率下的相角差的步骤包括：
    获取谐振电感电流和谐振电容电压在时域内的关系式；
    以原边桥臂中点电压的正值为基准值，对谐振电感电流的关系式和谐振电容电压的关系式的中的电压和电流进行标幺化，获得谐振电感电流与谐振电容电压的关系式；
    根据谐振电感电流与谐振电容电压的关系式以及每个正弦半波周期中实时的原边桥臂中点电压和副边桥臂中点电压获得不同时刻的谐振电感电流、谐振电容电压与电压增益之间的函数关系；
    将所述谐振电感电流与谐振电容电压的关系式以及谐振电感电流、谐振电容电压与电压增益之间的函数关系几何图形化，根据几何关系得到所述谐振频率下的桥内移相角和相角差。
  3. 根据权利要求1所述的控制方法，其特征在于，当电压增益小于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
    Figure PCTCN2022135959-appb-100001
    其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
  4. 根据权利要求1所述的控制方法，其特征在于，当电压增益大于或等于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
    Figure PCTCN2022135959-appb-100002
    其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
  5. 根据权利要求1所述的控制方法，其特征在于，根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号的步骤包括根据所述开关频率以及在谐振频率下的桥内移相角Φ 1和相角差Φ 2计算在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps，在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps分别为：
    Figure PCTCN2022135959-appb-100003
    其中，f r为谐振腔的谐振频率，f s为开关管的开关频率。
  6. 根据权利要求1所述的控制方法，其特征在于，所述根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率的步骤包括：
    将所述副边电流与所述参考电流进行比较，获得误差放大值，根据所述误差放大值计算所述开关频率。
  7. 根据权利要求1所述的控制方法，其特征在于：所述原边电路包括第一桥臂和第二桥臂，第一桥臂中的上开关管与第二桥臂中的下开关管导通信 号相差一个桥内移相角，第一桥臂中的下开关管与第二桥臂中的上开关管导通信号相差一个桥内移相角。
  8. 根据权利要求7所述的控制方法，其特征在于：所述副边电路包括第三桥臂和第四桥臂，所述第三桥臂中的上开关管和第四桥臂中的下开关管同时开关，所述第三桥臂中的下开关管和第四桥臂中的上开关管同时开关，所述第一桥臂的上开关管与所述第三桥臂的上开关管的导通信号相差一个桥间移相角。
  9. 一种控制器，用于控制谐振型双有源桥式变换电路，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及变压器，所述原边电路的交流侧通过谐振腔与变压器的原边电性连接，所述谐振腔包括串联的谐振电容和谐振电感，副边电路的交流侧与变压器的副边电性连接，其特征在于，所述控制器包括：
    采样单元，分别对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
    电压增益计算单元，根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
    移相角计算单元，利用电压增益计算谐振频率下的原边电路的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；
    开关频率计算单元，根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
    PWM生成单元，根据开关频率和谐振频率下的桥内移相角以及桥间移相角产生控制信号，所述控制信号用于控制原边电路和副边电路中开关管的开通与关断。
  10. 一种变换器，其特征在于，所述变换器包括：
    谐振型双有源桥式变换电路，所述谐振型双有源桥式变换电路包括原边电路、至少一个副边电路、谐振腔以及原副边分别包括至少一个绕组的变压器，所述原边电路的交流侧通过谐振腔与所述变压器的原边绕组电性连接， 所述谐振腔包括串联的谐振电容和谐振电感，所述副边电路与所述变压器的副边绕组一一对应，每个所述副边电路的交流侧与对应的副边绕组电性连接；
    控制器，所述控制器对所述谐振型双有源桥式变换电路执行如下控制方法：
    对原边电路直流侧的电压、副边电路直流侧的电压以及副边电流进行采样；
    根据原边电路直流侧的电压和副边电路直流侧的电压计算电压增益；
    利用所述电压增益计算所述原边电路在谐振频率下的桥内移相角，以及，开关频率下的桥间移相角与桥内移相角在谐振频率下的相角差；
    根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率；
    根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号，控制原边电路和副边电路中开关管的开通与关断。
  11. 根据权利要求10所述的变换器，其特征在于，
    利用所述电压增益计算谐振频率下所述原边电路的桥内移相角，以及，原边电路和副边电路之间的桥间移相角与桥内移相角在谐振频率下的相角差的步骤包括：
    获取谐振电感电流和谐振电容电压在时域内的关系式；
    以原边桥臂中点电压的正值为基准值，对谐振电感电流的关系式和谐振电容电压的关系式的中的电压和电流进行标幺化，获得谐振电感电流与谐振电容电压的关系式；
    根据谐振电感电流与谐振电容电压的关系式以及每个正弦半波周期中实时的原边桥臂中点电压和副边桥臂中点电压获得不同时刻的谐振电感电流、谐振电容电压与电压增益之间的函数关系；
    将所述谐振电感电流与谐振电容电压的关系式以及谐振电感电流、谐振电容电压与电压增益之间的函数关系几何图形化，根据几何关系得到所述谐振频率下的桥内移相角和相角差。
  12. 根据权利要求10所述的变换器，其特征在于，
    当电压增益小于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
    Figure PCTCN2022135959-appb-100004
    其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
  13. 根据权利要求10所述的变换器，其特征在于，
    当电压增益大于或等于1时，所述谐振频率下的桥内移相角Φ 1和相角差Φ 2为：
    Figure PCTCN2022135959-appb-100005
    其中，H 3的值由死区时间决定，为已知量，M为电压增益，Mc为谐振电容的电压峰值以原边桥臂中点电压的正值为基准值的标幺值。
  14. 根据权利要求10所述的控制方法，其特征在于，
    根据所述开关频率、谐振频率下的桥内移相角和相角差产生控制信号的步骤包括根据所述开关频率以及在谐振频率下的桥内移相角Φ 1和相角差Φ 2计算在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps，在所述开关频率下的桥内移相角Φ p和桥间移相角Φ ps分别为：
    Figure PCTCN2022135959-appb-100006
    其中，f r为谐振腔的谐振频率，f s为开关管的开关频率。
  15. 根据权利要求10所述的变换器，其特征在于，
    所述根据所述副边电流计算所述谐振型双有源桥式变换电路的开关频率的步骤包括：
    将所述副边电流与所述参考电流进行比较，获得误差放大值，根据所述误差放大值计算所述开关频率。
  16. 根据权利要求10所述的变换器，其特征在于，
    所述原边电路包括第一桥臂和第二桥臂，第一桥臂中的上开关管与第二桥臂中的下开关管导通信号相差一个桥内移相角，第一桥臂中的下开关管与第二桥臂中的上开关管导通信号相差一个桥内移相角。
  17. 根据权利要求16所述的变换器，其特征在于，
    所述副边电路包括第三桥臂和第四桥臂，所述第三桥臂中的上开关管和第四桥臂中的下开关管同时开关，所述第三桥臂中的下开关管和第四桥臂中的上开关管同时开关，所述第一桥臂的上开关管与所述第三桥臂的上开关管的导通信号相差一个桥间移相角。
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