WO2022206700A1 - 一种目标检测方法及装置 - Google Patents

一种目标检测方法及装置 Download PDF

Info

Publication number
WO2022206700A1
WO2022206700A1 PCT/CN2022/083456 CN2022083456W WO2022206700A1 WO 2022206700 A1 WO2022206700 A1 WO 2022206700A1 CN 2022083456 W CN2022083456 W CN 2022083456W WO 2022206700 A1 WO2022206700 A1 WO 2022206700A1
Authority
WO
WIPO (PCT)
Prior art keywords
sequence
signal
detection
target
correlation peak
Prior art date
Application number
PCT/CN2022/083456
Other languages
English (en)
French (fr)
Inventor
毛天奇
陈家璇
王昭诚
王琪
Original Assignee
华为技术有限公司
清华大学
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 华为技术有限公司, 清华大学 filed Critical 华为技术有限公司
Publication of WO2022206700A1 publication Critical patent/WO2022206700A1/zh

Links

Images

Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/36—Means for anti-jamming, e.g. ECCM, i.e. electronic counter-counter measures
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G01S7/411—Identification of targets based on measurements of radar reflectivity
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G01S7/417—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section involving the use of neural networks
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
    • H04B1/40—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04W—WIRELESS COMMUNICATION NETWORKS
    • H04W4/00—Services specially adapted for wireless communication networks; Facilities therefor
    • H04W4/02—Services making use of location information
    • H04W4/029—Location-based management or tracking services

Definitions

  • the present application relates to the field of communications, and in particular, to a target detection method and device based on terahertz communication detection integrated waveform.
  • the terahertz frequency band has abundant spectrum resources and can provide ultra-high available bandwidth, which can support high-speed data transmission of hundreds of gigabits per second (Gbps). This satisfies the demand for ultra-high transmission rates of many data services under the rapid development of wireless communication, and it can be seen that terahertz communication may become one of the key technologies of the 6th generation (generation) mobile communication technology.
  • the terahertz frequency band has a large amount of available bandwidth, it can be used for target detection or perception to effectively improve the target ranging resolution and improve the detection accuracy.
  • the beam of terahertz signal is extremely narrow, it can effectively reduce the clutter interference caused by multipath when it is used for radar detection.
  • terahertz signals in detection is very broad. Because both terahertz communication and detection systems need to occupy ultra-wide spectrum resources, and the production cost is high. Therefore, it is necessary to conduct in-depth research on terahertz communication detection, such as the integration of terahertz communication detection.
  • the integration of terahertz communication and detection can effectively reduce equipment cost, reduce equipment size and improve spectrum utilization by sharing hardware resources and spectrum resources for communication and detection.
  • terahertz detection requires ultra-high bandwidth to improve detection accuracy. Therefore, an analog-to-digital converter (ADC) with ultra-high sampling rate is required at the receiving end. to avoid signal distortion. Obviously, its production is difficult and expensive.
  • ADC analog-to-digital converter
  • the research on the integration of terahertz communication and detection is still in the exploratory stage. When the communication and detection functions are performed simultaneously, mutual interference will occur and the system performance will be deteriorated. Therefore, it is urgent to propose a new solution for the integrated system of terahertz communication detection to solve the above problems.
  • the embodiment of the present application provides a target detection method, by using a multi-subband quasi-perfect (multi-band quasi-perfect, MS-QP) sequence with multiple subbands as the transmitted communication coordination sequence signal, relying on the MS-QP sequence
  • the wider bandwidth is divided into multiple narrowbands.
  • each subband corresponds to a narrowband.
  • the narrowband Zadoff-Chu (ZC) sequence is transmitted on each self-contained signal, thereby ensuring that the communication cooperative sequence signal has strong autocorrelation characteristics, so that when the echo signal generated by the communication cooperative sequence signal is received
  • the radar detection component can be determined according to the echo signal, and the target detection can be realized by correlating the detection sequence component and the radar detection component. Since the transmitted communication coordination sequence signal divides the wider bandwidth into multiple narrow bands, the use of a high-cost high-sampling ADC can be avoided when the echo signal is received, thereby reducing the hardware cost.
  • a target detection method comprising: sending a communication coordination sequence signal.
  • a detection sequence component may be included in the communication cooperation sequence signal.
  • the detection sequence component is an MS-QP sequence with multiple subbands, and each subband may include a narrowband ZC sequence.
  • FFT fast Fourier transformation
  • each subband of the MS-QP sequence with multiple subbands includes a narrowband ZC sequence, thereby ensuring that the transmitted communication coordination sequence signal can have good autocorrelation characteristics so as to be used for target detection.
  • the large bandwidth occupied by the transmitted information is divided into multiple narrowbands, so that a low sampling rate ADC can be used when the echo signal is received, so as to reduce the hardware cost.
  • the method may further include: using an optimal phase factor set to perform phase adjustment on each ZC sequence in the MS-QP sequence.
  • the present application can also perform phase adjustment on each ZC sequence in the MS-QP sequence, thereby reducing the peak-to-average ratio of the signal.
  • the ZC sequence may include: sequence length L m and root index p. where the root exponent p satisfies L m represents the length of the ZC sequence on the mth subband, and m is a positive integer greater than or equal to 2.
  • the method may further include: a frequency domain guard interval (GI) is provided between the ZC sequences of every two adjacent subbands.
  • a frequency domain guard interval is set between adjacent subbands, so as to avoid mutual interference between different subbands, and to avoid data loss in the connecting part of the subbands during decoding.
  • the communication coordination sequence signal further includes: a data symbol component.
  • the data symbol component may include data sequences of multiple subbands, and the data sequences on each subband are transmitted on multiple zero-set frequency points on the subband.
  • the zero frequency point is obtained by retransmitting the ZC sequence on the subband M times in the time domain.
  • the present application retransmits the ZC sequence on each subband multiple times, so that multiple zero-setting frequency points can appear in the frequency domain, and digital symbol components are transmitted on the zero-setting frequency points, so that the present application can simultaneously realize communication and detection two functions, and there will be no interference between each other.
  • the method may further include: performing corresponding phase adjustment for the data sequence transmitted on each subband.
  • the present application can also perform corresponding phase adjustment on the data sequence, so that transmission can be performed at the corresponding zero-setting frequency point.
  • determining at least one target and at least one target distance according to at least one correlation peak and time delay information corresponding to at least one correlation peak may include: at least one correlation peak is in one-to-one correspondence with at least one target, so that The corresponding target can be determined from the correlation peak. And, the target distance of the target corresponding to each correlation peak is determined according to the time delay information corresponding to at least one correlation peak.
  • a target detection apparatus in a second aspect, includes: a transmitter for sending a communication coordination sequence signal, wherein the communication coordination sequence signal includes a detection sequence component, and the detection sequence component is an MS-QP sequence with multiple subbands , each subband includes a narrowband ZC sequence; a receiver for receiving the echo signal of the communication cooperating sequence signal; a processor for coupling with the memory, and reading and executing instructions stored in the memory; The instruction causes the processor to perform fast Fourier transform on the echo signal to determine the radar detection component of the echo signal in the frequency domain; correlate the detection sequence component and the radar detection component to determine at least one correlation peak and at least one correlation peak Time delay information corresponding to the peak; at least one target and at least one target distance are determined according to the at least one correlation peak and the time delay information corresponding to the at least one correlation peak.
  • each subband of the MS-QP sequence with multiple subbands includes a narrowband ZC sequence, thereby ensuring that the transmitted communication coordination sequence signal can have good autocorrelation characteristics so as to be used for target detection.
  • the large bandwidth occupied by the transmitted information is divided into multiple narrowbands, so that a low sampling rate ADC can be used when the echo signal is received, so as to reduce the hardware cost.
  • the processor is further configured to: perform phase adjustment on each ZC sequence in the MS-QP sequence by adopting the optimal phase factor set.
  • the present application can also perform phase adjustment on each ZC sequence in the MS-QP sequence, thereby reducing the peak-to-average ratio of the signal.
  • the ZC sequence includes: a sequence length L m and a root index p, wherein the root index p satisfies L m represents the length of the ZC sequence on the mth subband, and m is a positive integer greater than or equal to 2.
  • the processor is further configured to have a frequency domain guard interval between the ZC sequences of every two adjacent subbands.
  • a frequency domain guard interval is set between adjacent subbands, so as to avoid mutual interference between different subbands, and to avoid data loss in the connecting part of the subbands during decoding.
  • the communication coordination sequence signal further includes: a data symbol component.
  • the data symbol component includes data sequences of multiple subbands, and the data sequence on each subband is transmitted on multiple zeroing frequency points on the subband, and the zeroing frequency point is the ZC sequence on the subband in the time domain. It is obtained by performing M retransmissions.
  • the present application retransmits the ZC sequence on each subband multiple times, so that multiple zero-setting frequency points can appear in the frequency domain, and digital symbol components are transmitted on the zero-setting frequency points, so that the present application can simultaneously realize communication and detection two functions, and there will be no interference between each other.
  • the processor is further configured to: perform corresponding phase adjustment for the data sequence transmitted on each subband.
  • the present application can also perform corresponding phase adjustment on the data sequence, so that transmission can be performed at the corresponding zero-setting frequency point.
  • the processor is further configured to: one-to-one correspondence between the at least one correlation peak and the at least one target, so that the corresponding target can be determined through the correlation peak. And, the target distance of the target corresponding to the at least one correlation peak is determined according to the time delay information corresponding to the at least one correlation peak.
  • a computer-readable storage medium where instructions are stored in the computer-readable storage medium, and when the instructions are executed on a terminal, the terminal is made to execute any one of the methods in the first aspect.
  • a computer device containing an instruction, which, when running on a terminal, causes the terminal to execute any one of the methods in the first aspect.
  • a computer program product comprising instructions which, when run on a computer, cause the computer to perform any one of the methods of the first aspect.
  • the present application discloses a target detection method and device.
  • an MS-QP sequence with multiple subbands as the transmitted communication cooperative sequence signal, it not only perfectly inherits the autocorrelation characteristics of the ZC sequence, but also can convert the broadband required for communication It is divided into multiple narrowbands for transmission, thus avoiding the use of expensive high-sampling ADCs and reducing hardware costs when receiving echo signals.
  • FIG. 1 is a schematic diagram of an application scenario provided by an embodiment of the present application.
  • FIG. 2 is a schematic structural diagram of a communication detection integrated device provided by an embodiment of the present application.
  • FIG. 3 is a schematic diagram of a communication detection integrated device provided by an embodiment of the present application.
  • FIG. 4 is a flowchart of a method for integrating communication detection provided by an embodiment of the present application.
  • FIG. 5 is a schematic diagram of spectrum relocation of a sending device according to an embodiment of the present application.
  • FIG. 6 is a schematic diagram of another transmission device spectrum relocation provided by an embodiment of the present application.
  • FIG. 7 is a schematic time domain representation of an MS-QP sequence provided by an embodiment of the present application.
  • FIG. 8 is a schematic diagram of a frequency domain representation of an MS-QP sequence provided by an embodiment of the present application.
  • FIG. 9 is a schematic diagram of a narrowband ZC sequence autocorrelation characteristic
  • FIG. 10 is a schematic diagram of an MS-QP sequence autocorrelation characteristic provided by an embodiment of the present application.
  • FIG. 11 is a schematic diagram of a multi-subband communication system provided by an embodiment of the present application.
  • FIG. 12 is a schematic structural diagram of an MS-QP sequence provided by an embodiment of the present application.
  • FIG. 13 is a schematic diagram of a spectrum of an MS-QP sequence after time domain expansion provided by an embodiment of the present application.
  • FIG. 14 is a schematic diagram of a single-subband communication cooperative sequence signal provided by an embodiment of the present application.
  • 15 is a schematic diagram of a range image affected by hardware mismatch
  • 16 is a schematic diagram of a range image affected by a Doppler frequency shift
  • 17 is a schematic diagram of an autocorrelation change curve of a narrowband ZC sequence provided by an embodiment of the application.
  • 18 is a schematic diagram of a distance image before root index optimization provided by an embodiment of the application.
  • 19 is a schematic diagram of a distance image after root index optimization provided by an embodiment of the application.
  • FIG. 20 is a schematic diagram of constructing an echo signal of a receiving device according to an embodiment of the application.
  • 21 is a schematic diagram of signal processing of a receiving device in an mth subband according to an embodiment of the present application.
  • 22 is a schematic diagram of determining DRM from an echo signal according to an embodiment of the present application.
  • FIG. 23 is a schematic diagram of a single subband spectrum of a communication coordination sequence signal provided by an embodiment of the present application.
  • FIG. 24 is a schematic diagram of a DRM generation process signal waveform provided by an embodiment of the application.
  • FIG. 25 is a schematic diagram of a sequence ranging performance comparison provided by an embodiment of the present application.
  • Figure 26 is a schematic diagram of a sequence velocity measurement performance comparison provided in an embodiment of the application.
  • FIG. 27 is a schematic diagram of the relationship between frequency domain ratio and ranging performance according to an embodiment of the present application.
  • FIG. 28 is a schematic diagram of the relationship between frequency domain ratio and speed measurement performance according to an embodiment of the present application.
  • FIG. 29 is a schematic diagram of another integrated device for communication detection provided by an embodiment of the present application.
  • FIG. 30 is a schematic diagram of yet another integrated device for communication detection provided by an embodiment of the present application.
  • the present application is mainly applied to the scene of target detection.
  • the sending device 110 of the integrated communication detection device 100 sends a sending signal.
  • the transmitted signal meets the target 200 after passing a certain distance, and then the transmitted signal is reflected by the target 200 to form an echo signal.
  • the echo signal is received by the receiving device 120 of the integrated communication and detection apparatus 100 after passing through the same distance again.
  • the received echo signal is analyzed by the receiving device 120 to determine whether the target 200 exists and the distance between the target 200 and the integrated communication detection device 100 .
  • the integrated communication detection device 100 shown in FIG. 1 is an integrated device including a sending device 110 and a receiving device 120. In other scenarios, the sending device 110 and the receiving device 120 may also be two independent devices. installation.
  • the transmitting device 110 and the receiving device 120 can share the same set of hardware resources, thereby reducing the manufacturing cost.
  • the target 200 in this scene is described by taking a certain user as an example. It can be understood that the target 200 may also be any other object, which is not limited in this application.
  • the integrated communication detection device 100 having the sending device 110 and the receiving device 120 is used for communication and detection in most cases. Simultaneous communication and detection.
  • the transmitting device 110 and the receiving device 120 for communication and detection share a set of hardware resources, the location of the transmitting device 110 and the receiving device 120 can be approximately regarded as the same, and share the local oscillator (Can be referred to as the local oscillator).
  • the transmitting device 110 may also be referred to as a signal transmitting end
  • the receiving device 120 may also be referred to as a radar detecting end.
  • the local oscillator may be an inductor capacitor (LC) oscillator or the like. Due to the large attenuation of terahertz signal propagation, the scenario shown in Figure 1 is usually considered for short-range object detection. In order to effectively combat propagation attenuation, the communication detection coordination system can usually use high-gain directional antennas for signal transmission to generate extremely narrow beams, so that line-of-sight links and multipath effects can be ignored in this scenario.
  • the line-of-sight link refers to the signal transmitted between the communication and detection integrated device 100 and the target 200 in a straight line distance, ignoring the echo signal transmitted from other directions after the transmitted signal may be reflected by the target 200 .
  • the communication detection coordination system can still detect one or more targets. Human pose detection. Among them, different body parts can be regarded as different targets.
  • the integrated communication detection device 100 transmits a transmission signal, and the user 200 shown on the right side in FIG. 1 can hold a terminal device to receive the transmission signal.
  • the integrated communication detection device 100 can analyze the echo signal to realize the detection of the user's gesture and action.
  • the communication detection integrated apparatus 100 may include a sending device 110 and a receiving device 120 .
  • the transmitting device 110 first generates a baseband signal to be transmitted through the coding and modulation module 111, and sends the baseband signal to a digital-to-analog converter (DAC) 112 to convert the digital baseband signal into an analog signal. Signal.
  • DAC digital-to-analog converter
  • the transmission power is amplified by the high power amplifier 114 to form a transmission signal, and finally the transmission signal is sent out through a transmission antenna (TX ANT) 115.
  • TX ANT transmission antenna
  • the receiving device 120 in the integrated communication detection device 100 can receive the echo signal formed after the transmitted signal is reflected by the target 200 through the receiving antenna (receive antenna, RX ANT) 121 .
  • the received echo signal is processed by the low noise amplifier 122, and then the frequency of the processed echo signal is reduced by the down-conversion module 123, and the echo signal with the reduced frequency is converted from the frequency domain to the time by the ADC. domain to form the baseband signal corresponding to the echo signal.
  • the receiving device 120 can input the baseband signal corresponding to the echo signal to the radar detection module 125 for target detection, and input the baseband signal corresponding to the echo signal to the demodulation and decoding module 126 to obtain the transmitted information.
  • orthogonal frequency division multiplexing orthogonal frequency division multiplexing
  • single-carrier transmission waveforms lower-frequency communication detection coordination systems are usually used, such as OFDM signals, cyclic prefix-single carrier (CP-SC), preamble sequences in the data frame structure, and direct Sequence spread spectrum signal, etc. to perform communication detection coordination.
  • OFDM orthogonal frequency division multiplexing
  • CP-SC cyclic prefix-single carrier
  • preamble sequences in the data frame structure preamble sequences in the data frame structure
  • direct Sequence spread spectrum signal etc.
  • the advantage of the OFDM waveform is that it can be used for both communication and detection, and the maximum likelihood-based distance/velocity detection method is more robust to Doppler shift.
  • a[n'] is sent out through the transmitting antenna.
  • the OFDM time domain signal is sent to the communication receiving end, and part of the energy is received by the communication receiving end for signal demodulation, while the other part is reflected on the surface of the communication receiving end to form an echo signal and transmit it to the transmitting end.
  • the communication receiving end may be, for example, the target 200 in FIG.
  • the transmitting end adopts the integrated communication detection device
  • it can be received by the receiving device (or called the radar receiving end) at the same location, and the received echo signal can be expressed as y[n'].
  • the receiving device currently two detection algorithms can be generally used to detect y[n'].
  • the first way is that the receiving device and the sending device in the integrated communication detection device share a set of hardware, so the receiving device knows the sent a[n']. Therefore, by correlating y[n'] with a[n'], the distance image can be obtained. The position of the correlation peak in the range image is then analyzed and the target distance is estimated.
  • the operation of the detection algorithm in this method is relatively simple, but the autocorrelation characteristics of a[n'] are often not ideal.
  • the time domain sequence has ideal autocorrelation characteristics, especially in a[n'] In the case of randomness, its autocorrelation characteristics cannot be guaranteed accordingly, which affects the improvement of detection performance.
  • the second way is to do discrete Fourier transform (discrete fourier transform, DFT) on y[n'] to obtain Y[k] in the frequency domain. then according to The frequency domain response of the channel impulse response can be obtained. After that, an inverse discrete Fourier transform (IDFT) is performed on I[k] to obtain a range image for radar detection. For this method, by comparing the received signal and the transmitted signal on each subband, the shape of the range image is no longer affected by the randomness of the transmitted signal.
  • DFT discrete Fourier transform
  • the CP-SC is a single-carrier signal using a cyclic prefix (cyclic prefixed, CP).
  • CP cyclic prefixed
  • N" can represent the number of data in each group of signals, and dw' [n"] represents the n"th data of the wth group.
  • each group of signals dw' [n"] can use a preset constellation diagram For modulation, for example, quadrature phase-shift keying (QPSK) can be used for modulation.
  • QPSK quadrature phase-shift keying
  • the W' group CP-SC signal When the W' group CP-SC signal is modulated, it can be sent by the transmitting antenna as a transmitting signal, so that the target can receive it. It can be understood that a part of the energy of the transmitted signal is received by the target and the signal is demodulated, and at the same time, another part of the energy is reflected by the target surface to form an echo signal.
  • the receiving device For the echo signal, it can be received by the receiving device in the same position as the transmitting device, and the received echo signal can be expressed as y w' [n"].
  • the receiving device performs a cyclic correlation of length N” between the received data block y[n′′] of each group and the corresponding transmitted data d[n′′], so as to obtain the correlation result with the length of W' group of N′′, which can be written as r(n", w').
  • a point Z-point FFT can be performed on W' corresponding to each relevant position.
  • Z can typically take the value qW', where q is a positive integer.
  • R(n", k') a range-doppler matrix
  • R(n", k') a range-doppler matrix
  • the value range of k' is the same as that of w'.
  • Traverse R(n", k') to determine the horizontal and vertical coordinates corresponding to the maximum value.
  • the target distance and the target relative moving speed are obtained.
  • commonly used single carrier transmission waveforms may include linear frequency modulated (LFM) signals, pseudorandom spreading codes, perfect sequences, and the like.
  • LFM linear frequency modulated
  • pseudo-random spreading code may be, for example, the longest linear shift register sequence (may be referred to as an m sequence), a Gold sequence, or the like.
  • the perfect sequence can be, for example, a ZC sequence, a Frank sequence, or the like. It can be understood that when the CP-SC signal is directly used for radar detection, the implementation is relatively simple, and the peak average is relatively low. However, due to the randomness of the modulation symbols, the good autocorrelation properties of the sequence cannot be guaranteed, resulting in limited detection performance.
  • the data frame structure when a single-carrier preamble sequence is used as a transmission signal, the data frame structure includes a short training field (short training field, STF) and a channel estimation field (channel estimation field, CEF).
  • STF short training field
  • CEF channel estimation field
  • STF can be used for synchronization and frequency offset estimation
  • CEF can be used for channel estimation.
  • the single-carrier preamble sequence can be composed of multiple sets of Golay complementary sequences with good autocorrelation properties, so it is suitable for radar detection.
  • the transmitting device transmits the transmitted signal to the target by the transmitting antenna, and part of the energy of the transmitted signal is received by the target and demodulated, while another part of the energy is reflected by the surface of the target to form an echo signal.
  • the echo signal can be received by the receiving device in the same position as the transmitting device, and the target detection can be performed according to the preamble sequence in the echo signal.
  • the STF and the CEF can be taken as a whole to perform cross-correlation between the preamble sequence of the transmitted signal and the preamble sequence of the received signal during transmission.
  • the target distance is determined according to the target echo delay and the speed of light.
  • the carrier frequency offset estimation method of wireless local area network (WLAN) can also be used for estimation, so as to determine the relative moving speed of the target.
  • WLAN wireless local area network
  • the length of the single-carrier preamble sequence is usually short, it cannot resist the extremely high path loss of the terahertz channel. In other words, the single-carrier preamble sequence cannot resist the extremely low signal-to-noise ratio of the receiving device.
  • m-sequence may also be used to perform direct sequence spread spectrum on the single-carrier communication signal.
  • the single-carrier communication signal may be, for example, a phase shift keying (phase shift keying, PSK) symbol sequence.
  • PSK phase shift keying
  • a sufficiently large spreading ratio can be set, so that the spread spectrum can have good autocorrelation characteristics of the pseudo-random code m-sequence.
  • the transmitted signal is modulated using U-order PSK.
  • U represents the order of PSK, which is generally a positive integer.
  • N the length of the data symbol before spreading
  • N the corresponding bit of the i-th data symbol can be written as u is 0/1 bit (bit).
  • the first bits can be written as where L' can be a positive integer, After that, each element in the m sequence can be used and Each element in is added modulo 2, that is The obtained data is passed through the modulator to obtain L' M-PSK symbols subjected to direct sequence spread spectrum. Similarly, the above-described spread spectrum operation can be performed on each data symbol to generate a direct spread spectrum sequence signal of length N"L' as a transmit signal. In some examples, the transmitting device sends the transmit signal to the transmit antenna.
  • the target part of the energy of the transmitted signal is received by the target and demodulated, and at the same time, another part of the energy is reflected by the target surface to form an echo signal.
  • the echo signal it can be received by the receiving device in the same location as the transmitting device, And carry out target detection according to the echo signal.
  • Corresponding process can refer to the existing mode based on time domain, and the application will not repeat them here.
  • the spread spectrum signal has good autocorrelation characteristics, but when the spread spectrum ratio is set higher When , the communication rate will be reduced, the code length will be too large, and the influence of Doppler frequency shift will be aggravated.
  • the communication detection coordination system using terahertz needs to use an ultra-high signal bandwidth, so the receiving device also needs a very high sampling rate, which makes the ADC production difficult and costly.
  • the present application provides a target detection method by using an MS-QP sequence with multiple subbands as the transmitted communication coordination sequence signal, wherein each subband may include a narrowband ZC sequence. Then, after receiving the echo signal formed by the reflection of the communication coordination sequence signal, FFT is performed on the echo signal, so as to determine the radar detection component of the echo signal in the frequency domain. The detection sequence component and the radar detection component are correlated to determine at least one correlation peak and time delay information corresponding to the at least one correlation peak, and determine at least one target and target distances corresponding to each target.
  • the communication cooperation sequence signal sent by the present application can be guaranteed to have good autocorrelation characteristics so as to be used for target detection. At the same time, through multiple subbands, the large bandwidth occupied by the transmitted information is divided into multiple narrowbands, so that a low sampling rate ADC can be used when the echo signal is received, so as to reduce the hardware cost.
  • FIG. 3 is a schematic diagram of an integrated device for communication detection according to an embodiment of the present application.
  • the present application provides an integrated device 300 for communication detection.
  • the communication detection integrated device 300 can be applied to the signal transmission and reception scenarios during communication and detection, and may include: one or more processors 301, one or more memories 302, a transmitter 303, a receiver 304, one or more A plurality of antennas 305 and a bus 306.
  • the processor 301 , the memory 302 , the transmitter 303 and the receiver 304 in the integrated communication detection device 300 can establish a communication connection through the bus 306 .
  • the transmitter 303 and the receiver 304 can also be connected to the antenna 305 , so that the transmitter 303 transmits data through the antenna 305 , and the receiver 304 receives echo signals through the antenna 305 .
  • the transmitter 303 and the receiver 304 may share a set of hardware devices. It can be understood that the transmitter 303 is the transmitting device 110 in FIG. 1 and FIG. 2 , and the receiver 304 is the receiving device 120 in FIG. 1 and FIG. 2 .
  • the integrated communication detection device 300 may include more or less components than shown, or some components may be combined, or some components may be separated, or different components may be arranged.
  • the illustrated components may be implemented in hardware, software, or a combination of software and hardware.
  • the processor 301 may be an advanced reduced instruction set processor (advanced reduced instruction set computing machines, ARM), X86, a microprocessor without interlocked piped stages (MIPS) and other architectures .
  • the processor 301 may include one or more processing units, such as: application processor (application processor, AP), modem processor, GPU, image signal processor (image signal processor, ISP), controller, video codec processor, digital signal processor (digital signal processor, DSP), baseband processor and/or neural-network processing unit (neural-network processing unit, NPU), etc.
  • different processing units may be independent devices, or may be integrated in one or more processors.
  • Memory 302 is used to store instructions and data.
  • memory 302 is a cache memory.
  • the memory 302 may store instructions or data that have just been used or recycled by the processor 301 . If the processor 301 needs to use the instruction or data again, it can be called directly from the memory 302 . Repeated access is avoided, and the waiting time of the processor 301 is reduced, thereby improving the efficiency of the system.
  • the memory 302 may include a memory, and the memory may store an operating system, an image file, and the like.
  • the memory 302 may also include auxiliary memory or external memory, such as a non-removable memory or a removable memory card and the like.
  • the internal memory may be used to store computer executable program code, and the executable program code includes instructions.
  • the internal memory may include high-speed random access memory, and may also include non-volatile memory, such as at least one magnetic disk storage device, flash memory device, universal flash storage (UFS), and the like.
  • the processor 301 executes various functional applications and data processing of the integrated communication detection device 300 by executing the instructions stored in the memory.
  • the integrated communication detection device 300 may be a terminal device, for example, including but not limited to a mobile phone, a smart TV, a smart speaker, a wearable device, a tablet computer, a desktop computer, a handheld computer, a notebook computer, and a super mobile personal computer.
  • UMPC ultra-mobile personal computer
  • PDA personal digital assistant
  • laptop computer laptop
  • mobile computer augmented reality (augmented reality, AR) device
  • virtual reality (virtual reality, VR ) devices artificial intelligence (AI) devices
  • in-vehicle devices smart home devices and/or smart city devices and other terminal devices or portable terminal devices.
  • the integrated communication detection apparatus 300 may also be a network device, such as a wireless base station, a repeater, a network node, and the like.
  • FIG. 4 is a flowchart of a method for integrating communication detection according to an embodiment of the present application.
  • the present application further provides a method for integrating communication detection, which can be applied to the integrated communication detection device 300 shown in FIG. 3 .
  • the method includes the following steps:
  • the transmitter 303 in the integrated communication detection device 300 transmits the communication coordination sequence signal through the antenna 305 .
  • the communication cooperation sequence signal includes a detection sequence component.
  • the detection sequence component is an MS-QP sequence with multiple subbands, and each subband corresponds to a narrowband ZC sequence.
  • the MS-QP sequence is constructed by means of ZC sequence for the detection sequence component in the communication cooperative sequence signal. Since both the perfect sequence long and short codes have ideal autocorrelation characteristics, the perfect sequence is very suitable for target detection in the terahertz band. Therefore, the MS-QP sequence of the present application is constructed by the ZC sequence, thereby ensuring that the MS-QP sequence also has good detection performance.
  • the MS-QP sequence may be constructed using multiple narrowband ZC sequences. It can be understood that the construction of the MS-QP sequence can be implemented by the processor 301 . First, perform DFT on multiple narrowband ZC sequences to the frequency domain. Wherein, each narrowband ZC sequence in the plurality of narrowband ZC sequences may be the same or different. After that, the spectrum of multiple narrowband ZC sequences is moved and spliced, and the spectrum of each narrowband ZC sequence constitutes multiple adjacent subbands and forms an ultra-wideband spectrum. It can be understood that each subband corresponds to a narrowband ZC sequence.
  • the processor 301 performs IDFT on the UWB spectrum to obtain the time domain representation of the MS-QP sequence.
  • ultra-high bandwidth is required when transmitting the transmitted signal. It is understandable that when the bandwidth of the transmitted signal is higher, the corresponding ranging resolution will be more accurate. Therefore, when the MS-QP sequence has an ultra-high signal bandwidth, the ranging resolution can be effectively improved.
  • a frequency domain guard interval LG is inserted during splicing and splicing, so that there is a certain interval in the frequency domain between two adjacent subbands, so as to avoid interference between adjacent subbands.
  • the length of b m [n] is L m
  • f m represents the center frequency of the mth subband. For each narrowband ZC sequence its corresponding symbol interval is denoted as Tm .
  • x[n] is represented as an MS-QP sequence after frequency domain extension construction, and its length is ML G + ⁇ L m .
  • the length may be ⁇ L m .
  • T M time domain symbol interval
  • x(t) The analog signal output by the transmitting device in Fig. 5
  • the process of generating the analog signal x(t) can be, for each subband, first pass b m [n] through the shaping filter, and then convert the digital signal to an analog version via the DAC.
  • the shaping filter is mainly used to convert the clock rate of the signal from the symbol rate to the sampling rate of digital-to-analog (DA), so as to be subsequently input into the DAC for processing.
  • DA digital-to-analog
  • the spectrum is shifted after being converted into an analog signal, that is, each subband in the figure is multiplied by a multiplier
  • j is an imaginary unit
  • t is time. It can be understood that both f' m and f m are expressed as the center frequency of the mth subband, where f m is mainly used for computing discrete points in the digital domain, and f' m is mainly used for analog signals. domain to achieve spectrum shifting.
  • the analog signals after moving the multiple sub-bands are superimposed by an adder to form an output analog signal x(t).
  • the above method is to perform spectrum shifting on the analog signal side.
  • spectrum shifting can also be performed on the digital signal side.
  • FIG. 6 Another schematic diagram of spectrum shifting of a sending device shown in FIG. 6 is shown. It can be seen that this method first passes the shaping filter on different subbands b m [n], and performs spectrum shifting on the digital signal on each subband, that is, each subband in the figure is multiplied by the multiplier. To achieve spectrum shifting. After that, an adder is used to superimpose the spectrum-shifted digital signals to obtain a digital signal x[n] to be sent. After that, the digital signal x[n] is converted into an analog signal by a DAC, and the analog signal is raised to the terahertz frequency band, such as multiplying by a multiplier The realization of upscaling the analog signal to the terahertz band to obtain the transmit signal x(t).
  • f c can be expressed as the center frequency of the bandwidth required to transmit the signal.
  • the DAC shown in Figure 6 needs to use a high-sampling DAC to complete the digital-to-analog conversion of the ultra-wideband signal.
  • the time-domain symbol interval is TM at this time.
  • the digital signal is converted into an analog signal by a DAC before spectrum shifting. Therefore, the DAC in FIG. 5 can be implemented by selecting multiple low-sampling DACs, and the time-domain symbol interval is only T m .
  • the expression of the digital signal x[n] can be determined based on the manners provided in FIG. 5 and FIG. 6 , where the digital signal x[n] is the above MS-QP sequence.
  • the narrowband ZC sequence b m [n] on each subband can be expressed by Equation 1.
  • n"' is used to represent the serial number of the narrowband ZC sequence
  • n"' 0, 1, . . . , L m -1.
  • exp represents an exponential function with the base e
  • p m is the root exponent
  • L m may be an odd number
  • gcd is expressed as the greatest common divisor.
  • Equation 2 is used to represent the sequence number of the MS-QP sequence. Equation 2 can be further transformed into Equation 3.
  • the frequency domain guard interval when the frequency domain guard interval is set, although x[n] is composed of multiple narrowband ZC sequences, it is not a real perfect sequence, but it still has good autocorrelation characteristics. This is because the sequence has perfect autocorrelation property, which is equivalent to the transverse mode of the frequency domain amplitude spectrum of the sequence.
  • the frequency domain guard interval set between adjacent subbands accounts for a lower proportion of the overall bandwidth occupied by the transmitted signal, the frequency domain amplitude spectrum of x[n] is closer to the transverse mode, and the autocorrelation characteristic is better.
  • FIG. 7 shows a schematic diagram of a time domain representation of an MS-QP sequence.
  • the length of each narrowband ZC sequence is 1007, and a total of 4 subbands are set as an example, and the time domain diagram corresponding to the generated MS-QP sequence x[n] is shown in FIG. 7 .
  • the transverse model characteristic is no longer satisfied in the time domain, and still has a certain peak-to-average ratio.
  • the peak-to-average ratio can be reflected by the jitter amplitude of the signal in the time domain, for example, the higher the peak-to-average ratio, the greater the signal amplitude jitter (it can also be said to be more severe).
  • the x[n] spectrum is composed of multiple narrowband ZC sequence spectrums.
  • each block in Fig. 8 shows the spectrum of a narrowband ZC sequence.
  • the spectrum of x[n] is close to the transverse mode, so it shows that x[n] has good autocorrelation characteristics.
  • the amplitude value of the signal remains unchanged with the change of frequency, which means the transverse mode in this frequency band.
  • the frequency spectrum of x[n] will have the phenomenon that the amplitude value is 0 at some frequencies. So the spectrum of x[n] is close to the transverse mode, not the true transverse mode.
  • Figure 9 shows a schematic diagram of the autocorrelation characteristics of the narrowband ZC sequence. It can be seen that it has perfect autocorrelation properties for narrowband ZC sequences. Of course the same is true for other perfect sequences.
  • 10 shows a schematic diagram of the autocorrelation characteristics of the MS-QP sequence of the present application. It can be seen that the MS-QP sequence also has quasi-perfect autocorrelation characteristics, but compared with the narrow-band ZC sequence, small side lobes will be generated near the main peak, as shown in Figure 10, when the delay is close to 0 and other non-zero time delays , will have a smaller magnitude.
  • phase-frequency characteristics of the MS-QP sequence basically do not affect its autocorrelation characteristics, it can be considered to adjust the phase of the sequence to reduce the peak-to-average ratio of the signal.
  • different phase factors may be multiplied on different subbands of the MS-QP sequence respectively.
  • a finite set can be preset in, contains multiple phase factors. For example, when the number of subbands is M, it is possible to M phase factors are selected as the phase factor set.
  • phase factor set available from a finite set All possible phase factor sets are traversed in , and the peak-to-average ratio of the MS-QP sequence using the phase factor set is calculated. The phase factor set with the smallest peak-to-average ratio is selected as the optimal phase factor set. And use the optimal phase factor set to adjust the phase of the narrowband ZC sequence on each subband in the MS-QP sequence.
  • Equation 3 Equation 3
  • phase factor satisfies:
  • E() represents expectation
  • represents modulus
  • argmin represents when When taking the minimum value value of .
  • Equation 1 can also be equivalent to Equation 7.
  • the formula 6 is combined to obtain the MS-QP sequence with a smaller peak-to-average ratio.
  • FIG. 11 is a schematic diagram of a multi-subband communication system. It can be seen that the structure of the multi-subband communication system is similar to that of the MS-QP sequence, and the required frequency band can be divided into multiple subbands, such as M subbands. And a frequency domain guard interval is inserted between two adjacent subbands. Different subbands can be allocated to different data services, for example, subband 1 configures service 1, and subband 2 configures service 2. It can be understood that the structure of the MS-QP sequence can be as shown in FIG. 12 . The difference from FIG. 11 is that no corresponding service is configured on each subband, but each narrowband ZC sequence is configured.
  • the MS-QP sequence shown in FIG. 12 needs to transmit data for communication, that is, combining FIG. 11 with FIG. 12 , it can only be implemented by means of time division duplexing, which makes the efficiency extremely low. Therefore, in some examples, the above MS-QP sequence can be further improved, so that the transmitted communication coordination sequence signal can support target detection and communication at the same time without mutual interference.
  • the communication coordination sequence signal may also include a data symbol component.
  • the data symbol component includes data sequences of multiple subbands.
  • the narrowband ZC sequence on each subband in the MS-QP sequence can be repeatedly transmitted M' times, then in the time domain, it is extended to M' times the original length.
  • the transmitted sequence The length is L m M'
  • the time-domain extended sequence can be denoted as be ,m [n].
  • M' is a positive integer greater than or equal to 2.
  • the narrowband ZC sequence before the narrowband ZC sequence is extended in the time domain, there are almost no frequency points with an amplitude of 0, and even some frequency points have an amplitude of 0, but there will still be other data transmitted on this frequency point. For example, as shown by the solid line, the amplitude of part of the solid line is 0, but the frequency point where the amplitude is 0 still has an amplitude of 1.
  • the narrowband ZC sequence after time domain expansion is obtained as shown by the dotted line, and there will be multiple frequency points with a frequency of 0, that is, a zero frequency point.
  • the zero-setting frequency point is, for example, the position circled by the circle in FIG. 13 .
  • Equation 8 shows that the The communication coordination sequence signal after superimposition with be ,m [n]
  • n"" 0, 1, . . . , L m M'-1.
  • ⁇ e, m and ⁇ i', m are power distribution factors, which can be preset.
  • FIG. 14 is a schematic diagram of a single-subband communication coordination sequence signal provided by an embodiment of the present application. It can be understood that FIG. 14 only shows a process of generating a communication coordination sequence signal on a certain subband, and for each subband, the communication coordination sequence signal on each subband can be generated with reference to the manner shown in the figure. Thereby forming a communication cooperative sequence signal
  • the narrowband ZC sequence transmitted on the subband forms M' copies after M' retransmissions, such as narrowband ZC sequence copy 1, narrowband ZC sequence copy 2... narrowband ZC sequence copy M'.
  • the data sequence may also be repeatedly transmitted M' times in the time domain, for example, data sequence copy 1, data sequence copy 2...data sequence copy M'.
  • a cyclic prefix may also be added to the communication coordination sequence signal on each subband, so as to be used for data demodulation by the receiver of the communication coordination sequence signal.
  • formula 9 can be further obtained by performing spectrum shifting and frequency domain superposition on the communication cooperation sequence signals on multiple different subbands.
  • the multi-subband communication coordination sequence signal Its spectral efficiency can be
  • the unit may be bit/s/Hz (bit/s/Hz), L is the PSK modulation order, and ⁇ is the proportion of the cyclic prefix duration in the total duration of the sequence.
  • the signal-to-noise ratio of the echo signals generated by the transmitted communication coordination sequence signals is much lower than that of radars in the millimeter wave band.
  • the signal-to-noise ratio of the echo signal detected by the receiving device of the integrated communication detection device can even be as low as -56.6 decibels (dB). Therefore, when the communication coordination sequence signal is sent in S401, cyclic and repeated transmission may be performed, for example, the communication coordination sequence signal is repeatedly repeated in a frame structure.
  • Each frame can contain multiple word blocks, and each word block transmits a communication coordination sequence signal, thereby overcoming the low signal-to-noise ratio in the signal transmission process.
  • each word block transmits a communication coordination sequence signal, thereby overcoming the low signal-to-noise ratio in the signal transmission process.
  • S402 Receive an echo signal of the communication coordination sequence signal.
  • the receiver 304 in the integrated communication detection device 300 receives the echo signal of the communication coordination sequence signal through the antenna 305 . It can be understood that, when the echo signal is received, only a plurality of band-pass filters are needed to separate the signals in each sub-band, so an ADC with a low sampling rate can be used to collect the echo signal.
  • Signals in the terahertz band are usually affected by hardware mismatches generated by the transmitter 303 and the receiver 304, and produce a certain nonlinear mixing distortion.
  • the hardware mismatch mainly includes the amplitude and phase imbalance of the in-phase branch/quadrature branch in the transmitter 303 and the receiver 304 structure for the terahertz frequency band, that is, in-phase/quadrature (in-phase/quadrature, I /Q) out of balance.
  • the phase noise generated by the transceiver local oscillator and the nonlinearity of the high-gain power amplifier will cause nonlinear distortion of the signal, and affect the performance of terahertz communication and detection.
  • the transmitter 303 has compensated for transmit-side I/Q imbalance and non-linearity. Therefore the echo signal is only subject to the phase noise and I/Q imbalance of the receiver 304 . Of course, the echo signal may also be disturbed by other clutter signals besides the noise generated by the device. Because the beam of annunciators using the terahertz frequency band is usually narrow, the clutter interference is less, so that the clutter signal strength is relatively weak.
  • the received clutter in the echo signal can be jointly modeled with the thermal noise as additive complex white Gaussian noise and used as an equivalent noise term
  • its symbol period can be T s , assuming that the number of targets to be detected is Q', the echo delay of the q'-th target is ⁇ q' T s , and the corresponding channel fading
  • the coefficient is h q' , and the normalized Doppler is recorded as u q' is the target moving speed of the q'th target, c is the propagation speed of electromagnetic waves, the I/Q imbalance coefficient is recorded as ⁇ I/Q and ⁇ I/Q , and the phase noise of the local oscillator is recorded as ⁇ (n), so
  • the echo signal y[n] can be as shown in Equation 12.
  • the distance image refers to the correlation function image of the transmitted signal and the echo signal, which is used for target detection and time delay estimation. Therefore, the present application can ignore the hardware mismatch problem.
  • FIG. 15 is a schematic diagram of a range image affected by hardware mismatch. It can be seen that the correlation peak appears when the time delay is 1000. It can be understood that the correlation peak can also be called the main peak. On both sides of the correlation peak, that is, when the delay is around 1000, there are small side lobes.
  • p m satisfies 0 ⁇ p m ⁇ L m
  • q is 0.
  • the narrow-band ZC sequence has many good properties, such as the narrow-band ZC sequence is a transverse mode sequence, and the peak-to-average ratio is low; and the narrow-band ZC sequence has perfect autocorrelation characteristics, as shown in Equation 14,
  • n""' represents the point to be detected
  • m' represents the point to be detected after a time delay.
  • narrowband ZC sequences also include low cross-correlation properties.
  • two narrowband ZC sequences b 1 [n"'] and b 2 [n"'] of the same length have root exponents p 1 and p 2 respectively, then when gcd(
  • , When L m ) 1, is a constant
  • v q' L m can be approximated to 0, and the corresponding remainder when the corresponding n"' takes a non-zero value (ie (p m n"'-v q' L m )(mod L m )) are all non-zero integers, and the corresponding value of
  • approaches the highest value, resulting in a correlation peak.
  • the narrow-band ZC sequence has perfect autocorrelation characteristics.
  • p m can be optimized so that
  • the high side lobes and correlation peaks generated by the Doppler frequency shift can be concentrated as much as possible, thereby preventing the range side lobes caused by the Doppler frequency shift from being misjudged as the target.
  • Fig. 18 shows a schematic diagram of the distance profile before root index optimization.
  • the root exponent is 3
  • the high sidelobes generated by the Doppler frequency shift are well gathered near the correlation peak, so only one correlation peak is seen in Figure 19.
  • the target number is 1 for description.
  • the autocorrelation property of MS-QP sequence is equivalent to the superposition of autocorrelation on different subband components.
  • the distribution of its autocorrelation side lobes mainly depends on the autocorrelation of the narrowband ZC sequence on the subband. distributed. Therefore, for the Doppler frequency shift received by the MS-QP sequence, the sidelobe distribution generated in the range profile mainly depends on the narrowband ZC sequence in each subband.
  • the above-mentioned optimization for the root index is implemented when the communication cooperation sequence signal is sent in S401, which actually optimizes the root index in the communication cooperation sequence signal, so that the communication cooperation sequence is received in S402. After the echo signal of the signal, the target detection can be carried out very well.
  • FIG. 20 is a schematic diagram of constructing an echo signal of a receiving device according to an embodiment of the present application.
  • Fig. 20 shows a corresponding receiving device.
  • y(t) represents the analog signal of the echo signal
  • y[n] represents the baseband digital signal of the echo signal.
  • y[n] and x[n] in FIG. 5 and FIG. 6 are obviously the same as the symbol interval, and both are T M . It can be seen from FIG. 20 that, in the receiving device 120, multiple band-pass filters are first required to separate signals corresponding to different subbands.
  • the low-sampling ADC performs analog-to-digital conversion on the signal on each subband to obtain a digital baseband signal of the corresponding subband.
  • the sequence is reconstructed through the digital signal processing module, and y[n] is obtained.
  • the multiplier is multiplied by The separation of the signals on the individual subbands is achieved.
  • S403 will be executed after S402.
  • FFT is performed on the echo signal to determine the radar detection component of the echo signal in the frequency domain.
  • the receiver 304 can perform FFT on the echo signal y[n], so that the radar detection component of the echo signal can be determined in the frequency domain.
  • y[n] may have multiple subbands, so S403 may be to convert the analog signal on each subband into a baseband digital signal through an ADC. Then, for each subband, an FFT of L mm ' points can be performed to convert from the time domain to the frequency domain, for example, S2101 shown in FIG. 21 . At this time, the narrowband ZC sequence transmitted on the corresponding non-zero frequency point, that is, the radar detection component, can be taken out.
  • y[n] since y[n] is the echo signal formed after x[n] is reflected by the target, y[n] should also have the detection sequence component and the data symbol component, wherein, in y[n]
  • the detection sequence component is called the radar detection component, and is also the MS-QP sequence involved in the above. So that it can be used for channel estimation and target detection later, that is, S2102 in FIG. 21 .
  • the radar detection component since the cross-correlation between the radar detection component and the data symbol component is very small, the radar detection component may not be separated out by converting it to a baseband digital signal through ADC at y[n], but the data symbol carrying the data can be directly used.
  • the MS-QP sequence of the components is used for subsequent object detection.
  • the data sequence at the corresponding zeroed frequency point can also be taken out, that is, the data sequence in y[n] Data symbol components. Then, perform IFFT of Lmm' point on the data symbol components in y[ n ] to convert from frequency domain to time domain, that is, S2103 in Figure 21, so that signal demodulation can be continued subsequently, that is, S2104 in Figure 21. This process realizes the data transfer function.
  • one or more correlation peaks and time delay information of each correlation peak may be determined by correlating the detected sequence components at the time of transmission with the received radar detection components.
  • the received echo signal y[n] can be determined first, which can be shown by referring to formula 11.
  • formula 12 Equation 16.
  • K represents the number of sub-blocks included in a frame in the repeated frame structure, and K" is a positive integer greater than or equal to 2.
  • k 0,1,...,K"-1.
  • n 1 0, 1, . . . , N 1 -1, N 1 is the length of the MS-QP sequence transmitted on the k"th block.
  • the noise interference term It is understood that noise as well as data symbols may be included, where The reason for the generation of the data symbols in is due to the existence of Doppler frequency offset and some non-ideal factors, resulting in that the data symbol components and the detection sequence components are not perfectly orthogonal during data transmission, so a small amount of interference will still be generated.
  • "Point FFT to obtain RDM For example, Figure 22 shows a process of generating DRM. It can be clearly seen that for each element in DRM, its two-dimensional coordinates correspond to a possible target echo delay. and Doppler shift values. In some examples, for No special processing is required.
  • the target moving speed can also be estimated.
  • At least one correlation peak can be determined by means of constant false alarm probability detection.
  • the hypothesis testing judgment threshold is determined by a preset false alarm rate. And by comparing the element amplitude at each position in the DRM with the hypothesis test judgment threshold, it is determined whether there is a correlation peak.
  • the false alarm rate can be preset and unchanged, and each time the target is detected, the judgment threshold of this hypothesis test can be dynamically determined according to the false alarm value, for example, it can be determined by fully automatic calculation and simulation.
  • the judgment threshold of this hypothesis test is dynamically determined according to the false alarm value.
  • the DRM determined according to the echo signal is denoted as R(n 1 , k”)
  • the hypothesis test for determining whether the (n 1 , k”) th cell has a correlation peak can be expressed as formula 17.
  • H 1 means that the relevant peak (ie the target) is found
  • H 0 means that the relevant peak (ie the target) is not found.
  • the T threshold represents the judgment threshold of this hypothesis test
  • ⁇ 2 represents the noise power in the corresponding DRM unit.
  • the corresponding Doppler frequency offset parameter can be expressed as
  • the corresponding target speed u can be represented by formula 18, which includes two cases of positive speed and negative speed.
  • the absolute error range is The mean absolute error is where ⁇ is the wavelength of the terahertz signal, for example, it can be 3 ⁇ 10 8 /f c .
  • S405 may be continued.
  • S405 Determine at least one target and at least one target distance according to at least one correlation peak and time delay information corresponding to at least one correlation peak.
  • the at least one target and the corresponding target distance may be determined according to the at least one correlation peak and the time delay information of each correlation peak determined in S404.
  • the above-mentioned communication coordination sequence signal can gather the side lobes caused by Doppler frequency shift to the vicinity of the correlation peak, and in practice, the probability of multiple targets appearing in a small area is often extremely low. It can therefore be assumed that no other targets exist in the vicinity of the correlation peak. Thus, the influence of Doppler side lobes on the false alarm probability of the system is reduced.
  • the RDM Correlation peaks are detected in cells, it is considered that There is no target in the range except the target corresponding to the correlation peak.
  • n 0 can be understood as a preset dynamic range. Therefore, according to each correlation peak, a unique target corresponding to it can be determined.
  • the target distance it can be estimated as where c is the speed of electromagnetic wave propagation (also known as the speed of light).
  • the absolute error range of the d- target distance can be The mean absolute error is
  • the present application determines the communication cooperation sequence signal composed of the detection sequence component sum by extending based on the time domain.
  • the spectrum of the expanded detection sequence component generates multiple zero-value frequency points so as to transmit the data symbol component, thereby realizing the simultaneous operation of communication and detection without interference.
  • the above method ensures that correlation detection for long sequences can be supported with a low-complexity algorithm. Since the detection sequence component is a MS-QP sequence based on spectrum spreading, the problem that the terahertz ultra-wideband signal requires high sampling rate of the ADC in the receiving device is solved by dividing it into multiple subbands.
  • a narrowband ZC sequence is transmitted on each subband, so it has good autocorrelation characteristics. It is guaranteed that the receiving device can use the low sampling rate ADC to collect and restore the signal while using the ultra-high bandwidth, which effectively reduces the cost.
  • the present application also multiplies the different subband components of the sequence by different phase factors, and optimizes the phase factors to reduce the peak of the sequence. average ratio, and does not affect the sequence autocorrelation properties.
  • the root index in the MS-QP sequence is optimized, so that the high-range image sidelobes caused by Doppler frequency shift are gathered near the main peak, and A strategy of eliminating the adjacent area of the main peak is proposed to effectively reduce the interference of Doppler on target detection.
  • FIG. 23 is a schematic diagram of a single subband spectrum of a communication coordination sequence signal according to an embodiment of the present application.
  • FIG. 23 it is a schematic diagram of a single subband spectrum after substituting the above-mentioned communication coordination sequence signal for specific parameters.
  • the length L m of the narrowband ZC sequence in a single subband is 1007, and the other root exponent p m is 503 to minimize the influence of Doppler frequency shift on target detection.
  • the symbol interval of the narrow-band ZC sequence in a single subband can be set to be 1.1 nanoseconds (ns), and the total sequence length is 1.1077 microseconds ( ⁇ s).
  • a subband component can be formed with a bandwidth of 0.9GHz.
  • the number of times M' of MS-QP sequence expansion in the time domain can be set to be 2. That is to say, in an MS-QP sequence, a single subband corresponds to a narrow-band ZC sequence with a length of L m , and the time domain is repeated twice to form a sequence with a length of 2 L m . According to the corresponding video correspondence described in FIG. 13 , the 2L m -length sequence will only occupy even frequency points after the DFT is performed, so the data sequence can be modulated on the technical frequency points.
  • the detection sequence components and the data symbol components are represented by lines of different thicknesses.
  • multiple repeated transmissions can be performed when the transmitting device sends the communication coordination sequence signal.
  • the receiving device receives the echo signal
  • a cyclic correlation of length N is performed on the echo signal of each sub-block and the MS-QP sequence to form a correlation matrix r k" [n]. Take the nth in the cyclic correlation calculation result of the k" block. The correlation values of the two positions are combined to do qk" point FFT, so as to obtain the RDM for subsequent signal processing.
  • Figure 24 shows the received echo The waveform of the wave signal is shown. For example, the black vertical line represents a certain piece of echo signal. Obviously (a) is composed of multiple echo signals.
  • the performance test of the communication coordination sequence signal involved in the present application can be performed through simulation evaluation.
  • simulation evaluation refer to the terahertz system parameters shown in Table 2.
  • parameter value parameter value Center frequency 300GHz Maximum path loss (one way) -91.5dB communication bandwidth 10GHz Antenna and ambient temperature 285k transmit power 8.5dBm(PA) Receiving equipment noise figure 11dB Transceiver antenna gain 40dBi Ranging range within 3m Maximum ranging range 3m Speed range -20m/s ⁇ 20m/s reflection attenuation 25dB
  • the signal-to-noise ratio of the receiving device for terahertz detection within a detection range of 3m is usually higher than -56.5dB.
  • the phase noise is assumed to obey the symbol-by-symbol walk model during the signal transmission process.
  • the symbol-by-symbol walk model can be, for example, the phase noise ⁇ x1 of a certain symbol period can be expressed as the phase noise ⁇ x1-1 of the previous symbol plus a random increment ⁇ x1 , the increment is subject to ⁇ x1 ⁇ N 2 (0, (0.316°)) random variable.
  • the I/Q imbalance parameter of the receiving device can be calculated by formula 19 and formula 20.
  • the MS-QP sequence and the communication cooperation sequence signal involved in the present application are compared with the narrowband ZC sequence, the wideband ZC sequence and the chirp signal in the simulation.
  • the length of the narrowband ZC sequence is set to 1007
  • the root exponent is set to 503
  • the symbol interval of the narrowband ZC sequence is set to 1.1ns
  • the total sequence length is 1.1077 ⁇ s, which is the same length as the MS-QP sequence
  • the bandwidth is 0.9GHz.
  • the wideband ZC sequence let the length be 11077, the root exponent is 5538, the symbol interval is 0.1ns, the total sequence length is 1.1077 ⁇ s, and the bandwidth is 10GHz.
  • the pulse length is set to 1.1077 ⁇ s and the bandwidth is 10GHz.
  • the transmission is repeated 1024 times for both the MS-QP sequence and its reference sequence.
  • the communication cooperation sequence signal since its length is M' times that of the MS-QP sequence, for the sake of relatively fair comparison, the communication cooperation sequence signal is repeated 1024/M' times.
  • the specific parameters can refer to the values set in FIG. 23 .
  • Figure 25 shows the comparison of ranging performance of MS-QP sequence, communication cooperative sequence signal and narrowband ZC sequence, wideband ZC sequence and chirp signal.
  • the abscissa is the signal noise ratio (signal noise ratio, SNR)
  • the unit is dB
  • the ordinate is the average absolute ranging error, which is used to reflect the sequence ranging performance. It can be seen that, no matter what kind of sequence or signal, as the signal-to-noise ratio continues to increase, the average ranging error declines, and even a cliff-like decline occurs within a certain range of signal-to-noise ratio. Until the ranging error decreases to a certain extent due to the limitation of communication bandwidth, it tends to be stable.
  • the average absolute ranging error is constant at a signal-to-noise ratio of about -55dB, about 0.0035m, reaching centimeter-level ranging accuracy. Due to the limited bandwidth of the narrowband ZC sequence, the constant value of the average absolute ranging error will be higher, about 0.04m. It can be seen that the MS-QP sequence can achieve almost the same ranging accuracy as the equal-broadband ZC sequence, and can also use multi-subband parallel filtering for transmission and reception, which greatly reduces the requirements of the DAC in the transmitting device and the ADC in the receiving device. .
  • the same ranging accuracy as the fade-out detection sequence can be achieved while supporting communication.
  • the minimum signal-to-noise ratio is usually 3dB higher than that of MS-QP sequence.
  • the signal-to-noise ratio is higher than -52dB, the superposition of the data symbol components will not affect the ranging accuracy of the detection sequence components.
  • Figure 26 shows the comparison of the speed measurement performance of the above 5 sequences.
  • the speed measurement performance is usually determined by the length of the detection sequence. Therefore, the speed measurement performance that can be achieved by sequences of the same duration is basically the same.
  • the signal-to-noise ratio is higher than -55dB
  • the average absolute velocity measurement error of different detection sequences can basically reach a constant value.
  • the communication cooperative sequence signal that is, the integrated waveform of communication detection in the figure
  • the average absolute velocity measurement error of 3dB usually needs to be added to reach a constant value.
  • the signal-to-noise ratio is higher than -52dB, the superposition of the data symbol components will not affect the velocity measurement accuracy of the detection sequence components.
  • the detection sequence component and the data symbol component in the communication cooperation sequence signal can be transmitted simultaneously without interference, wherein the time domain spreading parameter M' determines the ratio of the data symbol component and the detection sequence component in the frequency domain resources.
  • M' determines the ratio of the data symbol component and the detection sequence component in the frequency domain resources.
  • the detection sequence component occupies 1/M' of the available frequency domain resources, and increasing M' can improve the transmission efficiency of communication data.
  • M' exceeds a certain threshold, it will affect the accuracy of ranging and speed measurement. Therefore, the value of M' can be dynamically adjusted according to the actual situation to achieve a perfect balance between communication performance and detection performance.
  • Fig. 27 shows the effect of different frequency domain ratios on the mean absolute ranging error.
  • FIG. 29 is a schematic diagram of another integrated device for communication detection provided by an embodiment of the present application.
  • the present application also provides another communication detection integrated device 2900, the device 2900 includes:
  • the sending module 2901 is configured to send a communication coordination sequence signal.
  • the signal generation module 2903 is used to determine the communication coordination sequence signal, the communication coordination sequence signal includes a detection sequence component, and the detection sequence component is an MS-QP sequence with multiple subbands, and each subband includes a narrowband ZC sequence.
  • the receiving module 2902 is configured to receive the echo signal of the communication coordination sequence signal.
  • the signal processing module 2904 is configured to perform fast Fourier transform on the echo signal to determine the radar detection component of the echo signal in the frequency domain.
  • the signal processing module 2904 is further configured to correlate the detection sequence component and the radar detection component to determine at least one correlation peak and time delay information corresponding to the at least one correlation peak.
  • the signal processing module 2904 is further configured to determine at least one target and the at least one target distance according to at least one correlation peak and time delay information corresponding to the at least one correlation peak.
  • the signal generation module 2903 is further configured to: use the optimal phase factor set to perform phase adjustment on each ZC sequence in the MS-QP sequence.
  • the ZC sequence includes: a sequence length L m and a root index p, wherein the root index p satisfies L m represents the length of the ZC sequence on the mth subband, and m is a positive integer greater than or equal to 2.
  • the signal generation module 2903 is further configured to: set a frequency domain guard interval between the ZC sequences of every two adjacent subbands.
  • the communication coordination sequence signal further includes: a data symbol component.
  • the data symbol component includes data sequences of multiple subbands.
  • the signal generation module 2903 is also used to transmit the data sequence on each subband on a plurality of zeroing frequency points on the subband, and the zeroing frequency point is the ZC sequence on the subband to repeat M times in the time domain. can be transmitted.
  • the signal generation module 2903 is further configured to: perform corresponding phase adjustment for the data sequence transmitted on each subband.
  • the signal processing module 2904 is further configured to: one-to-one correspondence between at least one correlation peak and at least one target; target distance.
  • the apparatus 2900 provided by the present application can implement any one of the methods described above in FIG. 1 to FIG. 28 .
  • FIG. 30 is a schematic diagram of yet another integrated device for communication detection provided by an embodiment of the present application.
  • FIG. 30 shows yet another communication detection integrated device 3000 .
  • the communication detection integrated device 3000 may be the communication detection integrated device in the solutions involved in the above-mentioned FIG. 1 to FIG. 29 .
  • the integrated communication detection device 3000 may include a processor 3010, an external memory interface 3020, an internal memory 3021, a universal serial bus (USB) interface 3030, a charging management module 3040, a power management module 3041, a battery 3042, an antenna 1. Antenna 2, mobile communication module 3050, wireless communication module 3060, display screen 3070, etc.
  • a processor 3010 an external memory interface 3020, an internal memory 3021, a universal serial bus (USB) interface 3030, a charging management module 3040, a power management module 3041, a battery 3042, an antenna 1.
  • Antenna 2 mobile communication module 3050, wireless communication module 3060, display screen 3070, etc.
  • the integrated communication detection apparatus 3000 may include more or less components than shown, or combine some components, or separate some components, or arrange different components.
  • the illustrated components may be implemented in hardware, software, or a combination of software and hardware.
  • the processor 3010 , the external memory interface 3020 and the internal memory 3021 may refer to the corresponding descriptions in the processor 301 and the memory 302 in FIG. 3 , which will not be repeated here.
  • the processor 3010 may include one or more interfaces.
  • the interface may include an integrated circuit (inter-integrated circuit, I2C) interface, an integrated circuit built-in audio (inter-integrated circuit sound, I2S) interface, a pulse code modulation (pulse code modulation, PCM) interface, a universal asynchronous transceiver (universal asynchronous transmitter) receiver/transmitter, UART) interface, mobile industry processor interface (MIPI), general-purpose input/output (GPIO) interface, subscriber identity module (SIM) interface, and / or universal serial bus (universal serial bus, USB) interface, etc.
  • I2C integrated circuit
  • I2S integrated circuit built-in audio
  • PCM pulse code modulation
  • PCM pulse code modulation
  • UART universal asynchronous transceiver
  • MIPI mobile industry processor interface
  • GPIO general-purpose input/output
  • SIM subscriber identity module
  • USB universal serial bus
  • the USB interface 3030 is an interface that conforms to the USB standard specification, which can be a Mini USB interface, a Micro USB interface, a USB Type C interface, and the like.
  • the USB interface 3030 can be used to connect a charger to charge the distributed recording device 1700, and can also be used to transmit data between the integrated device 3000 for communication and detection and peripheral devices. It can also be used to connect headphones to play or capture audio through the headphones.
  • the interface can also be used to connect other electronic devices, such as AR devices.
  • the interface connection relationship between the modules illustrated in the embodiment of the present invention is only a schematic illustration, and does not constitute a structural limitation on the integrated communication detection apparatus 3000 .
  • the integrated communication detection device 3000 may also adopt different interface connection manners in the foregoing embodiments, or a combination of multiple interface connection manners.
  • the wireless communication function of the integrated communication detection device 3000 can be realized by the antenna 1, the antenna 2, the mobile communication module 3050, the wireless communication module 3060, the modulation and demodulation processor, the baseband processor, and the like.
  • Antenna 1 and Antenna 2 are used to transmit and receive electromagnetic wave signals.
  • Each antenna in the integrated communication detection device 3000 can be used to cover a single or multiple communication frequency bands. Different antennas can also be reused to improve antenna utilization.
  • the antenna 1 can be multiplexed as a diversity antenna of the wireless local area network. In other embodiments, the antenna may be used in conjunction with a tuning switch.
  • the mobile communication module 3050 can provide a wireless communication solution including 2G/3G/4G/5G/6G, etc. applied on the communication detection integrated device 3000 .
  • the mobile communication module 3050 may include at least one filter, switch, power amplifier, low noise amplifier (LNA) and the like.
  • the mobile communication module 3050 can receive electromagnetic waves from the antenna 1, filter and amplify the received electromagnetic waves, and transmit them to the modulation and demodulation processor for demodulation.
  • the mobile communication module 3050 can also amplify the signal modulated by the modulation and demodulation processor, and then convert it into electromagnetic waves and radiate it out through the antenna 1 .
  • at least part of the functional modules of the mobile communication module 3050 may be provided in the processor 3010 .
  • at least part of the functional modules of the mobile communication module 3050 may be provided in the same device as at least part of the modules of the processor 3010 .
  • the modem processor may include a modulator and a demodulator.
  • the modulator is used to modulate the low frequency baseband signal to be sent into a medium and high frequency signal.
  • the demodulator is used to demodulate the received electromagnetic wave signal into a low frequency baseband signal. Then the demodulator transmits the demodulated low-frequency baseband signal to the baseband processor for processing.
  • the low frequency baseband signal is processed by the baseband processor and passed to the application processor.
  • the wireless communication module 3060 can provide wireless local area networks (WLAN) (such as WiFi network), Bluetooth, global navigation satellite system (GNSS), FM ( frequency modulation, FM), near field communication technology (near field communication, NFC), infrared technology (infrared, IR) and other wireless communication solutions.
  • WLAN wireless local area networks
  • GNSS global navigation satellite system
  • FM frequency modulation, FM
  • NFC near field communication technology
  • infrared technology infrared, IR
  • the wireless communication module 3060 may be one or more devices integrating at least one communication processing module.
  • the wireless communication module 3060 receives electromagnetic waves via the antenna 2 , frequency modulates and filters the electromagnetic wave signals, and sends the processed signals to the processor 3010 .
  • the wireless communication module 3060 can also receive the signal to be sent from the processor 3010 , perform frequency modulation on it, amplify it, and convert it into electromagnetic waves for radiation through the antenna 2 .
  • the antenna 1 of the integrated communication detection apparatus 3000 is coupled to the mobile communication module 3050, and the antenna 2 is coupled to the wireless communication module 3060, so that the integrated communication detection apparatus 3000 can communicate with the network and other devices through wireless communication technology.
  • Display screen 3070 is used to display images, videos, and the like.
  • the display screen 3070 includes a display panel.
  • the display panel can be a liquid crystal display (LCD), an organic light-emitting diode (OLED), an active-matrix organic light-emitting diode or an active-matrix organic light-emitting diode (active-matrix organic light).
  • LED organic light-emitting diode
  • AMOLED organic light-emitting diode
  • FLED flexible light-emitting diode
  • Miniled MicroLed, Micro-oLed
  • quantum dot light-emitting diode quantum dot light emitting diodes, QLED
  • the integrated communication detection device 3000 may include at least one display screen 3070 .
  • the data frame preamble sequence is used for radar detection
  • the direct sequence spread spectrum signal is used for communication detection coordination, etc.
  • there is a high peak-to-average ratio. non-ideal autocorrelation characteristics, limited code length, low data rate and many other problems.
  • the communication cooperation sequence signal involved in the present application subtly divides the ultra-high bandwidth of terahertz into multiple adjacent subbands, and loads the narrowband ZC sequence on different subbands. This makes it possible to obtain good autocorrelation characteristics, and only needs to use a band-pass filter and a low sampling rate ADC to process different subbands to complete broadband signal acquisition, achieve centimeter-level ranging accuracy, and effectively reduce hardware costs.
  • the waveform of the communication coordination sequence signal to be sent is involved, and the detection sequence is expanded in the time domain for multiple times, so that a zero-set frequency point appears in the component spectrum of the expanded detection sequence. Then, the data sequence components are loaded on the zero-setting frequency point, so that the detection and communication functions can be performed simultaneously without interference.
  • the design will not sacrifice the detection signal bandwidth, so it will not affect the ranging resolution.
  • it can also support correlation detection of long sequences with low-complexity algorithms.
  • the detection sequence component part of the communication cooperative sequence signal will not interfere with the data sequence component, and can be used for channel estimation to assist in the reception of communication data.
  • each sub-band transmits the narrow-band ZC sub-sequence, so as to realize the spectrum expansion of the narrow-band sequence.
  • the detection sequence components have good autocorrelation characteristics, can utilize ultra-high bandwidth to achieve high ranging resolution, and only need band-pass filter and low sampling rate ADC for reception and reconstruction.
  • the present application can also multiply the signals of different subbands with different phase factors in the frequency domain. After optimization, the peak-to-average ratio of the sequence can be effectively reduced, and the detection of the sequence is basically not affected. performance.
  • the root index of the narrowband ZC sequence on each subband in the detection sequence component is determined. optimize. After the root exponent is optimized, the high-range image sidelobes caused by the Doppler frequency shift can be gathered near the main peak. At the same time, the exclusion strategy of the adjacent area of the main peak is used, assuming that there is no other target in the vicinity of the target corresponding to the main peak. The probability of being misjudged as a target can effectively reduce the impact of Doppler frequency shift on target detection.
  • non-transitory English: non-transitory
  • the storage medium is non-transitory ( English: non-transitory) media, such as random access memory, read only memory, flash memory, hard disk, solid state disk, magnetic tape (English: magnetic tape), floppy disk (English: floppy disk), optical disc (English: optical disc) and any combination thereof.

Landscapes

  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Remote Sensing (AREA)
  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Signal Processing (AREA)
  • Artificial Intelligence (AREA)
  • Evolutionary Computation (AREA)
  • Radar Systems Or Details Thereof (AREA)

Abstract

一种目标检测方法，方法包括：发送包括检测序列分量的通信协同序列信号（S401）。其中，该检测序列分量为具有多个子带的MS-QP序列，每个子带包括一个窄带ZC序列。接收通信协同序列信号的回波信号（S402）并进行FFT，以在频域内确定雷达检测分量（S403）。对检测序列分量和雷达检测分量做相关，以确定至少一个相关峰及各个相关峰对应的时延信息（S404）。根据至少一个相关峰及各个相关峰对应的时延信息，确定出至少一个目标以及各个目标对应的目标距离。该方法发送的通信协同序列信号具有良好的自相关特性，以便用于目标检测。同时通过多个子带，将发送信息所占的大带宽划分为多个窄带，使得在接收回波信号时可以采用低采样率ADC，以降低硬件成本。

Description

一种目标检测方法及装置
本申请要求于2021年04月02日提交中国国家知识产权局、申请号为202110361611.9、申请名称为“一种目标检测方法及装置”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及通信领域，尤其涉及一种基于太赫兹通信检测一体化波形的目标检测方法及装置。
背景技术
太赫兹频段频谱资源十分丰富，可以提供超高可用带宽，从而可以支持上百吉比特每秒(gigabits per second，Gbps)的高速数据传输。这满足了无线通信高速发展下诸多数据业务对于超高传输速率的需求，可见太赫兹通信或将成为6代(generation)移动通信技术的关键技术之一。除通信功能之外，由于太赫兹频段具有大量可用带宽，其用于目标检测或感知可用有效提升目标测距分辨率，并提升检测精度。而另一方面，由于太赫兹信号的波束极窄，用于雷达检测时能够有效降低多径带来的杂波干扰。因此，太赫兹信号在检测方面的应用前景十分广阔。由于太赫兹通信与检测***均需要占用超宽频谱资源，且制作成本高。因此需要对太赫兹通信检测进行深入研究，例如太赫兹通信检测一体化。太赫兹通信检测一体化通过通信与检测共用硬件资源和频谱资源，可以有效降低设备成本、减小设备尺寸并提升频谱利用率。
目前对于太赫兹通信检测一体化技术存在诸多挑战，首先太赫兹检测需要超高带宽以提升检测精度，因此在接收端需要超高采样率的模数转换器(analog-to-digital converter，ADC)以避免信号失真。显然其制作难度较大，且成本很高。当前太赫兹通信检测一体化研究尚处于探索阶段，当通信与检测功能同时进行时，会产生相互干扰并恶化***性能。因此急需为太赫兹通信检测一体化***提出新的方案以解决上述问题。
发明内容
本申请实施例提供了一种目标检测方法，通过采用具有多个子带的多子带准完美(multi-band quasi-perfect，MS-QP)序列作为发送的通信协同序列信号，依托MS-QP序列具有的多个子带，将较宽的带宽划分为多个窄带。其中，每个子带对应一个窄带。同时在每个自带上传输窄带扎道夫-初(zadoff-chu，ZC)序列，从而保证了通信协同序列信号具有较强的自相关特性，以便当接收到通信协同序列信号产生的回波信号后，可以根据回波信号确定雷达检测分量，并通过对检测序列分量和雷达检测分量做相关实现对目标的检测。由于发送的通信协同序列信号将较宽的带宽划分为多个窄带，因此在接收回波信号时可以避免采用成本较高的高采样ADC，从而降低硬件成本。
第一方面，提供了一种目标检测方法，方法包括：发送通信协同序列信号。其中，通信协同序列信号中可以包括检测序列分量。该检测序列分量为具有多个子带的MS-QP序列，每个子带可以包括一个窄带ZC序列。之后，可以接收通信协同序列信号的回波信号，并对回波信号进行快速傅里叶变换(fast fourier transformation，FFT)，以在频域内确定该回波信号的雷达检测分量。之后，对检测序列分量和雷达检测分量做相关，以确定至少一个相关峰及至少一个相关峰对应的时延信息。然后，可以根据至少一个相关峰及至少一个相关峰对应的时延信息，确定出至少一个目标以及各个目标对应的目标距离。本申请通过具有多个子带的MS-QP序列中每个子带包含一个窄带ZC序列，从而保证了发送的通信协同序列信号可以具有良好的自相关特性，以便用于目标检测。同时通过多个子带，将发送的信息所占的大带宽划分为多个窄带，从而在接收回波信号时可以采用低采样率ADC，以降低硬件成本。
在一个可能的实施方式中，方法还可以包括：采用最优相位因子集合对MS-QP序列中的各个ZC序列进行相位调整。本申请还可以对MS-QP序列中的各个ZC序列进行相位调整，从而减小信号的峰均比。
在一个可能的实施方式中，ZC序列可以包括：序列长度L m和根指数p。其中，根指数p满足
Figure PCTCN2022083456-appb-000001
L m表示第m个子带上ZC序列的长度，m为大于或等于2的正整数。本申请通过对ZC序列的根指数进行优化，从而可以有效减小多普勒频移问题，使得在进行目标检测时由于多普勒产生的高旁瓣可以尽量集中在相关峰附近，从而避免误判目标。
在一个可能的实施方式中，方法还可以包括：每两个相邻子带的ZC序列之间具有频域保护间隔(guard interval，GI)。本申请通过在相邻子带之间设置频域保护间隔，从而避免不同子带之间产生相互干扰，以及避免解码时子带衔接部分的数据丢失。
在一个可能的实施方式中，通信协同序列信号还包括：数据符号分量。其中，数据符号分量可以包括多个子带的数据序列，每个子带上的数据序列在子带上的多个置零频点上进行传输。置零频点为该子带上的ZC序列在时域上进行M次重传得到。本申请通过对每个子带上的ZC序列进行多次重传，从而在频域上可以出现多个置零频点，并在置零频点上传输数字符号分量，使得本申请可以同时实现通信和检测两种功能，并且相互之间不会存在干扰。
在一个可能的实施方式中，方法还可以包括：针对每个子带上传输的数据序列进行相应的相位调整。本申请还可以对数据序列进行相应的相位调整，以便可以在对应的置零频点上进行传输。
在一个可能的实施方式中，根据至少一个相关峰及至少一个相关峰对应的时延信息，确定至少一个目标以及至少一个目标距离，可以包括：至少一个相关峰与至少一个目标一一对应，以便可以通过相关峰确定对应的目标。以及，根据至少一个相关峰对应的时延信息确定与各个相关峰对应的目标的目标距离。
第二方面，提供了一种目标检测装置，装置包括：发送器，用于发送通信协同序列信号，其中，通信协同序列信号包括检测序列分量，检测序列分量为具有多个子带的MS-QP序列，每个子带包括一个窄带ZC序列；接收器，用于接收通信协同序列信号的回波信号；处理器用于与存储器耦合，以及读取并执行存储在存储器中的指令；当处理器运行时执行指令，使得处理器用于对回波信号进行快速傅里叶变换，以在频域内确定回波信号的雷达 检测分量；对检测序列分量和雷达检测分量做相关，确定至少一个相关峰及至少一个相关峰对应的时延信息；根据至少一个相关峰及至少一个相关峰对应的时延信息，确定至少一个目标以及至少一个目标距离。本申请通过具有多个子带的MS-QP序列中每个子带包含一个窄带ZC序列，从而保证了发送的通信协同序列信号可以具有良好的自相关特性，以便用于目标检测。同时通过多个子带，将发送的信息所占的大带宽划分为多个窄带，从而在接收回波信号时可以采用低采样率ADC，以降低硬件成本。
在一个可能的实施方式中，处理器还用于：采用最优相位因子集合对MS-QP序列中的各个ZC序列进行相位调整。本申请还可以对MS-QP序列中的各个ZC序列进行相位调整，从而减小信号的峰均比。
在一个可能的实施方式中，ZC序列包括：序列长度L m和根指数p，其中，根指数p满足
Figure PCTCN2022083456-appb-000002
L m表示第m个子带上ZC序列的长度，m为大于或等于2的正整数。本申请通过对ZC序列的根指数进行优化，从而可以有效减小多普勒频移问题，使得在进行目标检测时由于多普勒产生的高旁瓣可以尽量集中在相关峰附近，从而避免误判目标。
在一个可能的实施方式中，处理器还用于：每两个相邻子带的ZC序列之间具有频域保护间隔。本申请通过在相邻子带之间设置频域保护间隔，从而避免不同子带之间产生相互干扰，以及避免解码时子带衔接部分的数据丢失。
在一个可能的实施方式中，通信协同序列信号还包括：数据符号分量。其中，数据符号分量包括多个子带的数据序列，每个子带上的数据序列在子带上的多个置零频点上进行传输，置零频点为子带上的ZC序列在时域上进行M次重传得到。本申请通过对每个子带上的ZC序列进行多次重传，从而在频域上可以出现多个置零频点，并在置零频点上传输数字符号分量，使得本申请可以同时实现通信和检测两种功能，并且相互之间不会存在干扰。
在一个可能的实施方式中，处理器还用于：针对每个子带上传输的数据序列进行相应的相位调整。本申请还可以对数据序列进行相应的相位调整，以便可以在对应的置零频点上进行传输。
在一个可能的实施方式中，处理器还用于：至少一个相关峰与至少一个目标一一对应，以便可以通过相关峰确定对应的目标。以及，根据至少一个相关峰对应的时延信息确定与至少一个相关峰对应的目标的目标距离。
第三方面，提供了一种计算机可读存储介质，计算机可读存储介质中存储有指令，当指令在终端上运行时，使得终端执行第一方面中的任意一项方法。
第四方面，提供了一种包含指令的计算机设备，当其在终端上运行时，使得终端执行第一方面中的任意一项方法。
第五方面，提供了一种包含指令的计算机程序产品，当其在计算机上运行时，使得计算机执行第一方面中的任意一项方法。
本申请公开了一种目标检测方法及装置，通过采用具有多个子带的MS-QP序列作为发送的通信协同序列信号，不仅完美继承了ZC序列的自相关特性，还可以将通信所需的宽带划分为多个窄带进行传输，从而在接收回波信号时可以避免采用成本较高的高采样ADC并降低硬件成本。
附图说明
图1为本申请实施例提供的一种应用场景示意图；
图2为本申请实施例提供的一种通信检测一体化装置的结构示意图；
图3为本申请实施例提供的一种通信检测一体化装置示意图；
图4为本申请实施例提供的一种通信检测一体化方法流程图；
图5为本申请实施例提供的一种发送设备频谱搬移示意图；
图6为本申请实施例提供的另一种发送设备频谱搬移示意图；
图7为本申请实施例提供的一种MS-QP序列时域表示示意图；
图8为本申请实施例提供的一种MS-QP序列频域表示示意图；
图9为一种窄带ZC序列自相关特性示意图；
图10为本申请实施例提供的一种MS-QP序列自相关特性示意图；
图11为本申请实施例提供的一种多子带通信***示意图；
图12为本申请实施例提供的一种MS-QP序列结构示意图；
图13为本申请实施例提供的一种经过时域扩展后的MS-QP序列频谱示意图；
图14为本申请实施例提供的一种单子带通信协同序列信号示意图；
图15为一种硬件失配影响的距离像示意图；
图16为一种多普勒频移影响的距离像示意图；
图17为本申请实施例提供的一种窄带ZC序列自相关变化曲线示意图；
图18为本申请实施例提供的一种根指数优化前距离像示意图；
图19为本申请实施例提供的一种根指数优化后距离像示意图；
图20为本申请实施例提供的一种接收设备回波信号构建示意图；
图21为本申请实施例提供的一种第m子带内接收设备信号处理示意图；
图22为本申请实施例提供的一种回波信号确定DRM示意图；
图23为本申请实施例提供的一种通信协同序列信号单子带频谱示意图；
图24为本申请实施例提供的一种DRM生成过程信号波形示意图；
图25为本申请实施例提供的一种序列测距性能对比示意图；
图26为本申请实施例提供的一种序列测速性能对比示意图；
图27为本申请实施例提供的一种频域占比与测距性能关系示意图；
图28为本申请实施例提供的一种频域占比与测速性能关系示意图；
图29为本申请实施例提供的另一种通信检测一体化装置示意图；
图30为本申请实施例提供的又一种通信检测一体化装置示意图。
具体实施方式
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行描述。
本申请主要应用于目标检测的场景，例如图1所示出的，通信检测一体化装置100的发送设备110发出发送信号。该发送信号经过一定距离后与目标200相遇，之后发送信号被目标200反射后形成回波信号。该回波信号再次经过相同的距离后被通信检测一体化装置100的接收设备120接收。通过接收设备120对接收到的回波信号进行解析后从而确定出是否存在目标200以及目标200与通信检测一体化装置100之间的距离。当然，图1 中示出的通信检测一体化装置100是一种包括了发送设备110和接收设备120的一体化装置，在其它的场景中，发送设备110和接收设备120还可以是两个独立的装置。显然，当发送设备110和接收设备120作为一个整体时，其可以共用同一组硬件资源，从而降低制造成本。该场景中的目标200以某个用户作为示例进行说明，可以理解的是，目标200还可以是其它任意物体，本申请并不做出限定。
通常情况下，为了降低制造成本大多数情况下都采用具有发送设备110和接收设备120的通信检测一体化装置100进行通信和检测，该通信检测一体化装置100上通常会运行通信检测协同***以便同时进行通信和检测。对于采用太赫兹的通信检测协同***，其通信与检测的发送设备110和接收设备120共用一套硬件资源，发送设备110和接收设备120位置可以近似看作是相同的，且共用本机振荡器(可简称本振)。可以理解的是，在一些情况下，发送设备110也可以称为信号发送端，而接收设备120还可以称为雷达检测端。在一些例子中，本振可以采用电感电容(inductor capacitor，LC)振荡器等。由于太赫兹信号传播的衰减较大，因此图1所示的场景通常考虑短距离目标检测。为了有可以有效对抗传播衰减，该通信检测协同***通常可以采用高增益定向天线进行信号传输，产生极窄波束，使得该场景下可以考虑视距链路以及忽略多径效应。其中，视距链路即表示通信检测一体化装置100与目标200直线距离之间传输的信号，而忽略发送信号可能经过目标200反射后而从其它方向上传递的回波信号。
可以理解的是，尽管发送信号的波束很窄，但在一些场景下通信检测协同***仍然可以对一个或多个目标进行检测，例如在全息视频会议中，为追踪人体动作姿态对变化，需要进行人体姿态检测。其中，对于不同的身体部位即可以看作为不同的目标。正如图1中，通信检测一体化装置100发送出发送信号，对于图1中右侧示出的用户200可以持有终端设备以便接收发送信号。当然，经过该用户200后，部分发送信号会被反射并形成回波信号，该回波信号会被通信检测一体化装置100接收。通信检测一体化装置100接收到回波信号后，可以对该回波信号进行分析从而实现对用户姿态动作检测。
如图2所示，则示出了一种通信检测一体化装置的结构示意图。可以看出，该通信检测一体化装置100中可以包括发送设备110和接收设备120。其中，发送设备110中首先经过编码调制模块111产生需要传输的基带信号，并将该基带信号发送至数模转换器(digital-to-analog converter，DAC)112将数字形式的基带信号转换为模拟信号。转换后的模拟信号经过上变频模块113将频率提升至太赫兹频段后，再经高功率放大器114放大发射功率后形成发送信号，最终经过发送天线(transport antenna，TX ANT)115将发送信号发出。同时，通信检测一体化装置100中的接收设备120则可以通过接收天线(receive antenna，RX ANT)121接收发送信号被目标200反射后形成的回波信号。之后，将接收到的回波信号经过低噪声放大器122进行处理，再经下变频模块123将处理后的回波信号的频率降低，再通过ADC将降低频率的回波信号从频域转换到时域，从而形成回波信号对应的基带信号。接收设备120可以将回波信号对应的基带信号输入至雷达检测模块125中进行目标的检测，以及将回波信号对应的基带信号输入至解调解码模块126中以获取传递的信息。
对于发送信号和回波信号的波形，在传统的检测波形中大多可以分为正交频分复用(orthogonal frequency division multiplexing，OFDM)波形和单载波传输波形。而在通信检 测一体化领域，目前通常采用较低频的通信检测协同***，例如使用OFDM信号、循环前缀-单载波(cyclic prefixed single carrier，CP-SC)、数据帧结构中的前导序列以及直接序列扩频信号等进行通信检测的协同。
对于OFDM波形的优势在于其可以同时用于通信与检测，并且基于最大似然的距离/速度检测方法对多普勒频移的鲁棒性较好。当使用OFDM符号作为发送信号时，其频域可以表示为A[k]，其中k＝1，2，...，N’，N’为正整数，可表示为OFDM符号的子载波个数。A[k]为第k个子载波上的数据符号。可以理解的是，A[k]的取值可以根据调制方案从预先设定的星座图中进行选取。A[k]经过快速傅里叶逆变换(invert fast fourier transformation，IFFT)后可以得到时域表示a[n’]，其中n’＝1，2，...，N’，n’为在时域上的采样点序号，采样点与子载波的个数可以相同。可以理解的是，n’与k的取值范围是相同的。然后a[n’]再经过发送天线发出。该OFDM的时域信号发送至通信接收端，其中一部分的能量被通信接收端接收用于信号解调，而另一部分则经过通信接收端表面反射，形成回波信号并传递至发送端。其中，通信接收端可以例如图1中的目标200。当发送端采用通信检测一体化装置时，则可以被同位置的接收设备(或称雷达接收端)接收，其接收到的回波信号可以表示为y[n’]。对于接收设备目前通常可以采用两种检测算法对y[n’]进行检测。第一种方式是介于通信检测一体化装置中的接收设备与发送设备共享一套硬件，因此接收设备已知发送的a[n’]。因此将y[n’]与a[n’]做相关，可以得到距离像。然后分析该距离像中相关峰的位置并估计出目标距离。此方式检测算法操作相对简单，但是a[n’]的自相关特性往往不理想，OFDM波形只有当频域符号恒模时，时域序列才具有理想自相关特性，尤其在a[n’]随机的情况下，其自相关特性得不到相应保障，从而影响到检测性能的提升。而第二种方式是将y[n’]做离散傅里叶变换(discrete fourier transform，DFT)得到频域上的Y[k]。然后根据
Figure PCTCN2022083456-appb-000003
可以得到信道冲击响应的频域响应。之后，再对I[k]做反离散傅里叶变换(inverse discrete fourier transform，IDFT)以得到用于雷达检测的距离像。对于该方式，通过对每个子带上对比接收信号与发送信号，使得距离像的形状不再受到发送信号随机性的影响。但是由于计算频域响应时采用除发送信号的方式会造成噪声的放大，对低信噪比下的检测性能会有一定限制。对于OFDM信号其峰均比较高，在经过太赫兹功率放大后将产生严重非线性效应。对于太赫兹频段的OFDM***其带宽大，因此需要超高采样率的ADC进行采样，显然对硬件要求较高。
而对于单载波信号，其峰均比通常比较低，可有效减小功放非线性效应，保证较好的输出功率。其中，CP-SC是采用了循环前缀(cyclic prefixed，CP)的单载波信号。目前一些方案中，采用了连续W’组传统CP-SC信号作为发送信号，每组信号的数据部分可以记作d w’[n″]，其中w’＝0，1，...，W’-1，n″＝0，1，...，N″-1，W’、N″为正整数。N″可以表示每组信号的数据个数，d w’[n″]则表示第w组的第n″个数据。其中，每组信号d w’[n″]可以使用预设的星座图进行调制，例如可以采用正交相移键控(quadrature phase-shift keying，QPSK)进行调制。当W’组CP-SC信号经过调制后可以作为发送信号由发送天线发出，以便目标进行接收。可以理解的是，发送信号的一部分能量被目标接收并进行信号解调，同时，另一部分能量则被目标表面反射形成回波信号。对于回波信号，则可以被与发送设备相同位置的接收设备所接收，其接收到的回波信号则可以表示为y w’[n″]。为了可以进行目标距离和目标相对移动速度进行检测，接收设备对每组的接收数据块y[n″]和对应的发送数据d[n″]做长度为N″的循环相关，以得到W’组长度为N″的相关结果，可以记作r(n″，w’)。之后，可以对每个相关位置对应的W’个做点Z点FFT。在一些例子中，Z通常可以取值为qW’，其中q为正整数。当经过FFT后可以得到距离-多普勒矩阵(range-doppler matrix，RDM)， 可记作R(n″，k’)。可以理解的是k’的取值范围与w’相同。并通过遍历R(n″，k’)确定最大值对应的横纵坐标。并根据对坐标进行简单换算得到目标距离和目标相对移动速度。在一些例子中，常用的单载波传输波形可以包括线性调频(linear frequency modulated，LFM)信号、伪随机扩频码、完美序列等。其中，伪随机扩频码例如可以是最长线性移位寄存器序列(可简称m序列)、戈尔德(gold)序列等。完美序列例如可以采用ZC序列、弗兰克(frank)序列等。可以理解的是，CP-SC信号直接用于雷达检测时实现较为简单，且峰均比较低。但由于调制符号的随机性，使得序列的良好自相关特性无法得到保障，从而导致检测性能的受限。
目前另一些方案中，当采用单载波前导序列作为发送信号时，其数据帧结构中包含有短时训练域(short training field，STF)和信道估计域(channel estimation field，CEF)。在通信方面，STF可以用于同步和频偏估计，CEF可以用于信道估计。单载波前导序列可以是由多组具有良好自相关特性的戈莱(golay)互补序列构成，因此适用于雷达检测。在一些例子中，发送设备由发送天线将发送信号发送至目标，发送信号的一部分能量被目标接收并进行信号解调，同时，另一部分能量则被目标表面反射形成回波信号。对于回波信号，则可以被与发送设备相同位置的接收设备所接收，并根据回波信号中的前导序列进行目标检测。在检测过程中可以将STF和CEF作为一个整体，对发送时发送信号的前导序列与接收信号的前导序列做互相关。通过查找互相关峰确定是否存在目标以及确定目标回波时延，并根据目标回波时延进一步换算得到目标距离。例如根据目标回波时延与光速确定出目标距离。同时还可以利用无线局域网(wireless local area network，WLAN)的载波频偏估计方法进行估计，从而确定目标相对移动速度。其具体实现过程可以参考现有方式进行，本申请在此不再赘述。但是由于单载波前导序列的长度通常较短，因此无法抵抗太赫兹信道极高的路径损耗，换句话说单载波前导序列无法抵抗接收设备的极低信噪比。
当然，在又一些方案中，还可以采用m序列对单载波通信信号进行直接序列扩频。其中，单载波通信信号例如可以是相移键控(phase shift keying，PSK)符号序列。在一个例子中，可以设定足够大的扩频比，从而使得扩频后的序列可以具有伪随机码m序列的良好自相关特性。例如，假设发送信号采用U阶PSK进行调制。U表示PSK的阶数，一般为正整数。其中，假设扩频前的数据符号长度为N”，其中第i个数据符号的对应比特可以记作
Figure PCTCN2022083456-appb-000004
u为0/1比特(bit)。假设m序列的长度为L’，或称扩频比为L’，则第
Figure PCTCN2022083456-appb-000005
个比特可以记为
Figure PCTCN2022083456-appb-000006
其中，L’可以为正整数，
Figure PCTCN2022083456-appb-000007
之后，可以使用m序列中的每个元素
Figure PCTCN2022083456-appb-000008
与
Figure PCTCN2022083456-appb-000009
中的各个元素进行模2加法，即
Figure PCTCN2022083456-appb-000010
并将所得数据流经调制器得到经过直接序列扩频的L’个M-PSK符号。同理，可以对每个数据符号进行上述扩频操作，以生成长度为N”L’的直接扩频序列信号，并作为发射信号。在一些例子中，发送设备由发送天线将发送信号发送至目标，发送信号的一部分能量被目标接收并进行信号解调，同时，另一部分能量则被目标表面反射形成回波信号。对于回波信号，则可以被与发送设备相同位置的接收设备所接收，并根据回波信号进行目标检测。相应过程可以参考现有基于时域的方式，本申请在此不再赘述。显然，扩频信号具有良好的自相关特性，但是当扩频比设定较高时，则会使得通信速率降低。同时还会导致码长过大并进一步导致多普勒频移的影响加重。
通过上述方案可以看出，在太赫兹通信检测协同***中，当通信和检测两个功能同时进行时，通信信号和检测信号(或称雷达信号)可能产生干扰，严重影响***的整体性能。同时为了满足检测的高分辨率，采用太赫兹的通信检测协同***需要使用超高信号带 宽，因此接收设备也需要极高采样率，从而导致ADC制作难度大且成本高。
因此，本申请提供了一种目标检测方法，通过采用具有多个子带的MS-QP序列作为发送的通信协同序列信号，其中，每个子带可以包括一个窄带ZC序列。之后，在接收通信协同序列信号反射形成的回波信号后对回波信号进行FFT，以在频域内确定该回波信号的雷达检测分量。对检测序列分量和雷达检测分量做相关，以确定至少一个相关峰及至少一个相关峰对应的时延信息，并确定出至少一个目标以及各个目标对应的目标距离。本申请所发送的通信协同序列信号可以保证具有良好的自相关特性，以便用于目标检测。同时通过多个子带，将发送的信息所占的大带宽划分为多个窄带，从而在接收回波信号时可以采用低采样率ADC，以降低硬件成本。
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行详细描述。
图3为本申请实施例提供的一种通信检测一体化装置示意图。
如图3所示，本申请提供了一种通信检测一体化装置300。该通信检测一体化装置300可以应用于进行通信和检测时的信号发送和接收场景，可以包括：一个或多个处理器301、一个或多个存储器302、发送器303、接收器304、一个或多个天线305以及总线306。通信检测一体化装置300中的处理器301、存储器302、发送器303和接收器304可以通过总线306建立通信连接。发送器303和接收器304还可以和天线305相连接，以便发送器303通过天线305发出发送数据，以及接收器304通过天线305接收回波信号。当然，在一些例子中，发送器303和接收器304可以共用一套硬件设备。可以理解的是，发送器303即图1和图2中的发送设备110，接收器304即图1和图2中的接收设备120。
可以理解的是，本发明实施例示意的结构并不构成对通信检测一体化装置300的具体限定。通信检测一体化装置300可以包括比图示更多或更少的部件，或者组合某些部件，或者拆分某些部件，或者不同的部件布置。图示的部件可以以硬件，软件或软件和硬件的组合实现。
其中，处理器301可以是高级精简指令集处理器(advanced reduced instruction set computing machines，ARM)、X86、无内部互锁流水级的微处理器(microprocessor without interlocked piped stages，MIPS)等架构的处理器。处理器301可以包括一个或多个处理单元，例如：应用处理器(application processor，AP)，调制解调处理器，GPU，图像信号处理器(image signal processor，ISP)，控制器，视频编解码器，数字信号处理器(digital signal processor，DSP)，基带处理器和/或神经网络处理器(neural-network processing unit，NPU)等。其中，不同的处理单元可以是独立的器件，也可以集成在一个或多个处理器中。
存储器302用于存储指令和数据。在一些实施例中，存储器302为高速缓冲存储器。该存储器302可以保存处理器301刚用过或循环使用的指令或数据。如果处理器301需要再次使用该指令或数据，可从存储器302中直接调用。避免了重复存取，减少了处理器301的等待时间，因而提高了***的效率。在一个例子中，存储器302可以包括内存，内存中可以存储有操作***、镜像文件等。存储器302还可以包括辅助存储器或称为外存储器，例如可以是不可拆卸存储器或可拆卸存储卡等等。
其中，内部存储器可以用于存储计算机可执行程序代码，可执行程序代码包括指令。内部存储器可以包括高速随机存取存储器，还可以包括非易失性存储器，例如至少一个磁盘存储器件，闪存器件，通用闪存存储器(universal flash storage，UFS)等。处理器301 通过运行存储在存储器的指令，执行通信检测一体化装置300的各种功能应用以及数据处理。
可以理解的是，通信检测一体化装置300可以是终端设备，例如包括但不限于手机、智能电视、智能音响、可穿戴设备、平板电脑、桌面型计算机、手持计算机、笔记本电脑、超级移动个人计算机(ultra-mobile personal computer，UMPC)、上网本、个人数字助理(personal digitalassistant，PDA)、膝上型计算机(laptop)、移动电脑、增强现实(augmented reality，AR)设备、虚拟现实(virtual reality，VR)设备、人工智能(artificial intelligence，AI)设备、车载设备、智能家居设备和/或智慧城市设备等任意终端设备或便携式终端设备。当然，在另一些例子中，通信检测一体化装置300也可以是网络设备，例如无线基站、中继器、网络节点等。
图4为本申请实施例提供的一种通信检测一体化方法流程图。
如图4所示，本申请还提供了一种通信检测一体化方法，该方法可以应用于图3所示出的通信检测一体化装置300。该方法包括以下步骤：
S401，发送通信协同序列信号。
通信检测一体化装置300中的发送器303通过天线305发送通信协同序列信号。其中，通信协同序列信号中包括检测序列分量。该检测序列分量为具有多个子带的MS-QP序列，每个子带对应一个窄带ZC序列。
在一个例子中，为了解决太赫兹通信检测协同***对于采样率要求高、ADC制作成本高且难度大等问题，借助ZC序列构造MS-QP序列用于通信协同序列信号中的检测序列分量。由于完美序列长短码均具有理想的自相关特性，因此完美序列非常适用于太赫兹频段的目标检测。因此本申请的MS-QP序列通过ZC序列进行构造，从而确保了MS-QP序列也具有良好的检测性能。
例如，MS-QP序列可以是采用多个窄带ZC序列进行构造的。可以理解的是，对于MS-QP序列的构造可以通过处理器301来实现。首先对多个窄带ZC序列做DFT到频域。其中，对于，多个窄带ZC序列中的每个窄带ZC序列可以相同或不同。之后，对多个窄带ZC序列的频谱进行搬移并进行拼接，每个窄带ZC序列的频谱构成了多个相邻子带，并形成超宽带频谱。可以理解的是，每个子带均对应一个窄带ZC序列。显然，相比于现有方式，若传输的数据所需带宽很宽时，则需要高采样率的ADC进行采样接收，而构造高采样率ADC难度大且成本高。而本申请中，将发送信号所需的大宽带划分为由多个窄带拼接而成。因此避免了接收时采用高采样率的ADC进行采样接收。然后，处理器301对超宽带频谱做IDFT以得到MS-QP序列的时域表示。通常情况下，为了提高目标检测的测距分辨率，在发送发送信号时会需要超高带宽，可以理解的是，当发送信号的带宽越高，对应的测距分辨率则精度越高。因此当MS-QP序列具有超高信号带宽时，可以有效提升测距分辨率。
在一个例子中，为了可以在进行信号滤波时避免相邻子带之间产生干扰，则对于每两个相邻子带之间，即相邻子带上的ZC序列之间，在进行频谱搬移和拼接时***频域保护间隔L G，使得相邻的两个子带之间在频域上具有一定的间隔，避免相邻子带之间产生干扰。
例如图5则示出了在硬件实现上的一种发送设备频谱搬移示意图，其中，b m[n]表示 第m个子带上的窄带ZC序列，m＝0，1，...，M-1，M为大于或等于2的正整数。可以理解的是，在一些极端条件下，例如发送信息所需带宽很小，此时MS-QP序列中仅包含一个子带来传输窄带ZC序列从而完成数据传输时，则M的数值可以为1。b m[n]的长度为L m，f m表示第m个子带的中心频率。对于每个窄带ZC序列其对应的是与符号间隔记为T m。x[n]则表示为进行了频域扩展构造后的MS-QP序列，其长度为ML G+∑L m。当然，可以理解的是，若没有设置频域保护间隔L G，则长度可以为∑L m。对于进行了频域扩展构造后的MS-QP序列，其对应的时域符号间隔可以记作T M，且
Figure PCTCN2022083456-appb-000011
图5中发送设备输出的模拟信号记为x(t)。可以看出，生成模拟信号x(t)的过程可以是针对每个子带，先将b m[n]通过成形滤波器后，再经由DAC将数字信号转换为模拟型号。其中，成形滤波器主要用于将信号的时钟速率由码元速率转换为模数(digital-to-analog，DA)采样速率，以便后续输入DAC中进行处理。对于每个子带，当转换为模拟信号后再进行频谱搬移，即图中每个子带均通过乘法器乘以
Figure PCTCN2022083456-appb-000012
实现频谱搬移。其中，j为虚数单位，t表示时间。可以理解的是，f’ m与f m均表示为第m个子带的中心频率，其中f m主要用于在数字域对各个离散的点进行计算时使用，而f’ m主要用于模拟信号域实现频谱搬移。然后在将多个子带搬移后的模拟信号采用加法器进行叠加，形成输出的模拟信号x(t)。显然，上述方式是通过在模拟信号侧进行频谱搬移。当然在另一些情况下，还可以在数字信号侧进行频谱搬移。
例如图6所示的另一种发送设备频谱搬移示意图。可以看出，该方式首先对不同子带b m[n]通过成形滤波器，对每个子带上的数字信号进行频谱搬移，即图中每个子带均通过乘法器乘以
Figure PCTCN2022083456-appb-000013
实现频谱搬移。之后，采用加法器将频谱搬移后的数字信号进行叠加得到待发送的数字信号x[n]。之后通过DAC将数字信号x[n]转换为模拟信号后将模拟信号提升至太赫兹频段，如利用乘法器乘以
Figure PCTCN2022083456-appb-000014
实现将模拟信号提升至太赫兹频段以得到发送信号x(t)。其中，f c可以表示为发送信号所需带宽的中心频率。显然，由于该方式是在数字信号侧频谱搬移并叠加后，在通过DAC将数字信号转换为模拟信号，因此图6所示的DAC需要借助高采样DAC完成超宽带信号的数模转换。可以理解为此时时域符号间隔为T M。而相对于图5而言，其是在进行频谱搬移前通过DAC将数字信号转换为模拟信号，因此图5中的DAC可以选择多个低采样DAC实现，此时时域符号间隔仅为T m。
在一个例子中，可以基于图5和图6所提供的方式确定数字信号x[n]的表达式，其中，数字信号x[n]即为上述MS-QP序列。首先对于每个子带上的窄带ZC序列b m[n]可以通过公式1表示。
Figure PCTCN2022083456-appb-000015
其中，n”’用于表示该窄带ZC序列的序号，n”’＝0，1，...，L m-1。exp表示以e为底的指数函数，p m为根指数，L m可以为奇数，且满足gcd(p m，L m)＝1。gcd表示为最大公约数。可以假设b m[n”’]的频域表示为B m[k]，其中k＝0，1，...，L m-1。若考虑设置频域保护间隔，则窄带ZC序列的序列总长可以为N＝ML G+∑L m。则x[n](n＝0，1，...，N-1)可以表示为
Figure PCTCN2022083456-appb-000016
其中，n用于表示MS-QP序列的序号。对于公式2还可以进一步变形为公式3。
Figure PCTCN2022083456-appb-000017
其中，
Figure PCTCN2022083456-appb-000018
可以理解的是，若未设置频域保护间隔，则上述公式中的L G可以为0。当然，当设置有频域保护间隔时，则x[n]虽然由多个窄带ZC序列组合而成，但是并不是真正的完美序列，不过其仍然具有良好的自相关特性。这是由于序列具有完美自相关特性等价于序列的频域幅度谱为横模。而当相邻子带之间设置的频域保护间隔占发送信号所占整体带宽的比重越低，则x[n]的频域幅度谱越接近于横模、自相关特性越好。
图7则示出了一种MS-QP序列时域表示示意图。其中，每个窄带ZC序列长度为1007，且一共设置4个子带为例，生成的MS-QP序列x[n]对应的时域图如图7所示。可以看出，在时域上不再满足横模特性，且仍然具有一定的峰均比。可以理解的是，峰均比可以通过时域上信号的抖动幅度体现，例如峰均比越高，信号幅度抖动越大(也可以称越剧烈)。而通过图8所示出的一种MS-QP序列频域表示示意图可以看出，在频域上，x[n]频谱由多个窄带ZC序列频谱构成。其中，图8中的每个方框示意出了一个窄带ZC序列的频谱。显然，x[n]的频谱接近横模，因此说明x[n]具有良好的自相关特性。其中，信号的幅度值随着频率的变化保持不变，则表示在该频段上横模。可以看出，由于设置了频域保护间隔，因此x[n]的频谱上会出现某些频率上幅度值为0的现象。所以x[n]的频谱接近横模，而并非真正意义上的横模。
图9则示出了窄带ZC序列的自相关特性示意图。可以看出，对于窄带ZC序列其具有完美的自相关特性。当然对于其它完美序列也是如此。而图10则示出了本申请的MS-QP序列自相关特性示意图。可以看出，MS-QP序列也具有准完美的自相关特征，但是相比窄带ZC序列，则会在主峰的附近产生小幅度旁瓣，如图10中在时延接近0以及其它非0时，均会具有较小的幅度。
则在另一些例子中，通过图7由于上述的MS-QP序列不在满足窄带ZC序列的横模特性，因此可以对MS-QP序列做出进一步的改进，从而减小信号的峰均比。考虑到MS-QP序列的相频特性基本上不会影响其自相关特性，因此可以考虑调整序列的相位以达到减小信号峰均比的目的。例如，可以在MS-QP序列的不同子带上分别乘不同的相位因子。在一个例子中，可以预先设定有限集合
Figure PCTCN2022083456-appb-000019
其中，
Figure PCTCN2022083456-appb-000020
中包含有多个相位因子。例如，当子带数量为M时，可以预先从有限集合
Figure PCTCN2022083456-appb-000021
中选择出M个相位因子作为相位因子集合。可以从有限集合
Figure PCTCN2022083456-appb-000022
中的遍历出所有可能的相位因子集合，并计算采用该相位因子集合的MS-QP序列的峰均比。选择峰均比最小的相位因子集合作为最优相位因子集合。并利用该最优相位因子集合对MS-QP序列中每个子带上的窄带ZC序列进行相位调整。
假设第m个子带的相位因子为
Figure PCTCN2022083456-appb-000023
则公式3可以进一步表示为公式5
Figure PCTCN2022083456-appb-000024
其中，相位因子满足：
Figure PCTCN2022083456-appb-000025
其中，E()表示期望，|| ||表示模，argmin表示当
Figure PCTCN2022083456-appb-000026
取最小值时
Figure PCTCN2022083456-appb-000027
的取值。
因此，基于公式5，公式1还可以等效为公式7。
Figure PCTCN2022083456-appb-000028
则基于公式5、公式7，其中
Figure PCTCN2022083456-appb-000029
L m为第m个窄带ZC序列的长度且设为奇数，p m为第m个窄带ZC序列的根指数且满足gcd(p m，L m)＝1。同时在结合公式6，以得到峰均比更小的MS-QP序列。
当然，在一些例子中，对于上述涉及到的MS-QP序列不仅可以应用于支持天盒子***高精度目标检测的通信检测协同***中，还可以兼容已有的多服务、多载波通信***，例如图11所示出的一种多子带通信***示意图。可以看出，该多子带通信***与MS-QP序列的结构相似，可以将所需频带划分为多个子带，如M个子带。并在相邻两个子带之间***频域保护间隔。对于不同的子带可以分配给不同的数据服务，例如子带1配置服务1，子带2配置服务2等。可以理解的是，对于MS-QP序列其结构可以如图12所示，与图11不同的是，对于每个子带上不再配置相应服务，而是配置各个窄带ZC序列。
但是，如图12所示出的MS-QP序列，当其需要传输数据进行通信时，即将图11与图12相结合，则只能通过时分双工的方式实现，使得效率极低。因此在一些例子中，还可以对上述MS-QP序列做出进一步改进，使得发送的通信协同序列信号可以同时支持目标检测和通信，且相互无干扰。
因此，在一些例子中，通信协同序列信号还可以包括数据符号分量。换句话说，就是还可以将数据符号分量嵌入至MS-QP序列中以形成通信协同序列信号。其中，数据符号分量包括多个子带的数据序列。
例如，可以对MS-QP序列中的每个子带上的窄带ZC序列进行M’次重复传输，则在时域上则拓展为原来长度的M’倍，对于每个子带而言其传输的序列长度为L mM’，并且可以将时域拓展序列记作b e，m[n]。其中，M’为大于或等于2的正整数。角标e用于表示扩展(expand)。将b e，m[n]进行L mM’点DFT后则只有在M’k(k＝0，1，...，L m-1)的频点(子载波)上有值，而其余频点数值均为0。当然，对于M’k频点上的值根据窄带ZC序列确定，因此可以为0或其它任意数值。而对于除M’k频点以外的其它频点，则可以称为置零频点。对于这些置零频点则可以用于调制信息，即用于传输数据，例如传输数据序列。图13则示出了一种经过时域扩展后的MS-QP序列频谱示意图，该图以某个子带上的窄带ZC序列经过M’＝2倍时域扩展为例进行的绘制。可以看出，在该窄带ZC序列进行时域扩展之前， 几乎不存在幅度为0的频点，及时部分频点上幅度为0，但是在该频点上仍然会存在传输其它数据。例如实线所示出的，部分实线的幅度为0，但是幅度为0的频点上仍然存在幅度为1的情况。而当该窄带ZC序列在时域上进行了重复传输，得到时域扩展后的窄带ZC序列如虚线所示，会出现多个频率为0的频点，即置零频点。其中，置零频点例如图13中圆圈所圈出的位置。同时，对于此类置零频点上则不会出现其它传输的数据，因此，此类置零频点上可以用于传输数据序列。显然，若在置零频点传输数据序列，可以保证数据符号分量与检测序列分量的波形相互正交，从而避免相互干扰。
可以理解的是，当窄带ZC序列在时域上重复传输的次数越多，对应的置零频点也就越多。可以理解的是，在某个子带上非置零频点仅为M’的整数倍。
在一些例子中，数据序列可以是PSK符号向量，其中某个子带上传输的PSK符号的长度可以与该子带上的窄带ZC序列长度相同，即长度为L m。因此，则第m个子带上传输的PSK符号向量可以记为S i’，m＝[s i’，m[0]，s i’，m[1]，...，s i’，m[L m-1]]，其中i’＝1，2，...，M’-1。为了可以使S i’，m落在M’k+i’(k＝0，1，...，L m-1)频点上，可以生成如下长度为L mM’的数据序列。如
Figure PCTCN2022083456-appb-000030
其中，
Figure PCTCN2022083456-appb-000031
之后，可以将
Figure PCTCN2022083456-appb-000032
与b e，m[n]进行叠加，从而实现与检测序列分量同时传输。
例如公式8则示出了将
Figure PCTCN2022083456-appb-000033
与b e，m[n]进行叠加后的通信协同序列信号
Figure PCTCN2022083456-appb-000034
Figure PCTCN2022083456-appb-000035
其中，n””＝0，1，...，L mM’-1。α e，m和α i’，m为功率分配因子，可以预先设定。
图14为本申请实施例提供的一种单子带通信协同序列信号示意图。可以理解的是图14仅仅示出了某一个子带上的通信协同序列信号生成过程，对于每个子带均可以参考该图中示出的方式生成各个子带上的通信协同序列信号。从而形成通信协同序列信号
Figure PCTCN2022083456-appb-000036
对于每个子带而言，在该子带上传输的窄带ZC序列经过了M’次重传后形成了M’个副本，如窄带ZC序列副本1、窄带ZC序列副本2……窄带ZC序列副本M’。同时，对于数据序列也可以进行在时域上重复传输M’次，例如数据序列副本1、数据序列副本2……数据序列副本M’。当然，为了保障各个数据序列副本可以落在置零频点上，因此需要对重复传输的数据序列副本进行相应的相位调整，例如将数据序列副本通过进行
Figure PCTCN2022083456-appb-000037
的相位调整与窄带ZC序列相叠加。显然，对于第一个数据序列副本由于M’＝1，因此M’-1＝0则无需进行相位调整。当对不同的数据序列副本进行了相应的相位调整后，与对应的窄带ZC序列副本相叠加，即可得到该子带上的通信协同序列信号。
当然，在一些例子中，对于每个子带上的通信协同序列信号还可以在信号前添加循环前缀，以便用于该通信协同序列信号的接收方进行数据解调。
在一些例子中，基于公式8所示出的
Figure PCTCN2022083456-appb-000038
和公式5，则对多个不同子带上的通信协同序列信号进行频谱搬移以及频域叠加，即可进一步得到公式9。
Figure PCTCN2022083456-appb-000039
以及，根据公式4还可以进一步得到公式10
Figure PCTCN2022083456-appb-000040
其中，
Figure PCTCN2022083456-appb-000041
L’ m＝L mM’为时域扩展后的序列长度。且根据公式6可以进一步得到公式11。
Figure PCTCN2022083456-appb-000042
此时，该多子带的通信协同序列信号
Figure PCTCN2022083456-appb-000043
其频谱效率可以为
Figure PCTCN2022083456-appb-000044
其单位可以是比特/秒/赫兹(bit/s/Hz)，L为PSK调制阶数，γ为循环前缀时长在该序列总时长中所占比例。
在又一些例子中，由于太赫兹信号的路径损耗远高于低频电磁波，因此使得发送的通信协同序列信号所产生的回波信号其信噪比远低于毫米波频段的雷达。例如当目标与通信检测一体化装置距离为3米时，通信检测一体化装置的接收设备检测到的回波信号的信噪比甚至可以低至-56.6分贝(dB)。因此，在S401发送通信协同序列信号时可以进行循环重复发送，例如采用通信协同序列信号多次重复帧结构。每一帧内可以包含多个字块，每个字块内传输一个通信协同序列信号，从而克服了信号传输过程中的低信噪比情况。显然，通过重复帧结构的设计也同时避免了发送长序列的相关复杂操作。
S402，接收通信协同序列信号的回波信号。
通信检测一体化装置300中的接收器304通过天线305接收通信协同序列信号的回波信号。可以理解的是，在接收回波信号时，只需要多个带通滤波器将各个子带中的信号分离即可，因此可以使用低采样率的ADC进行回波信号的采集。
太赫兹频段的信号通常会受到发送器303和接收器304产生的硬件失配影响，并产生一定的非线性混合失真。可以理解的是，除DAC和ADC之外，由于太赫兹频率介于微波与光之间，因此射频前端器件制作复杂度较高，从而导致发送器303和接收器304的硬件失配问题显著。其中，硬件失配主要包括用于太赫兹频段的发送器303和接收器304结构中的同相支路/正交支路幅度与相位不平衡，即同相/正交(in-phase/quadrature，I/Q)失衡。同时对于收发本振产生的相位噪音和高增益功率放大器的非线性等因素，均会导致信号产生非线性失真，并影响太赫兹通信性能和检测性能。
在一个例子中，可以假设发送器303已经对发送端I/Q失衡和非线性进行了补偿。因此回波信号只会受到接收器304的相位噪声和I/Q失衡。当然，对于回波信号还可能受到器件产生的噪声之外的其余杂波信号干扰。由于采用太赫兹频段的信号器波束通常较窄，因此受到的杂波干扰较小，使得杂波信号强度相对比较弱。因此对于回波信号中的受到的杂波可以与热噪声共同建模为加性复高斯白噪声，并作为等效噪声项
Figure PCTCN2022083456-appb-000045
对于发送设备发送的s[n]而言，其符号周期可以为T s，假设待检测的目标数量为Q’，第q’个目标的回波延迟为τ q’T s，对应的信道衰落系数为h q’，归一化多普勒记为
Figure PCTCN2022083456-appb-000046
u q’为第q’个目标的目标运动速度，c为电磁波传播速度，I/Q失衡系数记为μ I/Q和υ I/Q，收发本振相位噪声记作θ(n)，因此回波信号y[n]可以如公式12所示。
Figure PCTCN2022083456-appb-000047
其中，
Figure PCTCN2022083456-appb-000048
与h q’共轭，s *[n]与s[n]共轭。
对于y[n]而言，其会受到硬件失配和多普勒频移的影响，并导致检测性能下降。通过仿真可以知道，I/Q失衡系数对目标检测的影响较小，从而可以忽略。其原因在于通常情况下I/Q失衡系数记为μ I/Q近似为1，而υ I/Q近似为0，同时可以令共轭序列与原序列的互相关值较小。而对于相位噪声，对于发送器303和接收器304虽然共用本振，但是由于回波延迟的存在，使得收发相位噪声无法完全抵消。但经过证明可以知道，加载在信号上的总体相位噪声并不会随着时间无线累积，因此只会在距离像上主峰附近产生小幅度旁瓣。其中，距离像是指发送信号与回波信号的相关函数图像，用于目标检测和时延估计。因此，本申请可以忽略硬件失配问题。
可以理解是的，上述提到的y[n]、s[n]、s *[n]、θ(n)和
Figure PCTCN2022083456-appb-000049
中的n还可以替换为公式1中的n”’公式8中的n””。
图15为一种受硬件失配影响的距离像示意图。可以看出，当时延为1000时出现相关峰，可以理解的是相关峰也可以称为主峰。而在相关峰的两侧，即时延在1000左右时，则出现了小幅度旁瓣。
而在另一方面，由于采用太赫兹频段的信号其载波频率极高，因此会受到多普勒频移的显著影响。当多普勒频移较大且回波信号较长时，距离像上回产生明显的距离旁瓣，并降低相关峰的幅度。可以理解的是，其原因在于能量是守恒的，当其它时延段上出现了旁瓣能量，则响应相关峰的能量也必然会受到影响，甚至出现偏移。例如图16中则示出了多普勒频移影响下的距离像示意图，可以清楚看出，当时延为4000多时，出现了明显旁瓣，在时延将近8000的位置有出现了明显旁瓣。显然，对抗多普勒频移也是尤为重要的。
因此，在一些例子中，还可以对S401中的通信协同序列信号做出进一步的优化，当然可以理解的是，对于此部分的优化在S401中实现。对于S401中的MS-QP序列，以其中的一个子带为例进行描述，因此，对于某个子带上的窄带ZC序列，基于公式1，还可以等效为公式13。
Figure PCTCN2022083456-appb-000050
其中，p m满足0＜p m＜L m，且gcd(p m，Lm)＝1，
Figure PCTCN2022083456-appb-000051
Figure PCTCN2022083456-appb-000052
为整数。当然，对于公式1则是当q取值为0时的表示。由于窄带ZC序列具有诸多的良好性质，如窄带ZC序列为横模序列，且峰均比低；以及窄带ZC序列具有完美的自相关特性即公式14所示，
Figure PCTCN2022083456-appb-000053
可以理解的是，其中n””’表示为待检测点，m’表示经过时延后的待检测点。显然，当经过时延的回波信号的带监测点与发送信号的待检测点相同时，则输出L m，相应的则可以产生相关峰，从而检测出目标。
窄带ZC序列的良好性质还包括具有低互相关特性。例如可以假设同长的两个窄带ZC序列b 1[n”’]和b 2[n”’]，其根指数分别为p 1和p 2，则当gcd(|p 1-p 2|，L m)＝1时，
Figure PCTCN2022083456-appb-000054
为常数
Figure PCTCN2022083456-appb-000055
由于窄带ZC序列的完美自相关特性，当其作为检测序列时，理想情况下可以得到仅有单个主峰的距离像，以实现高精度的时延估计。当然，其假设为仅存在1个目标。当存在多个目标时，则可以得到多个主峰的距离像。但是，当多普勒频移影响下，则完美自相关特性将不再满足，具体而言，当q”＝0时，若只考虑回波信号受到多普勒频移影响，则公式14可以进一步表示为公式15。
Figure PCTCN2022083456-appb-000056
其中，将(p mn”’-v q’L m)(mod L m)，即p mn”’-v q’L m模L m的余数作为自变量，||R bb[n””’]||的图像可以如图17所示。可以看出多普勒频移对序列自相关的影响机理。当归一化多普勒参数v q’很小时，v q’L m可以近似于0，对应的n”’取非零值时对应的余数(即(p mn”’-v q’L m)(mod L m))均为非零整数，对应的||R bb[n””’]||的值为0。而当n”’取0时，则||R bb[n””’]||逼近最高值，产生相关峰，显然窄带ZC序列具有完美自相关特性。但是相对地，当多普勒频移不可忽略时，则对应(p mn”’-v q’L m)(mod L m)的取值将与整数点产生一定偏差，使得||R bb[n””’]||在n取非零整数值时不再为0，而是落在临近的旁瓣上。随着多普勒和序列的增大，当v q’L m≥1时，相关峰的位置都将会发生偏移，将会对测距精度产生严重影响。
显然，若假设v q’L m远小于1时，||(p mn”’-v q’L m)(mod L m)||越小，即||(p mn”’)(mod L m)||越小，v q’L m产生的偏差使得对应的||R bb[n””’]||越大。并且根据数论知识可以知道，
Figure PCTCN2022083456-appb-000057
构成了一个模L m的完全剩余类。因此，在一些例子中，可以通过对p m进行优化，使得||(p mn”’)(mod L m)||取值较小时所对应的n”’尽量集中在0值附近，这样可以使得由多普勒频移产生的高旁瓣与相关峰尽量集中，进而避免由于多普勒频移所产生的距离旁瓣被误判为目标。
在一些例子中，可以令
Figure PCTCN2022083456-appb-000058
以满足上述对抵抗多普勒频移的需求。在一个例子中，假设
Figure PCTCN2022083456-appb-000059
则(p m2n”’)(mod L m)＝L m-2n”’，以及
Figure PCTCN2022083456-appb-000060
Figure PCTCN2022083456-appb-000061
若假设L m＝10007时，则(p mn”’)(mod L m)的分布可以如表1所示。
n”’(mod L m) -4 -3 -2 -1 0 1 2 3 4
||(p mn”’)(mod L m)|| 2 -5002 1 -5003 0 5003 -1 5002 -2
表1
通过表1可以看出，显然当||(p mn”’)(mod L m)||取值较小时，所对应的n”’分布在0 值两侧的近邻。此时若进行目标检测，高旁瓣将会集中在相关峰附近，以减小多普勒频移的影响。
可以理解的是，上述对根指数进行优化中涉及到的参数n”’还可以替换为n和n””。
图18则示出了根指数优化前的距离像示意图。该图以L m＝10007，p m＝3为例进行绘制。显然，当根指数为3时，在时延为4000多以及将近8000的位置出现了明显旁瓣。对于此类明显旁瓣则很容易被误判为目标。但通过图19则可以看出，优化后的根指数为5003后，多普勒频移产生的高旁瓣很好的聚拢在相关峰附近，因此图19中仅看到一个相关峰。可以理解的是，对于图18和图19均以目标数量为1进行说明。
对于MS-QP序列中的多个子带，不同的子带分量相互正交(也可以说互相关为0)。因此MS-QP序列的自相关特性等价于不同子带分量上的自相关叠加，对于每个子带而言，其自相关旁瓣的分布主要取决于该子带上的窄带ZC序列的自相关分布。因此，对于MS-QP序列所收到的多普勒频移，在距离像上产生的旁瓣分布主要取决于各个子带上的窄带ZC序列。因此，可以理解的是，对于每个子带可以根据该子带上的窄带ZC序列进行各自根指数的独立优化。使得最终MS-QP序列对应的距离像上产生的旁瓣可以很好的聚拢在相关峰附近。
可以理解的是，上述所涉及的针对根指数的优化，在S401发送通信协同序列信号时实现，其实际上是对于通信协同序列信号中的根指数进行优化，以便在S402中接收到通信协同序列信号的回波信号后，可以很好的进行目标检测。
继续回到S402，图20为本申请实施例提供的一种接收设备回波信号构建示意图。与图5和图6相对应，图20示出了相应的接收设备。y(t)表示为回波信号的模拟信号和y[n]表示回波信号的基带数字信号。显然，y[n]与图5和图6中的x[n]的是与符号间隔显然是相同的，均为T M。通过图20可以看出，在接收设备120中首先需要多个带通滤波器分离出不同子带对应的信号。然后针对每个子带上的信号再经过低采样ADC进行模数转换得到对应子带的数字基带信号。之后，再通过数字信号处理模块进行序列重构，并得到y[n]。其中，可以看出在对各个子带进行分离时，可通过反向的频谱搬移实现，例如图20中通过乘法器乘以
Figure PCTCN2022083456-appb-000062
实现分离各个子带上的信号。
继续回到图4，在S402之后将执行S403。
S403，对回波信号进行FFT，以在频域内确定回波信号的雷达检测分量。
在一个例子中，当接收器304接收到回波信号y[n]后，可以对回波信号y[n]进行FFT，从而可以在频域内确定出该回波信号的雷达检测分量。可以理解的是，y[n]可以具有多个子带，因此S403可以是对每个子带上的模拟信号通过ADC转换为基带数字信号。然后，针对每个子带，可以做L mM’点的FFT从时域转换至频域，例如图21所示中的S2101。此时可以取出对应非置零频点上传输的窄带ZC序列，即雷达检测分量。
在一些例子中，由于y[n]是x[n]经过目标反射后形成的回波信号，因此y[n]中应当同样具有检测序列分量和数据符号分量，其中，y[n]中的检测序列分量则称为雷达检测分量，且同为上述涉及到的MS-QP序列。以便后续用于信道估计和目标检测，即图21中的S2102。当然，在又一些例子中，由于雷达检测分量和数据符号分量的互相关很小，因此在y[n]通过ADC转换为基带数字信号可以不分离出雷达检测分量，而是直接采用携带数据符号分量的MS-QP序列进行后续目标检测。
在另一些例子中，当针对每个子带，可以做L mM’点的FFT从时域转换至频域后，还可以取出对应置零频点上的数据序列，即y[n]中的数据符号分量。然后对y[n]中的数据符号分量做L mM’点的IFFT由频域转换至时域，即图21中的S2103，以便后续可以继续进行信号解调，即图21中的S2104。该过程实现了数据传输功能。
S404，对检测序列分量和雷达检测分量做相关，确定至少一个相关峰及至少一个相关峰对应的时延信息。
在一个例子中，可以采用发送时的检测序列分量和接收到的雷达检测分量做相关，以确定出一个或多个相关峰和每个相关峰的时延信息。
由于y[n]中的雷达检测分量和数据符号分量的互相关很小，因此通常可以直接采用y[n]与x[n]做相关。因此，首先可以确定出接收到的回波信号y[n]，可以参考公式11所示出的。当然，在一些例子中，为了克服传输的信号信噪比弱的问题，当S403中通过重复的帧结构进行多个字块上的重复传输时且带外泄漏，则基于公式12还可以进一步优化为公式16。
Figure PCTCN2022083456-appb-000063
其中，K”表示重复的帧结构中一帧内包含的子块个数，K”为大于或等于2的正整数。k”＝0，1，...，K”-1。n 1＝0，1，...，N 1-1，N 1为第k”个字块上传输的MS-QP序列长度。其中关于噪声干扰项
Figure PCTCN2022083456-appb-000064
可以理解的是可以包含有噪声以及数据符号，其中
Figure PCTCN2022083456-appb-000065
中的数据符号产生的原因在于存在多普勒频偏和一些非理想因素，导致数据传输时数据符号分量与检测序列分量并非完美的正交，因此则仍会产生小部分的干扰。
当确定出y k”[n 1]后，可以针对每个字块的y[n]可以与单个x[n]进行长度为L mM’的相关，并记作r k”[n 1]，假设给定一个时刻n 1，对K”块循环相关计算结果的相应位置取值r k”[n 1]，k”＝0，1，...，K”-1组成的向量进行qK”点FFT，从而得到RDM。例如图22则示出了一种生成DRM的过程。可以清晰的看出，对于DRM中的每一个元素，其二维坐标分别对应一个可能的目标回波时延以及多普勒频移取值。在一些例子中，对于
Figure PCTCN2022083456-appb-000066
则可以不进行特意处理。
可以理解的是，上述x[n]在一些情况下可以替换为
Figure PCTCN2022083456-appb-000067
当根据回波信号确定出DRM后，可以通过其中的各个元素的幅度确定是否存在相关峰。当然在一些例子中还可以对目标运动速度进行估计。
在一个例子中，可以采用恒虚警概率检测的方式确定出至少一个相关峰。例如通过预设的虚警率确定假设检验判定阈值。并通过DRM中每个位置上的元素幅值与该假设检验判定阈值进行对比，从而确定是否存在相关峰。可以理解的是，虚警率可以是预设不变的，而在每次进行目标检测时，可以根据虚警值动态确定本次的假设检验判定阈值，例如可以通过全自动计算仿真的方式确定根据虚警值动态确定本次的假设检验判定阈值。当然，还可也采用其它任意等效的方式实现，具体确定过程可以参考现有方式进行，本申请在此不再赘述。例如，可以假设根据回波信号确定出的DRM记为R(n 1，k”)，则确定第(n 1，k”)个单元格是否存在相关峰的假设检验可以表示为公式17。
Figure PCTCN2022083456-appb-000068
其中，H 1表示发现相关峰(即目标)，H 0表示没有发现相关峰(即目标)。T 门限表示本次假设检验判定阈值，σ 2表示对应DRM单元中的噪声功率。
在一个例子中，若第(n 1，k”)个单元格检测出相关峰，则对应目标的时延估计可以表示为τ＝n 1T s，演示估计出的τ即该相关峰对应的时延信息。对应的多普勒频偏参数可以表示为
Figure PCTCN2022083456-appb-000069
在一些例子中，则对应的目标速度u则可以通过公式18表示，其中包括有正速度和负速度两种情况。
Figure PCTCN2022083456-appb-000070
通过公式18所估计出的目标运动速度，在高信噪比环境下，绝对误差范围为
Figure PCTCN2022083456-appb-000071
平均绝对误差为
Figure PCTCN2022083456-appb-000072
其中λ表示为太赫兹信号的波长，例如可以是3×10 8/f c。
继续回到图4，当S404中确定出根据至少一个相关峰及至少一个相关峰对应的时延信息后，可以继续执行S405。
S405，根据至少一个相关峰及至少一个相关峰对应的时延信息，确定至少一个目标以及至少一个目标距离。
可以根据S404中确定的至少一个相关峰以及各个相关峰的时延信息，从而确定出至少一个目标以及对应的目标距离。
在一个例子中，考虑到上述采用的通信协同序列信号可以将多普勒频移导致的旁瓣聚拢至相关峰附近，并且实际中在小范围内出现多个目标的概率往往极低。因此可以假定相关峰临近区域不存在其它目标。从而降低多普勒旁瓣对***虚警概率的影响。例如，可以假设RDM中第
Figure PCTCN2022083456-appb-000073
个单元格检测出相关峰，则认为
Figure PCTCN2022083456-appb-000074
范围内除相关峰对应的目标外无其它目标。其中，n 0可以理解为预设的动态范围。因此，则可以根据每个相关峰，确定出与之对应的唯一目标。
在另一个例子中，对于目标距离的确定，可以估计为
Figure PCTCN2022083456-appb-000075
其中c为电磁波传播速度(也可以称为光速)。在高信噪比环境下，则d 目标距离的绝对误差范围可以是
Figure PCTCN2022083456-appb-000076
而平均绝对误差为
Figure PCTCN2022083456-appb-000077
至此，通过上述方式则实现了对目标的检测。本申请通过基于时域进行扩展，确定出由检测序列分量和构成的通信协同序列信号。通过对检测序列进行时域延拓，使扩展后的检测序列分量的频谱产生多个零值频点，以便传输数据符号分量，从而实现了通信、检测功能无干扰同时进行。当然，上述方式保证了可以支持以低复杂度算法对长序列进行相关检测。由于检测序列分量是一种基于频谱扩展的MS-QP序列，通过划分为多个子带解决了太赫兹超宽带信号对接收设备中ADC采样率要求高的问题。同时每个子带上传输窄带ZC序列，因此具有良好的自相关特性。保障了接收设备在利用超高带宽的同时，可使用低采样率ADC对信号进行采集和恢复，有效降低成本。
当然，本申请还同时为了解决针对多子带序列不再具有恒模特性，产生较高峰均比的问题，对序列不同子带分量乘不同相位因子，对相位因子进行最优化，以降低序列峰均比，且不影响序列自相关特性。以及，为了解决针对强多普勒频移对太赫兹检测性能的影响，对MS-QP序列中的根指数进行优化，使得多普勒频移导致的高距离像旁瓣聚拢在主峰附近，并提出主峰邻近区域排除策略，有效降低多普勒对目标检测的干扰。
图23为本申请实施例提供的一种通信协同序列信号单子带频谱示意图。
如图23所示，为具体参数进行代入上述所涉及的通信协同序列信号后的单子带频谱示意图。其中，通信协同序列信号中的MS-QP序列，取单个子带内的窄带ZC序列长度L m为1007，另根指数p m为503，以尽量减少多普勒频移对目标检测的影响。参考图5和图6所示出的方式，假设有10个子带，则将M＝10个子带在频谱上进行潘奕拼接，同时可以在相邻子带之间设置L G＝100的频域保护间隔。则可以得到长度N＝ML G+∑L m的MS-QP序列。可以设单个子带内窄带ZC序列的符号间隔为1.1纳秒(ns)，序列总长为1.1077微秒(μs)。当经过成形滤波器后可以形成一子带分量，带宽为0.9GHz，将不同子带分量在频谱上进行搬移和拼接，组成相邻M＝10个子带，子带间设置0.1GHz频域保护间隔，以形成总带宽为10GHz的MS-QP序列。
可以设定MS-QP序列在时域扩展次数M’＝2。也就是说在MS-QP序列中，单个子带上对应L m长度的窄带ZC序列，时域重复2次，以形成2L m长度的序列。根据上述图13相应描述的视频对应关系，该2L m长度的序列在进行DFT后只会占用偶数频点，因此可以在技术频点上调制数据序列。例如，取L m＝1007长的一组数据序列S i’，m＝[s i’，m[0]，s i’，m[1]，...，s i’，m[L m-1]](i’＝1，2，...，M’-1)，通过相位旋转同样将其扩展为2L m长的数据序列，表示为
Figure PCTCN2022083456-appb-000078
根据序列的视频对应关系，使得2L m长的数据序列仅占用奇数频点，因此与检测序列分量可以实现无干扰兼容。正如图23所示出的，以一个子带为例，其检测序列分量与数据符号分量叠加后相互无影响。可以每当频率取整数时，一个序列幅度为0，另一个序列幅度较高，相互不会产生干扰。相应的，该MS-QP序列的序列长度为扩展前的M’倍。图中以不同粗细的线条表示检测序列分量以及数据符号分量。
为了抵抗太赫兹频段的极低信噪比，可以在发送设备发送通信协同序列信号时进行多次重复传输。在接收设备接收回波信号时，则通过连续多个接收序列字块的相关检测实现低信噪比下的高精度测距和测速。例如，对单个接收字块，接收设备通过M＝10个带通滤波器，以1.1ns的采样周期分别采样每个子带的回波信号。每个子带上取长度为1007的序列，然后对各个子带回波信号在频谱上进行搬移和拼接，恢复出该字块的超宽带接收序列，即回波信号。长度为N＝ML G+∑L m，其中L G＝100。该回波信号等效于直接以T s＝0.1ns的采样周期采样活剥信号所得序列。接收设备采集K”块长度为N的回波信号进行处理，表示为y k”[n]，n＝0，1，...，N-1，k”＝0，1，...，K”-1。
在一个例子中，例如对每个子块的回波信号与MS-QP序列做长度为N的循环相关，形成相关矩阵r k”[n]。取k”块的循环相关计算结果中的第n个位置的相关值联合做qk”点FFT，从而获得RDM进行后续信号处理。例如图24中示出的DRM生成过程信号波形示意图。其中，图24中的(a)示出了接收到的回波信号其波形示意，例如其中的黑色竖线则表示为某一块回波信号，显然(a)则是由多块回波信号叠加构成的。基于(a)针对每块回波信号可以与发送的通信协同序列信号进行循环相关，则得到了(b)所示出的，可以看出， 无论是第几个子块，其相关值最高的位置几乎相同，即(b)中的白色横线。在针对(b)中横向做FFT，从而得到RDM，如(c)所示出的。可以看到(c)中的相关峰汇聚成了一个白点，即(c)中中间的白点。至此，可以根据RDM的横纵坐标分别确定出目标距离以及目标运动速度。
在一些例子中，可以通过仿真评估对本申请所涉及的通信协同序列信号进行性能测试。具体可参考表2示出的太赫兹***参数。
参数 取值 参数 取值
中心频率 300GHz 最大路径损耗(单程) -91.5dB
通信带宽 10GHz 天线及环境温度 285k
发送功率 8.5dBm(PA) 接收设备噪声系数 11dB
收发天线增益 40dBi 测距范围 3m以内
最大测距范围 3m 测速范围 -20m/s～20m/s
反射衰减 25dB    
表2
在一个例子中，以采用表2所示的太赫兹***参数进行测试，可以确定3m检测范围内太赫兹检测的接收设备信噪比通常高于-56.5dB。同时在信号传输过程中假设相位噪声服从逐符号游走模型。其中，逐符号游走模型例如可以是某个符号周期的相位噪声θ x1可以表示为前一个符号的相位噪声θ x1-1加上一随机增量Δθ x1，该增量为服从Δθ x1～N 2(0，(0.316°))的随机变量。则接收设备I/Q失衡参数可以通过公式19和公式20计算得到。
μ I/Q＝cos(φ)-jεsin(φ)
                      ……公式19
υ I/Q＝εcos(φ)-j sin(φ)
                        ……公式20其中，ε＝0.2，φ＝10°。
在一些例子中，进行仿真时，将本申请所涉及的MS-QP序列、通信协同序列信号与窄带ZC序列、宽带ZC序列和线性调频信号进行了对比。其中，窄带ZC序列长度设为1007，根指数设为503，窄带ZC序列符号间隔设为1.1ns，序列总长为1.1077μs，与MS-QP序列等长，带宽0.9GHz。对于宽带ZC序列，设长度为11077，根指数为5538，符号间隔为0.1ns，序列总长为1.1077μs，带宽为10GHz。对于现行调频信号，设脉冲长度为1.1077μs，带宽为10GHz。对于MS-QP序列及其基准序列均重复传输1024次。对于通信协同序列信号由于其长度为MS-QP序列的M’倍，因此为了对比相对公平，令通信协同序列信号重复为1024/M’次。对于MS-QP序列和通信协同序列信号，其具体参数可以参考图23中所设定的数值。
图25则示出了MS-QP序列、通信协同序列信号与窄带ZC序列、宽带ZC序列和线性调频信号的测距性能对比。其中，横坐标为信噪比(signal noise ratio，SNR)单位为dB，纵坐标为平均绝对测距误差，用于体现序列测距性能。可以看出，无论是那种序列或信号，在随着信噪比不断提升时，其平均测距误差均出现下滑，甚至在一定信噪比范围内出现了断崖式下降。直到由于通信带宽的限制导致测距误差在下降到一定程度后趋于稳定。对于 超宽带检测序列，例如MS-QP序列、宽带ZC序列和线性调频信号，在信噪比为-55dB左右平均绝对测距误差达到恒定，约为0.0035m，达到了厘米级测距精度。窄带ZC序列由于带宽受限，则平均绝对测距误差达到的恒定值会更高，约为0.04m。可以看出，MS-QP序列可以实现等宽带ZC序列几乎相同的测距精度，同时还可以利用多子带并行滤波进行发送和接收，极大降低了发送设备中DAC和接收设备中ADC的要求。
对于通信协同序列信号(即图中的通信检测一体化波形)能够在支持通信的同时，实现与淡出能检测序列相同的测距精度。但由于通信和检测对于发送能量的均分，在达到高精度定位时，最低信噪比通常会比MS-QP序列有3dB的增加。当信噪比高于-52dB时，数据符号分量的叠加将不会影响检测序列分量的测距精度。
图26则示出了上述5中序列在测速性能上的对比。其中，测速性能通常由检测序列的长度决定。因此，相同时长的序列可以达到的测速性能也基本一致。通过图26可以看出，当信噪比高于-55dB时，可以不同的检测序列的平均绝对测速误差基本可以达到恒定值。对于通信协同序列信号(即图中的通信检测一体化波形)，在加入数据符号分量后，通常需要增加3dB平均绝对测速误差才能达到恒定值。在信噪比高于-52dB时，数据符号分量的叠加将不会影响检测序列分量的测速精度。
在一个例子中，通信协同序列信号中的检测序列分量与数据符号分量可以实现无干扰同时传输，其中，时域扩展参数M’决定了数据符号分量与检测序列分量在频域资源的占比。例如，检测序列分量占可用频域资源的1/M’，增加M’可以提高通信数据传输效率。但是当M’超过某一阈值后，将会影响测距和测速精度。因此，可以根据实际情况动态调整M’的取值，以实现通信性能和检测性能到达完美平衡。例如图27示出了不同频域占比下对平均绝对测距误差的影响。可以看出，在相同信噪比情况下，检测序列分量的频域占比超过某一数值后，平均绝对测距误差将趋于恒定值，以实现厘米级定位精度。而当检测序列分量的频域占比小于该数值时，平均绝对测距误差将逐渐恶化。以-47dB为例，可以看出当检测序列分量的频域占比不小于1/10时，平均绝对测距误差将趋于恒定值。对于-44dB而言，由于其信噪比已经达到比较好的状态，因此检测序列分量的频域占比将不会影响平均绝对测距误差。又例如图28示出了不同频域占比下对平均绝对测速误差的影响。同样，在相同信噪比情况下，检测序列分量的频域占比超过某一数值后，平均绝对测速误差将趋于恒定值。而当检测序列分量的频域占比小于该数值时，平均绝对测速误差将逐渐恶化。仍以-47dB为例，可以看出当检测序列分量的频域占比不小于1/10时，平均绝对测距误差将趋于恒定值。对于-44dB而言，由于其信噪比已经达到比较好的状态，因此检测序列分量的频域占比将不会影响平均绝对测速误差。
图29为本申请实施例提供的另一种通信检测一体化装置示意图。
如图29所示，本申请还提供了另一种通信检测一体化装置2900，该装置2900包括：
发送模块2901，用于发送通信协同序列信号。其中，信号生成模块2903，用于确定通信协同序列信号，通信协同序列信号包括检测序列分量，检测序列分量为具有多个子带的MS-QP序列，每个子带包括一个窄带ZC序列。
接收模块2902，用于接收通信协同序列信号的回波信号。
信号处理模块2904，用于对回波信号进行快速傅里叶变换，以在频域内确定回波信号的雷达检测分量。
信号处理模块2904还用于，对检测序列分量和雷达检测分量做相关，确定至少一个相关峰及至少一个相关峰对应的时延信息。
信号处理模块2904还用于，根据至少一个相关峰及至少一个相关峰对应的时延信息，确定至少一个目标以及所述至少一个目标距离。
在一个可能的实施方式中，信号生成模块2903还用于：采用最优相位因子集合对MS-QP序列中的各个ZC序列进行相位调整。
在一个可能的实施方式中，ZC序列包括：序列长度L m和根指数p，其中，根指数p满足
Figure PCTCN2022083456-appb-000079
L m表示第m个子带上ZC序列的长度，m为大于或等于2的正整数。
在一个可能的实施方式中，信号生成模块2903还用于：每两个相邻所述子带的所述ZC序列之间设置频域保护间隔。
在一个可能的实施方式中，通信协同序列信号还包括：数据符号分量。其中，数据符号分量包括多个子带的数据序列。信号生成模块2903还用于，将每个子带上的数据序列在子带上的多个置零频点上进行传输，置零频点为子带上的ZC序列在时域上进行M次重传得到。
在一个可能的实施方式中，信号生成模块2903还用于：针对每个子带上传输的数据序列进行相应的相位调整。
在一个可能的实施方式中，信号处理模块2904还用于：至少一个相关峰与至少一个目标一一对应；以及，根据至少一个相关峰对应的时延信息确定与至少一个相关峰对应的目标的目标距离。
本申请所提供的装置2900可以实现上述图1至图28中描述的任意一种方法，具体实现方式可以参考述图1至图28的相应描述，在此不再赘述。
图30为本申请实施例提供的又一种通信检测一体化装置示意图。
如图30示出了又一种通信检测一体化装置3000。该通信检测一体化装置3000可以为上述图1至图29所涉及方案中的通信检测一体化装置。
该通信检测一体化装置3000可以包括处理器3010，外部存储器接口3020，内部存储器3021，通用串行总线(universal serial bus，USB)接口3030，充电管理模块3040，电源管理模块3041，电池3042，天线1，天线2，移动通信模块3050，无线通信模块3060以及显示屏3070等。
可以理解的是，本发明实施例示意的结构并不构成对通信检测一体化装置3000的具体限定。在本申请另一些实施例中，通信检测一体化装置3000可以包括比图示更多或更少的部件，或者组合某些部件，或者拆分某些部件，或者不同的部件布置。图示的部件可以以硬件，软件或软件和硬件的组合实现。
其中处理器3010、外部存储器接口3020和内部存储器3021可以参考图3中处理器301、存储器302中相应的描述，在此不再赘述。
在一些实施例中，处理器3010可以包括一个或多个接口。接口可以包括集成电路(inter-integrated circuit，I2C)接口，集成电路内置音频(inter-integrated circuit sound，I2S)接口，脉冲编码调制(pulse code modulation，PCM)接口，通用异步收发传输器(universal asynchronous receiver/transmitter，UART)接口，移动产业处理器接口(mobile industry processor interface，MIPI)，通用输入输出(general-purpose input/output，GPIO)接口， 用户标识模块(subscriber identity module，SIM)接口，和/或通用串行总线(universal serial bus，USB)接口等。
USB接口3030是符合USB标准规范的接口，具体可以是Mini USB接口，Micro USB接口，USB Type C接口等。USB接口3030可以用于连接充电器为分布式录音设备1700充电，也可以用于通信检测一体化装置3000与***设备之间传输数据。也可以用于连接耳机，通过耳机播放或采集音频。该接口还可以用于连接其他电子设备，例如AR设备等。
可以理解的是，本发明实施例示意的各模块间的接口连接关系，只是示意性说明，并不构成对通信检测一体化装置3000的结构限定。在本申请另一些实施例中，通信检测一体化装置3000也可以采用上述实施例中不同的接口连接方式，或多种接口连接方式的组合。
通信检测一体化装置3000的无线通信功能可以通过天线1，天线2，移动通信模块3050，无线通信模块3060，调制解调处理器以及基带处理器等实现。
天线1和天线2用于发射和接收电磁波信号。通信检测一体化装置3000中的每个天线可用于覆盖单个或多个通信频带。不同的天线还可以复用，以提高天线的利用率。例如：可以将天线1复用为无线局域网的分集天线。在另外一些实施例中，天线可以和调谐开关结合使用。
移动通信模块3050可以提供应用在通信检测一体化装置3000上的包括2G/3G/4G/5G/6G等无线通信的解决方案。移动通信模块3050可以包括至少一个滤波器，开关，功率放大器，低噪声放大器(low noise amplifier，LNA)等。移动通信模块3050可以由天线1接收电磁波，并对接收的电磁波进行滤波，放大等处理，传送至调制解调处理器进行解调。移动通信模块3050还可以对经调制解调处理器调制后的信号放大，经天线1转为电磁波辐射出去。在一些实施例中，移动通信模块3050的至少部分功能模块可以被设置于处理器3010中。在一些实施例中，移动通信模块3050的至少部分功能模块可以与处理器3010的至少部分模块被设置在同一个器件中。
调制解调处理器可以包括调制器和解调器。其中，调制器用于将待发送的低频基带信号调制成中高频信号。解调器用于将接收的电磁波信号解调为低频基带信号。随后解调器将解调得到的低频基带信号传送至基带处理器处理。低频基带信号经基带处理器处理后，被传递给应用处理器。
无线通信模块3060可以提供应用在通信检测一体化装置3000上的包括无线局域网(wireless local area networks，WLAN)(如WiFi网络)，蓝牙，全球导航卫星***(global navigation satellite system，GNSS)，调频(frequency modulation，FM)，近距离无线通信技术(near field communication，NFC)，红外技术(infrared，IR)等无线通信的解决方案。无线通信模块3060可以是集成至少一个通信处理模块的一个或多个器件。无线通信模块3060经由天线2接收电磁波，将电磁波信号调频以及滤波处理，将处理后的信号发送到处理器3010。无线通信模块3060还可以从处理器3010接收待发送的信号，对其进行调频，放大，经天线2转为电磁波辐射出去。
在一些实施例中，通信检测一体化装置3000的天线1和移动通信模块3050耦合，天线2和无线通信模块3060耦合，使得通信检测一体化装置3000可以通过无线通信技术与网络以及其他设备通信。
显示屏3070用于显示图像，视频等。显示屏3070包括显示面板。显示面板可以采用液晶显示屏(liquid crystal display，LCD)，有机发光二极管(organic light-emitting diode，OLED)，有源矩阵有机发光二极体或主动矩阵有机发光二极体(active-matrix organic light emitting diode的，AMOLED)，柔性发光二极管(flex light-emitting diode，FLED)，Miniled，MicroLed，Micro-oLed，量子点发光二极管(quantum dot light emitting diodes，QLED)等。在一些实施例中，通信检测一体化装置3000可以包括至少一个显示屏3070。
相比于现有通信检测一体化包括的单载波/多载波通信波形直接用于雷达检测、数据帧前导序列用于雷达检测以及直接序列扩频信号用于通信检测协同等，存在峰均比高、自相关特性非理想、码长受限、数据率低等诸多问题。以及无法解决超宽带太赫兹环境下通信检测一体化所面临的技术挑战，如高路损、高硬件成本、高多普勒频移等。本申请所涉及的通信协同序列信号巧妙地将太赫兹超高带宽划分为多个相邻子带，并在不同子带上加载窄带ZC序列。使得能够获得良好自相关特性，且只需要利用带通滤波器和低采样率ADC分别处理不同子带即可完成宽带信号的采集，实现厘米级测距精度，有效降低硬件成本。
本申请对需要发送的通信协同序列信号进行波形涉及，通过对检测序列进行多次时域扩展，使扩展后的检测序列分量频谱内部出现置零频点。然后将数据序列分量加载在置零频点上，从而实现了检测与通信功能无干扰同时进行。同时对于雷达检测而言，该设计不会牺牲检测信号带宽，因此不会对测距分辨率造成影响。同时还可支持以低复杂度算法对长序列进行相关检测。而对通信接收而言，通信协同序列信号中检测序列分量部分不会对数据序列分量产生干扰，且可用于信道估计，辅助通信数据的接收。
由于本申请通信协同序列信号中检测序列分量将超宽频带划分为多个子带，每个子带传输窄带ZC子序列，实现窄带序列的频谱扩展。使得检测序列分量具有良好自相关特性，可利用超高带宽实现高测距分辨率，并且只需带通滤波器和低采样率ADC即可进行接收和重建。
进一步的，本申请为了降低发送信号的峰均比，还可以对不同子带的信号在频域上与不同相位因子相乘，优化后可有效降低序列峰均比，同时基本不影响序列的检测性能。
进一步的，针对传统方式中普遍忽略的多普勒频移问题，本申请为了避免多普勒频移导致的高距离像旁瓣，则对检测序列分量中各个子带上窄带ZC序列的根指数进行优化。根指数优化后可使得多普勒频移导致的高距离像旁瓣聚拢在主峰附近，同时利用主峰邻近区域排除策略，假设主峰对应目标近邻区域内无其他目标，减小高距离像旁瓣被误判为目标的概率，有效降低多普勒频移对目标检测的影响。
本领域普通技术人员应该还可以进一步意识到，结合本文中所公开的实施例描述的各示例的单元及算法步骤，能够以电子硬件、计算机软件或者二者的结合来实现，为了清楚地说明硬件和软件的可互换性，在上述说明中已经按照功能一般性地描述了各示例的组成及步骤。这些功能究竟以硬件还是软件方式来执行，取决于技术方案的特定应用和设计约束条件。专业技术人员可以对每个特定的应用来使用不同方法来实现所描述的功能，但是这种实现不应认为超出本申请的范围。
本领域普通技术人员可以理解实现上述实施例方法中的全部或部分步骤是可以通过程序来指令处理器完成，所述的程序可以存储于计算机可读存储介质中，所述存储介质是非短暂性(英文：non-transitory)介质，例如随机存取存储器，只读存储器，快闪存储器， 硬盘，固态硬盘，磁带(英文：magnetic tape)，软盘(英文：floppy disk)，光盘(英文：optical disc)及其任意组合。
以上所述，仅为本申请较佳的具体实施方式，但本申请的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本申请揭露的技术范围内，可轻易想到的变化或替换，都应涵盖在本申请的保护范围之内。因此，本申请的保护范围应该以权利要求的保护范围为准。

Claims (16)

  1. 一种目标检测方法，其特征在于，所述方法包括：
    发送通信协同序列信号，其中，所述通信协同序列信号包括检测序列分量，所述检测序列分量为具有多个子带的多子带准完美MS-QP序列，每个所述子带包括一个窄带扎道夫-初ZC序列；
    接收所述通信协同序列信号的回波信号；
    对所述回波信号进行快速傅里叶变换，以在频域内确定所述回波信号的雷达检测分量；
    对所述检测序列分量和所述雷达检测分量做相关，确定至少一个相关峰及所述至少一个相关峰对应的时延信息；
    根据所述至少一个相关峰及所述至少一个相关峰对应的时延信息，确定至少一个目标以及所述至少一个目标距离。
  2. 如权利要求1所述的方法，其特征在于，所述方法还包括：
    采用最优相位因子集合对所述MS-QP序列中的各个所述ZC序列进行相位调整。
  3. 如权利要求1或2所述的方法，其特征在于，所述ZC序列包括：序列长度L m和根指数p，其中，所述根指数p满足
    Figure PCTCN2022083456-appb-100001
    L m表示第m个子带上ZC序列的长度，m为大于或等于2的正整数。
  4. 如权利要求1-3任意一项所述的方法，其特征在于，所述方法还包括：每两个相邻所述子带的所述ZC序列之间具有频域保护间隔。
  5. 如权利要求1-4任意一项所述的方法，其特征在于，所述通信协同序列信号还包括：数据符号分量；
    其中，所述数据符号分量包括多个子带的数据序列，每个所述子带上的数据序列在所述子带上的多个置零频点上进行传输，所述置零频点为所述子带上的所述ZC序列在时域上进行M次重传得到。
  6. 如权利要求5所述的方法，其特征在于，所述方法还包括：针对每个所述子带上传输的所述数据序列进行相应的相位调整。
  7. 如权利要求1-6任意一项所述的方法，其特征在于，所述根据所述至少一个相关峰及所述至少一个相关峰对应的时延信息，确定至少一个目标以及所述至少一个目标距离，包括：
    所述至少一个相关峰与所述至少一个目标一一对应；以及，根据所述至少一个相关峰对应的时延信息确定与所述至少一个相关峰对应的目标的目标距离。
  8. 一种目标检测装置，其特征在于，所述装置包括：
    发送器，用于发送通信协同序列信号，其中，所述通信协同序列信号包括检测序列分量，所述检测序列分量为具有多个子带的多子带准完美MS-QP序列，每个所述子带包括一个窄带扎道夫-初ZC序列；
    接收器，用于接收所述通信协同序列信号的回波信号；
    处理器用于与存储器耦合，以及读取并执行存储在所述存储器中的指令；
    当所述处理器运行时执行所述指令，使得所述处理器用于对所述回波信号进行快速傅里叶变换，以在频域内确定所述回波信号的雷达检测分量；对所述检测序列分量和所述雷达检测分量做相关，确定至少一个相关峰及所述至少一个相关峰对应的时延信息；根据所述至少一个相关峰及所述至少一个相关峰对应的时延信息，确定至少一个目标以及所述至少一个目标距离。
  9. 如权利要求8所述的装置，其特征在于，所述处理器还用于：
    采用最优相位因子集合对所述MS-QP序列中的各个所述ZC序列进行相位调整。
  10. 如权利要求8或9所述的装置，其特征在于，所述ZC序列包括：序列长度L m和根指数p，其中，所述根指数p满足
    Figure PCTCN2022083456-appb-100002
    L m表示第m个子带上ZC序列的长度，m为大于或等于2的正整数。
  11. 如权利要求8-10任意一项所述的装置，其特征在于，所述处理器还用于：每两个相邻所述子带的所述ZC序列之间具有频域保护间隔。
  12. 如权利要求8-11任意一项所述的装置，其特征在于，所述通信协同序列信号还包括：数据符号分量；
    其中，所述数据符号分量包括多个子带的数据序列，每个所述子带上的数据序列在所述子带上的多个置零频点上进行传输，所述置零频点为所述子带上的所述ZC序列在时域上进行M次重传得到。
  13. 如权利要求12所述的装置，其特征在于，所述处理器还用于：针对每个所述子带上传输的所述数据序列进行相应的相位调整。
  14. 如权利要求8-13任意一项所述的装置，其特征在于，所述处理器还用于：
    所述至少一个相关峰与所述至少一个目标一一对应；以及，根据所述至少一个相关峰对应的时延信息确定与所述至少一个相关峰对应的目标的目标距离。
  15. 一种计算机可读存储介质，所述计算机可读存储介质中存储有指令，其特征在于，当所述指令在终端上运行时，使得所述终端执行如权利要求1-7任意一项所述的方法。
  16. 一种包含指令的计算机设备，当其在终端上运行时，使得所述终端执行如权利要求1-7中的任意一项所述的方法
PCT/CN2022/083456 2021-04-02 2022-03-28 一种目标检测方法及装置 WO2022206700A1 (zh)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202110361611.9 2021-04-02
CN202110361611.9A CN115184888A (zh) 2021-04-02 2021-04-02 一种目标检测方法及装置

Publications (1)

Publication Number Publication Date
WO2022206700A1 true WO2022206700A1 (zh) 2022-10-06

Family

ID=83458022

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2022/083456 WO2022206700A1 (zh) 2021-04-02 2022-03-28 一种目标检测方法及装置

Country Status (2)

Country Link
CN (1) CN115184888A (zh)
WO (1) WO2022206700A1 (zh)

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117220802A (zh) * 2023-10-10 2023-12-12 白盒子(上海)微电子科技有限公司 一种用于5g低采样率场景的时延估计方法和装置

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN104580052A (zh) * 2013-10-20 2015-04-29 杨瑞娟 一种新的多载波直扩信号在雷达通信一体化中的应用
CN105676199A (zh) * 2015-12-31 2016-06-15 天津大学 基于通信/雷达一体化的单通道lte雷达***
CN107786480A (zh) * 2017-09-28 2018-03-09 清华大学 雷达通信一体化信号生成方法及装置
CN110794403A (zh) * 2019-10-30 2020-02-14 南京航空航天大学 一种汽车防撞雷达探测-通信一体化功能实现方法
CN111836296A (zh) * 2020-06-29 2020-10-27 西南电子技术研究所（中国电子科技集团公司第十研究所） 一体化波形通信测量***

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN104580052A (zh) * 2013-10-20 2015-04-29 杨瑞娟 一种新的多载波直扩信号在雷达通信一体化中的应用
CN105676199A (zh) * 2015-12-31 2016-06-15 天津大学 基于通信/雷达一体化的单通道lte雷达***
CN107786480A (zh) * 2017-09-28 2018-03-09 清华大学 雷达通信一体化信号生成方法及装置
CN110794403A (zh) * 2019-10-30 2020-02-14 南京航空航天大学 一种汽车防撞雷达探测-通信一体化功能实现方法
CN111836296A (zh) * 2020-06-29 2020-10-27 西南电子技术研究所（中国电子科技集团公司第十研究所） 一体化波形通信测量***

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
PAN CHENGKANG, WANG AILING;LIU JIANJUN;WANG QIXING;WANG YAJUAN;MA LIANG: "Technology Analysis of Integration of Wireless Sensing and Communication", RADIO COMMUNICATIONS TECHNOLOGY, vol. 47, no. 2, 28 February 2021 (2021-02-28), pages 143 - 148, XP055974474, ISSN: 1003-3114, DOI: 10.3969/j.issn.1003-3114.2021.02.003 *
SARIEDDEEN HADI; SAEED NASIR; AL-NAFFOURI TAREQ Y.; ALOUINI MOHAMED-SLIM: "Next Generation Terahertz Communications: A Rendezvous of Sensing, Imaging, and Localization", IEEE COMMUNICATIONS MAGAZINE., IEEE SERVICE CENTER, PISCATAWAY., US, vol. 58, no. 5, 9 June 2020 (2020-06-09), US , pages 69 - 75, XP011791995, ISSN: 0163-6804, DOI: 10.1109/MCOM.001.1900698 *

Also Published As

Publication number Publication date
CN115184888A (zh) 2022-10-14

Similar Documents

Publication Publication Date Title
US8441393B2 (en) Orthogonal frequency division multiplexing (OFDM) radio as radar
US11936477B2 (en) Sub-band and multi-band coded OFDM for high-resolution radar
Mao et al. Waveform design for joint sensing and communications in millimeter-wave and low terahertz bands
CN102411140B (zh) 一种基于多电台信号的无源雷达处理方法
CN105162493A (zh) 多普勒域和延迟域二维捕获方法及装置
WO2022171084A1 (zh) 接入方法、装置、通信设备及可读存储介质
Zeng et al. Joint radar-communication: Low complexity algorithm and self-interference cancellation
KR20080066122A (ko) 무선통신시스템을 위한 훈련 시퀀스
US8928524B1 (en) Method and system for enhancing data rates
Liu et al. Joint range and angle estimation for an integrated system combining MIMO radar with OFDM communication
WO2022052960A1 (zh) 线性调频通信***及其信道估计方法、装置、介质和芯片
CN104580052A (zh) 一种新的多载波直扩信号在雷达通信一体化中的应用
WO2022206700A1 (zh) 一种目标检测方法及装置
Gu et al. Waveform design for integrated radar and communication system with orthogonal frequency modulation
CN104836769A (zh) 一种基于共轭结构前导的联合定时与频率同步方法
CN115685115A (zh) 基于叠加式通信感知一体化波形的目标融合感知方法
Tang et al. Performance analysis of LoRa modulation with residual frequency offset
US20240072947A1 (en) Transmitter and receiver for high-resolution sub-band coded time frequency waveform radar system with bandpass sampling
Zhang et al. Application of LFM-CPM signal into a DFRC system based on circulating code array
Wu et al. A sensing integrated DFT-spread OFDM system for terahertz communications
CN114866124A (zh) 基于时延对齐调制的通感一体化信号设计与波束赋形方法
CN102171938A (zh) 移除经调制的单频干扰
Ni et al. Receiver Design in Full-Duplex Joint Radar-Communication Systems
An et al. A turbo coded LoRa-index modulation scheme for IoT communication
Qi et al. Circular Time Shift Modulation for robust underwater acoustic communications in doubly spread channels

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 22778882

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 22778882

Country of ref document: EP

Kind code of ref document: A1