WO2021036392A1 - 一种开关变换器及其控制方法 - Google Patents

一种开关变换器及其控制方法 Download PDF

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WO2021036392A1
WO2021036392A1 PCT/CN2020/094483 CN2020094483W WO2021036392A1 WO 2021036392 A1 WO2021036392 A1 WO 2021036392A1 CN 2020094483 W CN2020094483 W CN 2020094483W WO 2021036392 A1 WO2021036392 A1 WO 2021036392A1
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inductor
switch
current
switching tube
voltage
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PCT/CN2020/094483
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English (en)
French (fr)
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卢鹏飞
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广州金升阳科技有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel

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  • the invention relates to a switching power supply, in particular to a switching converter circuit and a control method thereof.
  • Figure 1 shows a traditional step-down circuit.
  • the effective value of the current is larger when the circuit works in discontinuous mode.
  • the MOS tube Q1 is hard-switched, and the conduction loss of the diode D1 is large.
  • Figure 2 is a step-down circuit with synchronous rectification function. Compared with the traditional step-down circuit, the conduction loss in the freewheeling phase is reduced. In FCCM mode, the ZVS of the MOS transistor Q1 can also be turned on, because the ZVS of the MOS transistor Q1 is turned on And the range of high-efficiency work is relatively narrow, so in the wide voltage input range, there is a problem of low overall efficiency in the full load range.
  • FIG 3 is an abstract drawing of the Taiwanese patent application number 100137357 and the invention titled "Control Method and Device for Improving the Light Load Efficiency of Synchronous Buck Converter". This patent solves the problem of reduced efficiency in Figure 2 at light load because In the FCCM mode, the driving of the MOS transistor Q1 is complementary to the driving of the MOS transistor Q2, and the negative current of the inductor L is very large.
  • Figure 4 is the timing diagram of the patent.
  • the MOS transistor Q2 when the current of the inductor L drops to zero at light load, the MOS transistor Q2 is turned off to avoid the energy loss caused by the negative current of the inductor L; next, the MOS transistor Q1 Before turning on, the MOS transistor Q2 is controlled to be turned on for a short time, so that the current of the inductor L is a suitable negative value, which creates conditions for realizing the ZVS turn-on of the MOS transistor Q1 and improves the light load efficiency.
  • MOS transistor Q2 instead of MOS transistor Q1
  • the current capacity of MOS transistor Q2 should be large during circuit design, making the input capacitance Ciss of MOS transistor Q2 Relatively large, MOS transistor Q2 switches twice in a cycle, driving loss is large, and in high frequency applications, in order to achieve short-term conduction of MOS transistor Q2, the driving time of MOS transistor Q2 is too short, which makes it difficult for the control IC to issue a too short drive Signal; and from t3 to t4, the drain of the MOS transistor Q2 resonates at a high frequency, and there is an EMI problem.
  • the present invention proposes a switching converter and its control method, which solves the problem of low comprehensive efficiency of the step-down circuit in a wide voltage input range and a full load range.
  • the patent "Control Method and Device for Improving the Light-Load Efficiency of Synchronous Buck Converter” the driving loss of MOS transistor Q2 is large, and the high frequency resonance of the drain of MOS transistor Q2 from t3 to t4 leads to EMI and narrow pulse driving from t4 to t5. Questions that are difficult to issue.
  • a switching converter including input power supply positive Vin, output voltage positive Vo, power supply common ground GND, switching tube Q1, switching tube Q2, switching tube Q3, unidirectional conducting device D1, inductor L1 and capacitor C1; switching tube Q1
  • the drain of the switch is connected to the positive Vin of the input power supply
  • the source of the switch Q1 and the drain of the switch Q2 are connected to one end of the inductor L1
  • the other end of the inductor L1 is connected to the source of the switch Q3 and the unidirectional conduction device
  • the cathode of D1, the drain of the switch Q3 and one end of the capacitor C1 are connected to the positive output voltage Vo
  • the anode of the unidirectional conducting device D1 the source of the switch Q2 and the other end of the capacitor C1 are connected to the power supply common ground GND.
  • the switching tube Q1, the switching tube Q2, and the switching tube Q3 are MOS tubes, triodes or IGBTs.
  • the unidirectional conduction device D1 is a diode, a MOS tube, a triode or an IGBT
  • the cathode of the diode, the drain of the MOS tube, the collector of the triode or the drain of the IGBT is the cathode of the unidirectional conduction device D1
  • the The anode, the drain of the MOS tube, the emitter of the triode, or the source of the IGBT is the anode of the unidirectional conducting device D1.
  • the control method of the above switching converter includes the following steps:
  • t0 ⁇ t1 stage at t0, the switch Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current i L of the inductor L1 rises, and the switch Q1 is turned off at the time t1;
  • Stage t1 ⁇ t2 After the switching tube Q1 is turned off, the current i L of the inductor L1 charges the output capacitance Coss1 of the switching tube Q1 and discharges the output capacitance Coss2 of the switching tube Q2. At t2, the voltage at one end of the inductor L1 is changed by Vin Reduce to 0V, switch Q2 realizes ZVS opening;
  • a voltage across the inductor L1 is Vo of, demagnetization of the inductor L1, the current i L decreases the inductor current i L L1 is zero at time t3, the switch Q3 is turned off, the switching transistor Q3 to achieve ZCS is off;
  • the output capacitor Coss3 of the switch Q3 and the inductor L1 form a resonant network and start to resonate.
  • the unidirectional conduction device D1 clamps the Coss3 voltage to resonate At the end, since the current of the inductor L1 is very small, it quickly drops to zero;
  • Stage t4 ⁇ t5 At t4, the switch Q3 turns on, the switch Q3 realizes ZCS turn on, and Vo gives the inductor L1 reverse excitation, and when the reverse excitation current i L satisfies the ZVS turn-on condition of the switch Q1, it turns off at t5 Switch tube Q2;
  • Stage t5 ⁇ t0+T The current i L of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1.
  • the voltage at one end of the inductor L1 rises from 0V to Vin ,
  • the switch tube Q1 realizes ZVS opening;
  • the current i L of the inductor L1 at t3 is characterized in that: at t3, the current i L of the inductor L1 drops to 0 ⁇ 1A, and the switching tube is turned off. Q3.
  • the present invention also provides a second switching converter with the same inventive concept, and the technical solution is as follows:
  • a switching converter including input power supply positive Vin, output voltage positive Vo, power supply common ground GND, switching tube Q1, switching tube Q2, switching tube Q3, inductor L1 and capacitor C1; the drain of switching tube Q1 is connected to The input power is positive Vin, the source of the switch Q1 and the drain of the switch Q2 are connected to one end of the inductor L1, the other end of the inductor L1 is connected to the source of the switch Q3, the drain of the switch Q3 and the capacitor C1 One end of is connected to the positive output voltage Vo, and the source of the switch Q2 and the other end of the capacitor C1 are connected to the power supply common ground GND.
  • the switching tube Q1, the switching tube Q2, and the switching tube Q3 are MOS tubes, triodes or IGBTs.
  • the control method of the above switching converter includes the following steps:
  • t0 ⁇ t1 stage at t0, the switch Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current i L of the inductor L1 rises, and the switch Q1 is turned off at the time t1;
  • Stage t1 ⁇ t2 After the switching tube Q1 is turned off, the current i L of the inductor L1 charges the output capacitance Coss1 of the switching tube Q1 and discharges the output capacitance Coss2 of the switching tube Q2. At t2, the voltage at one end of the inductor L1 is changed by Vin Reduce to 0V, switch Q2 realizes ZVS opening;
  • a voltage across the inductor L1 is Vo of, demagnetization of the inductor L1, the current i L decreases the inductor current i L L1 is zero at time t3, the switch Q3 is turned off, the switching transistor Q3 to achieve ZCS is off;
  • Stage t3 ⁇ t4 At t3, the output capacitor Coss3 of the switch Q3 and inductor L1 form a resonant network to start resonance, so that the voltage at the other end of the inductor L1 oscillates between Vo and -Vo, and then turns on at t4 according to the closed-loop control requirements Switch tube Q3;
  • Stage t4 ⁇ t5 Vo reversely excites the inductor L1, when the reverse excitation current i L meets the ZVS turn-on condition of the switch Q1, the switch Q2 is turned off at t5;
  • Stage t5 ⁇ t0+T The current i L of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1.
  • the voltage at one end of the inductor L1 rises from 0V to Vin ,
  • the switch tube Q1 realizes ZVS opening;
  • the current i L of the inductor L1 at t3 is characterized in that: at t3, the current i L of the inductor L1 drops to 0 ⁇ 1A, and the switching tube is turned off. Q3.
  • a unidirectional device refers to a device in which current can only flow from the anode to the cathode, but not from the cathode to the anode;
  • the gate of the switching tube For MOS tube, it refers to the gate, for the triode, it is the base, and for IGBT, it is the gate.
  • Other switching tubes can correspond according to the knowledge of those skilled in the art, no longer one by one. Enumerate
  • the drain of the switching tube For MOS tube, it refers to the drain, for the triode, it is the collector, and for IGBT, it is the drain.
  • Other switching tubes can correspond according to the knowledge of those skilled in the art, no longer one by one. Enumerate
  • the source of the switching tube For MOS tube, it refers to the source, for the triode, it is the emitter, and for IGBT, it is the source.
  • Other switching tubes can correspond to the knowledge of those skilled in the art, no longer one by one. Enumerate.
  • the present invention has the following beneficial effects:
  • Figure 1 is a schematic diagram of a traditional step-down circuit
  • Figure 2 is a schematic diagram of a step-down circuit with synchronous rectification function
  • Figure 3 is a schematic diagram of the step-down patented circuit with application number 100137357;
  • Figure 4 is a working sequence diagram of the step-down patented circuit with application number 100137357;
  • Figure 5 is a schematic diagram of the circuit of the first embodiment of the present invention.
  • Fig. 6 is a working sequence diagram of the first embodiment of the present invention.
  • Fig. 7 is a schematic circuit diagram of the second embodiment of the present invention.
  • Fig. 8 is a working sequence diagram of the second embodiment of the present invention.
  • Fig. 5 is a schematic circuit diagram of the first embodiment of the present invention. Including input power supply positive Vin, output voltage positive Vo, power supply common ground GND, MOS tube Q1, MOS tube Q2, MOS tube Q3, diode D1, inductor L1 and capacitor C1; the drain of MOS tube Q1 is connected to input power supply positive Vin ,
  • the source of MOS transistor Q1 and the drain of MOS transistor Q2 are connected to one end of inductor L1, the source of MOS transistor Q3 and the cathode of diode D1 are connected to the other end of inductor L1, and the drain of MOS transistor Q3 is connected to One end of the capacitor C1, the source of the MOS transistor Q2, the anode of the diode D1 and the other end of the capacitor C1 are connected to the power supply common ground GND.
  • Coss1, Coss2, and Coss3 in FIG. 5 are the output capacitors of the MOS tube Q1, the MOS tube Q2, and the MOS tube Q3, respectively.
  • the body diodes of the MOS tube Q1, the MOS tube Q2, and the MOS tube Q3 are also shown in FIG.
  • Diode D1 is only used to clamp the voltage of Coss3, and the current flowing through is very small, so diode D1 is a very small device, and diode D1 can even be removed according to actual applications.
  • the unidirectional conduction device is diode D1.
  • MOS tube, triode or IGBT can also be used.
  • the drain of MOS tube, the collector of triode or the drain of IGBT is the cathode of diode D1
  • the drain of MOS tube, triode is the anode of the diode D1.
  • FIG. 6 shows the working sequence of this embodiment, which is specifically as follows:
  • Stage t0 ⁇ t1 At t0, the MOS transistor Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current i L of the inductor L1 rises, and the MOS transistor Q1 is turned off at the time t1;
  • a voltage across the inductor L1 is Vo of, demagnetization of the inductor L1, the inductor current i L L1 is decreased, the inductor current i L L1 is zero at time t3, the MOS transistor Q3 to achieve ZCS Off Break
  • Stage t4 ⁇ t5 Vo reversely excites the inductor L1, and when the reverse excitation current i L meets the ZVS turn-on condition of the MOS transistor Q1, the MOS transistor Q2 is turned off at t5;
  • Stage t5 ⁇ t0+T The current i L of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2, and discharges the output capacitor Coss1 of the MOS transistor Q1.
  • the voltage of SW1 at the end of the inductor L1 rises from 0V to Vin, MOS tube Q1 realizes ZVS opening;
  • the T in the above t0+T represents the time length of one cycle.
  • the diode D1 Since the diode D1 flows a very small current, and the reverse withstand voltage is Vo, the Vo voltage is low, so the diode D1 is a small diode with a small current and a small withstand voltage.
  • MOS transistor Q3 The reverse withstand voltage of MOS transistor Q3 is Vo, and the reverse withstand voltage of MOS transistor Q2 is Vin.
  • MOS transistor Q3 has a much smaller withstand voltage, so under the same Rdson conditions
  • the input capacitance Ciss and output capacitance Coss3 of MOS transistor Q3 are much smaller, according to the formula The corresponding driving loss is much smaller.
  • the current i L of the inductor L1 at time t3 takes any value within 0 ⁇ 1A, and the MOS transistor Q3 is turned off.
  • the smaller the error value of the current i L the higher the turn-off efficiency of the MOS transistor Q3.
  • FIG. 7 is a schematic circuit diagram of the second embodiment of the present invention. On the basis of the first embodiment, the diode D1 is removed, and other connection relationships remain unchanged.
  • FIG 8 shows the working sequence of the second embodiment, which is specifically as follows:
  • Stage t0 ⁇ t1 At t0, the MOS transistor Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current i L of the inductor L1 rises, and the MOS transistor Q1 is turned off at the time t1;
  • a voltage across the inductor L1 is Vo of, demagnetization of the inductor L1, the current i L decreases to zero at time t3 L1 of the inductor current i L, the MOS transistor Q3 is turned off to achieve ZCS;
  • Stage t3 ⁇ t4 At t3, the output capacitor Coss3 of MOS transistor Q3 and inductor L1 form a resonant network and start to resonate, so that the voltage at the other end of the inductor L1 SW2 oscillates between Vo and -Vo, and then according to the closed-loop control requirements at time t4 Open MOS tube Q3;
  • Stage t4 ⁇ t5 Vo reversely excites the inductor L1, and when the reverse excitation current i L meets the ZVS turn-on condition of the MOS transistor Q1, the MOS transistor Q2 is turned off at t5;
  • Stage t5 ⁇ t0+T The current i L of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2, and discharges the output capacitor Coss1 of the MOS transistor Q1.
  • the voltage of SW1 at the end of the inductor L1 rises from 0V to Vin, MOS tube Q1 realizes ZVS opening;
  • Removing the diode D1 can reduce the cost, but there is resonance in the t3-t4 stage.
  • the impact of this resonance on EMI can be determined according to the actual application. If the impact is within an acceptable range, the diode D1 can be removed to reduce the cost.
  • the current i L of the inductor L1 at t3 takes any value within 0 ⁇ 1A, and the MOS transistor Q3 is turned off.
  • the smaller the error value of the current i L the higher the turn-off efficiency of the MOS transistor Q3, and when the current i When L is less than zero, the oscillation of the SW2 node during t3 to t4 will become larger.

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Abstract

本发明公开了一种开关变换器及其控制方法，开关变换器包括输入电源正、输出电压正、电源公共地、开关管Q1、开关管Q2、开关管Q3、二极管D1、电感器L1和电容器C1；开关管Q1的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和二极管D1的阴极连接到电感器L1的另一端，开关管Q3的漏极连接到电容器C1的一端，开关管Q2的源极、二极管D1的阳极和电容器C1的另一端连接到电源公共地。本发明解决了降压电路工作在宽输入电压和负载范围下综合效率偏低的问题，同时消除震荡改善EMI。

Description

一种开关变换器及其控制方法 技术领域
本发明涉及开关电源，特别涉及开关变换器电路及其控制方法。
背景技术
图1为传统的降压电路，电路工作在断续模式下电流有效值较大，MOS管Q1为硬开关，二极管D1的导通损耗大。
图2为具有同步整流功能的降压电路，相比传统的降压电路，降低了续流阶段的导通损耗，在FCCM模式还可以实现MOS管Q1的ZVS开通，由于MOS管Q1的ZVS开通及高效率工作的范围比较窄，所以在宽电压输入范围，全负载范围存在综合效率偏低的问题。
图3是申请号为100137357，发明名称为《提高同步降压转换器轻载效率之控制方法与装置》的中国台湾专利摘要附图，该专利解决了图2轻载时效率降低的问题，因为FCCM模式下MOS管Q1的驱动与MOS管Q2的驱动互补，电感器L负向电流很大。图4为该专利的时序图，如图所示，轻载时电感器L的电流下降到零时关断MOS管Q2，避免电感器L的电流为负造成能量损失；接下来在MOS管Q1导通前先控制MOS管Q2导通很短的时间，使电感器L的电流为合适的负值，为实现MOS管Q1的ZVS开通创造条件，提高轻载效率。但是对于输入输出电压之比较大和大电流输出场景，电流主要流过MOS管Q2，而不是MOS管Q1，所以在电路设计时MOS管Q2的通流能力要大，使得MOS管Q2的输入电容Ciss比较大，一个周期内MOS管Q2开关两次，驱动损耗大，同时高频应用时为了实现MOS管Q2短时导通，MOS管Q2的驱动时间太短，导致控制IC难以发出太短的驱动信号；且在t3到t4时刻MOS管Q2的漏极高频谐振，存在EMI问题。
发明内容
鉴于现有降压电路及其改进型专利电路和控制方式的技术缺陷，本发明提出一种开关变换器及其控制方法，解决了降压电路在宽电压输入范围，全负载范围综合效率偏低和专利《提高同步降压转换器轻载效率之控制方法与装置》存在MOS管Q2的驱动损耗大，t3到t4时刻MOS管Q2的漏极高频谐振导致 EMI和t4到t5时刻窄脉冲驱动难以发出的问题。
为了实现上述发明目的，本发明采用以下技术方案：
一种开关变换器，包括输入电源正Vin、输出电压正Vo、电源公共地GND、开关管Q1、开关管Q2、开关管Q3、单向导通器件D1、电感器L1和电容器C1；开关管Q1的漏极连接到输入电源正Vin，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，电感器L1的另一端连接到开关管Q3的源极和单向导通器件D1的阴极，开关管Q3的漏极与电容器C1的一端连接到输出电压正Vo，单向导通器件D1的阳极、开关管Q2的源极和电容器C1的另一端连接到电源公共地GND。
优选的，所述的开关管Q1、开关管Q2和开关管Q3为MOS管、三极管或者IGBT。
优选的，所述的单向导通器件D1为二极管、MOS管、三极管或者IGBT，二极管的阴极、MOS管的漏极、三极管的集电极或IGBT的漏极为单向导通器件D1的阴极，二极管的阳极、MOS管的漏极、三极管的发射极或IGBT的源极为单向导通器件D1的阳极。
上述开关变换器的控制方法包括如下步骤：
t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断开关管Q1；
t1～t2阶段：开关管Q1关断后，电感器L1的电流i L给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流i L下降，在t3时刻电感器L1的电流i L降为零，关断开关管Q3，开关管Q3实现ZCS关断；
t3～t4阶段：t3时刻开关管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，当电感器L1另一端的电压从Vo下降到0V，单向导通器件D1钳位了Coss3电压使谐振结束，由于电感器L1的电流很小，所以很快降为零；
t4～t5阶段：t4时刻开关管Q3导通，开关管Q3实现ZCS开通，Vo给电感器L1反向励磁，当反向励磁电流i L满足开关管Q1的ZVS开通条件时在t5时 刻关断开关管Q2；
t5～t0+T阶段：电感器L1的电流i L给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端的电压由0V上升到Vin，开关管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述开关变换器的控制方法在t3时刻电感器L1的电流i L的另一种具体实施方式，其特征在于：在t3时刻电感器L1的电流i L降为0±1A，关断开关管Q3。
本发明还提供具有相同发明构思的第二种开关变换器，技术方案如下：
一种开关变换器，包括包括输入电源正Vin、输出电压正Vo、电源公共地GND、开关管Q1、开关管Q2、开关管Q3、电感器L1和电容器C1；开关管Q1的漏极连接到输入电源正Vin，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，电感器L1的另一端连接到开关管Q3的源极，开关管Q3的漏极与电容器C1的一端连接到输出电压正Vo，开关管Q2的源极和电容器C1的另一端连接到电源公共地GND。
优选的，所述的开关管Q1、开关管Q2和开关管Q3为MOS管、三极管或者IGBT。
上述开关变换器的控制方法包括如下步骤：
t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断开关管Q1；
t1～t2阶段：开关管Q1关断后，电感器L1的电流i L给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流i L下降，在t3时刻电感器L1的电流i L降为零，关断开关管Q3，开关管Q3实现ZCS关断；
t3～t4阶段：t3时刻开关管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，使电感器L1另一端的电压在Vo到-Vo之间震荡，然后根据闭环控制要求在t4时刻开通开关管Q3；
t4～t5阶段：Vo给电感器L1反向励磁，当反向励磁电流i L满足开关管Q1的ZVS开通条件时在t5时刻关断开关管Q2；
t5～t0+T阶段：电感器L1的电流i L给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端的电压由0V上升到Vin，开关管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述开关变换器的控制方法在t3时刻电感器L1的电流i L的另一种具体实施方式，其特征在于：在t3时刻电感器L1的电流i L降为0±1A，关断开关管Q3。
术语含义说明：
单向导通器件是指电流仅能从阳极流向阴极，而不能从阴极流向阳极的器件；
开关管的栅极：对于MOS管指的是栅极、对于三极管指的是基极、对于IGBT指的是栅极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举；
开关管的漏极：对于MOS管指的是漏极、对于三极管指的是集电极、对于IGBT指的是漏极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举；
开关管的源极：对于MOS管指的是源极、对于三极管指的是发射极、对于IGBT指的是源极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举。
与现有技术相比，本发明具有如下有益效果：
1)即使在轻载时也实现了开关管Q1和开关管Q2的ZVS开通，开关管Q3的ZCS开关，且电感器L1的负向电流小，实现了全输入电压范围、全负载综合效率高；
2)开关管Q3耐压为Vo，比开关管Q2耐压小很多，所以相同Rdson下开关管Q3的输入电容Ciss比开关管Q2的输入电容Ciss小很多，一个周期内开关管Q2和开关管Q3各一次的驱动损耗低于图4中开关管Q2的两次驱动损耗；
3)消除了图4中t3～t4阶段的谐振，降低了EMI；而本发明第二种开关变 换器方案虽然没有消除t3～t4阶段的谐振，由于谐振的总能量来自输出电容Coss3，而图4中谐振的总能量来自Coss2，实际上Coss3小于Coss2，所以本发明第二种开关变换器方案谐振的能量小，衰减快，EMI也会改善；
4)取开关管Q2和开关管Q3同时导通时间段，即交集部分为电感器L1反向励磁，解决了图4中开关管Q2第二次驱动时间太短，导致控制IC难以发出太短的驱动信号问题。
附图说明
图1为传统的降压电路原理图；
图2为具有同步整流功能的降压电路原理图；
图3为申请号100137357的降压专利电路原理图；
图4为申请号100137357的降压专利电路工作时序图；
图5为本发明第一实施例电路原理图；
图6为本发明第一实施例工作时序图；
图7为本发明第二实施例电路原理图；
图8为本发明第二实施例工作时序图。
具体实施方式
第一实施例
图5为本发明的第一实施例的电路原理图。包括输入电源正Vin、输出电压正Vo、电源公共地GND、MOS管Q1、MOS管Q2、MOS管Q3、二极管D1、电感器L1和电容器C1；MOS管Q1的漏极连接到输入电源正Vin，MOS管Q1的源极和MOS管Q2的漏极连接到电感器L1的一端，MOS管Q3的源极和二极管D1的阴极连接到电感器L1的另一端，MOS管Q3的漏极连接到电容器C1的一端，MOS管Q2的源极、二极管D1的阳极和电容器C1的另一端连接到电源公共地GND。
图5中的Coss1、Coss2和Coss3分别为MOS管Q1、MOS管Q2和MOS管Q3的输出电容，此外，图5中还画出了MOS管Q1、MOS管Q2和MOS管Q3的体二极管。
二极管D1只是用来钳位Coss3的电压，流过的电流特别小，所以二极管D1是一个很小的器件，可根据实际应用甚至去掉二极管D1。
需要说明的是：
本实施例中单向导通器件为二极管D1，当然也可以采用MOS管、三极管或者IGBT，MOS管的漏极、三极管的集电极或IGBT的漏极为二极管D1的阴极，MOS管的漏极、三极管的发射极或IGBT的源极为二极管D1的阳极。
将MOS管Q1、MOS管Q2和MOS管Q3替换为三极管或IGBT等其它类型的开关管为本领域技术人员的惯用手段。
图6所示为本实施例的工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断MOS管Q1；
t1～t2阶段：MOS管Q1关断后，电感器L1的电流i L给MOS管Q1的输出电容Coss1充电，给MOS管Q2的输出电容Coss2放电，在t2时刻电感器L1一端SW1的电压由Vin降为0V，MOS管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电感器L1的电流i L下降，在t3时刻电感器L1的电流i L降为零，MOS管Q3实现ZCS关断；
t3～t4阶段：t3时刻MOS管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，当电感器L1另一端SW2的电压从Vo下降到0V，二极管D1钳位了Coss3电压使谐振结束，由于电感器L1的电流很小，所以电感器L1的电流i L很快降为零，在t4时刻MOS管Q3实现ZCS开通；
t4～t5阶段：Vo给电感器L1反向励磁，当反向励磁电流i L满足MOS管Q1的ZVS开通条件时在t5时刻关断MOS管Q2；
t5～t0+T阶段：电感器L1的电流i L给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端SW1的电压由0V上升到Vin，MOS管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
由于电路为周期性的工作，上述t0+T中的T代表的含义为一个周期的时间长度。
由于二极管D1流过很小的电流，且反向耐压为Vo，Vo电压低，所以二极管D1为一个小电流小耐压的小二极管。
MOS管Q3的反向耐压为Vo，MOS管Q2的反向耐压为Vin，从器件选型上来说，相对于MOS管Q2，MOS管Q3的耐压小很多，所以在相同Rdson的条件下，MOS管Q3的输入电容Ciss和输出电容Coss3要小很多，根据公式
Figure PCTCN2020094483-appb-000001
对应的驱动损耗小很多。
特别的，t3时刻电感器L1的电流i L取0±1A内的任意值，MOS管Q3关断，电流i L的误差值越小，MOS管Q3的关断效率越高。
第二实施例
图7为本发明的第二实施例的电路原理图，在第一实施例的基础上，去掉二极管D1，其它连接关系不变。
图8所示为第二实施例工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断MOS管Q1；
t1～t2阶段：MOS管Q1关断后，电感器L1的电流i L给MOS管Q1的输出电容Coss1充电，给MOS管Q2的输出电容Coss2放电，在t2时刻电感器L1一端SW1的电压由Vin降为0V，MOS管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流i L下降，在t3时刻电感器L1的电流i L降为零，MOS管Q3实现ZCS关断；
t3～t4阶段：t3时刻MOS管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，使电感器L1另一端SW2的电压在Vo到-Vo之间震荡，然后根据闭环控制要求在t4时刻开通MOS管Q3；
t4～t5阶段：Vo给电感器L1反向励磁，当反向励磁电流i L满足MOS管Q1的ZVS开通条件时在t5时刻关断MOS管Q2；
t5～t0+T阶段：电感器L1的电流i L给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端SW1的电压由0V上升到Vin，MOS管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
去掉二极管D1可降低成本，但在t3～t4阶段存在谐振，可根据实际应用来确定此谐振对EMI的影响，如果此影响在一个可接受的范围就可以去掉二极管D1降低成本。
特别的，t3时刻电感器L1的电流i L取0±1A内的任意值，MOS管Q3关断，电流i L的误差值越小，MOS管Q3的关断效率越高，且当电流i L小于零时t3～t4阶段的SW2结点震荡会变大一些。
上述实施方式不应视为对本发明的限制，本发明的保护范围应当以权利要求所限定的范围为准。对于本技术领域的普通技术人员来说，在不脱离本发明的精神和范围内，还可以做出若干等同替换、改进和润饰，如根据应用场合的不同，通过器件的简单串并联等手段对电路微调，这些改进和润饰也应视为本发明的保护范围。

Claims (9)

  1. 一种开关变换器，其特征在于：包括输入电源正Vin、输出电压正Vo、电源公共地GND、开关管Q1、开关管Q2、开关管Q3、单向导通器件D1、电感器L1和电容器C1；开关管Q1的漏极连接到输入电源正Vin，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，电感器L1的另一端连接到开关管Q3的源极和单向导通器件D1的阴极，开关管Q3的漏极与电容器C1的一端连接到输出电压正Vo，单向导通器件D1的阳极、开关管Q2的源极和电容器C1的另一端连接到电源公共地GND。
  2. 根据权利要求1所述的开关变换器，其特征在于：所述的开关管Q1、开关管Q2和开关管Q3为MOS管、三极管或者IGBT。
  3. 根据权利要求1所述的开关变换器，其特征在于：所述的单向导通器件D1为二极管、MOS管、三极管或者IGBT，二极管的阴极、MOS管的漏极、三极管的集电极或IGBT的漏极为单向导通器件D1的阴极，二极管的阳极、MOS管的漏极、三极管的发射极或IGBT的源极为单向导通器件D1的阳极。
  4. 一种权利要求1至3任一项所述的开关变换器的控制方法，其特征在于：
    t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断开关管Q1；
    t1～t2阶段：开关管Q1关断后，电感器L1的电流i L给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
    t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流i L下降，在t3时刻电感器L1的电流i L降为零，关断开关管Q3，开关管Q3实现ZCS关断；
    t3～t4阶段：t3时刻开关管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，当电感器L1另一端的电压从Vo下降到0V，二极管D1钳位了Coss3电压使谐振结束，由于电感器L1的电流很小，所以很快降为零；
    t4～t5阶段：t4时刻开关管Q3导通，开关管Q3实现ZCS开通，Vo给电感器L1反向励磁，当反向励磁电流i L满足开关管Q1的ZVS开通条件时在t5时 刻关断开关管Q2；
    t5～t0+T阶段：电感器L1的电流i L给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端的电压由0V上升到Vin，开关管Q1实现ZVS开通；
    本周期结束，下一个工作周期开始，重复上面的阶段。
  5. 根据权利要求4所述的控制方法，其特征在于：所述的t3时刻电感器L1的电流i L降为0±1A，关断开关管Q3。
  6. 一种开关变换器，其特征在于：包括包括输入电源正Vin、输出电压正Vo、电源公共地GND、开关管Q1、开关管Q2、开关管Q3、电感器L1和电容器C1；开关管Q1的漏极连接到输入电源正Vin，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，电感器L1的另一端连接到开关管Q3的源极，开关管Q3的漏极与电容器C1的一端连接到输出电压正Vo，开关管Q2的源极和电容器C1的另一端连接到电源公共地GND。
  7. 根据权利要求6所述的开关变换器，其特征在于：所述的开关管Q1、开关管Q2和开关管Q3为MOS管、三极管或者IGBT。
  8. 一种权利要求6或7所述的开关变换器的控制方法，其特征在于：
    t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流i L上升，在t1时刻关断开关管Q1；
    t1～t2阶段：开关管Q1关断后，电感器L1的电流i L给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
    t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流i L下降，在t3时刻电感器L1的电流i L降为零，关断开关管Q3，开关管Q3实现ZCS关断；
    t3～t4阶段：t3时刻开关管Q3的输出电容Coss3与电感器L1组成谐振网络开始谐振，使电感器L1另一端的电压在Vo到-Vo之间震荡，然后根据闭环控制要求在t4时刻开通开关管Q3；
    t4～t5阶段：Vo给电感器L1反向励磁，当反向励磁电流i L满足开关管Q1的ZVS开通条件时在t5时刻关断开关管Q2；
    t5～t0+T阶段：电感器L1的电流i L给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t0+T时刻电感器L1一端的电压由0V上升到Vin，开关管Q1实现ZVS开通；
    本周期结束，下一个工作周期开始，重复上面的阶段。
  9. 根据权利要求8所述的控制方法，其特征在于：所述的t3时刻电感器L1的电流i L降为0±1A，关断开关管Q3。
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