WO2020183553A1 - Dc power supply device, power conversion device, and refrigeration cycle device - Google Patents
Dc power supply device, power conversion device, and refrigeration cycle device Download PDFInfo
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- WO2020183553A1 WO2020183553A1 PCT/JP2019/009535 JP2019009535W WO2020183553A1 WO 2020183553 A1 WO2020183553 A1 WO 2020183553A1 JP 2019009535 W JP2019009535 W JP 2019009535W WO 2020183553 A1 WO2020183553 A1 WO 2020183553A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
- H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
- H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
- H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
- H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M5/4585—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only having a rectifier with controlled elements
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0003—Details of control, feedback or regulation circuits
- H02M1/0006—Arrangements for supplying an adequate voltage to the control circuit of converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0003—Details of control, feedback or regulation circuits
- H02M1/0009—Devices or circuits for detecting current in a converter
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
- H02M1/007—Plural converter units in cascade
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
- H02M1/4208—Arrangements for improving power factor of AC input
- H02M1/4233—Arrangements for improving power factor of AC input using a bridge converter comprising active switches
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
- H02M3/1586—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel switched with a phase shift, i.e. interleaved
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1588—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
- H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Definitions
- the present invention relates to a DC power supply device that converts alternating current into direct current and outputs it, a power conversion device including a direct current power supply device, and a refrigeration cycle device including a power conversion device.
- Patent Document 1 discloses a DC power supply device in which a technique called synchronous rectification is applied to reduce loss.
- Patent Document 1 is intended to reduce the switching loss due to the reverse recovery current generated in the high-side diode, and its application to a product having a wide range of power consumption has not been studied. That is, the technique shown in Patent Document 1 has a problem that the improvement of efficiency according to the power consumption is not sufficient.
- the present invention has been made in view of the above, and an object of the present invention is to obtain a DC power supply device that can be applied to a product having a wide range of power consumption and can improve efficiency according to the power consumption. ..
- the DC power supply device includes a plurality of metal oxide semiconductor field effect transistors having a parasitic diode inside, and the plurality of metal oxide semiconductor field effect transistors are bridged. It is equipped with a rectifier circuit that is connected and connected to an AC power supply via a reactor. Further, the DC power supply device includes a capacitor connected between the output terminals of the rectifier circuit, a first detector for detecting the polarity or phase of the power supply voltage which is the output voltage of the AC power supply, and a first current flowing through the rectifier circuit. A first current detector for detecting the above.
- the control unit controls the first operation mode in which the metal oxide semiconductor field effect transistor is turned on at the timing when the current flows in the same direction as the parasitic diode, and the pulse width modulation control of the metal oxide semiconductor field effect transistor. It has a second operation mode for boosting the voltage across the capacitor, and activates the first operation mode when the power consumption is low, and activates the second operation mode when the power consumption is high.
- the DC power supply device According to the DC power supply device according to the present invention, it can be applied to a product having a wide range of power consumption, and has an effect that efficiency can be improved according to the power consumption.
- FIG. 1 is a first diagram used for explaining the operation of a metal oxide semiconductor field effect transistor (Metal Oxide Semiconductor Field Effect Transistor: MOSFET) in the DC power supply device according to the first embodiment.
- FIG. 2 used to explain the operation of the MOSFET in the DC power supply device according to the first embodiment.
- a block diagram showing an example of a hardware configuration that realizes the functions of the control unit shown in FIG. A block diagram showing another example of a hardware configuration that realizes the function of the control unit shown in FIG.
- the figure used for the explanation of the operation mode in the control part of Embodiment 1. A circuit diagram showing a circuit configuration of a drive circuit in the rectifier circuit of the second embodiment.
- the figure used for the explanation of the detection timing of the direct current in Embodiment 4. The figure which shows the 1st operation waveform which operates the switching element of the rectifier circuit in Embodiment 4.
- FIG. 1 is a diagram showing a configuration of a motor drive device including a power conversion device according to the first embodiment.
- the motor drive device 100 according to the first embodiment includes a reactor 2, a rectifier circuit 3, a capacitor 4, an inverter circuit 5, a power supply voltage zero cross detection circuit 7, a DC voltage detection circuit 8, and a DC. It includes current detection circuits 9 and 10 and a control unit 11.
- the reactor 2, the rectifier circuit 3, the capacitor 4, the power supply voltage zero cross detection circuit 7, the DC voltage detection circuit 8, and the control unit 11 constitute the DC power supply device 50 according to the first embodiment.
- the DC power supply device 50, the inverter circuit 5, the DC current detection circuit 10, and the control unit 11 constitute the power conversion device 70 according to the first embodiment.
- the control unit 11 is configured as a common control unit that controls each of the rectifier circuit 3 and the inverter circuit 5, but two rectifier circuits 3 and the inverter circuit 5 are individually controlled. It may be configured to include a control unit.
- the reactor 2 may be an external component when viewed from the DC power supply device 50.
- the capacitor 4 is a component of the DC power supply device 50 here, it may be a component of the inverter circuit 5.
- the rectifier circuit 3 has switching elements SW3-1 to SW3-4 which are MOSFETs, and is connected to the AC power supply 1 via the reactor 2.
- the four switching elements SW3-1 to SW3-4 are bridge-connected, and each of the switching elements SW3-1 to SW3-4 has a parasitic diode D3-1 to D3-4 inside.
- the switching elements SW3-1 to SW3-4 the switching elements SW3-1 and SW3-2 may be referred to as "upper arm switching element” or simply "upper arm”.
- the switching elements SW3-3 and SW3-4 may be referred to as a "lower arm switching element” or simply a "lower arm”.
- each of the set of switching elements SW3-1 and SW3-3 and the set of switching elements SW3-2 and SW3-4 may be referred to as "upper and lower arms”.
- the rectifier circuit 3 further includes drive circuits 3a to 3d for driving each of the switching elements SW3-1 to SW3-4.
- the rectifier circuit 3 converts the AC power of the AC power supply 1 supplied via the reactor 2 into DC power.
- the reactor 2 is connected between the AC power supply 1 and the rectifier circuit 3 and smoothes the power supply current Is supplied from the AC power supply 1 to the rectifier circuit 3.
- the capacitor 4 is a smoothing capacitor.
- the capacitor 4 has a positive electrode terminal and a negative electrode terminal.
- the capacitor 4 is connected between the output terminals of the rectifier circuit 3 and smoothes the output voltage of the rectifier circuit 3.
- the inverter circuit 5 has six switching elements SW5-1 to SW5-6 connected by a bridge, and is connected to the output end of the rectifier circuit 3.
- the rectifier circuit 3 and the inverter circuit 5 are connected by a pair of DC bus lines 16a on the positive side and a DC bus line 16b on the negative side.
- the connection point on the positive electrode side of the capacitor 4 on the positive DC bus 16a is shown as P
- the connection point on the negative electrode side of the capacitor 4 on the negative DC bus 16b is shown as N.
- the inverter circuit 5 converts the DC power output by the rectifier circuit 3 into three-phase AC power and supplies it to the motor 6 which is a three-phase motor. That is, the motor 6 is driven by the three-phase electric power output from the inverter circuit 5.
- an example of the switching elements SW5-1 to SW5-6 is the MOSFET shown in the figure, but an insulated gate bipolar transistor (IGBT) may be used.
- Each of the switching elements SW5-1 to SW5-6 includes diodes D5-1 to D5-6 connected in antiparallel.
- the anti-parallel connection means that the drain of the MOSFET and the cathode of the diode are connected, and the source of the MOSFET and the anode of the diode are connected.
- the power supply voltage zero cross detection circuit 7 determines the polarity of the power supply voltage Vs based on the power supply voltage Vs which is the output voltage of the AC power supply 1.
- the DC voltage detection circuit 8 detects the voltage across the capacitor 4.
- the DC current detection circuit 9 detects the current flowing through the rectifier circuit 3.
- the DC current detection circuit 10 detects the current flowing through the inverter circuit 5.
- Each detection signal of the power supply voltage zero cross detection circuit 7, the DC voltage detection circuit 8, the DC current detection circuit 9, and the DC current detection circuit 10 is input to the control unit 11.
- the power supply voltage zero cross detection circuit 7 may be referred to as a "first detector”
- the DC voltage detection circuit 8 may be referred to as a "second detector”.
- the current flowing through the rectifier circuit 3 may be referred to as a "first current”, and the DC current detection circuit 9 for detecting the first current may be referred to as a "first current detector”.
- the current flowing through the inverter circuit 5 may be referred to as a "second current”, and the DC current detection circuit 10 for detecting the second current may be referred to as a "second current detector”.
- the DC current detection circuit 9 amplifies the voltage drop in the shunt resistor 9a caused by the DC current flowing in the shunt resistor 9a by the amplifier 9b and gives it to the control unit 11.
- the DC current detection circuit 10 amplifies the voltage drop in the shunt resistor 10a caused by the DC current flowing through the shunt resistor 10a by the amplifier 10b and gives it to the control unit 11.
- the amplifiers 9b and 10b can be configured by using an operational amplifier.
- the DC current detection circuits 9 and 10 of the first embodiment are configured to detect the DC current by amplifying the voltage drop in the shunt resistor 9a or the shunt resistor 10a, but the current transformer capable of detecting the DC current. (Curent Tracer: CT) may be used. Further, in FIG. 1, one resistor is shown as the shunt resistor 9a or the shunt resistor 10a, but two or three or more resistors may be provided.
- the control unit 11 includes analog-digital (AD) conversion units 12 to 14.
- the detection signal of the DC voltage detection circuit 8 is an analog signal, and is converted into a digital signal indicating a detection value by the AD conversion unit 12.
- Each of the detection signals of the DC current detection circuits 9 and 10 is also an analog signal, and each of them is converted into a digital signal indicating a detection value by the AD conversion units 13 and 14.
- the control unit 11 turns on or off the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 based on the polarity discrimination signal zc by the power supply voltage zero cross detection circuit 7 and the detection value of the DC current detection circuit 9.
- the control signals Sa to Sd to be controlled are generated.
- the control signals Sa to Sd are output to the corresponding drive circuits 3a to 3d.
- the drive circuits 3a to 3d drive the corresponding switching elements SW3-1 to SW3-4 based on the control signals Sa to Sd, respectively.
- control unit 11 controls the on / off of the switching elements SW5-1 to SW5-6 of the inverter circuit 5 based on the detected values of the DC voltage detection circuit 8 and the DC current detection circuit 10.
- the control signals UP, VP, WP, UN, VN, and WN are output to the corresponding drive circuits 5a to 5f.
- the drive circuits 5a to 5f drive the corresponding switching elements SW5-1 to SW5-6 based on the control signals UP, VP, WP, UN, VN, and WN, respectively.
- the motor 6 is connected to the output end of the inverter circuit 5.
- the inverter circuit 5 drives the motor 6 by supplying AC power to the motor 6.
- An example of the motor 6 is a permanent magnet electric motor.
- the motor 6 includes a stator 6a with a three-phase Y-type connection in which three-phase windings including U-phase, V-phase, and W-phase windings are connected in a Y shape, and a permanent magnet rotor 6b.
- One ends of the U-phase, V-phase, and W-phase windings are connected to each other, and the other ends of the respective windings form a U terminal, a V terminal, and a W terminal, respectively.
- the stator 6a of the motor 6 exemplifies a stator of a three-phase Y-type connection, but may be composed of a stator of a three-phase delta-type connection. Needless to say, the phase current information that can be detected from the DC current changes between the three-phase Y-type connection and the three-phase ⁇ -type connection.
- connection point between the switching element SW5-1 and the switching element SW5-4 of the inverter circuit 5 is connected to the terminal U of the motor 6, and the connection point between the switching element SW5-2 and the switching element SW5-5 is connected to the terminal V. It is connected, and the connection point between the switching element SW5-3 and the switching element SW5-6 is connected to the terminal W.
- the current flowing through the U-phase winding is called a U-phase current and is represented by "Iu”.
- V-phase current and W-phase current are referred to as V-phase current and W-phase current, respectively, and are represented by "Iv” and "Iw", respectively.
- FIG. 2 is a first diagram used for explaining the operation of the MOSFET in the DC power supply device 50 according to the first embodiment.
- FIG. 3 is a second diagram used for explaining the operation of the MOSFET in the DC power supply device 50 according to the first embodiment.
- FIG. 2 shows the current flow when the gate is off. Gate-off is a state in which a current flows through a parasitic diode formed inside the MOSFET. Further, FIG. 3 shows the current flow when the gate is on. Gate-on is a state in which a voltage is applied between the gate and the source of the MOSFET and a current is flowing to the FET main body side of the MOSFET. When the gate is turned on, the current flows on the FET main body side because the impedance on the FET main body side is lower than that on the parasitic diode.
- synchronous rectification is a control method in which the MOSFET is turned on at the timing when a current flows in the same direction as the parasitic diode to flow a current to the FET main body side.
- a forward voltage drop (so-called Vf) in the parasitic diode occurs by about 3V. Therefore, the voltage characteristics on the parasitic diode side are worse than those on the FET main body side, and the loss is large. Further, in the case of a MOSFET formed of a WBG semiconductor, the on-resistance of the FET body is also low. Therefore, by positively utilizing synchronous rectification, further reduction in loss can be achieved.
- the MOSFET formed of the WBG semiconductor has been described here, the present invention is not limited to this. Any element may be used as long as it can utilize synchronous rectification.
- FIG. 4 is a block diagram showing an example of a hardware configuration that realizes the function of the control unit 11 of FIG.
- FIG. 5 is a block diagram showing another example of a hardware configuration that realizes the function of the control unit 11 of FIG.
- the processor 300 that performs the calculation, the memory 302 that stores the program read by the processor 300, and the input / output of the signal are performed.
- the configuration can include the interface 304.
- the processor 300 may be a computing means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor).
- the memory 302 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EEPROM (Erasable Programmable ROM), or an EEPROM (registered trademark) (Electrically EPROM). It can be exemplified.
- the memory 302 stores a program that executes the control function of the control unit 11.
- the processor 300 sends and receives necessary information via the interface 304, the processor 300 executes a program stored in the memory 302, and the processor 300 refers to a table stored in the memory 302. Control can be performed.
- the calculation result by the processor 300 can be stored in the memory 302.
- the processor 300 and the memory 302 shown in FIG. 4 may be replaced with the processing circuit 305 as shown in FIG.
- the processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.
- the information input to the processing circuit 305 and the information output from the processing circuit 305 can be performed via the interface 304.
- FIG. 6 is a diagram showing a first operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment.
- FIG. 6A shows the waveform of the power supply voltage Vs which is the output voltage of the AC power supply 1
- FIG. 6B shows the polarity discrimination signal zc output from the power supply voltage zero cross detection circuit 7. ing.
- the polarity of the polarity discrimination signal zc is treated as positive in the direction of the arrow of the power supply voltage Vs shown in FIG. According to this definition, the polarity of the polarity discrimination signal zc is as shown in FIG.
- the polarity discrimination signal zc becomes “H” (high: High).
- the polarity discrimination signal zc becomes “L” (low).
- FIG. 6C shows the waveform of the direct current Idc1.
- the polarity of the direct current Idc1 is positive in the direction of the arrow shown in FIG.
- the current in the direction of the arrow is detected negatively in the shunt resistor 9a in relation to the ground (GND) position shown in FIG. 1, but is polar in the processing of the control unit 11.
- the reversal can be done freely.
- the waveform of the direct current Idc1 is shown as a positive electrode for ease of viewing.
- the cause of the overcurrent can be determined only by the polarity of the direct current Idc1.
- the direct current Idc1 flows in the direction of the arrow shown in FIG. 1, the negative electrode property is defined.
- a positive overcurrent it can be determined that the overcurrent is an overcurrent generated when the upper and lower arms are short-circuited. This is because when the upper and lower arms are short-circuited, a positive DC current Idc1 flows.
- a negative electrode overcurrent it can be determined that the overcurrent is an overcurrent generated at the time of overload. This is because the negative electrode direct current Idc1 flows when the load is overloaded.
- the DC current detection circuit 9 is inserted between the common GND terminal and the connection point on the source terminal side of the switching elements SW3 and SW3-4 which are MOSFETs.
- the power conversion device 70 shown in FIG. 1 can be configured as a non-isolated power conversion device using a common GND.
- the device can be configured without using an insulating circuit such as an insulating amplifier or a photocoupler, so that signal transmission / reception can be speeded up. Further, a highly responsive power conversion device can be provided at low cost.
- the switching elements SW3-1 to SW3-4 are controlled based on the polarity discrimination signal zc by the power supply voltage zero cross detection circuit 7 and the detection value of the DC current detection circuit 9.
- the control unit 11 turns on the switching element described below when the detected value of the direct current Idc1 is equal to or higher than a predetermined threshold value, and turns off the switching element described below when the detected value is equal to or lower than the threshold value.
- This threshold is indicated by "Idc_on" in FIG. 6 (c).
- the on threshold value and the off threshold value are shown with the same value, but the present invention is not limited to this. Hysteresis may be provided between the on threshold and the off threshold so that the on threshold and the off threshold are different. Further, although it has been described here that it is above the threshold value and below the threshold value, it goes without saying that the threshold value may be exceeded or below the threshold value.
- Two of SW3-1 to 3-4 are switching elements that are turned on when the detected value of the DC current Idc1 exceeds a predetermined threshold value. These two are determined based on the polarity of the AC power supply 1 detected by the power supply voltage zero cross detection circuit 7, that is, the polarity discrimination signal zc.
- the switching elements SW3-1 and SW3-4 are controlled to be ON as shown in FIGS. 6 (d) and 6 (g).
- the switching elements SW3 and SW3-3 are controlled to be ON as shown in FIGS. 6 (e) and 6 (f).
- the rectifier circuit 3 is composed of MOSFETs, as shown in FIG. 1, a parasitic diode is added to the MOSFETs. Therefore, even if the ON signal is not supplied to the gate of the MOSFET, the parasitic diode is turned on and a current called a capacitor input flows from the AC power supply 1. By turning on the MOSFET at the timing when this current flows, as shown in FIG. 2, the loss of the rectifier circuit 3 can be reduced by the synchronous rectification in which the current flows to the FET main body side having a low on-resistance.
- FIG. 7 is a diagram showing a second operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment.
- the arrangement of the signal waveforms in FIG. 7 is the same as that in FIG.
- the switching elements SW3-1 and SW3-3 are controlled by the control unit 11 for pulse width modulation (PWM).
- PWM control it is possible to generate a sine wave waveform of the current flowing from the AC power supply 1 to the rectifier circuit 3.
- boost control that raises the voltage between the connection point PN, which is the voltage across the capacitor 4, to ⁇ 2 times or more the effective value of the power supply voltage Vs is also possible.
- the power supply factor can be controlled to be substantially 1, and the power supply harmonic current can be controlled to be approximately 0.
- the voltage across the capacitor 4 can be boosted by PWM-controlling the switching elements SW3-1 and SW3-3. Therefore, by increasing the boosted voltage, an operating range with large power consumption is satisfied. be able to.
- the current capacities of the inverter circuit 5 and the motor 6 are fixed, and the current capacities of the switching elements SW5-1 to SW5-6 in the inverter circuit 5 are selected according to the current capacity of the motor 6. For example, when a switching element having a large current capacity is used in the inverter circuit 5, the conduction loss and the switching loss generated when the switching elements SW5-1 to SW5-6 are turned on tend to be larger than those of the switching element having a small current capacity. There is.
- the inverter circuit 5 becomes a very expensive circuit. Therefore, in order to provide a product with a wide range of power consumption to a user at an appropriate selling price, it is required to select a switching element having a small current capacity.
- the technique of the first embodiment is applied. Specifically, in the operating range where the power consumption is large, the switching elements SW3-1 and SW3-3 constituting the rectifier circuit 3 are PWM-controlled. The output voltage of the rectifier circuit 3 is boosted by PWM control. Since the output voltage of the rectifier circuit 3 is boosted, it is possible to increase the power consumption by the amount of the voltage increase without increasing the current capacity of the inverter circuit 5 and the motor 6. As a result, it is possible to obtain a power conversion device that can be applied to a highly efficient product.
- the technique of the first embodiment is applied, a switching element having a small current capacity can be used. As a result, the loss of the inverter circuit 5 can be reduced in the operating range where the power consumption is small. Then, it is possible to configure the power conversion device with less loss under any condition from the condition that the range of power consumption is narrow to the condition that the range of power consumption is wide. Further, it is possible to increase the maximum load amount by boosting the output voltage of the rectifier circuit 3. Therefore, according to the power conversion device according to the first embodiment, it is possible to achieve both the energy saving characteristic by improving the efficiency at the time of low load and the high power characteristic with the maximum load raised.
- FIG. 8 is a diagram showing a third operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment.
- FIG. 8 shows an operation waveform different from that of FIGS. 6 and 7.
- the signal waveforms of (a), (b), (d) to (g) excluding (c) are shown for 1.5 cycles.
- a waveform of the power supply current Is is shown between (b) and (d).
- the power supply short-circuit operation is performed for only one pulse in the power supply half cycle, but the present invention is not limited to this. Even if a power short-circuit operation of a plurality of pulses is performed in a power supply half cycle, it can be carried out without deviating from the object of the present invention.
- the threshold value Idc_on in FIG. 8 is set higher than that in FIG. In FIG. 8, when “H” is output from the polarity discrimination signal zc when the power supply voltage Vs is positive, the switching element SW3-3 is first turned on and the power supply is short-circuited after a preset time has elapsed. The state is formed. Then, the switching elements SW3-1 and SW3-4, which are positively operated, are turned on by the threshold value Idc_on. In this operation, if the switching elements SW3-1 and SW3-4 are turned off at the threshold value Idc_on, the synchronous rectification state ends early, and the time for the current to flow through the parasitic diode becomes long. Therefore, in FIG.
- the timing for ending the synchronous rectification that is, the timing for turning off the switching elements SW3-1 and SW3-4 is set by the threshold value Idc_off.
- the state of synchronous rectification in which a current flows through the FET main body can be maintained for a longer time than when the switching elements SW3-1 and SW3-4 are turned off at the threshold value Idc_on.
- FIG. 9 is a diagram showing a fourth operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment.
- FIG. 9 shows an operation waveform different from each of the drawings of FIGS. 6 to 8.
- the arrangement of the signal waveforms in FIG. 9 is the same as that in FIG.
- the value of the threshold value Idc_on in FIG. 9 is the same as that in FIG.
- the switching element SW3-1 is controlled to be OFF only when the switching element SW3-3 is controlled to be ON.
- the control unit 11 sets a threshold value different from the threshold value Idc_on, and controls the switching elements SW3-1 and SW3-3 so that the ON operation is switched from the switching element SW3-1 to the switching element SW3-3. become.
- the control of FIGS. 6 and 7 is selectively carried out by the control unit 11. Specifically, the control unit 11 performs the control shown in FIG. 6 when the power supplied from the AC power supply 1 is lower than the preset set value, and when it is higher than the preset set value, FIG. 7 The control shown in is performed.
- the case where the power supplied from the AC power supply 1 is lower than the preset set value means that the power consumption of a load (not shown) connected to the motor 6 is low. Such a state is called “low power consumption”.
- the case where the power supplied from the AC power source 1 is higher than the preset set value means the case where the power consumption of the load is high. Such a state is called "high power consumption”.
- an intermediate state between the time of low power consumption and the time of high power consumption is called “medium power consumption”.
- the power supplied from the rectifier circuit 3 to the inverter circuit 5 also becomes large. Therefore, each state according to the level of the power supplied from the rectifier circuit 3 to the inverter circuit 5 may be referred to as “low power consumption”, “medium power consumption”, and “high power consumption”. ..
- FIG. 10 is a diagram used for explaining the operation mode in the control unit 11 of the first embodiment.
- the control unit 11 has at least three operation modes, that is, a first operation mode, a second operation mode, and a third operation mode.
- the first operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the synchronous rectification shown in FIG.
- the second operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the PWM control shown in FIG. 7.
- the third operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the control shown in FIG.
- the control unit 11 activates any one of these three operation modes according to the power consumption of the load to control each switching element of the rectifier circuit 3.
- Ps_PS is a threshold value at the time of shifting from the first operation mode to the third operation mode.
- Ps_SP is a threshold value when shifting from the third operation mode to the first operation mode
- Ps_SF is a threshold value when shifting from the third operation mode to the second operation mode. It is a threshold value
- Ps_FS is a threshold value when shifting from the second operation mode to the third operation mode.
- the first operation mode is an operation mode carried out at the time of low power consumption
- the second operation mode is an operation mode carried out at the time of high and low power consumption
- the third operation mode is a medium power consumption. This is the operation mode that is sometimes implemented.
- the two threshold values "Ps_PS" and “Ps_SP” are provided with a hysteresis characteristic in order to stabilize the control. The same applies to the two thresholds "Ps_SF" and "Ps_FS".
- control shown in FIG. 8 is performed in the third operation mode, but the control shown in FIG. 9 may be performed instead of FIG.
- the power supplied from the AC power source 1 is used for determining whether the power consumption Ps of the load is low power consumption, medium power consumption, or high power consumption.
- the electric power obtained by calculation from the detected values detected by the DC current detection circuit 9 and the DC voltage detection circuit 8 may be used. Further, the determination may be made based on a detection value in a power detection circuit (not shown) that detects the power supplied from the AC power supply 1.
- switching may be performed based on the rotation speed of the motor 6 or the motor current flowing through the motor 6.
- any detected value may be used as long as it does not deviate from the object of the present invention.
- the first operation mode in which the MOSFET is turned on at the timing when the current flows in the same direction as the parasitic diode and the MOSFET are pulse-width. It has a second operation mode in which modulation control is performed to boost the voltage across the capacitor. Then, the control unit provided in the DC power supply device activates the first operation mode when the power consumption is low, and activates the second operation mode when the power consumption is high. As a result, it can be applied to products having a wide range of power consumption, and efficiency can be improved according to the power consumption.
- the DC power supply device when the power consumption is low, the number of switchings can be reduced as much as possible, and a low loss effect can be obtained by synchronous rectification of the MOSFET. Further, when the power consumption is high, the DC voltage can be boosted by PWM control of the MOSFET, and the current flowing through the inverter circuit and the motor can be reduced. This makes it possible to apply an inexpensive MOSFET with low loss. As a result, it is possible to provide a low-loss power conversion device.
- FIG. 11 is a circuit diagram showing a circuit configuration of drive circuits 3a to 3d in the rectifier circuit 3 of the second embodiment.
- the drive circuit 3a is a drive circuit that drives the switching element SW3-1 in which the source terminal of the MOSFET is electrically connected to the AC power supply 1 via the reactor 2.
- the drive circuit 3b is a drive circuit that drives the switching element SW3-2 in which the source terminal of the MOSFET is electrically connected to the AC power supply 1 via the reactor 2.
- the same or equivalent components as those in the first embodiment shown in FIG. 1 are designated by the same reference numerals, and redundant description will be omitted.
- the source terminal of the MOSFET is electrically connected to the common GND via the DC current detection circuit 9. Therefore, the drive circuits 3c and 3d can receive the drive power of the MOSFET directly from the control power supply 22 connected to the common GND.
- the switching elements SW3-1 and SW3-2 since the source terminal of the MOSFET is electrically connected to the AC power supply 1, the driving power cannot be directly supplied from the control power supply 22.
- the drive circuit 3a is connected to the control power supply 22 via a series circuit of the diode 20a and the resistor 21a.
- the control power supply 22 and the drive circuit 3a are electrically connected, and the drive power for driving the MOSFET at the gate can be supplied.
- the electric power supplied at this time is charged in the capacitor 23a. Therefore, even when the diode 20a is turned off and the driving power is not supplied, the driving power can be supplied to the switching element SW3-1.
- the drive circuit 3b is also connected to the control power supply 22 via a series circuit of the diode 20b and the resistor 21b.
- Diodes 20a and 20b and capacitors 23a and 23b are important key components for this bootstrap.
- the diodes 20a and 20b will be referred to as “boot diodes 20a and 20b”
- the capacitors 23a and 23b will be referred to as “boot capacitors 23a and 23b”.
- the operation will be described focusing on the boot diode 20a and the boot capacitor 23a.
- the negative electrode terminal of the boot capacitor 23a is connected to the source terminal of the switching element SW3-1 which is a MOSFET, and the positive electrode terminal of the boot capacitor 23a is connected to the control power supply 22 via a series circuit of the diode 20a and the resistor 21a.
- the boot capacitor 23a When the switching element SW3-3 connected to the negative electrode side of the boot capacitor 23a is turned on, the boot capacitor 23a is charged with electric charge and the driving power is secured.
- the rectifier circuit 3 of the first embodiment even if the MOSFET of the switching element SW3-3 is not turned on, the parasitic diode D3-3 is turned on when the charging current from the AC power supply 1 flows through the capacitor 4. , The charge of the boot capacitor 23a is realized. Charging of the boot capacitor 23b is also realized by the same operation.
- the voltage characteristics of the parasitic diode are poor and the voltage effect of the parasitic diode is large. Therefore, the charge voltage of the boot capacitor needs to be as high as the Vf of the parasitic diode. Due to this Vf, the drive of the MOSFET may become unstable.
- the term "unstable" as used herein means that, for example, the balance of switching characteristics such as dv / dt is lost between the switching element of the upper arm and the switching element of the lower arm connected in series. If the switching characteristics are out of balance, the behavior of the rectifier circuit 3 may become unstable.
- FIG. 12 is a diagram showing an operation waveform in the charge mode according to the second embodiment. The arrangement of the signal waveforms in FIG. 12 is the same as that in FIG.
- the switching element SW3-4 is controlled to be turned on in a half cycle in which the polarity of the power supply voltage Vs is positive.
- the negative electrode terminal of the boot capacitor 23b is electrically connected to the GND potential via the FET main body side of the MOSFET instead of the parasitic diode.
- the switching element SW3-3 In the half cycle when the polarity of the power supply voltage Vs is negative, the switching element SW3-3 is controlled to be ON, and the charging operation of the boot capacitor 23a is improved.
- the subsequent operation is the same as that of the boot capacitor 23b, and the duplicate description is omitted.
- the switching elements SW3-1 and SW3-2 can be driven by charging the boot capacitor with an electric charge. Further, since the switching elements SW3-1 and SW3-2 can be driven without providing a control power supply exclusively for the drive circuits 3a and 3b, the DC power supply device and the power conversion device can be miniaturized.
- FIG. 13 is a diagram showing an operation waveform of the rectifier circuit 3 according to the third embodiment.
- the operation waveform diagram corresponding to FIG. 13 is FIG. 7.
- the switching element that operates in PWM is replaced every power cycle, which is the cycle of the AC power supply 1. More specifically with reference to FIG. 13, first, the switching element SW3-1 and the switching element SW3-4 operate as a set, and the switching element SW3-2 and the switching element SW3-3 operate as a set. Then, when the switching element SW3-1 is in PWM operation, the switching element SW3-4 continues to operate independently of PWM in order to realize synchronous rectification. Further, after the PWM operation of the switching element SW3-1, the switching element SW3-4 becomes the PWM operation, and the switching element SW3-1 changes to the operation of continuing the on operation unrelated to the PWM. These operations are the same in the set of the switching element SW3-2 and the switching element SW3-3.
- an embodiment in which the PWM operations are not alternately replaced is also effective from another viewpoint.
- the heat dissipation design can be integrated into the two switching elements.
- the reliability of the entire device can be increased by increasing the reliability of the two switching elements. That is, in terms of product quality, the operation shown in FIG. 7 or the operation shown in FIG. 13 may be performed, and any operation may be performed.
- the set of two switching elements is exchanged for each cycle of the AC power supply 1, but the present invention is not limited to this.
- the replacement cycle may be replaced every two cycles, or may be replaced at a cycle of an integral multiple of 3 or more.
- the variation of the four switching elements causes an imbalance in which the positive and negative waveforms of the input current have an asymmetrical shape in the input current flowing from the AC power supply 1.
- the variation between the elements is made uniform. As a result, the waveform imbalance in the input current is suppressed.
- the switching element by PWM operation is used as a reference so that the relative imbalance is suppressed with reference to any of the four switching elements. You may adjust the on-time of. Even with such control, the imbalance of the waveform in the input current can be suppressed. As a result, it is possible to suppress the generation of harmonic currents that do not originally occur, such as the even-numbered order of the AC power supply 1. Further, since the harmonic current can be reduced, it is possible to provide a power conversion device that does not cause a failure to the AC power supply 1.
- the imbalance is not suppressed even if the correction is made to a certain threshold value, it is judged that one of the switching elements is open failure.
- An open failure cannot be detected by normal control because the path through which the current flows is open.
- the imbalance generation pattern changes. From this change, it is possible to determine which switching element has an open failure.
- the open failure of the switching element may be determined only by the amount of imbalance generated without performing the correction up to a certain threshold value. Further, in order to detect imbalance, an operation mode in which correction is not performed may be intentionally or forcibly created. In addition, although some other applications can be assumed, any method may be used as long as it does not deviate from the object of the present invention.
- FIG. 14 is a diagram showing an operation waveform of the rectifier circuit 3 according to the fourth embodiment.
- FIG. 14 shows the drive waveforms of the switching elements SW3-1 and SW3-3 when the PWM operation shown in FIG. 7 or 13 is performed, and the waveforms of the carriers used for generating the PWM signal.
- “Sa” is a control signal for controlling the switching element SW3-1
- “Sc” is a control signal for controlling the switching element SW3-3.
- a method called complementary PWM is adopted.
- control signals having an exclusive relationship are generated so that the switching elements of the upper and lower arms are not turned on at the same time.
- one or both of the peaks and valleys of the triangular wave which is the reference of the PWM cycle, is set as the control reference time. Specifically, the time when the control algorithm starts operating is based on a peak or a valley. Further, in the AD conversion units 12 to 14 that perform AD conversion of the detection signal, a peak or a valley is used as a reference for the time for holding the detection signal, the timing time for starting the AD conversion, and the like.
- the modulated wave signal is changed every carrier cycle.
- the duty of the modulated wave signal is set with reference to the peak of the carrier of the triangular wave.
- the control signal Sa that controls the switching element SW3-1 has a waveform that is symmetrical with respect to the dotted line in the figure with respect to the mountain.
- the control signal Sc that controls the switching element SW3-3 has a left-right asymmetric waveform with respect to the dotted line in the figure with respect to the valley. Further, the center of the control signal Sc is deviated from the carrier cycle.
- the detected current will not be at the center timing of turning on the switching element, and a detection error will occur.
- This detection error causes an imbalance in the input current and can also be a trigger to break the control algorithm.
- the switching elements SW3-1 and SW3-3 of the upper and lower arms connected in series always have a direct current at the center timing of the period during which the control signals Sa and Sc are turned on.
- the control unit 11 is configured to generate a timing signal for sampling in a time shorter than the half cycle of the carrier so as to be detected. As a result, an unintended detection error can be suppressed, and the operational stability of the rectifier circuit 3 can be improved.
- the central timing of the period during which the control signals Sa and Sc are turned on does not have to be the central timing of the period during which the control signals Sa and Sc are turned on.
- the time ratio between the first time between the time when the on-pulse rises and the time when the direct current is detected and the second time which is the width of the on-pulse is the control signal Sa As long as it is the same as Sc, it may be a pulse head portion or a pulse rear portion.
- the switching element SW3-1 and the switching element SW3-3 do not have the same duty, and the on-pulse widths of the two are different. Therefore, it is complicated to calculate the timing time at which the above time ratios are the same. Therefore, the method of detecting the central portion of the pulse may be simple.
- FIG. 15 is a diagram used for explaining the detection timing of the direct current Idc1 in the fourth embodiment.
- FIG. 15 shows a waveform of the direct current Idc1, and black circle dots representing the time of detection timing of the direct current Idc1 are plotted on the waveform.
- the plot interval of the black circle points is a half carrier period.
- FIG. 15 is an example of a case where the current detection sampling is relatively rough.
- the processing start timing in the processor is indicated by a downward arrow.
- the intervals between the arrows are two carrier cycles. These two carrier cycles are the cycles of the control algorithm, and thinning control is performed by thinning out some of the detected currents detected in the carrier half cycle.
- the current detection control is performed on the assumption that the sampling cycle of the current detection cannot be shortened. Specifically, the current detected current is estimated from the past detected current. Then, when it is predicted that the threshold will be exceeded between the previous sampling time and the current sampling time, the sampling interval is controlled so that the time predicted to exceed the threshold becomes the sampling time, and forced at the sampling time. The control is to detect the current. Alternatively, the switching element may be forcibly switched on and off at the predicted time. With these controls, even with the thinning control as shown in FIG. 15, current sampling can be performed at the required time, and the required current can be detected relatively accurately. This makes it possible to provide a power conversion device having a higher performance boosting function while using an inexpensive processor.
- FIG. 16 is a diagram showing a first operation waveform for operating the switching element of the rectifier circuit 3 according to the fourth embodiment.
- the operation waveform diagram corresponding to FIG. 16 is FIG. 6 or FIG.
- the waveform of the phase ⁇ s of the power supply voltage Vs is shown. That is, in the rectifier circuit 3 of the fourth embodiment that controls the operation of FIG. 16, a function of detecting or calculating the phase ⁇ s of the power supply voltage Vs is added.
- control signal Sa that controls the on / off of the switching element SW3-1 is controlled based on the phase ⁇ s of the power supply voltage Vs.
- a simultaneous on prevention phase angle ( ⁇ 5 degrees in the figure) is set in order to avoid simultaneous on of the switching element when the polarity is reversed. As a result, it is possible to reliably suppress the short circuit between the upper and lower arms.
- FIG. 17 is a diagram showing a second operation waveform for operating the switching element of the rectifier circuit 3 in the fourth embodiment.
- the control signal Sa that controls the switching element SW3-1 is controlled based on the phase ⁇ s of the power supply voltage Vs, but in FIG. 17, the control signal Sd that controls the switching element SW3-4 is controlled. ing.
- synchronous rectification can be performed while the parasitic diode of the MOSFET on the upper arm side maintains the reverse recovery characteristic. This makes it possible to operate the power conversion device more stably.
- FIG. 18 is a diagram showing a configuration of a motor drive device including the power conversion device according to the fifth embodiment.
- the DC power supply device 50 in the configuration of the motor drive device 100 according to the first embodiment shown in FIG. 1, the DC power supply device 50 is replaced with the DC power supply device 50A to convert power.
- the device 70 has been replaced by the power converter 70A.
- a relay 15 connected in parallel to the reactor 2 and a reactor 2a connected in series to the reactor 2 are added.
- the reactor 2a may be referred to as a "second reactor".
- other configurations are the same as or equivalent to the configuration of the first embodiment, and the same or equivalent components are designated by the same reference numerals, and duplicate description will be omitted.
- the relationship between the inductance value L0 of the reactor 2 and the inductance value La of the second reactor 2a is set to L0> La.
- the opening and closing of the relay 15 is controlled by the control unit 11 according to the operation modes of the switching elements SW3-1 to SW3-4.
- the relay 15 is controlled to the open state.
- the inductance value of the circuit is L0 + La.
- the relay 15 is controlled to the closed state. At this time, the inductance value of the circuit becomes the minimum necessary inductance value La. Thereby, the inductance value of the reactor suitable for the second operation mode can be set.
- the second operation mode is an operation mode in which PWM control is performed. If a reactor having an inductance value larger than necessary is used in PWM control, the voltage across the reactor becomes too large, which may cause insulation deterioration. Therefore, if a common reactor is used in the first operation mode and the second operation mode, more expensive parts are required.
- the relay 15 can control the inductance value of a suitable reactor according to the operation mode. As a result, the rectifier circuit 3 can be operated with higher efficiency and higher power factor. In addition, the rectifier circuit 3 can be operated with less generation of harmonic current. Moreover, the power conversion device can be realized with inexpensive and small parts.
- the relay 15 is used to switch the inductance value of the reactor, but the configuration is not limited to this. Any means may be used as long as it is a switch capable of short-circuiting both ends of the reactor 2.
- FIG. 19 is a diagram showing a configuration of a motor drive device including the power conversion device according to the sixth embodiment.
- the DC power supply device 50A in the configuration of the motor drive device 100A according to the fifth embodiment shown in FIG. 18, the DC power supply device 50A is replaced with the DC power supply device 50B to convert power.
- the device 70A has been replaced by the power converter 70B.
- the rectifier circuit 3 is replaced with the rectifier circuit 3A, and the reactor 2a'that is connected in parallel to the reactor 2 is added.
- the reactor 2a' may be referred to as a "third reactor".
- the rectifier circuit 3A is composed of 6 switching elements.
- a switching element SW3-1'having a parasitic diode D3-1'inside and a switching element SW3-3' having a parasitic diode D3-3'inside are added to the rectifier circuit 3A.
- drive circuits 3a'and 3c'that drive each of the switching elements SW3-1'and SW3-3' are added to the rectifier circuit 3A.
- the control unit 11 outputs control signals Sa'and Sc'for controlling each of the switching elements SW3-1'and SW3-3'.
- Other configurations are the same as or equivalent to the configuration of the fifth embodiment, and the same or equivalent components are designated by the same reference numerals, and duplicate description will be omitted.
- the reactor 2a is connected to the connection point of the switching elements SW3-1 and SW3-3 which are the element pairs connected in series, and the reactor 2a'is the switching element SW3-1 which is the element pair connected in series. It is connected to the connection point of', SW3-3'.
- the set of switching elements SW3-1 and SW3-3 may be referred to as a "first element pair", and the set of switching elements SW3-1'and SW3-3' may be referred to as a "second element pair".
- Reactor 2a' is composed of the same inductance value as reactor 2a. If the nominal inductance values are the same, the two inductance values can be regarded as equal, and even if there are measurement errors, errors due to aging, errors due to deterioration, etc., they may be treated as the same. Further, the switching element SW3-1 and the switching element SW3-1'operate in a state in which the phases are reversed by 180 degrees. The relationship with the switching elements SW3-3 and SW3-3'is the same. The control of operating in a state where the phase is inverted by 180 degrees is a control method known as interleaving, and the description thereof is omitted here.
- the DC power supply device by configuring the DC power supply device as an interleaved configuration, it is possible to configure it by using a reactor having a smaller inductance value.
- the number of inductors increases, since it can be configured with an inexpensive reactor, it is possible to suppress the harmonic current while suppressing the increase in cost.
- a general six-element module for an inverter can be applied as the rectifier circuit 3A.
- versatility and parts procurement can be improved, and a DC power supply device can be constructed at low cost.
- the current ripple during PWM operation can be substantially halved and noise can be reduced by adopting the interleaved configuration. Further, since the heat life of the parts can be equalized by dispersing the calorific value, the life of the DC power supply device can be extended at low cost.
- FIG. 20 is a diagram showing a configuration of a motor drive device including the power conversion device according to the seventh embodiment.
- the DC power supply device 50 is replaced with the DC power supply device 50C to convert power.
- the device 70 has been replaced by the power converter 70C.
- a thermistor 16 connected in series to the reactor 2 and a relay 15 connected in parallel to the thermistor 16 are added.
- Other configurations are the same as or equivalent to the configuration of the first embodiment, the same or equivalent components are designated by the same reference numerals, and duplicate description is omitted.
- the thermistor 16 is a PTC (Positive Temperature Coafficient) thermistor having a positive temperature coefficient.
- FIG. 21 is a diagram showing a configuration example of the refrigeration cycle device according to the eighth embodiment.
- the refrigeration cycle device 120 shown in FIG. 21 is an application example of the motor drive device according to the first to seventh embodiments.
- FIG. 21 illustrates a separate type air conditioner, it is not limited to the separate type.
- the refrigeration cycle device 120 is not limited to the air conditioner and can be applied to devices having a refrigeration cycle such as a refrigerator and a freezer.
- the motor drive device 100 according to the first to fourth embodiments is used for the refrigeration cycle device 120, but the present invention is not limited to this.
- any one of the motor drive devices 100A to 100C according to the fifth to seventh embodiments may be used for the refrigeration cycle device 120.
- the refrigeration cycle device 120 includes a compressor 101, a four-way valve 102, an outdoor heat exchanger 103, an expansion valve 104, an indoor heat exchanger 105, a refrigerant pipe 106, and a motor drive device. It has 100.
- the compressor 101, the four-way valve 102, the outdoor heat exchanger 103, the expansion valve 104, and the indoor heat exchanger 105 form a refrigeration cycle to which the compressor 101 is attached via the refrigerant pipe 106.
- a compression mechanism 107 for compressing the refrigerant and a motor 6 for operating the compression mechanism 107 are provided inside the compressor 101 in the refrigeration cycle device 120.
- the motor 6 of the compressor 101 is electrically connected to the motor drive device 100.
- the motor drive device 100 is used to drive the motor 6 used in the compressor 101 that compresses the refrigerant.
- the AC power supply 1 is described as a single-phase power supply, but the present invention is not limited to this. If the effects of each of the above-described embodiments can be obtained, a three-phase power supply or a multi-phase power supply may be used instead of the single-phase power supply. This kind of deformation can be done flexibly.
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Abstract
This DC power supply device (50) comprises: a rectifier circuit (3) which includes a plurality of MOSFETs each having a parasitic diode therein, the plurality of MOSFETs being bridge-connected, and which is connected to an AC power supply (1) via a reactor (2); a capacitor (4) connected between output terminals of the rectifier circuit (3); a power supply voltage zero cross detection circuit (7) which detects the polarity of a power supply voltage which is an output voltage of the AC power supply (1); and a direct current detection circuit (9) which detects a first current flowing through the rectifier circuit (3). The control unit (11) has a first operation mode in which synchronous rectification is performed to turn on the MOSFET at a timing when a current flows in the same direction as the parasitic diode, and a second operation mode in which the MOSFET is PWM-controlled to boost the voltage across the capacitor (4), activates the first operation mode during low power consumption, and activates the second operation mode during high power consumption.
Description
本発明は、交流を直流に変換して出力する直流電源装置、直流電源装置を備える電力変換装置、及び電力変換装置を備える冷凍サイクル装置に関する。
The present invention relates to a DC power supply device that converts alternating current into direct current and outputs it, a power conversion device including a direct current power supply device, and a refrigeration cycle device including a power conversion device.
下記特許文献1には、同期整流と称されている技術を適用して損失低減を図った直流電源装置が開示されている。
Patent Document 1 below discloses a DC power supply device in which a technique called synchronous rectification is applied to reduce loss.
しかしながら、上記特許文献1は、ハイサイドのダイオードに発生する逆回復電流によるスイッチング損失の低減を図るものであり、消費電力の範囲が広い製品への適用についての検討は為されていない。即ち、特許文献1に示される技術では、消費電力に応じた効率の改善が充分ではないという課題が存在する。
However, Patent Document 1 is intended to reduce the switching loss due to the reverse recovery current generated in the high-side diode, and its application to a product having a wide range of power consumption has not been studied. That is, the technique shown in Patent Document 1 has a problem that the improvement of efficiency according to the power consumption is not sufficient.
本発明は、上記に鑑みてなされたものであって、消費電力の範囲が広い製品に対して適用でき、消費電力に応じた効率の改善を可能とする直流電源装置を得ることを目的とする。
The present invention has been made in view of the above, and an object of the present invention is to obtain a DC power supply device that can be applied to a product having a wide range of power consumption and can improve efficiency according to the power consumption. ..
上述した課題を解決し、目的を達成するため、本発明に係る直流電源装置は、寄生ダイオードを内部に有する金属酸化物半導体電界効果トランジスタを複数備え、複数の金属酸化物半導体電界効果トランジスタがブリッジ接続され、リアクタを介して交流電源に接続される整流回路を備える。また、直流電源装置は、整流回路の出力端子間に接続されるコンデンサと、交流電源の出力電圧である電源電圧の極性又は位相を検出する第1の検出器と、整流回路に流れる第1電流を検出する第1の電流検出器と、を備える。制御部は、寄生ダイオードと同方向に電流が流れるタイミングで金属酸化物半導体電界効果トランジスタをオン動作させる同期整流を行う第1の動作モードと、金属酸化物半導体電界効果トランジスタをパルス幅変調制御してコンデンサの両端電圧を昇圧する第2の動作モードと、を有し、低消費電力時には、第1の動作モードを起動し、高消費電力時には、第2の動作モードを起動する。
In order to solve the above-mentioned problems and achieve the object, the DC power supply device according to the present invention includes a plurality of metal oxide semiconductor field effect transistors having a parasitic diode inside, and the plurality of metal oxide semiconductor field effect transistors are bridged. It is equipped with a rectifier circuit that is connected and connected to an AC power supply via a reactor. Further, the DC power supply device includes a capacitor connected between the output terminals of the rectifier circuit, a first detector for detecting the polarity or phase of the power supply voltage which is the output voltage of the AC power supply, and a first current flowing through the rectifier circuit. A first current detector for detecting the above. The control unit controls the first operation mode in which the metal oxide semiconductor field effect transistor is turned on at the timing when the current flows in the same direction as the parasitic diode, and the pulse width modulation control of the metal oxide semiconductor field effect transistor. It has a second operation mode for boosting the voltage across the capacitor, and activates the first operation mode when the power consumption is low, and activates the second operation mode when the power consumption is high.
本発明に係る直流電源装置によれば、消費電力の範囲が広い製品に対して適用でき、消費電力に応じた効率の改善が可能になるという効果を奏する。
According to the DC power supply device according to the present invention, it can be applied to a product having a wide range of power consumption, and has an effect that efficiency can be improved according to the power consumption.
以下に添付図面を参照し、本発明の実施の形態に係る直流電源装置、電力変換装置及び冷凍サイクル装置について詳細に説明する。なお、以下の実施の形態により、本発明が限定されるものではない。
The DC power supply device, the power conversion device, and the refrigeration cycle device according to the embodiment of the present invention will be described in detail with reference to the accompanying drawings below. The present invention is not limited to the following embodiments.
実施の形態1.
図1は、実施の形態1に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図1において、実施の形態1に係るモータ駆動装置100は、リアクタ2と、整流回路3と、コンデンサ4と、インバータ回路5と、電源電圧ゼロクロス検出回路7と、直流電圧検出回路8と、直流電流検出回路9,10と、制御部11とを備える。これらの構成部のうち、リアクタ2、整流回路3、コンデンサ4、電源電圧ゼロクロス検出回路7、直流電圧検出回路8、及び制御部11は、実施の形態1に係る直流電源装置50を構成する。また、直流電源装置50、インバータ回路5、直流電流検出回路10、及び制御部11は、実施の形態1に係る電力変換装置70を構成する。なお、図1において、制御部11は、整流回路3及びインバータ回路5のそれぞれを制御する共通の制御部として構成されているが、整流回路3及びインバータ回路5のそれぞれを個別に制御する2つの制御部を備えて構成されていてもよい。また、リアクタ2は、直流電源装置50から見て外部の構成要素であってもよい。また、ここでは、コンデンサ4を直流電源装置50の構成要素としているが、インバータ回路5の構成要素であってもよい。Embodiment 1.
FIG. 1 is a diagram showing a configuration of a motor drive device including a power conversion device according to the first embodiment. In FIG. 1, themotor drive device 100 according to the first embodiment includes a reactor 2, a rectifier circuit 3, a capacitor 4, an inverter circuit 5, a power supply voltage zero cross detection circuit 7, a DC voltage detection circuit 8, and a DC. It includes current detection circuits 9 and 10 and a control unit 11. Among these components, the reactor 2, the rectifier circuit 3, the capacitor 4, the power supply voltage zero cross detection circuit 7, the DC voltage detection circuit 8, and the control unit 11 constitute the DC power supply device 50 according to the first embodiment. Further, the DC power supply device 50, the inverter circuit 5, the DC current detection circuit 10, and the control unit 11 constitute the power conversion device 70 according to the first embodiment. In FIG. 1, the control unit 11 is configured as a common control unit that controls each of the rectifier circuit 3 and the inverter circuit 5, but two rectifier circuits 3 and the inverter circuit 5 are individually controlled. It may be configured to include a control unit. Further, the reactor 2 may be an external component when viewed from the DC power supply device 50. Further, although the capacitor 4 is a component of the DC power supply device 50 here, it may be a component of the inverter circuit 5.
図1は、実施の形態1に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図1において、実施の形態1に係るモータ駆動装置100は、リアクタ2と、整流回路3と、コンデンサ4と、インバータ回路5と、電源電圧ゼロクロス検出回路7と、直流電圧検出回路8と、直流電流検出回路9,10と、制御部11とを備える。これらの構成部のうち、リアクタ2、整流回路3、コンデンサ4、電源電圧ゼロクロス検出回路7、直流電圧検出回路8、及び制御部11は、実施の形態1に係る直流電源装置50を構成する。また、直流電源装置50、インバータ回路5、直流電流検出回路10、及び制御部11は、実施の形態1に係る電力変換装置70を構成する。なお、図1において、制御部11は、整流回路3及びインバータ回路5のそれぞれを制御する共通の制御部として構成されているが、整流回路3及びインバータ回路5のそれぞれを個別に制御する2つの制御部を備えて構成されていてもよい。また、リアクタ2は、直流電源装置50から見て外部の構成要素であってもよい。また、ここでは、コンデンサ4を直流電源装置50の構成要素としているが、インバータ回路5の構成要素であってもよい。
FIG. 1 is a diagram showing a configuration of a motor drive device including a power conversion device according to the first embodiment. In FIG. 1, the
整流回路3は、MOSFETであるスイッチング素子SW3-1~SW3-4を有し、リアクタ2を介して交流電源1に接続される。4つのスイッチング素子SW3-1~SW3-4はブリッジ接続され、スイッチング素子SW3-1~SW3-4のぞれぞれは、内部に寄生ダイオードD3-1~D3-4を有する。なお、スイッチング素子SW3-1~SW3-4のうち、スイッチング素子SW3-1,SW3-2を、「上アームのスイッチング素子」もしくは単に「上アーム」と呼ぶことがある。また、スイッチング素子SW3-3,SW3-4を、「下アームのスイッチング素子」もしくは単に「下アーム」と呼ぶことがある。また、スイッチング素子SW3-1,SW3-3の組、及びスイッチング素子SW3-2,SW3-4の組のそれぞれを「上下アーム」と呼ぶことがある。
The rectifier circuit 3 has switching elements SW3-1 to SW3-4 which are MOSFETs, and is connected to the AC power supply 1 via the reactor 2. The four switching elements SW3-1 to SW3-4 are bridge-connected, and each of the switching elements SW3-1 to SW3-4 has a parasitic diode D3-1 to D3-4 inside. Of the switching elements SW3-1 to SW3-4, the switching elements SW3-1 and SW3-2 may be referred to as "upper arm switching element" or simply "upper arm". Further, the switching elements SW3-3 and SW3-4 may be referred to as a "lower arm switching element" or simply a "lower arm". Further, each of the set of switching elements SW3-1 and SW3-3 and the set of switching elements SW3-2 and SW3-4 may be referred to as "upper and lower arms".
整流回路3は、更に、スイッチング素子SW3-1~SW3-4のそれぞれを駆動する駆動回路3a~3dを備える。整流回路3は、リアクタ2を介して供給される交流電源1の交流電力を直流電力に変換する。
The rectifier circuit 3 further includes drive circuits 3a to 3d for driving each of the switching elements SW3-1 to SW3-4. The rectifier circuit 3 converts the AC power of the AC power supply 1 supplied via the reactor 2 into DC power.
リアクタ2は、交流電源1と整流回路3との間に接続され、交流電源1から整流回路3へ供給される電源電流Isを平滑化する。コンデンサ4は、平滑コンデンサである。コンデンサ4は、正極端子及び負極端子を有する。コンデンサ4は、整流回路3の出力端子間に接続され、整流回路3の出力電圧を平滑化する。
The reactor 2 is connected between the AC power supply 1 and the rectifier circuit 3 and smoothes the power supply current Is supplied from the AC power supply 1 to the rectifier circuit 3. The capacitor 4 is a smoothing capacitor. The capacitor 4 has a positive electrode terminal and a negative electrode terminal. The capacitor 4 is connected between the output terminals of the rectifier circuit 3 and smoothes the output voltage of the rectifier circuit 3.
インバータ回路5は、ブリッジ接続される6つのスイッチング素子SW5-1~SW5-6を有し、整流回路3の出力端に接続される。整流回路3とインバータ回路5とは、一対の直流母線である正側直流母線16aと、負側直流母線16bとによって接続される。図1では、正側直流母線16aにおけるコンデンサ4の正極側の接続点をPとし、負側直流母線16bにおけるコンデンサ4の負極側の接続点をNとして示している。
The inverter circuit 5 has six switching elements SW5-1 to SW5-6 connected by a bridge, and is connected to the output end of the rectifier circuit 3. The rectifier circuit 3 and the inverter circuit 5 are connected by a pair of DC bus lines 16a on the positive side and a DC bus line 16b on the negative side. In FIG. 1, the connection point on the positive electrode side of the capacitor 4 on the positive DC bus 16a is shown as P, and the connection point on the negative electrode side of the capacitor 4 on the negative DC bus 16b is shown as N.
インバータ回路5は、整流回路3が出力する直流電力を三相交流電力に変換して、三相モータであるモータ6に供給する。即ち、モータ6は、インバータ回路5から出力される三相電力によって駆動される。インバータ回路5において、スイッチング素子SW5-1~SW5-6の一例は、図示のMOSFETであるが、絶縁ゲートバイポーラトランジスタ(Insulated Gate Bipolar Transistor:IGBT)を用いてもよい。スイッチング素子SW5-1~SW5-6のそれぞれは、逆並列に接続されるダイオードD5-1~D5-6を備えている。逆並列の接続とは、MOSFETのドレインとダイオードのカソードとが接続され、MOSFETのソースとダイオードのアノードとが接続されることを意味する。
The inverter circuit 5 converts the DC power output by the rectifier circuit 3 into three-phase AC power and supplies it to the motor 6 which is a three-phase motor. That is, the motor 6 is driven by the three-phase electric power output from the inverter circuit 5. In the inverter circuit 5, an example of the switching elements SW5-1 to SW5-6 is the MOSFET shown in the figure, but an insulated gate bipolar transistor (IGBT) may be used. Each of the switching elements SW5-1 to SW5-6 includes diodes D5-1 to D5-6 connected in antiparallel. The anti-parallel connection means that the drain of the MOSFET and the cathode of the diode are connected, and the source of the MOSFET and the anode of the diode are connected.
また、モータ駆動装置100において、電源電圧ゼロクロス検出回路7は、交流電源1の出力電圧である電源電圧Vsに基づいて、電源電圧Vsの極性を判別する。直流電圧検出回路8は、コンデンサ4の両端電圧を検出する。直流電流検出回路9は、整流回路3に流れる電流を検出する。直流電流検出回路10は、インバータ回路5に流れる電流を検出する。電源電圧ゼロクロス検出回路7、直流電圧検出回路8、直流電流検出回路9及び直流電流検出回路10の各検出信号は、制御部11に入力される。なお、以下の記載において、電源電圧ゼロクロス検出回路7を「第1の検出器」と呼び、直流電圧検出回路8を、「第2の検出器」と呼ぶ場合がある。また、整流回路3に流れる電流を「第1電流」と呼び、第1電流を検出する直流電流検出回路9を「第1の電流検出器」と呼ぶ場合がある。また、インバータ回路5に流れる電流を「第2電流」と呼び、第2電流を検出する直流電流検出回路10を「第2の電流検出器」と呼ぶ場合がある。
Further, in the motor drive device 100, the power supply voltage zero cross detection circuit 7 determines the polarity of the power supply voltage Vs based on the power supply voltage Vs which is the output voltage of the AC power supply 1. The DC voltage detection circuit 8 detects the voltage across the capacitor 4. The DC current detection circuit 9 detects the current flowing through the rectifier circuit 3. The DC current detection circuit 10 detects the current flowing through the inverter circuit 5. Each detection signal of the power supply voltage zero cross detection circuit 7, the DC voltage detection circuit 8, the DC current detection circuit 9, and the DC current detection circuit 10 is input to the control unit 11. In the following description, the power supply voltage zero cross detection circuit 7 may be referred to as a "first detector", and the DC voltage detection circuit 8 may be referred to as a "second detector". Further, the current flowing through the rectifier circuit 3 may be referred to as a "first current", and the DC current detection circuit 9 for detecting the first current may be referred to as a "first current detector". Further, the current flowing through the inverter circuit 5 may be referred to as a "second current", and the DC current detection circuit 10 for detecting the second current may be referred to as a "second current detector".
直流電流検出回路9は、シャント抵抗9aに流れる直流電流により生じるシャント抵抗9aにおける電圧降下を増幅器9bにより増幅して制御部11に与える。同様に、直流電流検出回路10は、シャント抵抗10aに流れる直流電流により生じるシャント抵抗10aにおける電圧降下を増幅器10bにより増幅して制御部11に与える。増幅器9b,10bは、オペアンプを用いて構成することができる。なお、実施の形態1の直流電流検出回路9,10では、直流電流をシャント抵抗9a又はシャント抵抗10aにおける電圧降下を増幅することで検出する構成としているが、直流電流を検出可能な変流器(Current Trasformer:CT)を用いた構成としてもよい。また、図1では、シャント抵抗9a又はシャント抵抗10aとして、1つの抵抗で記載しているが、2つもしくは3つ以上の抵抗を備えていてもよい。
The DC current detection circuit 9 amplifies the voltage drop in the shunt resistor 9a caused by the DC current flowing in the shunt resistor 9a by the amplifier 9b and gives it to the control unit 11. Similarly, the DC current detection circuit 10 amplifies the voltage drop in the shunt resistor 10a caused by the DC current flowing through the shunt resistor 10a by the amplifier 10b and gives it to the control unit 11. The amplifiers 9b and 10b can be configured by using an operational amplifier. The DC current detection circuits 9 and 10 of the first embodiment are configured to detect the DC current by amplifying the voltage drop in the shunt resistor 9a or the shunt resistor 10a, but the current transformer capable of detecting the DC current. (Curent Tracer: CT) may be used. Further, in FIG. 1, one resistor is shown as the shunt resistor 9a or the shunt resistor 10a, but two or three or more resistors may be provided.
制御部11は、アナログデジタル(Analog Digital:AD)変換部12~14を備える。直流電圧検出回路8の検出信号は、アナログ信号であり、AD変換部12によって検出値を示すデジタル信号に変換される。直流電流検出回路9,10の各検出信号もアナログ信号であり、それぞれは、AD変換部13,14によって検出値を示すデジタル信号に変換される。
The control unit 11 includes analog-digital (AD) conversion units 12 to 14. The detection signal of the DC voltage detection circuit 8 is an analog signal, and is converted into a digital signal indicating a detection value by the AD conversion unit 12. Each of the detection signals of the DC current detection circuits 9 and 10 is also an analog signal, and each of them is converted into a digital signal indicating a detection value by the AD conversion units 13 and 14.
制御部11は、電源電圧ゼロクロス検出回路7による極性判別信号zc、及び直流電流検出回路9の検出値に基づいて、整流回路3のスイッチング素子SW3-1~SW3-4におけるそれぞれのオン又はオフを制御する制御信号Sa~Sdを生成する。制御信号Sa~Sdは、それぞれが対応する駆動回路3a~3dに出力される。駆動回路3a~3dは、制御信号Sa~Sdのそれぞれに基づいて、対応するスイッチング素子SW3-1~SW3-4を駆動する。
The control unit 11 turns on or off the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 based on the polarity discrimination signal zc by the power supply voltage zero cross detection circuit 7 and the detection value of the DC current detection circuit 9. The control signals Sa to Sd to be controlled are generated. The control signals Sa to Sd are output to the corresponding drive circuits 3a to 3d. The drive circuits 3a to 3d drive the corresponding switching elements SW3-1 to SW3-4 based on the control signals Sa to Sd, respectively.
また、制御部11は、直流電圧検出回路8及び直流電流検出回路10の各検出値に基づいて、インバータ回路5のスイッチング素子SW5-1~SW5-6のオン又はオフを制御する制御信号UP,VP,WP,UN,VN,WNを生成する。制御信号UP,VP,WP,UN,VN,WNは、それぞれが対応する駆動回路5a~5fに出力される。駆動回路5a~5fは、制御信号UP,VP,WP,UN,VN,WNのそれぞれに基づいて、対応するスイッチング素子SW5-1~SW5-6を駆動する。
Further, the control unit 11 controls the on / off of the switching elements SW5-1 to SW5-6 of the inverter circuit 5 based on the detected values of the DC voltage detection circuit 8 and the DC current detection circuit 10. Generate VP, WP, UN, VN, WN. The control signals UP, VP, WP, UN, VN, and WN are output to the corresponding drive circuits 5a to 5f. The drive circuits 5a to 5f drive the corresponding switching elements SW5-1 to SW5-6 based on the control signals UP, VP, WP, UN, VN, and WN, respectively.
上記のように構成されたモータ駆動装置100において、インバータ回路5の出力端には、モータ6が接続されている。インバータ回路5は、モータ6に交流電力を供給することでモータ6を駆動する。モータ6の一例は、永久磁石電動機である。
In the motor drive device 100 configured as described above, the motor 6 is connected to the output end of the inverter circuit 5. The inverter circuit 5 drives the motor 6 by supplying AC power to the motor 6. An example of the motor 6 is a permanent magnet electric motor.
モータ6は、U相、V相及びW相巻線からなる三相巻線がY形に結線された三相Y形結線の固定子6aと、永久磁石回転子6bとを備える。U相、V相及びW相巻線の各一端同士は接続され、それぞれの巻線の他端は、それぞれU端子、V端子及びW端子を形成する。実施の形態1では、モータ6の固定子6aは、三相Y形結線の固定子を例示するが、三相Δ形結線の固定子で構成されていてもよい。なお、三相Y形結線のときと三相Δ形結線のときでは、直流電流から検出できる相電流情報が変わるのは言うまでもない。
The motor 6 includes a stator 6a with a three-phase Y-type connection in which three-phase windings including U-phase, V-phase, and W-phase windings are connected in a Y shape, and a permanent magnet rotor 6b. One ends of the U-phase, V-phase, and W-phase windings are connected to each other, and the other ends of the respective windings form a U terminal, a V terminal, and a W terminal, respectively. In the first embodiment, the stator 6a of the motor 6 exemplifies a stator of a three-phase Y-type connection, but may be composed of a stator of a three-phase delta-type connection. Needless to say, the phase current information that can be detected from the DC current changes between the three-phase Y-type connection and the three-phase Δ-type connection.
インバータ回路5のスイッチング素子SW5-1とスイッチング素子SW5-4との接続点は、モータ6の端子Uに接続され、スイッチング素子SW5-2とスイッチング素子SW5-5との接続点は、端子Vに接続され、スイッチング素子SW5-3とスイッチング素子SW5-6との接続点は、端子Wに接続される。実施の形態1では、U相巻線に流れる電流をU相電流と呼び、「Iu」で表す。以下、同様に、V相巻線及びW相巻線に流れる電流を、それぞれV相電流及びW相電流と呼び、それぞれを「Iv」、「Iw」で表す。
The connection point between the switching element SW5-1 and the switching element SW5-4 of the inverter circuit 5 is connected to the terminal U of the motor 6, and the connection point between the switching element SW5-2 and the switching element SW5-5 is connected to the terminal V. It is connected, and the connection point between the switching element SW5-3 and the switching element SW5-6 is connected to the terminal W. In the first embodiment, the current flowing through the U-phase winding is called a U-phase current and is represented by "Iu". Hereinafter, similarly, the currents flowing through the V-phase winding and the W-phase winding are referred to as V-phase current and W-phase current, respectively, and are represented by "Iv" and "Iw", respectively.
図2は、実施の形態1に係る直流電源装置50におけるMOSFETの動作説明に使用する第1の図である。また、図3は、実施の形態1に係る直流電源装置50におけるMOSFETの動作説明に使用する第2の図である。図2には、ゲートオフの場合の電流の流れが示されている。ゲートオフとは、MOSFETの内部に構成される寄生ダイオードに電流が流れる状態である。また、図3には、ゲートオンの場合の電流の流れが示されている。ゲートオンとは、MOSFETのゲートとソースとの間に電圧が印加されて、MOSFETにおけるFET本体側に電流が流れている状態である。ゲートオンの場合に、FET本体側に電流が流れるのは、寄生ダイオードよりもFET本体側の方が、インピーダンスが低いからである。
FIG. 2 is a first diagram used for explaining the operation of the MOSFET in the DC power supply device 50 according to the first embodiment. Further, FIG. 3 is a second diagram used for explaining the operation of the MOSFET in the DC power supply device 50 according to the first embodiment. FIG. 2 shows the current flow when the gate is off. Gate-off is a state in which a current flows through a parasitic diode formed inside the MOSFET. Further, FIG. 3 shows the current flow when the gate is on. Gate-on is a state in which a voltage is applied between the gate and the source of the MOSFET and a current is flowing to the FET main body side of the MOSFET. When the gate is turned on, the current flows on the FET main body side because the impedance on the FET main body side is lower than that on the parasitic diode.
上記の現象は、一般的に同期整流と呼ばれる。即ち、同期整流は、寄生ダイオードと同方向に電流が流れるタイミングでMOSFETをオン動作させてFET本体側に電流を流す制御手法である。
The above phenomenon is generally called synchronous rectification. That is, synchronous rectification is a control method in which the MOSFET is turned on at the timing when a current flows in the same direction as the parasitic diode to flow a current to the FET main body side.
特に、ワイドバンドギャップ(Wide Band Gap:WBG)半導体で形成されたMOSFETの場合、寄生ダイオードにおける順方向電圧降下(いわゆるVf)が3V程度発生する。このため、FET本体側よりも寄生ダイオード側の電圧特性が悪く、損失も大きくなる。また、WBG半導体で形成されたMOSFETの場合、FET本体のオン抵抗も低い。このため、同期整流を積極的に活用することで、更なる低損失化が図れる。なお、ここでは、WBG半導体で形成されたMOSFETについて説明したが、これに限定されない。同期整流が活用できる素子であればどのような素子を用いてもよい。
In particular, in the case of a MOSFET formed of a wide bandgap (WBG) semiconductor, a forward voltage drop (so-called Vf) in the parasitic diode occurs by about 3V. Therefore, the voltage characteristics on the parasitic diode side are worse than those on the FET main body side, and the loss is large. Further, in the case of a MOSFET formed of a WBG semiconductor, the on-resistance of the FET body is also low. Therefore, by positively utilizing synchronous rectification, further reduction in loss can be achieved. Although the MOSFET formed of the WBG semiconductor has been described here, the present invention is not limited to this. Any element may be used as long as it can utilize synchronous rectification.
図4は、図1の制御部11の機能を実現するハードウェア構成の一例を示すブロック図である。また、図5は、図1の制御部11の機能を実現するハードウェア構成の他の例を示すブロック図である。
FIG. 4 is a block diagram showing an example of a hardware configuration that realizes the function of the control unit 11 of FIG. Further, FIG. 5 is a block diagram showing another example of a hardware configuration that realizes the function of the control unit 11 of FIG.
下述する制御部11による制御機能を実現する場合には、図4に示すように、演算を行うプロセッサ300、プロセッサ300によって読みとられるプログラムが保存されるメモリ302、及び信号の入出力を行うインタフェース304を含む構成とすることができる。
When the control function by the control unit 11 described below is realized, as shown in FIG. 4, the processor 300 that performs the calculation, the memory 302 that stores the program read by the processor 300, and the input / output of the signal are performed. The configuration can include the interface 304.
プロセッサ300は、演算装置、マイクロプロセッサ、マイクロコンピュータ、CPU(Central Processing Unit)、又はDSP(Digital Signal Processor)といった演算手段であってもよい。また、メモリ302には、RAM(Random Access Memory)、ROM(Read Only Memory)、フラッシュメモリ、EPROM(Erasable Programmable ROM)、EEPROM(登録商標)(Electrically EPROM)といった不揮発性又は揮発性の半導体メモリを例示することができる。
The processor 300 may be a computing means such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 302 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EEPROM (Erasable Programmable ROM), or an EEPROM (registered trademark) (Electrically EPROM). It can be exemplified.
具体的に、メモリ302には、制御部11における制御機能を実行するプログラムが格納されている。プロセッサ300は、インタフェース304を介して必要な情報を授受し、メモリ302に格納されたプログラムをプロセッサ300が実行し、メモリ302に格納されたテーブルをプロセッサ300が参照することにより、下述するモータ制御を実行することができる。プロセッサ300による演算結果は、メモリ302に記憶することができる。
Specifically, the memory 302 stores a program that executes the control function of the control unit 11. The processor 300 sends and receives necessary information via the interface 304, the processor 300 executes a program stored in the memory 302, and the processor 300 refers to a table stored in the memory 302. Control can be performed. The calculation result by the processor 300 can be stored in the memory 302.
また、図4に示すプロセッサ300及びメモリ302は、図5のように処理回路305に置き換えてもよい。処理回路305は、単一回路、複合回路、ASIC(Application Specific Integrated Circuit)、FPGA(Field-Programmable Gate Array)、又は、これらを組み合わせたものが該当する。処理回路305に入力する情報、及び処理回路305から出力する情報は、インタフェース304を介して行うことができる。
Further, the processor 300 and the memory 302 shown in FIG. 4 may be replaced with the processing circuit 305 as shown in FIG. The processing circuit 305 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. The information input to the processing circuit 305 and the information output from the processing circuit 305 can be performed via the interface 304.
図6は、実施の形態1における整流回路3のスイッチング素子SW3-1~SW3-4を動作させる第1の動作波形を示す図である。図6(a)には、交流電源1の出力電圧である電源電圧Vsの波形が示され、図6(b)には、電源電圧ゼロクロス検出回路7から出力される極性判別信号zcが示されている。なお、極性判別信号zcの極性は、図1に示される電源電圧Vsの矢印の方向を正として扱う。この定義により、極性判別信号zcの極性は図6(b)に示されるものとなり、電源電圧Vsの極性が正の場合、極性判別信号zcは「H」(ハイ:High)となる。また、電源電圧Vsの極性が負の場合、極性判別信号zcは「L」(ロー:Low)となる。
FIG. 6 is a diagram showing a first operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment. FIG. 6A shows the waveform of the power supply voltage Vs which is the output voltage of the AC power supply 1, and FIG. 6B shows the polarity discrimination signal zc output from the power supply voltage zero cross detection circuit 7. ing. The polarity of the polarity discrimination signal zc is treated as positive in the direction of the arrow of the power supply voltage Vs shown in FIG. According to this definition, the polarity of the polarity discrimination signal zc is as shown in FIG. 6B, and when the polarity of the power supply voltage Vs is positive, the polarity discrimination signal zc becomes “H” (high: High). When the polarity of the power supply voltage Vs is negative, the polarity discrimination signal zc becomes “L” (low).
図6(c)には、直流電流Idc1の波形が示されている。直流電流Idc1の極性は、図1で示される矢印の向きを正とする。この定義の場合、図1に記載されたグラウンド(ground:GND)位置との関係で、矢印の向きの電流は、シャント抵抗9aにおいて負極性で検出されるが、制御部11の処理において、極性反転は自由に行うことができる。なお、図6(c)では、見やすさのため、直流電流Idc1の波形を正極性で示している。
FIG. 6C shows the waveform of the direct current Idc1. The polarity of the direct current Idc1 is positive in the direction of the arrow shown in FIG. In the case of this definition, the current in the direction of the arrow is detected negatively in the shunt resistor 9a in relation to the ground (GND) position shown in FIG. 1, but is polar in the processing of the control unit 11. The reversal can be done freely. In FIG. 6C, the waveform of the direct current Idc1 is shown as a positive electrode for ease of viewing.
上記では、直流電流Idc1の極性について説明したが、直流電流Idc1の極性のみで、過電流の原因を判別することができる。例えば、図1で示される矢印の向きに直流電流Idc1が流れるときを負極性とする。そして、正極性の過電流が検出された場合、その過電流は、上下アーム短絡時に発生した過電流であると判定できる。上下アーム短絡時では、正極性の直流電流Idc1が流れるからである。また、負極性の過電流が検出された場合、その過電流は、過負荷時に発生した過電流であると判定できる。過負荷時では、負極性の直流電流Idc1が流れるからである。このように、直流電流検出回路9における極性判別によって、上下アーム短絡による過電流であるのか、過負荷時の過電流であるのかを瞬時に判別することができる。これにより、高速での遮断が必要な上下アームの短絡保護を高速、且つ確実に実施することができる。また、極性判別だけで短絡保護を実施できるので、制御部11が、図4に示すようなソフトウェア処理で実現される場合でも、高速遮断を実施することができる。これにより、信頼性の高く、故障の少ない電力変換装置70を小型、且つ安価に提供することができる。
Although the polarity of the direct current Idc1 has been described above, the cause of the overcurrent can be determined only by the polarity of the direct current Idc1. For example, when the direct current Idc1 flows in the direction of the arrow shown in FIG. 1, the negative electrode property is defined. Then, when a positive overcurrent is detected, it can be determined that the overcurrent is an overcurrent generated when the upper and lower arms are short-circuited. This is because when the upper and lower arms are short-circuited, a positive DC current Idc1 flows. Further, when a negative electrode overcurrent is detected, it can be determined that the overcurrent is an overcurrent generated at the time of overload. This is because the negative electrode direct current Idc1 flows when the load is overloaded. In this way, by determining the polarity in the DC current detection circuit 9, it is possible to instantly determine whether the overcurrent is due to a short circuit between the upper and lower arms or an overcurrent at the time of overload. As a result, short-circuit protection of the upper and lower arms, which requires high-speed interruption, can be performed at high speed and reliably. Further, since short-circuit protection can be performed only by determining the polarity, high-speed shutoff can be performed even when the control unit 11 is realized by software processing as shown in FIG. As a result, the power conversion device 70 having high reliability and few failures can be provided in a small size and at low cost.
また、図1の構成において、直流電流検出回路9は、共通GND端子と、MOSFETであるスイッチング素子SW3-3,SW3-4のソース端子側の接続点との間に挿入されている。これにより、図1に示す電力変換装置70は、共通GNDによる非絶縁型の電力変換装置として構成することができる。これにより、絶縁アンプ又はフォトカプラといった絶縁回路を使用せずに装置を構成できるので、信号の授受を高速化することができる。また、高応答性の電力変換装置を安価に提供することができる。
Further, in the configuration of FIG. 1, the DC current detection circuit 9 is inserted between the common GND terminal and the connection point on the source terminal side of the switching elements SW3 and SW3-4 which are MOSFETs. As a result, the power conversion device 70 shown in FIG. 1 can be configured as a non-isolated power conversion device using a common GND. As a result, the device can be configured without using an insulating circuit such as an insulating amplifier or a photocoupler, so that signal transmission / reception can be speeded up. Further, a highly responsive power conversion device can be provided at low cost.
次に、実施の形態1における整流回路3の要部の動作について説明する。前述の通り、スイッチング素子SW3-1~SW3-4は、電源電圧ゼロクロス検出回路7による極性判別信号zc、及び直流電流検出回路9の検出値に基づいて制御される。
Next, the operation of the main part of the rectifier circuit 3 in the first embodiment will be described. As described above, the switching elements SW3-1 to SW3-4 are controlled based on the polarity discrimination signal zc by the power supply voltage zero cross detection circuit 7 and the detection value of the DC current detection circuit 9.
制御部11は、直流電流Idc1の検出値が予め定められた閾値以上であった場合、下述のスイッチング素子をオンさせ、閾値以下になった場合に下述のスイッチング素子をオフさせる。この閾値は、図6(c)において、「Idc_on」で示している。なお、図6では、オンの閾値とオフの閾値を同値で記載しているが、これに限定されない。オンの閾値とオフの閾値との間でヒステリシスを持たせ、オンの閾値とオフの閾値とを異ならせてもよい。また、ここでは、閾値以上、閾値以下と説明したが、閾値超過、閾値未満であってもよいことは言うまでもない。
The control unit 11 turns on the switching element described below when the detected value of the direct current Idc1 is equal to or higher than a predetermined threshold value, and turns off the switching element described below when the detected value is equal to or lower than the threshold value. This threshold is indicated by "Idc_on" in FIG. 6 (c). In FIG. 6, the on threshold value and the off threshold value are shown with the same value, but the present invention is not limited to this. Hysteresis may be provided between the on threshold and the off threshold so that the on threshold and the off threshold are different. Further, although it has been described here that it is above the threshold value and below the threshold value, it goes without saying that the threshold value may be exceeded or below the threshold value.
直流電流Idc1の検出値が予め定められた閾値以上になった場合にオンさせるスイッチング素子は、SW3-1~3-4のうちの2つである。この2つは、電源電圧ゼロクロス検出回路7で検出された交流電源1の極性、即ち極性判別信号zcに基づいて決定される。電源電圧ゼロクロス検出回路7が「H」を検出した場合、図6(d)、(g)に示すようにスイッチング素子SW3-1,SW3-4がオンに制御される。また、電源電圧ゼロクロス検出回路7が「L」を検出した場合、図6(e)、(f)に示すようにスイッチング素子SW3-2,SW3-3がオンに制御される。そして、直流電流Idc1の検出値が閾値以上から閾値未満となった場合、スイッチング素子SW3-1~SW3-4のうちでオンしているスイッチング素子は、オフに制御される。図6の(d)~(g)には、電源電圧Vsの1周期の間の様子が示されている。
Two of SW3-1 to 3-4 are switching elements that are turned on when the detected value of the DC current Idc1 exceeds a predetermined threshold value. These two are determined based on the polarity of the AC power supply 1 detected by the power supply voltage zero cross detection circuit 7, that is, the polarity discrimination signal zc. When the power supply voltage zero cross detection circuit 7 detects “H”, the switching elements SW3-1 and SW3-4 are controlled to be ON as shown in FIGS. 6 (d) and 6 (g). When the power supply voltage zero cross detection circuit 7 detects “L”, the switching elements SW3 and SW3-3 are controlled to be ON as shown in FIGS. 6 (e) and 6 (f). Then, when the detected value of the direct current Idc1 changes from the threshold value or more to the threshold value or less, the on switching element among the switching elements SW3-1 to SW3-4 is controlled to be turned off. 6 (d) to 6 (g) show the state of the power supply voltage Vs during one cycle.
前述したように、整流回路3はMOSFETで構成されているため、図1に示す通り、MOSFETには寄生ダイオードが付与されている。従って、MOSFETのゲートにオン信号を供給しなくとも、寄生ダイオードがオンして、コンデンサインプットと呼ばれる電流が交流電源1から流れる。この電流が流れるタイミングでMOSFETをオンすることで、図2で示した通り、オン抵抗の低いFET本体側に電流が流れる同期整流によって、整流回路3の損失を低減できる。
As described above, since the rectifier circuit 3 is composed of MOSFETs, as shown in FIG. 1, a parasitic diode is added to the MOSFETs. Therefore, even if the ON signal is not supplied to the gate of the MOSFET, the parasitic diode is turned on and a current called a capacitor input flows from the AC power supply 1. By turning on the MOSFET at the timing when this current flows, as shown in FIG. 2, the loss of the rectifier circuit 3 can be reduced by the synchronous rectification in which the current flows to the FET main body side having a low on-resistance.
次に、図6とは異なる動作について説明する。図7は、実施の形態1における整流回路3のスイッチング素子SW3-1~SW3-4を動作させる第2の動作波形を示す図である。なお、図7における信号波形の並びは、図6と同じである。
Next, an operation different from that shown in FIG. 6 will be described. FIG. 7 is a diagram showing a second operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment. The arrangement of the signal waveforms in FIG. 7 is the same as that in FIG.
図7の動作波形において、図6との差異点は、スイッチング素子SW3-1,SW3-3は、制御部11によってパルス幅変調(Pulse Width Modulation:PWM)制御される点である。PWM制御することで、交流電源1から整流回路3に流れる電流を正弦波に波形生成することが可能となる。加えて、コンデンサ4の両端電圧である接続点PN間の電圧を電源電圧Vsの実効値の√2倍以上に上昇させる昇圧制御も可能となる。なお、入力電流を正弦波に波形生成できるためできる、電源力率をほぼ1に制御でき、電源高調波電流をほぼ0に制御できることは言うまでもない。
In the operation waveform of FIG. 7, the difference from FIG. 6 is that the switching elements SW3-1 and SW3-3 are controlled by the control unit 11 for pulse width modulation (PWM). By PWM control, it is possible to generate a sine wave waveform of the current flowing from the AC power supply 1 to the rectifier circuit 3. In addition, boost control that raises the voltage between the connection point PN, which is the voltage across the capacitor 4, to √2 times or more the effective value of the power supply voltage Vs is also possible. Needless to say, since the input current can generate a sine wave waveform, the power supply factor can be controlled to be substantially 1, and the power supply harmonic current can be controlled to be approximately 0.
上述のように、スイッチング素子SW3-1,SW3-3をPWM制御することで、コンデンサ4の両端電圧を昇圧することができるので、昇圧電圧を高くすることで、消費電力の大きい動作範囲を満たすことができる。インバータ回路5及びモータ6は電流容量が決まっており、モータ6の電流容量に応じて、インバータ回路5におけるスイッチング素子SW5-1~SW5-6の電流容量が選定されている。例えば、電流容量の大きなスイッチング素子をインバータ回路5に用いると、スイッチング素子SW5-1~SW5-6がオンした場合に発生する導通損失及びスイッチング損失は、電流容量の小さいスイッチング素子よりも大きくなる傾向がある。仮に損失の小さい電流容量の大きいスイッチング素子を選定すると、インバータ回路5は非常に高価な回路となる。従って、消費電力の範囲が広い製品を使用者に適切な販売価格で提供するためには、電流容量の小さいスイッチング素子を選定することが要請される。
As described above, the voltage across the capacitor 4 can be boosted by PWM-controlling the switching elements SW3-1 and SW3-3. Therefore, by increasing the boosted voltage, an operating range with large power consumption is satisfied. be able to. The current capacities of the inverter circuit 5 and the motor 6 are fixed, and the current capacities of the switching elements SW5-1 to SW5-6 in the inverter circuit 5 are selected according to the current capacity of the motor 6. For example, when a switching element having a large current capacity is used in the inverter circuit 5, the conduction loss and the switching loss generated when the switching elements SW5-1 to SW5-6 are turned on tend to be larger than those of the switching element having a small current capacity. There is. If a switching element having a small loss and a large current capacity is selected, the inverter circuit 5 becomes a very expensive circuit. Therefore, in order to provide a product with a wide range of power consumption to a user at an appropriate selling price, it is required to select a switching element having a small current capacity.
上記の要請に対し、実施の形態1の技術を適用する。具体的に、消費電力の大きい動作範囲では、整流回路3を構成するスイッチング素子SW3-1及びSW3-3をPWM制御する。PWM制御により整流回路3の出力電圧は昇圧される。整流回路3の出力電圧は昇圧されるので、インバータ回路5及びモータ6の電流容量を大きくすることなく、電圧上昇分だけ消費電力を増大することが可能となる。これにより、高効率な製品に適用できる電力変換装置を得ることができる。
In response to the above request, the technique of the first embodiment is applied. Specifically, in the operating range where the power consumption is large, the switching elements SW3-1 and SW3-3 constituting the rectifier circuit 3 are PWM-controlled. The output voltage of the rectifier circuit 3 is boosted by PWM control. Since the output voltage of the rectifier circuit 3 is boosted, it is possible to increase the power consumption by the amount of the voltage increase without increasing the current capacity of the inverter circuit 5 and the motor 6. As a result, it is possible to obtain a power conversion device that can be applied to a highly efficient product.
更に言うと、実施の形態1の技術を適用すれば、電流容量が小さいスイッチング素子を用いることができる。これにより、消費電力の小さい動作範囲において、インバータ回路5の損失を小さくすることができる。そして、消費電力の範囲が狭いという条件から、消費電力の範囲が広いという条件までの何れの条件下においても、損失の少ない電力変換装置を構成することが可能となる。更に、整流回路3の出力電圧の昇圧により、最大負荷量を上昇させることも可能となる。従って、実施の形態1に係る電力変換装置によれば、低負荷時の効率改善による省エネルギー特性と、最大負荷を引き上げたハイパワー特性との両立が可能となる。
Furthermore, if the technique of the first embodiment is applied, a switching element having a small current capacity can be used. As a result, the loss of the inverter circuit 5 can be reduced in the operating range where the power consumption is small. Then, it is possible to configure the power conversion device with less loss under any condition from the condition that the range of power consumption is narrow to the condition that the range of power consumption is wide. Further, it is possible to increase the maximum load amount by boosting the output voltage of the rectifier circuit 3. Therefore, according to the power conversion device according to the first embodiment, it is possible to achieve both the energy saving characteristic by improving the efficiency at the time of low load and the high power characteristic with the maximum load raised.
また、図8は、実施の形態1における整流回路3のスイッチング素子SW3-1~SW3-4を動作させる第3の動作波形を示す図である。図8は、図6及び図7とは異なる動作波形を示すものである。図8では、図6及び図7に示す信号波形のうち、(c)を除く、(a),(b),(d)~(g)の信号波形を1.5周期分示している。また、(b)と(d)との間に、電源電流Isの波形を示している。
Further, FIG. 8 is a diagram showing a third operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment. FIG. 8 shows an operation waveform different from that of FIGS. 6 and 7. In FIG. 8, among the signal waveforms shown in FIGS. 6 and 7, the signal waveforms of (a), (b), (d) to (g) excluding (c) are shown for 1.5 cycles. Further, a waveform of the power supply current Is is shown between (b) and (d).
図8では、電源電圧Vsの半周期に1パルスだけ、図6で示した交流電源1の極性でオンさせないスイッチング素子をオンさせている。具体的に説明すると、電源電圧Vsの最初の半周期では、スイッチング素子SW3-1をオンにするために制御信号Saが「H」になっているが、その直前にスイッチング素子SW3-3をオンにするために制御信号Scが「H」になっている。また、電源電圧Vsの次の半周期では、スイッチング素子SW3-3をオンにするために制御信号Scが「H」になっているが、その直前にスイッチング素子SW3-1をオンにするために制御信号Saが「H」になっている。これらの動作は、交流電源1の出力を短絡させる、いわゆる「電源短絡」と呼ばれる動作である。これらの動作によって、力率を改善することができる。ここで、このような力率改善動作は、上述した実施の形態1の効果を阻害するものではなく、併用しても何ら問題は無い。逆に、図6と図7の動作の中間に、図8のような力率改善動作を挟んでもよい。このようにすることで、上述した効果に加え、力率改善効果をも得ることができる。
In FIG. 8, only one pulse is turned on in a half cycle of the power supply voltage Vs, and the switching element which is not turned on by the polarity of the AC power supply 1 shown in FIG. 6 is turned on. Specifically, in the first half cycle of the power supply voltage Vs, the control signal Sa is set to "H" in order to turn on the switching element SW3-1, but the switching element SW3-3 is turned on immediately before that. The control signal Sc is set to "H". Further, in the next half cycle of the power supply voltage Vs, the control signal Sc is set to "H" in order to turn on the switching element SW3-3, but to turn on the switching element SW3-1 immediately before that. The control signal Sa is "H". These operations are so-called "power short circuits" in which the output of the AC power supply 1 is short-circuited. These movements can improve the power factor. Here, such a power factor improving operation does not hinder the effect of the above-described first embodiment, and there is no problem even if it is used in combination. On the contrary, the power factor improving operation as shown in FIG. 8 may be inserted between the operations of FIGS. 6 and 7. By doing so, in addition to the above-mentioned effect, a power factor improving effect can also be obtained.
なお、図8では、電源半周期に1パルスだけ電源短絡動作を行わせているが、これに限定されない。電源半周期に複数パルスの電源短絡動作を行っても、本発明の目的を逸脱せずに実施することが可能である。
Note that, in FIG. 8, the power supply short-circuit operation is performed for only one pulse in the power supply half cycle, but the present invention is not limited to this. Even if a power short-circuit operation of a plurality of pulses is performed in a power supply half cycle, it can be carried out without deviating from the object of the present invention.
また、図8の閾値Idc_onは、図6のものよりも高く設定されている。図8では、電源電圧Vsの正極性時において、極性判別信号zcから「H」が出力されると、予め設定された時間の経過後に、まずは、スイッチング素子SW3-3がオン動作して電源短絡状態が形成されている。そして、閾値Idc_onによって、正極性での動作であるスイッチング素子SW3-1,SW3-4がオン動作している。この動作において、スイッチング素子SW3-1,SW3-4のオフを閾値Idc_onで実施させると、同期整流状態が早く終了してしまい、寄生ダイオードに電流が流れる時間が長くなる。そこで、図8では、同期整流を終了させるタイミング、即ちスイッチング素子SW3-1,SW3-4をオフにするタイミングは、閾値Idc_offで実施させている。これにより、スイッチング素子SW3-1,SW3-4のオフを閾値Idc_onで実施させるときと比べて、FET本体に電流が流れる同期整流の状態を長く維持できる。
Further, the threshold value Idc_on in FIG. 8 is set higher than that in FIG. In FIG. 8, when “H” is output from the polarity discrimination signal zc when the power supply voltage Vs is positive, the switching element SW3-3 is first turned on and the power supply is short-circuited after a preset time has elapsed. The state is formed. Then, the switching elements SW3-1 and SW3-4, which are positively operated, are turned on by the threshold value Idc_on. In this operation, if the switching elements SW3-1 and SW3-4 are turned off at the threshold value Idc_on, the synchronous rectification state ends early, and the time for the current to flow through the parasitic diode becomes long. Therefore, in FIG. 8, the timing for ending the synchronous rectification, that is, the timing for turning off the switching elements SW3-1 and SW3-4 is set by the threshold value Idc_off. As a result, the state of synchronous rectification in which a current flows through the FET main body can be maintained for a longer time than when the switching elements SW3-1 and SW3-4 are turned off at the threshold value Idc_on.
また、図9は、実施の形態1における整流回路3のスイッチング素子SW3-1~SW3-4を動作させる第4の動作波形を示す図である。図9は、図6~図8の各図面とは異なる動作波形を示すものである。なお、図9における信号波形の並びは、図6と同じである。また、図9における閾値Idc_onの値は、図6のものと同じである。
Further, FIG. 9 is a diagram showing a fourth operation waveform for operating the switching elements SW3-1 to SW3-4 of the rectifier circuit 3 in the first embodiment. FIG. 9 shows an operation waveform different from each of the drawings of FIGS. 6 to 8. The arrangement of the signal waveforms in FIG. 9 is the same as that in FIG. Further, the value of the threshold value Idc_on in FIG. 9 is the same as that in FIG.
図9では、スイッチング素子SW3-1,SW3-4がオンに制御された後、スイッチング素子SW3-3をオンに制御する場合のみ、スイッチング素子SW3-1がオフに制御されている。この場合、制御部11は、閾値Idc_onとは別の閾値を設け、スイッチング素子SW3-1からスイッチング素子SW3-3へオン動作が切り替わるように、スイッチング素子SW3-1,SW3-3を制御することになる。この動きは、図9中には示していないが、如何なる手法で実現してもよく、実現手法は問わない。
In FIG. 9, after the switching elements SW3-1 and SW3-4 are controlled to be ON, the switching element SW3-1 is controlled to be OFF only when the switching element SW3-3 is controlled to be ON. In this case, the control unit 11 sets a threshold value different from the threshold value Idc_on, and controls the switching elements SW3-1 and SW3-3 so that the ON operation is switched from the switching element SW3-1 to the switching element SW3-3. become. Although this movement is not shown in FIG. 9, it may be realized by any method, and the realization method does not matter.
実施の形態1の制御において、図6及び図7の制御は、制御部11によって、選択的に実施される。具体的に、制御部11は、交流電源1から供給される電力が予め設定された設定値より低い場合には図6に示す制御を行い、予め設定された設定値より高い場合には図7に示す制御を行う。交流電源1から供給される電力が予め設定された設定値より低い場合とは、モータ6に接続される図示しない負荷の消費電力が低い場合を意味する。このような状態を「低消費電力時」と呼ぶ。また、交流電源1から供給される電力が予め設定された設定値より高い場合とは、負荷の消費電力が高い場合を意味する。このような状態を「高消費電力時」と呼ぶ。また、低消費電力時と、高消費電力時との間の中間の状態を「中消費電力時」と呼ぶ。なお、負荷の消費電力が高い場合には、整流回路3からインバータ回路5に供給される電力も大きくなる。このため、整流回路3からインバータ回路5に供給される電力の高低に応じたそれぞれの状態を、「低消費電力時」、「中消費電力時」及び「高消費電力時」と称してもよい。
In the control of the first embodiment, the control of FIGS. 6 and 7 is selectively carried out by the control unit 11. Specifically, the control unit 11 performs the control shown in FIG. 6 when the power supplied from the AC power supply 1 is lower than the preset set value, and when it is higher than the preset set value, FIG. 7 The control shown in is performed. The case where the power supplied from the AC power supply 1 is lower than the preset set value means that the power consumption of a load (not shown) connected to the motor 6 is low. Such a state is called "low power consumption". Further, the case where the power supplied from the AC power source 1 is higher than the preset set value means the case where the power consumption of the load is high. Such a state is called "high power consumption". Further, an intermediate state between the time of low power consumption and the time of high power consumption is called "medium power consumption". When the power consumption of the load is high, the power supplied from the rectifier circuit 3 to the inverter circuit 5 also becomes large. Therefore, each state according to the level of the power supplied from the rectifier circuit 3 to the inverter circuit 5 may be referred to as "low power consumption", "medium power consumption", and "high power consumption". ..
図10は、実施の形態1の制御部11における動作モードの説明に使用する図である。図10に示すように、制御部11は、第1の動作モード、第2の動作モード及び第3の動作モードという3つの動作モードを少なくとも有している。第1の動作モードは、整流回路3の各スイッチング素子に対して、図6に示す同期整流を実施させる動作モードである。第2の動作モードは、整流回路3の各スイッチング素子に対して、図7に示すPWM制御を実施させる動作モードである。第3の動作モードは、整流回路3の各スイッチング素子に対して、図8に示す制御を実施させる動作モードである。制御部11は、負荷の消費電力に応じて、これらの3つの動作モードのうちの何れか1つを起動して、整流回路3の各スイッチング素子を制御する。
FIG. 10 is a diagram used for explaining the operation mode in the control unit 11 of the first embodiment. As shown in FIG. 10, the control unit 11 has at least three operation modes, that is, a first operation mode, a second operation mode, and a third operation mode. The first operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the synchronous rectification shown in FIG. The second operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the PWM control shown in FIG. 7. The third operation mode is an operation mode in which each switching element of the rectifier circuit 3 is subjected to the control shown in FIG. The control unit 11 activates any one of these three operation modes according to the power consumption of the load to control each switching element of the rectifier circuit 3.
図10の横軸は、負荷の消費電力Psを示しており、右方向に行くに連れて消費電力が大きくなっている。また、図10において、「Ps_PS」は、第1の動作モードから第3の動作モードに移行させるときの閾値である。以下同様に、「Ps_SP」は、第3の動作モードから第1の動作モードに移行させるときの閾値であり、「Ps_SF」は、第3の動作モードから第2の動作モードに移行させるときの閾値であり、「Ps_FS」は、第2の動作モードから第3の動作モードに移行させるときの閾値である。また、第1の動作モードは、低消費電力時に実施される動作モードであり、第2の動作モードは、高低消費電力時に実施される動作モードであり、第3の動作モードは、中消費電力時に実施される動作モードである。「Ps_PS」及び「Ps_SP」という2つの閾値は、制御を安定化させるためにヒステリシス特性を持たせたものである。「Ps_SF」及び「Ps_FS」という2つの閾値についても同様である。
The horizontal axis of FIG. 10 shows the power consumption Ps of the load, and the power consumption increases as it goes to the right. Further, in FIG. 10, “Ps_PS” is a threshold value at the time of shifting from the first operation mode to the third operation mode. Similarly, "Ps_SP" is a threshold value when shifting from the third operation mode to the first operation mode, and "Ps_SF" is a threshold value when shifting from the third operation mode to the second operation mode. It is a threshold value, and "Ps_FS" is a threshold value when shifting from the second operation mode to the third operation mode. Further, the first operation mode is an operation mode carried out at the time of low power consumption, the second operation mode is an operation mode carried out at the time of high and low power consumption, and the third operation mode is a medium power consumption. This is the operation mode that is sometimes implemented. The two threshold values "Ps_PS" and "Ps_SP" are provided with a hysteresis characteristic in order to stabilize the control. The same applies to the two thresholds "Ps_SF" and "Ps_FS".
なお、図10の例では、第3の動作モードにおいて、図8に示す制御を実施すると記載しているが、図8に代えて図9に示す制御を実施してもよい。
In the example of FIG. 10, it is described that the control shown in FIG. 8 is performed in the third operation mode, but the control shown in FIG. 9 may be performed instead of FIG.
また、上記では、負荷の消費電力Psが、低消費電力であるか、中消費電力であるか、高消費電力であるかの判定に、交流電源1から供給される電力を用いると説明しているが、この例に限定されない。例えば、直流電流検出回路9及び直流電圧検出回路8で検出した検出値から演算で求めた電力を用いてもよい。また、交流電源1から供給電力される電力を検出する図示しない電力検出回路での検出値に基づいて、判定してもよい。
Further, in the above description, it is described that the power supplied from the AC power source 1 is used for determining whether the power consumption Ps of the load is low power consumption, medium power consumption, or high power consumption. However, it is not limited to this example. For example, the electric power obtained by calculation from the detected values detected by the DC current detection circuit 9 and the DC voltage detection circuit 8 may be used. Further, the determination may be made based on a detection value in a power detection circuit (not shown) that detects the power supplied from the AC power supply 1.
また、負荷の消費電力Psに代えて、モータ6の回転数、又はモータ6に流れるモータ電流に基づいて切り替えてもよい。また、本発明の目的を逸脱しない範囲であれば、どのような検出値を用いてもよい。
Further, instead of the power consumption Ps of the load, switching may be performed based on the rotation speed of the motor 6 or the motor current flowing through the motor 6. Moreover, any detected value may be used as long as it does not deviate from the object of the present invention.
以上説明したように、実施の形態1に係る直流電源装置によれば、寄生ダイオードと同方向に電流が流れるタイミングでMOSFETをオン動作させる同期整流を行う第1の動作モードと、MOSFETをパルス幅変調制御してコンデンサの両端電圧を昇圧する第2の動作モードと、を有する。そして、直流電源装置に具備される制御部は、低消費電力時には、第1の動作モードを起動し、高消費電力時には、第2の動作モードを起動する。これにより、消費電力の範囲が広い製品に対して適用でき、消費電力に応じた効率の改善が可能になる。
As described above, according to the DC power supply device according to the first embodiment, the first operation mode in which the MOSFET is turned on at the timing when the current flows in the same direction as the parasitic diode and the MOSFET are pulse-width. It has a second operation mode in which modulation control is performed to boost the voltage across the capacitor. Then, the control unit provided in the DC power supply device activates the first operation mode when the power consumption is low, and activates the second operation mode when the power consumption is high. As a result, it can be applied to products having a wide range of power consumption, and efficiency can be improved according to the power consumption.
また、実施の形態1に係る直流電源装置によれば、低消費電力時においては、スイッチングの回数を極力少なくして、MOSFETの同期整流による低損失効果を得ることができる。また、高消費電力時においては、MOSFETをPWM制御することで直流電圧を昇圧し、インバータ回路及びモータに流れる電流を小さくすることができる。これにより、安価で損失の少ないMOSFETを適用することができる。これより、低損失な電力変換装置を提供することができる。
Further, according to the DC power supply device according to the first embodiment, when the power consumption is low, the number of switchings can be reduced as much as possible, and a low loss effect can be obtained by synchronous rectification of the MOSFET. Further, when the power consumption is high, the DC voltage can be boosted by PWM control of the MOSFET, and the current flowing through the inverter circuit and the motor can be reduced. This makes it possible to apply an inexpensive MOSFET with low loss. As a result, it is possible to provide a low-loss power conversion device.
実施の形態2.
図11は、実施の形態2の整流回路3における駆動回路3a~3dの回路構成を示す回路図である。図11において、駆動回路3aは、MOSFETのソース端子がリアクタ2を介して交流電源1と電気的に接続しているスイッチング素子SW3-1を駆動する駆動回路である。また、駆動回路3bは、MOSFETのソース端子がリアクタ2を介して交流電源1と電気的に接続しているスイッチング素子SW3-2を駆動する駆動回路である。なお、図1に示す実施の形態1と同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。Embodiment 2.
FIG. 11 is a circuit diagram showing a circuit configuration ofdrive circuits 3a to 3d in the rectifier circuit 3 of the second embodiment. In FIG. 11, the drive circuit 3a is a drive circuit that drives the switching element SW3-1 in which the source terminal of the MOSFET is electrically connected to the AC power supply 1 via the reactor 2. Further, the drive circuit 3b is a drive circuit that drives the switching element SW3-2 in which the source terminal of the MOSFET is electrically connected to the AC power supply 1 via the reactor 2. The same or equivalent components as those in the first embodiment shown in FIG. 1 are designated by the same reference numerals, and redundant description will be omitted.
図11は、実施の形態2の整流回路3における駆動回路3a~3dの回路構成を示す回路図である。図11において、駆動回路3aは、MOSFETのソース端子がリアクタ2を介して交流電源1と電気的に接続しているスイッチング素子SW3-1を駆動する駆動回路である。また、駆動回路3bは、MOSFETのソース端子がリアクタ2を介して交流電源1と電気的に接続しているスイッチング素子SW3-2を駆動する駆動回路である。なお、図1に示す実施の形態1と同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。
FIG. 11 is a circuit diagram showing a circuit configuration of
スイッチング素子SW3-3,SW3-4は、MOSFETのソース端子が直流電流検出回路9を介して、共通GNDに電気的に接続されている。このため、駆動回路3c,3dは共通GNDに接続されている制御電源22から直接、MOSFETの駆動電力の供給を受けることができる。これに対して、スイッチング素子SW3-1,SW3-2は、MOSFETのソース端子が交流電源1と電気的に接続されているため、制御電源22から直接、駆動電力の供給を受けることができない。
In the switching elements SW3 and SW3-4, the source terminal of the MOSFET is electrically connected to the common GND via the DC current detection circuit 9. Therefore, the drive circuits 3c and 3d can receive the drive power of the MOSFET directly from the control power supply 22 connected to the common GND. On the other hand, in the switching elements SW3-1 and SW3-2, since the source terminal of the MOSFET is electrically connected to the AC power supply 1, the driving power cannot be directly supplied from the control power supply 22.
そこで、駆動回路3aは、ダイオード20a及び抵抗21aの直列回路を介して制御電源22に接続される。これにより、SW3-3がオンしている場合に、ダイオード20aがオンして制御電源22と駆動回路3aは電気的に接続され、MOSFETをゲート駆動させる駆動電力の供給を受けることができる。このときに供給された電力は、コンデンサ23aにおいてチャージされる。このため、ダイオード20aがオフして駆動電力の供給が無い場合でもスイッチング素子SW3-1に駆動電力の供給が可能になる。駆動回路3bも、ダイオード20b及び抵抗21bの直列回路を介して制御電源22に接続される。これらは、一般的にブートストラップと呼ばれる技術であり、公知である。このため、これ以上の動作説明は割愛する。
Therefore, the drive circuit 3a is connected to the control power supply 22 via a series circuit of the diode 20a and the resistor 21a. As a result, when the SW3-3 is turned on, the diode 20a is turned on, the control power supply 22 and the drive circuit 3a are electrically connected, and the drive power for driving the MOSFET at the gate can be supplied. The electric power supplied at this time is charged in the capacitor 23a. Therefore, even when the diode 20a is turned off and the driving power is not supplied, the driving power can be supplied to the switching element SW3-1. The drive circuit 3b is also connected to the control power supply 22 via a series circuit of the diode 20b and the resistor 21b. These are techniques commonly referred to as bootstraps and are known. Therefore, further explanation of the operation is omitted.
このブートストラップは、ダイオード20a,20b、及びコンデンサ23a,23bが重要なキー部品となる。なお、以下の説明では、ダイオード20a,20bを「ブートダイオード20a,20b」と呼び、コンデンサ23a,23bを「ブートコンデンサ23a,23b」と呼ぶ。また、以下では、ブートダイオード20a及びブートコンデンサ23aを中心に動作説明を行う。
Diodes 20a and 20b and capacitors 23a and 23b are important key components for this bootstrap. In the following description, the diodes 20a and 20b will be referred to as " boot diodes 20a and 20b", and the capacitors 23a and 23b will be referred to as " boot capacitors 23a and 23b". Further, in the following, the operation will be described focusing on the boot diode 20a and the boot capacitor 23a.
ブートコンデンサ23aの負極端子は、MOSFETであるスイッチング素子SW3-1のソース端子に接続され、ブートコンデンサ23aの正極端子は、ダイオード20a及び抵抗21aの直列回路を介して制御電源22に接続される。
The negative electrode terminal of the boot capacitor 23a is connected to the source terminal of the switching element SW3-1 which is a MOSFET, and the positive electrode terminal of the boot capacitor 23a is connected to the control power supply 22 via a series circuit of the diode 20a and the resistor 21a.
ブートコンデンサ23aの負極性側に接続されているスイッチング素子SW3-3がオンすると、ブートコンデンサ23aに電荷がチャージされ、駆動電力が確保される。実施の形態1の整流回路3であれば、スイッチング素子SW3-3のMOSFETが特にオンしなくとも、コンデンサ4へ交流電源1からの充電電流が流れれば、寄生ダイオードD3-3がオンするので、ブートコンデンサ23aのチャージが実現される。ブートコンデンサ23bのチャージも同様な動作で実現される。
When the switching element SW3-3 connected to the negative electrode side of the boot capacitor 23a is turned on, the boot capacitor 23a is charged with electric charge and the driving power is secured. In the rectifier circuit 3 of the first embodiment, even if the MOSFET of the switching element SW3-3 is not turned on, the parasitic diode D3-3 is turned on when the charging current from the AC power supply 1 flows through the capacitor 4. , The charge of the boot capacitor 23a is realized. Charging of the boot capacitor 23b is also realized by the same operation.
その一方で、例えばワイドバンドギャップ半導体で形成されたMOSFETの場合、寄生ダイオードの電圧特性が悪く、寄生ダイオードでの電圧効果が大きい。このため、ブートコンデンサのチャージ電圧は、寄生ダイオードのVf分だけ高い電位が必要となる。このVfによって、MOSFETの駆動が不安定になる場合がある。ここで言う「不安定」とは、例えば、直列接続された上アームのスイッチング素子と下アームのスイッチング素子との間において、dv/dtなどのスイッチング特性のバランスが崩れることである。スイッチング特性のバランスが崩れると、整流回路3の挙動が不安定になることがある。
On the other hand, for example, in the case of a MOSFET formed of a wide bandgap semiconductor, the voltage characteristics of the parasitic diode are poor and the voltage effect of the parasitic diode is large. Therefore, the charge voltage of the boot capacitor needs to be as high as the Vf of the parasitic diode. Due to this Vf, the drive of the MOSFET may become unstable. The term "unstable" as used herein means that, for example, the balance of switching characteristics such as dv / dt is lost between the switching element of the upper arm and the switching element of the lower arm connected in series. If the switching characteristics are out of balance, the behavior of the rectifier circuit 3 may become unstable.
そこで、実施の形態2では、ブートコンデンサ23a,23bへのチャージを制御する新たな動作モードを用意する。ここでは、新たな動作モードを「チャージモード」と呼ぶ。また、このチャージモードを「第4の動作モード」と呼ぶ場合がある。図12は、実施の形態2におけるチャージモード時の動作波形を示す図である。なお、図12における信号波形の並びは、図6と同じである。
Therefore, in the second embodiment, a new operation mode for controlling the charge to the boot capacitors 23a and 23b is prepared. Here, the new operation mode is called a "charge mode". Further, this charge mode may be referred to as a "fourth operation mode". FIG. 12 is a diagram showing an operation waveform in the charge mode according to the second embodiment. The arrangement of the signal waveforms in FIG. 12 is the same as that in FIG.
図12の動作波形において、電源電圧Vsの極性が正の半周期では、電源電圧Vsのピーク値の前後において、コンデンサ4を充電する充電電流が流れる。このため、実施の形態2のチャージモード時においては、電源電圧Vsの極性が正の半周期では、スイッチング素子SW3-4がオンに制御される。このとき、ブートコンデンサ23bの負極端子は、寄生ダイオードではなく、MOSFETのFET本体側を介してGND電位に電気的に接続される。これによりブートコンデンサ23bに蓄えられる駆動電力の電位と制御電源22の電位との差がVf分抑制されるので、安定したMOSFETの駆動を実現できる。
In the operation waveform of FIG. 12, in a half cycle in which the polarity of the power supply voltage Vs is positive, a charging current for charging the capacitor 4 flows before and after the peak value of the power supply voltage Vs. Therefore, in the charge mode of the second embodiment, the switching element SW3-4 is controlled to be turned on in a half cycle in which the polarity of the power supply voltage Vs is positive. At this time, the negative electrode terminal of the boot capacitor 23b is electrically connected to the GND potential via the FET main body side of the MOSFET instead of the parasitic diode. As a result, the difference between the potential of the driving power stored in the boot capacitor 23b and the potential of the control power supply 22 is suppressed by Vf, so that stable MOSFET driving can be realized.
なお、電源電圧Vsの極性が負の半周期では、スイッチング素子SW3-3がオンに制御され、ブートコンデンサ23aのチャージ動作が改善される。以降の動作はブートコンデンサ23bと同様であり、重複する説明は割愛する。
In the half cycle when the polarity of the power supply voltage Vs is negative, the switching element SW3-3 is controlled to be ON, and the charging operation of the boot capacitor 23a is improved. The subsequent operation is the same as that of the boot capacitor 23b, and the duplicate description is omitted.
以上説明したように、実施の形態2に係る直流電源装置によれば、ブートコンデンサに電荷をチャージすることで、スイッチング素子SW3-1及びSW3-2を駆動することができるようになる。また、駆動回路3a,3b専用に制御電源を設けることなく、スイッチング素子SW3-1,SW3-2を駆動できるので、直流電源装置及び電力変換装置を小型化することができる。
As described above, according to the DC power supply device according to the second embodiment, the switching elements SW3-1 and SW3-2 can be driven by charging the boot capacitor with an electric charge. Further, since the switching elements SW3-1 and SW3-2 can be driven without providing a control power supply exclusively for the drive circuits 3a and 3b, the DC power supply device and the power conversion device can be miniaturized.
実施の形態3.
図13は、実施の形態3における整流回路3の動作波形を示す図である。図13に対応する動作波形図は、図7である。Embodiment 3.
FIG. 13 is a diagram showing an operation waveform of therectifier circuit 3 according to the third embodiment. The operation waveform diagram corresponding to FIG. 13 is FIG. 7.
図13は、実施の形態3における整流回路3の動作波形を示す図である。図13に対応する動作波形図は、図7である。
FIG. 13 is a diagram showing an operation waveform of the
図13の動作波形において、図7との差異点は、交流電源1の周期である電源周期ごとにPWM動作するスイッチング素子が入れ替わっている点にある。図13を用いて具体的に説明すると、まず、スイッチング素子SW3-1とスイッチング素子SW3-4とがセットで動作し、スイッチング素子SW3-2とスイッチング素子SW3-3とがセットで動作する。そして、スイッチング素子SW3-1がPWM動作しているときは、スイッチング素子SW3-4は同期整流を実現するために、PWMとは無関係にオン動作を継続する。また、スイッチング素子SW3-1のPWM動作の後に、スイッチング素子SW3-4がPWM動作となり、スイッチング素子SW3-1がPWMと無関係なオン動作を継続する動作に変化する。これらの動作は、スイッチング素子SW3-2とスイッチング素子SW3-3とのセットにおいても同様である。
In the operation waveform of FIG. 13, the difference from FIG. 7 is that the switching element that operates in PWM is replaced every power cycle, which is the cycle of the AC power supply 1. More specifically with reference to FIG. 13, first, the switching element SW3-1 and the switching element SW3-4 operate as a set, and the switching element SW3-2 and the switching element SW3-3 operate as a set. Then, when the switching element SW3-1 is in PWM operation, the switching element SW3-4 continues to operate independently of PWM in order to realize synchronous rectification. Further, after the PWM operation of the switching element SW3-1, the switching element SW3-4 becomes the PWM operation, and the switching element SW3-1 changes to the operation of continuing the on operation unrelated to the PWM. These operations are the same in the set of the switching element SW3-2 and the switching element SW3-3.
PWM動作を交互に入れ替えて実施することで、スイッチング損失が平均的に発生し、これら4つのスイッチング素子での発熱が均等化される。これにより、4つのスイッチング素子を同じ特性の部品で設計でき、部品標準化によるコストを抑えた低コストの製品を実現できる。
By alternately alternating the PWM operations, switching loss occurs on average, and the heat generated by these four switching elements is equalized. As a result, four switching elements can be designed with parts having the same characteristics, and a low-cost product can be realized with reduced costs due to parts standardization.
また、発熱量が分散されるので、放熱対策もスイッチング素子ごとに個別な設計を行う必要がなく、共通の設計が可能である。
In addition, since the amount of heat generated is dispersed, it is not necessary to individually design heat dissipation measures for each switching element, and a common design is possible.
なお、図7のように、PWM動作を交互に入れ替えない実施態様も別な観点での効果がある。例えば、発熱量を2つのスイッチング素子に多く集めることで、放熱設計を当該2つのスイッチング素子に集約することができる。これにより、当該2つのスイッチング素子の信頼性を高めることで、装置全体の信頼性を高めることができる。即ち、製品の品質面では、図7の動作であっても図13の動作であってもよく、何れの動作を実施させてもよい。
Note that, as shown in FIG. 7, an embodiment in which the PWM operations are not alternately replaced is also effective from another viewpoint. For example, by collecting a large amount of heat generated in the two switching elements, the heat dissipation design can be integrated into the two switching elements. As a result, the reliability of the entire device can be increased by increasing the reliability of the two switching elements. That is, in terms of product quality, the operation shown in FIG. 7 or the operation shown in FIG. 13 may be performed, and any operation may be performed.
なお、図13では、交流電源1の1周期ごとに2つのスイッチング素子のセットを入れ替えているが、これに限定されない。入れ替えの周期は、2周期ごとに入れ替えてもよいし、3以上の整数倍の周期で入れ替えてもよい。
Note that, in FIG. 13, the set of two switching elements is exchanged for each cycle of the AC power supply 1, but the present invention is not limited to this. The replacement cycle may be replaced every two cycles, or may be replaced at a cycle of an integral multiple of 3 or more.
また、4つのスイッチング素子の持つバラツキによって、交流電源1から流入する入力電流において、入力電流における正極性の波形と負極性の波形とが非対称形状となるアンバランスが生じることも想定される。しかしながら、上述したPWM動作を入れ替える制御を行うことにより、素子間のバラツキが均一化される。これにより、入力電流における波形のアンバランスは抑制される。
Further, it is assumed that the variation of the four switching elements causes an imbalance in which the positive and negative waveforms of the input current have an asymmetrical shape in the input current flowing from the AC power supply 1. However, by controlling the above-mentioned PWM operation to be replaced, the variation between the elements is made uniform. As a result, the waveform imbalance in the input current is suppressed.
なお、上記の制御でも、波形のアンバランスが生じる場合には、4つのスイッチング素子のうちの何れかのスイッチング素子を基準として、相対的なアンバランスが抑制されるように、PWM動作によるスイッチング素子のオン時間を調整してもよい。このような制御によっても、入力電流における波形のアンバランスを抑制できる。これにより、交流電源1の偶数次など、本来生じない高調波電流の発生を抑制することができる。また、高調波電流をより少なくできるので、交流電源1への障害を発生させない電力変換装置を提供することができる。
Even with the above control, if a waveform imbalance occurs, the switching element by PWM operation is used as a reference so that the relative imbalance is suppressed with reference to any of the four switching elements. You may adjust the on-time of. Even with such control, the imbalance of the waveform in the input current can be suppressed. As a result, it is possible to suppress the generation of harmonic currents that do not originally occur, such as the even-numbered order of the AC power supply 1. Further, since the harmonic current can be reduced, it is possible to provide a power conversion device that does not cause a failure to the AC power supply 1.
更に、ある閾値まで補正しても、アンバランスが抑制されない場合には、何れかのスイッチング素子の開放故障と判断する。開放故障は、電流が流れる経路がオープン状態であるため、通常の制御では検出することができない。ところが、PWM動作を入れ替えることで、アンバランスの発生パターンに変化が生じる。この変化により、何れのスイッチング素子が開放故障しているか否かを判別することができる。
Further, if the imbalance is not suppressed even if the correction is made to a certain threshold value, it is judged that one of the switching elements is open failure. An open failure cannot be detected by normal control because the path through which the current flows is open. However, by exchanging the PWM operation, the imbalance generation pattern changes. From this change, it is possible to determine which switching element has an open failure.
また、ある閾値までの補正を実施せずに、アンバランス発生量だけでスイッチング素子の開放故障を判別してもよい。また、アンバランス検出するため、補正を実施しない動作態様を意図的又は強制的に作成してもよい。なお、その他、幾つかの応用も想定できるが、本発明の目的を逸脱しない範囲であれば、どのような手法を用いてもよい。
Further, the open failure of the switching element may be determined only by the amount of imbalance generated without performing the correction up to a certain threshold value. Further, in order to detect imbalance, an operation mode in which correction is not performed may be intentionally or forcibly created. In addition, although some other applications can be assumed, any method may be used as long as it does not deviate from the object of the present invention.
実施の形態4.
図14は、実施の形態4における整流回路3の動作波形を示す図である。図14には、図7又は図13で示すPWM動作を行う場合でのスイッチング素子SW3-1,SW3-3の駆動波形と、PWM信号の生成に使用するキャリアの波形とが示されている。Embodiment 4.
FIG. 14 is a diagram showing an operation waveform of therectifier circuit 3 according to the fourth embodiment. FIG. 14 shows the drive waveforms of the switching elements SW3-1 and SW3-3 when the PWM operation shown in FIG. 7 or 13 is performed, and the waveforms of the carriers used for generating the PWM signal.
図14は、実施の形態4における整流回路3の動作波形を示す図である。図14には、図7又は図13で示すPWM動作を行う場合でのスイッチング素子SW3-1,SW3-3の駆動波形と、PWM信号の生成に使用するキャリアの波形とが示されている。
FIG. 14 is a diagram showing an operation waveform of the
図14において、「Sa」はスイッチング素子SW3-1を制御するための制御信号であり、「Sc」はスイッチング素子SW3-3を制御するための制御信号である。一般的なPWM制御を実現するプロセッサでは、相補PWMと呼ばれる方式が採用される。相補PWM方式では、上下アームのスイッチング素子が同時オンとならないように、排他関係となる制御信号が生成される。
In FIG. 14, “Sa” is a control signal for controlling the switching element SW3-1, and “Sc” is a control signal for controlling the switching element SW3-3. In a processor that realizes general PWM control, a method called complementary PWM is adopted. In the complementary PWM method, control signals having an exclusive relationship are generated so that the switching elements of the upper and lower arms are not turned on at the same time.
相補PWM方式では、PWM周期の基準である三角波の山もしくは谷、いずれか一方又は両方が制御の基準時刻とされる。具体的に、制御アルゴリズムが動作を開始する時刻は、山又は谷が基準とされる。また、検出信号のAD変換を行うAD変換部12~14では、検出信号をホールドする時刻、AD変換を開始するタイミング時刻などは、山又は谷が基準とされる。
In the complementary PWM method, one or both of the peaks and valleys of the triangular wave, which is the reference of the PWM cycle, is set as the control reference time. Specifically, the time when the control algorithm starts operating is based on a peak or a valley. Further, in the AD conversion units 12 to 14 that perform AD conversion of the detection signal, a peak or a valley is used as a reference for the time for holding the detection signal, the timing time for starting the AD conversion, and the like.
ここで、図14の例に示すように、変調波信号が毎キャリア周期で変更されるような動作になることも想定される。ここで、三角波のキャリアの山を基準に変調波信号のdutyを設定する。この場合、スイッチング素子SW3-1を制御する制御信号Saは、山を基準とする図中の点線に対して、左右対称の波形となる。一方、スイッチング素子SW3-3を制御する制御信号Scは、谷を基準とする図中の点線に対して、左右非対称の波形となる。また、制御信号Scの中心は、キャリア周期に対して、ズレが生じてしまう。
Here, as shown in the example of FIG. 14, it is assumed that the modulated wave signal is changed every carrier cycle. Here, the duty of the modulated wave signal is set with reference to the peak of the carrier of the triangular wave. In this case, the control signal Sa that controls the switching element SW3-1 has a waveform that is symmetrical with respect to the dotted line in the figure with respect to the mountain. On the other hand, the control signal Sc that controls the switching element SW3-3 has a left-right asymmetric waveform with respect to the dotted line in the figure with respect to the valley. Further, the center of the control signal Sc is deviated from the carrier cycle.
従って、キャリア周期の山もしくは谷の何れかで直流電流を検出し続けていると、検出電流は、スイッチング素子のオンの中央タイミングでは無くなり、検出誤差を生じさせる。この検出誤差は、入力電流のアンバランスを引き起こし、制御アルゴリズムを破綻させるトリガにもなり得る。
Therefore, if the DC current is continuously detected at either the peak or the valley of the carrier cycle, the detected current will not be at the center timing of turning on the switching element, and a detection error will occur. This detection error causes an imbalance in the input current and can also be a trigger to break the control algorithm.
上記の点を踏まえ、実施の形態4では、例えば直列接続された上下アームのスイッチング素子SW3-1,SW3-3は、必ず、制御信号Sa,Scがオンとなる期間の中心タイミングで直流電流が検出されるように、キャリアの半周期よりも短い時間でサンプリングするためのタイミング信号を発生するように制御部11を構成する。これにより、意図しない検出誤差を抑制でき、整流回路3の動作の安定性を向上させることができる。
Based on the above points, in the fourth embodiment, for example, the switching elements SW3-1 and SW3-3 of the upper and lower arms connected in series always have a direct current at the center timing of the period during which the control signals Sa and Sc are turned on. The control unit 11 is configured to generate a timing signal for sampling in a time shorter than the half cycle of the carrier so as to be detected. As a result, an unintended detection error can be suppressed, and the operational stability of the rectifier circuit 3 can be improved.
また、制御信号Sa,Scがオンとなる期間の中心タイミングでなくてもよい。例えば、制御信号Sa,Scにおけるオンパルスにおいて、オンパルスが立ち上がる時刻と直流電流の検出時刻との間の第1の時間と、オンパルスの幅である第2の時間との時比率が、制御信号Sa,Scとの間で同じであれば、パルス先頭部もしくはパルス後方部であってもよい。但し、中央部以外の場合、スイッチング素子SW3-1とスイッチング素子SW3-3とが同一dutyで無い場合がほとんどであり、両者のオンパルスの幅は異なる。従って、上記時比率を同じにするタイミング時刻を演算するのは複雑である。このため、パルス中央部を検出する手法が、簡単でよい。
Further, it does not have to be the central timing of the period during which the control signals Sa and Sc are turned on. For example, in the on-pulse of the control signals Sa and Sc, the time ratio between the first time between the time when the on-pulse rises and the time when the direct current is detected and the second time which is the width of the on-pulse is the control signal Sa, As long as it is the same as Sc, it may be a pulse head portion or a pulse rear portion. However, in most cases other than the central portion, the switching element SW3-1 and the switching element SW3-3 do not have the same duty, and the on-pulse widths of the two are different. Therefore, it is complicated to calculate the timing time at which the above time ratios are the same. Therefore, the method of detecting the central portion of the pulse may be simple.
図15は、実施の形態4における直流電流Idc1の検出タイミングの説明に使用する図である。図15には、直流電流Idc1の波形が示され、その波形上には、直流電流Idc1の検出タイミングの時刻を表す黒丸の点がプロットされている。図15において、黒丸の点のプロット間隔は、半キャリア周期である。
FIG. 15 is a diagram used for explaining the detection timing of the direct current Idc1 in the fourth embodiment. FIG. 15 shows a waveform of the direct current Idc1, and black circle dots representing the time of detection timing of the direct current Idc1 are plotted on the waveform. In FIG. 15, the plot interval of the black circle points is a half carrier period.
図15のケースは、電流検出サンプリングが比較的荒い場合の例である。また、図15では、プロセッサにおける処理開始タイミングを下向きの矢印で示している。図15において、矢印の間隔は、2キャリア周期である。この2キャリア周期は、制御アルゴリズムの周期であり、キャリア半周期で検出した検出電流の幾つかを間引いた間引き制御が実施される。
The case of FIG. 15 is an example of a case where the current detection sampling is relatively rough. Further, in FIG. 15, the processing start timing in the processor is indicated by a downward arrow. In FIG. 15, the intervals between the arrows are two carrier cycles. These two carrier cycles are the cycles of the control algorithm, and thinning control is performed by thinning out some of the detected currents detected in the carrier half cycle.
実施の形態4では、電流検出のサンプリング周期を短くできないことを想定した電流検出制御を行う。具体的には、過去の検出電流から現在の検出電流を推定する。そして、前回のサンプリング時刻と今回のサンプリング時刻との間に閾値を超えることを予測すると、閾値を超えると予測される時刻がサンプリング時刻となるようにサンプリング間隔を制御して、当該サンプリング時刻において強制的に電流を検出する制御とする。或いは、スイッチング素子のオンとオフを予測時刻において、強制的に切り替えてもよい。これらの制御により、図15に示されるような間引き制御であっても、必要時刻に電流サンプリングを実施でき、必要な電流を比較的正確に検出することができる。これにより、安価なプロセッサを使用しつつ、より高性能な昇圧機能を具備する電力変換装置を提供することができる。
In the fourth embodiment, the current detection control is performed on the assumption that the sampling cycle of the current detection cannot be shortened. Specifically, the current detected current is estimated from the past detected current. Then, when it is predicted that the threshold will be exceeded between the previous sampling time and the current sampling time, the sampling interval is controlled so that the time predicted to exceed the threshold becomes the sampling time, and forced at the sampling time. The control is to detect the current. Alternatively, the switching element may be forcibly switched on and off at the predicted time. With these controls, even with the thinning control as shown in FIG. 15, current sampling can be performed at the required time, and the required current can be detected relatively accurately. This makes it possible to provide a power conversion device having a higher performance boosting function while using an inexpensive processor.
図16は、実施の形態4における整流回路3のスイッチング素子を動作させる第1の動作波形を示す図である。図16に対応する動作波形図は、図6又は図12である。図16では、図6又は図12に示される動作波形に加え、電源電圧Vsの位相θsの波形が示されている。即ち、図16の動作を制御する実施の形態4の整流回路3では、電源電圧Vsの位相θsを検出又は演算する機能が付加されている。
FIG. 16 is a diagram showing a first operation waveform for operating the switching element of the rectifier circuit 3 according to the fourth embodiment. The operation waveform diagram corresponding to FIG. 16 is FIG. 6 or FIG. In FIG. 16, in addition to the operation waveform shown in FIG. 6 or FIG. 12, the waveform of the phase θs of the power supply voltage Vs is shown. That is, in the rectifier circuit 3 of the fourth embodiment that controls the operation of FIG. 16, a function of detecting or calculating the phase θs of the power supply voltage Vs is added.
図16において、スイッチング素子SW3-1のオン又はオフを制御する制御信号Saは、電源電圧Vsの位相θsに基づいて制御される。これにより、検出遅れなどで生じる同期整流範囲の縮小を抑制でき、より広い範囲で同期整流を実現することで高効率な電力変換装置を提供できる。
In FIG. 16, the control signal Sa that controls the on / off of the switching element SW3-1 is controlled based on the phase θs of the power supply voltage Vs. As a result, the reduction of the synchronous rectification range caused by the detection delay or the like can be suppressed, and the synchronous rectification can be realized in a wider range to provide a highly efficient power conversion device.
また、図16の例では、図示しているように、極性反転する際のスイッチング素子の同時オンを避けるため、同時オン防止位相角(図中では±5度)が設定されている。これにより、上下アーム短絡の抑制を確実に実施することができる。
Further, in the example of FIG. 16, as shown in the figure, a simultaneous on prevention phase angle (± 5 degrees in the figure) is set in order to avoid simultaneous on of the switching element when the polarity is reversed. As a result, it is possible to reliably suppress the short circuit between the upper and lower arms.
また、図17は、実施の形態4における整流回路3のスイッチング素子を動作させる第2の動作波形を示す図である。図16では、電源電圧Vsの位相θsに基づいて、スイッチング素子SW3-1を制御する制御信号Saが制御されていたが、図17では、スイッチング素子SW3-4を制御する制御信号Sdが制御されている。このとき、上アーム側のMOSFETの寄生ダイオードが逆回復特性を維持している状態で同期整流に入ることができる。これにより、電力変換装置をより安定に動作させることが可能となる。
Further, FIG. 17 is a diagram showing a second operation waveform for operating the switching element of the rectifier circuit 3 in the fourth embodiment. In FIG. 16, the control signal Sa that controls the switching element SW3-1 is controlled based on the phase θs of the power supply voltage Vs, but in FIG. 17, the control signal Sd that controls the switching element SW3-4 is controlled. ing. At this time, synchronous rectification can be performed while the parasitic diode of the MOSFET on the upper arm side maintains the reverse recovery characteristic. This makes it possible to operate the power conversion device more stably.
実施の形態5.
図18は、実施の形態5に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図18において、実施の形態5に係るモータ駆動装置100Aでは、図1に示す実施の形態1に係るモータ駆動装置100の構成において、直流電源装置50が直流電源装置50Aに置き替えられ、電力変換装置70が電力変換装置70Aに置き替えられている。直流電源装置50Aでは、リアクタ2に並列接続されるリレー15と、リアクタ2に直列接続されるリアクタ2aとが追加されている。なお、リアクタ2aを「第2のリアクタ」と呼ぶ場合がある。また、その他の構成については、実施の形態1の構成と同一又は同等であり、同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。Embodiment 5.
FIG. 18 is a diagram showing a configuration of a motor drive device including the power conversion device according to the fifth embodiment. In FIG. 18, in themotor drive device 100A according to the fifth embodiment, in the configuration of the motor drive device 100 according to the first embodiment shown in FIG. 1, the DC power supply device 50 is replaced with the DC power supply device 50A to convert power. The device 70 has been replaced by the power converter 70A. In the DC power supply device 50A, a relay 15 connected in parallel to the reactor 2 and a reactor 2a connected in series to the reactor 2 are added. The reactor 2a may be referred to as a "second reactor". Further, other configurations are the same as or equivalent to the configuration of the first embodiment, and the same or equivalent components are designated by the same reference numerals, and duplicate description will be omitted.
図18は、実施の形態5に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図18において、実施の形態5に係るモータ駆動装置100Aでは、図1に示す実施の形態1に係るモータ駆動装置100の構成において、直流電源装置50が直流電源装置50Aに置き替えられ、電力変換装置70が電力変換装置70Aに置き替えられている。直流電源装置50Aでは、リアクタ2に並列接続されるリレー15と、リアクタ2に直列接続されるリアクタ2aとが追加されている。なお、リアクタ2aを「第2のリアクタ」と呼ぶ場合がある。また、その他の構成については、実施の形態1の構成と同一又は同等であり、同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。
FIG. 18 is a diagram showing a configuration of a motor drive device including the power conversion device according to the fifth embodiment. In FIG. 18, in the
図18の構成において、リアクタ2のインダクタンス値L0と、第2のリアクタであるリアクタ2aのインダクタンス値Laとの関係は、L0>Laに設定される。リレー15の開閉は、制御部11により、スイッチング素子SW3-1~SW3-4の動作モードに応じて制御される。
In the configuration of FIG. 18, the relationship between the inductance value L0 of the reactor 2 and the inductance value La of the second reactor 2a is set to L0> La. The opening and closing of the relay 15 is controlled by the control unit 11 according to the operation modes of the switching elements SW3-1 to SW3-4.
具体的に、図6に示す第1の動作モードの場合、リレー15は開状態に制御される。このとき、回路のインダクタンス値は、L0+Laとなる。これにより、第1の動作モードに適したリアクタのインダクタンス値に設定することができる。
Specifically, in the case of the first operation mode shown in FIG. 6, the relay 15 is controlled to the open state. At this time, the inductance value of the circuit is L0 + La. Thereby, the inductance value of the reactor suitable for the first operation mode can be set.
また、図7に示す第2の動作モードの場合、リレー15は閉状態に制御される。このとき、回路のインダクタンス値は、必要最小限のインダクタンス値Laとなる。これにより、第2の動作モードに適したリアクタのインダクタンス値に設定することができる。
Further, in the case of the second operation mode shown in FIG. 7, the relay 15 is controlled to the closed state. At this time, the inductance value of the circuit becomes the minimum necessary inductance value La. Thereby, the inductance value of the reactor suitable for the second operation mode can be set.
なお、前述の通り、第2の動作モードはPWM制御を実施する動作モードである。PWM制御において、必要以上に大きいインダクタンス値のリアクタを使用すると、リアクタでの両端電圧が大きくなり過ぎて、絶縁劣化を起こす可能性がある。このため、第1の動作モードと第2の動作モードとで共通のリアクタを用いると、より高価な部品が必要となる。一方、実施の形態5では、リレー15により、動作モードに応じて、好適なリアクタのインダクタンス値で制御することができる。これにより、より高効率且つ高力率で整流回路3を動作させることができる。また、整流回路3を高調波電流の発生の少ない動作をさせることができる。また、安価且つ小型の部品で電力変換装置を実現できる。
As described above, the second operation mode is an operation mode in which PWM control is performed. If a reactor having an inductance value larger than necessary is used in PWM control, the voltage across the reactor becomes too large, which may cause insulation deterioration. Therefore, if a common reactor is used in the first operation mode and the second operation mode, more expensive parts are required. On the other hand, in the fifth embodiment, the relay 15 can control the inductance value of a suitable reactor according to the operation mode. As a result, the rectifier circuit 3 can be operated with higher efficiency and higher power factor. In addition, the rectifier circuit 3 can be operated with less generation of harmonic current. Moreover, the power conversion device can be realized with inexpensive and small parts.
更に、図18では、リレー15を用いてリアクタのインダクタンス値を切り替えて構成としているが、これに限定されない。リアクタ2の両端を短絡できる開閉器であればどのような手段を使用してもよい。
Further, in FIG. 18, the relay 15 is used to switch the inductance value of the reactor, but the configuration is not limited to this. Any means may be used as long as it is a switch capable of short-circuiting both ends of the reactor 2.
実施の形態6.
図19は、実施の形態6に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図19において、実施の形態6に係るモータ駆動装置100Bでは、図18に示す実施の形態5に係るモータ駆動装置100Aの構成において、直流電源装置50Aが直流電源装置50Bに置き替えられ、電力変換装置70Aが電力変換装置70Bに置き替えられている。直流電源装置50Bでは、整流回路3が整流回路3Aに置き替えられ、リアクタ2に並列接続されるリアクタ2a’が追加されている。なお、リアクタ2a’を「第3のリアクタ」と呼ぶ場合がある。Embodiment 6.
FIG. 19 is a diagram showing a configuration of a motor drive device including the power conversion device according to the sixth embodiment. In FIG. 19, in themotor drive device 100B according to the sixth embodiment, in the configuration of the motor drive device 100A according to the fifth embodiment shown in FIG. 18, the DC power supply device 50A is replaced with the DC power supply device 50B to convert power. The device 70A has been replaced by the power converter 70B. In the DC power supply device 50B, the rectifier circuit 3 is replaced with the rectifier circuit 3A, and the reactor 2a'that is connected in parallel to the reactor 2 is added. The reactor 2a'may be referred to as a "third reactor".
図19は、実施の形態6に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図19において、実施の形態6に係るモータ駆動装置100Bでは、図18に示す実施の形態5に係るモータ駆動装置100Aの構成において、直流電源装置50Aが直流電源装置50Bに置き替えられ、電力変換装置70Aが電力変換装置70Bに置き替えられている。直流電源装置50Bでは、整流回路3が整流回路3Aに置き替えられ、リアクタ2に並列接続されるリアクタ2a’が追加されている。なお、リアクタ2a’を「第3のリアクタ」と呼ぶ場合がある。
FIG. 19 is a diagram showing a configuration of a motor drive device including the power conversion device according to the sixth embodiment. In FIG. 19, in the
整流回路3Aは、6つのスイッチング素子で構成される。整流回路3Aには、内部に寄生ダイオードD3-1’を有するスイッチング素子SW3-1’と、内部に寄生ダイオードD3-3’を有するスイッチング素子SW3-3’とが追加されている。また、整流回路3Aには、スイッチング素子SW3-1’,SW3-3’のそれぞれを駆動する駆動回路3a’,3c’が追加されている。制御部11は、スイッチング素子SW3-1’,SW3-3’のそれぞれを制御するための制御信号Sa’,Sc’を出力する。その他の構成については、実施の形態5の構成と同一又は同等であり、同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。
The rectifier circuit 3A is composed of 6 switching elements. A switching element SW3-1'having a parasitic diode D3-1'inside and a switching element SW3-3' having a parasitic diode D3-3'inside are added to the rectifier circuit 3A. Further, drive circuits 3a'and 3c'that drive each of the switching elements SW3-1'and SW3-3' are added to the rectifier circuit 3A. The control unit 11 outputs control signals Sa'and Sc'for controlling each of the switching elements SW3-1'and SW3-3'. Other configurations are the same as or equivalent to the configuration of the fifth embodiment, and the same or equivalent components are designated by the same reference numerals, and duplicate description will be omitted.
図19において、リアクタ2aは、直列接続される素子対であるスイッチング素子SW3-1,SW3-3の接続点に接続され、リアクタ2a’は、直列接続される素子対であるスイッチング素子SW3-1’,SW3-3’の接続点に接続される。スイッチング素子SW3-1,SW3-3の組を「第1の素子対」と呼び、スイッチング素子SW3-1’,SW3-3’の組を「第2の素子対」と呼ぶ場合がある。
In FIG. 19, the reactor 2a is connected to the connection point of the switching elements SW3-1 and SW3-3 which are the element pairs connected in series, and the reactor 2a'is the switching element SW3-1 which is the element pair connected in series. It is connected to the connection point of', SW3-3'. The set of switching elements SW3-1 and SW3-3 may be referred to as a "first element pair", and the set of switching elements SW3-1'and SW3-3' may be referred to as a "second element pair".
リアクタ2a’は、リアクタ2aと同じインダクタンス値で構成される。なお、公称のインダクタンス値が等しければ、両者のインダクタンス値は等しいと見なすことができ、測定の誤差、経年変化による誤差、劣化による誤差等があっても同じと扱ってよい。また、スイッチング素子SW3-1と、スイッチング素子SW3-1’とは、互いに位相が180度反転した状態で動作する。スイッチング素子SW3-3,SW3-3’との関係も同様である。位相を180度反転した状態で動作させる制御は、インターリーブとして公知の制御方法であり、ここでの説明は割愛する。
Reactor 2a'is composed of the same inductance value as reactor 2a. If the nominal inductance values are the same, the two inductance values can be regarded as equal, and even if there are measurement errors, errors due to aging, errors due to deterioration, etc., they may be treated as the same. Further, the switching element SW3-1 and the switching element SW3-1'operate in a state in which the phases are reversed by 180 degrees. The relationship with the switching elements SW3-3 and SW3-3'is the same. The control of operating in a state where the phase is inverted by 180 degrees is a control method known as interleaving, and the description thereof is omitted here.
実施の形態6によれば、直流電源装置をインターリーブ化の構成とすることにより、インダクタンス値のより小さいリアクタを用いて構成することができる。インダクタの個数は増えるが、安価なリアクタで構成できるので、コストの増加を抑制しつつ、高調波電流の抑制が可能となる。
According to the sixth embodiment, by configuring the DC power supply device as an interleaved configuration, it is possible to configure it by using a reactor having a smaller inductance value. Although the number of inductors increases, since it can be configured with an inexpensive reactor, it is possible to suppress the harmonic current while suppressing the increase in cost.
また、実施の形態6によれば、整流回路3Aとして、一般的なインバータ用の6素子モジュールを適用することができる。これにより、汎用性及び部品調達性が向上でき、直流電源装置を安価に構成することができる。
Further, according to the sixth embodiment, a general six-element module for an inverter can be applied as the rectifier circuit 3A. As a result, versatility and parts procurement can be improved, and a DC power supply device can be constructed at low cost.
更に、実施の形態6に係る直流電源装置によれば、インターリーブ化の構成とすることで、PWM動作時の電流リップルをほぼ半減することができ、ノイズを低減することができる。また、発熱量の分散により、部品の熱寿命を均等化することができるので、直流電源装置の長寿命化を安価に実現できる。
Further, according to the DC power supply device according to the sixth embodiment, the current ripple during PWM operation can be substantially halved and noise can be reduced by adopting the interleaved configuration. Further, since the heat life of the parts can be equalized by dispersing the calorific value, the life of the DC power supply device can be extended at low cost.
実施の形態7.
図20は、実施の形態7に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図20において、実施の形態7に係るモータ駆動装置100Cでは、図1に示す実施の形態1に係るモータ駆動装置100の構成において、直流電源装置50が直流電源装置50Cに置き替えられ、電力変換装置70が電力変換装置70Cに置き替えられている。直流電源装置50Cでは、リアクタ2に直列接続されるサーミスタ16と、サーミスタ16に並列接続されるリレー15とが追加されている。その他の構成については、実施の形態1の構成と同一又は同等であり、同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。Embodiment 7.
FIG. 20 is a diagram showing a configuration of a motor drive device including the power conversion device according to the seventh embodiment. In FIG. 20, in themotor drive device 100C according to the seventh embodiment, in the configuration of the motor drive device 100 according to the first embodiment shown in FIG. 1, the DC power supply device 50 is replaced with the DC power supply device 50C to convert power. The device 70 has been replaced by the power converter 70C. In the DC power supply device 50C, a thermistor 16 connected in series to the reactor 2 and a relay 15 connected in parallel to the thermistor 16 are added. Other configurations are the same as or equivalent to the configuration of the first embodiment, the same or equivalent components are designated by the same reference numerals, and duplicate description is omitted.
図20は、実施の形態7に係る電力変換装置を含むモータ駆動装置の構成を示す図である。図20において、実施の形態7に係るモータ駆動装置100Cでは、図1に示す実施の形態1に係るモータ駆動装置100の構成において、直流電源装置50が直流電源装置50Cに置き替えられ、電力変換装置70が電力変換装置70Cに置き替えられている。直流電源装置50Cでは、リアクタ2に直列接続されるサーミスタ16と、サーミスタ16に並列接続されるリレー15とが追加されている。その他の構成については、実施の形態1の構成と同一又は同等であり、同一又は同等の構成部には同一の符号を付して、重複する説明は割愛する。
FIG. 20 is a diagram showing a configuration of a motor drive device including the power conversion device according to the seventh embodiment. In FIG. 20, in the
サーミスタ16は、正の温度係数を有するPTC(Positive Temperature Coefficient)サーミスタである。
The thermistor 16 is a PTC (Positive Temperature Coafficient) thermistor having a positive temperature coefficient.
図20において、スイッチング素子SW3-1~SW3-4のうちの少なくとも1つが短絡故障している場合、短絡故障していない他のスイッチング素子の寄生ダイオードを介して、電源短絡となり、大電流が整流回路3へ流れ込む。このとき、リアクタ2と直列にサーミスタ16が挿入されているので、サーミスタ16の発熱でサーミスタ16が高抵抗化し、短絡故障による過電流が抑制される。これにより、スイッチング素子SW3-1~SW3-4の短絡故障に起因する2次損傷が抑止されるので、スイッチング素子SW3-1~SW3-4の短絡故障が電力変換装置70C全体に及ぶのを抑止することができる。
In FIG. 20, when at least one of the switching elements SW3-1 to SW3-4 is short-circuited, the power supply is short-circuited via the parasitic diode of the other switching element that is not short-circuited, and a large current is rectified. It flows into the circuit 3. At this time, since the thermistor 16 is inserted in series with the reactor 2, the thermistor 16 increases its resistance due to the heat generated by the thermistor 16 and suppresses an overcurrent due to a short-circuit failure. As a result, secondary damage caused by the short-circuit failure of the switching elements SW3-1 to SW3-4 is suppressed, so that the short-circuit failure of the switching elements SW3-1 to SW3-4 is suppressed from extending to the entire power conversion device 70C. can do.
実施の形態8.
図21は、実施の形態8に係る冷凍サイクル装置の構成例を示す図である。図21に示す冷凍サイクル装置120は、実施の形態1~7に係るモータ駆動装置の応用例である。なお、図21では、セパレート形の空気調和機を例示しているが、セパレート形には限定されない。また、実施の形態8では、冷凍サイクル装置120が空気調和機を構成する例を説明するが、冷凍サイクル装置120は空気調和機に限定されず、冷蔵庫及び冷凍庫といった冷凍サイクルを備える機器に適用可能である。また、図21では、実施の形態1~4に係るモータ駆動装置100を冷凍サイクル装置120に使用しているが、これに限定されない。実施の形態1~4に係るモータ駆動装置100に代えて、実施の形態5~7に係るモータ駆動装置100A~100Cのうちの何れかを冷凍サイクル装置120に使用してもよい。Embodiment 8.
FIG. 21 is a diagram showing a configuration example of the refrigeration cycle device according to the eighth embodiment. Therefrigeration cycle device 120 shown in FIG. 21 is an application example of the motor drive device according to the first to seventh embodiments. Although FIG. 21 illustrates a separate type air conditioner, it is not limited to the separate type. Further, in the eighth embodiment, an example in which the refrigeration cycle device 120 constitutes an air conditioner will be described, but the refrigeration cycle device 120 is not limited to the air conditioner and can be applied to devices having a refrigeration cycle such as a refrigerator and a freezer. Is. Further, in FIG. 21, the motor drive device 100 according to the first to fourth embodiments is used for the refrigeration cycle device 120, but the present invention is not limited to this. Instead of the motor drive device 100 according to the first to fourth embodiments, any one of the motor drive devices 100A to 100C according to the fifth to seventh embodiments may be used for the refrigeration cycle device 120.
図21は、実施の形態8に係る冷凍サイクル装置の構成例を示す図である。図21に示す冷凍サイクル装置120は、実施の形態1~7に係るモータ駆動装置の応用例である。なお、図21では、セパレート形の空気調和機を例示しているが、セパレート形には限定されない。また、実施の形態8では、冷凍サイクル装置120が空気調和機を構成する例を説明するが、冷凍サイクル装置120は空気調和機に限定されず、冷蔵庫及び冷凍庫といった冷凍サイクルを備える機器に適用可能である。また、図21では、実施の形態1~4に係るモータ駆動装置100を冷凍サイクル装置120に使用しているが、これに限定されない。実施の形態1~4に係るモータ駆動装置100に代えて、実施の形態5~7に係るモータ駆動装置100A~100Cのうちの何れかを冷凍サイクル装置120に使用してもよい。
FIG. 21 is a diagram showing a configuration example of the refrigeration cycle device according to the eighth embodiment. The
図1に示すように、実施の形態8に係る冷凍サイクル装置120は、圧縮機101、四方弁102、室外熱交換器103、膨張弁104、室内熱交換器105、冷媒配管106及びモータ駆動装置100を備える。冷凍サイクル装置120では、圧縮機101、四方弁102、室外熱交換器103、膨張弁104および室内熱交換器105が冷媒配管106を介して取り付けられた冷凍サイクルを構成する。また、冷凍サイクル装置120における圧縮機101の内部には、冷媒を圧縮する圧縮機構107とこれを動作させるモータ6とが設けられている。圧縮機101のモータ6は、モータ駆動装置100と電気的に接続されている。モータ駆動装置100は、冷媒を圧縮する圧縮機101に用いられるモータ6を駆動するために用いられる。
As shown in FIG. 1, the refrigeration cycle device 120 according to the eighth embodiment includes a compressor 101, a four-way valve 102, an outdoor heat exchanger 103, an expansion valve 104, an indoor heat exchanger 105, a refrigerant pipe 106, and a motor drive device. It has 100. In the refrigeration cycle device 120, the compressor 101, the four-way valve 102, the outdoor heat exchanger 103, the expansion valve 104, and the indoor heat exchanger 105 form a refrigeration cycle to which the compressor 101 is attached via the refrigerant pipe 106. Further, inside the compressor 101 in the refrigeration cycle device 120, a compression mechanism 107 for compressing the refrigerant and a motor 6 for operating the compression mechanism 107 are provided. The motor 6 of the compressor 101 is electrically connected to the motor drive device 100. The motor drive device 100 is used to drive the motor 6 used in the compressor 101 that compresses the refrigerant.
なお、以上の実施の形態に示した構成は、本発明の内容の一例を示すものであり、別の公知の技術と組み合わせることも可能であるし、本発明の要旨を逸脱しない範囲で、構成の一部を省略、変更することも可能である。
The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.
例えば、上述した各実施の形態において、交流電源1は単相電源で説明しているが、これに限定されない。上述した各実施の形態の効果が得られるのであれば、単相電源に代えて、三相電源又は多相電源を用いてもよい。このような類の変形は、柔軟に行うことが可能である。
For example, in each of the above-described embodiments, the AC power supply 1 is described as a single-phase power supply, but the present invention is not limited to this. If the effects of each of the above-described embodiments can be obtained, a three-phase power supply or a multi-phase power supply may be used instead of the single-phase power supply. This kind of deformation can be done flexibly.
1 交流電源、2,2a,2a’ リアクタ、3,3A 整流回路、3a~3d,3a’,3c’,5a~5f 駆動回路、4 コンデンサ、5 インバータ回路、6 モータ、6a 固定子、6b 永久磁石回転子、7 電源電圧ゼロクロス検出回路、8 直流電圧検出回路、9,10 直流電流検出回路、9a,10a シャント抵抗、9b,10b 増幅器、11 制御部、12~14 AD変換部、15 リレー、16 サーミスタ、16a 正側直流母線、16b 負側直流母線、20a,20b ダイオード(ブートダイオード)、21a,21b 抵抗、22 制御電源、23a,23b コンデンサ(ブートコンデンサ)、50,50A,50B,50C 直流電源装置、70,70A,70B,70C 電力変換装置、100,100A,100B,100C モータ駆動装置、101 圧縮機、102 四方弁、103 室外熱交換器、104 膨張弁、105 室内熱交換器、106 冷媒配管、107 圧縮機構、120 冷凍サイクル装置、300 プロセッサ、302 メモリ、304 インタフェース、305 処理回路、D3-1~D3-4,D3-1’,D3-3’ 寄生ダイオード、D5-1~D5-6 ダイオード、SW3-1~SW3-4,SW3-1’,SW3-3’,SW5-1~SW5-6 スイッチング素子。
1 AC power supply, 2,2a, 2a'reactor, 3,3A rectifier circuit, 3a to 3d, 3a', 3c', 5a to 5f drive circuit, 4 capacitor, 5 inverter circuit, 6 motor, 6a stator, 6b permanent Magnet rotor, 7 power supply voltage zero cross detection circuit, 8 DC voltage detection circuit, 9, 10 DC current detection circuit, 9a, 10a shunt resistance, 9b, 10b amplifier, 11 control unit, 12-14 AD conversion unit, 15 relay, 16 thermista, 16a positive DC bus, 16b negative DC bus, 20a, 20b diode (boot diode), 21a, 21b resistor, 22 control power supply, 23a, 23b capacitor (boot capacitor), 50, 50A, 50B, 50C DC Power supply, 70, 70A, 70B, 70C power converter, 100, 100A, 100B, 100C motor drive, 101 compressor, 102 four-way valve, 103 outdoor heat exchanger, 104 expansion valve, 105 indoor heat exchanger, 106 Coolant piping, 107 compression mechanism, 120 refrigeration cycle device, 300 processor, 302 memory, 304 interface, 305 processing circuit, D3-1 to D3-4, D3-1', D3-3'parametric diode, D5-1 to D5 -6 Capacitors, SW3-1 to SW3-4, SW3-1', SW3-3', SW5-1 to SW5-6 Switching elements.
Claims (15)
- 寄生ダイオードを内部に有する金属酸化物半導体電界効果トランジスタを複数備え、複数の前記金属酸化物半導体電界効果トランジスタがブリッジ接続され、リアクタを介して交流電源に接続される整流回路と、
前記整流回路の出力端子間に接続されるコンデンサと、
前記交流電源の出力電圧である電源電圧の極性又は位相を検出する第1の検出器と、
前記整流回路に流れる第1電流を検出する第1の電流検出器と、
前記寄生ダイオードと同方向に電流が流れるタイミングで前記金属酸化物半導体電界効果トランジスタをオン動作させる同期整流を行う第1の動作モードと、
前記金属酸化物半導体電界効果トランジスタをパルス幅変調制御して前記コンデンサの両端電圧を昇圧する第2の動作モードと、を有し、
低消費電力時には、前記第1の動作モードを起動し、高消費電力時には、第2の動作モードを起動する制御部と、
を備えた直流電源装置。 A rectifier circuit having a plurality of metal oxide semiconductor field effect transistors having a parasitic diode inside, the plurality of metal oxide semiconductor field effect transistors being bridge-connected, and being connected to an AC power supply via a reactor.
A capacitor connected between the output terminals of the rectifier circuit and
A first detector that detects the polarity or phase of the power supply voltage, which is the output voltage of the AC power supply, and
A first current detector that detects the first current flowing through the rectifier circuit, and
A first operation mode in which synchronous rectification is performed to turn on the metal oxide semiconductor field effect transistor at a timing when a current flows in the same direction as the parasitic diode, and
It has a second operation mode in which the metal oxide semiconductor field effect transistor is pulse-width modulated and controlled to boost the voltage across the capacitor.
A control unit that activates the first operation mode when the power consumption is low, and activates the second operation mode when the power consumption is high.
DC power supply equipped with. - 前記制御部は、前記第1の動作モードで動作させる前記金属酸化物半導体電界効果トランジスタと異なる金属酸化物半導体電界効果トランジスタに電源短絡動作を行わせる第3の動作モードを有し、
前記第1の動作モードと前記第2の動作モードとの中間の消費電力時に前記第3の動作モードにて前記整流回路を制御する
請求項1に記載の直流電源装置。 The control unit has a third operation mode in which a metal oxide semiconductor field effect transistor different from the metal oxide semiconductor field effect transistor operated in the first operation mode is subjected to a power supply short-circuit operation.
The DC power supply device according to claim 1, wherein the rectifier circuit is controlled in the third operation mode when power consumption is intermediate between the first operation mode and the second operation mode. - 前記制御部は、前記第1電流を負極性で検出し、且つ、前記整流回路における上下アーム短絡電流を正極性で検出する
請求項1又は2に記載の直流電源装置。 The DC power supply device according to claim 1 or 2, wherein the control unit detects the first current with a negative electrode property and detects the upper and lower arm short-circuit current in the rectifier circuit with a positive electrode property. - 前記第2の動作モードにおいてパルス幅変調動作させる前記金属酸化物半導体電界効果トランジスタを前記交流電源の1周期又は複数周期ごとに入れ替える
請求項1から3の何れか1項に記載の直流電源装置。 The DC power supply device according to any one of claims 1 to 3, wherein the metal oxide semiconductor field effect transistor to be operated in pulse width modulation operation in the second operation mode is replaced every one cycle or a plurality of cycles of the AC power supply. - 前記第2の動作モードにおいて、前記交流電源から入力される電流にアンバランスが生じた場合に、前記金属酸化物半導体電界効果トランジスタのオン時間を調整することで前記アンバランスを抑制する
請求項4に記載の直流電源装置。 4. When an imbalance occurs in the current input from the AC power supply in the second operation mode, the imbalance is suppressed by adjusting the on-time of the metal oxide semiconductor field effect transistor. DC power supply device described in. - 前記アンバランスが抑制されない場合に前記金属酸化物半導体電界効果トランジスタが故障していると判定する
請求項5に記載の直流電源装置。 The DC power supply device according to claim 5, wherein it is determined that the metal oxide semiconductor field effect transistor has failed when the imbalance is not suppressed. - 前記第1の電流検出器は、直列接続された前記金属酸化物半導体電界効果トランジスタの双方がオンとなる期間の中心のタイミングで前記第1電流を検出する
請求項1から6の何れか1項に記載の直流電源装置。 The first current detector is any one of claims 1 to 6 that detects the first current at the center timing of the period during which both of the metal oxide semiconductor field effect transistors connected in series are turned on. DC power supply device described in. - 前記制御部は、前記第1電流が閾値を超える時刻を予測し、前記閾値を超えると予測される時刻がサンプリング時刻となるようにサンプリング間隔を制御する
請求項1から7の何れか1項に記載の直流電源装置。 The control unit predicts the time when the first current exceeds the threshold value, and controls the sampling interval so that the time when the first current exceeds the threshold value becomes the sampling time, according to any one of claims 1 to 7. The DC power supply described. - 前記リアクタと並列に接続された開閉器と、
前記リアクタと直列に接続され、前記リアクタよりも小さいインダクタンス容量の第2のリアクタを具備する
請求項1から8の何れか1項に記載の直流電源装置。 A switch connected in parallel with the reactor,
The DC power supply device according to any one of claims 1 to 8, which is connected in series with the reactor and includes a second reactor having an inductance capacity smaller than that of the reactor. - 前記第2のリアクタと同じインダクタンス値の第3のリアクタを備え、
前記第2のリアクタは、2つの前記金属酸化物半導体電界効果トランジスタが直列接続された第1の素子対の接続点に接続され、
前記第3のリアクタは、2つの前記金属酸化物半導体電界効果トランジスタが直列接続された第2の素子対の接続点に接続され、
前記第2のリアクタと前記第3のリアクタとは、互いに位相が180度反転した状態で動作する
請求項9に記載の直流電源装置。 A third reactor having the same inductance value as the second reactor is provided.
The second reactor is connected to the connection point of the first element pair in which the two metal oxide semiconductor field effect transistors are connected in series.
The third reactor is connected to the connection point of the second element pair in which the two metal oxide semiconductor field effect transistors are connected in series.
The DC power supply device according to claim 9, wherein the second reactor and the third reactor operate in a state in which the phases are reversed by 180 degrees. - 前記リアクタと直列接続されるサーミスタを有し、
前記サーミスタは、正の温度係数を有する
請求項1から10の何れか1項に記載の直流電源装置。 It has a thermistor connected in series with the reactor
The DC power supply device according to any one of claims 1 to 10, wherein the thermistor has a positive temperature coefficient. - 前記整流回路は、前記コンデンサの正極端子と接続された前記金属酸化物半導体電界効果トランジスタのソース端子に接続されるブートコンデンサを備え、
前記制御部は、前記ブートコンデンサを充電させる第4の動作モードを有する
請求項1から11の何れか1項に記載の直流電源装置。 The rectifier circuit comprises a boot capacitor connected to the source terminal of the metal oxide semiconductor field effect transistor connected to the positive electrode terminal of the capacitor.
The DC power supply device according to any one of claims 1 to 11, wherein the control unit has a fourth operation mode for charging the boot capacitor. - 請求項12に記載の直流電源装置と、
前記直流電源装置から供給される直流電力を交流電力に変換し、モータに前記交流電力を供給することで前記モータを駆動するインバータ回路と、
を備え
前記インバータ回路により消費電力量が変化する
電力変換装置。 The DC power supply device according to claim 12,
An inverter circuit that drives the motor by converting the DC power supplied from the DC power supply device into AC power and supplying the AC power to the motor.
A power conversion device whose power consumption is changed by the inverter circuit. - 前記制御部は、
前記ブートコンデンサに充電された電荷にて前記金属酸化物半導体電界効果トランジスタを駆動させることにより非絶縁で構成された
請求項13に記載の電力変換装置。 The control unit
The power conversion device according to claim 13, wherein the metal oxide semiconductor field effect transistor is driven by an electric charge charged in the boot capacitor to form non-insulation. - 請求項13又は14に記載の電力変換装置を備え、
前記モータは圧縮機に内蔵されたモータである
冷凍サイクル装置。 The power conversion device according to claim 13 or 14 is provided.
The motor is a refrigeration cycle device that is a motor built into a compressor.
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JP2021504633A JP7034373B2 (en) | 2019-03-08 | 2019-03-08 | DC power supply, power conversion device and refrigeration cycle device |
DE112019006992.6T DE112019006992T5 (en) | 2019-03-08 | 2019-03-08 | DC power supply device, power conversion device, and refrigeration cycle device |
CN201980093589.5A CN113544962A (en) | 2019-03-08 | 2019-03-08 | DC power supply device, power conversion device, and refrigeration cycle device |
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PCT/JP2019/009535 WO2020183553A1 (en) | 2019-03-08 | 2019-03-08 | Dc power supply device, power conversion device, and refrigeration cycle device |
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JP (1) | JP7034373B2 (en) |
CN (1) | CN113544962A (en) |
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EP4089897A1 (en) * | 2021-04-25 | 2022-11-16 | Huawei Digital Power Technologies Co., Ltd. | System and method for controlling totem pole power factor correction circuit, and power adapter |
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CN117748690B (en) * | 2024-02-19 | 2024-05-24 | 西安图为电气技术有限公司 | Charging and discharging circuit of battery and charging and discharging method of battery |
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- 2019-03-08 CN CN201980093589.5A patent/CN113544962A/en active Pending
- 2019-03-08 DE DE112019006992.6T patent/DE112019006992T5/en active Pending
- 2019-03-08 WO PCT/JP2019/009535 patent/WO2020183553A1/en active Application Filing
- 2019-03-08 JP JP2021504633A patent/JP7034373B2/en active Active
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EP4089897A1 (en) * | 2021-04-25 | 2022-11-16 | Huawei Digital Power Technologies Co., Ltd. | System and method for controlling totem pole power factor correction circuit, and power adapter |
US11923769B2 (en) | 2021-04-25 | 2024-03-05 | Huawei Digital Power Technologies Co., Ltd. | System and method for controlling totem pole power factor correction circuit, and power adapter |
Also Published As
Publication number | Publication date |
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JP7034373B2 (en) | 2022-03-11 |
JPWO2020183553A1 (en) | 2021-09-13 |
CN113544962A (en) | 2021-10-22 |
DE112019006992T5 (en) | 2021-11-18 |
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