WO2016154911A1 - 一种多载波时分复用调制/解调方法及*** - Google Patents

一种多载波时分复用调制/解调方法及*** Download PDF

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WO2016154911A1
WO2016154911A1 PCT/CN2015/075557 CN2015075557W WO2016154911A1 WO 2016154911 A1 WO2016154911 A1 WO 2016154911A1 CN 2015075557 W CN2015075557 W CN 2015075557W WO 2016154911 A1 WO2016154911 A1 WO 2016154911A1
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fourier transform
signal
point
frequency domain
inverse
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PCT/CN2015/075557
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English (en)
French (fr)
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王光宇
陈前斌
邵凯
庄陵
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重庆邮电大学
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Priority to PCT/CN2015/075557 priority Critical patent/WO2016154911A1/zh
Priority to US15/563,574 priority patent/US10541846B2/en
Publication of WO2016154911A1 publication Critical patent/WO2016154911A1/zh
Priority to US16/683,411 priority patent/US10826742B2/en
Priority to US17/086,486 priority patent/US11424974B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H04L27/2628—Inverse Fourier transform modulators, e.g. inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators
    • H04L27/263—Inverse Fourier transform modulators, e.g. inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators modification of IFFT/IDFT modulator for performance improvement
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H04L27/2634—Inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators in combination with other circuits for modulation
    • H04L27/2636—Inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators in combination with other circuits for modulation with FFT or DFT modulators, e.g. standard single-carrier frequency-division multiple access [SC-FDMA] transmitter or DFT spread orthogonal frequency division multiplexing [DFT-SOFDM]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H04L27/264—Pulse-shaped multi-carrier, i.e. not using rectangular window
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2649—Demodulators
    • H04L27/265—Fourier transform demodulators, e.g. fast Fourier transform [FFT] or discrete Fourier transform [DFT] demodulators
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2649—Demodulators
    • H04L27/26534—Pulse-shaped multi-carrier, i.e. not using rectangular window
    • H04L27/26538—Filtering per subband or per resource block, e.g. universal filtered multicarrier [UFMC] or generalized frequency division multiplexing [GFDM]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/0001—Arrangements for dividing the transmission path
    • H04L5/0003—Two-dimensional division
    • H04L5/0005—Time-frequency
    • H04L5/0007—Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/03401—PSK
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/03414—Multicarrier
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/0342—QAM

Definitions

  • the present invention relates to multi-carrier modulation and demodulation techniques, and more particularly to a multi-carrier time division multiplexing modulation/demodulation method and system (MC-TDMA).
  • MC-TDMA multi-carrier time division multiplexing modulation/demodulation method and system
  • the fourth generation mobile communication network represented by the LTE standard adopts a multi-carrier modulation technology.
  • the uplink uses single-carrier frequency division multi-user access technology (SC-FDMA), and the downlink uses orthogonal frequency division multiplexing (OFDMA) technology.
  • SC-FDMA single-carrier frequency division multi-user access technology
  • OFDMA orthogonal frequency division multiplexing
  • Multi-carrier modulation technology has the ability to resist multipath attenuation of wireless channels, because in multi-carrier modulation, high-speed signals are divided into multiple low-speed signals by IFFT, and then low-speed signals are modulated onto different sub-carriers to synthesize a signal with a long symbol period. Transfer.
  • IoT Internet of Things
  • M2M machine-to-machine communication
  • the fast Fourier transform in the multi-carrier time division multiplexing modulation method is further an NM-point fast Fourier transform, where N, M are positive integers greater than or equal to one.
  • the coefficient matrix H used in the multi-carrier time division multiplexing modulation method is further obtained by cyclically shifting M/2 of a matrix having 4N ⁇ 2N matrix element blocks.
  • the coefficient matrix H used in the multi-carrier time division multiplexing modulation method wherein the sub-matrices h i, 0 and the sub-matrices h i, 1 are arranged as follows:
  • the MDFT processing in the multi-carrier time division multiplexing modulation method includes an inverse Fourier transform, further a 2N-order M-point inverse Fu The middle leaf transform.
  • the IMDFT processing in the multi-carrier time division multiplexing demodulation method further includes an inverse interleave operation process, a Fourier transform process, and a sub-band synthesis filter process.
  • the fast inverse Fourier transform in the multi-carrier time division multiplexing demodulation method is further an NM-point fast inverse Fourier transform.
  • the IFFT processing in the multi-carrier time division multiplexing demodulation method includes a Fourier transform, further a 2N-order M-point Fourier transform.
  • the sub-band integrated filtering process included in the IMDFT process in the multi-carrier time division multiplexing demodulation method is further used for post-filtering the 2NM point frequency domain symbol signal, and then according to the coefficient matrix.
  • the transposition of H performs a right multiplication of the 2NM point frequency domain symbol signal to obtain an NM point frequency domain symbol signal.
  • a multi-carrier time division multiplexing modulation system comprising: a symbol mapping unit, a symbol sorting unit, a unit implementing fast Fourier transform, and an MDFT unit; the MDFT unit includes: a subband analysis filtering module, and a An inverse Fourier transform module, and an interleaving operation module.
  • the sub-band analysis filtering module included in the MDFT unit in the multi-carrier time division multiplexing modulation system is further configured to pre-filter the NM point frequency domain symbol signal, and then according to the prototype filter function.
  • the coefficient matrix H is constructed, and the NM point frequency domain symbol signal is right-multiplied by the coefficient matrix H to obtain a 2NM point frequency domain symbol signal.
  • the MDFT unit in the multi-carrier time division multiplexing modulation system includes an inverse Fourier transform module, and the inverse Fourier transform implemented by the MDFT unit is further a 2N-order M-point inverse Fourier Leaf transformation.
  • a multi-carrier time division multiplexing demodulation system includes an IMDFT unit, a unit implementing inverse fast Fourier transform, a symbol inverse sorting unit, and a symbol inverse mapping unit; and the IMDFT unit includes: a de-interleaving operation module , a Fourier transform module, a sub-band integrated filter module.
  • the multi-carrier time division multiplexing demodulation system includes a unit that implements an inverse fast Fourier transform, and the inverse fast Fourier transform implemented is further an NM-point reverse fast Fourier Leaf transformation, where N, M is a positive integer greater than or equal to 1.
  • the sub-band integrated filtering module included in the IMDFT unit in the multi-carrier time division multiplexing demodulation system is further used for post-filtering the 2NM point frequency domain symbol signal, and then according to the coefficient matrix H.
  • the transposition performs a right multiplication of the 2NM point frequency domain symbol signal to obtain an NM point frequency domain symbol signal.
  • FIG. 2 is a block diagram of multiple decomposition implementation of a frequency domain MDFT filter bank in MC-TDMA;
  • Figure 3 is a schematic diagram of the PAPR simulation results
  • Figure 4 shows the symbol error rate simulation result of the MC-TDMA system
  • Multi-carrier modulation techniques are widely used in fourth-generation mobile communication networks typified by the LTE standard, and in particular, the downlink uses Orthogonal Frequency Division Multiplexing (OFDMA) technology.
  • Multi-carrier modulation technology has natural anti-radio channel multipath attenuation capability, because in multi-carrier modulation, high-speed signals are divided into multiple low-speed signals by IFFT, and then low-speed signals are modulated onto different sub-carriers to synthesize a long symbol period. The signal is transmitted. Due to the extension of the symbol period, the ability of the signal to resist multipath fading is greatly improved. Therefore, the multi-carrier modulation technique is an indispensable part of high-speed wireless communication.
  • OFDMA and SC-FDMA are more sensitive to carrier drift.
  • the PAPR value of OFDMA is too large.
  • the PAPR value of SC-FDMA is small.
  • the LTE standard adopted in the current fourth generation mobile communication network adopts SC-FDMA in uplink communication. Since the multi-carrier characteristics of the SC-FDMA modulated signal are to be maintained, the LTE standard employs SC-FDMA (SC-LFDMA) in a centralized insertion mode.
  • SC-LFDMA SC-FDMA
  • the PAPR value of SC-LFDMA is lower than that of OFDMA, it still differs greatly from the theoretical minimum of PAPR, which means that its PAPR has the possibility of improvement.
  • OFDMA and SC-FDMA maintain strict synchronization of the subcarrier frequencies at the transmitting and receiving ends, which imposes high requirements on the crystal precision of the receiving and transmitting ends.
  • An alternative method to improve the anti-carrier drift capability of a multi-carrier modulation system is to replace the IFFT in the OFDMA system with a filter bank, because the frequency characteristics of the filter set prototype function are better than the frequency characteristics of the rectangular window function in the IFFT, which can better eliminate Interference between subcarriers (ICI).
  • good prototype function frequency characteristics can also improve the power spectral density of the system, reduce power leakage between sub-bands, and improve the effectiveness of signal transmission.
  • the characteristics of the filter bank multi-carrier modulation (FBMC) system are that the PAPR value is large and the delay is long. Because good frequency characteristics require very long prototype filter function coefficients. These characteristics restrict the application of FBMC multi-carrier modulation system in practice.
  • the FFT/IFFT transformed modulation symbol is a weighted sum of the input symbols (central mapping) or a repetition of the input symbols (interleaving mapping).
  • the PAPR value directly affects the battery life of the mobile terminal (mobile phone), so the lower the PAPR value for the uplink communication, the better.
  • the present invention combines the advantages of SC-FDMA and FBMC modulation techniques to successfully solve the above two problems that have plagued the physical layer of the wireless communication for many years in a modulation system.
  • the MC-TDMA multi-carrier modulation system proposed by the present invention can be used not only for high-speed communication but also for IoT and M2M communication.
  • a specific embodiment of the present invention is a multi-carrier time division multiplexing modulation and demodulation method, which performs interleaving allocation and FFT transformation on an input symbol before performing multi-carrier modulation on an input symbol, and transforms a time domain symbol into a frequency domain symbol signal. Then, the frequency domain symbol signal is subjected to MDFT processing.
  • MDFT processing the description herein is only the main process of a specific embodiment, and should not be regarded as the only embodiment. The steps are not necessary, and the whole process and its specific steps are not limited. The description in the figure and above. For example, some pre-processing steps can be performed before interleaving the input symbols, and the order of the FFT transform and the MDFT processing can be appropriately adjusted. It is apparent to those skilled in the art that various modifications and changes in form and detail may be made to the system without departing from the principles and the structure of the invention. These modifications and variations are still within the scope of the appended claims.
  • the conventional FMBC uses an integrated filter bank structure in the transmitting end, and in the present invention, transmits
  • the end uses the analysis filter bank structure, and the signal is pre-filtered to perform the IFFT transform instead of the traditional FFT transform.
  • the position of the pre-filter is between the NM point FFT and the M-point IFFT, which plays a dual role in MC-TDMA.
  • One is to reduce the PAPR value of the system by the symmetry of the filter coefficients, and the other is to divide the frequency domain symbol signal into two. Multi-carrier modulation to different sub-bands.
  • the description herein is only the main process of a specific embodiment, and should not be regarded as the only embodiment. The various steps are not necessary, and the whole process and its specific steps are not limited to the figure and the above. description of.
  • the equalizer here may use a frequency domain zero equalizer or a non-zero equalizer, and may use a blind equalizer or a non-blind equalizer.
  • Adaptive equalizers can also be used with non-adaptive equalizers.
  • the symbol mapping unit performs symbol mapping to obtain M time domain input symbol signals
  • the symbol sorting unit pairs
  • the M time domain input symbol signals are allocated to NM time points by using an interleaving allocation mode, and the FFT transform unit performs an FFT operation on the NM point time domain symbol signals to obtain an NM point frequency domain symbol signal.
  • the MDFT unit includes a subband analysis filtering portion for performing FFT transformation on the input symbol signal of the NM point transmitting end to obtain an NM point frequency domain symbol signal, constructing a coefficient matrix H according to a prototype filter function, and using a coefficient matrix H to NM.
  • the point frequency domain symbol signal is right-multiplied to obtain a 2NM point frequency domain symbol signal;
  • the IFFT transform part is used for performing 2N M-point IFFT transform on the 2NM point frequency domain symbol signal to obtain a 2NM point time domain signal;
  • the interleaving operation part for The 2NM point time domain signal is interleaved to obtain an NM point output complex symbol signal.
  • the interleaving operation part divides the 2NM point time-domain complex signal into upper and lower two channels, the previous signal has no delay, and the next signal has one delay. After the two signals of the upper and lower signals are sampled twice, they are alternately taken. The imaginary part and the imaginary part synthesize a complex symbol signal, so that the 2NM point complex signal becomes an NM point complex signal.
  • the coefficient matrix H is obtained by cyclically shifting the coefficients of the MDFT analysis filter bank by M/2 points.
  • the MDFT analysis filter bank coefficients are composed of a square root raised cosine RRC function, and the dimension of the H matrix is 2NM ⁇ NM.
  • the structure of the MC-TDMA system proposed by the invention is simple, the transmitting end is realized by input symbol interleaving and the MDFT filter bank, and the receiving end adopts an IMDFT integrated filter bank structure, and adopts an FFT transform unit.
  • the interleaving operation eliminates the interference between adjacent sub-bands, and after the interleaving operation, the 2NM point symbol signal becomes an NM point symbol signal.
  • the number of subcarriers is M, and N multicarrier modulation symbols are transmitted in N time periods respectively. By adjusting N and M, the system can obtain the optimal time-frequency resolution, so that the system has the ability to resist multipath attenuation of wireless channels. There is also the ability to resist carrier drift.
  • the MC-TDMA set has the advantage of all other modulation systems in one system, and the transmitting end uses an analysis filter bank structure.
  • the receiving end adopts an integrated filter bank structure, and the receiving end can recover the input signal of the NM point transmitting end at one time, the delay is small, and the resource occupied by the prefix can be saved, and the frequency domain zeroing equalizer is adopted, which is simple to implement.
  • FIG. 1 is a schematic structural block diagram of an MC-TDMA multi-carrier modulation system.
  • the MC-TDMA multi-carrier modulation system includes: a transmitting end and a receiving end, and the transmitting end includes a symbol mapping unit, a symbol sorting unit, and an NM-point FFT transform unit.
  • Sub-band analysis filtering, 2N-order M-point IFFT transform, interleaving operation unit; receiving end includes, de-interleaving operation unit, 2N-order M-point FFT transform unit, sub-band synthesis filter, NM-point IFFT transform unit, symbol Anti-sorting unit, symbol demapping unit.
  • the input binary bit sequence s(n) is symbol-mapped by a symbol mapping unit, and the symbol mapping can be performed by QPSK or QAM method to obtain a complex symbol signal that needs to be modulated.
  • the complex symbol signal is subjected to symbol sorting by a symbol sorting unit, and the symbol sorting adopts an interleaving sorting mode, that is, one symbol signal is inserted every N points, and for a single user (uplink transmission), between two non-zero symbol signals The value of the value is zero.
  • the value between the two non-zero symbol signals is the signal of other users, and the sorting unit processes to obtain the NM point symbol signal.
  • the subband is integrated and filtered to synthesize the full band frequency domain signal, and the synthesized frequency domain signal is sent to the NM-point IFFT transform unit to obtain the time domain signal, and the signal outputted by the IFFT transform unit is inversely coded, and finally reflected by the symbol.
  • the shot unit process obtains the reconstructed sender input symbol signal.
  • the subcarrier signals are mapped by the symbol mapping unit to obtain M symbol signals, and the M symbol signals are allocated to the L time points by the interleaving operation unit by using the interleaving symbol allocation mode.
  • the MDFT filter bank is a part before the FFT unit, and is composed of three parts.
  • the first part is a pre-filtering unit
  • the second part is an IFFT transform unit
  • the third part is a signal interleaving arithmetic unit.
  • the pre-filtering unit is composed of an analysis filter bank with a decimation rate of M/2 (the number of channels of the filter bank is M), and sub-band analysis processing is performed on the input signal x(n) to obtain M sub-band signals.
  • z -1 represents a one-bit delay.
  • N is determined by the number of users of the modulation system
  • M is determined by the number of subcarriers.
  • the M-channel output is subjected to an inverse Fourier transform (IFFT) to obtain an M-channel modulated signal.
  • IFFT inverse Fourier transform
  • the modulated signal needs to be interleaved before transmission to remove interference between sub-bands.
  • the interleaving operation includes two parts: the interleaving operation at the transmitting end and the deinterleaving operation at the receiving end.
  • the interleaving operation part of the transmitting end is composed of a decimator with a decimation value of 2, a real part of the complex number, and an imaginary unit.
  • the signal enters the interleaving operation part and is divided into upper and lower parts. There are no delays on the two roads, and there is a delay on the next road.
  • the 2NM point input becomes the NM point output, thereby ensuring that the input symbol number and output of the MC-TDMA system are identical.
  • a cyclic prefix is added to the synthesized complex symbol signal, and then transmitted to the wireless channel.
  • the method of constructing the coefficient matrix H according to the prototype filter function is as follows. If the prototype filter is an RRC (square root raised cosine function) function, the prototype filter function coefficient h(n) can be expressed as:
  • M is equal to the number of subcarriers
  • r is the roll-off factor of the RRC function, which determines RRC.
  • the stopband attenuation factor of the function filter, the range of the RRC function variable n is determined by the length NM of the prefilter.
  • the configuration of the coefficient matrix H is not limited to the above method and specific parameter settings.
  • H Constructing a coefficient matrix H, assuming that the number of access users is N, the number of subcarriers assigned by each user is M, and the H matrix is obtained by cyclically shifting M/2 of a matrix having 4N ⁇ 2N matrix element blocks, wherein
  • the size of H is 2NM ⁇ NM.
  • the matrix element blocks h 0,0 , h 0,1 , h 1,0 , h 1,1 are:
  • the NM point frequency domain signal X(k) is right-multiplied by the coefficient matrix H to obtain a 2NM point symbol signal, and then 2N times M-point IFFT (fast Fourier transform) operation is performed on the 2NM point symbol signal to obtain a 2NM point time domain. signal.
  • M-point IFFT fast Fourier transform
  • the receiving end performs the opposite operation to the transmitting end.
  • the receiving end first performs de-prefix processing on the received signal, and then performs equalization operation on the equalizer to remove channel interference.
  • the M-point FFT transform unit After being sent to the de-interleaving operation unit, the M-point FFT transform unit performs 2N M-point FFT transform to obtain the signal.
  • the frequency domain sub-band signal is synthesized by sub-band synthesis, and is sent to the NM-point IFFT transform unit to obtain the time domain signal, and the signal outputted by the transform unit is inversely coded, and the signal is reconstructed by the symbol demapping unit to obtain the transmission end reconstruction. Symbol signal.
  • the present invention can employ a frequency domain zeroing equalizer.
  • the frequency domain zero-equalizer performs FFT transformation on the de-prescribed signal to the frequency domain, and then divides the system function H(k) of the channel (the Fourier transform of the excitation response h(n) in the channel), and finally performs IFFT ( The inverse fast Fourier transform) transforms the signal that removes channel interference.
  • the simulation shows that the anti-carrier non-synchronization capability of the MC-TDMA system is more than 10 times that of OFDM.
  • the MC-TDMA multi-carrier modulation system can be used for both uplink (uplink) and downlink (downlink).
  • the MC-TDMA system can reduce the power consumption of the terminal device and reduce the power consumption of the base station device, and at the same time reduce the accuracy requirement of the terminal device clock frequency.
  • the MC-TDMA system is flexible in design and easy to implement, and can be used for both high-speed communication and IoT communication.
  • FIG. 4, and FIG. 5 are graphs showing simulation results of performance comparison of the MC-TDMA multi-carrier modulation system proposed by the present invention.
  • Figure 3 shows the simulation results of PAPR, where s represents symbol shift, solid lines represent OFDMA systems, dashed lines represent SC-FDMA systems, and dotted lines represent MC-TDMA systems.
  • the MC-TDMA system The PAPR value is the lowest, which is a significant improvement over OFDMA and SC-FDMA.
  • Figure 4 shows a comparison of symbol error rates. It can also be seen from the figure that MC-TDMA has the lowest bit error rate.
  • Figure 5 shows the performance comparison of the system's anti-subcarrier drift.
  • MC-TDMA still has good performance with 10% carrier drift, and then OFDMA and SC-FDMA systems Already unable to work. It is clear from the simulation results that the present invention proposes that the performance of the MC-TDMA system is better than that of OFDMA and SC-FDMA.
  • MC-TDMA can be used for both uplink communication and downlink communication, and can be used for both high-speed communication and non-synchronous low-speed communication.

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Abstract

本发明提出一种多载波时分复用调制解调方法以及***(MC-TDMA)。在对输入符号进行多载波调制之前先进行交织分配和FFT变换，把时域符号变换到频域符号信号进行MDFT处理。发送端采用分析滤波器组结构，信号进行前置滤波后进行IFFT变换。前置滤波器的位置在NM点FFT和M点IFFT之间，利用滤波器系数的对称性降低***的PAPR值，且把频域符号信号分到不同子带进行多载波调制。

Description

一种多载波时分复用调制/解调方法及*** 技术领域
本发明涉及多载波调制解调技术，特别涉及一种多载波时分复用调制/解调方法及***(MC-TDMA)。
背景技术
移动通信技术已经经历了四代的发展，第一代无线通信属于模拟通信，只能传语音信号，从第二代无线网开始，移动通信网进入了数字网时代，语音和数据同时可以进行传输。从第二代到***(LTE)移动通信，随着调制技术的发展，数据传送速率从14.4Kbps提高到了1Gbps。第二代无线通信采用的是高斯最小移位键控(GMSK)调制技术，第三代移动通信采用的是正交相移键控(QPSK)，这两种调制技术都属于单载波调制。
而为了提高数据传输速率，以LTE标准为代表的***移动通信网采用了多载波调制技术。在LTE标准中，上行采用单载波频分多用户接入技术(SC-FDMA)，下行采用正交频分复用多用户接入技术(OFDMA)。多载波调制技术具有抗无线信道多径衰减能力，因为在多载波调制中，高速信号通过IFFT被分成多个低速信号，然后把低速信号调制到不同子载波上，合成一个具有长符号周期的信号进行传输。
未来移动通信发展的主要动力是物联网(IoT)和机器对机器的通信(M2M)，IoT和M2M通信的特点是随机性、非同步、短数据、低延时及低功耗和低成本。因此有必要研究一种新的调制技术来满足未来无线通信发展的需求，本发明正是为了满足这些要求而提出的。
发明内容
一种多载波时分复用调制方法，其特征在于对输入符号进行交织分配；利用快速傅里叶变换进行FFT变换；把时域符号变换到频域符 号信号进行MDFT处理。
根据本发明的另一个实施例，该多载波时分复用调制方法中的MDFT处理进一步包括子带分析滤波处理，逆向傅里叶变换处理，以及交织运算处理。
根据本发明的另一个实施例，该多载波时分复用调制方法中的快速傅里叶变换进一步为一个NM-点快速傅里叶变换，这里N，M为大于或等于1的正整数。
根据本发明的另一个实施例，该多载波时分复用调制方法中的MDFT处理所包括的子带分析滤波处理进一步用于对NM点频域符号信号进行前置滤波，再根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信号。
根据本发明的另一个实施例，该多载波时分复用调制方法中所使用的系数矩阵H，进一步是由一个有4N×2N个矩阵元素块的矩阵循环右移M/2得到。
根据本发明的另一个实施例，该多载波时分复用调制方法中所使用的系数矩阵H，进一步包括如下的子矩阵hi，0和hi，1：通过把RRC原型函数系数h(n)(0＜＝n＜＝NM-1)分为N个子块(每子块包含M点)，分别由第i子块的前M/2点和后M/2点组成对角矩阵hi，0和对角矩阵hi，1，这里i为介于0到N-1之间的整数。根据本发明的另一个实施例，该多载波时分复用调制方法中所使用的系数矩阵H，其中子矩阵hi，0和子矩阵hi，1的排列方式如下：
Figure PCTCN2015075557-appb-000001
根据本发明的另一个实施例，该多载波时分复用调制方法中的MDFT处理包括的逆向傅里叶变换，进一步为一个2N次M-点逆向傅 里叶变换。
本发明还提出一种多载波时分复用解调方法，该方法包括：对接收信号去前缀，经均衡器均衡去前缀后的信号，进行IMDFT处理合成全带频域信号，合成的频域信号进行快速逆向傅里叶变换(IFFT)变换得到时域信号，再对输出的信号进行符号反排序，经符号反映射处理获得重建的发送端输入符号信号。
根据本发明的另一个实施例，该多载波时分复用解调方法中的IMDFT处理进一步包括反交织运算处理，傅里叶变换处理，以及子带综合滤波处理。
根据本发明的另一个实施例，该多载波时分复用解调方法中的快速逆向傅里叶变换，进一步为一个NM-点快速逆向傅里叶变换。
根据本发明的另一个实施例，该多载波时分复用解调方法中的IMDFT处理包括的傅里叶变换，进一步为一个2N次M-点傅里叶变换。
根据本发明的另一个实施例，该多载波时分复用解调方法中的IMDFT处理所包括的子带综合滤波处理，进一步用于对2NM点频域符号信号进行后置滤波，再根据系数矩阵H的转置对2NM点频域符号信号进行右乘得到NM点频域符号信号。
一种多载波时分复用调制***，包括：一个符号映射单元，一个符号排序单元，一个实施快速傅里叶变换的单元，一个MDFT单元；所述MDFT单元包括：一个子带分析滤波模块，一个逆向傅里叶变换模块，以及一个交织运算模块。
根据本发明的另一个实施例，该多载波时分复用调制***包括的实施快速傅里叶变换的单元，其所实施的傅里叶变换进一步为一个NM-点快速傅里叶变换，这里N，M为大于或等于1的正整数。
根据本发明的另一个实施例，该多载波时分复用调制***中的MDFT单元包括的子带分析滤波模块，进一步用于对NM点频域符号信号进行前置滤波，再根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信号。
根据本发明的另一个实施例，该多载波时分复用调制***中的MDFT单元包括的逆向傅里叶变换模块，其所实施的逆向傅里叶变换进一步为一个2N次M-点逆向傅里叶变换。
一种多载波时分复用解调***，包括一个IMDFT单元，一个实施逆向快速傅里叶变换的单元，一个符号逆排序单元，一个符号逆映射单元；所述IMDFT单元包括：一个反交织运算模块，一个傅里叶变换模块，一个子带综合滤波模块。
根据本发明的另一个实施例，该多载波时分复用解调***包括的实施逆向快速傅里叶变换的单元，其所实施的逆向快速傅里叶变换进一步为一个NM-点逆向快速傅里叶变换，这里N，M为大于或等于1的正整数。
根据本发明的另一个实施例，该多载波时分复用解调***中的IMDFT单元包括的子带综合滤波模块，进一步用于对2NM点频域符号信号进行后置滤波，再根据系数矩阵H的转置对2NM点频域符号信号进行右乘得到NM点频域符号信号。
附图说明
图1MC-TDMA多载波调制***原理结构框图；
图2MC-TDMA中的频域MDFT滤波器组多项分解实现框图；
图3PAPR模拟结果示意图；
图4MC-TDMA***符号误码率模拟结果；
图5MC-TDMA***抗载波漂移模拟结果。
具体实施方式
如本说明书和权利要求书中所示，除非上下文明确提示例外情形，“一”、“一个”、“一种”和/或“该”等词并非特指单数，也可包括复数。一般说来，术语“包括”与“包含”仅提示包括已明确标识的步骤和元素，而这些步骤和元素不构成一个排它性的罗列，方法或者设备也可能包含其它的步骤或元素。
在以LTE标准为代表的***移动通信网中广泛地使用了多载波调制技术，特别是下行链路采用了正交频分复用多用户接入(OFDMA)技术。多载波调制技术具有天然的抗无线信道多径衰减能力，因为在多载波调制中，高速信号通过IFFT被分成多个低速信号，然后把低速信号调制到不同子载波上，合成一个具有长符号周期的信号进行传输。由于符号周期的扩展，信号抵抗多径衰减的能力得到了很大提高，因此，多载波调制技术是高速无线通信不可缺少的部分。
OFDMA和SC-FDMA对载波漂移较为敏感。此外，OFDMA的PAPR值偏大。相比之下SC-FDMA的PAPR值小。在现在的***移动通信网中采用的LTE标准即在上行通信中采用SC-FDMA。由于要保持SC-FDMA调制信号的多载波特性，LTE标准采用的是集中***模式的SC-FDMA(SC-LFDMA)。尽管SC-LFDMA的PAPR值比OFDMA低，但仍然和PAPR的理论最小值有很大的差别，也就是说其PAPR还有改善的可能。OFDMA和SC-FDMA对发送和接收端的子载波频率需保持严格的同步，这对接收和发送端的晶振精度提出了很高的要求。
现有多载波调制***PAPR值大的原因是由于逆向傅里叶变换(IFFT变换)的使用。由于IFFT变换的基函数是复指数函数，而复指数函数通过随机符号的相乘叠加后幅度值变大，特别当相乘后的复指数的相位一致时，幅度值最大，这时的PAPR值也最大。降低OFDMA***中PAPR值的方法很多，但这些方法只能解决PAPR值高的问题，并不能完全解决OFDMA***子载波漂移对***性能带来的影响。
提高多载波调制***抗载波漂移能力的可选方法是用滤波器组代替OFDMA***中的IFFT，因为滤波器组原型函数频率特性比IFFT中的矩形窗口函数的频率特性好，能够较好的消除子载波之间的干扰(ICI)。此外，良好的原型函数频率特性还能提高***的功率谱密度，降低子带之间的功率泄露，提高信号传输的有效性。然而，滤波器组多载波调制(FBMC)***的特点，一是PAPR值较大，二是延时长， 因为良好的频率特性要求很长的原型滤波器函数系数。这些特点制约了FBMC多载波调制***在实践中的应用。
分析表明，降低PAPR值有两种基本方法，一是降低调制符号的功率峰值，二是缩短IFFT变换的长度。缩短IFFT变换的长度同时还能降低调制***的频率分辨率，提高***的抗载波漂移能力。由于理论最小PAPR值是原输入符号信号的PAPR，所以，如果调制输出符号能尽可能的逼近原输入符号，那么PAPR值就越低。在单载波调制中，输入符号首先进行FFT载波映射，然后再进行IFFT调制。根据载波映射的模式，经过FFT/IFFT变换后的调制符号是输入符号的加权和(集中映射)，或者是输入符号的重复(交织映射)。PAPR值直接影响移动终端(手机)的电池使用时间，因此，对于上行通信来说PAPR值越低越好。
目前所有的调制方法及***(包括已在标准中使用的***)都不能同时解决OFDMA中高PAPR和高载波漂移敏感度的问题。本发明结合了SC-FDMA和FBMC调制技术的优点，在一个调制***中成功的解决了上述两个困扰无线通信物理层多年的问题。本发明提出的MC-TDMA多载波调制***不仅可以用于高速通信也能用于IoT和M2M通信。
本发明的一个具体实施例是一种多载波时分复用调制解调方法，在对输入符号进行多载波调制之前先对输入符号进行交织分配和FFT变换，把时域符号变换到频域符号信号，然后对频域符号信号进行MDFT处理。需要注意的是，此处的描述仅仅是一个具体实施例的主要过程，不应被视为是唯一的实施例，其中的各个步骤并不是必须的，整个流程及其具体步骤也并不局限于图中和上文的描述。例如，对于输入符号进行交织分配前仍可以做相应的一些预处理步骤，而FFT变换以及MDFT处理的次序可以做适当的调整。显然，对于本领域的专业人员来说，在了解本发明内容和原理后，都可能在不背离本发明原理、结构的情况下，对此***进行形式和细节上的各种修正和改变，但是这些修正和改变仍在本发明的权利要求保护范围之内。
在MDFT处理部分，和传统的FBMC(OFDM/OQAM)调制方案与经典的MDFT处理结构有所区别的是，传统的FMBC中发送端采用的是综合滤波器组结构，而在本发明中，发送端采用分析滤波器组结构，信号进行前置滤波后进行的是IFFT变换而不是传统的FFT变换。前置滤波器的位置在NM点FFT和M点IFFT之间，其在MC-TDMA中起双重作用，一是利用滤波器系数的对称性降低***的PAPR值，二是把频域符号信号分到不同子带进行多载波调制。此处的描述仅仅是一个具体实施例的主要过程，不应被视为是唯一的实施例，其中的各个步骤并不是必须的，整个流程及其具体步骤也并不局限于图中和上文的描述。
本发明的一个进一步的具体实施例则提出一种多载波时分复用***，该***包括发送端和接收端，发送端包括：用于对二进制比特序列进行符号映射得到复数符号信号的符号映射单元；用于对输入符号进行交织分配的符号排序单元，用于把时域符号信号变换为频域符号信号的快速傅立叶变换(FFT变换)单元；作为前置滤波器对频域符号信号进行前置滤波后作IFFT变换的MDFT单元，MDFT单元采用分析滤波器组结构；接收端包括：对接收信号去前缀、均衡处理获得符号信号的去前缀单元、均衡器；用于进行反交织运算，对反交织运算获得信号进行MDFT逆变换的IMDFT单元，用于将获得的符号信号进行快速傅立叶反变换(IFFT变换)获得发送端重建信号的IFFT单元。此处的描述仅仅是一个具体实施例的主要过程，不应被视为是唯一的实施例，其中的各个单元并不是必须的，整个流程及其具体步骤也并不局限于图中和上文的描述。例如，取决于无线信道的通信质量好坏，这里的均衡器可以采用频域置零均衡器也可以采用非置零型均衡器，可以采用盲均衡器也可以采用非盲均衡器，可以采用自适应型均衡器也可以采用非自适应型均衡器。显然，类似的，对于本领域的专业人员来说，在了解本发明内容和原理后，都可能在不背离本发明原理、结构的情况下，对此***进行形式和细节上的各种修正和改变，但是这些修正和改变仍在本发明的权利要求保护范围之内。
更进一步，发送端前置滤波器的位置在NM点FFT和M点IFFT之间，所述滤波器系数具有对称性，频域符号信号被分到不同子带进行多载波调制。
对于接入用户数为N，每个用户的子载波数为M的***，总的子载波数为L＝NM，符号映射单元进行符号映射，获得M个时域输入符号信号，符号排序单元对M个时域输入符号信号采用交织分配模式分配到NM个时间点上，FFT变换单元对NM点时域符号信号进行FFT运算得到NM点频域符号信号。
更进一步，MDFT单元包括，子带分析滤波部分：用于对NM点发送端输入符号信号进行FFT变换得到NM点频域符号信号，根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信号；IFFT变换部分：用于对2NM点频域符号信号进行2N次M-点IFFT变换得到2NM点时域信号；交织运算部分：用于对2NM点时域信号进行交织运算得到NM点输出复数符号信号。交织运算部分将2NM点时域复数信号分为上、下两路，上一路信号没有延时，下一路信号有一位延时，对上、下两路信号进行2倍抽样后，分别交替取实部和虚部，合成复数符号信号，使2NM点复数信号变成NM点复数信号。系数矩阵H由MDFT分析滤波器组的系数通过M/2点循环移位得到，MDFT分析滤波器组系数由平方根升余弦RRC函数构成，H矩阵的维数是2NM×NM。所述构造系数矩阵H具体包括，由一个有M×NM个矩阵元素块的矩阵循环右移M/2得到大小为2NM×NM的矩阵H，右边移出的M/2点移进左边的M/2点，移位从第一个M×NM矩阵块开始直到第2N-1个矩阵块结束。
更进一步，接收端均衡器对符号信号去前缀后进行FFT变换获得频域信号，将频域信号除以信道的***函数，经IFFT变换得到去除信道干扰的信号。接收端IMDFT单元：用于把NM点去除信道干扰的符号信号还原为2NM点符号信号，对2NM点符号信号进行2N次M-点FFT变换得到2NM点频域信号，然后把2NM点频域信号右乘 系数矩阵H的转置矩阵H得到NM点信号。
本发明提出的MC-TDMA***结构实现简单，发送端通过输入符号交织分配和MDFT滤波器组实现，接收端采用IMDFT综合滤波器组结构，采用FFT变换单元。交织运算消除相邻子带间干扰，通过交织运算后，2NM点符号信号变成了NM点符号信号。子载波数为M，N个多载波调制符号分别在N个时间段传输，通过调整N和M可以让***得到最优的时频分辨率，从而让***既有抗无线信道多径衰减的能力也有抗载波漂移的能力，MC-TDMA集所有其他调制***的优点于一个***，发送端采用分析滤波器组结构。接收端采用综合滤波器组结构，接收端可以一次性恢复NM点发送端输入信号，延时小，并能节约前缀占用的资源，采用频域置零均衡器，实现简单。
以下结合附图和具体实例对本发明的实施作进一步详细说明。显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域的普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图将本发明应用于其它类似情景。除非从语言环境中显而易见或另做说明，图中相同标号代表相同结构和操作。
如图1所示为MC-TDMA多载波调制***原理结构框图，MC-TDMA多载波调制***包括：发送端和接收端，发送端包括，符号映射单元、符号排序单元、NM-点FFT变换单元、子带分析滤波、2N次M-点IFFT变换、交织运算单元；接收端包括，反交织运算单元，2N次M-点FFT变换单元，子带综合滤波器，NM-点IFFT变换单元，符号反排序单元，符号反映射单元。
输入二进制比特序列s(n)经符号映射单元进行符号映射，符号映射可以用QPSK或QAM方法，经过符号映射后得到需要进行调制的复数符号信号。复数符号信号经过符号排序单元对符号排序处理，符号排序采用交织排序模式，也就是说，每隔N点***一个符号信号，对于单用户(上行传输)来说，两个非零符号信号之间的值为零，对于多用户(下行传输)来说，两个非零符号信号之间的值为其他用户的信号，排序单元处理后得到NM点符号信号。获得的排序符号信号 经NM-点FFT变换单元进行处理，然后子带分析滤波单元对信号进行分析滤波后，送入2N次M-点IFFT变换单元进行变换，获得调制后的信号，所获得的信号进行交织运算，加前缀后送入信道发送到接收端。接收端对接收信号去前缀，经均衡器对去除前缀后的信号进行均衡处理，再送入反交织运算单元处理后，M-点FFT变换单元对其进行2N次M-点FFT变换获得频域子带信号，经子带综合滤波后合成全带频域信号，合成的频域信号送入NM-点IFFT变换单元得到时域信号，对IFFT变换单元输出的信号进行符号反排序，最后经符号反映射单元处理获得重建的发送端输入符号信号。
具体来说，对多用户接入调制***，假设总的子载波数为L＝NM，每个用户分到的子载波数为M。子载波信号经过符号映射单元映射后得到M个符号信号，M个符号信号经过交织运算单元采用交织符号分配模式分配到L个时间点上。对复数符号信号进行L点FFT变换，把时域符号信号变换到频域得到频域信号。
如图2所示，MDFT滤波器组为FFT单元前的部分，由三部分组成，第一部分为前置滤波单元，第二部分为IFFT变换单元，第三部分为信号交织运算单元。
MDFT滤波器组包括：子带分析滤波部分、IFFT变换部分、信号交织运算部分，子带分析滤波部分和IFFT变换部分完成前置滤波和IFFT变换后，信号交织运算部分对2NM点符号信号进行交织运算，得到NM点复数符号输出进行传输。
前置滤波单元由一个抽取率为M/2(取滤波器组的通道数为M)的分析滤波器组组成，对输入信号x(n)进行子带分析处理，获得M个子带信号。选取原型函数系数为h(n)的滤波器组，图2中z-1表示一位延时。每一帧NM点输入符号信号x(n)沿延时线进入前置滤波器，然后进行M/2抽样，得到NM路输出，NM路输出经过叠加后得到M路输出。其中，N由调制***的用户数决定，M由子载波数决定。M路输出经过傅立叶反变换(IFFT)后得到M路调制信号。调制信号在发送之前需进行交织运算，从而去除子带间的干扰。交织运算包括 发送端交织运算和接收端反交织运算两部分，发送端的交织运算部分由抽取值为2的抽取器、复数的实部和虚部运算单元组成，信号进入交织运算部分后分为上下两路，上面一路没有延时，下面一路有一位延时。上、下两路信号在进行2倍抽样后，交替取实部和虚部，提取的实部和虚部值合成一个新的复数符号信号值进行传输。相邻两通道的实部和虚部运算需要交织进行，如果前一通道上一路进行的是实部运算，那么当前通道的上一路需进行虚部运算。图2中，z-1表示一位延时，↓M/2表示M/2抽样，↑M/2表示M/2插值，h(n)表示RRC原型滤波器函数，Re{ }和IM{ }为实虚部运算，x(n)和
Figure PCTCN2015075557-appb-000002
分别表示发送端输入信号和接收端重建信号。
由于2倍抽样和交织运算，2NM点输入变成了NM点输出，从而保证MC-TDMA***的输入符号数和输出一致。经过上述处理后，对合成的复数符号信号添加循环前缀，然后进入无线信道发送。
MDFT滤波器组最优可采用如下方式，前置滤波部分根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信号，IFFT变换部分对2NM点频域符号信号进行2N次M-点IFFT变换，得到2NM点时域信号，交织运算部分对2NM点时域信号进行交织运算，得到NM点复数符号信号输出。
具体来说，根据原型滤波器函数构造系数矩阵H的方法如下，如原型滤波器为RRC(平方根升余弦函数)函数，则原型滤波器函数系数h(n)可表达为：
Figure PCTCN2015075557-appb-000003
Figure PCTCN2015075557-appb-000004
Figure PCTCN2015075557-appb-000005
其中，M等于子载波数，r表示RRC函数的滚降因子，决定RRC 函数滤波器的阻带衰减因子，RRC函数变量n的范围由前置滤波器的长度NM决定。对系数矩阵H的构造并不局限于上述方法及具体参数设定。
构造系数矩阵H，假设接入用户数为N，每个用户分到的子载波数为M，H矩阵是由一个有4N×2N个矩阵元素块的矩阵循环右移M/2得到，其中，矩阵元素块hi，0，hi，1(0＜＝i＜＝N-1)是两个对角阵，如果把RRC原型函数系数h(n)(0＜＝n＜＝NM-1)分为N个子块(每子块包含M点)，hi，0，hi，1分别由第i子块的前M/2点和后M/2点组成。H的大小为2NM×NM。在循环移位的过程中，右边移出的M/2点移进左边的M/2点。移位从第一个M×NM矩阵块开始到第2N-1个矩阵块结束。
Figure PCTCN2015075557-appb-000006
Figure PCTCN2015075557-appb-000007
Figure PCTCN2015075557-appb-000008
Figure PCTCN2015075557-appb-000009
下面我们给一个具体例子说明矩阵H，及hi，0和hi，1的构造。假如用户数N＝2，子载波数M＝8，滚降因子r＝0.5，根据RRC公式得到原型函数系数h(n)：
Figure PCTCN2015075557-appb-000010
Figure PCTCN2015075557-appb-000011
Figure PCTCN2015075557-appb-000012
矩阵元素块h0，0，h0，1，h1，0，h1，1分别为：
Figure PCTCN2015075557-appb-000013
Figure PCTCN2015075557-appb-000014
Figure PCTCN2015075557-appb-000015
Figure PCTCN2015075557-appb-000016
最后得到矩阵H等于
Figure PCTCN2015075557-appb-000017
用系数矩阵H对NM点频域信号X(k)进行右乘得到2NM点符号信号，然后，对2NM点符号信号进行2N次M-点IFFT(快速傅立叶反变换)运算，得到2NM点时域信号。
接收端进行与发送端相反的操作。接收端对接收到的信号首先进行去前缀处理，然后经均衡器进行均衡运算去除信道干扰，送入反交织运算单元处理后，M-点FFT变换单元对其进行2N次M-点FFT变换获得频域子带信号，经子带综合滤波合成全带信号，送入NM-点IFFT变换单元获得时域信号，对变换单元输出的信号进行符号反排序，经符号反映射单元处理获得发送端重建符号信号。
本发明可采用频域置零均衡器。频域置零均衡器将去前缀后的信号进行FFT变换到频域，然后除以信道的***函数H(k)(信道中激响应h(n)的傅里叶变换)，最后进行IFFT(快速傅立叶反变换)变换得到去除信道干扰的信号。
MC-TDMA是一种基于滤波器组的多载波调制***，使用修正的DFT(MDFT)滤波器组。MC-TDMA是在传统的滤波器组多载波调制***(FBMC)的基础上，通过引入单载波调制***的优点而得。MC-TDMA***具有多载波和单载波***的双重优点，既有非常低的功率峰均值比(PAPR)又有很强的抗无线信道多径衰减和载波非同步能。PAPR值接近理论最低值，比LTE标准中采用的单载波频分复用(SC-FDMA)***的PAPR值还要低。仿真表明，MC-TDMA***的抗载波非同步能力是OFDM的10倍以上。MC-TDMA多载波调制***既可用于上行通信(uplink)也可用于下行通信(downlink)。MC-TDMA***既能降低终端设备的功耗也能降低基站设备的功耗，同时可以降低对终端设备时钟频率的精度要求。MC-TDMA***具有设计灵活，实现容易的特点，既能用于高速通信也能用于物联网通信。
图3，图4，图5给出了对本发明提出的MC-TDMA多载波调制***性能比较的模拟结果图。图3给出PAPR的模拟结果，图中s表示符号移位，实线代表OFDMA***，虚线代表SC-FDMA***，点线代表MC-TDMA***。从图3中可清楚的看出，MC-TDMA*** 的PAPR值最低，比OFDMA和SC-FDMA有十分明显的改进。图4给出的是符号误码率的比较图，从图中也可看出，MC-TDMA的误码率最低。图5给出了***抗子载波漂移的性能比较，从图中可以看出，在10％的载波漂移的情况下，MC-TDMA仍然具有很好的性能，而这时OFDMA和SC-FDMA***已经不能工作。从模拟结果可以清楚的看出本发明提出MC-TDMA***比OFDMA和SC-FDMA的性能都好。MC-TDMA既可以用于上行通信也可以用于下行通信，既可以用于高速通信也可以用于非同步的低速通信。
本发明所列举的实施方式如上所述，但只是为了便于理解本发明而采用的一个案例，并非用以限定本发明。在不背离本发明思想以及实质的情况下，熟悉本领域的技术人员可根据本发明在实施的形式上或细节上做出各种相应的修改和变化，本发明的专利保护范围，仍以权利要求书所界定的范围为准。

Claims (20)

  1. 一种多载波时分复用调制方法，其特征在于：对输入符号进行交织分配；利用快速傅里叶变换进行FFT变换；把时域符号变换到频域符号信号进行MDFT处理。
  2. 根据权利要求1所述的调制方法，其特征在于：所述MDFT处理包括：子带分析滤波处理，逆向傅里叶变换处理，以及交织运算处理。
  3. 根据权利要求1所述的调制方法，其特征在于：所述快速傅里叶变换包括：其所实施的傅里叶变换为一个NM-点快速傅里叶变换，这里N，M为大于或等于1的正整数。
  4. 根据权利要求2所述的调制方法，其特征在于：所述子带分析滤波处理进一步包括，用于对NM点频域符号信号进行前置滤波，再根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信号。
  5. 根据权利要求4所述的调制方法，其特征在于：所述系数矩阵H，是由一个有4N×2N个矩阵元素块的矩阵循环右移M/2得到。
  6. 根据权利要求5所述的调制方法，其特征在于：所述系数矩阵H，包括如下的子矩阵hi，0和hi，1：通过把RRC原型函数系数h(n)(0＜＝n＜＝NM-1)分为N个子块，每子块包含M点，分别由第i子块的前M/2点和后M/2点组成对角矩阵hi，0和对角矩阵hi，1，这里i为介于0到N-1之间的整数。
  7. 根据权利要求6所述的调制方法，其特征在于：所述系数矩阵H包含的子矩阵hi，0和子矩阵hi，1的排列方式如下：
    Figure PCTCN2015075557-appb-100001
  8. 根据权利要求2所述的调制方法，其特征在于：所述逆向傅里叶 变换处理具体为：其所实施的逆向傅里叶变换为一个2N次M-点逆向傅里叶变换。
  9. 一种多载波时分复用解调方法，其特征在于：对接收信号去前缀，经均衡器均衡去前缀后的信号，进行IMDFT处理合成全带频域信号，合成的频域信号进行快速逆向傅里叶IFFT变换得到时域信号，再对输出的时域信号进行符号反排序，经符号反映射处理获得重建的发送端输入符号信号。
  10. 根据权利要求9所述的解调方法，其特征在于：所述IMDFT处理包括：反交织运算处理，傅里叶变换处理，以及子带综合滤波处理。
  11. 根据权利要求9所述的解调方法，其特征在于：所述快速逆向傅里叶变换，其实施的逆向傅里叶变换为一个NM-点快速逆向傅里叶变换。
  12. 根据权利要求10所述的解调方法，其特征在于：所述傅里叶变换，其实施的傅里叶变换为一个2N次M-点傅里叶变换。
  13. 根据权利要求10所述的解调方法，其特征在于：所述子带综合滤波处理具体为，对2NM点频域符号信号进行后置滤波，再根据系数矩阵H的转置对2NM点频域符号信号进行右乘得到NM点频域符号信号。
  14. 一种多载波时分复用调制***，其特征在于，包括：一个符号映射单元，一个符号排序单元，一个实施快速傅里叶变换的单元，一个MDFT单元，所述MDFT单元包括：一个子带分析滤波模块，一个逆向傅里叶变换模块，以及一个交织运算模块。
  15. 根据权利要求14所述的调制***，其特征在于，所述实施快速傅里叶变换的单元，其所实施的傅里叶变换为一个NM-点快速傅里叶变换，这里N，M为大于或等于1的正整数。
  16. 根据权利要求14所述的调制***，其特征在于，所述子带分析滤波模块，其用于对NM点频域符号信号进行前置滤波，再根据原型滤波器函数构造系数矩阵H，用系数矩阵H对NM点频域符号信号进行右乘得到2NM点频域符号信。
  17. 根据权利要求14所述的调制***，其特征在于，所述逆向傅 里叶变换模块，其所实施的逆向傅里叶变换为一个2N次M-点逆向傅里叶变换。
  18. 一种多载波时分复用解调***，其特征在于，包括一个IMDFT单元，一个实施逆向快速傅里叶变换的单元，一个符号逆排序单元，一个符号逆映射单元，所述IMDFT单元包括：一个反交织运算模块，一个傅里叶变换模块，一个子带综合滤波模块。
  19. 根据权利要求18所述的解调***，其特征在于，所述实施逆向快速傅里叶变换的单元，其所实施的逆向快速傅里叶变换为一个NM-点逆向快速傅里叶变换，这里N，M为大于或等于1的正整数。
  20. 根据权利要求18所述的解调***，其特征在于，所述子带综合滤波模块，其用于对2NM点频域符号信号进行后置滤波，再根据系数矩阵H的转置对2NM点频域符号信号进行右乘得到NM点频域符号信号。
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Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110971554A (zh) * 2018-09-29 2020-04-07 华为技术有限公司 数据传输方法及装置
CN113746775A (zh) * 2020-05-30 2021-12-03 华为技术有限公司 一种信号发送方法、信号接收方法与相关装置
US12034584B2 (en) 2020-05-30 2024-07-09 Huawei Technologies Co., Ltd. Signal sending method, signal receiving method, and related apparatus

Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2016154911A1 (zh) * 2015-03-31 2016-10-06 重庆邮电大学 一种多载波时分复用调制/解调方法及***
CN107926032A (zh) * 2015-09-11 2018-04-17 富士通株式会社 信息传输装置、方法以及通信***
WO2018059350A1 (zh) * 2016-09-30 2018-04-05 华为技术有限公司 一种数据处理方法、装置和***
WO2021053395A2 (en) 2019-09-18 2021-03-25 Rassini Suspensiones, S.A. De C.V. Composite spacer for leaf spring suspension

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20050047513A1 (en) * 2003-09-02 2005-03-03 Roman Vitenberg Method and system for transmission of information data over a communication line
CN102355273A (zh) * 2011-08-17 2012-02-15 清华大学 数字信道化方法及电路
CN103326972A (zh) * 2013-07-01 2013-09-25 重庆邮电大学 一种滤波器组多载频调制***及其设计方法
CN103441734A (zh) * 2013-07-02 2013-12-11 重庆邮电大学 Mdft滤波器组多载频调制***及其优化设计方法

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6658441B1 (en) * 1999-08-02 2003-12-02 Seung Pil Kim Apparatus and method for recursive parallel and pipelined fast fourier transform
US6885708B2 (en) * 2002-07-18 2005-04-26 Motorola, Inc. Training prefix modulation method and receiver
US8484272B2 (en) * 2004-08-20 2013-07-09 Qualcomm Incorporated Unified pulse shaping for multi-carrier and single-carrier waveforms
KR100989797B1 (ko) * 2008-06-09 2010-10-29 (주)에프씨아이 Ｆｆｔ/ｉｆｆｔ 연산코어
WO2016154911A1 (zh) * 2015-03-31 2016-10-06 重庆邮电大学 一种多载波时分复用调制/解调方法及***

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20050047513A1 (en) * 2003-09-02 2005-03-03 Roman Vitenberg Method and system for transmission of information data over a communication line
CN102355273A (zh) * 2011-08-17 2012-02-15 清华大学 数字信道化方法及电路
CN103326972A (zh) * 2013-07-01 2013-09-25 重庆邮电大学 一种滤波器组多载频调制***及其设计方法
CN103441734A (zh) * 2013-07-02 2013-12-11 重庆邮电大学 Mdft滤波器组多载频调制***及其优化设计方法

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110971554A (zh) * 2018-09-29 2020-04-07 华为技术有限公司 数据传输方法及装置
CN110971554B (zh) * 2018-09-29 2021-09-07 华为技术有限公司 数据传输方法及装置
CN113746775A (zh) * 2020-05-30 2021-12-03 华为技术有限公司 一种信号发送方法、信号接收方法与相关装置
CN113746775B (zh) * 2020-05-30 2023-08-04 华为技术有限公司 一种信号发送方法、信号接收方法与相关装置
US12034584B2 (en) 2020-05-30 2024-07-09 Huawei Technologies Co., Ltd. Signal sending method, signal receiving method, and related apparatus

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