WO2010092704A1 - 絶縁型スイッチング電源装置 - Google Patents
絶縁型スイッチング電源装置 Download PDFInfo
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- WO2010092704A1 WO2010092704A1 PCT/JP2009/065574 JP2009065574W WO2010092704A1 WO 2010092704 A1 WO2010092704 A1 WO 2010092704A1 JP 2009065574 W JP2009065574 W JP 2009065574W WO 2010092704 A1 WO2010092704 A1 WO 2010092704A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
- H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
Definitions
- the present invention comprises a main transformer for transmitting power from the primary side to the secondary side, and a power switch for intermittently passing a current flowing from the DC input power source to the primary coil of the main transformer, and a desired DC voltage or DC
- the present invention relates to an isolated switching power supply device that outputs current.
- FIG. 1 is a schematic circuit diagram of a DC-DC converter 58 disclosed in Patent Document 1.
- the DC-DC converter 58 is a circuit that is connected to the DC input power source Vin and outputs a voltage to the load resistor Rl.
- the DC-DC converter 58 includes a first power switch S1, a synchronous rectifier S2, an output smoothing capacitor Cout, resistors R1, R2, R3, a capacitor C1, a comparator (comparator) 52 having hysteresis characteristics, and a power switch drive circuit 54. It has.
- the DC-DC converter disclosed in Patent Document 1 is a step-down converter based on a control method obtained by improving a general “hysteresis control method”.
- the hysteresis control method is a control method that determines the on / off state of the power switch S1 by comparing the output voltage with a reference voltage with a comparator (comparator) having hysteresis characteristics. Performs switching operation.
- the general hysteresis control method has a drawback that it has a large dependency on the ripple caused by the switching operation.
- the ripple is reduced and the switching frequency is lowered.
- the control may become unstable due to the influence of noise, so it cannot be used in applications where the standard value of the ripple is severe.
- a low ESR capacitor such as a ceramic capacitor is used as the output smoothing capacitor, the ripple is not a ramp waveform but approximates a sinusoidal voltage waveform with a ⁇ / 2 phase delay. (Ie, when the power switch is turned on, the output voltage on the secondary side changes in the decreasing direction), and stable control cannot be performed.
- Bang-Bang control using the response delay times td1 and td2 of the comparator instead of using the hysteresis characteristic of the comparator is generally known.
- the output voltage increases due to the ripple in the on period of the power switch S1, exceeds the reference voltage, and then turns off the power switch S1 after the first response delay td1 (turns on the synchronous rectifier S2). After the output voltage decreases and falls below the reference voltage, the power switch S1 is turned off again (the synchronous rectifier S2 is turned on) after the second response delay td2.
- Bang-Bang control has an advantage of excellent transient response, it has a problem common to the hysteresis control method in that it has a defect that the switching operation is highly dependent on the ripple.
- a ramp wave is formed by an integrating circuit of a resistor R1 and a capacitor C1, and the output voltage is superimposed on the voltage divided by the resistors R2 and R3, thereby solving the above-described problem. Yes. Even if the ripple voltage is reduced by increasing the capacity of the output smoothing capacitor, or even if a low ESR capacitor is used as the output smoothing capacitor, stable control is possible.
- the hysteresis control method and the Bang-Bang control method are other than the step-down converter that is an on / on type non-insulated DC-DC converter (forward type DC-DC converter). Not applied. The reason is as follows.
- electromagnetic energy is stored in the excitation inductance of the inductor or main transformer during the on-time of the power switch, and the stored electromagnetic energy is stored in the power switch. It is discharged to the smoothing circuit through the rectifying element during the off period. Since the ripple voltage in the output smoothing capacitor decreases during the ON period and increases during the OFF period, the gain is inverted in the hysteresis control method and the Bang-Bang control method, and stable control cannot be performed. That is, since the output voltage on the secondary side of the main transformer is lowered during the period when the power switch is on, the control direction is reversed and it is difficult to control.
- a comparator for comparing the output voltage and the reference voltage is provided on the secondary side of the main transformer, and a power switch is provided on the primary side. It is necessary to provide both of them, and both are separated by sandwiching the main transformer.
- the voltage mode and current mode feedback control which is common in on / off type isolated DC-DC converters, is less responsive than the hysteresis control method and the Bang-Bang control method, and the error signal secondary side.
- the photocoupler used for transmission from the primary side to the primary side has a significant signal transmission delay, the maximum use temperature is limited to about 100 ° C., and there are many problems that the current transfer rate changes with time.
- the hysteresis control method and the Bang-Bang control method determine the on-timing of the power switch by comparing the output voltage with the reference voltage
- the hysteresis control method or Bang-Bang control is assumed for the on / off type DC-DC converter. Even if the method can be applied, zero voltage switching cannot be realized, and there is a problem that switching loss and noise increase.
- An object of the present invention is to provide an on / off type isolated DC-DC converter that stores electromagnetic energy in a main transformer during an on period of a power switch and releases electromagnetic energy to an output during an off period of the power switch. Switching loss is achieved by controlling the output voltage at high speed and with high stability without using a photocoupler whose current transfer rate changes with time, and by enabling zero voltage switching of the power switch. It is another object of the present invention to provide an insulating switching power supply device that reduces noise.
- the present invention is configured as follows. (1) a main transformer having a primary coil and a secondary coil and transmitting power from the primary side to the secondary side; At least one power switch connected in series to the primary coil of the main transformer and intermittently passing a current flowing from a DC input power source to the primary coil of the main transformer; A rectifying element that rectifies the voltage generated in the secondary coil of the main transformer; A smoothing circuit that smoothes the voltage rectified by the rectifying element, An insulated switching power supply device configured to store energy in the main transformer during a period when the power switch is on and to release the stored energy to a secondary side during a period when the power switch is off.
- a ramp wave generating circuit for generating a ramp wave based on a magnetic flux change generated in the main transformer;
- the ramp wave and the reference voltage are compared, or the reference voltage on which the ramp wave is superimposed is compared with the output voltage smoothed by the smoothing circuit, and a signal having a timing at which the magnitude relationship is inverted is output.
- a comparison means Timing signal transmission means for transmitting the timing signal output from the comparison means from the secondary side to the primary side; Switching control means for controlling the turn-off timing of the power switch by the timing signal transmitted by the timing signal transmission means, and controlling the on-time of the power switch; Is provided.
- the voltage to be compared with the ramp wave decreases as the output voltage decreases, and the voltage to be compared with the ramp wave increases as the output voltage increases.
- a timing signal for turning off the power switch can be generated during the ON period, and the stabilization control of the output voltage on the secondary side of the main transformer can be stably performed.
- the timing signal transmission means is a pulse transformer having a primary coil and a secondary coil.
- the ramp wave generation circuit includes an integration circuit that integrates a voltage generated in the secondary coil of the main transformer or a voltage generated at both ends of the rectifier element.
- an error voltage between the output voltage of the smoothing circuit and a predetermined reference voltage is amplified to generate an error amplification signal, and the error amplification signal is added to the ramp wave generated by the ramp wave generation circuit.
- a superimposed ramp wave correction circuit is provided. This improves the static output voltage accuracy.
- the switching control unit releases the electromagnetic energy stored in the main transformer, and detects the timing when the voltage across the power switch decreases to zero voltage or near zero voltage. And zero voltage switch control means for turning on the power switch.
- a series circuit including a switch element having a control terminal and a capacitor is connected in parallel to the primary coil of the main transformer or the power switch, Complementary on / off is repeated over a period in which both the power switch and the switch element are off, and immediately after the power switch is turned off, a surge voltage generated between the main current energizing terminals of the power switch is It has a voltage clamp circuit that absorbs and regenerates.
- the capacitor included in the series circuit is connected in series with the primary coil of the main transformer, and charging and discharging or discharging and charging are repeated in synchronization with the on / off operation of the switch element. To do.
- the voltage applied to the power switch is reduced by the voltage across the capacitor, and a switch with a lower withstand voltage can be used.
- the main transformer is provided with a switch element drive winding, and the switching control means has a voltage that is approximately proportional to the voltage generated in the primary coil generated in the switch element drive winding with the control terminal.
- a switch element drive circuit is provided that inputs to the switch elements and controls on / off of each.
- the switch element driving circuit is substantially proportional to a voltage of the switching means for turning off the switch element having the control terminal and the voltage of the primary coil for turning on the switch element having the control terminal in the switch element driving winding. And a time constant circuit for controlling the switch element having the control terminal to be turned off by the switch means after a predetermined time has elapsed.
- the driving time of the voltage clamp circuit can be arbitrarily set by the circuit constant of the time constant circuit.
- the switch means is constituted by a transistor, the transistor is connected to a control terminal of a switch element having the control terminal, and an impedance circuit and a capacitor constituting the time constant circuit are connected to the control terminal of the transistor. It is characterized by that.
- the rectifying element is a switching element that is switched by a control signal.
- the operating temperature range is relatively low. It is possible to perform output voltage control with high stability at high speed without requiring a photocoupler that is narrow and whose current transfer rate changes with time. Furthermore, by enabling zero voltage switching of the power switch, it is possible to configure an insulated switching power supply device that reduces switching loss and noise.
- FIG. 1 is a circuit diagram of an isolated switching power supply device 101 according to a first embodiment.
- FIG. It is a wave form diagram of each part of insulation type switching power supply device 101 concerning a 1st embodiment.
- It is a circuit diagram of the insulation type switching power supply device 104 based on 4th Embodiment.
- FIG. 2 is a circuit diagram of the isolated switching power supply device 101 according to the first embodiment.
- FIG. 3 is a waveform diagram of each part.
- an insulated switching power supply apparatus 101 constitutes an on / off insulated DC-DC converter (flyback DC-DC converter). Insulated switching power supply apparatus 101 is connected to DC input power supply Vin and supplies a predetermined constant voltage to load RL.
- the power conversion unit of the insulating switching power supply device 101 includes an input smoothing capacitor Cin, a power switch Q1, a main transformer T1 for power transmission, a rectifier diode RctD1, and an output smoothing capacitor Cout that constitutes a smoothing circuit.
- the control unit of the isolated switching power supply device 101 includes an overcurrent protection circuit OCP, a current detection resistor Rs, a diode D1, capacitors C1, C2, C3, and C4, a pulse transformer T2 for transmitting timing signals, and a comparator COMP having hysteresis characteristics. , A reference voltage source Vref, and resistors R1, R2, R3, and R4.
- the output voltage of the converter is divided by the resistors R3 and R4, and “voltage substantially proportional to the output voltage” Vo is input to the ( ⁇ ) input of the comparator COMP.
- the voltage of the secondary coil n2 of the main transformer T1 is integrated by the resistor R1 and the capacitor C3 to form a ramp wave, and the AC component is superimposed on the voltage (reference voltage) of the reference voltage source Vref via the capacitor C4. , Input to the (+) input of the comparator COMP.
- the integrating circuit using the resistor R1 and the capacitor C3 corresponds to a “ramp wave generating circuit” based on a change in magnetic flux generated in the main transformer.
- the comparator COMP has a hysteresis characteristic of a hysteresis width Hys.
- the COMP (+) input voltage exceeds Vo + (1/2) Hys
- the COMP output is inverted from the L level to the H level, and the COMP (+) input is performed.
- the voltage falls below Vo ⁇ (1/2) Hys
- the COMP output is inverted from the H level to the L level.
- This comparator COMP corresponds to the “comparison means” according to the present invention.
- the output voltage of the comparator COMP is applied between the gate and source of the first power switch Q1 (hereinafter referred to as “between GS”) via the pulse transformer T2, and the output voltage of the comparator COMP is at the H level.
- Q1 is on and the output voltage of the comparator COMP is at L level, Q1 is turned off.
- the pulse transformer T2 corresponds to “timing signal transmission means” according to the present invention, which transmits a signal of the inversion timing of the comparator COMP from the secondary side to the control circuit on the primary side.
- the (+) input voltage of the comparator COMP exceeds Vo + (1/2) Hys after a certain period, and the output of the comparator COMP changes from L level to H level. And the power switch Q1 is turned on.
- the circuit including the pulse transformer T2, the capacitor C1, and the diode D1 of the insulated switching power supply apparatus 101 shown in FIG. 2 corresponds to the “switching control means” according to the present invention.
- times ta to tb indicate a sudden increase in input voltage
- times tc to td indicate a sudden decrease in input voltage. Since the output voltage fluctuation due to the fluctuation of the input voltage is immediately reflected in the switching operation, that is, the on-time of the power switch Q1 changes immediately, the transient response is excellent as in the hysteresis control method. In addition, the slope of the ramp wave during the ON period of the power switch Q1 is proportional to the input voltage, so that a feedforward effect with respect to fluctuations in the input voltage occurs, and the responsiveness, output voltage accuracy, and noise resistance are improved.
- the ON period of the power switch Q1 continues until the voltage approximately proportional to the output voltage exceeds the reference voltage.
- the main transformer T1 is saturated when the ON period becomes excessive.
- the overcurrent protection circuit OCP turns off the power switch Q1, so that the magnetic saturation of the main transformer T1. Can be prevented.
- the isolated switching power supply device 101 operates in a continuous current mode, and the next on-period starts before all the electromagnetic energy stored in the main transformer T1 is released.
- the zero voltage state due to the LC resonance phenomenon between the exciting inductance of the later main transformer T1 and the parasitic capacitance parallel to the power switch Q1 does not appear. Therefore, the power switch Q1 is not a zero voltage switching operation but a hard switching operation.
- FIG. 4 is a circuit diagram of the isolated switching power supply apparatus 102 according to the second embodiment.
- FIG. 5 is a waveform diagram of each part.
- the isolated switching power supply apparatus 102 constitutes an on / off type isolated DC-DC converter (flyback DC-DC converter).
- the insulating switching power supply device 102 is connected to the DC input power supply Vin and supplies a predetermined constant voltage to the load RL.
- the power conversion unit of the isolated switching power supply 102 includes an input smoothing capacitor Cin, a first power switch Q1, a second power switch Q2, a clamp capacitor C5, a main transformer T1 for power transmission, a rectifier diode RctD1, and a smoothing circuit.
- An output smoothing capacitor Cout is provided.
- the control unit of the insulating switching power supply apparatus 102 includes a Q2 drive circuit Q2dr, a zero voltage detection circuit ZVdt, diodes D2 and D3, a multivibrator MV, inverters INV1 and INV2, capacitors C7, C8, and C9, resistors R5, R6, and R7. , R8, a switching element Q3, a pulse transformer T2 for transmitting a timing signal, an AND gate AND, a comparator COMP constituting a comparison means, and a reference voltage source Vref.
- the total excitation current of the main transformer T1 decreases linearly as shown by the dotted lines in FIGS. 5C and 5E, but the excitation current is divided into the second power switch Q2 and the rectifier diode RctD2. Therefore, each current has a waveform shown in FIGS. 5 (e) and 5 (f).
- the electromagnetic energy accumulated in the main transformer T1 is smoothed by the output smoothing capacitor Cout through the rectifier diode RctD2, converted into DC power, and supplied to the load RL.
- the surge voltage generated at both ends (between the main current energizing terminals) of the first power switch Q1 due to the leakage inductance of the main transformer T1 is absorbed.
- the absorbed energy is regenerated to the input / output of the converter by LC resonance between the exciting inductance of the main transformer T1 and the capacitor C5.
- the second power switch Q2 corresponds to a “switch element” according to the present invention.
- the inverters INV1 and INV2, the capacitor C7, and the resistor R5 constitute a multivibrator MV.
- the first power switch Q1 is driven by the oscillation operation of the multivibrator MV.
- the zero voltage detection circuit ZVdt connected to the tertiary coil n3 of the main transformer T1 is a zero voltage state or a state close to zero voltage that appears after all the electromagnetic energy stored in the main transformer is released (quasi-zero voltage state). Is detected, a trigger signal as shown in FIG. 5 (k) is generated. This trigger signal is applied to the INV1 input of the multivibrator MV via the diode D2, and inverts the multivibrator MV to turn on the first power switch Q1.
- the output voltage of the tertiary coil n3 of the main transformer T1 decreases to AC0V (zero cross). (1/4) Tr after that, the voltage across the first power switch Q1 becomes minimum. Therefore, zero voltage switching or quasi-zero voltage switching can be realized by turning on the first power switch Q1 after a delay time corresponding to (1/4) Tr after detecting the AC0V state.
- the zero voltage detection circuit ZVdt corresponds to “zero voltage switch control means” according to the present invention.
- the main transformer T1 is provided with a quaternary coil n4 corresponding to the “switch element drive winding” according to the present invention.
- the voltage of the quaternary coil n4 is applied to the “switch element drive circuit” according to the present invention.
- the voltage is applied to the corresponding Q2 drive circuit Q2dr.
- the Q2 drive circuit Q2dr includes a delay circuit composed of a series circuit of a resistor R13 and a capacitor C11 connected in series to the quaternary coil n4, a transistor Q4 as switch means for turning off the second power switch Q2, and the transistor Q4. And a time constant circuit comprising a resistor R14 which is an impedance circuit connected to the control terminal (base) and a capacitor C12 to be charged / discharged.
- the Q2 drive circuit Q2dr turns on Q2 immediately after the inversion of the transformer voltage of the main transformer T1, and the current flowing through the second power switch Q2 after a certain period of time after turning on is zero or near zero. Control to turn off with. Thereby, the switching element Q2 performs a zero current turn-off operation, and the switching loss and the switching surge at the time of turn-off are reduced.
- the Q2 drive circuit Q2dr does not need to exchange signals with the circuit that drives the first power switch Q1, it can be configured with a simple drive circuit.
- the voltage across the rectifier diode RctD2 is integrated by the resistor R6 and the capacitor C9, and the integrated voltage Vo is input to the (+) input of the comparator COMP.
- the integrated voltage includes a ramp-shaped ripple, and the average value of the integrated voltage is a value obtained by subtracting the voltage drop caused by the secondary coil n2 of the transformer T1 from the output voltage of the converter. It includes a ramp wave component that is “proportional voltage” and that gradually increases during the ON period of the first power switch Q1 and gradually decreases during the OFF period of the first power switch Q1.
- the integrating circuit including the resistor R6 and the capacitor C9 corresponds to the “ramp wave generating circuit” according to the present invention.
- the integrated voltage and the voltage (reference voltage) of the reference voltage source Vref are compared by the comparator COMP. As shown in FIG. 5I, when the integrated voltage is larger than the reference voltage, the output voltage of the comparator COMP is at the H level. It becomes. When the integrated voltage is smaller than the reference voltage, the output voltage of the comparator COMP becomes L level.
- the voltage obtained by dividing the voltage across the rectifier diode RctD2 by the resistors R7 and R8 is a voltage signal for detecting the ON period of the first power switch Q1.
- the AND gate AND generates a pulse signal that becomes H level when the output voltage of the comparator COMP is H level during the on period of Q1.
- the pulse signal is applied to the gate of the switch element Q3 via the capacitor C8 and the pulse transformer T2, the switch element Q3 is turned on, the multivibrator MV is inverted, and the first power switch Q1 is turned off.
- the pulse transformer T2 corresponds to timing signal transmission means for transmitting a timing signal at which the output of the comparator COMP is inverted from L level to H level from the secondary side to the primary side control circuit.
- the diode D3 resets the excitation state of the pulse transformer T2 after transmitting the pulse signal.
- the first power switch Q1 When the zero voltage detection circuit ZVdt detects a zero voltage state, the first power switch Q1 is turned on. When the voltage Vo that is approximately proportional to the output voltage input to the comparator COMP exceeds the voltage (reference voltage) of the reference voltage source Vref, the first power switch Q1 is turned off. By this operation, the duty ratio (on duty ratio) is controlled while maintaining zero voltage switching, and the output voltage of the converter is stabilized at a substantially constant value.
- the isolated switching power supply device 102 shown in FIG. 4 is excellent in transient response as in the hysteresis control because the change in the output voltage of the converter is immediately reflected in the off timing of the first power switch Q1.
- the slope of the ramp wave during the ON period of the first power switch Q1 is proportional to the input voltage, a feedforward effect with respect to fluctuations in the input voltage occurs, and the response, output voltage accuracy, and noise resistance are improved. .
- FIG. 6 is a circuit diagram of the isolated switching power supply device 103 according to the third embodiment.
- a voltage clamp circuit A1 formed by a second power switch Q2 and a clamp capacitor C5 is connected in parallel to the first power switch Q1.
- the other configuration is the same as that of FIG.
- FIG. 7 is a circuit diagram of the isolated switching power supply device 104 according to the fourth embodiment.
- FIG. 8 is a waveform diagram of each part.
- an insulating switching power supply device 104 constitutes an on / off type isolated DC-DC converter (flyback DC-DC converter).
- the insulating switching power supply device 104 is connected to the DC input power supply Vin and supplies a predetermined constant voltage to the load RL.
- the power conversion unit of the insulating switching power supply device 104 includes an input smoothing capacitor Cin, a first power switch Q1, a second power switch Q2, a clamp capacitor C5, a main transformer T1 for power transmission, and a rectifying element that is a rectifying element. Are provided, and the output smoothing capacitor Cout constituting the smoothing circuit.
- the control unit of the insulating switching power supply device 104 includes a Q2 drive circuit Q2dr, a zero voltage detection circuit ZVdt, diodes D2 and D3, a multivibrator MV, inverters INV1 and INV2, capacitors C7, C8, C9 and C10, resistors R5 and R6. , R7, R8, R9, R10, R11, R12, switch element Q3, pulse transformer T2 for timing signal transmission, synchronous rectifier driving circuit SRdr, AND gate AND, comparator COMP constituting comparison means, reference voltage source Vref, A ramp wave correction circuit INTG and an error amplifier AMP are provided.
- the power converter of the insulating switching power supply device 104 constitutes an asymmetrically controlled half bridge converter. That is, the first power switch Q1 and the second power switch Q2 are connected in series to the DC input power source Vin, the connection point between the first power switch Q1 and the second power switch Q2, and the DC input power source.
- a series circuit of a primary coil n1 of the main transformer T1 and a clamp capacitor C5 is connected between Vin.
- DC power input from the DC input power source Vin is converted to AC power by a complementary timing switching operation of the first power switch Q1 and the second power switch Q2.
- the first power switch Q1 When the first power switch Q1 is turned on, a linearly increasing current flows through the path of Vin (+) terminal ⁇ C5 ⁇ n1 ⁇ Q1 ⁇ Vin ( ⁇ ) terminal of the main transformer T1 (see FIG. 8C). Electromagnetic energy is accumulated in the exciting inductance of the main transformer T1.
- the transformer voltage of the main transformer T1 is inverted between the DS of Q1 and the parasitic voltage of the second power switch Q2.
- the diode and the synchronous rectifier element Rct are conducted.
- the total exciting current of the main transformer T1 decreases linearly as shown by dotted lines in FIGS. 8C, 8E, and 8F, but the exciting current is the second power switch Q2 and the synchronous rectifying element. Since the current is shunted to Rct, the respective currents have waveforms shown in FIGS. 8 (e) and 8 (g).
- the electromagnetic energy accumulated in the main transformer T1 is smoothed by the output smoothing capacitor Cout through the synchronous rectifying element Rct, converted into DC power, and supplied to the load RL.
- the Vin (+) terminal side is (+) on the clamp capacitor C5, and the primary coil n1 side of the main transformer T1 Charges having a polarity of ( ⁇ ) are accumulated and act as a pseudo DC voltage source.
- the inverters INV1, INV2, the capacitor C7, and the resistor R5 constitute a multivibrator MV.
- the first power switch Q1 is driven by the oscillation operation of the multivibrator MV.
- the zero voltage detection circuit ZVdt connected to the tertiary coil n3 of the main transformer T1 detects a zero voltage state or a state close to zero voltage (quasi-zero voltage state) that appears after the release of all electromagnetic energy stored in the main transformer T1. Then, a trigger signal as shown in FIG. This trigger signal is applied to the INV1 input of the multivibrator MV via the diode D2, and inverts the multivibrator MV to turn on the power switch Q1.
- Tr is the resonance period of the exciting inductance of the main transformer T1 and the parasitic capacitance existing in parallel with the first power switch Q1
- the output voltage of the tertiary coil n3 of the main transformer T1 decreases to AC0V after After 4Tr, the voltage across the first power switch Q1 is minimized. Therefore, zero voltage switching or quasi-zero voltage switching can be realized by turning on the first power switch Q1 after a delay time corresponding to 1/4 Tr after detecting the AC0V state.
- the main transformer T1 is provided with a quaternary coil n4 corresponding to the “switch element drive winding” according to the present invention.
- the voltage of the quaternary coil n4 is applied to the “switch element drive circuit” according to the present invention.
- the voltage is applied to the corresponding Q2 drive circuit Q2dr.
- the Q2 drive circuit Q2dr includes a delay circuit composed of a series circuit of a resistor R13 and a capacitor C11 connected in series to the quaternary coil n4, a transistor Q4 as switch means for turning off the second power switch Q2, and the transistor Q4. And a time constant circuit comprising a resistor R14 which is an impedance circuit connected to the control terminal (base) and a capacitor C12 to be charged / discharged.
- the Q2 drive circuit Q2dr turns on Q2 immediately after the inversion of the transformer voltage of the main transformer T1, and turns off when the current flowing through the second power switch Q2 is zero or near zero after a predetermined time from turning on. To control. Thereby, the switching element Q2 performs a zero current turn-off operation, and the switching loss and the switching surge at the time of turn-off are reduced.
- the Q2 drive circuit Q2dr does not need to exchange signals with the circuit that drives the first power switch Q1, it can be configured with a simple drive circuit.
- the voltage across the synchronous rectifier element Rct is integrated by the resistor R6 and the capacitor C9, and the integrated voltage Vo is input to the (+) input of the comparator COMP.
- the integrated voltage includes a ramp-like ripple, and the average value of the integrated voltage is a value obtained by subtracting a voltage drop caused by the secondary coil n2 of the transformer T1 from the output voltage of the converter. And a ramp wave component that gradually increases during the ON period of the first power switch Q1 and gradually decreases during the OFF period of the first power switch Q1.
- the integrated voltage and the voltage (reference voltage) of the reference voltage source Vref are compared by the comparator COMP. As shown in FIG. 7 (i), when the integrated voltage is larger than the reference voltage, the output voltage of the comparator COMP is set to the H level. It becomes. When the integrated voltage is smaller than the reference voltage, the output voltage of the comparator COMP becomes L level.
- the voltage obtained by dividing the voltage across the synchronous rectifier element Rct by the resistors R7 and R8 is a voltage signal for detecting the ON period of the first power switch Q1.
- the AND gate AND generates a pulse signal that becomes H level when the output voltage of the comparator COMP is H level during the on period of Q1.
- the pulse signal is applied to the gate of the switch element Q3 via the capacitor C8 and the pulse transformer T2, the switch element Q3 is turned on, the multivibrator MV is inverted, and the first power switch Q1 is turned off.
- the pulse transformer T2 corresponds to timing signal transmission means for transmitting a timing signal at which the output of the comparator COMP is inverted from L level to H level from the secondary side to the primary side control circuit.
- the diode D3 resets the excitation state of the pulse transformer T2 after transmitting the pulse signal.
- the first power switch Q1 When the zero voltage detection circuit ZVdt detects a zero voltage state, the first power switch Q1 is turned on. When the voltage Vo that is approximately proportional to the output voltage input to the comparator COMP exceeds the voltage (reference voltage) of the reference voltage source Vref, the first power switch Q1 is turned off. By this operation, the duty ratio (on duty ratio) is controlled while maintaining zero voltage switching, and the output voltage of the converter is stabilized at a substantially constant value.
- the transient response is excellent as in the hysteresis control.
- the slope of the ramp wave during the ON period of the first power switch Q1 is proportional to the input voltage, a feedforward effect with respect to fluctuations in the input voltage occurs, and the response, output voltage accuracy, and noise resistance are improved.
- the synchronous rectifier element Rct is a synchronous rectifier element using a MOSFET and is driven by a synchronous rectifier element drive circuit SRdr.
- the synchronous rectifying element driving circuit SRdr may perform an operation of turning on the synchronous rectifying element Rct immediately after the inversion of the transformer voltage of the main transformer T1, and turning off the synchronous rectifying element Rct after a lapse of a certain time after being turned on.
- the same circuit configuration as Q2dr can be applied.
- the amplitude of the ramp wave superimposed on the voltage or reference voltage that is substantially proportional to the output voltage changes due to fluctuations in the input voltage and output current, so the output voltage also varies slightly accordingly. . If the amplitude of the ramp wave is reduced, the output voltage accuracy is improved, but at the same time, the noise resistance is lowered. Therefore, there is a limit to reducing the amplitude (ripple) of the ramp wave. If the required output voltage accuracy cannot be ensured by control using a comparator, an integral correction means can be provided to improve the static output voltage accuracy by gently correcting the deviation between the output voltage and the target value. That's fine.
- the ramp wave correction circuit INTG provided in the insulating switching power supply device 104 shown in FIG. 7 is a circuit corresponding to the integral correction means.
- the ramp wave correction circuit INTG amplifies the difference between the converter output voltage divided by the error amplifier AMP with the resistors R11 and R12 and the reference voltage and superimposes it on the (+) input voltage of the comparator COMP. This improves the static output voltage accuracy. That is, since an error voltage proportional to the difference between the output voltage and the reference voltage is superimposed on the ramp wave, the error voltage is reduced and the accuracy of the output voltage is improved.
- a negative feedback circuit in which a capacitor C10 and a resistor R10 are connected in series is connected between the ( ⁇ ) input and the output of the error amplifier AMP.
- the operation of the error amplifier AMP is limited only to a low frequency region lower than the phase crossing frequency, in which the phase is delayed by 180 °, and an unnecessary response is suppressed.
- the present invention is not limited to the embodiments described above, and various applications are possible.
- other power conversion topologies such as a Cuk converter can be applied to the power conversion unit.
- a voltage obtained by rectifying and smoothing the tertiary coil of the main transformer provided on the primary side may be used instead of directly acquiring the voltage proportional to the output voltage from the output of the converter.
- the comparison means may also have a circuit configuration built in the primary side control circuit. Further, the comparison means may be constituted by a differential transistor instead of the comparator.
- the ramp wave is formed by integrating the voltages of the main transformer and the rectifying element, but the ramp wave may be formed by other methods.
- the switching frequency varies depending on the input / output conditions.
- a fixed-frequency oscillation circuit is provided in the primary control circuit, the on-timing of the power switch Q1 is determined by the fixed-frequency oscillation circuit, and the off-timing is compared. If determined by means, circuit operation at a fixed switching frequency is possible.
- “constant value control” in which the output voltage, which is the target value, is made constant, but may be applied to “follow-up control” in which the target value changes every moment.
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Abstract
Description
このようなヒステリシス制御法によるDC-DCコンバータは、制御回路の構成が簡易であり、誤差アンプによる位相遅れがないので過渡応答性に優れている。
(1)1次コイル及び2次コイルを有し、1次側から2次側へ電力を伝送する主トランスと、
前記主トランスの1次コイルに直列接続され、直流入力電源から前記主トランスの1次コイルに流れる電流を断続する、少なくとも1つの電力スイッチと、
前記主トランスの2次コイルに生じる電圧を整流する整流素子と、
前記整流素子によって整流された電圧を平滑する平滑回路と、を備え、
前記電力スイッチがオンである期間に前記主トランスにエネルギーを蓄積し、前記電力スイッチがオフである期間に前記蓄積したエネルギーを2次側に放出するように構成された絶縁型スイッチング電源装置であって、
前記主トランスに生じる磁束変化に基づくランプ波を生成するランプ波生成回路と、
前記ランプ波と基準電圧とを比較して、または前記ランプ波が重畳された基準電圧と、前記平滑回路により平滑された出力電圧とを比較して、大小関係が反転するタイミングの信号を出力する比較手段と、
前記比較手段から出力されるタイミング信号を2次側から1次側に伝送するタイミング信号伝送手段と、
前記タイミング信号伝送手段によって伝送されるタイミング信号によって、前記電力スイッチのターンオフタイミングを制御して、前記電力スイッチのオン時間を制御するスイッチング制御手段と、
を備える。
これにより、フォトカプラを用いる必要がないので、使用温度範囲が広く、電流伝達率の経時変化が小さな絶縁型スイッチング電源装置が構成できる。
これにより、前記ランプ波の傾きが前記出力電圧にほぼ比例することにより、フィードフォワード効果により、応答性、出力電圧精度、耐ノイズ性が改善される。
これにより、静的な出力電圧精度が向上する。
前記電力スイッチと前記スイッチ素子が共にオフである期間を挟んで相補的にオン・オフを繰り返し、前記電力スイッチがターンオフした直後に、前記電力スイッチの主電流通電端子間に生じるサージ電圧を前記コンデンサで吸収して回生する電圧クランプ回路を備える。
図2は、第1の実施形態に係る、絶縁型スイッチング電源装置101の回路図である。図3はその各部の波形図である。
図2において、絶縁型スイッチング電源装置101は、オン・オフ型絶縁DC-DCコンバータ(フライバック方式のDC-DCコンバータ)を構成している。絶縁型スイッチング電源装置101は、直流入力電源Vinに接続され、負荷RLへ所定の一定電圧を供給する。
まず、電力変換動作については、直流入力電源Vinから入力される直流電力は、電力スイッチQ1でスイッチングされて交流電力に変換される。電力スイッチQ1がオンすると、主トランスT1の1次コイルn1に、図3(d)に示す電流が流れ、励磁インダクタンスに電磁エネルギーが蓄積される。時刻t0において、電力スイッチQ1がオフすると、図3(c),(f)に示すように、Q1のドレイン-ソース間(以下、「D-S間」と表す。)、及び主トランスT1の電圧が反転し、整流ダイオードRctD1が導通して、図3(e)に示す電流が流れ、主トランスT1に蓄積された電磁エネルギーが平滑回路に放出される。整流ダイオードRctD1の出力は、出力平滑コンデンサCoutにより平滑されて直流電力に変換され、負荷RLに供給される。
図4は、第2の実施形態に係る、絶縁型スイッチング電源装置102の回路図である。また、図5はその各部の波形図である。
図4において、絶縁型スイッチング電源装置102は、オン・オフ型絶縁DC-DCコンバータ(フライバック方式のDC-DCコンバータ)を構成している。絶縁型スイッチング電源装置102は、直流入力電源Vinに接続され、負荷RLへ所定の一定電圧を供給する。
まず、電力変換動作については、直流入力電源Vinから入力される直流電力は、第1の電力スイッチQ1でスイッチングされて交流電力に変換される。第1の電力スイッチQ1がオンすると、主トランスT1の1次コイルn1に図5(c)に示すような直線状に増加する電流が流れ、励磁インダクタンスに電磁エネルギーが蓄積される。第1の電力スイッチQ1がオフすると、図5(b),(g)に示すように、Q1のD-S間、及び主トランスT1の電圧が反転し、第2の電力スイッチQ2の寄生ダイオードと、整流ダイオードRctD2とが導通する。主トランスT1の励磁電流の合計は、図5(c),(e)に点線で示すように直線的に減少するが、励磁電流が第2の電力スイッチQ2と、整流ダイオードRctD2とに分流するため、それぞれの電流は図5(e),(f)に示す波形になる。整流ダイオードRctD2を介して、主トランスT1に蓄積された電磁エネルギーが出力平滑コンデンサCoutにより平滑されて直流電力に変換され、負荷RLに供給される。
図6は、第3の実施形態に係る、絶縁型スイッチング電源装置103の回路図である。
この絶縁型スイッチング電源装置103は、第2の電力スイッチQ2とクランプコンデンサC5で形成される電圧クランプ回路A1を、第1の電力スイッチQ1に対して並列に接続したものである。それ以外の構成は図4と同様である。
図7は、第4の実施形態に係る、絶縁型スイッチング電源装置104の回路図である。図8はその各部の波形図である。
図7において、絶縁型スイッチング電源装置104は、オン・オフ型絶縁DC-DCコンバータ(フライバック方式のDC-DCコンバータ)を構成している。絶縁型スイッチング電源装置104は、直流入力電源Vinに接続され、負荷RLへ所定の一定電圧を供給する。
絶縁型スイッチング電源装置104の電力変換部は非対称制御のハーフブリッジコンバータを構成している。すなわち、直流入力電源Vinに、第1の電力スイッチQ1と第2の電力スイッチQ2とが直列接続され、前記第1の電力スイッチQ1と第2の電力スイッチQ2との接続点と、直流入力電源Vinとの間に主トランスT1の1次コイルn1と、クランプコンデンサC5との直列回路が接続されている。
A1…電圧クランプ回路
AMP…誤差アンプ
AND…ANDゲート
C5…クランプコンデンサ
Cin…入力平滑コンデンサ
COMP…コンパレータ
Cout…出力平滑コンデンサ
INTG…ランプ波補正回路
INV1,INV2…インバータ
MV…マルチバイブレータ
OCP…過電流保護回路
Q1…第1の電力スイッチ
Q2…第2の電力スイッチ
Q2dr…Q2駆動回路
Q3…スイッチ素子
RctD1,RctD2…整流ダイオード
Rct…同期整流素子
RL…負荷
Rs…電流検出抵抗
SRdr…同期整流素子駆動回路
T1…主トランス
T2…パルストランス
Vin…直流入力電源
Vref…基準電圧源
ZVdt…ゼロ電圧検出回路
Claims (11)
- 1次コイル及び2次コイルを有し、1次側から2次側へ電力を伝送する主トランスと、
前記主トランスの1次コイルに直列接続され、直流入力電源から前記主トランスの1次コイルに流れる電流を断続する、少なくとも1つの電力スイッチと、
前記主トランスの2次コイルに生じる電圧を整流する整流素子と、
前記整流素子によって整流された電圧を平滑する平滑回路と、を備え、
前記電力スイッチがオンである期間に前記主トランスにエネルギーを蓄積し、前記電力スイッチがオフである期間に前記蓄積したエネルギーを2次側に放出するように構成された絶縁型スイッチング電源装置であって、
前記主トランスに生じる磁束変化に基づくランプ波を生成するランプ波生成回路と、
前記ランプ波と基準電圧とを比較して、または前記ランプ波が重畳された基準電圧と、前記平滑回路により平滑された出力電圧とを比較して、大小関係が反転するタイミングの信号を出力する比較手段と、
前記比較手段から出力されるタイミング信号を2次側から1次側に伝送するタイミング信号伝送手段と、
前記タイミング信号伝送手段によって伝送されるタイミング信号によって、前記電力スイッチのターンオフタイミングを制御して、前記電力スイッチのオン時間を制御するスイッチング制御手段と、
を備えた絶縁型スイッチング電源装置。 - 前記タイミング信号伝送手段は、1次コイルと2次コイルとを有するパルストランスである、請求項1に記載の絶縁型スイッチング電源装置。
- 前記ランプ波生成回路は、前記主トランスの2次コイルに生じる電圧または前記整流素子の両端に生じる電圧を積分する積分回路である、請求項1または2に記載の絶縁型スイッチング電源装置。
- 前記平滑回路の出力電圧と所定の基準電圧との誤差電圧を増幅して誤差増幅信号を発生し、前記ランプ波生成回路が生成するランプ波に前記誤差増幅信号を重畳するランプ波補正回路を備えた、請求項1~3のいずれかに記載の絶縁型スイッチング電源装置。
- 前記スイッチング制御手段は、前記電力スイッチがターンオフした後、前記主トランスに蓄えられた電磁エネルギーを放出して、前記電力スイッチの両端電圧がゼロ電圧またはゼロ電圧付近まで低下するタイミングを検出して前記電力スイッチをターンオンするゼロ電圧スイッチ制御手段を備えた、請求項1~4のいずれかに記載の絶縁型スイッチング電源装置。
- 制御端子を有するスイッチ素子とコンデンサとを含む直列回路が、前記主トランスの1次コイルまたは前記電力スイッチに対して並列に接続され、
前記電力スイッチと前記スイッチ素子が共にオフである期間を挟んで相補的にオン・オフを繰り返し、前記電力スイッチがターンオフした直後に、前記電力スイッチの主電流通電端子間に生じるサージ電圧を前記コンデンサで吸収して回生する電圧クランプ回路を備えた、請求項1~5のいずれかに記載の絶縁型スイッチング電源装置。 - 前記直列回路に含まれる前記コンデンサは前記主トランスの1次コイルと直列に接続され、前記スイッチ素子のオン/オフ動作に同期して充電と放電、または放電と充電が繰り返されることを特徴とする、請求項6に記載の絶縁型スイッチング電源装置。
- 前記主トランスにスイッチ素子駆動巻線が設けられ、前記スイッチング制御手段は、前記スイッチ素子駆動巻線に発生する前記1次コイルに発生する電圧に略比例した電圧を前記制御端子を有するスイッチ素子に入力してそれぞれをオン/オフ制御するスイッチ素子駆動回路を備えた、請求項6または7に記載の絶縁型スイッチング電源装置。
- 前記スイッチ素子駆動回路は、前記制御端子を有するスイッチ素子をターンオフさせるスイッチ手段と、前記スイッチ素子駆動巻線に前記制御端子を有するスイッチ素子をターンオンさせる前記1次コイルの電圧に略比例した電圧を発生させてから、所定の時間後に前記制御端子を有するスイッチ素子を前記スイッチ手段によってターンオフさせるように制御する時定数回路とを備えた、請求項8に記載の絶縁型スイッチング電源装置。
- 前記スイッチ手段はトランジスタで構成され、前記トランジスタが前記制御端子を有するスイッチ素子の制御端子に接続され、前記トランジスタの制御端子に前記時定数回路を構成するインピーダンス回路及びコンデンサが接続された、請求項9に記載の絶縁型スイッチング電源装置。
- 前記整流素子は、制御信号によりスイッチングされるスイッチング素子である、請求項1~10のいずれかに記載の絶縁型スイッチング電源装置。
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Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
DE102011100644A1 (de) * | 2011-05-05 | 2012-11-08 | Minebea Co., Ltd. | Gleichspannungswandler |
WO2018179694A1 (ja) * | 2017-03-29 | 2018-10-04 | Fdk株式会社 | 絶縁型スイッチング電源 |
CN108933515A (zh) * | 2017-05-19 | 2018-12-04 | 英飞凌科技奥地利有限公司 | 反激式转换器控制器、反激式转换器及其操作方法 |
Families Citing this family (11)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US9007052B2 (en) * | 2012-07-26 | 2015-04-14 | Hamilton Sundstrand Space Systems International, Inc. | Voltage sensing in isolated converters |
KR101434049B1 (ko) * | 2012-12-07 | 2014-09-02 | 삼성전기주식회사 | 전원 공급 장치 |
DE102012223274A1 (de) * | 2012-12-14 | 2014-06-18 | Siemens Aktiengesellschaft | Anordnung mit potenzialgetrennter Stromversorgungseinrichtung |
US20150365003A1 (en) * | 2014-06-12 | 2015-12-17 | Laurence P. Sadwick | Power Conversion System |
CN105490545B (zh) * | 2014-09-19 | 2018-07-13 | 万国半导体(开曼)股份有限公司 | 固定导通时间切换式转换装置 |
US10483859B2 (en) * | 2015-11-12 | 2019-11-19 | Rohm Co., Ltd. | AC/DC converter including a bidirectional switch |
US10236777B2 (en) * | 2017-08-09 | 2019-03-19 | L3 Cincinnati Electronics Corporation | Magnetically isolated feedback circuits and regulated power supplies incorporating the same |
TWI688195B (zh) * | 2019-06-19 | 2020-03-11 | 宏碁股份有限公司 | 電源供應器 |
US11114945B2 (en) * | 2019-08-22 | 2021-09-07 | Cypress Semiconductor Corporation | Secondary-controlled active clamp implementation for improved efficiency |
US11165352B2 (en) | 2020-01-16 | 2021-11-02 | L3 Cincinnati Electronics Corporation | Capacitively isolated feedback circuits and regulated power supplies incorporating the same |
TWI726758B (zh) * | 2020-07-01 | 2021-05-01 | 宏碁股份有限公司 | 消除振鈴效應之電源供應器 |
Citations (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2002333653A (ja) * | 2001-05-08 | 2002-11-22 | Canon Inc | 発光装置及びカメラ |
JP2006006057A (ja) * | 2004-06-18 | 2006-01-05 | Murata Mfg Co Ltd | フォワードコンバータ |
WO2007018227A1 (ja) * | 2005-08-11 | 2007-02-15 | Murata Manufacturing Co., Ltd. | 絶縁型スイッチング電源装置 |
WO2007069403A1 (ja) * | 2005-12-16 | 2007-06-21 | Murata Manufacturing Co., Ltd. | 複合トランスおよび絶縁型スイッチング電源装置 |
Family Cites Families (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4315305A (en) * | 1979-09-12 | 1982-02-09 | Borg-Warner Corporation | Controlled D-C power supply |
US4465966A (en) * | 1982-04-06 | 1984-08-14 | Motorola, Inc. | Controlled ferroresonant voltage regulator providing immunity from sustained oscillations |
JPH04299070A (ja) | 1991-03-26 | 1992-10-22 | Hitachi Ltd | スイッチングレギュレータ |
US6147478A (en) | 1999-09-17 | 2000-11-14 | Texas Instruments Incorporated | Hysteretic regulator and control method having switching frequency independent from output filter |
US6563718B1 (en) * | 2001-12-06 | 2003-05-13 | Koninklijke Philips Electronics N.V. | Capacitively coupled power converter |
-
2009
- 2009-09-07 WO PCT/JP2009/065574 patent/WO2010092704A1/ja active Application Filing
- 2009-09-07 DE DE112009004383T patent/DE112009004383T5/de not_active Withdrawn
- 2009-09-07 JP JP2010513544A patent/JP5170241B2/ja not_active Expired - Fee Related
-
2011
- 2011-08-09 US US13/205,657 patent/US8335092B2/en active Active
Patent Citations (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2002333653A (ja) * | 2001-05-08 | 2002-11-22 | Canon Inc | 発光装置及びカメラ |
JP2006006057A (ja) * | 2004-06-18 | 2006-01-05 | Murata Mfg Co Ltd | フォワードコンバータ |
WO2007018227A1 (ja) * | 2005-08-11 | 2007-02-15 | Murata Manufacturing Co., Ltd. | 絶縁型スイッチング電源装置 |
WO2007069403A1 (ja) * | 2005-12-16 | 2007-06-21 | Murata Manufacturing Co., Ltd. | 複合トランスおよび絶縁型スイッチング電源装置 |
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DE102011100644A1 (de) * | 2011-05-05 | 2012-11-08 | Minebea Co., Ltd. | Gleichspannungswandler |
WO2018179694A1 (ja) * | 2017-03-29 | 2018-10-04 | Fdk株式会社 | 絶縁型スイッチング電源 |
US11038429B2 (en) | 2017-03-29 | 2021-06-15 | Fdk Corporation | Insulation-type switching power supply |
CN108933515A (zh) * | 2017-05-19 | 2018-12-04 | 英飞凌科技奥地利有限公司 | 反激式转换器控制器、反激式转换器及其操作方法 |
CN108933515B (zh) * | 2017-05-19 | 2023-08-18 | 英飞凌科技奥地利有限公司 | 反激式转换器控制器、反激式转换器及其操作方法 |
Also Published As
Publication number | Publication date |
---|---|
US8335092B2 (en) | 2012-12-18 |
US20110292691A1 (en) | 2011-12-01 |
JPWO2010092704A1 (ja) | 2012-08-16 |
DE112009004383T5 (de) | 2012-08-09 |
JP5170241B2 (ja) | 2013-03-27 |
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