WO2007108115A1 - 双方向昇降圧dcdcコンバータ装置 - Google Patents
双方向昇降圧dcdcコンバータ装置 Download PDFInfo
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- WO2007108115A1 WO2007108115A1 PCT/JP2006/305734 JP2006305734W WO2007108115A1 WO 2007108115 A1 WO2007108115 A1 WO 2007108115A1 JP 2006305734 W JP2006305734 W JP 2006305734W WO 2007108115 A1 WO2007108115 A1 WO 2007108115A1
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Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1582—Buck-boost converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/125—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means
- H02M3/135—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only
- H02M3/137—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/139—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1588—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
- H02M3/33584—Bidirectional converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0003—Details of control, feedback or regulation circuits
- H02M1/0012—Control circuits using digital or numerical techniques
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Definitions
- the present invention relates to a DCDC converter device used when connecting DC voltage sources to each other, and can be applied to, for example, an electric vehicle equipped with a power storage device.
- a step-down converter capable of bidirectional power control (hereinafter referred to as a bidirectional step-down DCDC converter) is connected to the DC overhead line and power. Used for connection to storage devices (for example, Patent Document 1).
- Patent Document 1 JP 2005-206111 A
- Patent Document 2 discloses a circuit configuration.
- Patent Document 2 Japanese Patent Laid-Open No. 2001-268900
- the bidirectional buck-boost DCDC converter disclosed in Patent Document 2 raises the primary side voltage higher than the secondary side voltage in the power flow from the primary side to the secondary side. If the primary side voltage is lower than the secondary side voltage, the power flow from the secondary side to the primary side.
- the operation pattern of the switching elements in the four operation modes is determined for each operation mode, when the primary side voltage is increased above the secondary side voltage and when the primary side voltage is decreased below the secondary side voltage. For example, it is not assumed that the primary side voltage and the secondary side voltage are the same, or the case where the power flow is zero. Therefore, it is not possible to continuously transition between the operation modes.
- the present invention has been made to solve such a problem, and in the state where different DC voltage sources are respectively connected to the primary side and the secondary side of the DCDC converter, the secondary side voltage and the primary side are connected. Regardless of the magnitude of the side voltage, the power flow in both directions is possible from the primary side to the secondary side and from the secondary side to the primary side.
- Bidirectional buck-boost DCDC converter device that can automatically control the value
- a bidirectional buck-boost DCDC converter device includes:
- a primary side converter connected to the input / output terminal of the primary side power supply for performing power conversion operation, a secondary side converter connected to the input / output terminal of the secondary side power supply for performing power conversion operation, and a primary side converter And a coupling unit for connecting the secondary side conversion unit,
- the detection values of the primary side conversion unit, the secondary side conversion unit, and the coupling unit force are input, and the primary side conversion unit or the secondary side conversion is made so that the detected value matches the given command value.
- the primary side voltage and the secondary side voltage are independent of the magnitude relationship. Enables bidirectional power flow from side to secondary side and from secondary side to primary side. It is possible to obtain a bidirectional buck-boost DCDC converter that can automatically control the direction and magnitude of power to the desired value continuously on an instantaneous value basis.
- FIG. 1 is a configuration diagram of a bidirectional buck-boost DCDC converter device in a first embodiment.
- FIG. 2 is a diagram showing a configuration example of a control unit 30a in the first embodiment.
- FIG. 3 is a diagram showing a configuration example of a current command conversion unit 31a in the first embodiment.
- FIG. 4 is a diagram showing a configuration example of a current command adjustment unit 32a in the first embodiment.
- FIG. 5 is a diagram showing a configuration example of a current control unit 33a in the first embodiment.
- FIG. 6 is a diagram showing a configuration example of a conduction rate command generation unit 34a in the first embodiment.
- FIG. 7 is a diagram illustrating a configuration example of a gate signal generation unit 35a in the first embodiment.
- FIG. 8 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 9 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 10 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 11 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 12 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 13 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 14 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 15 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the first embodiment.
- FIG. 16 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the second embodiment.
- FIG. 17 is a diagram illustrating a configuration example of a control unit 30b in the second embodiment.
- FIG. 18 is a diagram illustrating a configuration example of a current command adjustment unit 32b in the second embodiment.
- FIG. 19 is a diagram showing a configuration example of a primary side capacitor voltage upper limit operation amount calculation unit 60 in the second embodiment.
- FIG. 20 is a diagram showing a configuration example of a primary side capacitor voltage lower limit operation amount calculation unit 61 in the second embodiment.
- FIG. 21 is a diagram showing a configuration example of a secondary side capacitor voltage upper limit operation amount calculation unit 62 in the second embodiment.
- FIG. 22 is a diagram showing a configuration example of a secondary side capacitor voltage lower limit operation amount calculation unit 63 in the second embodiment.
- FIG. 23 is a diagram showing a configuration example of a primary side switching circuit current upper limit operation amount calculation unit 66 in the second embodiment.
- FIG. 24 is a diagram showing a configuration example of a primary side switching circuit current lower limit operation amount calculation unit 67 in the second embodiment.
- FIG. 25 is a diagram showing a configuration example of a secondary side switching circuit current upper limit operation amount calculation unit 68 in the second embodiment.
- FIG. 26 is a diagram showing a configuration example of a secondary side switching circuit current lower limit operation amount calculation unit 69 in the second embodiment.
- FIG. 27 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the third embodiment.
- FIG. 28 is a diagram illustrating a configuration example of a control unit 30c in the third embodiment.
- FIG. 29 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the fourth embodiment.
- FIG. 30 is a diagram illustrating a configuration example of a control unit 30d in the fourth embodiment.
- FIG. 31 is a diagram showing a configuration example of a current command conversion unit 31b in the fourth embodiment.
- FIG. 32 is a configuration diagram of the bidirectional buck-boost DCDC converter device in the fifth embodiment.
- FIG. 33 is a diagram illustrating a configuration example of a control unit 30e in the fifth embodiment.
- FIG. 34 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the sixth embodiment.
- FIG. 35 is a diagram showing a configuration example of a control unit 30f in the sixth embodiment.
- FIG. 36 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the seventh embodiment.
- FIG. 37 is a diagram showing a configuration example of a control unit 30g in the seventh embodiment.
- FIG. 38 is a diagram showing a configuration example of a conduction rate command generation unit 34b in the seventh embodiment.
- FIG. 39 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the seventh embodiment.
- FIG. 40 is a diagram showing a simulation result of operation waveforms of the bidirectional buck-boost DCDC converter device in the seventh embodiment.
- FIG. 41 is a diagram showing a simulation result of operation waveforms of the bidirectional buck-boost DCDC converter device in the seventh embodiment.
- FIG. 42 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the seventh embodiment.
- FIG. 43 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the eighth embodiment.
- FIG. 44 is a diagram showing a configuration example of a control unit 30h in the eighth embodiment.
- FIG. 45 is a diagram showing a configuration example of a conduction ratio command generation unit 34c in the eighth embodiment.
- FIG. 46 is a diagram showing a configuration example of a gate signal generation unit 35b in the eighth embodiment.
- FIG. 47 is a diagram showing a simulation result of an operation waveform of the bidirectional buck-boost DCDC converter device in the eighth embodiment.
- FIG. 48 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the eighth embodiment.
- FIG. 49 is a diagram showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device in the eighth embodiment.
- FIG. 50 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the ninth embodiment.
- FIG. 51 shows a configuration example of a control unit 30i in the ninth embodiment.
- FIG. 52 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the tenth embodiment.
- FIG. 53 shows a configuration example of a control unit 30j in the tenth embodiment.
- FIG. 54 is a diagram showing a configuration example of a conduction rate command generation unit 34d in the tenth embodiment.
- FIG. 55 is a diagram illustrating a configuration example of a gate signal generation unit 35c in the tenth embodiment.
- FIG. 56 is a configuration diagram of a bidirectional buck-boost DCDC converter device in an eleventh embodiment.
- FIG. 57 shows a configuration example of a control unit 30k in the eleventh embodiment.
- FIG. 58 is a configuration diagram of a bidirectional buck-boost DCDC converter device according to a twelfth embodiment.
- FIG. 59 is a diagram showing a configuration example of a control unit 30m in the twelfth embodiment.
- FIG. 60 is a diagram showing a configuration example of a current command conversion unit 31c in the twelfth embodiment.
- FIG. 61 is a diagram showing a configuration example of a current command adjustment unit 32c in the twelfth embodiment.
- FIG. 62 shows a configuration example of a current control unit 33b in the twelfth embodiment.
- FIG. 63 is a configuration diagram of a bidirectional buck-boost DCDC converter device in the thirteenth embodiment.
- FIG. 64 shows a configuration example of a control unit 30 ⁇ in the thirteenth embodiment.
- FIG. 65 is a diagram showing a configuration example of a current command conversion unit 31d in the thirteenth embodiment.
- FIG. 66 shows a configuration example of a current command adjustment unit 32d in the thirteenth embodiment.
- FIG. 67 shows a configuration example of a current control unit 33c in the thirteenth embodiment.
- FIG. 68 is a diagram showing an application example of the bidirectional buck-boost DCDC converter device in the fourteenth embodiment.
- FIG. 69 is a diagram showing an application example of the bidirectional buck-boost DCDC converter device in the fifteenth embodiment.
- FIG. 1 is a diagram showing a configuration of a bidirectional buck-boost DCDC converter device according to Embodiment 1 of the present invention.
- the primary side conversion unit la is connected to the input / output terminals 23a and 24a of the primary side power source 2a composed of the primary side power source impedance 21a and the primary side power source voltage source 22a.
- the secondary side power supply impedance 21b and the secondary side power supply voltage source 22b are connected to the input / output terminals 23b and 24b of the secondary side power supply 2b through the coupling part lc. It is connected to the side converter lb.
- the primary side conversion unit la includes a primary side switching circuit 10a in which switching elements lla and 12a are connected in series, a primary side capacitor 13a connected in parallel to the primary side switching circuit 10a, and a primary side
- the voltage detector 14a detects the voltage of the side capacitor 13a.
- the positive terminal of the switching element 1la on the upper arm side of the primary side switching circuit 10a is the first terminal 15a
- the negative terminal of the switching element 12a on the lower arm side of the primary side switching circuit 10a is the second terminal
- the first terminal 15a is connected to the positive side of the primary side capacitor 13a
- the second terminal 16a is connected to the negative side of the primary side capacitor 13a.
- the fourth terminal 18a is connected to the fourth terminal 18b in the similarly configured secondary side switching circuit 10b via the connection line 5, and the negative side and the lower side of the switching element 11a on the upper arm side
- the connection point on the positive side of the switching element 12a on the arm side is connected to the third terminal 17a and the third terminal 17b in the secondary-side switching circuit 10b having the same configuration by the coupling rear tuttle 3, and this coupling rear tuttle 3
- a first current detector 4 for detecting the current IL is provided.
- connection line 5 a voltage between the connection line 5 and an arbitrary position between the third terminal 17a of the primary side switching circuit 10a and the third terminal 17b of the secondary side switching circuit 10b is coupled.
- a voltage detector 6 is provided for detecting the coupling voltage VL as a partial voltage VL.
- the voltage detector 6 detects the voltage between the coupling reactor 3 and the connection line 5.
- the connection voltage VL may be, for example, the voltage between the third terminal 17a of the primary side switching circuit 10a and the connection line 5.
- it may be a voltage between the third terminal 17b of the secondary side switching circuit 10b and the connection line 5.
- the primary side capacitor voltage VI output from the primary side conversion unit la, the secondary side capacitor voltage V2 output from the secondary side conversion unit 1b, and the combined reactor power output from the coupling unit lc. IL and coupling unit voltage VL are input to control unit 30a.
- the control unit 30a performs on / off control of the switching elements lla, llb, 12a, and 12b so that the power PL flowing through the coupling unit lc from the primary side to the secondary side matches the given command value P *.
- the power command P * is input from another control device positioned higher than the control unit 30a of the DCDC converter that controls the power storage system including the DCDC converter device of the present invention, for example. It corresponds to a signal.
- the current of the first terminal 15a and the current of the second terminal 16a, the current of the coupling reactor 3 and the current of the connection line 5 in the primary side switching circuit 10a, the second side switching circuit 10b The current of the first terminal 15b and the current of the second terminal 16b are equal in magnitude and are simply in opposite directions.
- all the explanations in this specification are based on the current of the first terminal 15a in the primary side switching circuit 10a (hereinafter referred to as the primary side switching circuit current II), the current of the coupled reactor 3 ( The following description assumes that the current at the first terminal 15b of the secondary switching circuit (hereinafter referred to as secondary switching circuit current 12) is detected. .
- the negative side of the circuit (from the primary side input / output terminal 24a to the second terminal 16a of the primary side switching circuit 10a, the fourth terminal 18a, the connection line 5, the fourth terminal 18b of the secondary side switching circuit 10b, When the second terminal 16b and the line that leads to the secondary input / output terminal 24b) are grounded, the first terminal 15a and secondary of the primary side switching circuit 10a that is always at a high potential and fluctuates.
- -Side switching circuit 10b, first terminal 15b, secondary terminal 16a of primary-side switching circuit 10a which is at ground potential from the ground potential of coupled rear tuttle 3, secondary-side switching
- the ground potential of the second terminal 16b and connecting line 5 of the touching circuit 10b is lower and stable.
- the insulation withstand voltage required for the current detector may be lower, and the detection value may be obtained with less noise.
- FIG. 2 is a diagram showing a configuration example of the control unit 30a in the first embodiment of the present invention.
- the control unit 30a includes a current command conversion unit 3la, a current command adjustment unit 32a, a current control unit 33a, a conduction rate command generation unit 34a, and a gate signal generation unit 35a.
- the current command conversion unit 31a generates a coupled rear tuttle basic current command ILO * from the power command P * and the coupled unit voltage VL.
- the current command adjustment unit 32a adjusts the coupled rear tuttle basic current command ILO * input from the current command conversion unit 31a to generate a coupled rear tuttle current command IL *.
- the primary duty ratio command VREF is calculated from the current error DIL input from the current controller 33a, the primary side capacitor voltage VI, and the secondary side capacitor voltage V2.
- the switching elements l la, 12a, l ib are obtained from the primary side conduction rate command VREF1 and the secondary side conduction rate command VREF2 input from the conduction rate command generation unit 34a.
- the control unit 30a is configured to input the power command P * from the outside.
- the combined reactor main current command ILO * or the combined reactor current is used instead of the power command P *.
- a configuration may be adopted in which a signal corresponding to the command IL * is externally input to the control unit 30a.
- the current command conversion unit 31a and the current command adjustment unit 32a can be omitted.
- FIG. 3 is a diagram showing a configuration example of the current command conversion unit 31a in the first embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the divider 40 to remove unnecessary frequency components.
- the current command conversion unit 31a divides the given power command P * by the coupling unit voltage VL using the divider 40, thereby obtaining the coupled rear tuttle basic current command ILO *. Generate.
- FIG. 4 is a diagram showing a configuration example of the current command adjustment unit 32a in the first embodiment of the present invention. Although not shown, it may be configured to remove unnecessary frequency components by inserting a low-pass filter or the like at the input / output of limiter 70a.
- the current command upper limit value ILMTH and the current command are compared with the combined rear tuttle basic current command ILO * generated by the current command conversion unit 3la.
- Lower limit value The upper and lower limits are limited by the limiter 70a with the upper and lower limits set by ILMTL.
- the function of the limiter 70a will be described.
- the upper and lower limits of the combined rear tuttle basic current command ILO * to the combined rear tuttle current command IL * the upper and lower limits of the actual combined rear tuttle current IL that is controlled to match this are limited. It becomes possible. Since this coupled rear tutor current IL is a current that always flows through one of the switching elements 11a to 12b, by limiting the upper and lower limits of the coupled reactor power current IL, the current of the switching elements 1la to 12b can be reduced. Can be limited.
- the current command upper limit value ILMTH and the current command lower limit value ILMTL of the limiter 70a are appropriately set to be equal to or smaller than the current withstand capability of the switching elements 11a to 12b.
- FIG. 5 is a diagram showing a configuration example of the current control unit 33a in the first embodiment of the present invention. Although not shown in the figure, a configuration that removes unnecessary frequency components by inserting a low-pass filter or the like at the input / output of the functional block such as the subtractor 200 is also possible.
- the subtractor 200 in the current control unit 33a, the subtractor 200 generates a deviation between the combined rear tutor current command IL * generated by the current command adjusting unit 32a and the combined rear tuttle current IL, and proportionally integrates this. Input to the controller 201.
- the proportional-integral controller 201 obtains the current error DIL by the following equation.
- FIG. 6 is a diagram showing a configuration example of the conduction ratio command generation unit 34a in the first embodiment of the present invention. Although not shown in the figure, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the adder 21 la to remove unnecessary frequency components.
- the divider 210a divides the secondary capacitor voltage V2 by the primary capacitor voltage VI to obtain the secondary capacitor voltage V2 and the primary side voltage. Capacitor voltage VI ratio V2ZV1 is obtained. The value obtained by limiting the lower limit to zero and the upper limit to 1 by the limiter 213a is used as the primary side basic conduction rate command VREF1 A to the primary side conversion unit la.
- a value obtained by adding the current error DIL generated by the current control unit 33a to the primary side basic conduction rate command VREF1A by the adder 211a is the primary side conduction rate command of the primary side conversion unit la.
- the flow rate command is VREF1.
- the ratio V1ZV2 of the primary side capacitor voltage VI and the secondary side capacitor voltage V2 is obtained.
- the value obtained by limiting the lower limit to zero and the upper limit to 1 by the limiter 213b is set as the secondary side basic conduction ratio command VREF2A to the secondary side conversion unit lb.
- the secondary side conversion is performed by adding the DIL2 obtained by adding the sign DIL2 obtained by inverting the sign of the current error DIL generated by the current control unit 33a to the sign inverting circuit 212 to the adder 211b to the secondary side basic conduction ratio command VREF2A.
- FIG. 7 is a diagram showing a configuration example of the gate signal generation unit 35a in the first embodiment of the present invention. Although not shown, a configuration may be adopted in which unnecessary frequency components are removed by inserting a low-pass filter or the like at the input / output of the functional block such as the comparator 220a.
- a carrier signal CAR having a value of 1 is generated.
- the carrier signal CAR for example, a triangular wave, a sawtooth wave or the like is appropriate.
- the control unit 30a By configuring the control unit 30a as described above, when the power command P * is positive regardless of the magnitude and magnitude relationship of the primary-side capacitor voltage VI and the secondary-side capacitor voltage V2, the coupled rear Tuttle basic current command ILO * is positive, and power PL flowing through coupling part lc (hereinafter referred to as coupling part power PL) can flow from primary power supply 2a to secondary power supply 2b. Therefore, its size matches the size of the power command P *.
- the coupled rear tutor current basic command IL0 * is negative, and the joint power PL can flow from the secondary power source 2b to the primary power source 2a. This corresponds to the size of the power command P *.
- the coupling unit power PL is continuously instantaneously based on the arbitrary magnitude' direction. It is possible to control with.
- the current of the switching elements lla to 12b can be limited to an arbitrary value.
- the current of the switching elements lla to 12b can be limited to within the current withstand capability. It is possible to avoid the elements lla to 12b from being damaged by an overcurrent, and to obtain a bidirectional buck-boost DCDC converter device that is strong against disturbance such as an excessive power command input.
- the losses in the primary side conversion unit la, the coupling unit lc, and the secondary side conversion unit lb are very small, they are ignored, and fluctuations in the energy stored in the primary side capacitor 13a and the secondary side capacitor 13b are changed. Therefore, if this is ignored, the input / output power P10 of the primary power supply 2a, the coupling power PL, and the input / output power P20 of the secondary power supply 2b are equal on an instantaneous value basis.
- the coupling power PL By controlling the coupling power PL, the power flow between the primary power source 2a and the secondary power source 2b can be controlled.
- the power command P * or the coupled rear tutor current command IL is not shown.
- * By setting * to a value that includes the loss (usually a few percent of the total power input to and output from the DCDC converter), the power flow control accuracy can be further improved.
- the primary side capacitor 13a and the secondary side capacitor 13b are not shown.
- the power command P * or the combined rear tutor current command IL * according to the amount of energy fluctuation accumulated in the The passing control accuracy can be improved.
- FIGS. 8 to 11 and FIGS. 12 to 15 are diagrams showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter device to which the configuration of the control unit 30a is applied in the first embodiment of the present invention. is there.
- Figures 8 (a) and 12 (a) are diagrams showing the primary terminal voltage V10 and the secondary terminal voltage V20.
- Figures 8 (b) and 12 (b) show the primary-side conduction ratio command VREF1.
- FIGS. 9 (c) and 13 (c) are diagrams illustrating the secondary-side conduction ratio command VREF2
- FIGS. 9 (d) and 13 (d) are diagrams illustrating the combined rear tutor current command IL *.
- FIGS. 10 (e) and 14 (e) are diagrams showing the coupled rear tuttle current IL
- FIGS. 10 (f) and 14 (f) are diagrams showing the power command P *.
- FIG. 11 (g) and FIG. 15 (g) are diagrams showing the coupling portion power PL.
- the voltage source that changes the ramp of the primary terminal voltage V10 between 400V and 800V at 2Hz is connected as the primary power supply 2a, and a large-capacitance capacitor with an initial voltage of 600V is connected to the secondary power supply 2b.
- the primary side conduction ratio command VREF1 and the secondary side conduction ratio command VREF2 are optimally adjusted regardless of the magnitude relationship between the primary side terminal voltage V10 and the secondary side terminal voltage V20.
- the combined reactor current IL coincides with the combined reactor current command IL *. Since the coupled rear tutor current command IL * is less than ⁇ 1000A, it operates without being limited by the limiter 70a. As a result, it can be seen that the coupling power PL matches the power command P * in all regions.
- a voltage source that changes the primary side terminal voltage V10 between 400V and 800V at 2Hz is connected as the primary side power supply 2a, and a large capacity capacitor with an initial voltage of 600V is secondary. It is a figure showing the operation waveform when connected as the side power supply 2b and the power command P * is ramp-changed in the range of ⁇ 500KW at 1Hz. Note that the limiter 70a is set to ⁇ 500A to limit the coupled rear tutor current command IL * to ⁇ 500A.
- the primary side conduction rate command VREF1 and the secondary side conduction rate command VREF2 are optimally adjusted regardless of the magnitude relationship between the primary side terminal voltage V10 and the secondary side terminal voltage V20.
- Join The rear tuttle current IL matches the combined rear tuttle current command IL *, and its value is limited to ⁇ 500A or less.
- the coupled power PL matches the power command P *, and while the coupled rear tutor current IL is limited to ⁇ 500A, the coupled rear tutor current IL is insufficient. It is obvious that the coupling power PL is smaller than the power command P *.
- FIG. 16 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device in the second embodiment of the present invention.
- it is characterized by having a current detector 7a for detecting the primary-side switching circuit current II and a current detector 7b for detecting the secondary-side switching circuit current 12.
- the configuration of the part 30b has the following characteristics.
- FIG. 17 is a diagram showing a configuration example of the control unit 30b in the second embodiment of the present invention.
- the current command adjustment unit 32b is configured to receive the primary side switching circuit current II, the secondary side switching circuit 12, the primary side capacitor voltage VI, and the secondary side capacitor voltage V2, and has the following characteristics. .
- FIG. 18 is a diagram showing a configuration example of the current command adjustment unit 32b in the second embodiment of the present invention.
- the primary side capacitor voltage VI, the secondary side capacitor voltage V2, and the combined rear tutor current basic command ILO * input from the current command conversion unit 31a Primary side switching circuit current II, secondary side switching circuit current 12, primary side capacitor voltage upper limit value VILMTCOMH, and primary side capacitor voltage Lower limit value V1LMTCOML, Secondary capacitor voltage upper limit value V2LMTCOM H, Secondary capacitor voltage lower limit value V2LMTCOML, Primary switching circuit current upper limit value I1LMTCOMH, Primary switching circuit current lower limit value I1L MTCOML, secondary switching circuit current upper limit value I2LMTCOMH, secondary switching circuit current lower limit value I2LMTCOML, current command upper limit value THLMTH for temperature protection, current command lower limit value THLMTL for temperature protection, Using the current command upper limit value ILMTH and the current command lower limit value ILMTL,
- Primary capacitor voltage upper limit limit manipulated variable calculation unit 60 Calculated by primary side capacitor voltage upper limit limit manipulated variable V1LMTH, Primary side capacitor voltage lower limit limit manipulated variable calculation unit 61 calculated by primary side capacitor voltage lower limit limit manipulated variable V1LMTL
- the secondary side capacitor voltage upper limit limit manipulated variable calculator 62 calculates the secondary side capacitor voltage upper limit limit manipulated variable V 2LMTH, the secondary side capacitor voltage lower limit limit manipulated variable calculator 63 calculates the secondary side voltage limit Capacitor voltage lower limit operation amount V2LMTL,
- Primary switching circuit current upper limit operation amount calculation unit 66 Calculated by primary side switching circuit current upper limit operation amount I1LMTH, primary side switching circuit current lower limit operation amount primary operation switching circuit current lower limit calculated by operation amount calculation unit 67 Limiting operation amount I1LM TL, secondary-side switching circuit current upper limit operation amount calculation unit 68 Calculated by secondary-side switching circuit current upper limit operation amount I2LMTH, secondary-side switching circuit current lower limit operation amount calculation unit 69
- the calculated secondary side switching circuit current lower limit limit 12 LMTL is added by adders 59a to 59j, and after correction, temperature protection current command upper limit limit limit THLMTH, temperature protection current command lower limit limit limit THLMTL Limiter 71, current command upper limit limit value ILMTH, current command lower limit limit value ILMTL, limiter 70b. Is generated.
- the limiter 71 performs current command limitation for the purpose of over-temperature protection.
- the limiter 71 can detect the temperatures of the primary side power source 2a, the secondary side power source 2b, the switching elements lla to 12b, and the coupling rear tutor 3.
- the temperature protection current command upper limit limit THLMTH and the temperature protection current command lower limit limit THLMTL are determined according to the detection value of the temperature sensor (not shown).
- primary side capacitor voltage upper limit operation amount calculation unit 60 primary side capacitor voltage lower limit operation amount calculation unit 61, secondary side capacitor voltage upper limit operation amount calculation unit 62, secondary capacitor voltage lower limit operation limit Operation amount calculation unit 63, primary-side switching circuit current upper limit operation amount calculation unit 66, primary-side switching circuit current lower limit operation amount calculation unit 67, secondary-side switching circuit current upper limit operation amount calculation unit 68, secondary-side switching
- a configuration example of the circuit current lower limit operation amount calculation unit 69 will be described.
- FIG. 19 is a diagram illustrating a configuration example of the primary-side capacitor voltage upper limit operation amount calculation unit 60 in the second embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as subtractor 80 to remove unnecessary frequency components.
- a subtractor 80 subtracts the primary side capacitor voltage upper limit value V1LMTC OMH from the primary side capacitor voltage VI to obtain a deviation. This is amplified by the proportional-integral controller 81, and the value via the negative limiter 82 that cuts the negative side is output as the primary side capacitor voltage upper limit operation amount V1LMTH.
- the primary side capacitor voltage upper limit operation amount V1LMTH is output according to the deviation.
- the coupling power PL is increased, the primary capacitor voltage VI is suppressed from rising, and the primary capacitor voltage V1 is set to the primary capacitor voltage upper limit VI near LMTCOMH. Can be maintained.
- FIG. 20 is a diagram illustrating a configuration example of the primary-side capacitor voltage lower limit operation amount calculation unit 61 according to the second embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 90 to remove unnecessary frequency components.
- the subtractor At 90 subtract the primary capacitor voltage lower limit value V1LMTC OML from the primary capacitor voltage VI and take the deviation. This is amplified by the proportional-integral controller 91, and the value via the positive limiter 92 that cuts the positive side is output as the primary side capacitor voltage lower limit operation amount V1LMTL.
- the primary side capacitor voltage lower limit operation amount V1LMTL is output according to the deviation.
- the coupling power PL is decreased, the primary capacitor voltage VI is suppressed from decreasing, and the primary capacitor voltage VI is maintained near the primary capacitor voltage lower limit V1LMTCOML. It becomes possible to do.
- FIG. 21 is a diagram showing a configuration example of the secondary side capacitor voltage upper limit operation amount calculation unit 62 in the second embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted at the input / output of a functional block such as the subtractor 100 to remove unnecessary frequency components.
- the subtractor 100 subtracts the secondary side capacitor voltage V2 from the secondary side capacitor voltage upper limit value V2LMTCOMH, and calculates the deviation. take. This is amplified by the proportional integration controller 101, and the value via the positive limiter 102 that cuts the positive side is output as the secondary side capacitor voltage upper limit operation amount V2LMTH.
- the secondary side capacitor voltage upper limit operation amount V2LMTH is reduced according to the deviation.
- the coupling power PL is decreased, the rise of the secondary capacitor voltage V2 is suppressed, and the secondary capacitor voltage V2 is set to the upper limit value of the secondary capacitor voltage. It can be maintained near V2LMTCOMH.
- FIG. 22 is a diagram illustrating a configuration example of the secondary-side capacitor voltage lower limit operation amount calculation unit 63 in the second embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted at the input / output of a functional block such as the subtractor 110 to remove unnecessary frequency components. Yes.
- the subtractor 110 subtracts the secondary side capacitor voltage V2 from the secondary side capacitor voltage lower limit value V2LMTCOML, and calculates the deviation. take. This is amplified by the proportional integration controller 111, and the value via the negative limiter 112 that cuts the negative side is output as the secondary capacitor voltage lower limit operation amount V2LMTL.
- the secondary side capacitor voltage lower limit operation amount V2LMTL is set according to the deviation. Is output and the combined rear tutor current command IL * is increased to increase the coupling power PL, suppress the decrease of the secondary capacitor voltage V2, and limit the secondary capacitor voltage V2 to the secondary capacitor voltage lower limit. It can be maintained near the value V2LMTCOML.
- FIG. 23 is a diagram showing a configuration example of the primary-side switching circuit current upper limit operation amount calculation unit 66 in the second embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 130 to remove unnecessary frequency components.
- the subtractor 130 also subtracts the primary side switching circuit current II from the primary side switching circuit current upper limit value I1LMTCOMH force. take. This is amplified by the proportional-plus-integral controller 131, and the value via the positive-side limiter 132 that cuts the positive side is output as the primary-side switching circuit current upper limit operation amount I1LMTH.
- the primary side switching circuit current upper limit limiting operation is performed according to the deviation.
- the amount I1LMTH is output, and the coupled rear tutor current command IL * is decreased, thereby reducing the coupling power PL and suppressing the increase of the primary side switching circuit current I1 and the primary side switching circuit current II to the primary side switching circuit. It is possible to maintain the current limit value near I1LMTCOMH.
- FIG. 24 is a diagram showing a primary side switching circuit current lower limit limiting operation in Embodiment 2 of the present invention.
- FIG. 6 is a diagram illustrating a configuration example of a crop calculation unit 67. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 140 to remove unnecessary frequency components.
- the subtractor 140 subtracts the primary side switching circuit current II from the primary side switching circuit current lower limit value I1LMTCOML to obtain the deviation. take. This is amplified by the proportional-integral controller 141, and the value via the negative-side limiter 142 that cuts the negative side is output as the primary-side switching circuit current lower limit operation amount I1LMTL.
- FIG. 25 is a diagram showing a configuration example of the secondary side switching circuit current upper limit operation amount calculation unit 68 in the second embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 150 to remove unnecessary frequency components.
- the secondary side switching circuit current upper limit operation amount calculation unit 68 subtracts the secondary side switching circuit current 12 from the secondary side switching circuit current upper limit value I2LMTCOMH by the subtractor 150. Take the deviation. This is amplified by the proportional-plus-integral controller 151, and the value via the positive-side limiter 152 that cuts the positive side is output as the secondary-side switching circuit current upper limit operation amount I2LMTH.
- the secondary side switching circuit current upper limit is set according to the deviation.
- the limit operation amount I2LMTH is output, and the combined rear tutor current command IL * is decreased, so that the coupling power PL is decreased, the rise of the secondary side switching circuit current I2 is suppressed, and the secondary side switching circuit current 12 is reduced.
- the secondary side switching circuit current It becomes possible to maintain the limit value near I2LMTCOMH.
- FIG. 26 is a diagram illustrating a configuration example of the secondary-side switching circuit current lower limit operation amount calculation unit 69 in the second embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtracter 160 to remove unnecessary frequency components.
- the secondary switching circuit current lower limit operation amount calculation unit 69 subtracts the secondary switching circuit current 12 from the secondary switching circuit current lower limit value I2LMTCOML by the subtracter 160. Take the deviation. This is amplified by the proportional-integral controller 161, and the value via the negative limiter 162 that cuts the negative side is output as the secondary-side switching circuit current lower limit operation amount I2LMTL.
- the secondary side switching circuit current lower limit is set according to the deviation.
- the limit manipulated variable I2LMTL is output, and the coupled rear tutor current command IL * is increased, thereby increasing the coupling power PL, suppressing the decrease in the secondary switching circuit current 12 and reducing the secondary switching circuit current 12 to It becomes possible to maintain the switching current limit lower limit value near the I2LMTCOML.
- the primary side power source 2a, the secondary side power source 2b, the primary side conversion unit la, the secondary side conversion unit lb, and the coupling rear tutor 3 are overvoltage and overcurrent. It becomes possible to protect from over temperature.
- FIG. 27 is a diagram showing a configuration of a bidirectional buck-boost DCDC converter device according to Embodiment 3 of the present invention.
- the feature is that the current detector 7a for detecting the primary side switching circuit current II and the current detector 7b for detecting the secondary side switching circuit current 12 provided in the configuration in the second embodiment are omitted.
- the controller 30c has the following characteristics is there.
- FIG. 28 is a diagram showing a configuration example of the control unit 30c in the third embodiment of the present invention.
- the primary side switching circuit current I 1 and the secondary side switching circuit current 12 that are input to the current command adjustment unit 32b are combined with the combined rear tutor current IL, the coupling unit voltage VL, and the primary side capacitor voltage.
- the feature is that it is calculated from VI and the secondary capacitor voltage V2.
- the power passing through the first terminal 15a and the second terminal 16a of the primary side switching circuit 10a (hereinafter referred to as primary side switching circuit unit power P1) and the coupling unit It can be seen that the power PL is equal on an instantaneous value basis if the losses in the primary side conversion unit la and the coupling unit lc are ignored.
- a multiplier 37a uses this, as shown in FIG. 28, in the control unit 30c, a multiplier 37a generates a product of the coupling reactor current IL and the coupling unit voltage VL, and this is divided by the divider 36a into the primary side capacitor.
- the primary switching circuit current II can be obtained by dividing by the voltage VI.
- the multiplier 37a generates a product of the coupled rear tutor current IL and the coupled portion voltage VL, and the divider 36b divides the product by the secondary capacitor voltage V2, whereby the secondary switching is performed.
- the circuit current 12 is obtained as follows.
- the primary side switching circuit current II and the secondary side switching circuit current 12 are directly detected as a current detector. Control without using these values is possible without detection in 7a and 7b, and it is possible to configure a more sophisticated control unit without increasing the number of parts, size, and mass of the DCDC converter device. .
- FIG. 29 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device according to Embodiment 4 of the present invention.
- Control unit 30d has the following characteristics.
- FIG. 30 is a diagram showing a configuration example of the control unit 30d in the fourth embodiment of the present invention.
- the current command conversion unit 31b is characterized in that a primary side capacitor voltage VI and a secondary side capacitor voltage V2 are further input.
- FIG. 31 shows a configuration example of current command conversion unit 31b in the fourth embodiment of the present invention.
- the gain and phase are adjusted by passing the primary-side capacitor voltage VI and the secondary-side capacitor voltage V2 through the bandpass filters 120a and 120b, respectively.
- the primary side capacitor voltage oscillation suppression manipulated variable VI DMP and the secondary side capacitor voltage oscillation suppression manipulated variable V2DMP are obtained.
- This V1DMP is added to the power command P * by the calorimeter 121, V2DMP is subtracted from the power command P * by the subtractor 122, and the result obtained by dividing the result by the coupling voltage VL by the divider 41 is combined.
- the power command P * is adjusted so that the coupling power PL decreases, and when the secondary side capacitor voltage V2 tends to decrease.
- the power command P * is adjusted so that the coupling unit power PL increases.
- the bidirectional buck-boost DCDC converter device can suppress voltage oscillations of primary-side capacitor voltage VI and secondary-side capacitor voltage V2. It becomes possible, and more stable control becomes possible.
- FIG. 32 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device in the fifth embodiment of the present invention.
- the voltage detector 6 that detects the coupling unit voltage VL is omitted, and the control unit 30e has the following features.
- FIG. 33 is a diagram showing a configuration example of the control unit 30e in the fifth embodiment of the present invention. As shown in Fig. 33, the control unit 30e is characterized in that the multiplier 37b takes the product of the primary side capacitor voltage VI and the primary side conduction rate command VREF1 and uses this result as the coupling unit voltage VL. It is.
- voltage detector 6 for detecting coupling unit voltage VL can be omitted, and the device It becomes possible to make the whole smaller and lighter.
- FIG. 34 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device in the sixth embodiment of the present invention.
- the voltage detector 6 that detects the coupling unit voltage VL is omitted, and the control unit 30f has the following features.
- FIG. 35 is a diagram showing a configuration example of the control unit 30f in the sixth embodiment of the present invention.
- control unit 30f uses a multiplier 37c to calculate the product of the primary side capacitor voltage V2 and the secondary side conduction rate command VREF2, and uses this result as the coupling unit voltage VL. It is a feature.
- FIG. 36 shows the structure of the bidirectional buck-boost DCDC converter device according to Embodiment 7 of the present invention.
- the control unit 30g has the following characteristics.
- FIG. 37 is a diagram showing a configuration example of the control unit 30g in the seventh embodiment of the present invention.
- the duty ratio command generator 34b has the following characteristics.
- FIG. 38 is a diagram illustrating a configuration example of the conduction ratio command generation unit 34b in the seventh embodiment of the present invention.
- the value obtained by dividing the primary capacitor voltage VI by the secondary capacitor voltage V2 by the divider 21 Ob is used to limit the upper and lower limits by the limiter 214b, and the value obtained by multiplying the duty factor gain GREF by the multiplier 215b is As secondary side basic flow rate command VREF2A,
- the conductivity gain GREF can take any value from 0 to 1.
- the lower limit of the limiters 214a and 214b is set to zero, and the upper limit is set to 1.
- the primary side basic duty ratio command VREF1A is set to 0.9 for the value in which the upper limit of V2ZV1 is limited to 1.
- the secondary basic flow rate command VREF2A is a value obtained by multiplying the value obtained by limiting the upper limit of V1ZV2 to 1 by 0.9, and the maximum value is 0.9 for both.
- the primary-side conduction ratio command VREF1 and the secondary-side conduction ratio command VREF2 are respectively added to the VREF1A and VREF2A by the current error DIL and the value DIL2 obtained by inverting the sign thereof by the sign inversion circuit 212.
- the forces DIL and DIL2 added by the devices 21 la and 21 lb are small in the steady state, so if they are ignored, the maximum values of the primary-side flow rate command VREF1 and the secondary-side flow rate command VREF2 are , Equal to GREF, 0.9, no more.
- the primary-side conduction voltage command VREF1 and the secondary-side conduction ratio command VREF2 have the maximum primary-side capacitor voltage VI and secondary-side capacitor voltage V2, and so on.
- the maximum value of both the conduction rate command VREF1 and the secondary side conduction rate command VREF2 is 0.9, which is equal to the conduction rate gain GREF, and does not exceed this value.
- the ON / OFF pulse widths of the switching elements lla to 12b are determined by the magnitude relationship between the primary-side conduction ratio command VREF1, the secondary-side conduction ratio command VREF2, and the carrier signal CAR as described above.
- Limiting the maximum value of the primary-side conduction ratio command VREF1 and the secondary-side conduction ratio command VREF2 is equivalent to limiting the minimum value of the on / off pulse width of the switching elements 1 la to 12b. In other words, it means that the minimum pulse width of the switching elements lla to 12b can be arbitrarily limited by the value of the conductivity ratio gain GREF.
- the switching element has a limit on the minimum pulse width that can be accurately turned on / off due to the delay of the on / off operation, and a gate signal having a pulse width narrower than a few seconds to several tens of seconds. Even if it is given, it is difficult to perform the on / off operation as it is, and the pulse width according to the given gate signal cannot be output accurately.
- control performance deteriorates such that the combined rear tuttle current IL causes a combined rear tutor current command IL * force minute error.
- the minimum pulse width of the switching elements lla to 12b can be set to an arbitrary value by the flow rate gain GREF. Therefore, the value of GREF is switched.
- the primary-side conduction rate command VREF1 and the secondary-side conduction rate command VREF2 are maximized. Even when the primary-side capacitor voltage VI and the secondary-side capacitor voltage V2 are equal, it can be avoided that the switching elements 1 la to 12b operate with a narrow pulse width exceeding the limit.
- the switching elements lla to 12b can accurately output the pulse width according to the given gate signal, so that the coupled rear tutor current IL is controlled by the coupled rear tuttle current command IL * force and a slight error. Performance degradation can be avoided.
- FIG. 39 to FIG. 42 are diagrams showing the results of simulating the operation of the bidirectional buck-boost DCDC converter when the configuration of the control unit 30g in the seventh embodiment is applied.
- Fig. 39 (a) is a diagram showing the primary side terminal voltage V10 and the secondary side terminal voltage V20
- Fig. 39 (b) is a diagram showing the primary side conduction rate command VREF1
- Fig. 40 (c) is Secondary side flow rate command VRE
- Fig. 40 (d) is a diagram showing the coupled rear tutor current command IL *
- Fig. 41 (e) is a diagram showing the coupled rear tuttle current IL
- Fig. 41 (f) is a diagram showing the power command P
- FIG. 42 (g) is a diagram showing the coupling unit power PL.
- a voltage source that changes the ramp of the primary side terminal voltage V10 between 400V and 800V at 2Hz is connected as the primary power supply 2a, and a large-capacitance capacitor with an initial voltage of 600V is connected as the secondary power supply 2b.
- This is the operating waveform when P * is changed by ramping at a rate of ⁇ 500KW at 1Hz.
- the limiter 70a is set to ⁇ 1000A, thereby limiting the coupled rear tutor current command IL * to within ⁇ 1000A.
- the primary side conduction ratio command VREF1 and the secondary side conduction ratio command VREF2 are optimally adjusted, and the coupled rear tutor current IL is Matches the command IL *. Since the coupled rear tutor current command IL * is less than ⁇ 1000A, it operates without being limited by the limiter 70a. As a result, it can be seen that the coupling power PL is consistent with the power command P * in all regions.
- the maximum values of the primary-side conduction rate command VREF1 and the secondary-side conduction rate command VREF2 at the point where the primary-side terminal voltage V10 and the secondary-side terminal voltage V20 are equal are set by the conduction factor gain GREF. It can be confirmed that the switching elements 11a to 12b can be prevented from operating with a narrow pulse width exceeding the limit.
- the duty ratio gain GREF may be changed to an arbitrary value at an arbitrary timing during operation.
- the duty factor gain GREF is set to a narrow pulse width where the switching elements 1 la to 12b exceed the limit.
- FIG. 43 is a diagram showing the configuration of the bidirectional step-up / step-down DCDC converter device according to Embodiment 8 of the present invention.
- Control unit 30h has the following characteristics.
- Fig. 44 is a diagram illustrating a configuration example of the control unit 30h in the eighth embodiment of the present invention.
- the signal output from the duty ratio command generator 34c has been changed to VREF, and the duty ratio command generator 34c and the gate signal generator 35b have the following characteristics.
- FIG. 45 is a diagram showing a configuration example of the conduction ratio command generation unit 34c in the eighth embodiment of the present invention.
- adder 232 calculates the sum of primary-side capacitor voltage VI and secondary-side capacitor voltage V2.
- the secondary side capacitor voltage V2 is divided by the sum of the primary side capacitor voltage VI and the secondary side capacitor voltage V2 by the secondary divider 230.
- the ratio V2Z (V1 + V2) of the sensor voltage V2 and the sum of the primary capacitor voltage VI and the secondary capacitor voltage V2 is obtained. This is the basic duty ratio command VREF0 common to the primary and secondary converters la and lb.
- a value obtained by adding the current error DIL to the basic duty ratio command VREF0 by the adder 231 is defined as a duty ratio command VREF common to the primary side and secondary side conversion units la and lb.
- FIG. 46 is a diagram showing a configuration example of the gate signal generation unit 35b in the eighth embodiment of the present invention.
- carrier signal generation section 241 generates carrier signal CAR that takes a value between 0 and 1.
- carrier signal CAR a triangular wave, a sawtooth wave or the like is appropriate.
- the comparator 240 and the inverting circuit 242 turn on / off the gate signals Gla to G2 b of the switching elements lla to 12b according to the following logic, based on the magnitude relationship between the duty ratio command VREF and the carrier signal CAR. To decide.
- Gla is turned off, and gate signal G2a of switching element 12a is turned on.
- the gate signal G2b of the switching element 12b of the secondary conversion unit lb is turned off and the gate signal Gib of the switching element 1 lb is turned on.
- duty ratio command VREF is 0.5
- switching elements 11a and 12a switching element l ib
- the on / off duty ratio of 12b is 50% respectively.
- the on / off duty ratio of switching elements 11a and 12a and switching elements l ib and 12b is around 50%, respectively, according to the ratio. Will increase or decrease.
- the switching elements lla to 12b operate with a narrow pulse width exceeding the limit. Therefore, the switching elements lla to 12b can accurately output the norse width according to the given gate signal, and as a result, the combined rear tutor current IL becomes the combined rear tutor current command IL * force. Can be avoided.
- FIGS. 47 to 49 are diagrams showing simulation results of operation waveforms of the bidirectional buck-boost DCDC converter to which the configuration of the control unit 30h is applied according to Embodiment 8 of the present invention.
- Fig. 47 (a) shows the primary terminal voltage V10 and secondary terminal voltage V20
- Fig. 47 (b) shows the conduction ratio command VREF
- Fig. 48 (c) shows the coupled rear tutor current.
- Fig. 48 (d) is a diagram showing the coupled rear tutor current IL
- Fig. 49 (e) is a diagram showing the power command P *
- Fig. 49 (f) is a diagram showing the coupling IL *. It is a figure which shows electric power PL.
- a voltage source that changes the ramp of the primary side terminal voltage V10 between 400V and 800V at 2Hz is connected as the primary power supply 2a, and a large-capacitance capacitor with an initial voltage of 600V is connected as the secondary power supply 2b.
- the limiter 70a is set to ⁇ 2000A, which limits the coupled rear tutor current command IL * to within ⁇ 2000A.
- the conduction ratio command VREF is optimally adjusted, and the coupled rear tutor current IL matches the coupled rear tuttle current command IL *. Since the coupled rear tutor current command IL * is less than ⁇ 2000A, it operates without being limited by the limiter 70a. As a result, it can be seen that the coupling power PL is consistent with the power command P * in all regions.
- the conduction ratio command VREF is 0.5 when the primary side capacitor voltage VI and the secondary side capacitor voltage V2 are equal, and when the primary side capacitor voltage VI and the secondary side capacitor voltage V2 are different, It can be confirmed that the ratio increases or decreases around 0.5 according to the ratio. As a result, it can be divided that the switching elements 1 la to 12 b can avoid the operation with a narrow pulse width exceeding the limit.
- Embodiment 9 the configuration of the bidirectional buck-boost DCDC converter according to Embodiment 9 of the present invention will be described in detail with reference to the drawings.
- the configuration of the ninth embodiment is based on the configuration of the eighth embodiment. Only the differences from the configuration of the bidirectional buck-boost DCDC converter device according to Embodiment 8 of the present invention will be described below.
- FIG. 50 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device according to the ninth embodiment of the present invention.
- the voltage detector 6 that detects the coupling unit voltage VL is omitted, and the control unit 30i has the following features.
- FIG. 51 shows a configuration example of the control unit 30i in the ninth embodiment of the present invention.
- the product of the value obtained by subtracting the conduction ratio command VREF from 1.0 by the subtractor 39a, which is calculated by the multiplier 37e, and the product of the secondary capacitor voltage V2 are added by the adder 38a, and the result is added to the multiplier 37f.
- the characteristic is that the value obtained by multiplying 0.5 is used as the coupling part voltage VL.
- the voltage detector 6 that detects the coupling unit voltage VL can be omitted, and the DCDC converter device can be made smaller and lighter.
- FIG. 52 is a diagram showing a configuration of the bidirectional step-up / step-down DCDC converter device according to the tenth embodiment of the present invention.
- the control unit 30j has the following characteristics.
- FIG. 53 is a diagram showing a configuration example of the control unit 30j in the tenth embodiment of the present invention.
- the signal output from the duty ratio command generator 34d has been changed to VREF, and the duty ratio command generator 34d and the gate signal generator 35c have the following characteristics.
- Fig. 54 is a diagram illustrating a configuration example of the conduction ratio command generation unit 34d in the tenth embodiment of the present invention. Although not shown in the figure, it is possible to remove unnecessary frequency components by inserting a low-pass filter or the like at the input / output of the functional block such as the adder 252.
- the adder 252 calculates the sum of the primary side capacitor voltage VI and the secondary side capacitor voltage V2.
- the primary side capacitor voltage VI is divided by the sum of the primary side capacitor voltage VI and the secondary side capacitor voltage V2 in the divider 250, and the primary side capacitor voltage VI, the primary side capacitor voltage VI, and the secondary side are divided.
- the ratio VIZ (V1 + V2) to the sum of the side capacitor voltage V2 is obtained. This is the basic flow rate command VREF0 common to the primary and secondary converters la and lb.
- a value obtained by adding the current error DIL to the basic conduction rate command VREF0 by the adder 251 is used as a conduction rate command VREF common to the primary side and secondary side conversion units la and lb. .
- FIG. 55 is a diagram showing a configuration example of the gate signal generation unit 35c in the tenth embodiment of the present invention.
- carrier signal generation section 261 generates carrier signal CAR that takes a value between 0 and 1.
- carrier signal CAR a triangular wave, a sawtooth wave or the like is appropriate.
- the comparator 260 and the inverting circuit 262 determine the ON / OFF state of the gate signals Gla to G2 b of the switching elements lla to 12b according to the following logic, based on the magnitude relationship between the duty ratio command VREF and the carrier signal CAR. To do.
- Gla is turned on, and gate signal G2a of switching element 12a is turned off.
- the gate signal G2b of the switching element 12b of the secondary converter lb is turned on, and the gate signal Gib of the switching element ib is turned off.
- the primary side capacitor voltage VI and the secondary side capacitor voltage V2 are reduced. If they are equal, the duty ratio command VREF is 0.5, and the on / off duty ratios of the switching elements 11a and 12a and the switching elements l ib and 12b are 50%.
- the on / off duty ratio of switching elements 11a and 12a and switching elements l ib and 12b is around 50%, respectively, according to the ratio. Will increase or decrease.
- the switching elements lla to 12b operate with a narrow pulse width exceeding the limit. Therefore, the switching elements lla to 12b can accurately output the norse width according to the given gate signal, and as a result, the combined rear tutor current IL becomes the combined rear tutor current command IL * force. Can be avoided.
- the configuration of the bidirectional buck-boost DCDC converter device according to Embodiment 11 of the present invention will be described in detail with reference to the drawings.
- the configuration of the eleventh embodiment is based on the configuration of the tenth embodiment. Only the differences from the configuration of the bidirectional buck-boost DCDC converter device according to Embodiment 10 of the present invention will be described below.
- FIG. 56 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device according to Embodiment 11 of the present invention.
- the feature is that the voltage detector 6 for detecting the coupling unit voltage VL is omitted, and the control unit 30k has the following features.
- FIG. 57 shows a configuration example of the control unit 30k in the eleventh embodiment of the present invention.
- the product of the value obtained by subtracting the conduction ratio command VREF from 1.0 by the subtractor 39b, calculated by the multiplier 37h, and the product of the primary capacitor voltage VI are added by the adder 38b, and the multiplier 37i
- the characteristic is that the value obtained by multiplying 0.5 is used as the coupling part voltage VL.
- the voltage detector 6 that detects the coupling unit voltage VL can be omitted, and the DCDC converter device can be configured to be smaller and lighter. [0169] Embodiment 12.
- FIG. 58 is a diagram showing a configuration of the bidirectional buck-boost DCDC converter device according to the twelfth embodiment of the present invention.
- the current detector 4 for detecting the coupled rear tutor current IL and the voltage detector 6 for detecting the coupling unit voltage VL are omitted, and a current detector 7a for detecting the primary side switching circuit current II is added to the primary side conversion unit la.
- the control section 30m has the following characteristics.
- FIG. 59 shows a configuration example of the control unit 30m in the twelfth embodiment of the present invention.
- the primary side capacitor voltage VI is input to the current command conversion unit 31c, and the output signal from the current command conversion unit 31c is the primary side switching circuit basic current command 110 *.
- the output of the current command adjustment unit 32c is the primary switching circuit current command II *, and the configuration is such that the primary switching circuit current II is input to the current control unit 33b.
- the current command conversion unit 31c, current command adjustment unit 32c, and current control unit 33b have the following characteristics.
- FIG. 60 is a diagram showing a configuration example of the current command conversion unit 31c in the twelfth embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted at the input / output of the divider 42 to remove unnecessary frequency components.
- the divider 42 generates the primary switching circuit basic current command 110 * by dividing the power command P * by the primary capacitor voltage VI.
- the primary side capacitor voltage VI is input instead of the coupling unit voltage VL, and the primary side switching circuit basic current command 110 * is output instead of the combined rear tutor basic current command IL0 *. Is different.
- FIG. 61 is a diagram showing a configuration example of the current command adjustment unit 32c in the twelfth embodiment of the present invention. Although not shown, it is not necessary to insert a low-pass filter at the input / output of limiter 70c. It can also be configured to remove frequency components.
- the limit command 70c is applied to the primary side switching circuit basic current command 110 * by the limiter 70c, and the current command upper limit value ILMTL.
- the value with the upper and lower limits restricted by is output as the primary side switching circuit current command II *.
- limiter 70c The effect brought about by limiter 70c is the same as that of limiter 70a in the first embodiment, and thus the description thereof is omitted.
- primary switching circuit basic current command 110 * is input instead of coupled rear tuttle basic current command IL0 *, and primary switching circuit current is substituted for coupled rear tuttle current command IL *.
- command II * is output.
- FIG. 62 is a diagram showing a configuration example of the current control unit 33b in the twelfth embodiment of the present invention. Although not shown, a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 202 to remove unnecessary frequency components.
- the subtractor 202 generates a deviation between the primary side switching circuit current command II * and the primary side switching circuit current II, and inputs this to the proportional-plus-integral controller 203.
- a current error DIL is obtained as an output of the proportional integral controller 203.
- the primary switching circuit current command II * is input instead of the coupled rear tuttle current command IL *, and the primary switching circuit current is substituted for the coupled rear tuttle current IL.
- the difference is that II is input.
- the control method shown in the twelfth embodiment focuses on the primary side switching circuit unit power P1 and performs control so as to match the power command P *.
- the power command P * is converted into the corresponding primary-side switching circuit current command II *, and control is performed so that the actual primary-side switching circuit current II matches this.
- the loss in the primary side conversion unit la, the coupling unit lc, and the secondary side conversion unit lb and the fluctuation of the energy accumulated in the primary side capacitor 13a and the secondary side capacitor 13b are very small. Therefore, if this is ignored, the input / output power P10 of the primary power supply 2a, the primary switching circuit power P1 and the input / output power P20 of the secondary power supply 2b are equal on an instantaneous value basis.
- Primary side power supply by controlling the primary side switching circuit power P1 It is possible to control the power flow between 2a and the secondary power supply 2b.
- the force that ignores the fluctuation of the energy accumulated in the primary side capacitor 13a and the secondary side capacitor 13b as being minute is not negligible. However, it is stored in the primary side capacitor 13a and the secondary side capacitor 13b !, and by adjusting the power command P * or the primary side switching circuit current command ⁇ * according to the amount of energy fluctuation, the power flow transients Control accuracy can be improved
- the control unit 30m is configured to input the power command P * from the outside.
- the primary side switching circuit basic current command 110 * or the primary side switching is used instead of the power command P *.
- the current command conversion unit 31c and the current command adjustment unit 32c can be omitted.
- a signal corresponding to the circuit current command II * may be input to the control unit 30m from the outside.
- FIG. 63 shows the configuration of the bidirectional buck-boost DCDC converter device according to the thirteenth embodiment of the present invention.
- the current detector 4 for detecting the coupled rear tutor current IL and the voltage detector 6 for detecting the coupled voltage VL are omitted, and the secondary side switching circuit lb is switched to the secondary side switching circuit.
- a feature is that a current detector 7b for detecting the path current 12 is added, and the control unit 30 ⁇ has the following features.
- FIG. 64 shows a configuration example of the control unit 30 ⁇ in the thirteenth embodiment of the present invention.
- the configuration is such that the secondary-side capacitor voltage V2 is input to the current command converter 31d, and the output signal from the current command converter 3 Id is the secondary switching circuit basic current.
- Command 120 * the output of the current command adjustment unit 32d is the secondary switching circuit current command 12 *, and the secondary switching circuit current 12 is input to the current control unit 33c.
- the current command conversion unit 31d, the current command adjustment unit 32d, and the current control unit 33c have the following features.
- FIG. 65 is a diagram showing a configuration example of the current command conversion unit 3 Id in the thirteenth embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted at the input / output of the divider 43 to remove unnecessary frequency components.
- the divider 43 generates a secondary switching circuit basic current command 120 * by dividing the power command P * by the secondary capacitor voltage V2.
- the secondary side capacitor voltage V2 is input instead of the coupling unit voltage VL, and the secondary side switching circuit basic current command ILO * is replaced with the secondary side switching circuit basic current command 120 *. Is different in that is output.
- FIG. 66 is a diagram showing a configuration example of the current command adjustment unit 32d in the thirteenth embodiment of the present invention. Although not shown, it may be configured to remove unnecessary frequency components by inserting a low-pass filter or the like at the input / output of limiter 70d.
- the limit value 70d is used to limit the current command upper limit value ILMTH and the current command lower limit value for the secondary switching circuit basic current command 120 *. Secondary side switching circuit current command with upper and lower limits limited by ILMTL
- limiter 70d is the same as that of limiter 70a in the first embodiment, and thus the description thereof is omitted.
- FIG. 67 shows a configuration example of the current control unit 33c in the thirteenth embodiment of the present invention.
- a configuration may be adopted in which a low-pass filter or the like is inserted into the input / output of a functional block such as the subtractor 204 to remove unnecessary frequency components!
- the subtracter 204 generates a deviation between the secondary side switching circuit current command 12 * and the secondary side switching circuit current 12, and inputs this to the proportional-plus-integral controller 205.
- a current error DIL is obtained as an output of the proportional-integral controller 205.
- the secondary switching circuit current command 12 * is input instead of the coupled rear tutor current command IL *, and the secondary switching is performed instead of the coupled rear tuttle current IL.
- circuit current 12 is input.
- the power passing through the first terminal 15b and the second terminal 16b of the secondary side switching circuit 10b (hereinafter referred to as secondary side switching circuit section power P2 This is controlled so as to match the power command P *. V.
- the power command P * is converted into the corresponding secondary-side switching circuit current command 12 *, and control is performed so that the actual secondary-side switching circuit current 12 matches this. It is.
- the loss in the primary side conversion unit la, the coupling unit lc, and the secondary side conversion unit lb and the fluctuation of the energy stored in the primary side capacitor 13a and the secondary side capacitor 13b are very small. Therefore, if this is ignored, the input / output power P10 of the primary power supply 2a, the secondary switching circuit power P2, and the input / output power P20 of the secondary power supply 2b are equal on an instantaneous value basis.
- the secondary-side switching circuit unit power P2 By controlling the secondary-side switching circuit unit power P2, the power flow between the primary-side power source 2a and the secondary-side power source 2b can be controlled.
- control unit 30 ⁇ is configured to receive the power command ⁇ * from the outside, and instead of the power command ⁇ *, the secondary switching circuit basic current command 120 * or In this case, the command corresponding to the secondary side switching circuit current command 12 * may be input to the control unit 30 ⁇ from the outside. In this case, the current command conversion unit 31d and the current command adjustment unit 32d can be omitted.
- Embodiments 1 to 13 described above are examples of the embodiment of the present invention and the configuration thereof, and are not limited to this. It goes without saying that the contents of the present invention can be implemented even if the configuration is modified within a range not impairing the technical meaning.
- FIG. 68 is a diagram showing an application example of the bidirectional buck-boost DCDC converter device according to Embodiment 14 of the present invention.
- the bidirectional step-up / step-down DCDC converter device 285 operates to release an appropriate amount of electric power from the power storage device 186 at an appropriate timing such as when the vehicle is running, and vice versa.
- the power storage device 286 operates to absorb an appropriate amount of power at an appropriate timing such as during braking.
- Bidirectional buck-boost DCDC converter device 285 is configured to realize a power flow that matches power command P * input from drive control inverter device 282. It is controlled by the means indicated by 1-13.
- the power command P * may be input from a device other than the drive control inverter device 282 (for example, a vehicle information management device, not shown). Further, although it has a function of transmitting the operation state to the drive control inverter device 282, it may be transmitted to a device other than the drive control inverter device 282 (for example, a vehicle information management device, not shown).
- the configuration may be good.
- FIG. 69 is a diagram showing an application example of the bidirectional buck-boost DCDC converter device according to the fifteenth embodiment of the present invention.
- the overhead line 280 and the rail 284 are connected.
- the power storage device 286 has a function of releasing the power to the overhead wire 280 side, and conversely, the overhead wire 280 side force also absorbs the power.
- the bidirectional buck-boost DCDC converter device 285 operates so as to release an appropriate amount of power from the power storage device 286 when the voltage of the overhead line 280 drops, and vice versa. When the power rises, the power storage device 286 operates to absorb an appropriate amount of power.
- control for realizing a power flow that matches the power command P * from the system controller 289 may be performed.
- Bidirectional buck-boost DCDC converter apparatus 285 allows bidirectional power control while setting the terminal voltage of power storage device 286 to an optimum value related to the overhead wire voltage. Is possible. As a result, the voltage of the power storage device 286 can be increased from the voltage of the overhead line 280, and the current of the bidirectional buck-boost DCDC converter device 285 and the power storage device 286 can be reduced. This makes it possible to build an electric railway system.
- Embodiments 14 and 15 are merely examples of applications of the bidirectional buck-boost DCDC converter device, and are not limited to this. Needless to say, the present invention can be applied to various fields that handle DC power, such as devices, hybrid cars, electric cars, and DC power supplies.
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- Dc-Dc Converters (AREA)
- Electric Propulsion And Braking For Vehicles (AREA)
Abstract
Description
Claims
Priority Applications (8)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
CN2006800539415A CN101401287B (zh) | 2006-03-22 | 2006-03-22 | 双向升降压dcdc转换器装置、铁路车辆驱动控制***及电气铁路供电*** |
KR1020087023115A KR100975485B1 (ko) | 2006-03-22 | 2006-03-22 | 쌍방향 승강압 dcdc 컨버터 장치, 철도 차량 구동 제어 시스템, 전기 철도 급전 시스템 |
CA 2646226 CA2646226C (en) | 2006-03-22 | 2006-03-22 | Bidirectional buck boost dc-dc converter, railway coach drive control system, and railway feeder system |
JP2006552392A JP4094649B2 (ja) | 2006-03-22 | 2006-03-22 | 双方向昇降圧dcdcコンバータ装置、鉄道車両駆動制御システム、電気鉄道き電システム |
US12/293,828 US7723865B2 (en) | 2006-03-22 | 2006-03-22 | Bidirectional buck boost DC-DC converter, railway coach drive control system, and railway feeder system |
PCT/JP2006/305734 WO2007108115A1 (ja) | 2006-03-22 | 2006-03-22 | 双方向昇降圧dcdcコンバータ装置 |
EP06729701.0A EP1998428B9 (en) | 2006-03-22 | 2006-03-22 | Bidirectional buck boost dc/dc converter, railway coach drive control system, and railway feeder system |
HK09106977A HK1128998A1 (en) | 2006-03-22 | 2009-07-29 | Bidirectional step-up/step-down dc/dc converter apparatus,railway coach drive control system and railway feeder system |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
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PCT/JP2006/305734 WO2007108115A1 (ja) | 2006-03-22 | 2006-03-22 | 双方向昇降圧dcdcコンバータ装置 |
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WO2007108115A1 true WO2007108115A1 (ja) | 2007-09-27 |
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Family Applications (1)
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PCT/JP2006/305734 WO2007108115A1 (ja) | 2006-03-22 | 2006-03-22 | 双方向昇降圧dcdcコンバータ装置 |
Country Status (8)
Country | Link |
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US (1) | US7723865B2 (ja) |
EP (1) | EP1998428B9 (ja) |
JP (1) | JP4094649B2 (ja) |
KR (1) | KR100975485B1 (ja) |
CN (1) | CN101401287B (ja) |
CA (1) | CA2646226C (ja) |
HK (1) | HK1128998A1 (ja) |
WO (1) | WO2007108115A1 (ja) |
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KR20080102228A (ko) | 2008-11-24 |
KR100975485B1 (ko) | 2010-08-11 |
EP1998428B1 (en) | 2021-03-17 |
US20100045102A1 (en) | 2010-02-25 |
CN101401287A (zh) | 2009-04-01 |
EP1998428A4 (en) | 2011-08-24 |
HK1128998A1 (en) | 2009-11-13 |
JP4094649B2 (ja) | 2008-06-04 |
EP1998428B9 (en) | 2021-08-11 |
CN101401287B (zh) | 2013-05-01 |
US7723865B2 (en) | 2010-05-25 |
CA2646226A1 (en) | 2007-09-27 |
JPWO2007108115A1 (ja) | 2009-07-30 |
CA2646226C (en) | 2011-05-24 |
EP1998428A1 (en) | 2008-12-03 |
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