US20160056808A9 - Switching power converter input voltage approximate zero crossing determination - Google Patents

Switching power converter input voltage approximate zero crossing determination Download PDF

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US20160056808A9
US20160056808A9 US14/104,137 US201314104137A US2016056808A9 US 20160056808 A9 US20160056808 A9 US 20160056808A9 US 201314104137 A US201314104137 A US 201314104137A US 2016056808 A9 US2016056808 A9 US 2016056808A9
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voltage
zero crossing
input voltage
controller
input
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US20150171853A1 (en
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Eric J. King
John L. Melanson
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Koninklijke Philips NV
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Cirrus Logic Inc
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/13—Modifications for switching at zero crossing
    • H05B33/0815—
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/10—Controlling the intensity of the light
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/30—Driver circuits
    • H05B45/37—Converter circuits
    • H05B45/3725—Switched mode power supply [SMPS]
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps
    • Y02B20/30—Semiconductor lamps, e.g. solid state lamps [SSL] light emitting diodes [LED] or organic LED [OLED]

Definitions

  • the present invention relates in general to the field of electronics, and more specifically to a method and system for utilizing a switching power converter and determining an approximate zero crossing of an input voltage to the switching power converter.
  • Many electronic systems include circuits, such as switching power converters that interface with a dimmer.
  • the interfacing circuits deliver power to a load in accordance with the dimming level set by the dimmer.
  • dimmers provide an input signal to a lighting system.
  • the input signal represents a dimming level that causes the lighting system to adjust power delivered to a lamp, and, thus, depending on the dimming level, increase or decrease the brightness of the lamp.
  • dimmers generate a digital or analog coded dimming signal that indicates a desired dimming level.
  • some analog based dimmers utilize a triode for alternating current (“triac”) device to modulate a phase angle of each cycle of an alternating current (“AC”) supply voltage.
  • Triac alternating current
  • AC alternating current
  • “Modulating the phase angle” of the supply voltage is also commonly referred to as “phase cutting” the supply voltage. Phase cutting the supply voltage reduces the average power supplied to a load, such as a lighting system, and thereby controls the energy provided to the load.
  • the switching power converter draws a minimum current referred to as a “hold current”.
  • a hold current As long as an input current to the switching power converter is greater than or equal to the hold current, the triac-based dimmer should not prematurely disconnect.
  • a premature disconnect occurs when the dimmer begins conducting and stops conducting prior to reaching a zero crossing of the supply voltage. Premature disconnects can cause problems with the lighting system, such as flicker and instability.
  • FIG. 1 depicts a lighting system 100 that includes a leading edge, phase-cut dimmer 102 .
  • FIG. 2 depicts ideal, exemplary voltage graphs 200 associated with the lighting system 100 .
  • the lighting system 100 receives an AC supply voltage V IN from voltage supply 104 .
  • the supply voltage V IN indicated by voltage waveform 202 , is, for example, a nominally 60 Hz/110 V line voltage in the United States of America or a nominally 50 Hz/220 V line voltage in Europe.
  • a leading edge dimmer 102 phase cuts leading edges, such as leading edges 204 and 206 , of each half cycle of supply voltage V IN .
  • the leading edge dimmer 102 phase cuts the supply voltage V IN at an angle greater than 0 degrees and less than 180 degrees.
  • the voltage phase cutting range of a leading edge dimmer 102 is 10 degrees to 170 degrees.
  • the input signal voltage V ⁇ — IN to the lighting system 100 represents a dimming level that causes the lighting system 100 to adjust power delivered to a lamp 122 , and, thus, depending on the dimming level, increase or decrease the brightness of the lamp 122 .
  • the leading edge dimmer 102 can be any type of leading edge dimmer, such as a triac-based leading edge dimmer available from Lutron Electronics, Inc. of Coopersberg, Pa. (“Lutron”). A triac-based leading edge dimmer is described in the Background section of U.S. patent application Ser. No. 12/858,164, entitled Dimmer Output Emulation, filed on Aug. 17, 2010, and inventor John L. Melanson.
  • the phase cut dimmer 102 supplies the input voltage V ⁇ — IN as modified by the phase cut dimmer 102 to a full bridge diode rectifier 106 .
  • the full bridge rectifier 106 supplies an AC rectified voltage V ⁇ R — IN to the switching power converter 108 .
  • the rectified input voltage V ⁇ R — IN is also derived from the input supply voltage V IN .
  • Capacitor 110 filters high frequency components from rectified voltage V ⁇ R — IN .
  • controller 110 generates a control signal CS 0 to control conductivity of field effect transistor (FET) switch 112 .
  • FET field effect transistor
  • the control signal CS 0 is a pulse width modulated signal.
  • Control signal CS 0 waveform 114 represents an exemplary control signal CS 0 .
  • Each pulse of control signal CS 0 turns switch 112 ON (i.e. conducts), and the inductor current i L increases, as shown in the exemplary inductor current waveform 115 , to charge inductor 116 during a charging phase T C .
  • Diode 118 prevents current flow from link capacitor 120 into switch 112 .
  • the inductor 116 reverses voltage polarity (commonly referred to as “flyback”), and the inductor current i L decreases during the flyback phase T FB , as shown inductor current waveform 115 .
  • the inductor current i L boosts the link voltage across the link capacitor 120 through diode 118 .
  • the switching power converter 108 is a boost-type converter, and, thus, the link voltage V LINK is greater than the rectified input voltage V ⁇ R — IN .
  • Controller 110 senses the rectified input voltage V ⁇ R — IN at node 124 and senses the link voltage V LINK at node 126 .
  • Controller 110 operates the switching power converter 108 to maintain an approximately constant link voltage V LINK for lamp 122 , provide power factor correction, and correlate the output current i OUT with the phase cut angle of the rectified input voltage V ⁇ R — IN .
  • Lamp 132 includes one or more light emitting diodes.
  • an apparatus in one embodiment, includes a controller having an input to sense a leading edge, phase cut alternating current (AC) input voltage to a switching power converter at least at a first time during a cycle of the AC input voltage.
  • the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by the controller.
  • the controller is configured to determine the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
  • a method in another embodiment, includes receiving a sense signal indicating a leading edge, phase cut alternating current (AC) input voltage to a switching power converter.
  • the sense signal is received at least at a first time during a cycle of the AC input voltage, the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by a controller of a switching power converter.
  • the method further includes determining the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
  • a method in another embodiment, includes exposing a leading edge, phase cut alternating current (AC) input voltage supplied to a switching power converter for part of a cycle of the AC input voltage.
  • the exposed AC input voltage is used to supply current to a load, and the AC input voltage is derived from an AC input supply voltage.
  • the method further includes sensing the AC input voltage and ceasing exposure of the AC input voltage after sensing the AC input voltage.
  • the method further includes determining an approximate zero crossing of the AC input supply voltage based on a value of the sensed AC input voltage.
  • an apparatus in a further embodiment, includes means for receiving a sense signal indicating a leading edge, phase cut alternating current (AC) input voltage to a switching power converter.
  • the sense signal is received at least at a first time during a cycle of the AC input voltage, the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by a controller of a switching power converter.
  • the apparatus further includes means for determining the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
  • FIG. 1 (labeled prior art) depicts a lighting system that includes a triac-based dimmer.
  • FIG. 2 (labeled prior art) depicts exemplary voltage graphs associated with the lighting system of FIG. 1 .
  • FIG. 3 depicts an electronic system that includes a controller having a zero crossing calculator.
  • FIG. 4 depicts exemplary input supply voltage and a rectified input voltage for the system of FIG. 3 .
  • FIG. 5 depicts a controller that represents an embodiment of the controller of FIG. 3 .
  • FIG. 6 depicts an exemplary approximate zero crossing determination process.
  • FIGS. 7A and 7B depict exemplary approximate zero crossing determination code.
  • FIG. 8 depicts an exemplary capacitor/current model utilized by the approximate zero crossing determination code of FIG. 7 .
  • FIG. 9 depicts an electronic system that represents an embodiment of the electronic system of FIG. 3 .
  • an electronic system includes a controller, and the controller determines an approximate zero crossing of an alternating current (AC) input supply voltage to a switching power converter based on a voltage value of the AC input voltage.
  • AC alternating current
  • the term “approximate” as used herein means exact or sufficiently exact. A zero crossing is sufficiently exact if the zero crossing can be used by the controller in lieu of the actual zero crossing. Actually ‘detecting’ a zero crossing of the AC input supply voltage, as opposed to ‘determining’an approximate zero crossing of the AC input supply voltage, can be problematic in some circumstances. For example, triac-based dimmers have conventionally been used with incandescent lamps.
  • Incandescent lamps are generally immune from disturbances in the phase cut voltage from the triac-based dimmer, such as premature disconnection and electronic noise generated by a triac-based dimmer.
  • disturbances of a supply voltage by a dimmer such as premature disconnection, premature conduction, and electronic noise, can be problematic to relatively low-power, actively controlled electronic systems, such as light emitting diode (LED) based lighting systems.
  • the controller utilizes the zero crossing of the AC input voltage to begin one or more operations, such as providing a sufficiently low input impedance for the dimmer at the zero crossing to hold a dimmer output voltage at approximately zero volts as described in, for example, U.S. patent application Ser. No.
  • the controller senses a leading edge, phase cut AC input voltage value to a switching power converter during a cycle of the AC input voltage.
  • the controller senses the voltage value at a time prior to a zero crossing of the AC input voltage.
  • the controller utilizes the voltage value to determine an approximate zero crossing of the AC input supply voltage.
  • the controller by determining an approximate zero crossing of the AC input voltage, the controller is unaffected by any disturbances of the dimmer that could otherwise make detecting the zero crossing from sensing the actual AC input voltage problematic. The particular way of determining an approximate zero crossing is a matter of design choice.
  • the controller approximates the AC input voltage using a function that estimates a waveform of the AC input voltage and determines the approximate zero crossing of the AC input voltage from the approximation of the AC input voltage.
  • the particular function can be any type of function, such as a polynomial function or a trigonometric function.
  • the controller includes dedicated circuits to determine the approximate zero crossing.
  • the controller includes a processor and a memory, and the memory includes code that is executable by the processor to determine the approximate zero crossing.
  • the controller includes a look-up table that identifies when the zero crossing will occur based on the sensed voltage value.
  • the controller senses the voltage value during a portion of the AC input voltage when the dimmer has a high probability of providing a relatively undisturbed input voltage to the switching power converter. Sensing the voltage value during a relatively undisturbed portion of the AC input voltage allows the controller to utilize a voltage value that accurately represents a voltage value of a supply voltage to the dimmer. Additionally, in at least one embodiment, the controller senses the voltage value when the phase cut voltage is relatively undisturbed and when the switching power converter has received sufficient power to meet power demands by a load.
  • the controller after the controller senses the voltage value, the controller causes the electronic system to dissipate excess energy as, for example, (i) described in U.S. patent application Ser. No. 13/289,845, filed Nov. 4, 2011, entitled “Controlled Power Dissipation in a Switch Path in a Lighting System”, and inventors John L. Melanson and Eric J. King, (ii) U.S. patent application Ser. No. 13/289,931, filed Nov. 4, 2011, entitled “Controlled Power Dissipation in a Lighting System”, and inventors John L. Melanson and Eric J. King, (iii) Ser. No. 13/289,967 filed Nov.
  • FIG. 3 depicts an electronic system 300 that includes a zero crossing calculator 302 of controller 304 that determines the approximate zero crossing of the rectified input voltage V ⁇ R — IN .
  • FIG. 4 depicts exemplary voltage waveforms 400 of the input supply voltage V IN and rectified input voltage V ⁇ R — IN .
  • dimmer 306 is a leading edge, phase cut dimmer.
  • Dimmer 306 can be any type of leading edge, phase cut dimmer including a triac-based dimmer or a field effect transistor (FET) based dimmer.
  • FET field effect transistor
  • the rectified input voltage V ⁇ R — IN depicts two cycles, cycle A and cycle B, which are derived from the cycle 401 of the input supply voltage V IN .
  • Cycle A is a phase cut version of the first half cycle 402 of the input supply voltage V IN
  • cycle B is a rectified, phase cut version of the second half cycle 404 of the input supply voltage V IN .
  • Cycle A occurs from time t 0 until the zero crossing of the input supply voltage V IN at time t 3 .
  • Cycle B occurs from time t 3 until the next zero crossing at time t 6 of the input supply voltage V IN .
  • the dimmer 306 phase cuts the input supply voltage V IN from voltage supply 104 to generate the phase cut input voltage V ⁇ — IN .
  • the full-bridge diode rectifier rectifies the phase cut input voltage V ⁇ — IN to generate the rectified input voltage V ⁇ R — IN .
  • the dimmer 306 does not conduct current from the voltage supply 104 and, thus, phase cuts the supply voltage V IN until time t 1 .
  • the dimmer 306 conducts so that the rectified input voltage V ⁇ R — IN equals the input voltage V IN .
  • the controller 304 senses the voltage value v(0) A of rectified input voltage V ⁇ R — IN during cycle A of the rectified input voltage V ⁇ R — IN .
  • the controller 304 senses the voltage value v(0) A downstream (right side) of the rectifier 106 from the phase-cut input voltage V ⁇ — IN . In at least one embodiment, the controller 304 senses the voltage value v(0) A upstream (left side) of the rectifier 106 from the phase-cut input voltage V ⁇ — IN .
  • the controller 304 includes a zero crossing calculator 302 to determine an approximate zero crossing of the rectified input voltage V ⁇ R — IN . For cycle A of the rectified input voltage V ⁇ R — IN , the zero crossing occurs at time t 3 .
  • the particular implementation of the zero crossing calculator 302 is a matter of design choice. Various exemplary embodiments of the zero crossing calculator 302 are subsequently described in more detail.
  • the time t 2 is selected as the time to sense the voltage value v(0) A because the power converter 308 has received sufficient power from voltage supply 104 to maintain an approximately constant link voltage V LINK and meet power demands of load 310 .
  • the controller 304 maintains the rectified input voltage V ⁇ R — IN at the voltage value v(0) A until reaching the zero crossing at time t 3 . Maintaining the voltage of the rectified input voltage V ⁇ R — IN effectively stops the current flow i IN into the power converter 308 .
  • the controller 304 causes the rectified input voltage V ⁇ R — IN to rapidly decrease to approximately 0 volts. How to dissipate the energy associated with decreasing the rectified input voltage V ⁇ R — IN to approximately 0 volts is also a matter of design choice.
  • the power is dissipated as, for example, described in any or all of the Power Dissipation Applications and/or through another power dissipation circuit.
  • Controller 304 continues to determine the approximate zero crossing of the rectified input voltage V ⁇ R — IN in subsequent cycles of the rectified input voltage V ⁇ R — IN as, for example, shown in cycle B.
  • cycle B of the rectified input voltage V ⁇ R — IN the phase cut dimmer 306 phase cuts the rectified input voltage V ⁇ R — IN from time t 3 until time t 4 .
  • the controller 304 senses the voltage value v(0) B of the rectified input voltage V ⁇ R — IN at time t 5 .
  • the zero crossing calculator 302 determines the approximate zero crossing time that occurs at time t 6 as previously described.
  • the actual zero crossings of the rectified input voltage V ⁇ R — IN are not directly observable by the controller 304 , and, thus, are not actually detectable by the controller 304 .
  • the sensing times of the rectified input voltage V ⁇ R — IN are within a range 0.5-5 ms of the approximate zero crossing time, such as respective times t 3 and t 6 .
  • the sensing times of the rectified input voltage V ⁇ R — IN are within a range of 0.25-5 ms of the approximate zero crossing time, such as respective times t 3 and t 6 .
  • the range sensing times are inversely linearly related with respect to the frequency of the rectified input voltage V ⁇ R — IN , e.g. for a 240 Hz rectified input voltage V ⁇ R — IN , the sensing times are reduced by 50%.
  • the sensed voltage such as sensed voltage v(0) A and v(0) B , are greater than or equal to 50V or, in at least one embodiment, greater than 0.3 times an RMS peak value of the rectified input voltage V ⁇ R — IN .
  • the controller 304 controls the power converter 308 .
  • the particular type of power converter 308 is a matter of design choice.
  • the power converter 308 can be a boost-type switching power converter such as switching power converter 108 , a buck type switching power converter, a boost-buck type switching power converter, or a C ⁇ k type switching power converter.
  • controller 304 controls the power converter 308 as described in, for example, U.S. patent application Ser. No. 11/967,269, entitled “Power Control System Using a Nonlinear Delta-Sigma Modulator With Nonlinear Power Conversion Process Modeling”, filed on Dec. 31, 2007, inventor John L. Melanson (referred to herein as “Melanson II”), U.S. patent application Ser. No.
  • FIG. 5 depicts a controller 500 , which represents one embodiment of controller 304 .
  • Controller 500 includes zero crossing calculator 502 , which represents one embodiment of the zero crossing calculator 302 .
  • FIG. 6 depicts an approximate zero crossing determination process 600 , which represents one embodiment of an approximate zero crossing determination process utilized by the controller 500 to determine an approximate zero crossing of the rectified input voltage V ⁇ R — IN .
  • the controller 500 observes the rectified input voltage V ⁇ R — IN .
  • the controller selects a value v(0) of the rectified input voltage V ⁇ R — IN and stores the selected value v(0) in a register 504 .
  • the zero crossing calculator 500 includes a memory 506 that stores approximate zero crossing determination code 508 .
  • the processor 510 communicates with the memory 506 and, in operation 606 , executes the approximate zero crossing determination code 508 to synthesize the rectified input voltage V ⁇ R — IN and determine an approximate zero crossing of the synthesized rectified input voltage V ⁇ R — IN .
  • the processor 510 utilizes the sensed voltage value V(0) as an initial value to determine an approximate zero crossing of the rectified input voltage V ⁇ R — IN .
  • the particular implementation of the approximate zero crossing determination code 508 is a matter of design choice.
  • the approximate zero crossing determination code 508 implements a parabolic function as, for example, subsequently describedError! Reference source not found.
  • the approximate zero crossing determination code 508 implements other polynomial functions or trigonometric functions, such as an actual sine function, to synthesize the rectified input voltage V ⁇ R — IN to determine zero crossings of the rectified input voltage V ⁇ R — IN .
  • the approximate zero crossing determination code 508 includes code to access a look up table, and the look up table includes zero crossing times corresponding to possible values of the sensed voltage value V(0).
  • FIGS. 7A and 7B collectively depict zero crossing code 700 , which represents one embodiment of the zero crossing code 508 .
  • the zero crossing code 700 begins in FIG. 7A and continues in FIG. 7B .
  • the zero crossing code 700 implements an iterative approximate zero crossing determination process using a parabolic function based on modeling changes in voltage across a capacitor resulting from current flow from the capacitor.
  • FIG. 8 depicts an exemplary capacitor/current model 800 utilized by the zero crossing code 700 .
  • the capacitor/current model 800 models the input supply voltage V IN using a parabolic function to synthesize the input supply voltage V IN .
  • the zero crossing code 700 utilizes an initial sample V(0) of the rectified input voltage V ⁇ R — IN .
  • the capacitor/current model 800 models the sampling with a switch 802 that momentarily closes at a time t SENSE and then immediately opens.
  • the time t SENSE at which the rectified input voltage V ⁇ R — IN is sensed is a matter of design choice and, in at least one embodiment, is selected on a cycle-by-cycle basis of the rectified input voltage V ⁇ R — IN .
  • the sensing time t 0 occurs when the voltage supply 104 ( FIG. 3 ) has supplied sufficient power to meet the power demand of load 310 ( FIG. 3 ).
  • the time t SENSE corresponds to time t 2 for cycle A and time t 5 for cycle B of the rectified input voltage V ⁇ R — IN as depicted in FIG. 4 .
  • the time t SENSE is within 0.5-5 ms of the zero crossing time t ZC .
  • the time t SENSE is within 0.25-5 ms of the zero crossing time t zc .
  • the voltage-current graph 804 presents a linear relationship between the voltage V(n) across the capacitor 806 and the current I(V(n)) as, for example, represented by Equation [1]:
  • V(n) represents the voltage across capacitor 806 , which represents the input supply voltage V IN .
  • I(V(n)) represents the current discharged from capacitor 806 and is modeled by a varying current source 810 .
  • m is the slope of the V(n)/I(V(n)) relationship line 808
  • b is the y-intercept of the V(n)/I(V(n)) relationship line 808 .
  • the values of “m” and “b” are a matter of design choice and are, in at least one embodiment, chosen to best approximate an actual relationship between the input supply voltage V IN and the input current i IN ( FIG. 3 ) for the modeled capacitor 806 .
  • “dV(n)/dt” represents the change in voltage with respect to time
  • “C” represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • C represents a capacitance of capacitor 806 .
  • dt represents a change in time
  • f CALC is a constant time period at which the zero crossing indicator value ZC is updated.
  • the value of dt is chosen based on a desired accuracy of the zero crossing indicator value ZC.
  • the zero crossing value ZC is updated at a frequency f CALC , and
  • Equation [2] can be rearranged as Equation [3]:
  • Each subsequent value V(n+1) is related to the immediately preceding voltage value V(n) by Equation [4]:
  • V ( n+ 1) V ( n ) ⁇ dV ( n ) [4].
  • the initial value V(0) is provided by an actual sensed value of rectified input voltage V ⁇ R — IN at time t SENSE .
  • the voltage-current graph 804 provides a value of I(V) for each sample or calculation of V(n), and the value of each dV(n) for each increment of dt can, thus, be determined from Equation [3] and the value of I(V) from the voltage-current graph 804 . Since the relationship between V(n) and I(V) is linear in voltage-current graph 804 , the combination of Equations [1]-[4] result in a parabolic function, and the values of V(n+1) will decrease in accordance with the parabolic function of Equation [4].
  • the input supply voltage V IN is, in at least one embodiment, a sine wave.
  • the parabolic function of Equation [4] is relatively fast and easy to calculate and closely models a sine wave.
  • Comparator 812 compares the voltage value V(n) with a reference value V ZC — REF , and the zero crossing value ZC represents the result of the comparison.
  • the reference value V ZC — REF is chosen so that when the voltage value V(n) is less than the reference value V ZC — REF , the zero crossing value ZC changes state from a logical 0 to a logical 1 to indicate a zero crossing of the input supply voltage V IN .
  • the controller 304 FIG.
  • Equation [5] represents an approximation equation that can be used to iteratively determine an approximate zero crossing of the AC input supply voltage V IN :
  • V APPROX ( i ) V APPROX ( i ⁇ 1) ⁇ [ k 1 ⁇ V APPROX [i ⁇ 1]) ⁇ k 2 [5].
  • V APPROX (i) is the i th approximate zero crossing of the AC input supply voltage V IN
  • i is an integer index
  • V APPROX (i ⁇ 1) is the approximate zero crossing value that immediately precedes the value of V APPROX (i).
  • k 1 ” and “k 2 ” are scaling factors. k 1 relates to the peak value of the input supply voltage and k 2 relates to the frequency and, thus, the step-size of each iteration of Equation [5].
  • V APPROX (i) rounded to the nearest volt for Equation [5] are:
  • FIG. 9 depicts an electronic system 900 , which represents one embodiment of the electronic system 300 .
  • Controller 902 senses the rectified input voltage V ⁇ R — IN at node 124 .
  • Controller 902 can sense the rectified input voltage V ⁇ R — IN in any desired manner, such as through a resistor divider circuit 905 .
  • Zero crossing calculator 904 which represents one embodiment of the zero crossing calculator 302 ( FIG. 3 ), synthesizes the rectified input voltage V ⁇ R — IN as, for example, previously described in conjunction with zero crossing calculator 302 , 500 ( FIG. 5 ), or 700 ( FIG. 7 ).
  • controller 902 controls the switching power converter as, for example, Melanson II, Melanson III, Melanson IV, or Melanson V.
  • a controller senses a leading edge, phase cut alternating current (AC) input voltage to a switching power converter at least at a time t 0 during a cycle of the AC input voltage.
  • the time t 0 is prior to a zero crossing of the AC input voltage and at least some zero crossings of the AC input voltage are not directly observable by the controller.
  • the controller is configured to determine an approximate zero crossing of the AC input voltage based on a voltage value of the AC input voltage at time t 0 .

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Abstract

In at least one embodiment, the controller senses a leading edge, phase cut AC input voltage value to a switching power converter during a cycle of the AC input voltage. The controller senses the voltage value at a time prior to a zero crossing of the AC input voltage and utilizes the voltage value to determine the approximate zero crossing. In at least one embodiment, by determining an approximate zero crossing of the AC input voltage, the controller is unaffected by any disturbances of the dimmer that could otherwise make detecting the zero crossing problematic. The particular way of determining an approximate zero crossing is a matter of design choice. In at least one embodiment, the controller approximates the AC input voltage using a function that estimates a waveform of the AC input voltage and determines the approximate zero crossing of the AC input voltage from the approximation of the AC input voltage.

Description

    CROSS-REFERENCE TO RELATED APPLICATION
  • This application claims the benefit under 35 U.S.C. §119(e) and 37 C.F.R. §1.78 of U.S. Provisional Application No. 61/410,269, filed on Nov. 4, 2010, and is incorporated by reference in its entirety.
  • This application is a continuation of U.S. patent application Ser. No. 13/290,032, filed on Nov. 4, 2011 and now U.S. Pat. No. 8,610,365.
    • BACKGROUND OF THE INVENTION
  • 1. Field of the Invention
  • The present invention relates in general to the field of electronics, and more specifically to a method and system for utilizing a switching power converter and determining an approximate zero crossing of an input voltage to the switching power converter.
  • 2. Description of the Related Art
  • Many electronic systems include circuits, such as switching power converters that interface with a dimmer. The interfacing circuits deliver power to a load in accordance with the dimming level set by the dimmer. For example, in a lighting system, dimmers provide an input signal to a lighting system. The input signal represents a dimming level that causes the lighting system to adjust power delivered to a lamp, and, thus, depending on the dimming level, increase or decrease the brightness of the lamp. Many different types of dimmers exist. In general, dimmers generate a digital or analog coded dimming signal that indicates a desired dimming level. For example, some analog based dimmers utilize a triode for alternating current (“triac”) device to modulate a phase angle of each cycle of an alternating current (“AC”) supply voltage. “Modulating the phase angle” of the supply voltage is also commonly referred to as “phase cutting” the supply voltage. Phase cutting the supply voltage reduces the average power supplied to a load, such as a lighting system, and thereby controls the energy provided to the load.
  • Once a triac-based dimmer begins conducting during a cycle of an alternating current (“AC”) supply voltage, to prevent the triac from disadvantageously, prematurely disconnecting during mid-cycle of the supply voltage, the switching power converter draws a minimum current referred to as a “hold current”. As long as an input current to the switching power converter is greater than or equal to the hold current, the triac-based dimmer should not prematurely disconnect. For a leading edge dimmer, a premature disconnect occurs when the dimmer begins conducting and stops conducting prior to reaching a zero crossing of the supply voltage. Premature disconnects can cause problems with the lighting system, such as flicker and instability.
  • FIG. 1 depicts a lighting system 100 that includes a leading edge, phase-cut dimmer 102. FIG. 2 depicts ideal, exemplary voltage graphs 200 associated with the lighting system 100. Referring to FIGS. 1 and 2, the lighting system 100 receives an AC supply voltage VIN from voltage supply 104. The supply voltage VIN, indicated by voltage waveform 202, is, for example, a nominally 60 Hz/110 V line voltage in the United States of America or a nominally 50 Hz/220 V line voltage in Europe. A leading edge dimmer 102 phase cuts leading edges, such as leading edges 204 and 206, of each half cycle of supply voltage VIN. Since each half cycle of supply voltage VIN is 180 degrees of the supply voltage VIN, the leading edge dimmer 102 phase cuts the supply voltage VIN at an angle greater than 0 degrees and less than 180 degrees. Generally, the voltage phase cutting range of a leading edge dimmer 102 is 10 degrees to 170 degrees.
  • The input signal voltage Vφ — IN to the lighting system 100 represents a dimming level that causes the lighting system 100 to adjust power delivered to a lamp 122, and, thus, depending on the dimming level, increase or decrease the brightness of the lamp 122. The leading edge dimmer 102 can be any type of leading edge dimmer, such as a triac-based leading edge dimmer available from Lutron Electronics, Inc. of Coopersberg, Pa. (“Lutron”). A triac-based leading edge dimmer is described in the Background section of U.S. patent application Ser. No. 12/858,164, entitled Dimmer Output Emulation, filed on Aug. 17, 2010, and inventor John L. Melanson.
  • The phase cut dimmer 102 supplies the input voltage Vφ — IN as modified by the phase cut dimmer 102 to a full bridge diode rectifier 106. The full bridge rectifier 106 supplies an AC rectified voltage VφR — IN to the switching power converter 108. Thus, since the input voltage Vφ — IN is derived from the input supply voltage VIN, the rectified input voltage VφR — IN is also derived from the input supply voltage VIN. Capacitor 110 filters high frequency components from rectified voltage VφR — IN. To control the operation of switching power converter 108, controller 110 generates a control signal CS0 to control conductivity of field effect transistor (FET) switch 112. The control signal CS0 is a pulse width modulated signal. Control signal CS0 waveform 114 represents an exemplary control signal CS0. Each pulse of control signal CS0 turns switch 112 ON (i.e. conducts), and the inductor current iL increases, as shown in the exemplary inductor current waveform 115, to charge inductor 116 during a charging phase TC. Diode 118 prevents current flow from link capacitor 120 into switch 112. When the pulse ends, the inductor 116 reverses voltage polarity (commonly referred to as “flyback”), and the inductor current iL decreases during the flyback phase TFB, as shown in inductor current waveform 115. The inductor current iL boosts the link voltage across the link capacitor 120 through diode 118.
  • The switching power converter 108 is a boost-type converter, and, thus, the link voltage VLINK is greater than the rectified input voltage VφR — IN. Controller 110 senses the rectified input voltage VφR — IN at node 124 and senses the link voltage VLINK at node 126. Controller 110 operates the switching power converter 108 to maintain an approximately constant link voltage VLINK for lamp 122, provide power factor correction, and correlate the output current iOUT with the phase cut angle of the rectified input voltage VφR — IN. Lamp 132 includes one or more light emitting diodes.
  • It is desirable to improve interfacing with triac-based dimmers.
    • SUMMARY OF THE INVENTION
  • In one embodiment of the present invention, an apparatus includes a controller having an input to sense a leading edge, phase cut alternating current (AC) input voltage to a switching power converter at least at a first time during a cycle of the AC input voltage. The cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by the controller. The controller is configured to determine the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
  • In another embodiment of the present invention, a method includes receiving a sense signal indicating a leading edge, phase cut alternating current (AC) input voltage to a switching power converter. The sense signal is received at least at a first time during a cycle of the AC input voltage, the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by a controller of a switching power converter. The method further includes determining the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
  • In another embodiment of the present invention, a method includes exposing a leading edge, phase cut alternating current (AC) input voltage supplied to a switching power converter for part of a cycle of the AC input voltage. The exposed AC input voltage is used to supply current to a load, and the AC input voltage is derived from an AC input supply voltage. The method further includes sensing the AC input voltage and ceasing exposure of the AC input voltage after sensing the AC input voltage. The method further includes determining an approximate zero crossing of the AC input supply voltage based on a value of the sensed AC input voltage.
  • In a further embodiment of the present invention, an apparatus includes means for receiving a sense signal indicating a leading edge, phase cut alternating current (AC) input voltage to a switching power converter. The sense signal is received at least at a first time during a cycle of the AC input voltage, the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, and the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage. At least some zero crossings of the AC input supply voltage are not directly observable by a controller of a switching power converter. The apparatus further includes means for determining the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
    • BRIEF DESCRIPTION OF THE DRAWINGS
  • The present invention may be better understood, and its numerous objects, features and advantages made apparent to those skilled in the art by referencing the accompanying drawings. The use of the same reference number throughout the several figures designates a like or similar element.
  • FIG. 1 (labeled prior art) depicts a lighting system that includes a triac-based dimmer.
  • FIG. 2 (labeled prior art) depicts exemplary voltage graphs associated with the lighting system of FIG. 1.
  • FIG. 3 depicts an electronic system that includes a controller having a zero crossing calculator.
  • FIG. 4 depicts exemplary input supply voltage and a rectified input voltage for the system of FIG. 3.
  • FIG. 5 depicts a controller that represents an embodiment of the controller of FIG. 3.
  • FIG. 6 depicts an exemplary approximate zero crossing determination process.
  • FIGS. 7A and 7B (collectively referred to as “FIG. 7”) depict exemplary approximate zero crossing determination code.
  • FIG. 8 depicts an exemplary capacitor/current model utilized by the approximate zero crossing determination code of FIG. 7.
  • FIG. 9 depicts an electronic system that represents an embodiment of the electronic system of FIG. 3.
    •  DETAILED DESCRIPTION
  • In at least one embodiment, an electronic system includes a controller, and the controller determines an approximate zero crossing of an alternating current (AC) input supply voltage to a switching power converter based on a voltage value of the AC input voltage. The term “approximate” as used herein means exact or sufficiently exact. A zero crossing is sufficiently exact if the zero crossing can be used by the controller in lieu of the actual zero crossing. Actually ‘detecting’ a zero crossing of the AC input supply voltage, as opposed to ‘determining’an approximate zero crossing of the AC input supply voltage, can be problematic in some circumstances. For example, triac-based dimmers have conventionally been used with incandescent lamps. Incandescent lamps are generally immune from disturbances in the phase cut voltage from the triac-based dimmer, such as premature disconnection and electronic noise generated by a triac-based dimmer. However, disturbances of a supply voltage by a dimmer, such as premature disconnection, premature conduction, and electronic noise, can be problematic to relatively low-power, actively controlled electronic systems, such as light emitting diode (LED) based lighting systems. In at least one embodiment, the controller utilizes the zero crossing of the AC input voltage to begin one or more operations, such as providing a sufficiently low input impedance for the dimmer at the zero crossing to hold a dimmer output voltage at approximately zero volts as described in, for example, U.S. patent application Ser. No. 12/858,164, filed Aug. 17, 2010, entitled: “Dimmer Output Emulation”, and inventor: John L. Melanson (referred to herein as “Melanson I”) and U.S. patent application Ser. No. 13/217,174, filed Aug. 24, 2011, entitled: “Multi-Mode Dimmer Interfacing Including Attach State Control”, and inventors: Eric J. King and John L. Melanson, which are both incorporated by reference in their entireties.
  • In at least one embodiment, the controller senses a leading edge, phase cut AC input voltage value to a switching power converter during a cycle of the AC input voltage. The controller senses the voltage value at a time prior to a zero crossing of the AC input voltage. The controller utilizes the voltage value to determine an approximate zero crossing of the AC input supply voltage. In at least one embodiment, by determining an approximate zero crossing of the AC input voltage, the controller is unaffected by any disturbances of the dimmer that could otherwise make detecting the zero crossing from sensing the actual AC input voltage problematic. The particular way of determining an approximate zero crossing is a matter of design choice. In at least one embodiment, the controller approximates the AC input voltage using a function that estimates a waveform of the AC input voltage and determines the approximate zero crossing of the AC input voltage from the approximation of the AC input voltage. The particular function can be any type of function, such as a polynomial function or a trigonometric function. In at least one embodiment, the controller includes dedicated circuits to determine the approximate zero crossing. In at least one embodiment, the controller includes a processor and a memory, and the memory includes code that is executable by the processor to determine the approximate zero crossing. In at least one embodiment, the controller includes a look-up table that identifies when the zero crossing will occur based on the sensed voltage value.
  • Determining when to sense the voltage value during a cycle of the AC input voltage is a matter of design choice. In at least one embodiment, the controller senses the voltage value during a portion of the AC input voltage when the dimmer has a high probability of providing a relatively undisturbed input voltage to the switching power converter. Sensing the voltage value during a relatively undisturbed portion of the AC input voltage allows the controller to utilize a voltage value that accurately represents a voltage value of a supply voltage to the dimmer. Additionally, in at least one embodiment, the controller senses the voltage value when the phase cut voltage is relatively undisturbed and when the switching power converter has received sufficient power to meet power demands by a load. In at least one embodiment, after the controller senses the voltage value, the controller causes the electronic system to dissipate excess energy as, for example, (i) described in U.S. patent application Ser. No. 13/289,845, filed Nov. 4, 2011, entitled “Controlled Power Dissipation in a Switch Path in a Lighting System”, and inventors John L. Melanson and Eric J. King, (ii) U.S. patent application Ser. No. 13/289,931, filed Nov. 4, 2011, entitled “Controlled Power Dissipation in a Lighting System”, and inventors John L. Melanson and Eric J. King, (iii) Ser. No. 13/289,967 filed Nov. 4, 2011, entitled “Controlled Power Dissipation in a Link Path in a Lighting System”, and inventors John L. Melanson and Eric J. King, and/or (iv) dissipate power through another power dissipation circuit. The U.S. patent applications identified in (i), (ii), and (iii) are collectively referred to as the “Power Dissipation Applications”.
  • FIG. 3 depicts an electronic system 300 that includes a zero crossing calculator 302 of controller 304 that determines the approximate zero crossing of the rectified input voltage VφR — IN. FIG. 4 depicts exemplary voltage waveforms 400 of the input supply voltage VIN and rectified input voltage VφR — IN. Referring to FIGS. 3 and 4, dimmer 306 is a leading edge, phase cut dimmer. Dimmer 306 can be any type of leading edge, phase cut dimmer including a triac-based dimmer or a field effect transistor (FET) based dimmer. One cycle of the input supply voltage VIN is depicted in FIG. 4. The rectified input voltage VφR — IN depicts two cycles, cycle A and cycle B, which are derived from the cycle 401 of the input supply voltage VIN. Cycle A is a phase cut version of the first half cycle 402 of the input supply voltage VIN, and cycle B is a rectified, phase cut version of the second half cycle 404 of the input supply voltage VIN. Cycle A occurs from time t0 until the zero crossing of the input supply voltage VIN at time t3. Cycle B occurs from time t3 until the next zero crossing at time t6 of the input supply voltage VIN. In at least one embodiment, the dimmer 306 phase cuts the input supply voltage VIN from voltage supply 104 to generate the phase cut input voltage Vφ — IN. The full-bridge diode rectifier rectifies the phase cut input voltage Vφ — IN to generate the rectified input voltage VφR — IN. Between times t0 and t1, the dimmer 306 does not conduct current from the voltage supply 104 and, thus, phase cuts the supply voltage VIN until time t1. At time t1, the dimmer 306 conducts so that the rectified input voltage VφR — IN equals the input voltage VIN. At time t2, the controller 304 senses the voltage value v(0)A of rectified input voltage VφR — IN during cycle A of the rectified input voltage VφR — IN. The place and manner of sensing a voltage value of the rectified input voltage VφR — IN that represents a voltage value of the input supply voltage VIN is a matter of design choice. In at least one embodiment, the controller 304 senses the voltage value v(0)A downstream (right side) of the rectifier 106 from the phase-cut input voltage Vφ — IN. In at least one embodiment, the controller 304 senses the voltage value v(0)A upstream (left side) of the rectifier 106 from the phase-cut input voltage Vφ — IN.
  • The controller 304 includes a zero crossing calculator 302 to determine an approximate zero crossing of the rectified input voltage VφR — IN. For cycle A of the rectified input voltage VφR — IN, the zero crossing occurs at time t3. The particular implementation of the zero crossing calculator 302 is a matter of design choice. Various exemplary embodiments of the zero crossing calculator 302 are subsequently described in more detail. In at least one embodiment, the time t2 is selected as the time to sense the voltage value v(0)A because the power converter 308 has received sufficient power from voltage supply 104 to maintain an approximately constant link voltage VLINK and meet power demands of load 310. In at least one embodiment, at time t2 the controller 304 maintains the rectified input voltage VφR — IN at the voltage value v(0)A until reaching the zero crossing at time t3. Maintaining the voltage of the rectified input voltage VφR — IN effectively stops the current flow iIN into the power converter 308. At the zero crossing, in at least one embodiment, the controller 304 causes the rectified input voltage VφR — IN to rapidly decrease to approximately 0 volts. How to dissipate the energy associated with decreasing the rectified input voltage VφR — IN to approximately 0 volts is also a matter of design choice. In at least one embodiment, the power is dissipated as, for example, described in any or all of the Power Dissipation Applications and/or through another power dissipation circuit.
  • Controller 304 continues to determine the approximate zero crossing of the rectified input voltage VφR — IN in subsequent cycles of the rectified input voltage VφR — IN as, for example, shown in cycle B. In cycle B of the rectified input voltage VφR — IN, the phase cut dimmer 306 phase cuts the rectified input voltage VφR — IN from time t3 until time t4. At time t5, the controller 304 senses the voltage value v(0)B of the rectified input voltage VφR — IN at time t5. The zero crossing calculator 302 then determines the approximate zero crossing time that occurs at time t6 as previously described. In at least one embodiment, the actual zero crossings of the rectified input voltage VφR — IN are not directly observable by the controller 304, and, thus, are not actually detectable by the controller 304. In at least one embodiment, for a 120 Hz rectified input voltage VφR — IN, the sensing times of the rectified input voltage VφR — IN, such as times t2 and t5, are within a range 0.5-5 ms of the approximate zero crossing time, such as respective times t3 and t6. The sensing times of the rectified input voltage VφR — IN, such as times t2 and t5, are within a range of 0.25-5 ms of the approximate zero crossing time, such as respective times t3 and t6. In at least one embodiment, the range sensing times are inversely linearly related with respect to the frequency of the rectified input voltage VφR — IN, e.g. for a 240 Hz rectified input voltage VφR — IN, the sensing times are reduced by 50%. In at least one embodiment, the sensed voltage, such as sensed voltage v(0)A and v(0)B, are greater than or equal to 50V or, in at least one embodiment, greater than 0.3 times an RMS peak value of the rectified input voltage VφR — IN.
  • The controller 304 controls the power converter 308. The particular type of power converter 308 is a matter of design choice. For example, the power converter 308 can be a boost-type switching power converter such as switching power converter 108, a buck type switching power converter, a boost-buck type switching power converter, or a Cúk type switching power converter. In at least one embodiment, controller 304 controls the power converter 308 as described in, for example, U.S. patent application Ser. No. 11/967,269, entitled “Power Control System Using a Nonlinear Delta-Sigma Modulator With Nonlinear Power Conversion Process Modeling”, filed on Dec. 31, 2007, inventor John L. Melanson (referred to herein as “Melanson II”), U.S. patent application Ser. No. 11/967,275, entitled “Programmable Power Control System”, filed on Dec. 31, 2007, and inventor John L. Melanson (referred to herein as “Melanson III”), U.S. patent application Ser. No. 12/495,457, entitled “Cascode Configured Switching Using at Least One Low Breakdown Voltage Internal, Integrated Circuit Switch to Control At Least One High Breakdown Voltage External Switch”, filed on Jun. 30, 2009 (“referred to herein as “Melanson IV”), and inventor John L. Melanson, and U.S. patent application Ser. No. 12/174,404, entitled “Constant Current Controller With Selectable Gain”, filing date Jun. 30, 2011, and inventors John L. Melanson, Rahul Singh, and Siddharth Maru (referred to herein as “Melanson V”), which are all incorporated by reference in their entireties.
  • FIG. 5 depicts a controller 500, which represents one embodiment of controller 304. Controller 500 includes zero crossing calculator 502, which represents one embodiment of the zero crossing calculator 302. FIG. 6 depicts an approximate zero crossing determination process 600, which represents one embodiment of an approximate zero crossing determination process utilized by the controller 500 to determine an approximate zero crossing of the rectified input voltage VφR — IN.
  • Referring to FIGS. 5 and 6, in operation 602, the controller 500 observes the rectified input voltage VφR — IN. In operation 604, the controller selects a value v(0) of the rectified input voltage VφR — IN and stores the selected value v(0) in a register 504. The zero crossing calculator 500 includes a memory 506 that stores approximate zero crossing determination code 508. The processor 510 communicates with the memory 506 and, in operation 606, executes the approximate zero crossing determination code 508 to synthesize the rectified input voltage VφR — IN and determine an approximate zero crossing of the synthesized rectified input voltage VφR — IN. The processor 510 utilizes the sensed voltage value V(0) as an initial value to determine an approximate zero crossing of the rectified input voltage VφR — IN. The particular implementation of the approximate zero crossing determination code 508 is a matter of design choice. In at least one embodiment, the approximate zero crossing determination code 508 implements a parabolic function as, for example, subsequently describedError! Reference source not found. In other embodiments, the approximate zero crossing determination code 508 implements other polynomial functions or trigonometric functions, such as an actual sine function, to synthesize the rectified input voltage VφR — IN to determine zero crossings of the rectified input voltage VφR — IN. In at least one embodiment, the approximate zero crossing determination code 508 includes code to access a look up table, and the look up table includes zero crossing times corresponding to possible values of the sensed voltage value V(0).
  • FIGS. 7A and 7B (collectively referred to as “FIG. 7”) collectively depict zero crossing code 700, which represents one embodiment of the zero crossing code 508. The zero crossing code 700 begins in FIG. 7A and continues in FIG. 7B. The zero crossing code 700 implements an iterative approximate zero crossing determination process using a parabolic function based on modeling changes in voltage across a capacitor resulting from current flow from the capacitor. FIG. 8 depicts an exemplary capacitor/current model 800 utilized by the zero crossing code 700. The capacitor/current model 800 models the input supply voltage VIN using a parabolic function to synthesize the input supply voltage VIN.
  • Referring to FIGS. 7 and 8, the zero crossing code 700 utilizes an initial sample V(0) of the rectified input voltage VφR — IN. The capacitor/current model 800 models the sampling with a switch 802 that momentarily closes at a time tSENSE and then immediately opens. The time tSENSE at which the rectified input voltage VφR — IN is sensed is a matter of design choice and, in at least one embodiment, is selected on a cycle-by-cycle basis of the rectified input voltage VφR — IN. In at least one embodiment, the sensing time t0 occurs when the voltage supply 104 (FIG. 3) has supplied sufficient power to meet the power demand of load 310 (FIG. 3). Thus, in at least one embodiment, the time tSENSE corresponds to time t2 for cycle A and time t5 for cycle B of the rectified input voltage VφR — IN as depicted in FIG. 4. In at least one embodiment, for a 120 Hz rectified input voltage VφR — IN, the time tSENSE is within 0.5-5 ms of the zero crossing time tZC. In at least one embodiment, for a 120 Hz rectified input voltage VφR — IN, the time tSENSE is within 0.25-5 ms of the zero crossing time tzc.
  • The voltage-current graph 804 models a current I(V(n)) as a function of the voltage V(n) across the capacitor 806, i.e. I(V(n))=f(V(n)). The voltage-current graph 804 presents a linear relationship between the voltage V(n) across the capacitor 806 and the current I(V(n)) as, for example, represented by Equation [1]:

  • I(V(n))=m·V(n)+b  [1]
  • “V(n)” represents the voltage across capacitor 806, which represents the input supply voltage VIN. “I(V(n))” represents the current discharged from capacitor 806 and is modeled by a varying current source 810. “m” is the slope of the V(n)/I(V(n)) relationship line 808, and “b” is the y-intercept of the V(n)/I(V(n)) relationship line 808. The values of “m” and “b” are a matter of design choice and are, in at least one embodiment, chosen to best approximate an actual relationship between the input supply voltage VIN and the input current iIN (FIG. 3) for the modeled capacitor 806.
  • The change in voltage V(n) with respect to time is represented by Equation [2]:

  • dV(n)/dt=I(V)/C  [2].
  • “dV(n)/dt” represents the change in voltage with respect to time, and “C” represents a capacitance of capacitor 806. In at least one embodiment, for ease of calculation, C=1. “dt” represents a change in time, and, in at least one embodiment, is a constant time period at which the zero crossing indicator value ZC is updated. The particular value of dt is a matter of design choice. In at least one embodiment, the value of dt is chosen based on a desired accuracy of the zero crossing indicator value ZC. In at least one embodiment, the zero crossing value ZC is updated at a frequency fCALC, and fCALC is at least 10 kHz. Thus, dt, which equals 1/fCALC, is less than or equal to 0.0001 secs.
  • Equation [2] can be rearranged as Equation [3]:

  • dV(n)=(I(V)/C)·dt  [3].
  • Each subsequent value V(n+1) is related to the immediately preceding voltage value V(n) by Equation [4]:

  • V(n+1)=V(n)−dV(n)  [4].
  • The initial value V(0) is provided by an actual sensed value of rectified input voltage VφR — IN at time tSENSE. The voltage-current graph 804 provides a value of I(V) for each sample or calculation of V(n), and the value of each dV(n) for each increment of dt can, thus, be determined from Equation [3] and the value of I(V) from the voltage-current graph 804. Since the relationship between V(n) and I(V) is linear in voltage-current graph 804, the combination of Equations [1]-[4] result in a parabolic function, and the values of V(n+1) will decrease in accordance with the parabolic function of Equation [4]. The input supply voltage VIN is, in at least one embodiment, a sine wave. The parabolic function of Equation [4] is relatively fast and easy to calculate and closely models a sine wave.
  • Comparator 812 compares the voltage value V(n) with a reference value VZC — REF, and the zero crossing value ZC represents the result of the comparison. The reference value VZC — REF is chosen so that when the voltage value V(n) is less than the reference value VZC — REF, the zero crossing value ZC changes state from a logical 0 to a logical 1 to indicate a zero crossing of the input supply voltage VIN. In at least one embodiment, when the zero crossing of the input supply voltage VIN has been reached, the controller 304 (FIG. 3) transitions to hold or “glue” the rectified input voltage VφR — IN at a low value to prevent the phase cut dimmer 306 from prematurely firing during the next cycle of the input supply voltage VIN as, for example, described in Melanson I.
  • Equation [5] represents an approximation equation that can be used to iteratively determine an approximate zero crossing of the AC input supply voltage VIN:

  • V APPROX(i)=V APPROX(i−1)−[k 1 −V APPROX [i−1])·k 2  [5].
  • “VAPPROX(i)” is the ith approximate zero crossing of the AC input supply voltage VIN, “i” is an integer index, “VAPPROX(i−1)” is the approximate zero crossing value that immediately precedes the value of VAPPROX(i). “k1” and “k2” are scaling factors. k1 relates to the peak value of the input supply voltage and k2 relates to the frequency and, thus, the step-size of each iteration of Equation [5].
  • For a 120 Vrms, 60 Hz input supply voltage, k1=220, k2=0.38, and starting at a 112 degree phase cut, the values of VAPPROX(i) rounded to the nearest volt for Equation [5] are:
      • {157, 155, 152, 150, 147, 145, 142, 139, 136, 132, 129, 126, 122, 118, 114, 110, 106, 102, 97, 93, 88, 83, 78, 72, 67, 61, 55, 49, 42, 35, 28, 21, 13, 5, 0}
        The actual sine wave values are:
      • {157, 155, 153, 150, 147, 144, 141, 137, 134, 130, 126, 122, 118, 114, 109, 104, 100, 95, 90, 85, 80, 74, 69, 64, 58, 52, 47, 41, 35, 29, 24, 18, 12, 6, 0}
        For a 120 Vrms, 60 Hz input supply voltage, k1=220, k2=0.38, and starting at a 130 degree phase cut, the values of VAPPROX(i) rounded to the nearest volt for Equation [5] are:
      • {130, 127, 123, 119, 116, 112, 107, 103, 99, 94, 89, 84, 79, 74, 68, 63, 57, 50, 44, 37, 30, 23, 16, 8, 0, −9}
        The actual sine wave values are:
      • {130, 126, 122, 118, 114, 109, 104, 100, 95, 90, 85, 80, 74, 69, 64, 58, 52, 47, 41, 35, 29, 24, 18, 12, 6, 0}
  • Thus, there is a slight error between the actual zero crossing as indicated by the sine wave data and the determined approximate zero crossing determined using Equation [5]. However, the approximation of the zero crossing is sufficient to allow the controller 506 to accurately control the power converter 308 and maintain compatibility with the phase-cut dimmer 306. More accurate functions can be used if higher accuracy is desired.
  • FIG. 9 depicts an electronic system 900, which represents one embodiment of the electronic system 300. Controller 902 senses the rectified input voltage VφR — IN at node 124. Controller 902 can sense the rectified input voltage VφR — IN in any desired manner, such as through a resistor divider circuit 905. Zero crossing calculator 904, which represents one embodiment of the zero crossing calculator 302 (FIG. 3), synthesizes the rectified input voltage VφR — IN as, for example, previously described in conjunction with zero crossing calculator 302, 500 (FIG. 5), or 700 (FIG. 7). In at least one embodiment, controller 902 controls the switching power converter as, for example, Melanson II, Melanson III, Melanson IV, or Melanson V.
  • Thus, a controller senses a leading edge, phase cut alternating current (AC) input voltage to a switching power converter at least at a time t0 during a cycle of the AC input voltage. The time t0 is prior to a zero crossing of the AC input voltage and at least some zero crossings of the AC input voltage are not directly observable by the controller. The controller is configured to determine an approximate zero crossing of the AC input voltage based on a voltage value of the AC input voltage at time t0.
  • Although embodiments have been described in detail, it should be understood that various changes, substitutions, and alterations can be made hereto without departing from the spirit and scope of the invention as defined by the appended claims.

Claims (2)

1. An apparatus comprising:
a controller having an input to sense a leading edge, phase cut alternating current (AC) input voltage to a switching power converter at least at a first time during a cycle of the AC input voltage, wherein the cycle of the AC input voltage is derived from a cycle of the AC input supply voltage, the first time is prior to an approximate zero crossing of the cycle of the AC input supply voltage, and at least some zero crossings of the AC input supply voltage are not directly observable by the controller, and the controller is configured to:
determine the approximate zero crossing of the AC input supply voltage based on a voltage value of the phase cut AC input voltage sensed at the first time.
2.-31. (canceled)
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