TWI449321B - High frequency power source - Google Patents

High frequency power source Download PDF

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TWI449321B
TWI449321B TW099124111A TW99124111A TWI449321B TW I449321 B TWI449321 B TW I449321B TW 099124111 A TW099124111 A TW 099124111A TW 99124111 A TW99124111 A TW 99124111A TW I449321 B TWI449321 B TW I449321B
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inverter
charge recovery
switching element
power supply
reactor
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TW099124111A
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Chinese (zh)
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TW201112604A (en
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Hiroyasu Iwabuki
Hirohisa Kuwano
Hiroshi Kurushima
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Mitsubishi Electric Corp
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4811—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode having auxiliary actively switched resonant commutation circuits connected to intermediate DC voltage or between two push-pull branches
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01S—DEVICES USING THE PROCESS OF LIGHT AMPLIFICATION BY STIMULATED EMISSION OF RADIATION [LASER] TO AMPLIFY OR GENERATE LIGHT; DEVICES USING STIMULATED EMISSION OF ELECTROMAGNETIC RADIATION IN WAVE RANGES OTHER THAN OPTICAL
    • H01S3/00—Lasers, i.e. devices using stimulated emission of electromagnetic radiation in the infrared, visible or ultraviolet wave range
    • H01S3/09—Processes or apparatus for excitation, e.g. pumping
    • H01S3/097—Processes or apparatus for excitation, e.g. pumping by gas discharge of a gas laser
    • H01S3/09702—Details of the driver electronics and electric discharge circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/342—Active non-dissipative snubbers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Lasers (AREA)

Description

高頻率電源裝置High frequency power supply unit

本發明係有關一種輸出電壓脈波(pulse)的高頻率電源裝置。The present invention relates to a high frequency power supply device that outputs a voltage pulse.

就輸出電壓脈波的高頻率電源裝置而言，有在利用例如交流放電進行雷射振盪時所使用的高頻率電源裝置。近年，就以印刷基板開孔加工為代表的微細加工用的雷射裝置而言，雷射脈波輸出的脈波寬度為1微秒(micro second)至數百微秒、能夠進行脈波振盪的裝置之需求正增加中。For a high-frequency power supply device that outputs a voltage pulse wave, there is a high-frequency power supply device used for performing laser oscillation using, for example, an AC discharge. In recent years, in the laser device for microfabrication represented by the processing of the printed circuit board, the pulse width of the laser pulse output is from 1 microsecond to hundreds of microseconds, and pulse wave oscillation can be performed. The demand for devices is increasing.

在上述的雷射裝置(例如，二氧化碳雷射裝置)所使用的高頻率電源裝置中，為了激發雷射介質，施加至雷射振盪器的放電產生用電極的放電電壓必須為數kV(千伏特)，放電時流通的放電電流的峰值(peak)必須為數十A(安培)，放電時的交流頻率(放電頻率)必須為數百kHz(千赫茲)至數MHz(十萬赫茲)以上。以下，將用以使放電以上述條件產生的電力稱為放電電力。In the high-frequency power supply device used in the above-described laser device (for example, a carbon dioxide laser device), in order to excite the laser medium, the discharge voltage applied to the discharge generating electrode of the laser oscillator must be several kV (kV) The peak of the discharge current flowing during discharge must be several tens of ampere (amperes), and the alternating current frequency (discharge frequency) at the time of discharge must be several hundred kHz (kilohertz) to several MHz (hundred thousand Hz). Hereinafter, the electric power for causing the discharge to be generated under the above conditions is referred to as discharge electric power.

在習知技術中，為了將如上述的放電電力供給給雷射裝置，係使用使用有反相器(inverter，有多種名稱，如反相器、反向器、變換器、變頻器、反相器等等。本文中稱為反相器)的高頻率電源裝置。藉由對反相器的開關(switching)元件進行導通(on)/切斷(off)控制，從而將大致正弦波的電流供給給雷射振盪器，而將電力供給至充滿雷射介質的放電空間而產生放電。接著，因該放電而被激發的雷射介質進行雷射振盪形成雷射脈波並輸出(參照例如下述的專利文獻1)。In the prior art, in order to supply the discharge power as described above to the laser device, an inverter is used, which has various names such as an inverter, an inverter, a converter, a frequency converter, and an inverter. High-frequency power supply devices, etc., referred to herein as inverters. By supplying on/off control to the switching element of the inverter, a substantially sinusoidal current is supplied to the laser oscillator, and power is supplied to the discharge filled with the laser medium. The space generates a discharge. Then, the laser medium excited by the discharge is subjected to laser oscillation to form a laser pulse wave and output (see, for example, Patent Document 1 below).

而在印刷基板等微細加工用的雷射裝置中，依加工物的材料與加工方法，存在有雷射的最佳能量值。因此，設定各種控制參數用來控制雷射振盪器的雷射輸出。On the other hand, in a laser device for microfabrication such as a printed circuit board, depending on the material and processing method of the workpiece, there is an optimum energy value of the laser. Therefore, various control parameters are set to control the laser output of the laser oscillator.

就上述的控制參數而言，有例如第20圖所示，(i)代表雷射脈波的放電電力的峰值之峰值輸出、(ii)代表雷射脈波的輸出頻率的重複脈波頻率、及(iii)代表雷射脈波的脈波寬度等。For the above control parameters, for example, as shown in Fig. 20, (i) peak output of the peak value of the discharge power representing the laser pulse wave, (ii) the repetition pulse frequency representing the output frequency of the laser pulse wave, And (iii) represents the pulse width of the laser pulse wave and the like.

為了獲得適合於上述控制參數的雷射脈波，在高頻率電源裝置中例如進行如下述的控制。In order to obtain a laser pulse wave suitable for the above control parameters, for example, control is performed in the high frequency power supply device.

(i)雷射脈波的放電電力的峰值輸出主要是根據施加至雷射振盪器的電極的放電電壓及因電極間產生的放電而流通的放電電流的峰值而所決定。因此，為了控制雷射脈波的放電電力的峰值輸出，採用調整針對反相器的直流電源的電壓而控制放電電壓、或者藉由以PWM(Pulse Width Modulation；脈波寬度調變)控制反相器的工作週期(duty)，而控制放電電流的峰值之方法。例如，如第21圖所示，在第20圖的條件1、2中，當峰值輸出為p1>p2時，反相器的直流電源的電壓被設定為E1>E2。(i) The peak output of the discharge power of the laser pulse wave is mainly determined based on the discharge voltage applied to the electrode of the laser oscillator and the peak value of the discharge current flowing through the discharge generated between the electrodes. Therefore, in order to control the peak output of the discharge power of the laser pulse wave, the discharge voltage is controlled by adjusting the voltage of the DC power source for the inverter, or by inverting by PWM (Pulse Width Modulation). The duty cycle of the device, and the method of controlling the peak value of the discharge current. For example, as shown in Fig. 21, in the conditions 1 and 2 of Fig. 20, when the peak output is p1 > p2, the voltage of the DC power source of the inverter is set to E1 > E2.

(ii)重複脈波頻率乃是相當於每單位時間照射的雷射脈波數的雷射脈波的輸出頻率。藉由增加此重複脈波頻率，便能夠使微細化加工時的加工速度(加工次數)提升。並且，為了控制此重複脈波頻率，係控制對反相器的開關元件進行導通/切斷控制的一連串控制脈波群的每單位時間的個數。例如，如第21圖所示，在第20圖的條件1、2中，當重複脈波頻率為f1>f2時，對反相器的開關元件進行導通/切斷控制的一連串控制脈波群的每單位時間T的個數被設定為N1>N2。在第21圖中係設定N1=3，N2=2。(ii) The repetition pulse frequency is an output frequency of a laser pulse corresponding to the number of laser pulses per unit time. By increasing the frequency of the repeated pulse wave, the processing speed (the number of times of processing) during the microfabrication processing can be improved. Further, in order to control the frequency of the repeated pulse wave, the number of per unit time of a series of control pulse wave groups for controlling the on/off control of the switching elements of the inverter is controlled. For example, as shown in FIG. 21, in the conditions 1 and 2 of FIG. 20, when the repetition pulse frequency is f1>f2, a series of control pulse groups for performing on/off control of the switching elements of the inverter are performed. The number of T per unit time is set to N1>N2. In Fig. 21, N1 = 3 and N2 = 2 are set.

(iii)雷射脈波的脈波寬度係比例於對反相器的開關元件進行導通/切斷控制的一連串控制脈波群的個數，亦即比例於開關元件的開關次數n。因此，為了控制雷射的脈波寬度，控制開關元件的開關次數n。例如，如第21圖所示，在第20圖的條件1、2中，當脈波寬度為t1<t2時，開關元件的開關次數被設定為n1<n2。在第21圖中，2.n1=n2，將一連串的開關次數設定為2倍(2n)，藉此能夠將雷射脈波寬度設定為2倍。(iii) The pulse width of the laser pulse is proportional to the number of series of control pulse groups for controlling the on/off of the switching elements of the inverter, that is, the number n of switching times of the switching elements. Therefore, in order to control the pulse width of the laser, the number of switching n of the switching element is controlled. For example, as shown in Fig. 21, in the conditions 1 and 2 of Fig. 20, when the pulse width is t1 < t2, the number of switching of the switching elements is set to n1 < n2. In Fig. 21, 2.n1 = n2, and the number of switching times is set to 2 times (2n), whereby the laser pulse width can be set to 2 times.

而近年來，隨著IT(Information Technology；資訊科技)相關機器的普及，要求微細加工用的雷射裝置要有更高生產性。此外，微細化加工中的加工材料與加工種類也走向多樣化。因此，為了提高加工速度(加工次數)、或者為了輸入適合於加工材料與加工種類的加工能量，必須設定適當的控制參數。In recent years, with the popularization of IT (Information Technology) related equipment, laser devices for microfabrication are required to be more productive. In addition, the processing materials and processing types in the miniaturization process are also diversified. Therefore, in order to increase the processing speed (the number of times of processing) or to input the processing energy suitable for the material to be processed and the type of processing, it is necessary to set appropriate control parameters.

此時，為了提高加工速度(加工次數)，如前述，必須設定控制參數使雷射振盪的重複脈波頻率增加。此外，為了輸入適合於加工材料與加工種類的加工能量，係要求設定控制參數使雷射的脈波寬度大幅地變化(例如，從1微秒以下變化至數百微秒)。舉聚醯亞胺(polyimide)系樹脂的情形為例，採用峰值輸出高但雷射照射時間短(亦即脈波寬度小)的方式較能進行品質佳的加工。又，在含有玻璃纖維的玻璃環氧(glass fiber filled epoxy)材的情形中，採用峰值輸出小但雷射照射時間比較長(亦即脈波寬度大)的方式較能進行品質佳的加工。At this time, in order to increase the processing speed (the number of times of processing), as described above, it is necessary to set the control parameter to increase the frequency of the repeated pulse wave of the laser oscillation. In addition, in order to input processing energy suitable for the material to be processed and the type of processing, it is required to set the control parameters so that the pulse width of the laser greatly changes (for example, from 1 microsecond or less to hundreds of microseconds). In the case of a polyimide resin, for example, a method in which the peak output is high but the laser irradiation time is short (that is, the pulse width is small) can be processed with higher quality. Further, in the case of a glass fiber-filled epoxy material containing glass fibers, a method in which the peak output is small but the laser irradiation time is relatively long (that is, the pulse width is large) is more preferable.

如上述，考慮生產性與加工材質特性，而使屬於控制參數的脈波雷射的重複脈波頻率數增加、或使雷射的脈波寬度變化乃係重要的。然而，若使脈波雷射的重複脈波頻率增加、或使雷射的脈波寬度增加，則如上述必然增加開關元件的每單位時間的開關次數，結果，開關元件的開關損失亦增加，導致高頻率電源裝置的發熱增大。As described above, it is important to increase the number of repeated pulse wave frequencies of the pulse laser which is a control parameter or to change the pulse width of the laser in consideration of productivity and processing material characteristics. However, if the frequency of the repetitive pulse wave of the pulse laser is increased or the pulse width of the laser is increased, the number of switching per unit time of the switching element is inevitably increased as described above, and as a result, the switching loss of the switching element is also increased. This causes an increase in heat generation of the high frequency power supply unit.

在習知技術中，就用以一邊抑制反相器的開關損失一邊使雷射的脈波寬度增加的對策而言，已提案一種手法，係將對反相器的開關元件進行導通/切斷控制的一連串控制脈波有規則地進行疏緩，藉此一邊確保所期望的雷射脈波寬度一邊相對地降低開關元件的開關次數，從而抑制開關損失(參照例如下述的專利文獻2)。In the conventional technique, in order to suppress the switching loss of the inverter while increasing the pulse width of the laser, a technique has been proposed in which the switching element of the inverter is turned on/off. The series of control pulse waves to be controlled is regularly reduced, and the number of switching of the switching elements is relatively reduced while ensuring the desired laser pulse width, thereby suppressing the switching loss (see, for example, Patent Document 2 below).

專利文獻1：日本特開2000-4059號公報Patent Document 1: Japanese Patent Laid-Open Publication No. 2000-4059

專利文獻2：日本特開2004-22696號公報Patent Document 2: Japanese Patent Laid-Open Publication No. 2004-22696

如上述，考慮生產性與加工材質特性，而使屬於控制參數的脈波雷射的重複脈波頻率數增加、或使脈波寬度增加，則必然增加開關元件的每單位時間的開關次數。結果，開關元件的開關損失亦增加，導致高頻率電源裝置的發熱增大。As described above, considering the productivity and the material properties of the processing, the number of repetition pulse waves of the pulse laser which belongs to the control parameter is increased, or the pulse width is increased, and the number of switching per unit time of the switching element is inevitably increased. As a result, the switching loss of the switching element also increases, resulting in an increase in heat generation of the high frequency power supply device.

此種伴隨著反相器的開關元件之導通/切斷而有的開關損失係下述損失的總和：於開關動作時開關元件自身之施加在通道(channel)的電壓與電流之積所產生的切通損失(turn-on power loss)及切斷損失(turn-off power loss)、以及於開關元件切通時，開關元件的雜散電容(stray capacity)的充電電荷通過開關元件的通道進行放電時由雜散電容產生的充放電損失。Such a switching loss accompanying the on/off of the switching elements of the inverter is the sum of the following losses: the product of the voltage and current applied to the channel by the switching element itself during the switching operation. Turn-on power loss and turn-off power loss, and when the switching element is cut, the charging charge of the stray capacity of the switching element is discharged through the channel of the switching element Charge and discharge losses due to stray capacitance.

上述中，雜散電容所產生的充放電損失W如下式(1)所式。In the above, the charge and discharge loss W generated by the stray capacitance is as follows by the formula (1).

W=(1/2)CV2 f　(1)W=(1/2)CV 2 f (1)

其中，C為雜散電容，V為雜散電容的充電電壓，f為雜散電容的放電造成的損失發生時的開關元件的開關次數。該雜散電容C乃係包含下述電容等的雜散電容整體：寄生在各開關元件的汲極-源極間的雜散電容、雜散於閘極供電線而等效地產生在開關元件的汲極-源極間的雜散電容、及為了使各開關元件的串聯分壓均等化而外接的電容器。Where C is the stray capacitance, V is the charging voltage of the stray capacitance, and f is the number of switching times of the switching element when the loss due to the discharge of the stray capacitance occurs. The stray capacitance C is a stray capacitance including a capacitor or the like as follows: a stray capacitance which is parasitic between the drain and the source of each switching element, and a stray capacitance applied to the gate supply line is equivalently generated in the switching element The stray capacitance between the drain and the source and the capacitor externally connected to equalize the series voltage division of each switching element.

從上述式(1)可知，雜散電容產生的充放電損失W主要是起因於雜散電容充放電的放電損失。因此，在雜散電容的充電電壓高、開關元件的開關次數多的高頻率反相器電路中，雜散電容產生的充放電損失會明顯變大。因此，為了使反相器以高頻率動作，必須使充放電損失W的降低，亦即使雜散電容的充電電壓V降低或者使雜散電容本身降低。As is clear from the above formula (1), the charge and discharge loss W due to the stray capacitance is mainly caused by the discharge loss of the stray capacitance charge and discharge. Therefore, in the high-frequency inverter circuit in which the charging voltage of the stray capacitance is high and the number of switching of the switching elements is large, the charge and discharge loss due to the stray capacitance is remarkably large. Therefore, in order to operate the inverter at a high frequency, it is necessary to lower the charge and discharge loss W, and even if the charging voltage V of the stray capacitance is lowered or the stray capacitance itself is lowered.

在習知技術(專利文獻2)中，係將對反相器的開關元件進行導通/切斷控制的一連串控制脈波有規則地進行疏緩，藉此一邊確保所期望的雷射脈波寬度一邊相對地降低開關元件的開關次數，從而抑制電源裝置的發熱。然而，在將一連串的控制脈波有規則地進行疏緩(減少次數或產生脈波的頻度的動作，本文中簡稱為疏緩)時，由於反相器的停止/啟動之動作頻繁地重複，導致起因於雜散電容的充放電之開關元件的開關損失顯著地增加。結果，依然存在有電源裝置的發熱增大之問題。In the prior art (Patent Document 2), a series of control pulse waves for performing on/off control of the switching elements of the inverter are regularly delayed, thereby ensuring a desired laser pulse width. The number of times of switching of the switching elements is relatively reduced, thereby suppressing heat generation of the power supply device. However, when a series of control pulse waves are regularly slowed down (the number of times of reduction or the frequency of generating pulse waves, referred to herein as "slowdown"), since the stop/start action of the inverter is frequently repeated, The switching loss of the switching element resulting in charging and discharging due to stray capacitance is remarkably increased. As a result, there is still a problem that the heat generation of the power supply device increases.

亦即，在未將導通/切斷控制開關元件之一連串的控制脈波進行疏緩而連續地使開關元件導通/切斷動作之情形中，雜散電容的充電電荷係藉由再生動作而放電。然而，在將一連串的控制脈波有規則地進行疏緩時，由於反相器的停止/啟動之動作頻繁地重複，因此若是在反相器停止後雜散電容的充電電荷未充分放電的狀態下接著啟動反相器，則無法充分地滿足共振條件，反相器啟動瞬後會流通大電流而造成損失變大。因此，上述的疏緩對開關損失的降低效果幾乎不明顯。In other words, in a case where the control pulse wave of one of the on/off control switching elements is not slowed down and the switching element is continuously turned on/off, the charged electric charge of the stray capacitance is discharged by the regenerative action. . However, when a series of control pulse waves are regularly delayed, since the stop/start operation of the inverter is frequently repeated, if the charge of the stray capacitance is not sufficiently discharged after the inverter is stopped, When the inverter is started next, the resonance condition cannot be sufficiently satisfied, and a large current flows when the inverter is turned on, causing a loss to increase. Therefore, the above-mentioned relief effect on the switching loss is hardly noticeable.

此外，為了使加工速度(加工次數)提升而使雷射脈波的重複脈波頻率增加，是無法以將對開關元件進行導通/切斷控制的一連串控制脈波進行疏緩這樣的手法來處理。因此，開關元件的每單位時間的開關次數依然增加，結果，高頻率電源裝置的發熱增加。Further, in order to increase the processing pulse speed (the number of processing) and increase the repetition pulse frequency of the laser pulse wave, it is not possible to deal with a series of control pulse waves for turning on/off the switching element. . Therefore, the number of switching per unit time of the switching element is still increased, and as a result, the heat generation of the high frequency power supply device is increased.

本發明乃是為了解決上述課題而研創者，其目的在於提供一種高頻率電源裝置，係在以反相器的開關元件的導通/切斷控制來實現高頻率電力供給的高頻率電源裝置中，能夠大幅地降低因構成反相器的各開關元件的開關次數之增加所產生的開關損失。The present invention has been made in order to solve the above problems, and an object of the present invention is to provide a high-frequency power supply device in a high-frequency power supply device that realizes high-frequency power supply by on/off control of a switching element of an inverter. It is possible to greatly reduce the switching loss caused by the increase in the number of switching of the switching elements constituting the inverter.

本發明的高頻率電源裝置係輸出電壓脈波至負載的高頻率電源裝置，係具備：電力轉換用的反相器，具有開關元件；第1電抗器(reactor)，與上述負載連接；電荷回收電路，與上述反相器的輸出端連接，用以回收上述開關元件的雜散電容的電荷；及控制手段，控制上述反相器與上述電荷回收電路的動作；上述電荷回收電路係具有：開關元件，用以啟動該電荷回收電路；第2電抗器，用以在與上述雜散電容之間產生LC共振；及電容器，用以回收電荷。The high-frequency power supply device of the present invention is a high-frequency power supply device that outputs a voltage pulse to a load, and includes: an inverter for power conversion, having a switching element; a first reactor connected to the load; and charge recovery a circuit connected to the output end of the inverter to recover the charge of the stray capacitance of the switching element; and a control means for controlling the operation of the inverter and the charge recovery circuit; the charge recovery circuit has a switch a component for activating the charge recovery circuit; a second reactor for generating LC resonance between the stray capacitance; and a capacitor for recovering charge.

依據本發明，於必要的時序(timing)將構成電荷回收電路的開關元件導通/切斷，使該電路的第2電抗器與反相器的開關元件的雜散電容之間產生LC共振，藉此，能夠使充電至構成反相器的各開關元件的雜散電容的電荷輸出/輸入於電荷回收電路的電容器。藉此，能夠降低反相器的開關元件開關時的雜散電容的充電電壓，因此即使開關次數增加仍能夠大幅降低反相器的開關損失。藉此，高頻率電源裝置的冷卻機構得以簡化，能夠謀求裝置小型化、成本降低化、設置空間縮小化等。According to the present invention, the switching element constituting the charge recovery circuit is turned on/off at a necessary timing to cause LC resonance between the second reactor of the circuit and the stray capacitance of the switching element of the inverter. Thereby, the electric charge charged to the stray capacitance of each of the switching elements constituting the inverter can be output/inputted to the capacitor of the charge recovery circuit. Thereby, the charging voltage of the stray capacitance at the time of switching of the switching elements of the inverter can be reduced, so that even if the number of switching increases, the switching loss of the inverter can be greatly reduced. Thereby, the cooling mechanism of the high-frequency power supply device can be simplified, and the size of the device can be reduced, the cost can be reduced, and the installation space can be reduced.

實施形態1Embodiment 1

第1圖係顯示本發明實施形態1的高頻率電源裝置可適用的氣體雷射裝置的概要之構成圖，第2圖係本發明實施形態1的高頻率電源裝置的構成圖。1 is a view showing a schematic configuration of a gas laser device to which the high-frequency power supply device according to the first embodiment of the present invention is applied, and FIG. 2 is a configuration diagram of the high-frequency power supply device according to the first embodiment of the present invention.

本實施形態1的高頻率電源裝置1係連接至構成氣體雷射裝置的雷射振盪器2。如第1圖所示，雷射振盪器2具備：相對向配置的一對電極3、形成在各電極3的對向表面側的介電體4、及由用以進行雷射振盪的部分反射鏡(mirror)5與全反射鏡6所構成的共振鏡。於形成在介電體4間的放電空間7充滿雷射介質(混合氣體)。並且，供給電力給放電空間7而產生放電，藉由因該放電而被激發的雷射介質的受激發射，在鏡5、6間雷射振盪形成雷射光而輸出。此時，如第2圖所示，介電體4與放電空間7係形成高頻率電源裝置1的電容性負載19，其等效電路係以放電電阻器191與介電體電容器192的串聯電路表示。The high-frequency power supply device 1 of the first embodiment is connected to the laser oscillator 2 constituting the gas laser device. As shown in Fig. 1, the laser oscillator 2 includes a pair of electrodes 3 arranged to face each other, a dielectric body 4 formed on the opposite surface side of each electrode 3, and a partial reflection for performing laser oscillation. A mirror composed of a mirror 5 and a total reflection mirror 6. The discharge space 7 formed between the dielectric bodies 4 is filled with a laser medium (mixed gas). Then, electric power is supplied to the discharge space 7 to generate a discharge, and the stimulated emission of the laser medium excited by the discharge is outputted by laser light between the mirrors 5 and 6 to form laser light. At this time, as shown in FIG. 2, the dielectric body 4 and the discharge space 7 form a capacitive load 19 of the high-frequency power supply device 1, and the equivalent circuit thereof is a series circuit of the discharge resistor 191 and the dielectric capacitor 192. Said.

如第2圖所示，高頻率電源裝置1的主體構成具有：全橋(full bridge)型的反相器11、直流電源12、升壓變壓器13、第1電抗器14、反相器控制電路15、疏緩電路16、脈波控制電路17及電荷回收電路18。並且，高頻率電源裝置1中，反相器11的輸出側係經由升壓變壓器13及第1電抗器14而連接至成為電容性負載19之氣體雷射裝置的放電電阻器191與介電體電容器192的串聯電路。As shown in FIG. 2, the main body of the high-frequency power supply device 1 has a full bridge type inverter 11, a DC power supply 12, a step-up transformer 13, a first reactor 14, and an inverter control circuit. 15. The mitigation circuit 16, the pulse wave control circuit 17, and the charge recovery circuit 18. Further, in the high-frequency power supply device 1, the output side of the inverter 11 is connected to the discharge resistor 191 and the dielectric body of the gas laser device serving as the capacitive load 19 via the step-up transformer 13 and the first reactor 14. A series circuit of capacitors 192.

另外，上述第1電抗器14常會利用高頻率變壓器的漏電感，並非一定要是安裝於外部者。此外，在反相器11能夠輸出高電壓的情形中，亦可不使用升壓變壓器13而直接經由第1電抗器14連接至負載19。此外，該高頻率電源裝置1並非僅適用於如第1圖所示構成的氣體雷射裝置，亦可適用於如下述用途等之具有以高頻率交流電力予以驅動的負載的所有裝置：半導體製造裝置、液晶製造裝置、放電加工裝置、臭氧產生裝置及微波產生裝置等的電漿產生用途；板金加工用與印刷基板開孔用的雷射裝置等的雷射產生用途；熱可塑性樹脂等熔著用的高頻率熔著機、金屬加工用的感應加熱壓入機及烹調用的IH電磁爐(Induction Heating cooking heater；感應加熱調理器)等的高頻率感應過熱用途。Further, the first reactor 14 often uses the leakage inductance of the high-frequency transformer, and is not necessarily required to be attached to the outside. Further, in the case where the inverter 11 can output a high voltage, it is also possible to directly connect to the load 19 via the first reactor 14 without using the step-up transformer 13. Further, the high-frequency power supply device 1 is not only applicable to the gas laser device configured as shown in Fig. 1, but may be applied to all devices having a load driven by high-frequency alternating current power, such as the following: semiconductor manufacturing Plasma generating applications such as a device, a liquid crystal manufacturing device, an electric discharge machining device, an ozone generating device, and a microwave generating device; a laser generating device for sheet metal processing and a laser device for opening a printing substrate; and a thermoplastic resin or the like High frequency induction superheating applications such as a high frequency melting machine, an induction heating press machine for metal processing, and an induction heating cooking heater (Induction Heating cooking heater).

上述反相器11具有：由MOSFET(Metal Oxide Semiconductor Field Effect Transistor；金屬氧化物半導體場效應電晶體)與IGBT(Insulated Gate Bipolar Transistor；絕緣閘雙極性電晶體)等所構成的各開關元件S1至S4、以及個別驅動各開關元件S1至S4的閘極驅動電路111至114。此外，於各開關元件S1至S4，並聯連接有各二極體D11至D14及雜散電容C11至C14。The inverter 11 has a switching element S1 composed of a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) and an IGBT (Insulated Gate Bipolar Transistor). S4, and gate drive circuits 111 to 114 that individually drive the respective switching elements S1 to S4. Further, in each of the switching elements S1 to S4, the respective diodes D11 to D14 and the stray capacitances C11 to C14 are connected in parallel.

另外，各二極體D11至D14乃係反相器11進行再生與回流時所使用者，當各開關元件S1至S4的內部有寄生二極體時可以省略。此外，在此的各雜散電容C11至C14係意指包含下述電容等的雜散電容整體：寄生在各開關元件S1至S4的汲極-源極間的雜散電容、雜散於閘極供電線而等效地產生在開關元件的汲極-源極間的雜散電容、及為了使各開關元件S1至S4的串聯分壓均等化而另外外接的電容器的電容。Further, each of the diodes D11 to D14 is a user who performs regeneration and reflow of the inverter 11 and can be omitted when there is a parasitic diode inside each of the switching elements S1 to S4. In addition, each of the stray capacitances C11 to C14 herein means a stray capacitance including a capacitor or the like as a whole: a stray capacitance parasitic between the drain and the source of each of the switching elements S1 to S4, and a stray capacitance The pole supply line equivalently generates a stray capacitance between the drain and the source of the switching element, and a capacitance of the externally connected capacitor in order to equalize the series division voltage of each of the switching elements S1 to S4.

另一方面，反相器控制電路15乃係依據第20圖所示的各種控制參數的設定而輸出適合的反相器控制用的指令脈波A1至A4者。On the other hand, the inverter control circuit 15 outputs the appropriate command pulse waves A1 to A4 for inverter control in accordance with the setting of various control parameters shown in FIG.

疏緩電路16乃係依據對反相器11的疏緩設定而將指令脈波A1至A4以一定間隔予以疏緩的電路。在疏緩電路16中可自由設定疏緩的間隔及疏緩的次數，例如每隔1脈波疏緩1脈波、或每隔2脈波疏緩2脈波等。此外，亦可不實施疏緩處理，此時使該疏緩電路16的動作停止或者省略疏緩電路16，而能夠將反相器控制電路15的指令脈波A1至A4直接提供給脈波控制電路17。The mitigation circuit 16 is a circuit that delays the command pulse waves A1 to A4 at regular intervals in accordance with the mitigation setting of the inverter 11. In the sluice circuit 16, the interval between the stagnation and the number of stagnations can be freely set, for example, one pulse wave is delayed every one pulse wave, or two pulse waves are delayed every two pulse waves. Further, the mitigation processing may not be performed. At this time, the operation of the sluice circuit 16 is stopped or the snubber circuit 16 is omitted, and the command pulse waves A1 to A4 of the inverter control circuit 15 can be directly supplied to the pulse wave control circuit. 17.

脈波控制電路17乃係對應於申請專利範圍中的控制手段者，如第3圖所示，由反相器控制脈波作成電路171與電荷回收用控制脈波作成電路172所構成。反相器控制脈波作成電路171係根據通過疏緩電路16後的疏緩處理後的指令脈波B1至B4，作成反相器控制脈波G1至G4，輸出至反相器11的各閘極驅動電路111至114。電荷回收用控制脈波作成電路172係根據通過疏緩電路16後的疏緩處理後的指令脈波B1至B4，作成具有預定脈波寬度tw(後述)的電荷回收控制脈波G11至G44，輸出至電荷回收電路18。The pulse wave control circuit 17 is a control device in accordance with the scope of the patent application. As shown in FIG. 3, the pulse wave control circuit 171 and the charge recovery control pulse wave circuit 172 are formed by an inverter. The inverter control pulse wave generation circuit 171 is configured to generate inverter control pulse waves G1 to G4 based on the command pulse waves B1 to B4 after the mitigation processing after the sluice circuit 16, and output to the gates of the inverter 11. The pole drive circuits 111 to 114. The charge recovery control pulse wave generation circuit 172 is configured to generate charge recovery control pulse waves G11 to G44 having a predetermined pulse width tw (described later) based on the command pulse waves B1 to B4 after the relief processing by the relief circuit 16. The output is to the charge recovery circuit 18.

此時，反相器控制脈波G1至G4的輸出時序(timing)係比各電荷回收控制脈波G11至G44的輸出時序被延遲達一定時間Δt。因此，在構成反相器11的各開關元件S1至S4藉由反相器控制脈波G1至G4而進行導通動作之前，電荷回收電路18會藉由電荷回收控制脈波G11至G44而啟動。該延遲時間Δt乃係電荷回收電路18的雜散電容電荷的最大電荷回收時間。At this time, the output timing of the inverter control pulse waves G1 to G4 is delayed by a certain time Δt than the output timing of each of the charge recovery control pulse waves G11 to G44. Therefore, before the switching elements S1 to S4 constituting the inverter 11 are turned on by the inverter controlling the pulse waves G1 to G4, the charge recovery circuit 18 is activated by the charge recovery control pulses G11 to G44. The delay time Δt is the maximum charge recovery time of the stray capacitance of the charge recovery circuit 18.

另外，作成上述反相器控制脈波G1至G4與電荷回收控制脈波G11至G44的脈波控制電路17係可使用如第3圖所示的類比IC電路來進行作成，但並不以此為限，例如亦可構成為將必要的脈波模式(pattern)預先登錄在ROM(Read Only Memory；唯讀記憶體)與FPGA(Field Programmable Gate Array；可規劃邏輯閘陣列)、微電腦(microcomputer)等的記憶設備中，依據各設定來輸出各脈波。Further, the pulse wave control circuit 17 for forming the inverter control pulse waves G1 to G4 and the charge recovery control pulse waves G11 to G44 can be created using the analog IC circuit as shown in Fig. 3, but this is not For example, it is also possible to preliminarily register a necessary pulse wave pattern in a ROM (Read Only Memory) and an FPGA (Field Programmable Gate Array) and a microcomputer (microcomputer). In a memory device such as the above, each pulse wave is output in accordance with each setting.

電荷回收電路18具備例如第4圖或第5圖所示的構成。該電荷回收電路18係在構成反相器11的各開關元件S1至S4藉由反相器控制脈波G1至G4而進行導通動作之前，藉由電荷回收控制脈波G11至G44而啟動。並且，電荷回收電路18係進行將存在於各開關元件S1至S4的離散電容C11至C14的電荷放電至電荷回收電路18、將電荷從電荷回收電路18充電至離散電容C11至C14的動作。另外，該電荷回收電路18的構成及作用係於後文進一步詳述。The charge recovery circuit 18 has a configuration as shown in FIG. 4 or FIG. 5, for example. The charge recovery circuit 18 is activated by the charge recovery control pulse waves G11 to G44 before the switching elements S1 to S4 constituting the inverter 11 are turned on by the inverter control pulse waves G1 to G4. Further, the charge recovery circuit 18 performs an operation of discharging the electric charges of the discrete capacitances C11 to C14 existing in the respective switching elements S1 to S4 to the charge recovery circuit 18, and charging the electric charges from the charge recovery circuit 18 to the discrete capacitances C11 to C14. The configuration and operation of the charge recovery circuit 18 will be described in further detail below.

第6圖係顯示與第2圖所示的各信號對應的時序圖(timing chart)之一例者，在此係顯示相當於雷射脈波(電壓脈波)的1脈波份的輸出期間Tp中的各信號波形。Fig. 6 is a view showing an example of a timing chart corresponding to each signal shown in Fig. 2, in which an output period Tp of one pulse portion corresponding to a laser pulse wave (voltage pulse wave) is displayed. Each signal waveform in .

反相器控制電路15係將依據控制參數(例如脈波寬度指令)，將相應的指令脈波A1至A4輸出至疏緩電路16。疏緩電路16係將該指令脈波A1至A4疏緩成所需要的狀態，輸出疏緩處理後的指令脈波B1至B4，該疏緩處理後的指令脈波B1至B4係輸入至脈波控制電路17。The inverter control circuit 15 outputs the corresponding command pulses A1 to A4 to the sever circuit 16 in accordance with control parameters (e.g., pulse width command). The mitigation circuit 16 delays the command pulse waves A1 to A4 into a desired state, and outputs the pulsing processed pulse waves B1 to B4, and the swash processed pulse waves B1 to B4 are input to the pulse. Wave control circuit 17.

輸入至脈波控制電路17的電荷回收用控制脈波作成電路172的疏緩處理後的指令脈波B1至B4在這係轉換成具有比疏緩處理後的指令脈波B1至B4短的脈波寬度tw的電荷回收控制脈波G11至G44並予以輸出。此外，輸入至反相器控制脈波作成電路171的疏緩處理後的指令脈波B1至B4在這係以比電荷回收控制脈波G11至G44延遲達一定時間Δt的狀態作為反相器控制脈波G1至G4輸出。The command pulse waves B1 to B4 after the delay processing of the charge recovery control pulse wave creation circuit 172 input to the pulse wave control circuit 17 are converted into pulses having a pulse pulse B1 to B4 shorter than the command pulse waves B1 to B4 after the relief processing. The charge recovery control pulse waves G11 to G44 of the wave width tw are output. Further, the command pulse waves B1 to B4 which are input to the inverter control pulse wave generating circuit 171 are subjected to the inverter control by the state of being delayed by the predetermined time Δt by the charge recovery control pulse waves G11 to G44. Pulse waves G1 to G4 are output.

從脈波控制電路17輸出的各反相器控制脈波G1至G4係輸入至對應反相器11的各開關元件S1至S4而設置的閘極驅動電路111至114。閘極驅動電路111至114係依據各控制脈波G1至G4，輸出必要的閘極驅動脈波以驅動開關元件S1至S4，而驅動各開關元件S1至S4導通/切斷。The inverter control pulse waves G1 to G4 output from the pulse wave control circuit 17 are input to the gate drive circuits 111 to 114 provided corresponding to the respective switching elements S1 to S4 of the inverter 11. The gate drive circuits 111 to 114 output necessary gate drive pulse waves to drive the switching elements S1 to S4 in accordance with the respective control pulse waves G1 to G4, and drive the respective switching elements S1 to S4 to be turned on/off.

藉此，於反相器11的輸出產生矩形的高頻率電壓，該輸出係藉由高頻變壓器13而升壓，經由第1電抗器14而供給至負載19。此時，由於第1電抗器14與介電體電容器192所構成的LC電路，消除輸出電流的高頻率成分，於負載19係流通大致正弦波的電流。結果，電力供給至如第1圖所示的充滿雷射介質(混合氣體)的放電空間7，產生放電，藉由因該放電而被激發的雷射介質的受激發射，雷射振盪器2進行雷射振盪輸出雷射脈波。另外，在反相器11能夠輸出高電壓的情形中，亦可不使用升壓變壓器13而直接經由第1電抗器14連接至負載19。Thereby, a rectangular high frequency voltage is generated at the output of the inverter 11, and the output is boosted by the high frequency transformer 13, and supplied to the load 19 via the first reactor 14. At this time, the LC circuit including the first reactor 14 and the dielectric capacitor 192 eliminates a high-frequency component of the output current, and a substantially sinusoidal current flows through the load 19. As a result, electric power is supplied to the discharge space 7 filled with the laser medium (mixed gas) as shown in Fig. 1, and a discharge is generated, and the laser is excited by the discharge of the laser medium excited by the discharge, the laser oscillator 2 Perform a laser oscillation to output a laser pulse. Further, in the case where the inverter 11 can output a high voltage, it is also possible to directly connect to the load 19 via the first reactor 14 without using the step-up transformer 13.

接著，詳細說明前述電荷回收電路18的構成及動作。Next, the configuration and operation of the charge recovery circuit 18 will be described in detail.

在本實施形態1中，電荷回收電路18係採用例如第4圖所示的構成。該電荷回收電路18係具有利用LC共振使存在於反相器11的各開關元件S1至S4的雜散電容C11至C14的電荷充放電之任務。為此，電荷回收電路18具有兩個充放電電路18a、18b，而各充放電電路18a、18b係具備：用以啟動該充放電電路18a、18b的開關元件S11、S44及S22、S33；用以在與反相器11的雜散電容C11至C14之間進行LC共振的第2電抗器182；及電荷回收用的電容器183。In the first embodiment, the charge recovery circuit 18 has a configuration as shown in Fig. 4, for example. This charge recovery circuit 18 has a task of charging and discharging the electric charges of the stray capacitances C11 to C14 of the respective switching elements S1 to S4 of the inverter 11 by LC resonance. Therefore, the charge recovery circuit 18 has two charge and discharge circuits 18a and 18b, and each of the charge and discharge circuits 18a and 18b includes switching elements S11 and S44 and S22 and S33 for activating the charge and discharge circuits 18a and 18b. The second reactor 182 that performs LC resonance between the stray capacitances C11 and C14 of the inverter 11 and the capacitor 183 for charge recovery.

此外，在使用MOSFET作為構成各充放電電路18a、18b的開關元件S11至S44的情形中，由於各MOSFET的內部存在寄生二極體184，因此為了阻止該寄生二極體184的順向通電，而針對寄生二極體184以相反極性的方式串聯連接有二極體185。此外，要能夠充放電係必須進行雙向的開關動作，因此在各充放電電路18a、18b中係分別將連接各開關S11至S44與二極體185之串聯電路顛倒並聯連接，且於其一端側連接第2電抗器182，於另一端側連接電容器183。Further, in the case where the MOSFET is used as the switching elements S11 to S44 constituting the respective charging and discharging circuits 18a and 18b, since the parasitic diode 184 is present inside each MOSFET, in order to prevent the forward energization of the parasitic diode 184, The diode 185 is connected in series to the parasitic diode 184 in a reverse polarity manner. In addition, in order to enable the charging and discharging system to perform the bidirectional switching operation, in each of the charging and discharging circuits 18a and 18b, the series circuits connecting the switches S11 to S44 and the diode 185 are connected in parallel and connected in parallel, and at one end side thereof. The second reactor 182 is connected, and the capacitor 183 is connected to the other end side.

並且，一方(例如圖的上側)的充放電電路18a的第2電抗器182側係連接至反相器11的一方的輸出端(第2圖的輸出端X)，另一方(例如圖的下側)的充放電電路18b的第2電抗器182側係連接至反相器11的另一方的輸出端(第2圖的輸出端Y)，此外，這些各充放電電路18a、18b的電容器183側連接至反相器11的負側(N側)端子。Further, the second reactor 182 side of the charge/discharge circuit 18a of one (for example, the upper side of the figure) is connected to one output terminal (output terminal X of FIG. 2) of the inverter 11, and the other side (for example, the lower side of the figure) The side of the second reactor 182 of the charging/discharging circuit 18b of the side is connected to the other output end of the inverter 11 (the output terminal Y of FIG. 2), and the capacitor 183 of each of the charging and discharging circuits 18a and 18b. The side is connected to the negative side (N side) terminal of the inverter 11.

在該電荷回收電路18中，對電容器183、開關元件S11至S44及二極體185，施加至少反相器11的母線電壓E的1/2之電壓。因此，在電荷回收電路18中，必須選擇具有額定值為母線電壓E的1/2以上的元件、或者串聯連接達必要數的元件俾使電路整體耐壓成為母線電壓的1/2。此外，於電容器183輸入有來自反相器11的雜散電容C11至C14的電荷。因此，避免因該電荷的輸出/輸入造成電容器183的電壓大幅地變化，較佳為連接比雜散電容C11至C14更大電容量的電容器183。例如，雜散電容C11至C14的電容量為數百pF時，電容器183的電容量可選定為數十nF以上。In the charge recovery circuit 18, at least 1/2 of the bus voltage E of the inverter 11 is applied to the capacitor 183, the switching elements S11 to S44, and the diode 185. Therefore, in the charge recovery circuit 18, it is necessary to select an element having a rated value of 1/2 or more of the bus voltage E or a component connected in series to a necessary number so that the overall withstand voltage of the circuit becomes 1/2 of the bus voltage. Further, charges from the stray capacitances C11 to C14 of the inverter 11 are input to the capacitor 183. Therefore, it is avoided that the voltage of the capacitor 183 is largely changed by the output/input of the charge, and it is preferable to connect the capacitor 183 having a larger capacitance than the stray capacitances C11 to C14. For example, when the capacitance of the stray capacitances C11 to C14 is several hundred pF, the capacitance of the capacitor 183 can be selected to be several tens of nF or more.

另外，在各開關S11至S44的內部不存在寄生二極體184的理想狀況中，就電荷回收電路18而言，能夠使用第5圖所示的構成。亦即，該電荷回收電路18具有兩個充放電電路18a、18b，而各充放電電路18a、18b係依序將第2電抗器182、開關元件186、電容器183串聯連接。並且，一方(例如圖的上側)的充放電電路18a的第2電抗器182側係連接至反相器11的一方的輸出端(第2圖的輸出端X)，另一方(例如圖的下側)的充放電電路18b的第2電抗器182側係連接至反相器11的另一方的輸出端(第2圖的輸出端Y)。此外，這些各充放電電路18a、18b的電容器183側連接至反相器11的負側(N側)端子。Further, in the ideal state in which the parasitic diode 184 is not present inside each of the switches S11 to S44, the charge recovery circuit 18 can be configured as shown in FIG. That is, the charge recovery circuit 18 has two charge and discharge circuits 18a and 18b, and each of the charge and discharge circuits 18a and 18b sequentially connects the second reactor 182, the switching element 186, and the capacitor 183 in series. Further, the second reactor 182 side of the charge/discharge circuit 18a of one (for example, the upper side of the figure) is connected to one output terminal (output terminal X of FIG. 2) of the inverter 11, and the other side (for example, the lower side of the figure) The side of the second reactor 182 of the charge/discharge circuit 18b of the side is connected to the other output end of the inverter 11 (the output terminal Y of FIG. 2). Further, the capacitor 183 side of each of the charge and discharge circuits 18a and 18b is connected to the negative side (N side) terminal of the inverter 11.

第4圖或第5圖所示的電荷回收電路18係利用LC共振使電荷對反相器11的雜散電容C11至C14充放電。因此，如下式(2)所示，最理想為以使由電荷回收電路18的第2電抗器182的電感值L與雜散電容C11至C14的總電容量C所決定的共振週期的1/2成為比最大電荷回收時間Δt更小的方式來設定第2電抗器182的電感值L。並且，此時產生的因雜散電容C11至C14的充放電而致的開關損失成為最小。The charge recovery circuit 18 shown in Fig. 4 or Fig. 5 charges and discharges the stray capacitances C11 to C14 of the inverter 11 by the LC resonance. Therefore, as shown in the following formula (2), it is most preferable to make 1/1 of the resonance period determined by the inductance value L of the second reactor 182 of the charge recovery circuit 18 and the total capacitance C of the stray capacitances C11 to C14. 2 The inductance value L of the second reactor 182 is set to be smaller than the maximum charge recovery time Δt. Further, the switching loss due to charge and discharge of the stray capacitances C11 to C14 generated at this time is minimized.

Δt≧π√LCΔt≧π√LC

∴L≦(Δt/π)2 ×1/C　(2)∴L≦(Δt/π) 2 ×1/C (2)

而在反相器11的開關元件S1至S4與二極體185通電時的電路阻抗(impedance)為非常小的理想條件中，流通於電荷回收電路18的電流峰值I係由下式(3)所定。On the other hand, in the ideal condition in which the impedance of the circuit when the switching elements S1 to S4 of the inverter 11 and the diode 185 are energized are extremely small, the current peak I flowing through the charge recovery circuit 18 is expressed by the following equation (3). Set.

I=V√(L/C)　(3)I=V√(L/C) (3)

因此，若縮短電荷回收時間Δt，第2電抗器182的電感值L會減小，但電流峰值I會變大，構成各充放電電路18a、18b的開關元件S11至S44(186)與二極體185必須選定能夠流通比較高的峰值電流的元件。反之，若增長電荷回收時間Δt，必須增大第2電抗器182的電感值L，但由於電流峰值I小，因此沒必要為能夠流通高峰值電流的元件。如上述，由於流通於電荷回收電路18的電流是由電荷回收時間Δt與第2電抗器182的電感值L所決定，故依據所選擇的元件的額定值與電路尺寸來選擇適當的電抗器。Therefore, when the charge recovery time Δt is shortened, the inductance value L of the second reactor 182 is decreased, but the current peak value I is increased, and the switching elements S11 to S44 (186) and the diodes constituting the respective charge and discharge circuits 18a and 18b are formed. Body 185 must select an element that is capable of flowing a relatively high peak current. On the other hand, if the charge recovery time Δt is increased, the inductance value L of the second reactor 182 must be increased. However, since the current peak I is small, it is not necessary to be an element capable of circulating a high peak current. As described above, since the current flowing through the charge recovery circuit 18 is determined by the charge recovery time Δt and the inductance L of the second reactor 182, an appropriate reactor is selected in accordance with the rated value of the selected component and the circuit size. .

另外，若將由第2電抗器182與雜散電容C11至C14所決定的串聯共振頻率設定為充分高於第2電抗器182的電感值及第1電抗器14的電感值的電感值和與介電體電容器192的串聯共振頻率，便能夠降低從電荷回收電路18流入負載19的流入電流。據此，自雜散電容C11至C14的電荷回收效果變大，並且能夠減小構成各充放電電路18a、18b的開關元件S11至S44(186)與二極體185的損失。Further, the series resonance frequency determined by the second reactor 182 and the stray capacitances C11 to C14 is set to be sufficiently higher than the inductance value of the second reactor 182 and the inductance value of the first reactor 14 and the inductance value. The series resonance frequency of the electric capacitor 192 can reduce the inflow current flowing from the charge recovery circuit 18 into the load 19. According to this, the charge recovery effect from the stray capacitances C11 to C14 becomes large, and the loss of the switching elements S11 to S44 (186) and the diodes 185 constituting the respective charge and discharge circuits 18a and 18b can be reduced.

此外，即使將第2電抗器182的電感值設定為比第1電抗器14的電感值小，自雜散電容C11至C14的電荷回收效果仍變大，並且仍能夠減少構成各充放電電路18a、18b的開關元件S11至S44(186)與二極體185的損失。或者，以使雜散電容C11至C14的阻抗成為比第1電抗器14與負載19的串聯阻抗還小的方式來設定第1電抗器14的電感值，以使該第1電抗器14的阻抗值與負載19的介電體電容器192發生共振的方式設定反相器11的頻率亦能夠得到上述相同的效果。Further, even if the inductance value of the second reactor 182 is set smaller than the inductance value of the first reactor 14, the charge recovery effect from the stray capacitances C11 to C14 becomes large, and the charge and discharge circuits 18a can be reduced. Loss of the switching elements S11 to S44 (186) of 18b and the diode 185. Alternatively, the inductance of the first reactor 14 is set such that the impedance of the stray capacitances C11 to C14 is smaller than the series impedance of the first reactor 14 and the load 19, so that the impedance of the first reactor 14 is made. The same effect as described above can be obtained by setting the frequency of the inverter 11 such that the value resonates with the dielectric capacitor 192 of the load 19.

另外，構成反相器11的各開關元件S1至S4的雜散電容C11至C14的電荷之回收係必須以制定為該上下開關元件(S1與S4、S2與S3)不同時導通的空檔時間(dead time)期間中的極短時間進行。例如，在以數百kHz至數MHz驅動反相器11時，該空檔時間為數十毫微秒(nano second)至數百毫微秒左右，電荷的回收必須在該範圍以下的時間進行。因此，藉由使用具有比習知的使用Si(矽)半導體的開關元件的開關速度(103 V/微秒)快了1位數以上的高速開關(104 V/微秒至105 V/微秒)之寬帶隙半導體(wide bandgap semiconductor)，電荷回收的效果變得更大，能夠更降低反相器11的各開關元件S1至S4的損失。就此時的寬帶隙半導體而言，例如有SiC(碳化矽)、氮化鎵系材料或鑽石。Further, the charge recovery of the stray capacitances C11 to C14 of the respective switching elements S1 to S4 constituting the inverter 11 must be set to a neutral time when the upper and lower switching elements (S1 and S4, S2 and S3) are not turned on. (dead time) is carried out in a very short time. For example, when the inverter 11 is driven at several hundred kHz to several MHz, the neutral time is from about nanoseconds to several hundred nanoseconds, and charge recovery must be performed at a time below the range. . Therefore, a high-speed switch (10 4 V / microsecond to 10 5 V) having a switching speed (10 3 V / microsecond) faster than a conventional switching element using a Si (矽) semiconductor is used. In the case of a wide bandgap semiconductor, the effect of charge recovery becomes larger, and the loss of each of the switching elements S1 to S4 of the inverter 11 can be further reduced. For the wide band gap semiconductor at this time, for example, SiC (tantalum carbide), gallium nitride-based material, or diamond is used.

接著，參照第7圖及第8至13圖，說明使用具有第4圖所示構成的電荷回收電路18時對反相器11的雜散電容C11至C14的充放電動作。第7圖係顯示在不對指令脈波A1至A4進行疏緩的情形下根據反相器控制脈波G1至G4使反相器11連續動作時的各開關元件S1至S4與電荷回收電路18的動作波形、以及伴之而從反相器11輸出的電壓Viv、Iiv，此外，第8至13圖係顯示在第7圖的情形中流通於反相器11的電流路徑(圖中的虛線)。Next, the charge and discharge operation of the stray capacitances C11 to C14 of the inverter 11 when the charge recovery circuit 18 having the configuration shown in Fig. 4 is used will be described with reference to Fig. 7 and Figs. 8 to 13. Fig. 7 shows the switching elements S1 to S4 and the charge recovery circuit 18 when the inverter 11 is continuously operated in accordance with the inverter control pulse waves G1 to G4 without delaying the command pulse waves A1 to A4. The operation waveform and the voltages Viv and Iiv which are outputted from the inverter 11 in addition, and the eighth to thirteenth drawings show the current path (broken line in the figure) flowing through the inverter 11 in the case of FIG. .

如第7圖所示，在反相器11的開關元件S3、S4於時刻t0完全切斷後到反相器11的開關元件S1、S2於時刻t2導通為止的期間中的成為t0<t1<t2之時刻t1的時序，將電荷回收電路18的開關元件S11、S22導通。此外，在反相器11的開關元件S1、S2於時刻t2導通到於時刻t4切斷為止的期間中的成為t2<t3<t4之時刻t3的時序，電荷回收電路18的開關元件S11、S22切斷。As shown in Fig. 7, when the switching elements S3 and S4 of the inverter 11 are completely turned off at time t0 and the switching elements S1 and S2 of the inverter 11 are turned on at time t2, t0 < t1 < t2. At the timing of time t1, the switching elements S11 and S22 of the charge recovery circuit 18 are turned on. In addition, the switching elements S1 and S22 of the charge recovery circuit 18 are turned on at the timing t3 when t2 < t3 < t4 in the period from the time t2 to the time t4 when the switching elements S1 and S2 of the inverter 11 are turned on. Cut off.

如上述，將電荷回收電路18的開關元件S11、S22導通的時序時刻t1設定為t0<t1<t2，此外，將開關元件S11、S22切斷的時序時刻t3設定為t2<t3<t4，如此設定的理由如下。亦即，為了防止因電荷回收電路18的開關元件S11與反相器11的開關元件S4及電荷回收電路18的開關元件S22與反相器11的開關元件S3同時導通而形成橋臂短路(arm short)造成各開關元件故障之故。這些脈波的適當輸出時序係由脈波控制電路17決定。As described above, the timing t1 at which the switching elements S11 and S22 of the charge recovery circuit 18 are turned on is set to t0 < t1 < t2, and the timing t3 at which the switching elements S11 and S22 are turned off is set to t2 < t3 < t4. The reasons for setting are as follows. That is, in order to prevent the bridge arm short circuit (arm) from being caused by the switching element S11 of the charge recovery circuit 18 and the switching element S4 of the inverter 11 and the switching element S22 of the charge recovery circuit 18 and the switching element S3 of the inverter 11 being simultaneously turned on. Short) causes the failure of each switching element. The appropriate output timing of these pulse waves is determined by the pulse wave control circuit 17.

如上述，由於電荷回收控制脈波G11、G22於時刻t1輸出，因此如第8圖所示，藉由該脈波G11、G22，電荷回收電路18的開關元件S11、S22同時導通。藉此，在(t1至t2)的期間中，電荷從電荷回收電路18的電容器183轉移至反相器11的開關元件S4的雜散電容C14進行充電，並且反相器11的開關元件S1的雜散電容C11的蓄積電荷被放電。與此同時，反相器11的開關元件S2的雜散電容C12的蓄積電荷被放電，電荷轉移至電荷回收電路18的電容器183，並且電荷充電至反相器11的開關元件S3的雜散電容C13。As described above, since the charge recovery control pulse waves G11 and G22 are outputted at time t1, as shown in FIG. 8, the switching elements S11 and S22 of the charge recovery circuit 18 are simultaneously turned on by the pulse waves G11 and G22. Thereby, in the period of (t1 to t2), the electric charge is transferred from the capacitor 183 of the charge recovery circuit 18 to the stray capacitance C14 of the switching element S4 of the inverter 11 to be charged, and the switching element S1 of the inverter 11 is charged. The accumulated electric charge of the stray capacitance C11 is discharged. At the same time, the accumulated electric charge of the stray capacitance C12 of the switching element S2 of the inverter 11 is discharged, the electric charge is transferred to the capacitor 183 of the charge recovery circuit 18, and the electric charge is charged to the stray capacitance of the switching element S3 of the inverter 11. C13.

如此，在反相器11的開關元件S1、S2的雜散電容C11、C12所蓄積的電荷大致全被放電的狀態下，於下個時刻t2輸出反相器控制脈波G1、G2使反相器11的兩開關元件S1、S2導通。因此，無通過兩開關元件S1、S2的通道的來自雜散電容C11、C12的電荷，開關元件S1、S2的開關損失大幅降低。In the state where the electric charges accumulated in the stray capacitances C11 and C12 of the switching elements S1 and S2 of the inverter 11 are substantially completely discharged, the inverter control pulse waves G1 and G2 are outputted at the next time t2 to be inverted. The two switching elements S1, S2 of the device 11 are turned on. Therefore, the electric charge from the stray capacitances C11 and C12 passing through the passages of the two switching elements S1 and S2 does not significantly reduce the switching loss of the switching elements S1 and S2.

接著，在(t2至t4)的期間，反相器11的開關元件S1、S2皆導通，藉此，如第9圖所示，於開關元件S1、S2流通電流。Then, during the period from (t2 to t4), the switching elements S1 and S2 of the inverter 11 are all turned on, whereby as shown in Fig. 9, current flows through the switching elements S1 and S2.

接著，在從反相器11的開關元件S1、S2於時刻t4完全切斷後到反相器11的開關元件S3、S4於時刻t6導通為止的期間的(t4至t6)的期間乃係反相器11的空檔時間。接著，如後述，電荷回收電路18的開關元件S33、S34於此空檔時間期間中的時刻t5的時序導通，而在到目前為止(t4至t5)的期間中，如第10圖所示，反相器11的電流開始再生，與此同時，開關元件S3、S4的雜散電容C13、C14的電荷開始被放電，另一方面，開關元件S1、S2的雜散電容C11、C12開始被充電。雜散電容C13、C14的電荷一旦被放電，充電電壓便逐漸降低，但由於再生時間的長短使放電不夠完全，成為在雜散電容C13、C14殘留有電荷的狀態。Then, the period from (t4 to t6) in the period from when the switching elements S1 and S2 of the inverter 11 are completely turned off at the time t4 to when the switching elements S3 and S4 of the inverter 11 are turned on at the time t6 is inverted. The neutral time of the device 11. Next, as will be described later, the switching elements S33 and S34 of the charge recovery circuit 18 are turned on at the timing of time t5 in the neutral time period, and as shown in FIG. 10, in the period from now on (t4 to t5), The current of the inverter 11 starts to regenerate, and at the same time, the electric charges of the stray capacitances C13 and C14 of the switching elements S3 and S4 are discharged, and on the other hand, the stray capacitances C11 and C12 of the switching elements S1 and S2 are charged. . When the electric charges of the stray capacitances C13 and C14 are discharged, the charging voltage is gradually lowered. However, the discharge is not sufficiently completed due to the length of the regeneration time, and the electric charge remains in the stray capacitances C13 and C14.

接著，於反相器11的開關元件S1至S4皆成為切斷的空檔時間期間中的成為t4<t5<t6的時刻t5的時序，電荷回收電路18的開關元件S33、S44導通。此外，反相器11的開關元件S3、S4於時刻t6導通，在反相器11的開關元件S3、S4於時刻t8切斷為止的期間的成為t6<t7<t8的時刻t7的時序，電荷回收電路18的開關元件S33、S34切斷。Then, the switching elements S1 to S4 of the inverter 11 are turned on at the timing t5 of t4 < t5 < t6 in the cut neutral time period, and the switching elements S33 and S44 of the charge recovery circuit 18 are turned on. In addition, the switching elements S3 and S4 of the inverter 11 are turned on at time t6, and the timing of the period t7 at which t6 < t7 < t8 during the period in which the switching elements S3 and S4 of the inverter 11 are turned off at time t8 is charged. The switching elements S33 and S34 of the recovery circuit 18 are turned off.

如上述，將電荷回收電路18的開關元件S33、S44導通的時序時刻t5設定為t4<t5<t6的期間，此外，將開關元件S33、S44切斷的時序時刻t7設定為t6<t7<t8的期間，如此設定的理由如下。亦即，為了防止因電荷回收電路18的開關元件S33與反相器11的開關元件S2及電荷回收電路18的開關元件S44與反相器11的開關元件S1同時導通而形成橋臂短路(arm short)造成各開關元件故障之故。這些脈波的適當輸出時序係由脈波控制電路17決定。As described above, the timing t5 at which the switching elements S33 and S44 of the charge recovery circuit 18 are turned on is set to a period in which t4 < t5 < t6, and the timing t7 at which the switching elements S33 and S44 are turned off is set to t6 < t7 < t8. The reason for this setting is as follows. That is, in order to prevent the bridge arm short circuit (arm) from being caused by the switching element S33 of the charge recovery circuit 18 and the switching element S2 of the inverter 11 and the switching element S44 of the charge recovery circuit 18 and the switching element S1 of the inverter 11 being simultaneously turned on. Short) causes the failure of each switching element. The appropriate output timing of these pulse waves is determined by the pulse wave control circuit 17.

如上述，由於電荷回收控制脈波G33、G44於時刻t5輸出，因此如第11圖所示，藉由該脈波G33、G44，電荷回收電路18的開關元件S33、S44同時導通。藉此，在(t5至t6)的期間中，電荷從電荷回收電路18的電容器183轉移至反相器11的開關元件S2的雜散電容C12進行充電，並且從反相器11的開關元件S3的雜散電容C13的電荷被放電。與此同時，反相器11的開關元件S4的雜散電容C14被放電，電荷轉移至電荷回收電路18的電容器183，並且電荷充電至反相器11的開關元件S1的雜散電容C11。As described above, since the charge recovery control pulse waves G33 and G44 are output at time t5, as shown in Fig. 11, the switching elements S33 and S44 of the charge recovery circuit 18 are simultaneously turned on by the pulse waves G33 and G44. Thereby, in the period of (t5 to t6), the electric charge is transferred from the capacitor 183 of the charge recovery circuit 18 to the stray capacitance C12 of the switching element S2 of the inverter 11 to be charged, and the switching element S3 of the inverter 11 is charged. The charge of the stray capacitance C13 is discharged. At the same time, the stray capacitance C14 of the switching element S4 of the inverter 11 is discharged, the charge is transferred to the capacitor 183 of the charge recovery circuit 18, and the electric charge is charged to the stray capacitance C11 of the switching element S1 of the inverter 11.

如此，在反相器11的開關元件S3、S4的雜散電容C13、C14所蓄積的電荷大致全被放電的狀態下，於時刻t6輸出反相器控制脈波G3、G4使反相器11的兩開關元件S3、S4導通。因此，無通過兩開關元件S3、S4的通道的來自雜散電容C13、C14的電荷，這些開關元件S3、S4的開關損失大幅降低。In this state, in a state where the electric charges accumulated in the stray capacitances C13 and C14 of the switching elements S3 and S4 of the inverter 11 are substantially completely discharged, the inverter control pulse waves G3 and G4 are outputted at time t6 to turn on the inverter 11 . The two switching elements S3, S4 are turned on. Therefore, the switching loss of the switching elements S3 and S4 is greatly reduced without the electric charges from the stray capacitances C13 and C14 passing through the channels of the two switching elements S3 and S4.

接著，在(t6至t8)的期間，反相器11的開關元件S3、S4皆導通，藉此，如第12圖所示，於開關元件S3、S4流通電流。Next, during the period from (t6 to t8), the switching elements S3 and S4 of the inverter 11 are all turned on, whereby as shown in Fig. 12, current flows through the switching elements S3 and S4.

接著，在從反相器11的開關元件S3、S4於時刻t8完全切斷後到反相器11的開關元件S1、S2於時刻t10導通為止的期間的t8<t<t10的期間乃係反相器11的空檔時間期間。接著，電荷回收電路18的開關元件S11、S22於此空檔時間期間中的時刻t9的時序導通。接著，在(t8至t9)的期間中，如第13圖所示，反相器11的電流開始再生，與此同時，開關元件S1、S2的雜散電容C11、C12的電荷開始被放電，另一方面，開關元件S3、S4的雜散電容C13、C14的雜散電容開始被充電。雜散電容C11、C12的電荷一旦被放電，充電電壓便逐漸降低，但由於再生時間的長短使放電不夠完全，成為在雜散電容C11、C12殘留有電荷的狀態。Then, the period from the time when the switching elements S3 and S4 of the inverter 11 are completely turned off at the time t8 to the period in which the switching elements S1 and S2 of the inverter 11 are turned on at the time t10 is inverted. During the neutral time of the device 11. Next, the switching elements S11 and S22 of the charge recovery circuit 18 are turned on at the timing of time t9 during the neutral time period. Then, in the period from (t8 to t9), as shown in Fig. 13, the current of the inverter 11 starts to be regenerated, and at the same time, the electric charges of the stray capacitances C11 and C12 of the switching elements S1 and S2 are discharged. On the other hand, the stray capacitances of the stray capacitances C13, C14 of the switching elements S3, S4 are initially charged. When the electric charges of the stray capacitances C11 and C12 are discharged, the charging voltage is gradually lowered. However, the discharge is not sufficiently completed due to the length of the regeneration time, and the electric charge remains in the stray capacitances C11 and C12.

之後，同樣的動作繼續下去。After that, the same action continues.

如上述，於必要的時序將構成電荷回收電路18的開關元件S11至S44導通/切斷，使電荷回收電路18的電抗器182與反相器11的開關元件S1至S4的雜散電容C11至C14共振，藉此，能夠使充電至反相器11的開關元件S1至S4的雜散電容C11至C14的電荷輸出/輸入於電荷回收電路18的電容器183。並且，藉此，尤其在反相器11的開關元件S1至S4即將導通之前能夠降低開關元件S1至S4的雜散電容C11至C14的充電電壓。因此，在增加雷射脈波的重複頻率與脈波寬度時，即使開關次數增加仍能夠大幅降低開關損失。As described above, the switching elements S11 to S44 constituting the charge recovery circuit 18 are turned on/off at necessary timings, so that the reactor 182 of the charge recovery circuit 18 and the stray capacitance C11 of the switching elements S1 to S4 of the inverter 11 are turned to C14 resonates, whereby charges of the stray capacitances C11 to C14 charged to the switching elements S1 to S4 of the inverter 11 can be output/inputted to the capacitor 183 of the charge recovery circuit 18. And, by this, the charging voltages of the stray capacitances C11 to C14 of the switching elements S1 to S4 can be lowered, particularly before the switching elements S1 to S4 of the inverter 11 are turned on. Therefore, when the repetition frequency and the pulse width of the laser pulse wave are increased, the switching loss can be greatly reduced even if the number of switching times is increased.

此外，如本實施形態1，在使用電荷回收電路18的情形中，係如第7圖所示，在反相器11的各開關元件S1至S4導通/切斷時係流通脈波狀的電流，因此於構成電荷回收電路18的開關元件S11至S44與二極體185產生的通電損失極少。然而，大部分的電荷係被再利用於反相器11的開關元件S1至S4的雜散電容C11至C14的充放電。因此，電荷回收電路18本身產生的損失極少，高頻率電源裝置1整體的損失降低，因此能夠構成非常高效率的電源。結果，能夠構成一種在進行裝置的大容量化時，能夠避免裝置的冷卻機構的增設與裝置本身的構成大型化，也不需要進行許多元件、電路的並列，成本面的優點自不待言、機械的設置空間的考量上也非常有利的電源裝置。Further, in the first embodiment, when the charge recovery circuit 18 is used, as shown in Fig. 7, when the switching elements S1 to S4 of the inverter 11 are turned on/off, a pulse-like current flows. Therefore, the conduction loss generated by the switching elements S11 to S44 constituting the charge recovery circuit 18 and the diode 185 is extremely small. However, most of the charge is reused for charge and discharge of the stray capacitances C11 to C14 of the switching elements S1 to S4 of the inverter 11. Therefore, the loss generated by the charge recovery circuit 18 itself is extremely small, and the loss of the entire high-frequency power supply device 1 is reduced, so that a very high-efficiency power supply can be constructed. As a result, it is possible to prevent the increase in the size of the device and the increase in the size of the device itself when the device is increased in capacity, and it is not necessary to carry out the juxtaposition of many components and circuits, and the advantages of the cost surface are self-evident. The consideration of the setting space is also very advantageous for the power supply unit.

此外，在使用此種電荷回收電路18的情形中，於反相器11的動作前，雜散電容C11至C14的蓄積電荷被充放電於電荷回收用的電容器183，因此能夠使反相器11的開關元件S1至S4的開關速度很快速，結果如第7圖的(t1至t2)、(t5至t6)、(t9至t10)的各期間所示，反相器輸出的工作比(Duty)擴大達有Δt量，每一脈波的反相器電流Iiv變大，輸入電力增加。在進行脈波驅動雷射時，於放電在放電空間7開始進行的放電初期，電流非常不易流通，難以使雷射輸出增加。然而，藉由使用本方式，能夠從脈波的初期使反相器電流Iiv增加，因此能夠使1脈波1脈波的雷射輸出增加，並且能夠獲得極易啟動的雷射脈波輸出。Further, in the case of using such a charge recovery circuit 18, the accumulated charges of the stray capacitances C11 to C14 are charged and discharged to the capacitor 183 for charge recovery before the operation of the inverter 11, so that the inverter 11 can be made. The switching speeds of the switching elements S1 to S4 are very fast, and as a result, as shown in the respective periods (t1 to t2), (t5 to t6), and (t9 to t10) of Fig. 7, the operation ratio of the inverter output (Duty) The expansion reaches an amount of Δt, the inverter current Iiv of each pulse wave becomes large, and the input power increases. When the pulse-driven laser is applied, the current is extremely difficult to flow at the initial stage of discharge in which the discharge starts in the discharge space 7, and it is difficult to increase the laser output. However, by using this embodiment, the inverter current Iiv can be increased from the initial stage of the pulse wave, so that the laser output of one pulse wave and one pulse wave can be increased, and the laser pulse wave output that is extremely easy to start can be obtained.

另外，在第7圖所示的例子中，雖然係顯示反相器11的各相對角的開關元件S1、S2及S3、S4分別同時導通/切斷的情形，但在如藉由PWM等使開關元件S1、S2及S3、S4不同時導通/切斷的情形中亦是藉由於上述的時序將與各開關元件S1至S4對應的電荷回收電路18的各開關元件S11至S44予以導通/切斷而能夠獲得相同的效果。Further, in the example shown in Fig. 7, the switching elements S1, S2, S3, and S4 of the respective opposite angles of the inverter 11 are simultaneously turned on/off, but by PWM or the like. In the case where the switching elements S1, S2, and S3, and S4 are not turned on or off at the same time, the switching elements S11 to S44 of the charge recovery circuit 18 corresponding to the respective switching elements S1 to S4 are turned on/off by the above-described timing. The same effect can be obtained by breaking.

雖然第7圖所示的例子為不疏緩指令脈波A1至A4而是依據反相器控制脈波G1至G4使反相器11連續性地動作之情形，然而在藉由疏緩電路16疏緩指令脈波A1至A4而產生反相器控制脈波G1至G4及電荷回收控制脈波G11至G44之情形中，亦能獲得與使反相器11連續性地動作之情形相同的效果。Although the example shown in FIG. 7 is a case where the inverter pulses 11 are continuously operated in accordance with the inverter control pulse waves G1 to G4 without slowing down the command pulse waves A1 to A4, the slowdown circuit 16 is used. In the case where the command pulse waves A1 to A4 are generated to generate the inverter control pulse waves G1 to G4 and the charge recovery control pulse waves G11 to G44, the same effect as in the case where the inverter 11 is continuously operated can be obtained. .

第14圖係顯示藉由疏緩電路16對指令脈波A1至A4進行疏緩處理而產生反相器控制脈波G1至G4及電荷回收控制脈波G11至G44時的動作。亦即，在第14圖中，在於(t1至t8)的期間中輸出反相器控制脈波G1至G4與電荷回收控制脈波G11至G44後，由於(t8至t9)的期間的指令脈波被疏緩，因此這些控制脈波未被輸出，於下個時刻t9之後，再度輸出反相器控制脈波G1至G4與電荷回收控制脈波G11至G44。此時，放電電流及雷射脈波的輸出峰值係依指令脈波A1至A4的疏緩間隔而變化，因此考慮雷射振盪器2輸出的輸出峰值來決定要每隔幾個脈波疏緩多少脈波。Fig. 14 is a view showing an operation when the inverter pulse waves G1 to G4 and the charge recovery control pulse waves G11 to G44 are generated by the slack processing of the command pulse waves A1 to A4 by the delay circuit 16. That is, in Fig. 14, after the inverter control pulse waves G1 to G4 and the charge recovery control pulse waves G11 to G44 are output in the period of (t1 to t8), the command pulse due to the period of (t8 to t9) The waves are unsettled, so these control pulse waves are not output, and after the next time t9, the inverter control pulse waves G1 to G4 and the charge recovery control pulse waves G11 to G44 are again output. At this time, the output peak of the discharge current and the laser pulse wave changes according to the intermittent interval of the command pulse waves A1 to A4. Therefore, considering the output peak value of the output of the laser oscillator 2, it is determined that every pulse wave is to be slowed down. How many pulse waves.

在以疏緩電路16對指令脈波A1至A4進行疏緩處理而進行動作的情形中，經疏緩的(t8至t9)的期間乃係反相器11沒有流通電流(與連續動作不同，雜散電容的電荷藉由再生而未被放電)的期間。在習知技術的情形中，雖然充電電壓會因反相器停止瞬後的共振振動而有某程度的降低，但由於雜散電容C11至C14被維持在反相器即將停止前的充電電壓(大致反相器11的母線電壓)，因此開關損失相較於連續動作時變得非常大，難以藉由脈波的疏緩條件來擴大雷射脈波的脈波寬度。In the case where the slack circuit 16 operates to delay the command pulse waves A1 to A4, the period of the delay (t8 to t9) is that the inverter 11 does not have a current flowing (unlike the continuous operation, The period during which the charge of the stray capacitance is not discharged by regeneration. In the case of the prior art, although the charging voltage is somewhat reduced due to the instantaneous resonance vibration of the inverter, the stray capacitances C11 to C14 are maintained at the charging voltage immediately before the inverter is stopped ( Since the bus voltage of the inverter 11 is substantially the same, the switching loss becomes very large compared to the continuous operation, and it is difficult to expand the pulse width of the laser pulse by the pulse wave relaxation condition.

相對於此，在本發明中，在對脈波進行疏緩處理而進行動作的情形中，藉由使用電荷回收電路18而能夠使開關損失大幅地降低，因此能夠使雷射脈波的重複頻率與脈波寬度大幅地增加。此外，由於能夠大幅降低開關損失，因此高頻率電源裝置的冷卻機構得以簡化，能夠謀求裝置小型化、成本降低化、設置空間縮小化等。On the other hand, in the present invention, when the pulse wave is subjected to the mitigation processing and the operation is performed, the switching loss can be greatly reduced by using the charge recovery circuit 18, so that the repetition frequency of the laser pulse wave can be made. The pulse width is greatly increased. Further, since the switching loss can be greatly reduced, the cooling mechanism of the high-frequency power supply device can be simplified, and the size of the device can be reduced, the cost can be reduced, and the installation space can be reduced.

此外，在第14圖所示的例子乃係針對藉由疏緩電路16對指令脈波A1至A4進行疏緩處理而單純地產生反相器控制脈波G1至G4及電荷回收控制脈波G11至G44的情形進行說明。然而並不以此為限，亦可如第15圖所示，針對從脈波控制電路17輸出的反相器控制脈波G1至G4之中的例如與反相器11的兩個開關元件S3、S4對應而輸出的反相器控制脈波G3、G4，採取在疏緩期間(t8至t9)中仍繼續反相器控制脈波G3、G4的輸出的方式，而使開關元件S3、S4的導通狀態繼續。Further, the example shown in FIG. 14 is for simply generating the inverter control pulse waves G1 to G4 and the charge recovery control pulse wave G11 by the delay circuit 16 for the command pulse waves A1 to A4. The case to G44 will be explained. However, it is not limited thereto, and as shown in FIG. 15, for the inverter switching signals G1 to G4 output from the pulse wave control circuit 17, for example, the two switching elements S3 of the inverter 11 may be used. The inverter control pulse waves G3 and G4 outputted in response to S4 take the mode of continuing the inverter control pulse waves G3 and G4 during the sluggish period (t8 to t9), thereby causing the switching elements S3 and S4. The conduction state continues.

如上述，在疏緩期間(t8至t9)中繼續開關元件S3、S4的導通狀態，藉此，寄生在該開關元件S3、S4的雜散電容C13、C14的充電電壓係成為0。另一方面，另一方的開關元件S1、S2的雜散電容C11、C12係被維持在被以大致母線電壓E充電的狀態。在此時點，電荷回收電路18的電容器183係處於被充電成為母線電壓的1/2的電壓(1/2E)的狀態。該差距部分的電壓1/2E乃係LC共振時的電壓。在電荷回收電路18中，該電壓乃係LC共振的最大電壓，在以脈波方式驅動反相器11時，最能夠獲得電荷回收的效果。As described above, the conduction state of the switching elements S3 and S4 is continued during the stagnation period (t8 to t9), whereby the charging voltages of the stray capacitances C13 and C14 parasitic on the switching elements S3 and S4 become zero. On the other hand, the stray capacitances C11 and C12 of the other switching elements S1 and S2 are maintained in a state of being charged by the substantially bus voltage E. At this time, the capacitor 183 of the charge recovery circuit 18 is in a state of being charged to a voltage (1/2E) of 1/2 of the bus voltage. The voltage 1/2E of the difference portion is the voltage at the time of LC resonance. In the charge recovery circuit 18, this voltage is the maximum voltage of the LC resonance, and when the inverter 11 is driven by the pulse wave method, the effect of charge recovery is most obtained.

藉此，能夠使雷射脈波寬度與雷射脈波的重複頻率大幅地增加。此外，由於能夠大幅降低產生的開關損失，因此高頻率電源裝置的冷卻機構得以簡化，能夠謀求裝置小型化、成本降低化、設置空間縮小化等。Thereby, the laser pulse width and the repetition frequency of the laser pulse wave can be greatly increased. Further, since the switching loss generated can be greatly reduced, the cooling mechanism of the high-frequency power supply device can be simplified, and the size of the device can be reduced, the cost can be reduced, and the installation space can be reduced.

另外，除了如上述將反相器11的開關元件S3、S4繼續予以導通的動作之外，當使電荷回收電路18動作時，有反相器電流的峰值上升的情形。在此種情形中，只要減少反相器11的開關元件S1、S4的導通工作比(on duty)即可。能夠藉由減少導通工作比來降低反相器電流的峰值，從而能夠調整放電電力，而能夠更細微地調整雷射輸出。Further, in addition to the operation of continuing to turn on the switching elements S3 and S4 of the inverter 11 as described above, when the charge recovery circuit 18 is operated, the peak value of the inverter current rises. In this case, it is only necessary to reduce the on duty of the switching elements S1, S4 of the inverter 11. The peak of the inverter current can be reduced by reducing the on-operation ratio, so that the discharge power can be adjusted, and the laser output can be finely adjusted.

此外，在第7圖、第14圖、第15圖中係顯示在更早於各個反相器控制脈波G1至G4的輸出時序之前必定輸出電荷回收控制脈波G11至G44的例子。然而，當對於反相器11的上下的開關元件S1、S4及S2、S3無法取得足夠的空檔時間期間時，則亦可如第16圖所示，採取僅輸出與反相器控制脈波G1、G2對應的電荷回收控制脈波G11、G22，而與反相器控制脈波G3、G4對應的電荷回收控制脈波G33、G44則不予以輸出的方式。Further, in FIGS. 7 , 14 , and 15 , an example in which the charge recovery control pulse waves G11 to G44 are necessarily output before the output timings of the respective inverter control pulse waves G1 to G4 is displayed. However, when the upper and lower switching elements S1, S4, and S2, and S3 of the inverter 11 cannot obtain a sufficient neutral time period, as shown in FIG. 16, only the output and the inverter control pulse wave can be taken. The charge recovery control pulse waves G11 and G22 corresponding to G1 and G2 are not outputted by the charge recovery control pulse waves G33 and G44 corresponding to the inverter control pulse waves G3 and G4.

如上述，僅在反相器11從停止狀態移行至動作狀態時的最初的反相器控制脈波G1、G2的輸出時序之前輸出電荷回收控制脈波G11、G22亦能夠對開關損失有不錯的效果。As described above, outputting the charge recovery control pulses G11 and G22 only before the output timing of the first inverter control pulse waves G1 and G2 when the inverter 11 is shifted from the stop state to the operation state can also have a good switching loss. effect.

此係如先前所述，由於在反相器11連續動作時，於再生期間電荷被放電，損失變小，但在從停止狀態移行至動作狀態時，在雜散電容C11、C12受高電壓充電的狀態下開始進行開關，開關損失變得非常大之故。因此，藉由採用如第16圖所示的開關模式亦能夠將大幅增加雷射脈波的重複頻率與脈波寬度時的開關損失予以大幅地降低，高頻率電源裝置的冷卻機構得以簡化，能夠謀求裝置小型化、成本降低化、設置空間縮小化等。As described above, since the electric charge is discharged during the regenerative operation when the inverter 11 is continuously operated, the loss becomes small, but when the shift state is stopped from the stopped state to the operating state, the stray capacitances C11 and C12 are charged by the high voltage. When the switch is started, the switching loss becomes very large. Therefore, by adopting the switching mode as shown in Fig. 16, it is possible to greatly reduce the switching loss when the repetition frequency of the laser pulse wave and the pulse width are greatly increased, and the cooling mechanism of the high-frequency power supply device can be simplified. The device is miniaturized, the cost is reduced, and the installation space is reduced.

實施形態2Embodiment 2

第17圖係本發明實施形態2的高頻率電源裝置的構成圖，與第2圖至第5圖所示的實施形態1對應或相當的構成部分標註相同的符號。Fig. 17 is a configuration diagram of a high-frequency power supply device according to a second embodiment of the present invention, and components corresponding to or corresponding to those of the first embodiment shown in Figs. 2 to 5 are denoted by the same reference numerals.

在上述的實施形態1中，係在反相器11裝設單一直流電源12，且以該直流電源12作為反相器11的母線電壓E之構成。在本實施形態2中，輸入至反相器11的直流電源分為兩個，且將兩直流電源12a、12b的中點設置在地線(earth)，將±E/2的電壓輸入至反相器11。此外，在各直流電源12a、12b與反相器11之間，各直流電源12a、12b有電容器C1、C2並聯連接。並且，該反相器11與電荷回收電路18連接。In the first embodiment described above, the single DC power supply 12 is provided in the inverter 11, and the DC power supply 12 is used as the bus voltage E of the inverter 11. In the second embodiment, the DC power source input to the inverter 11 is divided into two, and the midpoints of the two DC power sources 12a and 12b are set at the earth (earth), and the voltage of ±E/2 is input to the opposite. Phaser 11. Further, between the DC power supplies 12a and 12b and the inverter 11, the DC power supplies 12a and 12b are connected in parallel with the capacitors C1 and C2. Further, the inverter 11 is connected to the charge recovery circuit 18.

此情形中，電荷回收電路18係能夠採用如第18圖或第19圖所示的構成。與實施形態1的第4圖、第5圖的構成相比較，在各充放電電路18a、18b中，省略掉電荷回收用的電容器183，電抗器182側與反相器11的各輸出端X、Y連接，此外，開關元件S11至S44(186)側與屬於直流電源12a、12b的中點的地線連接。In this case, the charge recovery circuit 18 can adopt a configuration as shown in Fig. 18 or Fig. 19. In comparison with the configurations of the fourth and fifth embodiments of the first embodiment, in each of the charge and discharge circuits 18a and 18b, the capacitor 183 for charge recovery, the reactor 182 side and the output terminal X of the inverter 11 are omitted. The Y connection is connected, and the switching elements S11 to S44 (186) side are connected to the ground line belonging to the midpoint of the DC power sources 12a, 12b.

其餘構成係與實施形態1相同，因此在此省略詳細說明。The rest of the configuration is the same as that of the first embodiment, and thus detailed description thereof is omitted here.

若採用如上述的構成，由於將分離的兩個直流電源12a、12b的中點接地，因此電路各部的與地線間的最大電壓被壓低在實施形態1的直流電源12的電壓的一半。藉此，反相器11的耐壓設計變得容易，可謀求電路的小型化。此外，在實施形態1的情形中，電荷回收用的電容器183係需要充電有母線電壓的1/2的電壓者，而在本實施形態2中，能夠以連接至直流電源12a、12b的中點與反相器11的負側(N側)的電容器C2代用，因此電荷回收電路18不需要電容器，能夠使電荷回收電路18更加小型化。According to the above configuration, since the midpoints of the separated DC power supplies 12a and 12b are grounded, the maximum voltage between the respective lines of the circuit and the ground is reduced to half the voltage of the DC power supply 12 of the first embodiment. Thereby, the withstand voltage design of the inverter 11 becomes easy, and the circuit can be downsized. Further, in the case of the first embodiment, the capacitor 183 for charge recovery needs to be charged with a voltage of 1/2 of the bus voltage, and in the second embodiment, it is possible to connect to the midpoint of the DC power sources 12a and 12b. Since the capacitor C2 on the negative side (N side) of the inverter 11 is substituted, the charge recovery circuit 18 does not require a capacitor, and the charge recovery circuit 18 can be further miniaturized.

關於反相器11與電荷回收電路18的控制動作係與實施形態1相同，藉此，能夠使雷射脈波的重複頻率與脈波寬度大幅地增加。此外，由於能夠大幅降低開關損失，因此高頻率電源裝置的冷卻機構得以簡化，能夠謀求裝置小型化、成本降低化、設置空間縮小化等。The control operation of the inverter 11 and the charge recovery circuit 18 is the same as that of the first embodiment, whereby the repetition frequency of the laser pulse wave and the pulse width can be greatly increased. Further, since the switching loss can be greatly reduced, the cooling mechanism of the high-frequency power supply device can be simplified, and the size of the device can be reduced, the cost can be reduced, and the installation space can be reduced.

1．．．高頻率電源裝置1. . . High frequency power supply unit

2．．．雷射振盪器2. . . Laser oscillator

3．．．電極3. . . electrode

4．．．介電體4. . . Dielectric body

5．．．部分反射鏡5. . . Partial mirror

6．．．全反射鏡6. . . Total reflection mirror

7．．．放電空間7. . . Discharge space

11．．．反相器11. . . inverter

12、12a、12b．．．直流電源12, 12a, 12b. . . DC power supply

13．．．升壓變壓器13. . . Step-up transformer

14．．．第1電抗器14. . . First reactor

15．．．反相器控制電路15. . . Inverter control circuit

16．．．疏緩電路16. . . Suppression circuit

17．．．脈波控制電路17. . . Pulse wave control circuit

18．．．電荷回收電路18. . . Charge recovery circuit

18a、18b．．．充放電電路18a, 18b. . . Charge and discharge circuit

19．．．負載19. . . load

111至114．．．閘極驅動電路111 to 114. . . Gate drive circuit

171．．．反相器控制脈波作成電路171. . . Inverter control pulse wave making circuit

172．．．電荷回收用控制脈波作成電路172. . . Charge recovery control pulse wave making circuit

182．．．第2電抗器182. . . Second reactor

183．．．電容器183. . . Capacitor

184．．．寄生二極體184. . . Parasitic diode

185．．．二極體185. . . Dipole

186、S11至S44．．．開關186, S11 to S44. . . switch

191．．．放電電阻器191. . . Discharge resistor

192．．．介電體電容器192. . . Dielectric capacitor

A1至A4．．．指令脈波A1 to A4. . . Command pulse

B1至B4．．．指令脈波B1 to B4. . . Command pulse

C．．．電容器C. . . Capacitor

C1、C2．．．電容器C1, C2. . . Capacitor

C11至C14．．．雜散電容C11 to C14. . . Stray capacitance

D11至D14．．．二極體D11 to D14. . . Dipole

E．．．母線電壓E. . . bus voltage

G1至G4．．．反相器控制脈波G1 to G4. . . Inverter control pulse wave

G11至G44．．．電荷回收控制脈波G11 to G44. . . Charge recovery control pulse wave

R．．．電阻器R. . . Resistor

S1至S4．．．開關元件S1 to S4. . . Switching element

t0至t10．．．時刻T0 to t10. . . time

tw．．．脈波寬度Tw. . . Pulse width

Tp．．．輸出期間Tp. . . Output period

X、Y．．．輸出端X, Y. . . Output

Δt．．．延遲時間Δt. . . delay

第1圖係顯示氣體雷射裝置的概要之構成圖。Fig. 1 is a view showing the outline of a gas laser device.

第2圖係本發明實施形態1的高頻率電源裝置之構成圖。Fig. 2 is a view showing the configuration of a high-frequency power supply device according to a first embodiment of the present invention.

第3圖係顯示本發明實施形態1的高頻率電源裝置的脈波控制電路構成之方塊圖。Fig. 3 is a block diagram showing the configuration of a pulse wave control circuit of the high-frequency power supply device according to the first embodiment of the present invention.

第4圖係顯示本發明實施形態1的高頻率電源裝置的電荷回收電路的一例之構成圖。Fig. 4 is a block diagram showing an example of a charge recovery circuit of the high-frequency power supply device according to the first embodiment of the present invention.

第5圖係顯示本發明實施形態1的高頻率電源裝置的電荷回收電路的另一例之構成圖。Fig. 5 is a block diagram showing another example of the charge recovery circuit of the high-frequency power supply device according to the first embodiment of the present invention.

第6圖係顯示本發明實施形態1的高頻率電源裝置的控制動作的一例之時序圖。Fig. 6 is a timing chart showing an example of the control operation of the high-frequency power supply device according to the first embodiment of the present invention.

第7圖係顯示本發明實施形態1的高頻率電源裝置的驅動時的反相器控制脈波與電荷回收控制脈波以及因該些脈波而流通於各開關元件的電流之時序圖。Fig. 7 is a timing chart showing the inverter control pulse wave and the charge recovery control pulse wave and the current flowing through the respective switching elements due to the pulse waves in the driving of the high-frequency power supply device according to the first embodiment of the present invention.

第8圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 8 is an explanatory view showing a path of a current flowing through each unit of the inverter during driving of the high-frequency power supply device according to the first embodiment of the present invention.

第9圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 9 is an explanatory view showing a path of a current flowing through each unit of the inverter during driving of the high-frequency power supply device according to the first embodiment of the present invention.

第10圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 10 is an explanatory view showing a path of a current flowing through each unit of the inverter during driving of the high-frequency power supply device according to the first embodiment of the present invention.

第11圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 11 is an explanatory view showing a path of a current flowing through each unit of the inverter when the high-frequency power supply device according to the first embodiment of the present invention is driven.

第12圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 12 is an explanatory view showing a path of a current flowing through each unit of the inverter during driving of the high-frequency power supply device according to the first embodiment of the present invention.

第13圖係顯示本發明實施形態1的高頻率電源裝置的驅動時流通於反相器的各部的電流的路徑之說明圖。Fig. 13 is an explanatory view showing a path of a current flowing through each unit of the inverter during driving of the high-frequency power supply device according to the first embodiment of the present invention.

第14圖係顯示使本發明實施形態1的高頻率電源裝置進行疏緩控制動作時的一例之時序圖。Fig. 14 is a timing chart showing an example of a case where the high-frequency power supply device according to the first embodiment of the present invention performs a sluggish control operation.

第15圖係顯示使本發明實施形態1的高頻率電源裝置進行疏緩控制動作時的另一例之時序圖。Fig. 15 is a timing chart showing another example of the case where the high-frequency power supply device according to the first embodiment of the present invention performs the slack control operation.

第16圖係顯示使本發明實施形態1的高頻率電源裝置進行疏緩控制動作時的又一例之時序圖。Fig. 16 is a timing chart showing still another example of the case where the high-frequency power supply device according to the first embodiment of the present invention performs the slack control operation.

第17圖係本發明實施形態2的高頻率電源裝置的構成圖。Figure 17 is a configuration diagram of a high-frequency power supply device according to a second embodiment of the present invention.

第18圖係顯示本發明實施形態2的高頻率電源裝置的電荷回收電路的一例之構成圖。Fig. 18 is a block diagram showing an example of a charge recovery circuit of the high-frequency power supply device according to the second embodiment of the present invention.

第19圖係顯示本發明實施形態2的高頻率電源裝置的電荷回收電路的另一例之構成圖。Fig. 19 is a block diagram showing another example of the charge recovery circuit of the high-frequency power supply device according to the second embodiment of the present invention.

第20圖係為了控制雷射振盪器的雷射輸出而設定的控制參數之說明圖。Figure 20 is an explanatory diagram of control parameters set to control the laser output of the laser oscillator.

第21圖係顯示伴隨第20圖的控制參數設定的反相器的開關元件驅動狀態與雷射脈波輸出之關係之說明圖。Fig. 21 is an explanatory view showing the relationship between the switching element driving state of the inverter and the laser pulse wave output in accordance with the control parameter setting of Fig. 20.

1．．．高頻率電源裝置1. . . High frequency power supply unit

11．．．反相器11. . . inverter

12．．．直流電源12. . . DC power supply

13．．．升壓變壓器13. . . Step-up transformer

14．．．第1電抗器14. . . First reactor

15．．．反相器控制電路15. . . Inverter control circuit

16．．．疏緩電路16. . . Suppression circuit

17．．．脈波控制電路17. . . Pulse wave control circuit

18．．．電荷回收電路18. . . Charge recovery circuit

19．．．負載19. . . load

111至114．．．閘極驅動電路111 to 114. . . Gate drive circuit

191．．．放電電阻器191. . . Discharge resistor

192．．．介電體電容器192. . . Dielectric capacitor

A1至A4．．．指令脈波A1 to A4. . . Command pulse

B1至B4．．．指令脈波B1 to B4. . . Command pulse

C．．．電容器C. . . Capacitor

C11至C14．．．雜散電容C11 to C14. . . Stray capacitance

D11至D14．．．二極體D11 to D14. . . Dipole

E．．．母線電壓E. . . bus voltage

G1至G4．．．反相器控制脈波G1 to G4. . . Inverter control pulse wave

G11、G22、G33、G44．．．電荷回收控制脈波G11, G22, G33, G44. . . Charge recovery control pulse wave

R．．．電阻器R. . . Resistor

S1至S4．．．開關元件S1 to S4. . . Switching element

X、Y．．．輸出端X, Y. . . Output

Claims (10)

一種高頻率電源裝置，係輸出電壓脈波至負載的高頻率電源裝置，其係包括：電力轉換用的反相器，具有開關元件；第1電抗器，與前述負載連接；電荷回收電路，與前述反相器的輸出端連接，用以回收前述開關元件的雜散電容的電荷；及控制手段，控制前述反相器與前述電荷回收電路的動作；前述電荷回收電路係包括：開關元件，用以啟動該電荷回收電路；第2電抗器，用以在與前述雜散電容之間產生LC共振；及電容器，用以回收電荷；前述控制手段乃係在即將輸出使前述反相器的開關元件導通的反相器控制脈波之前，輸出使前述電荷回收電路的開關元件導通的電荷回收控制脈波者。 A high-frequency power supply device is a high-frequency power supply device that outputs a voltage pulse to a load, and includes: an inverter for power conversion, having a switching element; a first reactor connected to the load; and a charge recovery circuit; The output end of the inverter is connected to recover the charge of the stray capacitance of the switching element; and the control means controls the operation of the inverter and the charge recovery circuit; the charge recovery circuit includes: a switching element To activate the charge recovery circuit; a second reactor for generating LC resonance between the stray capacitance; and a capacitor for recovering charge; the control means is to output a switching element for the inverter Before the turned-on inverter controls the pulse wave, the charge recovery control pulse wave that turns on the switching element of the charge recovery circuit is output. 一種高頻率電源裝置，係輸出電壓脈波至負載的高頻率電源裝置，其係包括：電力轉換用的反相器，具有開關元件；第1電抗器，與前述負載連接；直流電源，供給直流電力至該反相器；電荷回收電路，與前述反相器的輸出端連接，用以回收前述開關元件的雜散電容的電荷；及 控制手段，控制前述反相器與前述電荷回收電路的動作；前述直流電源係分離成兩個且其中點接地至地線，此外，在前述直流電源與反相器之間，於各直流電源並聯連接電容器；前述電荷回收電路係具有：開關元件，用以啟動該電荷回收電路；及第2電抗器，用以在與前述雜散電容之間產生LC共振；前述控制手段乃係在即將輸出使前述反相器的開關元件導通的反相器控制脈波之前，輸出使前述電荷回收電路的開關元件導通的電荷回收控制脈波者。 A high-frequency power supply device is a high-frequency power supply device that outputs a voltage pulse to a load, and includes: an inverter for power conversion, having a switching element; a first reactor connected to the load; a DC power supply, supplying a DC Power is supplied to the inverter; a charge recovery circuit is coupled to the output of the inverter to recover the charge of the stray capacitance of the switching element; and Control means for controlling the operation of the inverter and the charge recovery circuit; the DC power source is separated into two and the midpoint thereof is grounded to the ground, and further, between the DC power source and the inverter, the DC power sources are connected in parallel Connecting the capacitor; the foregoing charge recovery circuit has: a switching element for activating the charge recovery circuit; and a second reactor for generating LC resonance between the stray capacitance; and the foregoing control means is to output Before the inverter in which the switching element of the inverter is turned on controls the pulse wave, the charge recovery control pulse wave that turns on the switching element of the charge recovery circuit is output. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，前述第2電抗器的值係設定為使由前述電荷回收電路的第2電抗器與前述反相器的開關元件的雜散電容所決定的LC共振的週期的1/2成為較前述電荷回收電路的開關元件導通的電荷回收時間小。 The high frequency power supply device according to claim 1 or 2, wherein the value of the second reactor is set to a switching element of the second reactor of the charge recovery circuit and the inverter The half of the period of the LC resonance determined by the stray capacitance becomes smaller than the charge recovery time of the switching element of the charge recovery circuit. 如申請專利範圍第1項所述之高頻率電源裝置，其中，前述電荷回收電路的電容器的電容值係設定為比前述反相器的開關元件的雜散電容的電容值更大。 The high-frequency power supply device according to claim 1, wherein a capacitance value of a capacitor of the charge recovery circuit is set to be larger than a capacitance value of a stray capacitance of a switching element of the inverter. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，由前述第2電抗器與前述反相器的開關元件的雜散電容所決定的串聯共振頻率係設定為比由前述第1電抗器與前述第2電抗器的電感值和與前述負載的 介電體電容器所決定的串聯共振頻率更高。 The high frequency power supply device according to claim 1 or 2, wherein the series resonance frequency determined by the stray capacitance of the switching element of the second reactor and the inverter is set to be The inductance values of the first reactor and the second reactor and the load The series resonant frequency determined by the dielectric capacitor is higher. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，前述第2電抗器的電感值係設定為比前述第1電抗器的電感值更小。 The high-frequency power supply device according to the first or second aspect of the invention, wherein the inductance of the second reactor is set to be smaller than an inductance value of the first reactor. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，以使前述反相器的開關元件的雜散電容的阻抗比前述第1電抗器與前述負載的串聯阻抗更小的方式來設定前述第1電抗器的電感值，且前述反相器的頻率係設定為使該第1電抗器的電感值與前述負載的介電體電容器產生串聯共振。 The high-frequency power supply device according to claim 1 or 2, wherein the impedance of the stray capacitance of the switching element of the inverter is smaller than the series impedance of the first reactor and the load. The inductance of the first reactor is set in such a manner that the frequency of the inverter is set such that the inductance of the first reactor is in series resonance with the dielectric capacitor of the load. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，包括有疏緩電路，該疏緩電路係依據根據前述電壓脈波的輸出特性而定的控制參數，對前述反相器控制脈波及電荷回收控制脈波進行疏緩處理，以變更前述反相器與電荷回收電路的各開關元件的開關次數。 The high-frequency power supply device according to claim 1 or 2, further comprising a sluice circuit, wherein the severance circuit is based on a control parameter according to an output characteristic of the voltage pulse wave, The phase control pulse wave and the charge recovery control pulse wave are subjected to a mitigation process to change the number of times of switching of each of the switching elements of the inverter and the charge recovery circuit. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，前述控制手段乃係以使在前述反相器的最初動作的相對角的開關元件之後動作的另一方的相對角的開關元件的導通期間比該最初動作的相對角的開關元件的導通期間更長的方式輸出前述反相器控制脈波者。 The high-frequency power supply device according to claim 1 or 2, wherein the control means is a relative angle of the other one that operates after the switching element at the relative angle of the initial operation of the inverter The inverting control pulse wave is outputted in a manner that the on period of the switching element is longer than the on period of the switching element of the opposite angle of the initial operation. 如申請專利範圍第1項或第2項所述之高頻率電源裝置，其中，前述電荷回收電路的開關元件係以寬帶隙半導體所形成。The high frequency power supply device according to claim 1 or 2, wherein the switching element of the charge recovery circuit is formed of a wide band gap semiconductor.
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