TWI399062B - Interference estimating apparatus, method and apparatus for determining guard interval length and bit loading - Google Patents

Description

干擾估測裝置、決定保護區間與位元負載的方法及其 裝置 Interference estimation device, method for determining protection interval and bit load, and Device

本發明是有關於一種多載波通訊系統，且特別是有關於一種干擾估測裝置、決定保護區間與位元負載的方法以及裝置。 The present invention relates to a multi-carrier communication system, and more particularly to an interference estimation apparatus, a method and apparatus for determining a guard interval and a bit load.

在通訊技術不斷進步的時代裡，能夠在有限的頻寬之下，乘載越來越多的資料位元，已經變成不可避免的趨勢。為了更有效的利用頻譜(spectrum)，目前已發展出利用多載波的通訊技術，例如分頻多工(frequency division multiplexing,FDM)與正交分頻多工(orthogonal frequency division multiplexing,OFDM)等等。以OFDM技術為例，其主要是將有限的頻寬分為多個子通道(sub-channel)，並運用多個子載波(sub-carrier)平行傳輸，而每個子載波之間維持正交性(orthogonality)，其中每個子載波透過不同的調變，使得上述子載波可分別乘載不同的資料位元量。 In an era of continuous advancement in communication technology, it has become an inevitable trend to be able to carry more and more data bits under a limited bandwidth. In order to utilize the spectrum more effectively, communication technologies using multiple carriers, such as frequency division multiplexing (FDM) and orthogonal frequency division multiplexing (OFDM), etc., have been developed. . Taking OFDM technology as an example, it mainly divides the limited bandwidth into multiple sub-channels and uses multiple sub-carriers to transmit in parallel, while maintaining orthogonality between each sub-carrier (orthogonality) ), wherein each subcarrier transmits different modulations, so that the above subcarriers can respectively carry different data bit quantities.

在OFDM技術中，為了防止符元間干擾(Inter Symbol Interference,ISI)以及載波間干擾(Inter Carrier Interference,ICI)，一般的符元是由一信號加上防護區間(Guard Interval,GI)，例如循環前置(Cyclic Prefix,CP)，所構成。一般利用的正交分頻多工技術的通訊系統大多數都會在規格中有明確規定一或多個固定長度的GI，以因應各種不同的環境下，該採用何種長度的GI。然而GI的長度選擇不當，將會很直接的影響到傳送資料的效率。以下利用Home-Plug AV系統來說明防護區間對效率的影響。 In OFDM technology, in order to prevent Inter Symbol Interference (ISI) and Inter Carrier Interference (ICI), a general symbol is a signal plus a Guard Interval (GI), for example Cyclic Prefix (CP), composed of. Most of the communication systems that use the orthogonal frequency division multiplexing technology generally specify one or more fixed length GIs in the specification to determine which length of GI to use in various environments. However, the improper selection of the length of the GI will directly affect the efficiency of transmitting data. Use Home-Plug below The AV system is used to illustrate the impact of the guard zone on efficiency.

圖1繪示為在Home-Plug AV系統中所規定之長的(long)媒體存取協定資料單元(Media access control Protocol Data Unit,MPDU)的示意圖。請參考圖1，一種Home-Plug AV規格的MPDU可以包括128位元的框控制區塊AVFC以及N個520位元組的物理層(PHY block)區塊PB520。每一個PB520包括4位元組的物理層區塊標頭檔H520、512位元組的資料B520以及4位元組的物理層區塊檢查碼CS520。 FIG. 1 is a schematic diagram showing a long media access control protocol data unit (MPDU) defined in a Home-Plug AV system. Referring to FIG. 1, a Home-Plug AV specification MPDU may include a 128-bit block control block AVFC and N 520-bit PHY block blocks PB520. Each PB 520 includes a 4-byte physical layer block header file H520, a 512-bit data B520, and a 4-byte physical layer block check code CS520.

一般Home-Plug AV系統中所使用的OFDM調變技術包括1536個子載波(tone或者是sub-carrier)。每一個子載波根據其調變的不同，可以乘載1至10位元的負載(bit loading)。若上述的子載波中，其中一特定子載波使用64-QAM調變，則此特定子載波的位元負載就是6 bit；若此特定子載波使用QPSK調變，則此特定子載波的位元負載就是2 bit。然而，在Home-Plug AV系統的規範中，真實可用的子載波不見得是1536個子載波；在各國的法規可能會規定某些頻帶必須受到管制，例如以北美來說，在Home-Plug AV系統僅917個子載波可使用，這個規範稱為tone mask。 The OFDM modulation technique used in the general Home-Plug AV system includes 1536 subcarriers (tone or sub-carrier). Each subcarrier can carry a bit loading of 1 to 10 bits depending on its modulation. If one of the above subcarriers uses 64-QAM modulation, the bit load of the specific subcarrier is 6 bits; if the specific subcarrier uses QPSK modulation, the bit of the specific subcarrier is used. The load is 2 bits. However, in the specification of the Home-Plug AV system, the truly available subcarriers are not necessarily 1536 subcarriers; regulations in various countries may stipulate that certain frequency bands must be regulated, for example, in North America, in the Home-Plug AV system. Only 917 subcarriers are available. This specification is called a tone mask.

接下來，便以北美的規定，並配合Home-Plug AV系統中所使用的OFDM調變來舉例。首先，假設傳送端依照Home-Plug AV系統所規定的標準強健傳送模式(standard robust,STD-ROBO)，要傳送一個PB520，其中STD-ROBO 規定每一個tone必須使用QPSK調變，且欲傳送之資料必須重複傳送4次，此資料的編碼率為1/2。如此，便可以簡單計算出利用STD-ROBO規範傳送一個PB520所需要傳送的位元數：Bit number of PB520=520(位元組)×8(轉為位元)×2(編碼率)×4(重複4次)=33280 bits。 Next, the North American regulations are used in conjunction with the OFDM modulation used in the Home-Plug AV system. First, assume that the transmitter is transmitting a PB520 according to the standard robust transmission mode (STD-ROBO) specified by the Home-Plug AV system, of which STD-ROBO It is stipulated that each tone must use QPSK modulation, and the data to be transmitted must be transmitted four times. The coding rate of this data is 1/2. In this way, the number of bits to be transmitted by transmitting a PB520 using the STD-ROBO specification can be simply calculated: Bit number of PB520=520 (bytes)×8 (transferred to bits)×2 (coding rate)×4 (repeated 4 times) = 33280 bits.

一個具有917個子載波，且所有子載波皆利用QPSK的符元可以傳送的位元數：Bit number of Symbol=917×2=1834 bits。 A number of bits that have 917 subcarriers and all subcarriers can be transmitted using QPSK symbols: Bit number of Symbol=917×2=1834 bits.

因此，若要完整的傳送一個PB520，理想的情況下必須要有33280/1834=18.146個符元才能傳送完成。接下來，再舉一例子，假設使用傳收雙方溝通好的bit loading，且每個tone的平均傳送位元數為5 bits，編碼率為1/2，則傳送一個PB520需要的符元個數為：(520×8×2×4)/(917×5)=7.258個符元。 Therefore, in order to completely transmit a PB520, ideally there must be 33280/1834=18.146 symbols to complete the transfer. Next, let's take another example. Assume that the bit loading with good communication is used, and the average number of transmitted bits per tone is 5 bits, and the coding rate is 1/2, then the number of symbols required for transmitting a PB520 is transmitted. It is: (520 × 8 × 2 × 4) / (917 × 5) = 7.258 symbols.

在Home-Plug AV系統中，使用3072點的反傅立葉轉換(IFFT)為調變，並規定3種GI的長度。分別是GI-417、GI-567以及GI-3534，其中GI-417表示一個符元是由417個循環前置，加上原本的OFDM信號所構成，GI-567與GI3534則可以依此類推。以HomePlug AV規範的75MHz clock rate，GI-417本身所佔的時間為5.56μs，GI-567為7.56μs，GI-3534為47.12μs。在習知的技術中，為了防止符元間干擾(Inter Symbol Interference,ISI)以及載波間干擾(Inter Carrier Interference,ICI)，通常會根據通道的脈衝響 應的長度，來決定使用多長的GI。若選擇GI-417，則一個符元所需的傳送時間為沒有加上GI的符元之1.14倍；若選擇GI-567，則一個符元所需的傳送時間為沒有加上GI的符元之1.18倍；然而若選擇GI-3534，則一個符元所需的傳送時間為沒有加上GI的符元之2.15倍。在習知的技術中，若不使用時域等化器(Time-domain Equalizer)縮短等效通道長度，則一般須選擇GI長度比通道的脈衝響應之長度(簡稱通道長度)來的長，以避免符間干擾與載波間干擾，但此方法並無法保證將整體的傳輸效率最大化。舉例來說，在某些情況下，等效通道長度可能僅略大於GI-517，但是卻必須採用GI-3534以避免GI長度不足產生ISI與ICI，影響原本bit loading的估測結果，這顯然的對資料的傳送/接收效率會造成很大的影響。 In the Home-Plug AV system, the inverse Fourier transform (IFFT) of 3072 points is used for modulation, and the lengths of the three GIs are specified. They are GI-417, GI-567, and GI-3534, respectively. GI-417 indicates that a symbol is composed of 417 loop preambles, plus the original OFDM signal, and GI-567 and GI3534 can be deduced by analogy. At the 75MHz clock rate of the HomePlug AV specification, the time occupied by the GI-417 itself is 5.56 μs, the GI-567 is 7.56 μs, and the GI-3534 is 47.12 μs. In the prior art, in order to prevent Inter Symbol Interference (ISI) and Inter Carrier Interference (ICI), the pulse of the channel is usually generated. The length should be determined to determine how long to use the GI. If GI-417 is selected, the transmission time required for one symbol is 1.14 times that of the symbol without GI; if GI-567 is selected, the transmission time required for one symbol is the symbol without GI. 1.18 times; however, if GI-3534 is selected, the transmission time required for one symbol is 2.15 times that of the symbol without GI. In the prior art, if the equivalent channel length is shortened without using a Time-domain Equalizer, it is generally necessary to select the length of the GI length to be longer than the length of the channel impulse response (referred to as the channel length). Avoid inter-symbol interference and inter-carrier interference, but this method does not guarantee the overall transmission efficiency. For example, in some cases, the equivalent channel length may only be slightly larger than GI-517, but GI-3534 must be used to avoid the lack of GI length to produce ISI and ICI, which affects the estimation of the original bit loading. The transmission/reception efficiency of data can have a great impact.

本發明在提供一種決定保護區間與位元負載的方法，用以根據通道長度以及干擾之大小，以決定循環前置長度以及各個子載波所乘載之資料位元量。 The present invention provides a method for determining a guard interval and a bit load for determining a loop preamble length and a data bit amount to be carried by each subcarrier according to the channel length and the magnitude of the interference.

本發明在提供一種決定保護區間長度與位元負載的裝置，用以增加位元產量(throughput)。 The present invention provides a means for determining the length of a guard interval and the load of a bit to increase the bit throughput.

本發明另提供一種干擾估測裝置，用以有效估測ISI、ICI等干擾。 The invention further provides an interference estimation device for effectively estimating interferences such as ISI and ICI.

本發明提出一種適應性決定保護區間長度與位元負載的方法，首先，提供多載波調變系統，其規格規定第一保護區間以及第二保護區間，其中第一保護區間的長度小於 第二保護區間的長度。接著，在通道閒置期間，估測每一子通道的雜訊能量。然後，估測通道脈衝響應(impulse channel response)。之後，計算每一子載波所能乘載的第一位元負載。當通道的長度小於第一保護區間時，則決定使用第一位元負載與第一保護區間。當通道長度大於或等於第一保護區間的長度時，估測符元加上第一保護區間時的子載波間干擾能量與符元間干擾能量，並計算出此時符元加上第一保護區間的信號雜訊比，以得到每一子載波所能乘載的第二位元負載。最後，當使用第一保護區間並配合第二位元負載的符元之位元產量大於使用第二保護區間並配合第一位元負載的符元之位元產量時，決定用第一保護區間與第二位元負載，否則，則使用第二保護區間與第一位元負載。 The present invention provides a method for adaptively determining the length of a guard interval and a bit load. First, a multi-carrier modulation system is provided, the specification of which defines a first guard interval and a second guard interval, wherein the length of the first guard interval is less than The length of the second guard interval. Next, the noise energy of each subchannel is estimated during the idle period of the channel. Then, the impulse channel response is estimated. Thereafter, the first bit load that can be carried by each subcarrier is calculated. When the length of the channel is less than the first guard interval, it is decided to use the first bit load and the first guard interval. When the channel length is greater than or equal to the length of the first guard interval, the inter-subcarrier interference energy and the inter-symbol interference energy of the estimated symbol plus the first guard interval are calculated, and the symbol plus the first protection is calculated at this time. The signal to noise ratio of the interval is obtained to obtain the second bit load that can be carried by each subcarrier. Finally, when the first guard interval is used and the bit yield of the symbol of the second bit load is greater than the bit yield of the symbol using the second guard interval and the first bit load, the first guard interval is determined. With the second bit load, otherwise, the second guard interval is used with the first bit load.

依照本發明的實施例所述之適應性決定保護區間長度與位元負載的方法，上述「計算該多載波調變系統中之每一子載波所能乘載的一第一位元負載」，此步驟包括：估測每一子通道的信號能量；以及計算每一子載波的信號雜訊比，並藉此決定每一子載波所能乘載的該第一位元負載。在一特定實施例中，其中「計算每一子載波的信號雜訊比」是指在沒有符元間干擾與子載波間干擾時的信號雜訊比。 The method for determining the length of the guard interval and the bit load according to the adaptability of the embodiment of the present invention, the above “calculating a first bit load that can be carried by each subcarrier in the multi-carrier modulation system”, The step includes: estimating signal energy of each subchannel; and calculating a signal to noise ratio of each subcarrier, and thereby determining the first bit load that each subcarrier can carry. In a particular embodiment, wherein "calculating the signal to noise ratio of each subcarrier" is the signal to noise ratio when there is no inter-symbol interference and inter-subcarrier interference.

依照本發明的實施例所述之適應性決定保護區間長度與位元負載的方法，上述第q個子通道的信號能量表示為，第q個子通道的雜訊能量表示為，則「符元加 上第一保護區間之後，第q個子通道的信號雜訊比SNR _q 」表示如下： 其中P _ISI(q)為在第q個子通道的符元間干擾的能量，P _ICI(q)為在第q個子通道的子載波間干擾的能量。而且對於任何子通道q，P _ISI(q)=P_ICI(q)。 The method according to the embodiment of the present invention determines a method for protecting a length of a section and a bit load, and the signal energy of the qth subchannel is expressed as , the noise energy of the qth subchannel is expressed as Then, after the sign is added to the first guard interval, the signal-to-noise ratio SNR _{q of} the q- th sub-channel is expressed as follows: Where P _ISI ( q ) is the energy that interferes between the symbols of the qth subchannel, and P _ICI ( q ) is the energy of the interference between the subcarriers of the qth subchannel. And for any subchannel q , P _ISI ( q ) = P _I CI( q ).

本發明提出一種決定保護區間長度與位元負載的裝置，用於一多載波調變系統，該多載波調變系統的規格至少規定一第一保護區間以及一第二保護區間，其中第一保護區間的長度小於第二保護區間的長度，此裝置包括通道估測裝置、雜訊估測裝置、干擾估測裝置、位元負載決定裝置以及判斷裝置。通道估測裝置用以估測一通道脈衝響應；雜訊估測裝置用以估測第q個子通道不含ISI與ICI的雜訊能量；干擾估測裝置用以估測第q個子通道的干擾能量P _ISI(q)；位元負載決定裝置首先利用第q個子通道的信號能量以及第q個子通道的雜訊能量，根據第q個子通道的信號雜訊比，以計算第q個子通道所乘載的位元負載，完成所需的通道位元負載計算後所的到的結果，稱為第一位元負載。位元負載決定裝置其次利用所得之第q個子通道的干擾能量P _ISI(q)、第q個子通道的信號能量以及第q個子通道的雜訊能量，根據第q個子通道的信號雜訊比×P(q)]，以決定第q個子通道所乘載的位元負載，完成所需 的通道位元負載計算後所的到的結果，稱為第二位元負載；判斷裝置根據第一位元負載與第二保護區間的組合以判斷一第一位元產量，並根據第二位元負載與第一保護區間的組合，以判斷一第二位元產量，比較第一位元產量與第二位元產量以決定使用第一保護區間與第二位元負載或第二保護區間與第一位元負載。 The present invention provides a device for determining a guard interval length and a bit load, which is used in a multi-carrier modulation system. The specification of the multi-carrier modulation system defines at least a first protection interval and a second protection interval, wherein the first protection The length of the interval is smaller than the length of the second guard interval. The device includes a channel estimation device, a noise estimation device, an interference estimation device, a bit load determining device, and a determining device. The channel estimation device is used to estimate a channel impulse response; the noise estimation device is used to estimate the noise energy of the qth subchannel without ISI and ICI. ; Interference energy for estimating the q-th channel interference estimation means P _ISI (q); bit load signal energy using the apparatus first determines the q th channel And the noise energy of the qth subchannel According to the signal noise ratio of the qth subchannel To calculate the bit load carried by the qth subchannel, and complete the required channel bit load calculation, which is called the first bit load. Second load device using a bit decision of the interference energy of the resulting subchannels, q P _ISI (q), q-th signal channel energy And the noise energy of the qth subchannel According to the signal noise ratio of the qth subchannel × P ( q )], to determine the bit load carried by the qth sub-channel, to complete the required channel bit load calculation, the result is called the second bit load; the judging device according to the first The combination of the bit load and the second guard interval to determine a first bit yield, and according to the combination of the second bit load and the first guard interval, to determine a second bit yield, compare the first bit yield with The second bit yield is used to determine the use of the first guard interval and the second bit load or the second guard interval and the first bit load.

依照本發明的實施例所述之適應性決定保護區間長度與位元負載的裝置，上述干擾估測裝置包括傅立葉轉換器、頻帶選擇器以及干擾能量估測器。傅立葉轉換器用以將通道脈衝響應指定部分作傅立葉轉換，以輸出指定部分的頻率響應；頻帶選擇器耦接傅立葉轉換器，選擇所需之子頻帶q指定部分的頻率響應；干擾能量估測器耦接頻帶選擇器以及通道估測裝置，接收第q個子頻帶指定部分之頻率響應，以及通道估測裝置產生的通道響應估測值，以估測出第q個子頻帶的干擾能量P(q)。 The apparatus for determining the length of the guard interval and the bit load according to the adaptation described in the embodiment of the present invention, the interference estimating apparatus comprising a Fourier converter, a band selector, and an interference energy estimator. The Fourier converter is used to perform Fourier transform on the specified portion of the channel impulse response to output the frequency response of the specified portion; the band selector is coupled to the Fourier converter to select the frequency response of the specified portion of the desired subband q ; the interference energy estimator is coupled The band selector and the channel estimating device receive the frequency response of the specified portion of the qth subband and the channel response estimate generated by the channel estimating device to estimate the interference energy P ( q ) of the qth subband.

依照本發明的實施例所述之適應性決定保護區間長度與位元負載的裝置，上述干擾能量估測器包括選擇電路、減法器、第一乘法器、延遲電路、絕對值平方運算器、累加器以及第二乘法器。選擇電路包括一第一接收部分、一第二接收部分以及一輸出部分，第一接收部分耦接頻帶選擇器，接收第q個子頻帶之頻率響應的指定部份，用以選擇第一、第二接收部分其中之一輸出；減法器耦接輸出部分以及通道估測裝置，用以將輸出部分的所輸出之一特定值與通道估測裝置所輸出的一估測值相減後輸出一減法 值，其中特定值與估測值互相對應；第一乘法器用以將減法值乘以，以得到第一乘法值；延遲電路耦接第一乘法器與第二接收部分，接收第一乘法值，將第一乘法值延遲一預設時間後輸出；絕對值平方運算器耦接選擇電路輸出部分，將輸出部分所輸出的特定值取絕對值並且平方後輸出一頻率響應能量值；累加器耦接絕對值平方運算器，將所接收到的頻率響應能量值作累加運算，輸出一累加頻率響應能量值；第二乘法器將累加頻率響應能量值乘上一子載波平均傳送能量，輸出第q個子頻帶的干擾能量P(q)。 The apparatus according to the embodiment of the present invention determines a guard interval length and a bit load, and the interference energy estimator includes a selection circuit, a subtractor, a first multiplier, a delay circuit, an absolute value square operator, and an accumulation. And a second multiplier. The selection circuit includes a first receiving portion, a second receiving portion and an output portion, the first receiving portion is coupled to the band selector, and receives a designated portion of the frequency response of the qth sub-band for selecting the first and second Receiving one of the output portions; the subtractor is coupled to the output portion and the channel estimating device for subtracting one of the output values of the output portion from an estimated value output by the channel estimating device, and outputting a subtraction value Where the specific value corresponds to the estimated value; the first multiplier is used to multiply the subtraction value by Obtaining a first multiplication value; the delay circuit is coupled to the first multiplier and the second receiving portion, receiving the first multiplication value, delaying the first multiplication value by a predetermined time, and outputting; the absolute value square operator is coupled to the selection circuit In the output part, the specific value outputted by the output part is taken as an absolute value and squared to output a frequency response energy value; the accumulator is coupled to the absolute value square operator, and the received frequency response energy value is accumulated, and the output is accumulated. The frequency response energy value; the second multiplier multiplies the accumulated frequency response energy value by a subcarrier average transmission energy, and outputs the interference energy P ( q ) of the qth subband.

依照本發明的實施例所述之適應性決定保護區間長度與位元負載的裝置，上述傅立葉轉換器僅需運作一次並由頻帶選擇棄輸出指定部分的頻率響應H _K+1 (q)，其餘H _m (q)，m=K+2~L-1皆由干擾能量估測器運算而得，可減少硬體運算量與功耗。 The apparatus for determining the length of the guard interval and the bit load according to the embodiment of the present invention, the Fourier converter only needs to be operated once and the frequency response H _{K + 1} (q) of the specified portion is discarded by the frequency band selection, and the rest H _m (q) , m = K +2~ L -1 are all calculated by the interference energy estimator, which can reduce the amount of hardware calculation and power consumption.

依照本發明的實施例所述之決定保護區間長度與位元負載的裝置，第一保護區間長度為K，通道脈衝響應的長度為L，通道脈衝響應為h(0)~h(L-1)，L>K，第q個子頻帶之指定部分頻率響應為H _K+1 (q)，其中H _K+1 (q)為h(K+1)~h(L-1)作傳立葉轉換後的第q個子頻帶的頻率響應。P(q)為符元間干擾的能量P _ISI(q)，且符元間干擾的能量等於載波間干擾的能量P _ICI(q)。 According to the embodiment of the present invention, the apparatus for determining the length of the guard interval and the bit load, the length of the first guard interval is K , the length of the channel impulse response is L , and the channel impulse response is h ( 0 )~ h ( L - 1 ), L > K , the specified partial frequency response of the qth sub-band is H _{K + 1} (q) , where H _{K + 1} (q) is h ( K +1)~ h ( L -1) for the Fourier transform The frequency response of the qth sub-band after. P ( q ) is the energy P _ISI ( q ) of the inter-symbol interference, and the energy of the inter-symbol interference is equal to the energy P _ICI ( q ) of the inter-carrier interference.

本發明提出一種干擾估測裝置，用於一多載波調變系統，此多載波調變系統的規格至少規定一第一保護區間，其中第一保護區間的長度為K，此裝置包括通道估測單元、傅立葉轉換器、頻帶選擇器以及干擾能量估測器。通 道估測單元用以估測通道脈衝響應h(0)~h(L-1)；傅立葉轉換器用以將通道脈衝響應的指定部分h(K+1)~h(L-1)作傅立葉轉換，以輸出多個子頻帶指定部分之頻率響應H _K+1(0)~H _K+1(N-1)；頻帶選擇器耦接傅立葉轉換器，用以選擇第q個子頻帶指定部分之頻率響應H _K+1(q)；干擾能量估測器耦接頻帶選擇器以及通道估測裝置，接收第q個子頻帶指定部分之頻率響應H _K+1(q)以及通道脈衝響應h(K+1)~h(L-2)，據以計算出H _K+2(q)~H _L-1(q)，並估測出第q個子頻帶的干擾能量P(q)。 The present invention provides an interference estimation apparatus for a multi-carrier modulation system. The specification of the multi-carrier modulation system defines at least a first protection interval, wherein the length of the first protection interval is K , and the device includes channel estimation. Unit, Fourier converter, band selector, and interference energy estimator. The channel estimation unit is used to estimate the channel impulse response h ( 0 )~ h ( L -1); the Fourier converter is used to perform the Fourier transform on the specified portion h ( K +1)~ h ( L -1) of the channel impulse response. a frequency response H _{K +1} ( 0 )~ H _{K +1} ( N - 1 ) for outputting a plurality of sub-band designated portions; the band selector is coupled to the Fourier converter for selecting a frequency response of the designated portion of the q- th sub-band H _{K +1} ( q ); the interference energy estimator is coupled to the band selector and the channel estimation device, and receives the frequency response H _{K +1} ( q ) of the q- th sub-band specified portion and the channel impulse response h ( K +1 ) ~ h ( L -2), from which H _{K +2} ( q )~ H _{L -1} ( q ) is calculated, and the interference energy P ( q ) of the qth sub-band is estimated.

依照本發明的實施例所述之干擾估測裝置，上述干擾能量估測器包括選擇電路、減法器、第一乘法器、延遲電路、絕對值平方運算器、累加器以及第二乘法器。選擇電路包括一第一接收部分、一第二接收部分以及一輸出部分，第一接收部分耦接頻帶選擇器，接收H _K+1(q)，用以選擇第一、該第二接收部分其中之一輸出；減法器耦接輸出部分以及通道估測裝置，用以將輸出部分的所輸出之一特定值與通道估測裝置所輸出的一估測值相減後輸出一減法值，其中特定值與估測值互相對應；第一乘法器用以將減法值乘以，以得到一第一乘法值；延遲電路耦接第一乘法器與第二接收部分，接收第一乘法值，將第一乘法值延遲一預設時間後輸出；絕對值平方運算器耦接選擇電路輸出部分，將輸出部分所輸出的特定值取絕對值並且平方後輸出一頻率響應能量值；累加器耦接絕對值平方運算器，將所接收到的頻率響應能量值作累加運算，輸出一累加頻率響應能量值；第二乘法器將累加頻率響應能量值乘 上一子載波平均傳送能量，輸出第q個子頻帶的干擾能量P(q)。 According to the interference estimating apparatus according to the embodiment of the present invention, the interference energy estimator includes a selection circuit, a subtractor, a first multiplier, a delay circuit, an absolute value square operator, an accumulator, and a second multiplier. The selection circuit includes a first receiving portion, a second receiving portion and an output portion, the first receiving portion is coupled to the band selector, and receives H _{K +1} ( q ) for selecting the first and the second receiving portion. One output; the subtractor is coupled to the output portion and the channel estimating device for subtracting one of the output values of the output portion from an estimated value output by the channel estimating device, and outputting a subtraction value, wherein the specific value is specified The value corresponds to the estimated value; the first multiplier is used to multiply the subtraction value by And obtaining a first multiplication value; the delay circuit is coupled to the first multiplier and the second receiving portion, receives the first multiplication value, delays the first multiplication value by a predetermined time, and outputs; the absolute value square operator is coupled and selected. The output part of the circuit takes an absolute value of the output value of the output part and squares it to output a frequency response energy value; the accumulator is coupled with the absolute value square operator, and accumulates the received frequency response energy value, and outputs one The frequency response energy value is accumulated; the second multiplier multiplies the accumulated frequency response energy value by a subcarrier average transmission energy, and outputs the interference energy P ( q ) of the qth subband.

依照本發明的實施例所述之干擾估測裝置，上述P(q)為符元間干擾的能量P _ISI(q)，且符元間干擾的能量等於載波間干擾的能量P _ICI(q)。 According to the interference estimating apparatus according to the embodiment of the present invention, the P ( q ) is the inter-symbol interference energy P _ISI ( q ), and the energy of the inter-symbol interference is equal to the inter-carrier interference energy P _ICI ( q ) .

本發明在通道脈衝響應的長度介於一第一保護區間與第二保護區間的長度之間的情況下，採用判斷使用第一保護區間的符元之位元產量與使用第二保護區間的符元之位元產量，以決定使用第一保護區間或第二保護區間作為符元之保護區間；另外，本發明實施例透過數學與模擬分析出符元間干擾以及載波間干擾，藉以作出決定保護區間長度與位元負載的裝置以及干擾估測裝置，因此除了可以達到「根據通道長度以及干擾之大小，以決定循環前置長度以及各個子載波所乘載之資料位元量」以及「估測符元間干擾以及載波間干擾」，還可以達到增加位元產量與節省成本的目的。 In the case where the length of the channel impulse response is between the lengths of the first guard interval and the second guard interval, the bit yield of the symbol using the first guard interval and the sign using the second guard interval are used. The bit yield of the element is determined to use the first guard interval or the second guard interval as the guard interval of the symbol; in addition, the embodiment of the present invention analyzes the inter-symbol interference and the inter-carrier interference through mathematical and analog, thereby making a decision protection Interval length and bit load devices and interference estimation devices, in addition to "according to the channel length and the size of the interference, to determine the loop pre-length and the amount of data bits carried by each sub-carrier" and "estimate Inter-symbol interference and inter-carrier interference can also achieve the goal of increasing bit yield and cost.

為讓本發明之上述和其他目的、特徵和優點能更明顯易懂，下文特舉實施例，並配合所附圖式，作詳細說明如下。 The above and other objects, features and advantages of the present invention will become more <RTIgt;

為了解決習知的問題，並且增加資料的傳送/接收效率，本發明提出了一種適應性決定保護區間長度與位元負載的方法與裝置。在說明此方法與裝置之前，先分析通道長度比GI長的情況下所產生之ISI以及ICI。 In order to solve the conventional problems and increase the transmission/reception efficiency of data, the present invention proposes a method and apparatus for adaptively determining the length of the guard interval and the load of the bit. Before describing the method and device, the ISI and ICI generated when the channel length is longer than GI are analyzed.

圖2A以及圖2B繪示為符元的防護區間201之長度小 於通道脈衝響應202的長度時所發生的ISI示意圖。當符元加上防護區間後，透過通道傳送到接收端，接收端所接收到的信號會等於符元與通道脈衝響應的線性捲積(linear convolution)。由於接收端所接收到的信號會等於符元與通道脈衝響應的線性捲積，當防護區間比通道長度短時，則正在接收的符元便會有前次所接收的符元之成分存在，因而會造成ISI。在圖2A中可以看出，防護區間201的長度為K，通道脈衝響應202的長度為L，符元的時域信號(time-domain signal)表示為x(0)~x(-K-1)，通道脈衝響應202表示為h(0)~h(L-1)，因此在T0時間捲積的結果「y(0)」可表示如下：y(0)=x(0)×h(0)+x(-1)×h(1)+...+x(-K-1)×h(K+1)+...+x(-L+1)×h(L-1) 2A and 2B are schematic diagrams showing the ISI that occurs when the length of the guard interval 201 of the symbol is less than the length of the channel impulse response 202. When the symbol is added to the guard interval, it is transmitted to the receiving end through the channel, and the signal received by the receiving end is equal to the linear convolution of the symbol and the channel impulse response. Since the signal received by the receiving end is equal to the linear convolution of the symbol and the channel impulse response, when the guard interval is shorter than the channel length, the symbol being received will have the component of the previously received symbol. This will result in ISI. As can be seen in FIG. 2A, the length of the guard interval 201 is K , the length of the channel impulse response 202 is L , and the time-domain signal of the symbol is represented as x(0) ~ x(-K-1 ) , the channel impulse response 202 is represented as h(0) ~ h(L-1) , so the result of convolution at time T0 " y (0)" can be expressed as follows: y (0) = x (0) × h ( 0)+ x (-1)× h (1)+...+ x (- K -1)× h ( K +1)+...+ x (- L +1)× h ( L -1 )

上述式子中，具有h(K+1)~h(L-1)的部分便為受到前一時間的符元所影響的部分，因此在T0時間的ISI便可表示如下：y _ISI(0)=x(-K-1)×h(K+1)+x(-K-2)×h(K+2)+...+x(-L+1)×h(L-1) In the above formula, the part with h ( K +1)~h( L -1) is the part affected by the symbol of the previous time, so the ISI at time T0 can be expressed as follows: y _ISI (0 ) = x (- K -1) × h ( K +1) + x (- K -2) × h ( K +2) +... + x (- L +1) × h ( L -1)

同樣的，在圖2B中可以看出，在T1時間的ISI可表示如下：y _ISI(1)=x(-K-1)×h(K+2)+x(-K-2)×h(K+3)+...+x(-L+2)×h(L-1) Similarly, as can be seen in Figure 2B, the ISI at time T1 can be expressed as follows: y _ISI (1) = x (- K -1) × h ( K +2) + x (- K -2) × h ( K +3)+...+ x (- L +2)× h ( L -1)

在此，假設所欲接收的符元有N個子載波，因此接收信號在不同時間點n的ISI可以表示如下：y _ISI(n)=x(-K-1)×h(K+1+n)+x(-K-2)×h(K+2+n)+...+x(-L+1+n)×h(L-1),n=0,1,...,N-1 Here, it is assumed that the symbol to be received has N subcarriers, so the ISI of the received signal at different time points n can be expressed as follows: y _ISI ( n ) = x (- K -1) × h ( K +1 + n ) + x (- K -2) × h ( K +2+ n )+...+ x (- L +1+n)× h ( L -1), n =0,1,..., N -1

將上述傳送符元表示為x _i ，干擾x _i 的符元表示為x _i-1，因此接收信號y _i 便可以表示如下：y _i =C _i x _i +C _i-1 x _i-1，其中C _i-1與C _i 分別表示如下的N×N矩陣： 上述數學式中，h ₀ 相當於上述的h(0)，h _K 相當於上述的h(K)，以此類推；y _i =[y _i (0),y _i (1),...,y _i (N-1)]^T為第i個接收符元，對應傳送符元的信號部分為x _i =[x _i (0),x _i (1),...,x _i (N-1)]^T，ISI部分為x _i-1=[x _i-1(0),x _i-1(1),...,x _i-1(N-1)]^T。根據上述的數學式可以看出，由於通道脈衝響應過長，導致線性捲積的結果會有一部分捲積到符元x _i-1，因而使得接收信號y _i 會有ISI。上述的數學式可以明顯看出接收信號y _i 的ISI成分y _ISI=C _i-1 x _i-1。 The above-mentioned transmission symbol is represented as x _i , and the symbol of the interference x _i is represented as x _{i -1} , so the received signal y _i can be expressed as follows: y _i = C _i x _i + C _{i -1} x _{i -1} , Where C _{i -1} and C _i represent the following N × N matrices, respectively: In the above mathematical formula, h ₀ corresponds to the above h ( 0 ), h _K corresponds to the above h ( K ), and so on; y _i =[ y _i (0), y _i (1),... , y _i ( N -1)] ^T is the ith received symbol, and the signal portion corresponding to the transmitted symbol is x _i =[ x _i (0), x _i (1),..., x _i ( N -1)] ^T , the ISI part is x _{i -1} =[ x _{i -1} (0), x _{i -1} (1),..., x _{i -1} ( N -1)] ^T . It can be seen from the above mathematical formula that since the channel impulse response is too long, a result of the linear convolution will be partially convoluted to the symbol x _{i -1} , so that the received signal y _i will have ISI. The above formula can clearly see the ISI component y _ISI = C _{i -1} x _{i -1 of the} received signal y _i .

接下來，請仔細觀察矩陣C _i ，在矩陣C _i 的第一列缺少了h _K+1~h _L-1，其第二列缺少了h _K+2~h _L-1...。而在理想的情況下，也就是當防護區間足夠長的情況，接收信號的線性捲積應當等於通道脈衝響應與符元x _i 的循環捲積(circular convolution)，表示如下：理想的y _i =C _CYCL x _i ，其中因此， y _i =C _i x _i +C _i-1 x _i-1=C_CYCL x _i +C _i-1 x _i-1-C_ICI x _i ，其中， 而C _ICI x _i 便是接收信號y _i 的載波間干擾(ICI)的成分。 Next, please carefully observe the matrix C _i , the first column of the matrix C _i lacks h _{K +1} ~ h _{L -1} , and the second column lacks h _{K +2} ~ h _{L -1} .... In the ideal case, that is, when the guard interval is long enough, the linear convolution of the received signal should be equal to the circular convolution of the channel impulse response and the symbol x _i , expressed as follows: ideal y _i = C _CYCL x _i , of which, therefore, y _i = C _i x _i + C _{i -1} x _{i -1} = C _CYCL x _i + C _{i -1} x _{i -1} -C _ICI x _i , where C _ICI x _i is the component of the inter-carrier interference (ICI) of the received signal y _i .

接下來，對y _i =C _CYCL x _i +C _i-1 x _i-1-C _ICI x _i 作傅立葉轉換，得到Y _i =W _N C _CYCL x _i +W _N C _ISI x _i-1-W _N C _ICI x _i =Y _Signal+Y _ISI+Y _ICI，其中W _N 為傅立葉矩陣，表示如下： 而第q個子載波(tone)在 頻域上的函數可以表示如下： Y(q)=w _q C _CYCL x _i +w _q C _ISI x _i-1-w _q C _ICI x _i Next, Fourier transform is performed on y _i = C _CYCL x _i + C _{i -1} x _{i -1} - C _ICI x _i to obtain Y _i = W _N C _CYCL x _i + W _N C _ISI x _{i -1} - W _N C _ICI x _i = Y _Signal + Y _ISI + Y _ICI , where W _N is a Fourier matrix, expressed as follows: The function of the qth subcarrier in the frequency domain can be expressed as follows: Y ( q ) = w _q C _CYCL x _i + w _q C _ISI x _{i -1} - w _q C _ICI x _i

接下來，利用上述的Y(q)，將其取共扼之後，取期望值，假設每一傳送符元中各點互不相依且平均能量為，並假設不同符元間各點互不相依(independent)， 由此可得第q個子載波(tone)在頻譜上的能量為： 其中，{˙} ^H 表示矩陣或向量的Hermitian。上式中，表示接收端所接收到第q個子載波信號部分的能量；表示第q個子載波收到的ISI能量，表示第q個子載波收到的ICI能量；其餘的表示C _ICI與C _CYCL交互影響的能量，其中上述交互影響的能量可以忽略不計(在此先假設可以忽略不計，詳細的原因容後詳述)。接下來，上述的W _N C _ICI可以表示如下： 其中[H _m (0),...H _m (N-1)]^T表示通道脈衝響應[h _m ,...,h _L-1,0,...,0]^T經傅立葉轉換的結果。故ICI在第q個子載波的干擾能量為： 同樣的，ISI在第q個子載波的干擾能量為： 從上式得知在每一個子載波q的ISI能量會與ICI的能量相同，即P _ISI(q)=P _ICI(q)。另外，依照上述的數學證明可以看出，只要估測出通道脈衝響應，便可以估測出在防護區間小於通道脈衝響應長度時，每一個子載波受到ISI與ICI干擾P _ISI(q)與P _ICI(q)。 Next, using the above Y ( q ), after taking the total, take the expected value, assuming that each point in each transmission symbol is not dependent and the average energy is And assume that the points between different symbols are independent of each other, Thus, the energy of the qth subcarrier (tone) in the spectrum is: Where { ̇ } ^H represents the Hermitian of the matrix or vector. In the above formula, Representing the energy of the qth subcarrier signal portion received by the receiving end; Indicates the ISI energy received by the qth subcarrier, Indicates the ICI energy received by the qth subcarrier; the rest represents the energy affected by the interaction between C _ICI and C _CYCL , where the energy of the above interaction is negligible (the assumption is negligible here, the detailed reason is detailed later) . Next, the above W _N C _ICI can be expressed as follows: Where [ H _m (0),... H _m ( N -1)] ^T represents the channel impulse response [ h _m ,..., h _{L -1} ,0,...,0] ^T is Fourier transformed result. Therefore, the interference energy of the ICI on the qth subcarrier is: Similarly, the interference energy of ISI on the qth subcarrier is: It is known from the above equation that the ISI energy at each subcarrier q will be the same as the ICI energy, that is, P _ISI ( q ) = P _ICI ( q ). In addition, according to the above mathematical proof, it can be seen that as long as the channel impulse response is estimated, it can be estimated that each subcarrier is subjected to ISI and ICI interference P _ISI ( q ) and P when the guard interval is smaller than the channel impulse response length. _ICI ( q ).

因此，在本發明的實施例中，提出了決定保護區間長度與位元負載的裝置。如圖3所示，此決定保護區間長度與位元負載的裝置包括通道估測裝置301、本發明實施例的干擾估測裝置302、位元決定裝置303以及判斷裝置304。在此便以上述Home-Plug AV的規格為基礎以說明以下實施例，其中Home-Plug AV具有3種長度的保護區間， 分別是GI-417、GI-567以及GI-3534。 Therefore, in an embodiment of the present invention, an apparatus for determining a guard interval length and a bit load is proposed. As shown in FIG. 3, the apparatus for determining the length of the guard interval and the bit load includes the channel estimating means 301, the interference estimating means 302 of the embodiment of the present invention, the bit determining means 303, and the determining means 304. Herein, the following embodiment will be described based on the specifications of the Home-Plug AV described above, wherein the Home-Plug AV has three lengths of protection intervals. They are GI-417, GI-567 and GI-3534.

通道估測裝置301會先估測通道之通道脈衝響應。在此假設此通道估測裝置301所估測出的通道脈衝響應之長度(通道長度)介於GI-567與GI-3534之間，干擾估測裝置302便開始估測在使用GI-567作為防護區間的情況下，第q個子通道的符元間干擾能量P _ISI(q)大小。位元負載決定裝置303接收第q個子通道的符元間干擾能量P _ISI(q)、第q個子通道的接收信號能量以及第q個子通道的雜訊能量，根據第q個子通道的信號雜訊比，以決定第q個子通道所能乘載的位元負載。透過上述的裝置303，便可以估測出每一子通道在受到ISI與ICI干擾的情況下，可以乘載的位元量。 The channel estimation device 301 first estimates the channel impulse response of the channel. It is assumed here that the channel impulse response estimated by the channel estimation device 301 (channel length) is between GI-567 and GI-3534, and the interference estimation device 302 starts to estimate the use of GI-567 as In the case of the guard interval, the inter-symbol interference energy P _ISI ( q ) of the qth sub-channel. Symbol energy difference between the received signal power P _ISI interference bit decision means 303 receives the load q-th channel (q), q-th sub-channel And the noise energy of the qth subchannel According to the signal noise ratio of the qth subchannel To determine the bit load that the qth subchannel can ride. Through the above-mentioned device 303, it is possible to estimate the amount of bits that can be carried in each sub-channel under the interference of ISI and ICI.

一般來說，本領域具有通常知識者若要實施位元負載決定裝置303，最基本的方法就是利用查找表以及比大小的方式。底下是Home-Plug AV中，編碼率為1/2時的信號雜訊比與子載波調變的查找表。 In general, if the person skilled in the art has to implement the bit load decision means 303, the most basic method is to use a lookup table and a size ratio. Below is a lookup table for signal-to-noise ratio and subcarrier modulation at a coding rate of 1/2 in Home-Plug AV.

上述的查找表主要是表示，當信號雜訊比SNR大於21dB時，則子載波所使用的調變可以使用1024QAM，當信號雜訊比SNR介於16.25dB與21dB之間時，則子載波所使用的調變使用256QAM，以此類推。當子載波使用1024QAM調變時，即表示此子載波可乘載10bits位元量； 當子載波使用BPSK調變時，即表示此子載波可乘載1bit位元量。 The above lookup table mainly indicates that when the signal noise ratio SNR is greater than 21 dB, the modulation used by the subcarrier can use 1024QAM, and when the signal noise ratio SNR is between 16.25 dB and 21 dB, the subcarrier is used. Modulation uses 256QAM, and so on. When the subcarrier is modulated by 1024QAM, it means that the subcarrier can be multiplied by 10 bits. When the subcarrier uses BPSK modulation, it means that the subcarrier can be multiplied by 1 bit.

最後，透過判斷裝置304評估在使用GI-567作為防護區間的情況下的位元產量以及在使用GI-3534作為防護區間的情況下的位元產量，以決定使用GI-567或GI-3534作為符元之保護區間。然而，基於上述的情況，使用GI-567時每個符元所能乘載的總位元量顯然會小於使用GI-3534的符元所能乘載的總位元量，但是，使用GI-3534的符元的傳輸時間將近使用GI-567的符元的傳輸時間的2倍。因此，在考慮位元產量，除了必須考慮到符元所能乘載的總位元量之外，還必須考慮到符元的傳輸時間等其他因素。 Finally, through the judging means 304, the bit yield in the case of using the GI-567 as the guard interval and the bit yield in the case of using the GI-3534 as the guard interval are evaluated to decide whether to use the GI-567 or the GI-3534 as the bit yield. The protection interval of the symbol. However, based on the above situation, the total number of bits that can be multiplied by each symbol when using GI-567 is obviously smaller than the total number of bits that can be multiplied by symbols using GI-3534, but using GI- The transmission time of the symbol of 3534 is nearly twice that of the symbol of GI-567. Therefore, in consideration of the bit yield, in addition to the total number of bits that the symbol can be loaded, other factors such as the transmission time of the symbol must be considered.

圖4繪示為本發明實施例圖3的干擾估測裝置302之電路方塊圖。請參考圖4，此干擾估測裝置302包括傅立葉轉換器401、頻帶選擇器402、干擾能量估測器403。首先，傅立葉轉換器401會先將通道脈衝響應h(K+1)~h(L-1)作傅立葉轉換，得到H _K+1(0)~H _K+1(N-1)，在此實施例中，上述H _K+1(0)~H _K+1(N-1)為在第K+1取樣點之後的通道脈衝響應在個別子通道上的頻率響應。之後，頻帶選擇器402便會選擇其中一個子頻帶(在此實施例以第q個子頻帶作為例子)的頻率響應H _K+1(q)給干擾能量估測器403。干擾能量估測器403利用H _K+1(q)計算得到H _K+2(q)~H _L-1(q)，並將H _K+1(q)~H _L-1(q)分別取絕對值並平方運算，之後累加|H _m (q)|²,m=K+1,...,L-1，最後再乘上每個子載波的平均能量 ，便得到第q個子頻帶的干擾能量(P _ISI(q)或P _ICI(q))。 4 is a circuit block diagram of the interference estimation apparatus 302 of FIG. 3 according to an embodiment of the present invention. Referring to FIG. 4, the interference estimation device 302 includes a Fourier converter 401, a band selector 402, and an interference energy estimator 403. First, the Fourier converter 401 first performs a Fourier transform on the channel impulse response h ( K +1)~ h ( L -1) to obtain H _{K +1} (0)~ H _{K +1} ( N -1), where In an embodiment, the above H _{K +1} (0)~ H _{K +1} ( N -1) is the frequency response of the channel impulse response after the K +1th sampling point on the individual subchannels. Thereafter, the band selector 402 selects the frequency response H _{K +1} ( q ) of one of the sub-bands (in this embodiment, the q- th sub-band as an example) to the interference energy estimator 403. Interference energy estimator 403 using H _{K +1} (q) calculated _{H K +2 (q) ~ H} L -1 (q), and _{H K +1 (q) ~ H} L -1 (q) , respectively Take the absolute value and square the operation, then accumulate | H _m ( q )| ² , m = K +1,..., L -1, and finally multiply the average energy of each subcarrier The interference energy ( P _ISI ( q ) or P _ICI ( q )) of the qth sub-band is obtained.

在上面圖4的例子中，值得注意的是在傅立葉轉換器401所輸出的頻率響應僅第H _K+1(0)~H _K+1(N-1)的部分，另外，通道估測裝置301有輸出h(l)給干擾能量估測器403。由於第q個子頻帶的通道脈衝響應H _m (q)可以如下式表示： 接下來，我們把上面式子的前三項表示如下： 因此可以看出： 也就是說，H _K+2 (q)可以透過H _K+1 (q)與未經轉換的通道脈衝響應h(K+1)相減後乘上得到，故上述的干擾能量估測器403之電路便可以如圖5的方式實施，圖5繪示為本發明實施例之干擾能量估測器403的較詳細電路方塊圖。請參考圖5，此電路包括減法器501、第一乘法器502、延遲電路503、選擇電路504、絕對值平方運算器505、累加器506以及第二乘法器507。上述干擾能量估測器403內部元 件的耦接請參照圖5的繪示。 In the example of FIG. 4 above, it is worth noting that the frequency response outputted by the Fourier converter 401 is only the portion of the H _{K +1} (0) to H _{K +1} ( N -1), and the channel estimation device 301 has an output h ( l ) to the interference energy estimator 403. Since the channel impulse response H _m ( q ) of the qth sub-band can be expressed as follows: Next, we put the first three terms of the above expression as follows: So it can be seen that: In other words, H _{K + 2} (q) can be subtracted from the unconverted channel impulse response h ( K +1) by H _{K + 1} (q) The circuit of the interference energy estimator 403 can be implemented as shown in FIG. 5. FIG. 5 is a block diagram showing a more detailed circuit of the interference energy estimator 403 according to an embodiment of the present invention. Referring to FIG. 5, the circuit includes a subtractor 501, a first multiplier 502, a delay circuit 503, a selection circuit 504, an absolute value square operator 505, an accumulator 506, and a second multiplier 507. For the coupling of the internal components of the interference energy estimator 403, please refer to FIG. 5 .

在初始時，選擇電路504的第一接收部會從傅立葉轉換器401接收到，之後選擇電路504會選擇其第一端所接收的H _K+1 (q)輸出。之後H _K+1 (q)會送到減法器501以及絕對值平方運算器505。減法器501將H _K+1 (q)和通道估測裝置301所送出的h(K+1)相減後，將結果送給第一乘法器502。之後，乘法器502便將上述減法器501所輸出的結果與相乘，也就得到了H _K+2 (q)。接下來，延遲電路503將所接收到的H _K+2 (q)延遲一預定時間後，送給選擇電路504的第二接收部，因此選擇電路可以直接選擇第二接收部所輸出的H _K+2 (q)給之後的絕對值平方運算器505、累加器506以及第二乘法器507作之後的運算。重複上述的動作，便可以得到第q個子頻帶的干擾能量(P _ISI(q)或P _ICI(q))。 Initially, the first receiving portion of the selection circuit 504 is received from the Fourier converter 401, after which the selection circuit 504 selects the H _{K + 1} (q) output received by its first end. H _{K + 1} (q) is then sent to subtractor 501 and absolute square operator 505. The subtracter 501 subtracts H _{K + 1} (q) from h ( K +1) sent from the channel estimation device 301, and sends the result to the first multiplier 502. Then, the multiplier 502 outputs the result output by the subtractor 501 described above. Multiply, you get H _{K + 2} (q) . Next, the delay circuit 503 delays the received H _{K + 2} (q) for a predetermined time and then sends it to the second receiving portion of the selection circuit 504. Therefore, the selection circuit can directly select the H _K output by the second receiving unit _{. + 2} (q) The subsequent operations of the absolute value square operator 505, the accumulator 506, and the second multiplier 507 are performed. By repeating the above actions, the interference energy ( P _ISI ( q ) or P _ICI ( q )) of the qth sub-band can be obtained.

值得一提的是，由於上述實施例的干擾能量估測器403只需要接收H _K+1 (q)，便可以運算出H _K+2 (q)~H _L-1 (q)，因此使用此干擾能量估測器403來達成本發明之實施例可以減少傅立葉轉換器401的使用頻率，更可以進一步地與其他模組(例如OFDM解調器)共用傅立葉轉換器401的設計，減少電路面積與功率消耗。 It is worth mentioning that since the interference energy estimator 403 of the above embodiment only needs to receive H _{K + 1} (q) , H _{K + 2} (q) ~ H _{L - 1} (q) can be calculated, so The interference energy estimator 403 to achieve the embodiment of the present invention can reduce the frequency of use of the Fourier converter 401, and can further share the design of the Fourier converter 401 with other modules (such as an OFDM demodulator) to reduce the circuit area. With power consumption.

另外，本發明提出一種決定保護區間長度與位元負載的方法，其流程圖繪示於圖6。請參考圖6，首先，提供一多載波調變系統，此多載波調變系統的規格規定至少一第一保護區間以及一第二保護區間，其中第一保護區間的長 度小於第二保護區間的長度，例如：GI-567與GI-3534(步驟601)。下一步驟，在通道閒置期間，估測每一子通道之雜訊能量(步驟602)。接著，於信號來臨時，估測通道的脈衝響應(步驟603)。之後，計算每一子載波的第一位元負載(步驟604)。接著，判斷通道長度是否小於第一保護區間(步驟605)。若通道長度小於第一保護區間，則決定使用第一位元負載與第一保護區間(步驟606)。若通道長度大於或等於第一保護區間，則估測每一個子通道使用第一保護區間時的ICI與ISI之能量，並得出信號雜訊比與每一個子載波所能乘載的第二位元負載(步驟607)。之後，判斷使用第一保護區間配合第二位元負載之符元的位元產量是否大於第二保護區間配合第一位元負載的位元產量(步驟608)。若第一保護區間配合第二位元負載之符元的位元產量大於第二保護區間配合第一位元負載的位元產量，則決定使用第二位元負載與第一保護區間(步驟609)。若第一保護區間配合第二位元負載之符元的位元產量小於或等於第二保護區間配合第一位元負載的位元產量，則決定使用第二位元負載與第一保護區間(步驟610)。 In addition, the present invention proposes a method for determining the length of the guard interval and the load of the bit, the flow chart of which is shown in FIG. Please refer to FIG. 6. First, a multi-carrier modulation system is provided. The specification of the multi-carrier modulation system defines at least a first protection interval and a second protection interval, wherein the length of the first protection interval is long. The degree is less than the length of the second guard interval, for example: GI-567 and GI-3534 (step 601). In the next step, the noise energy of each subchannel is estimated during the idle period of the channel (step 602). Next, the impulse response of the channel is estimated as soon as the signal is present (step 603). Thereafter, the first bit payload of each subcarrier is calculated (step 604). Next, it is determined whether the channel length is smaller than the first guard interval (step 605). If the channel length is less than the first guard interval, then the first bit load and the first guard interval are determined to be used (step 606). If the channel length is greater than or equal to the first guard interval, estimate the energy of ICI and ISI when each subchannel uses the first guard interval, and obtain a signal noise ratio and a second carrier that can be multiplied by each subcarrier. The bit load (step 607). Thereafter, it is determined whether the bit yield of the symbol using the first guard interval and the second bit load is greater than the bit yield of the second guard interval in conjunction with the first bit load (step 608). If the bit yield of the symbol of the first guard interval and the symbol of the second bit load is greater than the bit yield of the second guard interval and the first bit load, then determining to use the second bit load and the first guard interval (step 609) ). If the bit yield of the symbol of the first guard interval and the symbol of the second bit load is less than or equal to the bit yield of the second guard interval and the first bit load, then determining to use the second bit load and the first guard interval ( Step 610).

步驟604主要是估測在沒有ISI或ICI的情況下，符元可以乘載的最大位元負載，也就是估測每一個子載波在上述沒有ISI或ICI時的信號雜訊比(SNR)來決定可以乘載的最大位元負載，其中，每一個子通道的雜訊能量與通道的脈衝響應是在前些步驟602與603時獲得。 Step 604 is mainly to estimate the maximum bit load that the symbol can be loaded without ISI or ICI, that is, to estimate the signal to noise ratio (SNR) of each subcarrier without the ISI or ICI mentioned above. The maximum bit load that can be loaded is determined, wherein the noise energy of each subchannel and the impulse response of the channel are obtained in the previous steps 602 and 603.

在一個OFDM系統中，接收端與傳送端為了要做同步時，傳送端可能連續傳送相同且接收端已知的符元，因此可估測出通道頻率響應與接收信號能量，並利用通道閒置期間所估得的雜訊能量，此時所估測出的SNR可以說是「沒有ISI或ICI時的SNR」。另外，習知技術中也有使用前導信號(PILOT)來估測SNR，故本發明不限於此，且符合本發明之精神者，當在本發明的保護範圍內。 In an OFDM system, when the receiving end and the transmitting end are to synchronize, the transmitting end may continuously transmit the same symbol known to the receiving end, so that the channel frequency response and the received signal energy can be estimated, and the channel idle period is utilized. The estimated noise energy, the estimated SNR at this time can be said to be "the SNR without ISI or ICI". In addition, the prior art also uses a preamble signal (PILOT) to estimate the SNR, and the present invention is not limited thereto, and is within the protection scope of the present invention.

上述步驟604在本發明的一特定實施例包括下列子步驟：估測每一子通道的信號能量(步驟604.1)。根據子通道的雜訊能量(於步驟602時獲得)以及子通道的信號能量，計算出每一子載波的信號雜訊比，並決定每一子載波所能乘載的該第一位元負載(步驟604.2)。上述步驟雖然是以通道閒置(silent)期間所估測到的雜訊能量來計算信號雜訊比，然本領域具有通常知識者若要做到估測信號雜訊比，並不一定要在通道閒置期間估測，之前已經提過幾種方法，故在此不予贅述。 The above-described step 604 includes a sub-step of estimating a signal energy of each sub-channel (step 604.1) in a particular embodiment of the invention. Calculating the signal to noise ratio of each subcarrier according to the noise energy of the subchannel (obtained at step 602) and the signal energy of the subchannel, and determining the first bit load that can be carried by each subcarrier. (Step 604.2). Although the above steps calculate the signal noise ratio based on the noise energy estimated during the silent period of the channel, the general knowledge in the field is not necessarily in the channel if the estimated signal noise ratio is to be estimated. Estimated during the idle period, several methods have been mentioned before, so I will not repeat them here.

於步驟607中，會先估測出每個子通道使用第一保護區間時的ICI與ISI之能量，並在ICI與ISI的情況下，估計一符元加上第一保護區間之後的信號雜訊比，其計算方法表示如下： ，其中P _ISI(q)為第q個子通道的一符元間干擾能量，P _ICI(q) 為第q個子通道的一子載波間干擾能量，q為子通道編號，第q個子通道的信號能量表示為，且第q個子通道的雜訊能量表示為。接著，藉由計算出之信號雜訊比SNR _q 與前述之查找表便能計算出第二位元負載。 In step 607, the energy of ICI and ISI when each subchannel uses the first guard interval is estimated, and in the case of ICI and ISI, the signal noise after adding the first guard interval is estimated. The calculation method is as follows: Wherein a symbol P _ISI (q) is the q th channel intersymbol interference power, P _ICI (q) is between a subcarrier q-th channel interference energy, q is the subchannel number, the signal q-th channel Energy is expressed as And the noise energy of the qth subchannel is expressed as . Then, the second bit load can be calculated by calculating the signal noise ratio SNR _q and the aforementioned lookup table.

於步驟607中，判斷使用第一保護區間(例如GI-567)並配合第二位元負載的符元之位元產量是否大於使用第二保護區間(GI-3534)並配合第一位元負載的符元之位元產量(步驟604)。當使用GI-567的符元之位元產量比使用GI-3534的符元之位元產量大，便決定用該第一保護區間以作為該符元之保護區間(步驟605)，否則使用GI-3534(步驟606)。 In step 607, it is determined whether the bit yield of the symbol using the first guard interval (for example, GI-567) and the second bit load is greater than the second guard interval (GI-3534) and the first bit load is used. The bit yield of the symbol is (step 604). When the bit yield of the symbol using GI-567 is larger than the bit yield using the symbol of GI-3534, it is decided to use the first guard interval as the guard interval of the symbol (step 605), otherwise the GI is used. -3534 (step 606).

在另一特定實施例中，可以在步驟607之前加入一判斷步驟：判斷通道的脈衝響應的長度是否大於一預設門檻值，其中此預設門檻值介於GI-567與GI-3534的長度之間。此預設門檻值的定義標準一般來說：只要通道脈衝響應的長度超過上述預設門檻值長度時，且防護區間是使用GI-567，就絕對不可能使符元之位元產量超過使用GI-3534的符元。因此，只要通道脈衝響應的長度大於此預設門檻值，便可以直接決定使用GI-3534作為符元的防護區間。一般來說，此預設門檻值的設計可以透過整個系統的預先模擬或是數學的分析而得。另外，上述實施例的步驟雖然以上述的順序排列，但本領域具有通常知識者應當可以作些許的修正，例如將步驟602與步驟603互相交換，同樣 可以達到本發明之功效，故本發明不限於上述特定實施例。 In another specific embodiment, a determining step may be added before step 607: determining whether the length of the impulse response of the channel is greater than a predetermined threshold, wherein the preset threshold is between GI-567 and GI-3534 between. The definition of the preset threshold is generally: as long as the length of the channel impulse response exceeds the preset threshold value, and the guard interval is GI-567, it is absolutely impossible to make the bit yield of the symbol exceed the use of GI. -3534 symbol. Therefore, as long as the length of the channel impulse response is greater than the preset threshold, the guard interval using the GI-3534 as a symbol can be directly determined. In general, the design of this preset threshold can be obtained through pre-simulation or mathematical analysis of the entire system. In addition, although the steps of the above embodiments are arranged in the above-described order, those skilled in the art should be able to make some modifications, for example, the steps 602 and 603 are exchanged, and the same. The effects of the present invention can be attained, and the present invention is not limited to the specific embodiments described above.

為了證明上述的方法與裝置之運作可以有效的達到產量增加，並且證明上述干擾最主要是ISI以及ICI，其餘的交互影響的能量可以忽略不計，底下列舉了幾個模擬結果。圖7A繪示一有效通道長度為1410的通道脈衝響應圖。圖7B繪示為當通道脈衝響應為圖7A的情況時，使用GI-3534的符元之信號雜訊比701以及使用GI-417的符元之信號雜訊比702。圖7C繪示為量測出使用GI-417與GI-3534的符元之信號雜訊比的差值703以及利用本發明實施例所估測出的使用GI-417與GI-3534的符元之信號雜訊比的差值704。 In order to prove that the above methods and devices can effectively achieve the increase in output, and prove that the above interference is mainly ISI and ICI, the energy of the remaining interactions can be neglected, and several simulation results are listed below. FIG. 7A is a diagram showing a channel impulse response of an effective channel length of 1410. FIG. 7B illustrates the signal-to-noise ratio 701 of the symbol using GI-3534 and the signal-to-noise ratio 702 of the symbol using GI-417 when the channel impulse response is the case of FIG. 7A. FIG. 7C is a diagram showing the difference 703 between the signal and noise ratios of the symbols using GI-417 and GI-3534, and the symbols using GI-417 and GI-3534 estimated by the embodiment of the present invention. The difference 704 of the signal to noise ratio.

請參照圖7B中的701以及702，由圖7B的701與702可以看出701是沒有ISI與ICI干擾的信號雜訊比，702則是有ISI與ICI干擾以及上述交互影響的能量的信號雜訊比。接下來，請參照圖7C中的703與704，由圖7C中可以看到，703是701與702兩者實際的差值，704則是利用本發明實施例的方法所估測出701與702之差值。值得注意的是，量測出的信號雜訊比差值703與估測出信號雜訊比差值704兩者非常接近。因此，本發明的實施例可以準確的估測出ISI與ICI能量，另外，也證實了上述交互影響的能量是可以忽略的。 Referring to 701 and 702 in FIG. 7B, it can be seen from 701 and 702 of FIG. 7B that 701 is a signal to noise ratio without ISI and ICI interference, and 702 is a signal miscellaneous with ISI and ICI interference and energy of the above interaction. News ratio. Next, please refer to 703 and 704 in FIG. 7C. As can be seen from FIG. 7C, 703 is the actual difference between 701 and 702, and 704 is estimated by 701 and 702 by using the method of the embodiment of the present invention. The difference. It is worth noting that the measured signal noise ratio difference 703 is very close to the estimated signal noise ratio difference 704. Therefore, the embodiment of the present invention can accurately estimate the ISI and ICI energy, and also confirms that the energy of the above interaction is negligible.

圖8A繪示為通道長度為1890的通道脈衝響應圖。圖8B繪示為當通道脈衝響應為圖8A的情況時，使用GI-3534 的符元之信號雜訊比801以及使用GI-417的符元之信號雜訊比802。圖8C繪示為量測出使用GI-417與GI-3534的符元之信號雜訊比的差值803以及利用本發明實施例所估測出的使用GI-417與GI-3534的符元之信號雜訊比的差值804。同樣的，在這三張圖也可以得到上述的結論。另外，由上述圖7C與8C可以看出，低頻子載波部分信號雜訊比落差較大，即ISI與ICI的影響大部分只在較低頻的子載波，較高頻的子載波所受到的ISI與ICI之影響較小。因此在實施本發明時，亦可以不用估測所有的子頻帶之信號雜訊比，只需估測較低頻帶的信號雜訊比即可達到本發明的目的。 FIG. 8A is a diagram showing a channel impulse response of a channel length of 1890. Figure 8B illustrates the use of GI-3534 when the channel impulse response is the case of Figure 8A. The signal noise of the symbol is higher than that of the 801 and the signal noise of the symbol using GI-417 is 802. FIG. 8C is a diagram showing the difference 803 between the signal and noise ratios of the symbols using GI-417 and GI-3534, and the symbols using GI-417 and GI-3534 estimated by the embodiment of the present invention. The difference 804 of the signal noise ratio. Similarly, the above conclusions can be obtained in these three figures. In addition, as can be seen from the above FIGS. 7C and 8C, the low-frequency subcarrier portion signal noise ratio is large, that is, the influence of ISI and ICI is mostly only on the lower frequency subcarrier, and the higher frequency subcarrier is received. The impact of ISI and ICI is small. Therefore, in implementing the present invention, it is also possible to estimate the signal-to-noise ratio of all sub-bands without estimating the signal-to-noise ratio of the lower frequency band.

圖9A為通道長度為1100的通道脈衝響應圖。圖9B繪示使用GI-3534的符元之信號雜訊比901以及使用GI-567的符元之信號雜訊比902，配合使用北美的tone mask的結果，每個使用GI-567的符元可以乘載6043個bits，使用GI-3534時，每個符元可以乘載6228個bits。顯然選擇GI-567搭配較低位元負載的位元產量，會高於GI-3534搭配6228 bits位元負載的位元產量，因此證明透過本發明的方法可以達到更佳的傳輸效率。 Figure 9A is a channel impulse response diagram with a channel length of 1100. FIG. 9B illustrates the signal noise ratio 901 of the symbol using GI-3534 and the signal noise ratio 902 of the symbol using GI-567, and the result of using the tone mask of North America, each symbol using GI-567. You can ride 6043 bits. When using GI-3534, each symbol can carry 6228 bits. Obviously, selecting the bit yield of GI-567 with lower bit load will be higher than the bit yield of GI-3534 with 6228 bits bit load, thus demonstrating better transmission efficiency through the method of the present invention.

綜上所述，本發明在通道脈衝響應的長度介於一第一保護區間與第二保護區間的長度之間的情況下，採用判斷使用第一保護區間的符元之位元產量與使用第二保護區間的符元之位元產量，以決定使用第一保護區間或第二保護 區間作為符元之保護區間，以及第一位元負載或第二位元負載作為符元之位元負載；另外，本發明實施例透過數學與模擬分析出符元間干擾以及載波間干擾，藉以作出適應性決定保護區間長度與位元負載的裝置以及干擾估測裝置，因此本發明除了可以達到「根據通道長度以及干擾之大小，以決定循環前置長度以及各個子載波所乘載之資料位元量」以及「估測符元間干擾以及載波間干擾」之外，在本發明的特定實施例中，還可以達到增加位元產量與節省成本的目的。 In summary, in the case where the length of the channel impulse response is between the lengths of the first guard interval and the second guard interval, the bit yield and use of the symbol using the first guard interval are determined. The yield of the symbol of the second guard interval to determine the use of the first guard interval or the second protection The interval is used as the guard interval of the symbol, and the first bit payload or the second bit payload is used as the bit load of the symbol. In addition, the embodiment of the present invention analyzes the inter-symbol interference and the inter-carrier interference through mathematical and analog. The apparatus for adaptively determining the interval length and the bit load and the interference estimating device are made. Therefore, the present invention can achieve "determining the cyclic pre-length and the data bits occupied by each sub-carrier according to the channel length and the size of the interference. In addition to the "quantity" and "estimation of inter-symbol interference and inter-carrier interference", in a particular embodiment of the invention, the goal of increasing bit yield and cost savings can also be achieved.

雖然本發明已經以實施例揭露如上，然上述實施例並非用以限定本發明之精神，任何所屬技術領域具有通常知識者，在不脫離本發明之精神和範圍內，當可作些許之更動與潤飾，因此本發明之保護範圍當視後附之申請專利範圍所界定者為準。 Although the present invention has been disclosed in the above embodiments, the above embodiments are not intended to limit the spirit of the present invention, and those skilled in the art can make a few changes without departing from the spirit and scope of the invention. The scope of protection of the present invention is therefore defined by the scope of the appended claims.

AVFC‧‧‧框控制區塊 AVFC‧‧‧ box control block

PB520‧‧‧物理層區塊 PB520‧‧‧ physical layer block

H520‧‧‧物理層區塊標頭檔 H520‧‧‧Physical layer block header file

B520‧‧‧資料區塊 B520‧‧‧Information block

CS520‧‧‧物理層區塊檢查碼 CS520‧‧‧Physical layer block check code

201‧‧‧防護區間 201‧‧‧Protection zone

202、h(0)~h(L-1)‧‧‧通道脈衝響應 202, h(0) ~ h(L-1) ‧‧‧ channel impulse response

K‧‧‧防護區間201的長度 K ‧‧‧ Length of guard zone 201

L‧‧‧通道脈衝響應202的長度 L ‧‧‧ channel impulse response 202 length

x(0)~x(-K-1)‧‧‧傳送出的時域信號 x(0) ~ x(-K-1) ‧‧‧ transmitted time domain signals

301‧‧‧通道估測裝置 301‧‧‧channel estimation device

302‧‧‧干擾估測裝置 302‧‧‧Interference estimation device

303‧‧‧位元決定裝置 303‧‧‧ bit decision device

304‧‧‧判斷裝置 304‧‧‧Judgement device

401‧‧‧傅立葉轉換器 401‧‧‧Fourier converter

402‧‧‧頻帶選擇器 402‧‧‧ Band Selector

403‧‧‧干擾能量估測器 403‧‧‧Interference energy estimator

501‧‧‧減法器 501‧‧‧Subtractor

502‧‧‧第一乘法器 502‧‧‧First multiplier

503‧‧‧延遲電路 503‧‧‧Delay circuit

504‧‧‧選擇電路 504‧‧‧Selection circuit

505‧‧‧絕對值平方運算器 505‧‧‧Absolute Square Operator

506‧‧‧累加器 506‧‧‧ accumulator

507‧‧‧第二乘法器 507‧‧‧Second multiplier

601~610‧‧‧步驟流程 601~610‧‧‧Step process

604.1、604.2‧‧‧子步驟流程 604.1, 604.2‧‧‧ substep process

701、801‧‧‧使用GI-3534的符元之信號雜訊比 701, 801‧‧‧Signal noise ratio using the symbol of GI-3534

702、802‧‧‧使用GI-417的符元之信號雜訊比 702, 802‧‧‧ Signal to noise ratio using GI-417

703、803‧‧‧使用GI-417與GI-3534的符元之信號雜訊比的差值 703, 803‧‧‧ using the difference between the signal-to-noise ratio of the GI-417 and GI-3534

704、804‧‧‧利用本發明實施例所估測出的使用GI-417與GI-3534的符元之信號雜訊比的差值 704, 804 ‧ ‧ the difference between the signal and noise ratios of the symbols using GI-417 and GI-3534 estimated by the embodiment of the present invention

901‧‧‧使用GI-3534的符元之信號雜訊比並配合使用北美的tone mask 901‧‧‧Use the signal noise ratio of the GI-3534 symbol and use the tone mask of North America

902‧‧‧使用GI-567的符元之信號雜訊比並配合使用北美的tone mask 902‧‧‧Use the GI-567 symbolic noise ratio and use the North American tone mask

圖1繪示為在Home-Plug AV系統中所規定之長的媒體存取協定資料單元的示意圖。 1 is a schematic diagram of a long media access protocol data unit defined in a Home-Plug AV system.

圖2A以及圖2B繪示為符元的防護區間201之長度小於通道脈衝響應202的長度時所發生的ISI示意圖。 2A and 2B are schematic diagrams showing the ISI that occurs when the length of the guard interval 201 of the symbol is less than the length of the channel impulse response 202.

圖3繪示為適應性決定保護區間與位元負載的裝置。 FIG. 3 illustrates an apparatus for adaptively determining a guard interval and a bit load.

圖4繪示為本發明實施例圖3的干擾估測裝置302之電路方塊圖。 4 is a circuit block diagram of the interference estimation apparatus 302 of FIG. 3 according to an embodiment of the present invention.

圖5繪示為本發明實施例圖3的干擾能量估測器403之電路方塊圖。 FIG. 5 is a circuit block diagram of the interference energy estimator 403 of FIG. 3 according to an embodiment of the present invention.

圖6繪示為本發明實施例之適應性決定保護區間長度與位元負載的方法的流程圖。 FIG. 6 is a flow chart showing a method for determining the length of a guard interval and a bit load according to an embodiment of the present invention.

圖7A繪示為通道長度為1410的通道脈衝響應圖。 FIG. 7A is a diagram showing a channel impulse response of a channel length of 1410.

圖7B繪示為當通道脈衝響應為圖7A的情況時，使用GI-3534的符元之信號雜訊比701以及使用GI-417的符元之信號雜訊比702。 FIG. 7B illustrates the signal-to-noise ratio 701 of the symbol using GI-3534 and the signal-to-noise ratio 702 of the symbol using GI-417 when the channel impulse response is the case of FIG. 7A.

圖7C繪示為量測出使用GI-417與GI-3534的符元之信號雜訊比的差值703以及利用本發明實施例所估測出的使用GI-417與GI-3534的符元之信號雜訊比的差值704。 FIG. 7C is a diagram showing the difference 703 between the signal and noise ratios of the symbols using GI-417 and GI-3534, and the symbols using GI-417 and GI-3534 estimated by the embodiment of the present invention. The difference 704 of the signal to noise ratio.

圖8A繪示為通道長度為1890的通道脈衝響應圖。 FIG. 8A is a diagram showing a channel impulse response of a channel length of 1890.

圖8B繪示為當通道脈衝響應為圖7A的情況時，使用GI-3534的符元之信號雜訊比801以及使用GI-417的符元之信號雜訊比802。 FIG. 8B illustrates the signal-to-noise ratio 801 of the symbol using GI-3534 and the signal-to-noise ratio 802 of the symbol using GI-417 when the channel impulse response is the case of FIG. 7A.

圖8C繪示為量測出使用GI-417與GI-3534的符元之信號雜訊比的差值803以及利用本發明實施例所估測出的使用GI-417與GI-3534的符元之信號雜訊比的差值804。 FIG. 8C is a diagram showing the difference 803 between the signal and noise ratios of the symbols using GI-417 and GI-3534, and the symbols using GI-417 and GI-3534 estimated by the embodiment of the present invention. The difference 804 of the signal noise ratio.

圖9A為通道長度為1100的通道脈衝響應圖。 Figure 9A is a channel impulse response diagram with a channel length of 1100.

圖9B繪示使用GI-3534的符元之信號雜訊比901以及使用GI-567的符元之信號雜訊比902，配合使用北美的tone mask的結果。 FIG. 9B shows the result of using the signal noise ratio 901 of the symbol of GI-3534 and the signal noise ratio 902 of the symbol using GI-567, in conjunction with the tone mask of North America.

601~610‧‧‧步驟流程 601~610‧‧‧Step process

604.1、604.2‧‧‧子步驟流程 604.1, 604.2‧‧‧ substep process

Claims (13)

一種適應性決定保護區間長度與位元負載的方法，包括：提供一多載波調變系統，該多載波調變系統的規格規定至少一第一保護區間以及一第二保護區間，其中該第一保護區間的長度小於該第二保護區間的長度；在一通道閒置期間，估測該通道之每一子通道的雜訊能量；估測該通道之一通道脈衝響應；計算該多載波調變系統中之每一子載波所能乘載的一第一位元負載；當該通道的長度小於該第一保護區間時，則決定使用該第一位元負載與該第一保護區間；以及當該通道長度大於或等於該第一保護區間的長度時：估測一符元加上該第一保護區間時的一子載波間干擾能量與一符元間干擾能量，並計算出此時該符元加上該第一保護區間的信號雜訊比，以得到每一子載波所能乘載的一第二位元負載；以及當使用該第一保護區間並配合該第二位元負載的該符元之位元產量大於使用該第二保護區間並配合該第一位元負載的該符元之位元產量時，決定用該第一保護區間與該第二位元負載，否則，則使用該第二保護區間與該第一位元負載。 A method for adaptively determining a guard interval length and a bit load, comprising: providing a multi-carrier modulation system, wherein the specification of the multi-carrier modulation system defines at least a first guard interval and a second guard interval, wherein the first The length of the guard interval is less than the length of the second guard interval; during a period of idle period, the noise energy of each subchannel of the channel is estimated; and the channel impulse response of one channel of the channel is estimated; and the multi-carrier modulation system is calculated a first bit load that can be carried by each of the subcarriers; when the length of the channel is less than the first guard interval, then determining to use the first bit load and the first guard interval; and when the When the length of the channel is greater than or equal to the length of the first guard interval: estimating a inter-carrier interference energy and a symbol interference energy when the first guard interval is added to the first guard interval, and calculating the symbol at this time And adding a signal to noise ratio of the first guard interval to obtain a second bit load that can be carried by each subcarrier; and when the first guard interval is used and matched with the second bit load When the bit yield is greater than the bit yield of the symbol using the second guard interval and matching the first bit load, determining to use the first guard interval and the second bit load, otherwise, using the first The second guard interval is loaded with the first bit. 如申請專利範圍第1項所述之適應性決定保護區間 長度與位元負載的方法，其中「計算該多載波調變系統中之每一子載波所能乘載的一第一位元負載」，此步驟包括：估測每一子通道的信號能量；以及計算每一子載波的信號雜訊比，並藉此決定每一子載波所能乘載的該第一位元負載。 The adaptive decision protection interval as described in item 1 of the patent application scope A method of length and bit load, wherein "calculating a first bit load that can be carried by each subcarrier in the multicarrier modulation system" includes: estimating a signal energy of each subchannel; And calculating a signal to noise ratio of each subcarrier, and thereby determining the first bit load that each subcarrier can ride. 如申請專利範圍第2項所述之適應性決定保護區間長度與位元負載的方法，其中「計算每一子載波的信號雜訊比」是指在沒有符元間干擾與子載波間干擾時的信號雜訊比。 The method for determining the length of the guard interval and the bit load as described in the second paragraph of the patent application scope, wherein "calculating the signal-to-noise ratio of each sub-carrier" means that there is no inter-symbol interference and inter-subcarrier interference. Signal noise ratio. 如申請專利範圍第1項所述之適應性決定保護區間長度與位元負載的方法，其中一第q個子通道的信號能量表示為，該第q個子通道的雜訊能量表示為，則「該符元加上該第一保護區間之後，該第q個子通道的信號雜訊比SNR _q 」表示如下： ，其中P _ISI(q)為第q個子通道的一符元間干擾能量，P _ICI(q)為第q個子通道的一子載波間干擾能量，q為子通道編號。 The method of determining the length of the protection interval and the bit load according to the adaptation described in the first claim of the patent scope, wherein the signal energy of a qth subchannel is expressed as The noise energy of the qth subchannel is expressed as , "The signal-to-noise ratio SNR _{q of} the q- th sub-channel after the symbol is added to the first guard interval" is expressed as follows: Wherein a symbol P _ISI (q) is the q th energy channel intersymbol interference, P _ICI (q) is the q subcarriers between a subchannel of the interference energy, q is the sub-channel number. 如申請專利範圍第4項所述之適應性決定保護區間長度與位元負載的方法，其中P _ISI(q)=P _ICI(q)。 The method of determining the length of the guard interval and the bit load as described in the fourth paragraph of the patent application, wherein P _ISI ( q ) = P _ICI ( q ). 一種適應性決定保護區間長度與位元負載的裝置，用於一多載波調變系統，該多載波調變系統的規格至少規定一第一保護區間以及一第二保護區間，其中該第一保護區間的長度小於該第二保護區間的長度，此裝置包括： 一通道估測裝置，用以估測一通道脈衝響應；一干擾估測裝置，用以估測一第q個子通道的干擾能量P(q)；一位元負載決定裝置，接收該第q個子通道的干擾能量P(q)、該第q個子通道的信號能量以及該第q個子通道的雜訊能量，根據該第q個子通道的信號雜訊比，以決定該第q個子通道所乘載的一位元負載；以及一判斷裝置，根據以及一第二保護區間以判斷一第一位元產量，並根據 P(q)]以及一第二保護區間以判斷一第二位元產量，比較該第一位元產量與該第二位元產量以決定使用該第一保護區間或該第二保護區間，其中q為子通道編號。 A device for determining a guard interval length and a bit load for use in a multi-carrier modulation system, wherein the specification of the multi-carrier modulation system defines at least a first protection interval and a second protection interval, wherein the first protection The length of the interval is less than the length of the second guard interval, and the device comprises: a channel estimating device for estimating a channel impulse response; and an interference estimating device for estimating the interference energy P of the qth subchannel (q); one yuan load determining means receives the second interference energy subchannels q P (q), the signal energy of the q-th channel And the noise energy of the qth subchannel According to the signal noise ratio of the qth subchannel To determine a one-bit load carried by the qth sub-channel; and a judging device, according to And a second protection interval to determine a first bit yield, and according to P(q) ] and a second guard interval to determine a second bit yield, comparing the first bit yield with the second bit yield to determine to use the first guard interval or the second guard interval, wherein q is the subchannel number. 如申請專利範圍第6項所述之適應性決定保護區間長度與位元負載的裝置，其中該干擾估測裝置包括：一傅立葉轉換器，用以將該通道脈衝響應作傅立葉轉換，以輸出多數個子頻帶之頻率響應的一指定部份；一頻帶選擇器，耦接該傅立葉轉換器，接收該些子頻帶之頻率響應的該指定部份，選擇該第q個子頻帶之頻率響應的該指定部份輸出；以及一干擾能量估測器，耦接該頻帶選擇器以及該通道估測裝置，接收該第q個子頻帶之頻率響應的該指定部分，以估測出該第q個子頻帶的干擾能量P(q)。 The apparatus for determining the length of the protection interval and the bit load according to the adaptation described in claim 6 wherein the interference estimation apparatus comprises: a Fourier converter for performing Fourier transform on the impulse response of the channel to output a majority a specified portion of the frequency response of the sub-bands; a band selector coupled to the Fourier converter, receiving the designated portion of the frequency response of the sub-bands, and selecting the designated portion of the frequency response of the q- th sub-band parts output; and a designated portion of the interference energy estimator, coupled to the band selector and channel estimation means for receiving the q-th frequency band of the response, in order to estimate the energy of the interference of the q subbands P(q) . 如申請專利範圍第7項所述之適應性決定保護區間 長度與位元負載的裝置，其中該干擾能量估測器包括：一選擇電路，包括一第一接收部分、一第二接收部分以及一輸出部分，該第一接收部分耦接該頻帶選擇器，接收該第q個子頻帶之頻率響應的該指定部份，用以選擇該第一、該第二接收部分其中之一輸出；一減法器，耦接該輸出部分以及該通道估測裝置，用以將該輸出部分的所輸出之一特定值與該通道估測裝置所輸出的一估測值相減後輸出一減法值，其中該特定值與該估測值互相對應；一第一乘法器，用以將該減法值乘以一第q個子載波，以得到一第一乘法值；一延遲電路，耦接該第一乘法器與該第二接收部分，接收該第一乘法值，將該第一乘法值延遲一預設時間後輸出；一絕對值平方運算器，耦接該輸出部分，將該輸出部分所輸出的該特定值取絕對值並且平方後輸出一頻率響應能量值；一累加器，耦接該絕對值平方運算器，將所接收到的該頻率響應能量值作累加運算，輸出一累加頻率響應能量值；以及一第二乘法器，將該累加頻率響應能量值乘上一子載波平均能量，輸出該第q個子頻帶的干擾能量P(q)。 The apparatus for determining the length of the protection interval and the bit load, as described in claim 7, wherein the interference energy estimator comprises: a selection circuit comprising a first receiving portion, a second receiving portion, and a An output portion, the first receiving portion is coupled to the band selector, and receives the designated portion of the frequency response of the qth sub-band for selecting one of the first and second receiving portions to output; a subtractor And the channel estimating device is configured to subtract a value of the output of the output portion from an estimated value output by the channel estimating device, and output a subtraction value, wherein the specific And a value corresponding to the estimated value; a first multiplier for multiplying the subtraction value by a qth subcarrier to obtain a first multiplication value; a delay circuit coupled to the first multiplier and the The second receiving portion receives the first multiplication value, delays the first multiplication value by a predetermined time, and outputs the first multiplication value; an absolute value square operator is coupled to the output portion, and the specific value output by the output portion is taken Absolutely And summing a value of the frequency response energy value; an accumulator coupled to the absolute value square operator, accumulating the received frequency response energy value, outputting an accumulated frequency response energy value; and a second The multiplier multiplies the accumulated frequency response energy value by a subcarrier average energy, and outputs the interference energy P(q) of the qth subband. 如申請專利範圍第8項所述之適應性決定保護區間長度與位元負載的裝置，其中該第一通道長度為K，該通 道脈衝響應的長度為L，通道脈衝響應為h(0)~h(L-1)，L>K，該第q個子頻帶之頻率響應的該指定部份為H _K+1 (q)，其中H _K+1 (q)為h(K+1)~h(L-1)作傅立葉轉換後的該第q個子頻帶的頻率響應。 The apparatus for determining the length of the protection interval and the bit load as described in claim 8 of the patent scope, wherein the length of the first channel is K , the length of the impulse response of the channel is L , and the channel impulse response is h(0) ~ h(L-1) , L > K , the specified portion of the frequency response of the qth sub-band is H _{K + 1} (q) , where H _{K + 1} (q) is h(K + 1) ~ h (L-1) The frequency response of the qth sub-band after Fourier transform. 如申請專利範圍第6項所述之適應性決定保護區間長度與位元負載的裝置，其中P(q)為一符元間干擾能量P _ISI (q)，且該符元間干擾能量等於一載波間干擾能量P _ICI (q)。 The apparatus for determining the length of the guard interval and the bit load as described in claim 6 of the patent scope, wherein P(q) is an inter-symbol interference energy P _ISI (q) , and the inter-symbol interference energy is equal to one Inter-carrier interference energy P _ICI (q) . 一種干擾估測裝置，用於一多載波調變系統，該多載波調變系統的規格至少規定一第一保護區間，其中該第一保護區間的長度為K，此裝置包括：一通道估測單元，用以估測通道脈衝響應h(0)~h(L-1)，其中L為該通道脈衝響應的長度，且L>K；一傅立葉轉換器，用以將通道脈衝響應的h(K+1)~h(L-1)作傅立葉轉換，以輸出多數個子頻帶之頻率響應H _K+1 (0)~H _K+1 (N)；一頻帶選擇器，耦接該傅立葉轉換器，用以選擇一第q個子頻帶之頻率響應H _K+1 (q)；以及一干擾能量估測器，耦接該頻帶選擇器以及該通道估測裝置，接收該第q個子頻帶之頻率響應H _K+1 (q)以及通道脈衝響應h(K+1)，據以計算出H _K+2 (q)~H _L-1 (q)，以估測出該第q個子頻帶的干擾能量P(q)。 An interference estimation apparatus is used for a multi-carrier modulation system. The specification of the multi-carrier modulation system defines at least a first protection interval, wherein the length of the first protection interval is K , and the device includes: a channel estimation a unit for estimating a channel impulse response h(0) ~ h(L-1) , where L is the length of the impulse response of the channel, and L>K; a Fourier transformer, the h of the channel impulse response ( K + 1) ~ h(L-1) for Fourier transform to output the frequency response H _{K + 1} ( 0 )~ H _{K + 1} ( N ) of a plurality of sub-bands; a band selector coupled to the Fourier converter a frequency response H _{K + 1} ( q ) for selecting a qth sub-band; and an interference energy estimator coupled to the band selector and the channel estimation device to receive the frequency response of the q- th sub-band H _{K + 1} ( q ) and the channel impulse response h(K + 1) , from which H _{K + 2} ( q )~ H _{L - 1} ( q ) is calculated to estimate the interference energy of the qth sub-band P(q) . 如申請專利範圍第11項所述之干擾估測裝置，其中該干擾能量估測器包括： 一選擇電路，包括一第一接收部分、一第二接收部分以及一輸出部分，該第一接收部分耦接該頻帶選擇器，接收H _K+1 (q)，用以選擇該第一、該第二接收部分其中之一輸出；一減法器，耦接該輸出部分以及該通道估測裝置，用以將該輸出部分的所輸出之一特定值與該通道估測裝置所輸出的一估測值相減後輸出一減法值，其中該特定值與該估測值互相對應；一第一乘法器，用以將該減法值乘以第q個子載波，以得到一第一乘法值；一延遲電路，耦接該第一乘法器與該第二接收部分，接收該第一乘法值，將該第一乘法值延遲一預設時間後輸出；一絕對值平方運算器，耦接該輸出部分，將該輸出部分所輸出的該特定值取絕對值並且平方後輸出一頻率響應能量值；一累加器，耦接該絕對值平方運算器，將所接收到的該頻率響應能量值作累加運算，輸出一累加頻率響應能量值；以及一第二乘法器，將該累加頻率響應能量值乘上一子載波平均能量，輸出該第q個子頻帶的干擾能量P(q)。 The interference estimation device of claim 11, wherein the interference energy estimator comprises: a selection circuit comprising a first receiving portion, a second receiving portion, and an output portion, the first receiving portion Coupling the band selector, receiving H _{K + 1} ( q ) for selecting one of the first and second receiving portions; a subtractor coupled to the output portion and the channel estimating device Subtracting a specific value of the output of the output portion and an estimated value output by the channel estimating device, and outputting a subtraction value, wherein the specific value and the estimated value correspond to each other; a first multiplier And multiplying the subtraction value by the qth subcarrier to obtain a first multiplication value; a delay circuit coupled to the first multiplier and the second receiving portion, receiving the first multiplication value, the first a multiplication value is delayed after a predetermined time output; an absolute value square operator is coupled to the output portion, and the specific value outputted by the output portion is taken as an absolute value and squared to output a frequency response energy value; an accumulator , coupled to the a value square operator, which accumulates the received frequency response energy value to output an accumulated frequency response energy value; and a second multiplier that multiplies the accumulated frequency response energy value by a subcarrier average energy, The interference energy P(q) of the qth sub-band is output. 如申請專利範圍第11項所述之干擾估測裝置，其中P(q)為一符元間干擾能量P _ISI (q)，且該符元間干擾能量等於一載波間干擾能量P _ICI (q)。 The interference estimation apparatus according to claim 11, wherein P(q) is an inter-symbol interference energy P _ISI (q) , and the inter-symbol interference energy is equal to an inter-carrier interference energy P _ICI (q) ) .