201216629 六、發明說明： 【發明所屬之技術領域】 β本揭示内容是有關於一種通訊裝置及其方法，且特別 疋有關於-種雙模射頻收發H及其㈣訊號接收方法。 【先前技術】 一零中頻架構為現今經常被使絲接收射頻訊辣的電路 =计。不需要鏡像抑制亦不需要中輯波器的優點盆 t於單晶：整合。然而，在雙模式的射頻收發器上，常； ，入正交信號不匹配（J/Q mismatch)的問題。因為輸入 正^父信號不匹配會引起直流偏置現象，不論是由於與射頻 讯號進行混頻的本地振盪訊號之頻率與所接收的射頻訊號 頻率^目近造成的拉扯（puUing)效應，或是在本地振盪訊 號為高頻時的設計所造成功率消耗之劇增，都將使零中頻 架構的表現變差。 〇 因此，如何設計一個新的雙模射頻收發器及其射頻訊 號接收方法，以改善上述缺失，乃為此一業界亟待解決 問題。 【發明内容】 因此’本揭示内容之一態樣是在提供一種雙模射頻收 發器，包含：天線、差動低雜訊放大模組、本地振盪器以 及雙模差動混頻模組。一差動低雜訊放大模組，包含：差 動放大電路以及差動電感電路。差動放大電路包含輸入 201216629 端，用以根據天線接收於第一頻段接收模式時位於第一頻 段且於第二頻段接收模式時位於第二頻段之射頻訊號，其 中第二頻段約為第一頻段之兩倍。差動電感電路輕合於差 動放大電路之差動輸出端，並於第一頻段接收模式具有第 一電感值，於第二頻段接收模式具有第二電感值，俾根據 射頻訊號產生差動放大射頻訊號。本地振盪器產生多相位 • 本地振盪訊號’頻率約位於第一頻段。雙模差動混頻模組 包含：切換模組、複數差動本頻混頻電路以及複數差動次 Φ 諧波混頻電路。差動本頻混頻電路於第一頻段接收模式時 為切換模組致能’以接收多相位本地振盡訊號中為反相之 二相位’俾根據位於第一頻段之差動放大射頻訊號混頻產 生第一差動基頻訊號。差動次諧波混頻電路於第二頻段接 枚模式時為切換模組致能，以接收多相位本地振盪訊號中 為正交之四相位’俾疊加以根據位於第二頻段之差動放大 射頻訊號混頻產生第二差動基頻訊號。 依據本揭示内容一實施例，差動輸出端包含第一輸出 Φ 端以及第二輸出端。差動放大電路包含：第一放大支路以 及第二放大支路。第一放大支路包含輸入端以及第一輸出 端。第二放大支路包含耦合至耦合電位之耦合電位輸入端 以及第二輸出端。其中差動電感電路為中間抽頭對稱電感 電路。 依據本揭示内容另一實施例，第—及第二放大支路更 分別包含：輸入金氧半電晶體、輸出金氧半電晶體以及輸 入匹配模組。輸入金氧半電晶體包含用以接收射頻訊號或 耦合電位之閘極。輸出金氧半電晶體包含源極以及汲極， 201216629 源極搞接於輸入金氧半電晶體之汲極，沒極用以產生差動 放大射頻訊號。輸入匹配模組連接於輪入金氧半電晶體之 間極及没極，俾提供輸入匹配。其中輪入金氧半電晶體更 包3源極，藉由電感連接至耦合電位，電感之電感值大於 特定準位。耦合電位為接地電位。 • 依據本揭示内容又一實施例，差動本頻混頻 電路及次 讀波混頻電路分別包含吉伯特混頻器。差動本頻混頻電路 之吉伯特混頻器包含複數混頻單元，各為金氧半電晶體。 • 次諧波混頻電路之吉伯特混頻器包含複數混頻單元，各為 一並聯之金氧半電晶體。差動本頻混頻電路更包含冗餘混 頻器，俾使多相位本地振盪訊號中未為吉伯特混頻器使用 以產生第一及第二差動基頻訊號之部份，藉由冗餘混頻器 抵消。 依據本揭示内容再一實施例，其中第一差動基頻訊號 及第二差動基頻訊號為同相差動基頻訊號或為正交相差動 基頻訊號。 鲁 依據本揭示内容更具有之一實施例，其中多相位本地 振盪訊號為八相位本地振後訊號，包含〇度、45度、90度、 135度、180度、225度、270度以及315度之相位。其中 於第一頻段接收模式時，該等差動本頻混頻電路其中之一 者接收多相位本地振盪訊號中為〇度及180度之相位，差 動本頻混頻電路其中另一者接收多相位本地振盪訊號中為 90度及270度之相位，以分別根據差動放大射頻訊號混頻 產生第一差動基頻訊號。於第二頻段接收模式時，差動次 諧波混頻電路其中之一者接收多相位本地振盪訊號中為〇 201216629 度、90度、180度及270度之相位，差動次諧波混頻電路 其中另一者接收多相位本地振盪訊號中為45度、135度、 225度及315度之相位’以分別根據差動放大射頻訊號混 頻產生第二差動基頻訊號。 依據本揭示内容再具有之一實施例，其中多相位本地 振盪嬈由振盪訊號源除頻產生，振盪訊號源之頻率約為 • 多相位本地振盪訊號之四倍。 本揭示内容之另一態樣是在提供一種射頻訊號接收方 • 法，應用於雙模射頻收發器中，射頻訊號接收方法包含下 列步驟：根據天線接收於第一頻段接收模式時位於第一頻 段且於第二頻段接收模式時位於第二頻段之射頻訊號，其 中第二頻段約為第一頻段之兩倍。根據射頻訊號’於第一 頻段接收模式時使差動電感電路具有第一電感值，並於第 二頻段接收模式時使差動電感電路具有第二電感值，俾與 差動放大電路耦合產生差動放大射頻訊號。產生多相位本 地振盪訊號，其頻率約位於第一頻段。當位於第一頻段接 φ 收模式時，致能複數差動本頻混頻電路，以分別接收多相 位本地振盪訊號中為反相之二相位’俾根據差動放大射頻 訊號混頻產生第一差動基頻訊號。當位於第二頻段接收模 式時’致能複數差動次諧波混頻電路，以接收多相位本地 振盡訊號中為正交之四相位，俾疊加以根據差動放大射頻 訊號混頻產生第二差動基頻訊號。 -依據本揭示内容一實施例，其中當位於第一頻段接收 模式以致能差動本頻混頻電路，更包含使多相位本地振還 訊號中未使用以產生第一及第二差動基頻訊號之部份，藉 S, 7 201216629 由冗餘混頻器抵消。 依據本揭示内容另一實施例，其中第一差動基頻訊號 及第二差動基頻訊號為同相差動基頻訊號或為正交相差動 基頻訊號。 依據本揭示内容又一實施例，其中產生多相位本地振 盪訊號之步驟更包含由振盪訊號源除頻產生，振盪訊號源 之頻率約為多相位本地振盪訊號之四倍。 以、 應用本揭示内容之優點係在於藉由可切換的差動本頻 混頻電路以及差動次諧波混頻電路，可使用一個多相位本 地振盪訊號達到雙模式的射頻訊號接收，達到節省功率與 面積之功效，並可避免振盪訊號源為射頻訊號干擾，輕易、 地達到上述之目的。 【實施方式] 請參照第1圖。第1圖為本揭示内容之一實施例中， 一種雙模射頻收發器1之方塊圖。雙模射頻收發器1包含： 天線10、差動低雜訊放大模組12、本地振盪器14以及雙 模差動混頻模組16。 天線10可操作於第一頻段接收模式以及第二頻段接 收模式。舉例來說，於第一頻段接收模式，天線丨〇可接收 約為2GHz (F〇)的射頻訊號11，而於第二頻段接收模式， 天線10可接收約為5GHz (2Fo)的射頻訊號11。於本實 施例中，第二頻段約為第一頻段之兩倍。 請同時參照第2圖。第2圖為一實施例中，第1圖之 201216629 差動低雜訊放大模組12更詳細之電路圖。差動低雜訊放大 模組12包含：差動放大電路20以及差動電感電路22。 差動放大電路20包含第一放大支路以及第二放大支 路。其中第一放大支路包含輸入金氧半電晶體200a、輸出 金氧半電晶體202a以及輸入匹配模組204a，第二放大支路 包含輸入金氧半電晶體200b、輸出金氧半電晶體202b以 • 及輸入匹配模組204b。 於本實施例中，輸入金氧半電晶體200a、200b與輸出 • 金氧半電晶體202a、202b均為N型金氧半電晶體。輸入金 氧半電晶體200a之閘極為輸入端，用以自A點與天線10 連接’並接收來自天線10的射頻訊號11。輸出金氧半電 晶體202a之源極耦接於輸入金氧半電晶體2〇〇a之汲極。 輸入金氧半電晶體200b之閘極為耦合至耦合電位之耦合 電位輸入端，於本實施例中，此耦合電位為接地電位 gnd°輸入金氧半電晶體200b與耦合電位間具有負載，如 電阻或電容’於一實施例中’這些負載之阻抗係愈小愈好， •甚至趨近於零為佳。輸出金氧半電晶體202b之源極耦接於 輸入金氧半電晶體2〇〇b之汲極。輸出金氧半電晶體202a 與輸出金氧半電晶體202b之汲極連接至用以產生差動放 大射頻訊號RF-與RF+的第一輸出端及第二輸出端。 ^第一放大支路以及第二放大支路更分別包含輸入匹配 板組204a及204b。輪入匹配模組204a及204b於本實施例 中各包含電容與電阻，以提供一輸入匹配之作用。 - 輸入金氧半電晶體200a與200b之源極更藉由一電感 2〇6連接至前述為接地電位GND之耦合電位。電感206之 201216629 電感值大於一特定準位。於一實施例中，電感206之電感 值可大於500歐姆，甚至更高，以使單端轉雙端之機制更 為完善。 差動電感電路22實質上包含兩組電感，一組為電感 220a、222a’ 另一組為電感 220b、222b。電感 220a、222a、 220b、222b為中間抽頭對稱電感之形式，並藉由電感222a • 與222b耦合於差動放大電路20的第一及第二輸出端。其 中，差動電感電路22更包含控制電路224，用以接收一控 • 制訊號223。控制訊號223可設置為在第一頻段接收模式 時為〇,使控制電路224在第一頻段接收模式使電感220a、 222a以及電感220b、222b均導通，提供較大的第一電感 值，以針對較低頻率的射頻訊號n (Fq)進行處理。而在 第二頻段接收模式時控制訊號223則為丨，使控制電路224 在第二頻段接收模式時僅使電感222a以及電感222b導 通，提供相對較低的第二電感值，以針對較高頻率的射頻 訊號11 (2F。）進行處理。 • 因此，在經過差動放大電路20以及差動電感電路22 的處理後，射頻訊输】1收 ..._ ” 口丨以U U肘由早端轉雙端，而在包含第一及 差動輸出端產生差動放大射頻訊號RF_與 於第1圖中’差動放大射頻訊號繪示為201216629 VI. Description of the Invention: [Technical Field of the Invention] The present disclosure relates to a communication device and a method thereof, and particularly to a dual mode RF transceiver H and a method for receiving the same. [Prior Art] A zero-IF architecture is a circuit that is often used to receive RF signals. There is no need for image suppression or the advantages of the medium wave filter. However, on dual-mode RF transceivers, the problem of incoming quadrature signal mismatch (J/Q mismatch). Because the input positive parent signal mismatch will cause DC bias, whether due to the frequency of the local oscillation signal mixed with the RF signal and the puUing effect caused by the frequency of the received RF signal, or The dramatic increase in power consumption caused by the design of the local oscillator signal at high frequencies will degrade the performance of the zero-IF architecture. 〇 Therefore, how to design a new dual-mode RF transceiver and its RF signal receiving method to improve the above-mentioned defects is an urgent problem for the industry. SUMMARY OF THE INVENTION Accordingly, one aspect of the present disclosure is to provide a dual mode RF transceiver comprising: an antenna, a differential low noise amplification module, a local oscillator, and a dual mode differential mixing module. A differential low noise amplification module includes: a differential amplifier circuit and a differential inductor circuit. The differential amplifying circuit includes an input 201216629 terminal for receiving an RF signal in the second frequency band when the antenna is received in the first frequency band receiving mode and in the second frequency band receiving mode, wherein the second frequency band is about the first frequency band. Doubled. The differential inductor circuit is lightly coupled to the differential output end of the differential amplifying circuit, and has a first inductance value in the first frequency band receiving mode, a second inductance value in the second frequency band receiving mode, and a differential amplification according to the RF signal. RF signal. The local oscillator generates multiple phases. • The local oscillator signal 'frequency is approximately in the first band. The dual-mode differential mixing module includes: a switching module, a complex differential frequency mixing circuit, and a complex differential Φ harmonic mixing circuit. The differential local frequency mixing circuit is enabled for the switching module when receiving the mode in the first frequency band to receive the two phases of the multi-phase local vibration signal as the inversion phase 俾 according to the differential amplification RF signal in the first frequency band. The frequency generates a first differential fundamental frequency signal. The differential subharmonic mixing circuit is enabled for the switching module in the second frequency band mode to receive the quadrature four phase '俾 superposition in the multi-phase local oscillation signal to be differentially amplified according to the second frequency band. The RF signal mixing produces a second differential fundamental frequency signal. According to an embodiment of the present disclosure, the differential output includes a first output Φ terminal and a second output terminal. The differential amplifying circuit includes a first amplifying branch and a second amplifying branch. The first amplification branch includes an input and a first output. The second amplification branch includes a coupled potential input coupled to the coupled potential and a second output. The differential inductor circuit is a center-tap symmetrical inductor circuit. According to another embodiment of the present disclosure, the first and second amplification branches further comprise: an input MOS semi-transistor, an output MOS semi-transistor, and an input matching module. The input MOS transistor includes a gate for receiving an RF signal or a coupling potential. The output MOS transistor contains a source and a drain. The 201216629 source is connected to the drain of the input MOS transistor, and the immersion is used to generate a differential amplified RF signal. The input matching module is connected to the pole and the pole of the turn-in MOS transistor, and provides input matching. The turn-in MOS transistor also includes 3 sources, which are connected to the coupling potential by an inductor, and the inductance value of the inductor is greater than a specific level. The coupling potential is the ground potential. • According to still another embodiment of the present disclosure, the differential local frequency mixing circuit and the secondary read wave mixing circuit respectively comprise a Gilbert mixer. The differential local frequency mixing circuit of the Gilbert mixer comprises a plurality of mixing units, each of which is a gold-oxygen semi-transistor. • The Gilbert mixer of the subharmonic mixing circuit consists of a complex mixing unit, each of which is a parallel MOS transistor. The differential local frequency mixing circuit further includes a redundant mixer, so that the multi-phase local oscillation signal is not used by the Gilbert mixer to generate the first and second differential fundamental signals. The redundant mixer cancels. According to still another embodiment of the present disclosure, the first differential fundamental frequency signal and the second differential fundamental frequency signal are in-phase differential fundamental frequency signals or quadrature phase differential fundamental frequency signals. According to the disclosure, there is an embodiment in which the multi-phase local oscillation signal is an eight-phase local post-oscillation signal, including twist, 45 degrees, 90 degrees, 135 degrees, 180 degrees, 225 degrees, 270 degrees, and 315 degrees. The phase. Wherein in the first frequency band receiving mode, one of the differential local frequency mixing circuits receives a phase of a multi-phase local oscillation signal and a phase of 180 degrees, and the other one of the differential local frequency mixing circuits receives The multi-phase local oscillation signal has phases of 90 degrees and 270 degrees to generate a first differential fundamental frequency signal according to differential amplification of the RF signal respectively. In the second frequency band receiving mode, one of the differential subharmonic mixing circuits receives the phase of the 201216629 degree, 90 degree, 180 degree, and 270 degrees in the multiphase local oscillation signal, and the differential subharmonic mixing The other of the circuits receives the phase of 45 degrees, 135 degrees, 225 degrees, and 315 degrees in the multi-phase local oscillation signal to generate a second differential fundamental frequency signal according to the differential amplification RF signal mixing, respectively. According to still another embodiment of the present disclosure, the multi-phase local oscillator is generated by dividing the frequency of the oscillating signal source, and the frequency of the oscillating signal source is about four times that of the multi-phase local oscillating signal. Another aspect of the present disclosure is to provide an RF signal receiving method for use in a dual mode RF transceiver. The RF signal receiving method includes the following steps: when the antenna is received in the first frequency band receiving mode, it is located in the first frequency band. And in the second frequency band receiving mode, the RF signal in the second frequency band, wherein the second frequency band is about twice the first frequency band. According to the RF signal 'in the first frequency band receiving mode, the differential inductance circuit has a first inductance value, and in the second frequency band receiving mode, the differential inductance circuit has a second inductance value, and the differential amplifier circuit is coupled to generate a difference. Amplify the RF signal. A multi-phase local oscillation signal is generated, the frequency of which is approximately in the first frequency band. When the first frequency band is connected to the φ receiving mode, the differential differential frequency mixing circuit is enabled to respectively receive the two phases of the multi-phase local oscillation signal as the inversion phase 俾 according to the differential amplification RF signal mixing to generate the first Differential fundamental frequency signal. When the second frequency band receiving mode is located, the multi-phase differential sub-harmonic mixing circuit is enabled to receive four phases of orthogonality in the multi-phase local vibration signal, and the 俾 is superimposed to generate the first frequency according to the differential amplification RF signal mixing. Two differential fundamental frequency signals. According to an embodiment of the present disclosure, when the first frequency band receiving mode is enabled to enable the differential local frequency mixing circuit, the method further includes: causing the multi-phase local oscillator signal to be unused to generate the first and second differential fundamental frequencies. The signal part is offset by S, 7 201216629 by the redundant mixer. According to another embodiment of the present disclosure, the first differential baseband signal and the second differential baseband signal are in-phase differential fundamental frequency signals or quadrature phase differential fundamental frequency signals. According to still another embodiment of the present disclosure, the step of generating the multi-phase local oscillation signal further comprises frequency division by the oscillating signal source, and the frequency of the oscillating signal source is about four times of the multi-phase local oscillation signal. The advantage of applying the present disclosure is that a multi-phase local oscillation signal can be used to achieve dual-mode RF signal reception by means of a switchable differential local frequency mixing circuit and a differential sub-harmonic mixing circuit, thereby achieving savings. The power and area functions, and can avoid the oscillating signal source as the RF signal interference, and easily achieve the above purpose. [Embodiment] Please refer to Fig. 1. 1 is a block diagram of a dual mode RF transceiver 1 in an embodiment of the present disclosure. The dual mode RF transceiver 1 includes an antenna 10, a differential low noise amplification module 12, a local oscillator 14, and a dual mode differential mixing module 16. The antenna 10 is operable in a first band receive mode and a second band receive mode. For example, in the first frequency band receiving mode, the antenna port can receive the RF signal 11 of about 2 GHz (F〇), and in the second band receiving mode, the antenna 10 can receive the RF signal 11 of about 5 GHz (2Fo). . In this embodiment, the second frequency band is approximately twice the first frequency band. Please also refer to Figure 2. Fig. 2 is a more detailed circuit diagram of the 201216629 differential low noise amplification module 12 of Fig. 1 in an embodiment. The differential low noise amplification module 12 includes a differential amplifier circuit 20 and a differential inductor circuit 22. The differential amplifying circuit 20 includes a first amplifying branch and a second amplifying branch. The first amplification branch includes an input MOS transistor 200a, an output MOS transistor 202a, and an input matching module 204a. The second amplification branch includes an input MOS transistor 200b and an output MOS transistor 202b. And the input matching module 204b. In the present embodiment, the input MOS transistors 200a, 200b and the output MOS transistors 202a, 202b are N-type MOS transistors. The input terminal of the MOS transistor 200a is connected to the antenna 10 from point A and receives the RF signal 11 from the antenna 10. The source of the output MOS transistor 202a is coupled to the drain of the input MOS transistor 2A. The gate of the input MOS transistor 200b is coupled to the coupling potential input terminal of the coupling potential. In this embodiment, the coupling potential is the ground potential gnd°. The input MOS transistor 46b has a load between the coupling potential, such as a resistor. Or the capacitance 'in one embodiment' the impedance of these loads is as small as possible, and even closer to zero is preferred. The source of the output MOS transistor 202b is coupled to the drain of the input MOS transistor 2〇〇b. The output MOS transistor 202a is coupled to the drain of the output MOS transistor 202b to a first output and a second output for generating differential amplification RF signals RF- and RF+. The first amplification branch and the second amplification branch further include input matching board groups 204a and 204b, respectively. The wheel-in matching modules 204a and 204b each include a capacitor and a resistor in this embodiment to provide an input matching function. - The source of the input MOS transistors 200a and 200b is further connected to the aforementioned coupling potential of the ground potential GND by an inductor 2?6. The inductance of 201216629 of inductor 206 is greater than a certain level. In one embodiment, the inductance of the inductor 206 can be greater than 500 ohms or even higher to improve the single-ended to double-ended mechanism. The differential inductor circuit 22 essentially comprises two sets of inductors, one set being the inductors 220a, 222a' and the other set being the inductors 220b, 222b. Inductors 220a, 222a, 220b, 222b are in the form of intermediate tap symmetric inductors and are coupled to first and second outputs of differential amplifier circuit 20 by inductors 222a and 222b. The differential inductor circuit 22 further includes a control circuit 224 for receiving a control signal 223. The control signal 223 can be set to be 〇 in the first frequency band receiving mode, so that the control circuit 224 can turn on the inductors 220a, 222a and the inductors 220b, 222b in the first frequency band receiving mode to provide a larger first inductance value for The lower frequency RF signal n (Fq) is processed. In the second frequency band receiving mode, the control signal 223 is 丨, so that the control circuit 224 only turns on the inductor 222a and the inductor 222b in the second frequency band receiving mode, providing a relatively low second inductance value for the higher frequency. The RF signal 11 (2F.) is processed. • Therefore, after the differential amplifier circuit 20 and the differential inductor circuit 22 are processed, the RF signal is transmitted to the end of the UU elbow from the early end to the double end, and the first and the difference are included. The differential output generates a differential amplification RF signal RF_ and the differential amplification RF signal in Figure 1 is shown as
以表示其為具有RF+及RF-兩個訊號。 刖述之輸入金氧半電晶體200a、200b 、200b與輸出金氧半電To indicate that it has two signals, RF+ and RF-. The input gold oxide semi-transistors 200a, 200b, 200b and the output gold oxide semi-electric
201216629 本地振盪器14產生多相位本地振盪訊號l〇。於本實 施例中’本地振盪器14是自頻率約為4Fo的振盪訊號源經 過二次除頻後產生’並為八相位之本地振盪訊號，因此於 第1圖中繪示為LO[7:0]。八相位包含〇度、45度、90度、 135度、180度、225度、270度以及315度之相位，並於 第 1 圖中繪示為訊號 LOj)。、l〇_45。、LO_90。、LO_135。、 • LO_180。、LO_225。、LO—270。以及 LO_315。。由於是由 4Fo 的振盪訊號源經過二次除頻後產生，因此多相位本地振盪 φ 訊號L〇之頻率約位於第一頻段，即約為Fo。 請同時參照第3圖。第3圖為本揭示内容一實施例中， 雙模差動混頻模組16之示意圖。雙模差動混頻模組16包 含：切換模組30、差動本頻混頻電路32a、32b以及差動 次譜波混頻電路34a、34b。 切換模組30可由傳輸閘或是其他方式實現。切換模組 30接收八相位之本地振盪訊號L0_〇。、l〇_45。、LO_90。、 LO—135°、LO_18〇°、LO 225。、LO 270。以及 LO 315。， — — _ • 並在第一頻段接收模式時，傳送本地振盪訊號至差動本頻 混頻電路32a及32b ’因此將這些傳輸至差動本頻混頻電 路32a及32b的本地振盪訊號表示為F()_〇。、F〇_45。、Fq_90 °、Fo一135°、Fo_180°、Fo 225。、Fo 270。以及 F〇 315。。 * — _ 差動本頻混頻電路32a及32b於本實施例中分別為一 吉伯特混頻器。請同時參照第4圖，為差動本頻混頻電路 32a放大之示意圖。此吉伯特混頻器包含四個混頻單元， • 於本實施例中各個混頻單元為一個N型金氧半電晶體 320、322、324及326。其中，差動本頻混頻電路32a之四 201216629 個N型金氧半電晶體320、322、324及326的閘極由於在 第一頻段接收模式時分別自切換模組30接收本地振盪訊 號中反相之二相位Fo_0。及Fo_180°而被致能，並根據由源 極來的差動放大射頻訊號RF+及RF-進行混頻，在汲極產 生第一差動基頻訊號。於本實施例中，差動本頻混頻電路 32a所產生的第一差動基頻訊號為同相差動基頻訊號1+及 需注意的是，差動本頻混頻電路32a於本實施例中更 φ 包含冗餘混頻器321、323、325及327之設置。冗餘混頻 器俾使多相位本地振盪訊號中，未為吉伯特混頻器使用以 產生第一及第二差動基頻訊號之部份，如Fo_45。、Fo_135 °、F〇_225°以及Fo_315。，可以藉由冗餘混頻器321、323、 325及327抵消。由第4圖可以得知，差動放大射頻訊號 RF+與Fg_45°混頻出的部份，可與差動放大射頻訊號RF+ 與Fo_225°混頻出的部份相抵消，差動放大射頻訊號RF-與F〇_225°混頻出的部份，可與差動放大射頻訊號RF-與 φ Fo-4r混頻出的部份相抵消。因此，於差動本頻混頻電路 32a中未使用到的本地振盡訊號之相位，可以藉由抵消而 不會對實際需要進行混頻的部份造成干擾。 同樣地’差動本頻混頻電路32b之四個N型金氧半電 晶體的閘極由於在第一頻段接收模式時分別自切換模組30 接收本地振盪訊號中反相之二相位F()_9〇。及Fo_270。而被 致能，並根據由源極來的差動放大射頻訊號RF+及RF-進 行混頻’在汲極產生第一差動基頻訊號。於本實施例中， 差動本頻混頻電路32b所產生的第一差動基頻訊號為正交 201216629 相差動基頻訊號Q+及Q-。而未使用到的Fo_225°及Fq_315 。之相位亦將藉由冗餘混頻器之設置而抵消。 在第二頻段接收模式時，切換模組30亦可接收八相位 之本地振盪訊號 LO_0、LO_45、LO_90、LO_135、LO_180、 LO_225、LO_270以及LO_315，並傳送本地振盪訊號至差 動本頻混頻電路32a及32b，因此將這些傳輸至差動本頻 混頻電路32a及32b的本地振盪訊號表示為2Fg_0、 2Fo_45、2Fo_90、2Fo_135、2Fo_180、2Fo_225、2Fo—270 以 及 2Fo_315。 差動次諧波混頻電路34a及34b於本實施例中分別為 一吉伯特混頻器。請同時參照第5圖，為差動次諧波混頻 電路34a放大之示意圖。此吉伯特混頻器包含四個混頻單 元，於本實施例中各個混頻單元為二個N型金氧半電晶體 340 與 341、342 與 343、344 與 345、346 與 347。其中， 差動次諧波混頻電路34a之八個N型金氧半電晶體340-347 的閘極由於在第一頻段接收模式時分別自切換模組30接 收本地振盪訊號中正交之四相位2Fo_0 °、2Fo_90 °、2Fo_l 80 "及2Fo_270 °而被致能，並根據由源極來的差動放大射頻訊 號RF+及RF-進行混頻，在汲極產生第一差動基頻訊號。 由於本地振盪訊號具有相差90度相位之本地振盪訊號，因 此可自原本為頻率為Fo之本地振盪訊號藉由疊加的方式產 生2Fg之本地振盪訊號並與差動放大射頻訊號RF+及RF-進行混頻。於本實施例中，差動次諧波混頻電路34a所產 生的第二差動基頻訊號為同相差動基頻訊號R及I-。 同樣地，差動次諧波混頻電路34b之八個N型金氧半 201216629 電晶體的閘極由於在第二頻段接收模式 30接收本地振盪訊號中正交之四相位邛77別自切換模組 2F0—225。及 2F〇-315。而被致能，並根據 一 135°、 大射頻訊號RF+及RF.進行混頻，在=極來的差動放 頻訊號。於本實施例中，差動次譜波_$=差，基 的第二差動基頻訊號為正交相差動基頻訊所。生 因此，在本實施例中，可使用同一 訊號達到雙模式_頻職接收，不需c本地振盪 設置額外的本地振盪訊號轉換電路，不同模式而 率與面積之功效。設置在較射頻訊號之頻功 號源’也可避免為射頻訊號干擾。再^⑤的振盪訊 振盪訊號’可藉由較低的頻率達到較低的走:j二：夕=立 的走線損耗，縣進-步使輸人正交、、、減。較低 得改善。 父1〇唬不匹配的現象獲 可切換於雙模式的差動低雜訊放大模組，將 =積節省。差動本頻混頻電路以及差動次料混頻電=之 =桓，可根據同—個多相位本地縫訊號針對不同頻段 接收射頻訊號以進行直接降m意的是，上 遂差動本頻混頻電路以及差動次諧波混頻電路中，以 形成之實施方式，是以N型金氧半電晶體為:】。 混頻單元亦可為P型金氧半電晶體形成，而並非 必需由Ν型金氧半電晶體形成。 請參照第6圖。第6圖為本揭示内容—實施例中，射 =接收方法之流程圖。射頻訊號接收方法可應用於如 第1圖所示之雙模射頻收發器i卜射頻訊號接收方法包 201216629 含下列步驟。 於步驟601，根據天線10接收於第一頻段接收模式時 位於第一頻段且於第二頻段接收模式時位於第二頻段之射 頻訊號11，其中第二頻段約為第一頻段之兩倍。於步驟 602 ’根據射頻訊號11，於第一頻段接收模式時使差動電 感電路22具有第一電感值，並於第二頻段接收模式時使差 •動電感電路22具有第二電感值，俾與差動放大電路20輕 合產生差動放大射頻訊號RF+及RF-。於步驟603，產生多 φ 相位本地振盪訊號LO[7:0]，其頻率約位於第一頻段。 當在步驟604位於第一頻段接收模式時，執行步驟 605 ’致能差動本頻混頻電路32a及32b，以分別接收多相 位本地振盪訊號LO[7:0]中為反相之二相位，俾根據差動放 大射頻訊號11混頻產生第一差動基頻訊號I+、I-或是Q+、 Q·。 當在步驟606位於第二頻段接收模式時，執行步驟 607，致能差動次諧波混頻電路34a及34b，以接收多相位 φ 本地振盪訊號LO[7:0]中為正交之四相位，俾疊加以根據差 動放大射頻訊號Η混頻產生第二差動基頻訊號1+、I-或是 Q+、Q_。 雖然本揭示内容已以實施方式揭露如上，然其並非用 以限定本揭示内容，任何熟習此技藝者，在不脫離本揭示 内容之精神和範圍内，當可作各種之更動與潤飾，因此本 揭示内容之保護範圍當視後附之申請專利範圍所界定者為 準〇 15 201216629 【圖式簡單說明】 為讓本揭示内容之上述和其他目的、特徵、優點與實 施例能更明顯易懂，所附圖式之說明如下： 第1圖為本揭示内容之一實施例中，一種雙模射頻收 發器之方塊圖。 第2圖為第1圖之差動低雜訊放大模組更詳細之電 圖。 一第3圖為本揭示内容一實施例中，雙模差動混頻模組 之示意圖。 第4圖為第3圖中之差動本頻混頻電路放大之示意圖 第5圖為第3圖中之差動次諧波混頻電路放大之示专 圖 μ 第6圖為本揭示内容一實施例中，射頻訊號接收 ^ ^ /ΖΓ 之流程圖。 【主要元件符號說明】 10 :天線 12 :差動低雜訊放大模組 16 :雙模差動混頻模組 200a、200b :輸入金氧半電晶 體 I :雙模射頻收發器 II :射頻訊號 14 :本地振盪器 20 :差動放大電路 202a、202b :輸出金氧半電晶 體 204a、204b :輸入匹配模組 206 :電感 220a、222a、220b、222b :電 201216629 223:控制訊號 感 224 :控制電路 30 :切換模組 32a、32b :差動本頻混頻電路320、322、324、326 : N型金 32卜323、325、327 :冗餘混氧半電晶體 頻器 34a、34b:差動次諧波混頻電 340-347 :N型金氧半電晶體路 601-607 :步驟201216629 The local oscillator 14 generates a multi-phase local oscillation signal. In the present embodiment, the local oscillator 14 is a local oscillation signal that is generated by the secondary frequency division of the oscillating signal source having a frequency of about 4Fo, and is eight phases. Therefore, it is shown as LO[7 in FIG. 0]. The eight phases include phases of twist, 45 degrees, 90 degrees, 135 degrees, 180 degrees, 225 degrees, 270 degrees, and 315 degrees, and are shown as signal LOj in Figure 1. , l〇_45. , LO_90. , LO_135. , • LO_180. , LO_225. , LO-270. And LO_315. . Since the 4Fo oscillating signal source is generated after the second frequency division, the frequency of the multi-phase local oscillation φ signal L 约 is located in the first frequency band, that is, approximately Fo. Please also refer to Figure 3. FIG. 3 is a schematic diagram of a dual mode differential mixing module 16 according to an embodiment of the disclosure. The dual mode differential mixing module 16 includes a switching module 30, differential local frequency mixing circuits 32a, 32b, and differential sub-spectral mixing circuits 34a, 34b. Switching module 30 can be implemented by a transfer gate or by other means. The switching module 30 receives the eight-phase local oscillation signal L0_〇. , l〇_45. , LO_90. , LO—135°, LO_18〇°, LO 225. , LO 270. And LO 315. And — _ • and transmitting the local oscillation signal to the differential local frequency mixing circuits 32a and 32b when in the first frequency band receiving mode. Therefore, the local oscillation signals are transmitted to the differential local frequency mixing circuits 32a and 32b. For F()_〇. , F〇_45. , Fq_90 °, Fo 135 °, Fo_180 °, Fo 225. , Fo 270. And F〇 315. . * - _ The differential local frequency mixing circuits 32a and 32b are respectively a Gilbert mixer in this embodiment. Please refer to Fig. 4 at the same time, and the schematic diagram of the differential main frequency mixing circuit 32a is enlarged. This Gilbert mixer contains four mixing units. • In this embodiment, each mixing unit is an N-type MOS transistor 320, 322, 324, and 326. The gates of the 201216629 N-type MOS transistors 320, 322, 324 and 326 of the differential local frequency mixing circuit 32a receive the local oscillation signal from the switching module 30 respectively in the receiving mode of the first frequency band. Inverted two phase Fo_0. It is enabled with Fo_180°, and is mixed according to the differential amplified RF signals RF+ and RF- from the source to generate the first differential fundamental frequency signal at the drain. In this embodiment, the first differential fundamental frequency signal generated by the differential local frequency mixing circuit 32a is the in-phase differential fundamental frequency signal 1+. It should be noted that the differential local frequency mixing circuit 32a is implemented in the present embodiment. In the example, φ includes the settings of the redundant mixers 321, 323, 325, and 327. The redundant mixer 俾 makes the multi-phase local oscillator signal not used by the Gilbert mixer to generate portions of the first and second differential fundamental signals, such as Fo_45. , Fo_135 °, F〇_225°, and Fo_315. Can be offset by redundant mixers 321, 323, 325, and 327. It can be seen from Fig. 4 that the portion of the differentially amplified RF signal RF+ and Fg_45° can be offset by the portion of the differentially amplified RF signal RF+ and Fo_225°, and the differentially amplified RF signal RF-and The F混_225° mixed portion can be offset by the portion of the differential amplified RF signal RF- and φ Fo-4r. Therefore, the phase of the local oscillating signal that is not used in the differential local frequency mixing circuit 32a can be canceled without causing interference to the portion that is actually required to be mixed. Similarly, the gates of the four N-type MOS transistors of the differential frequency mixing circuit 32b receive the inverted phase F of the local oscillation signal from the switching module 30 due to the receiving mode in the first frequency band. )_9〇. And Fo_270. And being enabled, and based on the differential amplification of the RF signal RF+ and RF- from the source, the first differential baseband signal is generated at the drain. In this embodiment, the first differential fundamental frequency signal generated by the differential local frequency mixing circuit 32b is orthogonal to the 201216629 phase differential fundamental frequency signals Q+ and Q-. The unused Fo_225° and Fq_315 are used. The phase will also be offset by the setting of the redundant mixer. In the second frequency band receiving mode, the switching module 30 can also receive the eight-phase local oscillation signals LO_0, LO_45, LO_90, LO_135, LO_180, LO_225, LO_270, and LO_315, and transmit the local oscillation signal to the differential local frequency mixing circuit. 32a and 32b, therefore, the local oscillation signals transmitted to the differential local frequency mixing circuits 32a and 32b are represented as 2Fg_0, 2Fo_45, 2Fo_90, 2Fo_135, 2Fo_180, 2Fo_225, 2Fo-270, and 2Fo_315. The differential subharmonic mixing circuits 34a and 34b are respectively a Gilbert mixer in this embodiment. Please also refer to Fig. 5 for a schematic diagram of the amplification of the differential subharmonic mixing circuit 34a. The Gilbert mixer comprises four mixing units. In this embodiment, each mixing unit is two N-type MOS transistors 340 and 341, 342 and 343, 344 and 345, 346 and 347. The gates of the eight N-type MOS transistors 340-347 of the differential sub-harmonic mixing circuit 34a receive the orthogonal four of the local oscillation signals from the switching module 30 respectively in the first frequency band receiving mode. The phases 2Fo_0 °, 2Fo_90 °, 2Fo_l 80 " and 2Fo_270 ° are enabled, and are mixed according to the differential amplified RF signals RF+ and RF- from the source to generate the first differential fundamental frequency signal at the drain. . Since the local oscillation signal has a local oscillation signal with a phase difference of 90 degrees, the local oscillation signal of the original frequency of Fo can be generated by superimposing a 2Fg local oscillation signal and mixed with the differential amplification RF signal RF+ and RF-. frequency. In the present embodiment, the second differential fundamental frequency signal generated by the differential subharmonic mixing circuit 34a is the in-phase differential fundamental frequency signals R and I-. Similarly, the gates of the eight N-type MOS-2216629 transistors of the differential sub-harmonic mixing circuit 34b receive the four-phase 邛77 of the local oscillation signal in the second frequency band receiving mode 30. Group 2F0-225. And 2F〇-315. It is enabled and mixed according to a 135°, large RF signal RF+ and RF. The differential amp signal is at the pole. In this embodiment, the differential sub-spectral wave _$= is poor, and the second differential fundamental frequency signal of the base is a quadrature phase differential base frequency. Therefore, in this embodiment, the same signal can be used to achieve dual mode _ frequency reception, without the need for c local oscillation to set an additional local oscillator signal conversion circuit, different modes and rate and area effects. Radio frequency signal interference can also be avoided by setting the frequency source of the RF signal. Then, the oscillation signal of the ^5 can reach a lower pass by the lower frequency: j: the eve = the trace loss of the line, and the county advances the step to make the input orthogonal, and subtract. Lower improvement. The phenomenon that the parent 1〇唬 does not match can be switched to the dual mode differential low noise amplification module, which will save = product. The differential local frequency mixing circuit and the differential secondary mixing power===, according to the same multi-phase local sewing signal, the RF signal is received for different frequency bands to directly reduce the m meaning, the upper differential is In the frequency mixing circuit and the differential subharmonic mixing circuit, the embodiment is formed by using an N-type metal oxide semi-transistor as: The mixing unit may also be formed of a P-type MOS transistor, and is not necessarily formed of a ruthenium-type MOS transistor. Please refer to Figure 6. Figure 6 is a flow chart of the method of receiving = receiving in the disclosure - the embodiment. The RF signal receiving method can be applied to the dual-mode RF transceiver as shown in FIG. 1 . The RF signal receiving method package 201216629 includes the following steps. In step 601, the RF signal 11 is located in the second frequency band when the antenna 10 is in the first frequency band receiving mode and is in the second frequency band receiving mode, wherein the second frequency band is about twice the first frequency band. In step 602', according to the RF signal 11, the differential inductor circuit 22 has a first inductance value in the first frequency band receiving mode, and the differential inductance circuit 22 has a second inductance value in the second frequency band receiving mode. The differential amplifying circuit 20 is coupled with the differential amplifying circuit 20 to generate differential amplified RF signals RF+ and RF-. In step 603, a multi-φ phase local oscillation signal LO[7:0] is generated, the frequency of which is approximately in the first frequency band. When the first frequency band receiving mode is in step 604, step 605' is enabled to enable the differential frequency mixing circuits 32a and 32b to respectively receive the two phases of the multiphase local oscillation signal LO[7:0]. , 混 according to the differential amplification RF signal 11 mixing to generate the first differential fundamental frequency signal I+, I- or Q+, Q·. When it is in the second frequency band receiving mode in step 606, step 607 is executed to enable the differential subharmonic mixing circuits 34a and 34b to receive the quadrature of the multiphase φ local oscillation signal LO[7:0]. The phase, 俾 is superimposed to generate a second differential fundamental frequency signal 1+, I- or Q+, Q_ according to the differential amplification RF signal Η mixing. The present disclosure has been disclosed in the above embodiments, but it is not intended to limit the disclosure, and any person skilled in the art can make various changes and refinements without departing from the spirit and scope of the disclosure. The scope of the disclosure is defined by the scope of the appended claims. The following is a description of the above and other objects, features, advantages and embodiments of the present disclosure. The description of the drawings is as follows: Figure 1 is a block diagram of a dual mode radio frequency transceiver in an embodiment of the disclosure. Figure 2 is a more detailed diagram of the differential low noise amplification module of Figure 1. FIG. 3 is a schematic diagram of a dual mode differential mixing module according to an embodiment of the disclosure. Fig. 4 is a schematic diagram showing the amplification of the differential frequency mixing circuit in Fig. 3. Fig. 5 is an enlarged view of the differential subharmonic mixing circuit in Fig. 3. Fig. 6 is a disclosure of Fig. In the embodiment, the radio frequency signal receives a flow chart of ^^/ΖΓ. [Main component symbol description] 10: Antenna 12: Differential low noise amplification module 16: Dual-mode differential mixing module 200a, 200b: Input MOS semi-transparent crystal I: Dual-mode RF transceiver II: RF signal 14: Local oscillator 20: differential amplifier circuits 202a, 202b: output MOS semi-transistors 204a, 204b: input matching module 206: inductors 220a, 222a, 220b, 222b: electricity 201216629 223: control signal sense 224: control Circuit 30: switching modules 32a, 32b: differential local frequency mixing circuits 320, 322, 324, 326: N-type gold 32 323, 325, 327: redundant mixed oxygen semiconductor transistors 34a, 34b: poor Dynamic Hybrid Mixing 340-347: N-type Gold Oxygen Semi-Crystalline 601-607: Step
17 5；17 5;