TW201014291A

TW201014291A - Receiver

Info

Publication number: TW201014291A
Application number: TW98109891A
Authority: TW
Inventors: Mitsuru Tanabe; Yukio Okada; Mitsuru Maeda
Original assignee: Panasonic Elec Works Co Ltd
Priority date: 2008-03-26
Filing date: 2009-03-26
Publication date: 2010-04-01
Also published as: CN101981842A; WO2009119736A1; CN101981842B; KR101209501B1; KR20100126568A; JP5215704B2; TWI385990B; HK1154131A1; JP2009239545A

Abstract

A receiver (10) is provided with a circuit (16) for discriminating multilevel modulation system, a circuit (17) for selecting phase correction system, a phase correction circuit (14), and a judgment circuit (15). The circuit (16) discriminates a multilevel modulation system used for a modulation signal on the basis of the modulation signal received from a transmitter. The circuit (17) selects a phase correction system to be used for the phase correction of the symbol of the modulation signal from a plurality of prepared phase correction systems on the basis of the multilevel degree of the multilevel modulation system discriminated by the circuit (16). The circuit (14) corrects the phase of the symbol using the phase correction system selected by the circuit (17). The circuit (15) judges the bit string of the symbol phase-corrected by the circuit (14) on the basis of the multilevel modulation system discriminated by the circuit (16).

Description

201014291 六、發明說明：【發明所屬之技術領域】本發明係有關於受信機，尤指利用可調適調變 (Adaptive Modulati〇n )方式之通訊系統所用的受信機。【先前技術】以往，有使用可調適調變方式的通訊系統。此 β 通訊系統具有多值度（位元率）不同的多數個多層次調變方式，例如端視本身所處的環境（線路品質）’切換所使用的多層次調變方式。此時，可依照線路品質取得最佳傳送效率。這裡的多層次調變方式，例如從位元率低的方式開始依次為Β p s κ (Binary Phase Shift Keying ) ^ Q p S K ( Quadrature Phase Shift Keying )，1 6 Q A M ( Quadrature ❼ Amphtude Modulation )、6 4 Q Λ M 等（例如參照曰本公開專利公報2 0 〇 7 — i 5 〇 g 〇 6、j E EE 802-lla-1999)〇不過，構成通訊系統的發信機及受信機分別具有基準訊號來源。基準訊號來源係使用水晶發信 • 器。發信機及受信機之各基準訊號來源的發振頻率 . (以下稱基準頻率）依據水晶發信器的精度略有誤差。其結果在進行調變處理的發信機端、及進行復調處理的受信機端兩者間的基準頻率將產生p p爪 201014291 指令的誤差。此基準頻率的誤差將造成受信機接收數據之相位旋轉的原因。此相位旋轉的出現將大大影響復調後的位元錯誤率（Bit Err〇r Rate:以下稱 B E R )。因此’受信機在復調時將進行接收數據的相位旋轉補償。尤以使用〇 F DM (0rthogonal Frequency Division Multiplexing) ^ 變方式的多層次調變時’其相位旋轉將大受復調時的頻率誤差影響。此是因為每個符號所佔的時間變大，導致相位補償的間隔變長的關係。以彺，作為進行符號相位旋轉補償的相位補償方式’有使用引示副载波（pil〇t Sub_Carrie。的相位補償方式（例如參照日本公開專利公報2 〇〇 8 ^ 2 3 3 9)或使用5丨示符號的相位補償方式（例 ^照曰本公開專利公報2 〇〇 6 — 3 5 2 7 6 )。在使用引示副載波的相位補償方式中，退（施―叫）影響，訊號傳播特性存有較大頻率選擇性的話，補償誤差將變大。 :此’在頻率選擇性強的傳播環境下，以使用引不符號的相位補償方式較有效。於使方面’使用引示符號的相位補償方式不同 D二:副載波的相位補償方式，不會每個OF 相都逐次補償。故多層次調變方式的多值度，就必須將引示符號嵌入調變訊號的間隔縮 1¾ 201014291 J詞適調變方式中有多層次調變方式的相位補：中滿足所層次調變方式，則傳送效率將變差/又低的多此外’使用弓丨示符號的相位號後才會進行相位補償則貞在取传引不符佣頂故進仃相位補償的pg h 長。因此，若在高多值产 A B較復調處理，相位旋轉的“，調變方式下進行焚轉的誤差恐將超過容許範圍。201014291 VI. Description of the Invention: [Technical Field of the Invention] The present invention relates to a receiver, and more particularly to a communication machine for a communication system using an adaptive modulation mode (Adaptive Modulating). [Prior Art] In the past, there was a communication system using an adaptable modulation method. This β communication system has a plurality of multi-level modulation methods with different multi-valued (bit rate), for example, a multi-level modulation method used for switching the environment (line quality) in which the end view itself is located. In this case, the best transmission efficiency can be achieved according to the line quality. Here, the multi-level modulation method, for example, from the low bit rate, is Β ps κ (Binary Phase Shift Keying ) ^ Q p SK ( Quadrature Phase Shift Keying ), 1 6 QAM ( Quadrature ❼ Amphtude Modulation ), 6 4 Q Λ M, etc. (for example, refer to Japanese Laid-Open Patent Publication No. 20-7-i 5 〇g 〇6, j E EE 802-lla-1999), however, the transmitter and the receiver constituting the communication system respectively have a reference Source of the signal. The source of the reference signal is the use of a crystal transmitter. The vibration frequency of each reference signal source of the transmitter and the receiver (hereinafter referred to as the reference frequency) is slightly different depending on the accuracy of the crystal transmitter. As a result, the reference frequency between the transmitter end that performs the modulation process and the receiver end that performs the reset process produces an error of the p p-claw 201014291 command. This error in the reference frequency will cause the phase of the data received by the receiver to rotate. The appearance of this phase rotation will greatly affect the bit error rate after the polyphonation (Bit Err〇r Rate: hereinafter referred to as B E R ). Therefore, the receiver will perform phase rotation compensation of the received data during the polyphony. In particular, when the multi-level modulation is performed using the 〇 F DM (0 rthogonal Frequency Division Multiplexing), the phase rotation is greatly affected by the frequency error at the time of the polyphonation. This is because the time occupied by each symbol becomes large, resulting in a relationship in which the interval of phase compensation becomes long. In the case of the phase compensation method for performing symbol phase rotation compensation, there is a phase compensation method using a pilot subcarrier (see, for example, Japanese Laid-Open Patent Publication No. ^8^2 3 3 9) or use 5 The phase compensation method of the symbol (example is disclosed in Japanese Laid-Open Patent Publication No. 〇〇6 - 3 5 2 7 6 ). In the phase compensation method using the pilot subcarrier, the effect of retreating, signal propagation If the characteristic has a large frequency selectivity, the compensation error will become larger. This: In the propagation environment with strong frequency selectivity, it is more effective to use the phase compensation method of the unsigned symbol. The phase compensation method is different. D: The phase compensation method of subcarriers does not compensate each OF phase successively. Therefore, the multi-value modulation method must embed the pilot symbol in the interval of the modulation signal by 13⁄4 201014291. In the J word suitable modulation method, there is a multi-level modulation method for the phase complement: in the middle of the modulation mode, the transmission efficiency will be worse/lower, and the use of the phase number of the symbol is used. If the phase compensation is performed, then the pg h length of the phase compensation is not satisfied. Therefore, if the high-multi-value production AB is more complex, the phase rotation is "the error of the combustion in the modulation mode." Will exceed the allowable range.

此類相位補償方式不適合多層次二在的受信機對於來自發信機的調變訊號，使用：同的相位補償方式。因此，以往的受信機發信機所用的多層次調變方式，進行最佳相位補償。【發明内容】本發明的目所用的多層本發明係有鑑於前述緣由而製成。的在提供一種受信機，其能端視發信機次調變方式’進行最佳相位補償。 /、賊獨:一起用於可調適者’該發信機細以發送，變訊號，該調變訊號係使用端視線路品質從多數個夕層次調變方式選擇的多層次調變方式所生成者夕述調變訊號具有符號列，其係用以顯示發送至前$ 受信機的數據。前述符號係根據前述發信機所選擇的多層次調變方式，決定與位元列的對應關係：述受信機備有多層次調變方式判別部、相位補償方 201014291 j擇部、相位補償部、及決策部。前述多層，方式_部的構成是，根據接收自前述發信機的 :::::號’判别前述調變訊號所用的前述多層根攄！。則述相位補償方式選擇部的構成是，次調式判別部所判別的前述多層 .,式的夕值度，從多數個預備的相位補償方，，選擇用於前述調變訊號之符號相位補償的相 =償方式。前述相位補償部的構成是，使用前述相位=償方歧擇部所選擇的前述相位補償方式， =償則述符號的相位。前述決策部的構成是，前述調變方式判別部所判別的前述多層次調變方 ^ ’判定對應於已由相位補償部補償相位後之前述付就的位元列。依據本發明，即可根據前述發信機使用之前述多層次調變方式的多值度’選擇相位補償方式。因此，就能端視前述發信機使用的前述多層次調變方式’進行最佳相位補償。較佳為，前述發信機具有一次調變方式與二次調變方式。前述-次調變方式係從多值度不二多數個多層次調變方式中以預定基準選擇的多層次調變方式，生成顯示前述符號的一次調變符號;前述二次調變方式係多載波調變方式。前述二次調變方式係根據前述-次調變符號，而將構成複數振幅的多個副載波重疊，生成二次調變符號，構成由多數 201014291 調變：二調變符號所構成的前述調變訊號。前述號係二t母預定時間具有引示符號。前述引示符述已知ί，信機而言為已知的二次調變符號。前波所椹a人凋變付號係以具有已知複數振幅的副載 :構：截前述二次調變符號包含引示副載波。前 :田載波係對前述受信機而言為已知的副載相位==的副載波具有已知的複數振幅。前述 ”選擇部的構成是’在前述多層次調變 :別部：判別之前述多層次調變方式的前述多預定值的話，即選擇用前述弓丨示符號補償 :述付5虎之相位的第-相位補償方式，而在前述多值度達到既定值以上的話’即選擇用波補償前述符號之相位的第二相位補償方式。 Φ 此時，前述發信機使用之前述多層次調變方式的多值度小於職值的話，即使頻率選擇性強的傳播環境’亦能使用相位補償效果大的前進行相位補償。另-方面，前述發信機使^前^ 多層次調變方式的多值度達到預定值以上的用容易實現高傳送效率之前述引示副載波進行相位補償。因此’能端視前述發信機使用的前述多調變方式’進行最佳相位補償。更佳為，前述相位補償方式選擇部的構成a , 在前述多層次調變方式判別部所判別之前述多=欠調變方式的前述多值度小於預定值的話，僅選^前 201014291 述第一相位補償方式，以上的話，選擇前述第而f則述夕值度達到預定值相位補償方式兩者。飞〃、則述第二此時，經常使用前述引位補償。因此，俞、/ 疋仃刖述第一相再：卜述相位補償部的控制將變簡單。在則述第一相位補償方彳 — 副載波執行相位補償。因此；、在母個前述的傳播環境下:亦能經常加大相位補=效選果擇性強位補償方式選的:冓成是’在前述相述第二相位補俨古斗工女征顸俏方式與别相位補償方犬情況下，遵照前述第二償方式制㈣述符號的相位後，再遵照前述目位補償方式補償前述符號的相位。位補ί時’絲使Μ示副餘將每個符號進行相誤員，且使用引示副載波進行相位補償所產生的因’將因使用引示符號進行的相位補償而消除。，能針對多值度大的多層次調變方式，進行最佳相位補償。償立較佳為，前述預定值之設定係根據前述相位補用則述相位補償方式選擇部所選擇的前述相彳員方式進行前述符號之相位補償之情況下的傳廷致率。式匕時’連端視前述發信機使用的多層次調變方進行最佳相位補償，且提高傳送效率。 201014291 【實施方式】 (第一實施形態） r機的叉仏機1〇如第二圖所示，與發、:適應調變通訊系統（以下稱為工通訊系統依據發信機20調變（本實施 t為0FDM調變）的〇FDM訊號，進行封匕L訊此夕卜冑關發信機傳送調變This type of phase compensation is not suitable for multi-level two-way receivers. For the modulation signal from the transmitter, use the same phase compensation method. Therefore, the multi-level modulation method used by the conventional receiver transmitter performs optimal phase compensation. SUMMARY OF THE INVENTION The present invention has been made in view of the foregoing. In providing a receiver, it can perform optimal phase compensation by looking at the transmitter sub-modulation mode. /, thief alone: together for the adaptable person 'the transmitter is fine to send, change signal, the modulation signal is generated by using the multi-level modulation method of the end line line quality selected from most tier-level modulation methods. The modulating signal has a symbol column that is used to display the data sent to the previous receiver. The symbol is determined according to the multi-level modulation method selected by the transmitter, and the correspondence relationship with the bit column is determined: the receiver is provided with a multi-level modulation method determination unit, a phase compensation unit 201014291, a selection unit, and a phase compensation unit. And the decision-making department. The above-described multi-layer, mode_section is configured to discriminate the aforementioned multi-layered roots for the modulated signal based on the ::::: number received from the transmitter! . The phase compensation method selection unit is configured to select the mid-level value of the multi-layer equation determined by the sub-modulation determination unit, and select a phase phase compensation for the modulation signal from a plurality of preliminary phase compensation parties. Phase = compensation method. The phase compensating unit is configured to use the phase compensation method selected by the phase=compensation determining unit to compensate the phase of the symbol. The decision unit is configured such that the multi-level modulation unit determined by the modulation method determination unit corresponds to the bit line sequence after the phase compensation unit has compensated the phase. According to the present invention, the phase compensation method can be selected in accordance with the multi-valued degree of the multi-level modulation method used by the aforementioned transmitter. Therefore, the optimum phase compensation can be performed by looking at the aforementioned multi-level modulation method used by the aforementioned transmitter. Preferably, the transmitter has a primary modulation mode and a secondary modulation mode. The first-order modulation method generates a first-order modulation symbol for displaying the symbol from a multi-level modulation method selected from a plurality of multi-level modulation methods with a predetermined value; the second modulation method is Multi-carrier modulation mode. The quadratic modulation method superimposes a plurality of subcarriers constituting a complex amplitude according to the above-described submodulation symbol to generate a quadratic modulation symbol, and constitutes the aforementioned modulation composed of a majority of 201014291 modulation: two modulation symbols. Change signal. The aforementioned number two t-mother predetermined time has a sign. The foregoing indications are known to be known as secondary modulation symbols. The front wave is a sub-carrier with a known complex amplitude: the truncation of the aforementioned quadratic modulation symbol includes the pilot subcarrier. The subcarriers of the subcarriers that are known to the receiver as known for the subcarrier phase == have a known complex amplitude. The configuration of the selection unit is: in the multi-level modulation: the other part: the plurality of predetermined values of the multi-level modulation method are determined, that is, the symbol compensation is selected by the aforementioned arrow: the phase of the 5 tiger is described. In the first-phase compensation method, when the multi-valued degree is equal to or greater than a predetermined value, the second phase compensation method for compensating the phase of the symbol by using the wave is selected. Φ At this time, the multi-level modulation method used by the transmitter is used. If the multi-valued value is less than the job value, even if the frequency selective environment is strong, the phase compensation can be performed before the phase compensation effect is large. On the other hand, the aforementioned transmitter enables the multi-level modulation method. If the value is equal to or greater than a predetermined value, phase compensation is performed by using the above-described pilot subcarrier which is easy to realize high transmission efficiency. Therefore, it is possible to perform optimum phase compensation by looking at the multi-modulation method used by the transmitter. More preferably, In the configuration a of the phase compensation method selection unit, when the multi-value degree of the multi-under-modulation method determined by the multi-level modulation method determination unit is smaller than a predetermined value, only the selection is performed. ^前201014291 The first phase compensation method is described above. In the above case, the above-mentioned fth is selected, and the value of the eigenvalue is equal to the predetermined value phase compensation method. In the second case, the above-mentioned plunging compensation is often used. , Yu, / narrate the first phase again: the control of the phase compensation unit will be simpler. In the first phase compensation method, the subcarrier performs phase compensation. Therefore, in the aforementioned propagation environment of the parent Bottom: It is also possible to increase the phase compensation = effect selection and selective selection of strong position compensation: 冓成 is 'in the second phase of the above-mentioned phase, the 相位俨俨俨俨俨与与与与与与Then, according to the second compensation mode (4), the phase of the symbol is described, and then the phase of the symbol is compensated according to the above-mentioned target compensation method. When the bit is compensated, the wire causes the symbol to be misaligned, and The phase compensation caused by the use of the pilot subcarrier for phase compensation is eliminated by the phase compensation using the pilot symbol. The optimum phase compensation can be performed for the multi-level modulation method with large multi-valued degrees. For, before The predetermined value is set based on the phase compensation when the phase compensation method selected by the phase compensation method selection unit performs the phase compensation of the symbol. The multi-level modulation used by the machine performs the optimal phase compensation and improves the transmission efficiency. 201014291 [Embodiment] (The first embodiment) The forklift 1 of the r machine is as shown in the second figure. Modulated communication system (hereinafter referred to as the industrial communication system according to the transmitter 20 modulation (this implementation t is 0FDM modulation) 〇 FDM signal, the sealing of the L message, the transfer of the transmitter transmission modulation

波至受信機1 0所需的傳送路徑3 Q，以有線或益線任一者皆可。The transmission path 3 Q required to reach the receiver 10 can be either wired or profitable.

發域2 0採用多層次調變方式作為一次調變〇FDM調變作為二次調變。發信機2 〇將發达至受信機1 Q的數據（f訊位元列）轉成錯誤更正碼。且發信機2 〇將錯誤更正碼化後的數據進行串並聯轉換。發信機2 0根據與多層次調變方式決定之符號與位元列的對應關係，從串並聯轉換後的數據形成（符號對應）調變副載波所需的複數符號 (一次調變符號）。發信機2 〇依次將複數符號進行反離散傅立葉轉換（二次調變）後，藉由串並聯轉換形成數位形式的複數基頻OFDM訊號（OFD Μ符號或二次調變符號）。發信機2 〇將複數基頻〇 F DM訊號進行數位/類比轉換（〇 Α轉換）。發信機2 0以去除轉換D A產生之影像訊號的濾波器過濾前述D A轉換後的〇F D Μ訊號，乘上載波（進行頻率轉換），再進行既定訊號增幅，以形成〇 F D 201014291 發信機2 〇將如此形成的前述The hair domain 20 uses a multi-level modulation method as a primary modulation 〇FDM modulation as a secondary modulation. Transmitter 2 converts the data (f-bit column) that was sent to the receiver 1 Q into an error correction code. And the transmitter 2 〇 converts the error-corrected data into serial-parallel conversion. The transmitter 20 forms a complex symbol (primary modulation symbol) required for the modulated subcarrier from the data converted in series and parallel according to the correspondence between the symbol determined by the multi-level modulation method and the bit column. . The transmitter 2 反 performs inverse discrete Fourier transform (secondary modulation) on the complex symbols in turn, and forms a complex fundamental OFDM signal (OFD Μ symbol or quadratic modulation symbol) in digital form by serial-parallel conversion. Transmitter 2 数 performs digital/analog conversion (〇 Α conversion) of the complex fundamental frequency 〇 F DM signal. The transmitter 20 filters the DA-converted 〇FD signal by removing the signal of the image signal generated by the conversion DA, multiplies the carrier (for frequency conversion), and then performs the predetermined signal amplification to form the 〇FD 201014291 transmitter. 2 前述 will be formed as described above

OFDM Μ調變波。發信機2 〇調變波發送至傳送路徑3 〇 ΟOFDM Μ 变变。. Transmitter 2 调 Transmitted wave is sent to transmission path 3 〇 Ο

這裡的發信機2 〇具有多值度不同的多數徊之B 次調變方式’使數據傳送速度變最快。且發信機2 0亦可端視傳送路徑3 〇的狀態（線路品質）或數據容量’選擇多層次調變方式’使傳送速度達到一定值以上。此外’發信機2 〇備有具石英振i器的基準訊號來源（未以圖示）。發信機2 〇使用基準訊號來源發出的基準頻率，進行反離散傅立葉轉換（〇fd Μ調變）或頻率轉換。此外，受信機i 〇亦備有基準訊號來源。 & 此類發#機2 0從多個多層次調變方式（1 6 QAM，6 4QAM)使用以既定基準選取的多層次調變方式’發送已生成的〇F DM調變波。這裡的6 4 QAM等高多值度的多層次調變方式中，使用複數平面上的複數符號決策符號位元列時’其容許誤差角度彳艮小。因此，每個符號都必須 201014291 進行相位補償。以下將以1 6 QAM為例，說明Q AM的容許誤差角度。第三圖表示複數平面上對於丄6 QAM位元列〔0 0 0 ◦〕〜〔1 1 1 1〕的符號配置（訊號點 1己置）、。廷裡是以格雷碼為前提。第三圖中的區隔線 ^以Q (Quadrature-Phase)軸方向，通過連結位几列〔1 1 1 〇〕與位元列〔丄〇丄〇〕各符號點間之線的中間。區隔線L2W (In，ase)軸方向’通過連結位Μ〔1010〕與位元列〔！ on〕各符號點間之線的中間。受信機1〇接收 =複數符號若存在於包含位元列〔i Q丄◦〕之區 jL 1、L 2所包圍的領域A丄2 —的話，此複數付號將被推測表示位元列〔1G1G〕的機率較 :H實際上’依據基準頻率的誤差造成相位旋轉（參照箭頭Q 1 )，肩太& Λ… ）原本在發信機2 0存在於領 =2的複數符號’在受信機1〇的話恐會變成頁錯誤。此蛉，位兀列的決策將出現對應於受信機1 〇接收之位_〔 i 〇的複數符號’若基準頻率無誤差的話，將以表示丄兀列〔1 0 1 0〕符號點為中心進行常態分布此，以符號點不脫離原本領域為條件，因號點的容許誤差角度01時㈣會出現 ^ 以16QAM的情況，容許誤差角度q〔de^ 11 201014291 〕為 1 6 . 8 8。表1顯示具有代表性之QAM的容許誤差角度 Θ 1。從表1明顯可知，隨多值度變大，容許誤差角度0 1將變小。【表1】多層次調變方式 16QAM 64QAM 256QAM 10240AM 容許誤差角度0 1 ( degree) 16.88 10.55 7.69 6.06Here, the transmitter 2 has a multi-valued B-modulation method that makes the data transmission speed the fastest. Further, the transmitter 20 can also view the state of the transmission path 3 (the line quality) or the data capacity 'select the multi-level modulation method' to make the transmission speed reach a predetermined value or more. In addition, the transmitter 2 has a reference signal source (not shown) with a quartz oscillator. Transmitter 2 反 Performs inverse discrete Fourier transform (〇fd Μ modulation) or frequency conversion using the reference frequency emitted by the reference signal source. In addition, the receiver i 〇 also has a source of reference signals. & This type of machine 20 transmits the generated 〇F DM modulated wave from a plurality of multi-level modulation modes (1 6 QAM, 6 4QAM) using a multi-level modulation method selected by a predetermined reference. Here, in the multi-level modulation method of the 6 4 QAM contour multi-valued degree, when the complex symbol is used to determine the symbol bit column on the complex plane, the allowable error angle is small. Therefore, each symbol must be phase compensated with 201014291. The following is an example of the Q 6 tolerance error angle with 1 6 QAM as an example. The third figure shows the symbol configuration (signal point 1) for the 丄6 QAM bit column [0 0 0 ◦]~[1 1 1 1] on the complex plane. Tingri is premised on Gray code. The section line in the third figure is in the middle of the line between the symbol points of the number of columns [1 1 1 〇] and the bit column [丄〇丄〇] in the direction of the Q (Quadrature-Phase) axis. The section line L2W (In, ase) axis direction 'through the joint position Μ [1010] and the bit column [! On] The middle of the line between the symbol points. Receiver 1〇Receive=If the complex symbol exists in the field A丄2 surrounded by the region jL 1 and L 2 including the bit column [i Q丄◦], the complex payout will be inferred to represent the bit column [ The probability of 1G1G] is: H actually 'phase rotation according to the error of the reference frequency (refer to arrow Q 1 ), shoulder too & Λ... ) originally in the transmitter 2 0 in the collar = 2 complex symbol 'in the letter If you are a machine, you may become a page fault. In this case, the decision of the bit array will appear corresponding to the bit received by the receiver 1 _[the complex symbol of i 〇 if the reference frequency has no error, it will be centered on the symbol point of the queue [1 0 1 0] The normal distribution is performed, and the symbol point does not deviate from the original field. The tolerance angle of the sign point is 01 (4). When the angle is 16QAM, the allowable error angle q[de^ 11 201014291] is 16.68. Table 1 shows the allowable error angle Θ 1 of a representative QAM. It is apparent from Table 1 that as the multi-value becomes larger, the tolerance error angle 0 1 becomes smaller. [Table 1] Multi-level modulation method 16QAM 64QAM 256QAM 10240AM Tolerance angle 0 1 (degree) 16.88 10.55 7.69 6.06

接著’在OFDM調變中，評估每個OFDM 符號因基準頻率誤差造成的相位旋轉。因〇Fdm © 符號之相位旋轉造成的相位誤差，其要因為二：〇 FDM進行復調處理與所需的載波頻率同步（頻率轉換）時的誤差（第一誤差）’及與取樣頻率同步（快速傅利葉轉換）時的誤差（第二誤差）。若將快速傅利葉轉換的取樣頻率視為f s、快速傅利葉轉換的大小視為（F F T大小）N點、保護區間的時間視為T g i，則每個〇 f D Μ符號的佔有時間丁 a可以下列公式（1 )表示。 ❿ 【數1】 T^ = ~ + Tgi fs ( 1 ) 第一誤差及第二誤差造成的相位誤差為加法性。因此，令載波頻率為f c、調變及復調兩處理 . 間的基準頻率誤差視為e，則每個〇 F D Μ符號的相位誤差角度Θ 2〔 degree〕可以下列公式（2 ) 12 201014291 表示。【數2】 I 2j U ~ τ ( 2 ) 如根據 IEEE 802.11a — 1999 的規格（I EEE (美國電氣電子工程師學會）規定之無線LAN的規格IEEE 802.11a) 只要將各基準訊號來源的基準頻率誤差容許在2 0 ❹ PPm，因調變及復調兩處理後，則變成4〇pp m的基準頻率誤差e。載波頻率f c的誤差根據受信機自動頻率補償電路，通常會收斂至f s/2的頻率誤差。因此，上述公式（2 )將變形成以下公式（3 )。 Ν^τ λ 180 ~T + Tgi β . 【數列3】 ΘΊ ~ Τ.7ζ· fs ❹ 3 — 1 此外’根據I Ε Ε Ε 802 9的規格，快速傅利葉轉換的取樣頻率f s為2丨 MH z、〇 F DM符號的佔有時間？ a為$ ( c (其中保護區間的時間丁 g 土為〇 · 8 c )、快速傅利葉轉換的大小^^為6 4點·。 8 ( 只要遵照I Ε Ε E 8 0 2.li — 9的規格計算上述公式（3)，變成2.88。鋏而，β/ΐηΛΛ 左月度Θ2 …、、而’ 64QAM因四個符號將超 13 201014291 /“〜：度Θ1。且，根據IEee 802 〇〇〇位元？99的規！、，每一封包將要求最大1 餘位元的与、'且：因此’若沒有因錯誤訂正而增加冗成二十七：符:個封包可傳送的0FDM符號約變誤差角；^ 目此’在®四個符號而超過容許 / 1的情況下’將無法正確復調一個封包。〜？，IEEE 802.lla-1999 規疋如第四圖yy規作為與數^，將全部五十二條副載波中的四條 C 據傳送無關的引示副載波PSC1〜Ps s C 0 2下的四十八條作為用於數據傳送的副載波 C 4 7。因此，就能使用引示副載波p 補償。〜P S C 4，進行每個0F DM符號的相位大頻：Z路徑衰減影響’在訊號傳播特性存有較、選擇性的話，補償誤差將變大。被引示副載波嵌入頻率的S/N比恐將變得極差。此時，使用引不副载波的相位補償方式，其補償誤差將變大。因此，尤以高多值度的多層次調變方式，將因實施相位補償’反倒出現B E R變差的情況。例如第四圖中’在引示副載波p S C 1附近，因頻率特性工〇〇〇惡化，以致使用引示副載波P s C 1的相位補償精度變差。、在頻率選擇性強的傳播環境使用引示符號的相位補償方式是有效的。引示符號由受信機丄〇及發 201014291 信機2 0兩者已知的符號構成。且引示符號以既定時間的間隔被嵌入由〇F DM符號構成的調變訊號 (封包）中。因此，即可使用引示符號，進行每個° 副載波的相位補償。然而，使用引示符號的相位補償方式並非每個〇F DM符號都進行補償。因此，多層次調變方式的多值度越大，就必須將引示符號嵌入調變訊號中的間隔縮小。例如1 6 q AM以每五個符號將引示符號嵌入調變訊號中即十分足夠。相較於此，6 4 QAΜ的話，就必須將引示符號嵌入調變訊號中的間隔設定為每三個符號。因此，可調適調變方式中， =欲進行足以滿足所有多層次調變方式的相位補償，尤以多值度低的多層次調變方式，則傳送效率將變差。此外，使用引示符號的相位補償只有取得引示苻號後，才會進行相位補償，故進行相位補償的間隔車乂長。因此，若在高多值度的多層次調變方式下，可能在進行復調處理途中超過容許誤差角度Θ1。因此，發信機2 0將形成調變訊號，以讓受信機10能選擇性的進行使用引示符號的相位補償，及使用引示副載波的相位補償。调變訊號（封包）如第五圖所示由短前置碼s Ρ、長前置碼L Ρ及數據部D構成。短則置碼S Ρ為確立與符號時序同步，由發信 15 201014291 機2 0、受乜機1 〇及已知的同步圖形（特定圖形） X，依照各個基本周期τ工（=〇 · 8 # s e c ) 反覆進行十次（x1〜x1 ο)構成。意即，短前置碼S P由基本周期τ 1的反覆訊號構成。長前置碼L P為能推測頻道，由發信機2 〇、受乜機1 0及已知的同步圖形γ，依照各個基本周期T2 ( = e c)反覆進行兩:欠（γ丄、 Y 2 )構成。Then in the OFDM modulation, the phase rotation caused by the reference frequency error is evaluated for each OFDM symbol. The phase error caused by the phase rotation of the Fdm © symbol is due to the error (first error) of the 复FDM polyphonic processing and the required carrier frequency synchronization (frequency conversion) and synchronization with the sampling frequency (fast Fourier transform) error (second error). If the sampling frequency of the fast Fourier transform is regarded as fs, the size of the fast Fourier transform is regarded as (FFT size) N point, and the time of the guard interval is regarded as T gi, the occupation time of each 〇f D Μ symbol can be as follows Formula (1) is indicated. ❿ [Number 1] T^ = ~ + Tgi fs ( 1 ) The phase error caused by the first error and the second error is additive. Therefore, the reference frequency error between the carrier frequency fc, the modulation and the polyphonation is regarded as e, and the phase error angle Θ 2 [degree] of each 〇 F D Μ symbol can be expressed by the following formula (2) 12 201014291. [Number 2] I 2j U ~ τ ( 2 ) According to the IEEE 802.11a - 1999 specification (I EEE (American Institute of Electrical and Electronics Engineers) wireless LAN specification IEEE 802.11a), as long as the reference frequency of each reference signal source The error is allowed to be 20 ❹ PPm, and after the two processes of modulation and resetting, it becomes the reference frequency error e of 4 〇 pp m. The error of the carrier frequency f c is usually converged to the frequency error of f s/2 according to the automatic frequency compensation circuit of the receiver. Therefore, the above formula (2) will be changed to the following formula (3). Ν^τ λ 180 ~T + Tgi β . [Number 3] ΘΊ ~ Τ.7ζ· fs ❹ 3 — 1 In addition, according to the specifications of I Ε Ε 802 802 9, the sampling frequency fs of the fast Fourier transform is 2丨MH z , 〇F DM symbol possession time? a is $ (c (where the time of the guard interval is 〇·8 c), and the size of the fast Fourier transform ^^ is 6 4 points. 8 (as long as I Ε Ε E 8 0 2.li - 9 The specification calculates the above formula (3), which becomes 2.88. 铗, β/ΐηΛΛ left month Θ 2 ..., and '64QAM because four symbols will exceed 13 201014291 / "~: degree Θ 1. And, according to IEee 802 〇〇〇 Yuan?99's rules!,, each packet will require a maximum of 1 bit, and 'and: therefore' if there is no error due to error correction, add 27: symbol: 0FDM symbol change that can be transmitted by a packet Error angle; ^ This is 'in the case of ® four symbols and exceeds the allowable / 1' will not be able to properly modulate a packet. ~?, IEEE 802.lla-1999 regulations as the fourth figure yy rules as the number ^, Four of the total of fifty-two subcarriers are transmitted as the subcarrier C 47 under the transmission-independent pilot subcarriers PSC1 to Ps s C 0 2 as a subcarrier C 47 for data transmission. Therefore, the pilot can be used. Subcarrier p compensation. ~PSC 4, performing phase multiplication of each 0F DM symbol: Z path attenuation affects 'in signal propagation If the performance is more selective and selective, the compensation error will become larger. The S/N ratio of the subcarrier embedded frequency will be extremely poor. At this time, the phase compensation method of subcarriers is used, and the compensation error is used. Therefore, especially in the multi-level modulation method with high multi-value, the BER will be worsened due to the implementation of phase compensation. For example, in the fourth figure, near the subcarrier p SC 1 , The frequency characteristic process is deteriorated, so that the phase compensation accuracy using the pilot subcarrier P s C 1 is deteriorated. It is effective to use the phase compensation method of the pilot symbol in a propagation environment with strong frequency selectivity. The receiver is configured with symbols known to both the 201014291 signal machine 20, and the pilot symbols are embedded in the modulation signal (packet) composed of the 〇F DM symbols at regular time intervals. The sign is used to perform phase compensation for each ° subcarrier. However, the phase compensation method using the pilot symbol is not compensated for each 〇F DM symbol. Therefore, the multi-level modulation method has a higher degree of multi-valued Must be The spacing of the sign embedding in the modulation signal is reduced. For example, 1 6 q AM is sufficient to embed the sign in the modulation signal every five symbols. In contrast, if 6 4 QAΜ, it must be introduced. The interval in the symbol embedding modulation signal is set to every three symbols. Therefore, in the adaptive modulation method, = is necessary to perform phase compensation sufficient for all multi-level modulation methods, especially multi-level modulation with low multi-valued In this way, the transmission efficiency will be worse. In addition, the phase compensation using the pilot symbol will only perform phase compensation after obtaining the index nickname, so the phase compensation is long. Therefore, in the multi-level modulation mode with high multi-value, it is possible to exceed the allowable error angle Θ1 in the middle of the polyphonic processing. Therefore, the transmitter 20 will form a modulation signal to allow the receiver 10 to selectively perform phase compensation using the pilot symbols and phase compensation using the pilot subcarriers. The modulation signal (packet) is composed of a short preamble s Ρ, a long preamble L Ρ, and a data portion D as shown in the fifth figure. The short code S Ρ is established to synchronize with the symbol timing, and is sent according to each basic period (= 〇 · 8) by sending a signal to the machine 1 2, the receiver 1 and the known synchronization pattern (specific graphics) X. # sec ) Repeat ten times (x1~x1 ο). That is, the short preamble S P is composed of a repetition signal of the basic period τ 1 . The long preamble LP is a speculative channel, and the transmitter 2 〇, the receiver 10 and the known synchronization pattern γ are repeatedly performed according to each basic period T2 (= ec): owe (γ丄, Y 2 ) constitutes.

數據部D是用以傳送儲存數據位元、調變方式等資訊數據的區域。調變訊號以短前置碼S P、長前置碼l P、數The data portion D is an area for transmitting information data such as a data bit and a modulation method. Modulation signal with short preamble S P, long preamble l P, number

據部D、短前置碼S P、長前置碼L P、數據部D 依次配置。隹長則置碼L· P及數據部The data unit D, the short preamble S P , the long preamble L P , and the data unit D are sequentially arranged.隹Long is coded L·P and data department

已複製部分各區域後半的保護區間G 1 1、〇 2。藉由保護區間G I 1、G I 2降低多路秤的The protection intervals G 1 1 and 〇 2 of the second half of each part have been copied. Reducing the multi-way scale by protecting the intervals G I 1 , G I 2

響。工J 又L機10如第一圖所示，備有自動頻率電路（AFC) 1 1、保護區間去除電路1 2、速傅利葉轉換電路（”T) 1 3、相位補償相位補償部）1 4、決策電路（決策部）丄 ::-人調變方式判別電路(多層次調變方式判別^ 、二位補償方式選擇電路（相位補償方式選 ° 。圖中將類比部的訊號增幅、頻率轉換（ 16 201014291 轉換）、去除干涉波濾波器、類比/數位轉換（AD 轉換）等類比訊號處理電路予以省略。自動頻率補償電路11將基頻訊號進行類比/ 數位轉換（AD轉換）後，使用短前置碼”及: 則置碼LP ’進行每個〇Fdm符號的相位旋轉補目動頻率補償電路11ring. As shown in the first figure, the machine J and the L machine 10 are provided with an automatic frequency circuit (AFC) 1 1 , a guard interval removal circuit 1 2, a fast Fourier transform circuit ("T) 13 , and a phase compensation phase compensation unit) 1 4 , decision circuit (decision department) 丄::- human modulation mode discriminating circuit (multi-level modulation mode discrimination ^, two-bit compensation mode selection circuit (phase compensation mode select °. In the figure, analog signal amplification, frequency conversion (16 201014291 conversion), removal of interference wave filter, analog/digital conversion (AD conversion), etc. The analog signal processing circuit is omitted. The automatic frequency compensation circuit 11 performs analog/digital conversion (AD conversion) of the fundamental frequency signal and uses it short. The preamble" and: then the code LP' performs a phase rotation supplemental frequency compensation circuit 11 for each 〇Fdm symbol

川观刖置瑪SChuan Guanyu Ma Ma S

P、，檢測出發信機2 〇的基準頻率與受信機工〇的基準頻率間較大的頻率誤差。頻率誤差的檢測是夢由例如僅將基本周期τ1延遲之調變訊號的共輛複宋上基本周期Τ1後的調變訊號進行。接著’自動頻率補償電路！ i使用長前置碼L ?檢測出頻率誤差。使用長前置碼L P進行頻率誤 ί的檢測是以與使用短前置碼SP進行頻率誤差的檢測相冋程序進行。使用長前置碼檢測出1/ { 下〇 w , 就月b 印丄/ 2 f { 2·6 4 n 小的頻率誤差。二自動頻率補償電路i i使用短前置碼s p及長刚置碼LP，將檢測出的頻率誤差逆相位乘在收的調變錢。藉此，自動頻率補償電相位補償（頻率補償）。進仃保護區間去除電路i 2將由發調變訊號的保護區間…、…去除㈣加在快速傅利葉轉換電路i 3以基準頻率為根據的 17 201014291 取樣頻率，冑0 F DM符號進行離散傅禾J葉轉換。藉此，快速傅利葉轉換電路1 3在多個副載波訊號進仃分波多載波復調。藉此，將各副載波之複數符號的成分篩選出來。 —相位補償電路丄4進行因頻率誤差造成一次調 =符號之相位旋轉補償。相位補償電路i 4備有推疋邛1 4 1、等化部丄4 2、相位誤差去除部丄4 3 〇、、推定部1 4 1使用引示符號，推測每個副載波傳送路控3 Q之頻率領域的脈衝響應。脈衝響應顯示每個副載波的傳播特性。推定部i i將有關接 f於前置碼的調變訊號，其前置碼的已知數據（短前置碼s P的同步圖形X或長前置碼L P同步圖形 :)視為引示符號，推測已知數據的相位旋轉及振幅誤差（每個副載波的脈衝響應）。有關之後的調變讯號’每隔既定時間就有引示符號，故從已知數據推測相位旋轉及振幅誤差。推測的相位旋轉及振幅誤差可在下—引示符號進行補償前視為有效，亦可將下一引示符號的補償量與現有引示符號的補償量附上適虽加權後，作為下—引示符號的補償值。此外，引示符號亦可將所有副载波的複數振幅視為已知數據，但即使僅在頻率選擇性大的頻率領域的副載波嵌人已知數據亦可。此時，未嵌人已知數據之副載波的傳播特性只要根據嵌人已知數據之副载波 201014291 的傳播特性導出即可。等：部！42對於繼前置碼付旒’乘上由推定部丄4 双疚的複數衝響應逆特性。藉此，等化c副载波的脈波的頻率領域歪斜，進行 2補償每個副載轉制員。此外’在振幅變動大的傳送替麵位旋轉，亦可補償振幅誤差。不/、相 ❿ 如此，推定部141及等化付说進行補償符號相位的第一相位補償方：用不引示符號的相位補償將依工使用，故即使在頻率選擇性強的二效果也很大。相位補償且，本實施形態如第四圖所示，副载波中的四條是與數據傳十一條像得送無關的引示副載波p 〜 C4,剩下的四十八條是 Π載波SC〇〜SC47。引示副載波心 1〜PSC4上的符號是已知數據（已知符號）。P, detecting a large frequency error between the reference frequency of the departure signal 2 〇 and the reference frequency of the receiver machine. The detection of the frequency error is performed by, for example, a modulated signal after the fundamental period Τ1 of the common vehicle of the modulated signal delayed by only the fundamental period τ1. Then 'automatic frequency compensation circuit! i uses a long preamble L to detect the frequency error. The frequency error detection using the long preamble L P is performed in contrast to the detection of the frequency error using the short preamble SP. Using a long preamble, 1 / { 〇 w is detected, and the frequency error of the month b 丄 / 2 f { 2·6 4 n is small. The second automatic frequency compensation circuit i i uses the short preamble s p and the long rigid code LP to multiply the detected frequency error by the inverse phase multiplied by the modulated variable. Thereby, the automatic frequency compensation electric phase compensation (frequency compensation). The 仃 protection interval removal circuit i 2 removes the guard interval ..., ... from the tuned signal (4) and adds it to the fast Fourier transform circuit i 3 based on the reference frequency 17 201014291 sampling frequency, 胄 0 F DM symbol for discrete Fu Wo Leaf conversion. Thereby, the fast Fourier transform circuit 13 performs multi-carrier multiplexing on a plurality of subcarrier signals. Thereby, the components of the complex symbols of the respective subcarriers are filtered out. - The phase compensation circuit 丄4 performs phase rotation compensation for the primary modulation = sign due to the frequency error. The phase compensation circuit i 4 includes a pusher 1 4 1 , an equalization unit 丄 4 2 , a phase error removal unit 丄 4 3 〇 , and an estimation unit 1 4 1 using pilot symbols to estimate each subcarrier transmission path 3 The impulse response of the frequency domain of Q. The impulse response shows the propagation characteristics of each subcarrier. The estimating unit ii regards the modulation signal related to the preamble, the known data of the preamble (the synchronization pattern X of the short preamble s P or the long preamble LP synchronization pattern:) as the pilot symbol Predict the phase rotation and amplitude error of the known data (the impulse response of each subcarrier). The subsequent modulation signal 'has a sign at every predetermined time, so the phase rotation and the amplitude error are estimated from the known data. The presumed phase rotation and amplitude error can be regarded as valid before the next-indicator symbol is compensated, and the compensation amount of the next index symbol and the compensation amount of the existing index symbol can be attached as appropriate to the lower-reference. The compensation value of the symbol. In addition, the index symbol can also regard the complex amplitude of all subcarriers as known data, but even if the data is known to be embedded in the subcarriers of the frequency domain having a large frequency selectivity. At this time, the propagation characteristics of the subcarriers to which the known data is not embedded may be derived from the propagation characteristics of the subcarrier 201014291 in which the known data is embedded. Etc: Department! 42 is the inverse of the complex impulse response of the preamble 旒' multiplied by the presump 丄4 double 疚. Thereby, the frequency domain of the pulse wave of the equalization c subcarrier is skewed, and 2 sub-commissioners are compensated. In addition, the amplitude error can be compensated for by rotating the transfer surface with a large amplitude variation. In this case, the estimating unit 141 and the first phase compensating party that compensates for the phase of the symbol are used: the phase compensation using the unindicated symbol is used in accordance with the work, so that even in the second effect of the frequency selectivity Very big. In the present embodiment, as shown in the fourth figure, four of the subcarriers are the subcarriers p to C4 which are independent of the transmission of the eleven images, and the remaining forty eight are the carrier SC〇. SC47. The symbols on the subcarrier cores 1 to PSC4 are known as known data (known symbols).

相位誤差去除部143使用四條引示副载波P 位補ms去二據每個0fdm符號進行相位誤差去除部143根據引示副載波？已知符號，檢測出各引示副载波 =員率誤差。相位誤差去除部143使用檢測出的 :率:疾差’根據同一〇 FDM符號，算出經由離散利葉轉換之各複數符號的相位誤差。接著，相位 19 201014291 誤差去除部1 4 3將算出之相位誤差的逆相位乘在各複數符號。藉此’相位誤差絲部丄4 3將進行因頻率誤差造成符號的相位旋轉補償。如此，相位誤差去除部143使用引示副載波，進行補償符號相位的第二相位補償方式。調變方式判別電路16根據發信機2〇接收的調變訊號’判別調變訊號所用的多層次調變方式。本實施、下，根據發信機£ Q接收之調變訊號的數據部D所含的調變方式資訊，決策每個〇聊· 符號的（從同一〇F η Μ炫„备 υ t U Μ符唬經由離散傅利葉轉換的各複數符號的）多層次調變方式是16QAM或 6 4 QAM任-者。補償方式選擇電路丄7根據調變方式判別電路工6所判別之多層次調變方式的多值度，從多個預制於觀訊號之符號相位補償的相位補償方式中選擇。本實施形態下，補償方式選擇電路1 7在調變方式判別電路丄6判別結果是i 6QAM時，選擇第— > 日你_ 圮伴弟相位補償方式；判別結果是 6 4QAM時，選擇第二相位補償方式。相位補该電路1 4使用由補償方式選擇電路丄 7選取之前述相位補償方式’補償符號的相位。決策電路15根據多層次調變方式判別電路丄 6所判別的多層次調變方式而業已由相位補償電路 1 4補償相位的符號，決策數據的位元列。進行更詳盡的說明：決策電路Η根據多層次調變方式判 20 201014291 別電路1 6判別的多層次調變方式，利用解映射，將已由相位補償電路14補償相位的各複數符號轉換成軟決策值。藉此’決策電路丄5從發信機2 〇將接收的數據之位元列輸出至受信機工〇内或受信機10外未以圖示的數據處理電路。每個OF DM符號的相位誤差角度0 2如前述所言為 2.88。。表 2 在 qpsk、16QAM，The phase error removing unit 143 performs the phase error removing unit 143 based on the reference subcarriers by using four derivation subcarriers P bits to complement ms to denoise each 0fdm symbol. The symbol is known, and each of the pilot subcarriers = the rate error is detected. The phase error removing unit 143 calculates the phase error of each complex symbol converted via the discrete Fourier transform using the detected rate: the disease difference based on the same 〇 FDM symbol. Next, the phase 19 201014291 error removing unit 1 4 3 multiplies the inverse phase of the calculated phase error by each complex symbol. Thereby, the phase error wire portion 3 4 3 performs phase rotation compensation of the symbol due to the frequency error. In this manner, the phase error removing unit 143 performs a second phase compensation method for compensating the phase of the symbol by using the subcarrier. The modulation mode discriminating circuit 16 discriminates the multi-level modulation method used for the modulation signal based on the modulation signal received by the transmitter 2A. In this implementation, according to the modulation method information contained in the data portion D of the modulation signal received by the transmitter Q Q, each slogan symbol is determined (from the same 〇F η Μ „ υ υ υ υ U U The multi-level modulation method of the complex symbols via the discrete Fourier transform is 16QAM or 6 4 QAM. The compensation mode selection circuit 丄7 discriminates the multi-level modulation method discriminated by the circuit operator 6 according to the modulation method. The multi-valued value is selected from a plurality of phase compensation methods pre-fabricated in the symbol phase compensation of the signal. In the present embodiment, the compensation mode selection circuit 17 selects when the modulation mode determination circuit 丄6 determines that the result is i 6QAM. The first -> _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ The phase of the symbol. The decision circuit 15 discriminates the sign of the phase by the phase compensation circuit 14 according to the multi-level modulation mode discriminating the multi-level modulation mode determined by the circuit 丄6, and determines the bit sequence of the data. More detailed description: The decision circuit 判 is based on the multi-level modulation method to determine the multi-level modulation mode of the circuit of the 1614 14291 circuit, and uses the demapping to convert the complex symbols that have been compensated by the phase compensation circuit 14 into soft decisions. The value is output by the 'decision circuit 丄5 from the transmitter 2 位 to the bit row of the received data to the data processing circuit in the receiver or outside the receiver 10. The phase of each OF DM symbol The error angle 0 2 is 2.88 as described above. Table 2 is in qpsk, 16QAM,

6 4QAM等各多層次調變方式下，顯示θ 2的值。且表2顯示在復調處理途中，將引示符號嵌入調變訊號的最小符號間隔M ( Θ丄/θ 2以下為最大正整数），以避免超過容許誤差角度0 ;[。再者’表2顯示在以最小符號間隔μ將引示符號嵌入调變訊號的情況，傳送效率ρ丄=Μ/ (Μ +丄）。6 In the multi-level modulation mode such as 4QAM, the value of θ 2 is displayed. And Table 2 shows that during the polyphonic processing, the minimum symbol interval M (where Θ丄/θ 2 is the largest positive integer) is embedded in the modulation signal to avoid exceeding the tolerance angle 0; Further, Table 2 shows a case where the pilot symbol is embedded in the modulation signal at the minimum symbol interval μ, and the transmission efficiency ρ 丄 = Μ / (Μ + 丄).

【表2【Table 2

如第六圖所示’各多層次調變方式中，係在每個Μ符號將引示符號p s嵌入調變訊號。藉此，能避免復調處理途中，相位誤差超過容許誤差角度^ 1 ’且傳送效率將變最高。使用引示符號的第一相位補償方式中，補償間隔將變得比使用引示副載波者還長。這是因為等化 4 1 4 2的4化參數更新時機為每個μ符號。因 21 201014291 此，高多值度的多層次調變方式為64QAM的情況下（最小符號間隔M = 3 )，引示符號之間隔為^ 時，復調處理途中，相位誤差恐將超過容許誤差度 Θ 1。 'As shown in the sixth figure, in each multi-level modulation method, the modulating symbol p s is embedded in the modulation signal in each Μ symbol. Thereby, it is possible to avoid the phase error exceeding the allowable error angle ^ 1 ' during the polyphonic processing and the transmission efficiency will be the highest. In the first phase compensation method using the pilot symbols, the compensation interval will become longer than those using the pilot subcarriers. This is because the 4th parameter update timing of the equalization 4 1 4 2 is for each μ symbol. Because 21 201014291 Therefore, when the multi-level modulation method with high multi-value is 64QAM (minimum symbol interval M = 3), when the interval of the sign is ^, the phase error will exceed the tolerance when the polyphonic processing is in progress. Θ 1. '

另一方面’使用引示副載波PSC1〜Psc 4的第二相位補償方式是每個〇F D M符號逐次補償。因此，每個〇 F D Μ符號可（針對從同一〇F DM符號進行離散傅利葉轉換的各複數符號）進行最佳相位補償。因此，即使是高多值度的多層次調變方式，亦可在進行復調處理途中避免相位誤差超過容許誤差角度Θ 1。第一相位補償方式中’必須在多值度越大，越將嵌入調變訊號之引示符號的間隔縮小。因此，傳送效率比較容易變低。如表2所示，傳送效率p 1 係在1 6 Q AM下Ρ丄=〇. 8 3 ;在6 4 Q AM下On the other hand, the second phase compensation method using the pilot subcarriers PSC1 to Psc 4 is that each 〇F D M symbol is successively compensated. Therefore, each 〇 F D Μ symbol can be optimally compensated for each complex symbol that performs discrete Fourier transform from the same 〇F DM symbol. Therefore, even in the case of a multi-level modulation method of high multi-value, it is possible to prevent the phase error from exceeding the allowable error angle Θ 1 during the polyphonic processing. In the first phase compensation method, the larger the multi-value degree must be, the smaller the interval between the pilot symbols embedded in the modulation signal is reduced. Therefore, the transmission efficiency is relatively easy to become low. As shown in Table 2, the transmission efficiency p 1 is 1=〇. 8 3 at 16 Q AM; at 6 4 Q AM

ρ1 = 〇·75。另一方面，第二相位補償方式中，係將所有五十二條副載波中的四條用於引示副載波 P S C 1〜p s C 4。因此，使用引示副載波時的傳送效率ρ 2是〇·9 2 (=四十八條/五十二條）°因此’第二相位補償方式比第一相位補償方式更易實現而傳送效率。如以上所言’根據本實施形態下的受信機1 0 ’將根據發信機2 〇所用之多層次調變方式的多值度’選擇相位補償方式。因此，能端視發信機2 22 201014291 0所用的多層次調變方式，進行最佳相位補償。尤以發信機2 0所用之多層次調變方式的多值度在小於預定值（對應6 4 QAM的多值度）的情况下，即使在頻率選擇性強的傳播環境，亦可使用相位補償效果大的引示符號，進行相位補償。另一方面，發信機2 0所用的多層次調變方式之多值度在預定值（對應6 4 QAM的多值度）以上的情二下’將使用容易達到高傳送效率的引示副載波，進行相位補償。因此’能端視發信機2 0所用的多層次調變方式，進行最佳相位補償。、唯，相位補償方式選擇電路丄7亦可由以下構成。、意即相位補償方式選擇電路1 7在多層次調變方式判別電路1 6所判別的多層次調變方式之多值度小於财值（對應6 4QAM的多值度）的話，則只會選擇第一相位補償方式；判別的多值度在預Ρ1 = 〇·75. On the other hand, in the second phase compensation method, four of all fifty-two subcarriers are used to derive subcarriers P S C 1 to p s C 4 . Therefore, the transmission efficiency ρ 2 when the subcarriers are used is 〇·9 2 (= forty-eight/fifty-two). Therefore, the second phase compensation method is easier to implement than the first phase compensation method and the transmission efficiency is improved. As described above, the receiver 1 0 ' in the present embodiment selects the phase compensation method based on the multi-valued degree of the multi-level modulation method used by the transmitter 2 。. Therefore, the optimal phase compensation can be performed by looking at the multi-level modulation method used by the transmitter 2 22 201014291 0. In particular, in the case where the multi-value modulation method of the multi-level modulation method used by the transmitter 20 is smaller than a predetermined value (corresponding to the multi-value degree of 6 4 QAM), the phase can be used even in a propagation environment with high frequency selectivity. The compensation symbol with large compensation effect is used for phase compensation. On the other hand, the multi-level modulation method used by the transmitter 20 is above the predetermined value (corresponding to the multi-value degree of 6 4 QAM), and will be used to introduce the high transmission efficiency. Carrier, phase compensation. Therefore, the optimum phase compensation can be performed by looking at the multi-level modulation method used by the transmitter 20. The phase compensation mode selection circuit 丄7 can also be constructed as follows. In other words, the multi-level modulation method determined by the multi-level modulation method discriminating circuit 16 is less than the financial value (corresponding to the multi-value degree of 6 4QAM), and only the selection is made. First phase compensation mode; the multi-valued degree of discrimination is in advance

對應“QAM的多值度）以上的話，則選擇第-相位補償方式及第二相位補償方式兩者。匕夺將經常使用引示符號進行相位補償。因 j ’相位補償電路工4的控制將變簡單。再者，第補秘方式將依每個副載波執行相位補償。因，即使在頻率選擇性強的傳播環加大相位補償的效果。疋了灶吊以-4 1將由前置碼的已知符號及疋間隔肷入調變訊號的引示符號，推定傳送路 23 201014291Corresponding to "multiple value of QAM" or more, both the first phase compensation method and the second phase compensation method are selected. The capture will often use the pilot symbol for phase compensation. The control of j 'phase compensation circuit 4 will be In addition, the first method of compensating will perform phase compensation according to each subcarrier. Therefore, even in the frequency selective strong propagation loop, the effect of phase compensation is increased. 疋 The stove hoist is -4 1 will be the preamble Known symbols and 疋 intervals into the sign of the modulation signal, presumed transmission path 23 201014291

:以’恐將超過副载波複數振幅的一次調變符號之容許誤差角度01。波推定傳送路徑3 0的脈衝響應。〇 F dm符號更新傳送路徑3 〇 , 而，有關引示副載波以外的副载波另一方面，相位誤差去除部丄43以引示副載❹ 。因此，能依每個的脈衝響應。然支（意即用以傳送值包含誤差。然而，一數據的副載波）之脈衝響應的推定值，可根據來自引示副載波之脈衝響應，利用外插法或内插法算出。因此，用以傳送數據之副載波的脈衝響應推定 _ ”、、、而，一旦只有相位誤差去除部1 4 3更新脈衝響應’等化部1 4 2由更新後的脈衝響應所計算的逆特性進行等化，故每次脈衝響應更新都會累積誤差。因此，只要過了某個時間，誤差累積量超過容許量（容許誤差各Θ i )，相位補償的效果就會消失。尤其多值度越大的多層次調變方式，容許誤差角度β 1越小，因此，相位補償效果消失的可能性極高。 24 201014291 因此，多yf次調變方式判別多層次調變方柄多值度在 6所判別之 Μ的多值度）以值（對應6 4 QΑ 7同時選取第一相你心士割貝方式選擇電路1 .,, 相位補彳員方式與第二相位補償方 :)’相位補償電路i 4最好由以。： :，相位補償電路工4遵照第二相位補償；: The allowable error angle 01 of the first modulation symbol that exceeds the complex amplitude of the subcarrier. The wave estimates the impulse response of the transmission path 30. 〇 F dm symbol update transmission path 3 〇 , and subcarriers other than the reference subcarriers On the other hand, the phase error removal unit 丄 43 is used to introduce subcarriers. Therefore, it is possible to respond to each pulse. The estimated value of the impulse response (that is, the transmitted value contains the error. However, the subcarrier of a data) can be calculated by extrapolation or interpolation based on the impulse response from the pilot subcarrier. Therefore, the impulse response of the subcarrier for transmitting data is estimated to be _", and, if only the phase error removing portion 1 4 3 updates the impulse response, the inverse characteristic calculated by the updated impulse response is equalized by the equalizing portion 1 4 2 Equalization is performed, so the error is accumulated every time the impulse response is updated. Therefore, as long as the error accumulation exceeds the allowable amount (the tolerance is Θ i ) after a certain time, the effect of phase compensation disappears. Large multi-level modulation method, the smaller the allowable error angle β 1 , therefore, the possibility of phase compensation effect disappearing is extremely high. 24 201014291 Therefore, the multi-yf submodulation method discriminates the multi-level modulation square handle multi-valued degree in 6 The multi-valued degree of the determined ) is based on the value (corresponding to 6 4 Q Α 7 simultaneously selects the first phase of your heart-cutting mode selection circuit 1 ,, phase complement mode and second phase compensation side:) 'phase compensation The circuit i 4 is preferably composed of :: :, the phase compensation circuit 4 follows the second phase compensation;

符號相位後，再遵昭第一彳 iM 位。理…弟才目位補償方式補償符號相After the symbol phase, follow the first iM bit. The younger brother’s position compensation method compensates for the symbol phase

這樣來就月b使用引示副載波，依每個〇ρ DM符號進行相位補償。且因使用引示副載波進行相位補償所產生的誤差，能藉由使用引示符號進行的相位補償而消除。因此’能針對多值度大的次調變方式，進行最佳相位補償。另外，前述範例將相位補償方式選擇電路工7 預疋值視為對應6 4 Q A Μ的多值度。然而，亦可根據相位補償電路i 4使用相位補償方式選擇電路 17所選取的相位補償方式進行符號相位補償時的傳送效率，來設定預定值。例如’考慮發信機20具有QPSK、16Q AM、6 4QAM等多層次調變方式的情況。此時，多層次調變方式判別電路1 6係根據接收自發信機2 0之調變訊號的數據部D内含之調變方式的資訊，判別用於調變訊號的多層次調變方式是QPSK、1 6QAM、64QAM任一者。 25 201014291 在第一相位補償方式下，則In this way, the subcarriers are used for the month b, and the phase compensation is performed for each 〇ρ DM symbol. Moreover, the error caused by phase compensation using the pilot subcarrier can be eliminated by phase compensation using the pilot symbol. Therefore, the best phase compensation can be performed for a multi-valued submodulation method. In addition, the foregoing example considers the phase compensation mode selection circuit 7 to be a multi-valued value corresponding to 6 4 Q A Μ. However, the phase compensation circuit i 4 can also set the predetermined value by performing the transmission efficiency at the symbol phase compensation using the phase compensation method selected by the phase compensation mode selection circuit 17. For example, it is considered that the transmitter 20 has a multi-level modulation method such as QPSK, 16Q AM, or 6 4QAM. At this time, the multi-level modulation method discriminating circuit 16 determines that the multi-level modulation method for the modulation signal is based on the information of the modulation method included in the data portion D of the modulation signal received from the transmitter 20. Any of QPSK, 1 6QAM, and 64QAM. 25 201014291 In the first phase compensation mode,

^6〇Q u. 7 5 (參照表 2 )。 υ· ”=方9面2，在第二相位補償方式下，傳送效率此時，相位補償方式選擇率”比傳送效率Ρ2( = 0擇電9路2 :針對傳送效變方式，選擇0.92)兩的多層次調 m 位補償方雷1的相位補償。意即，相 7保方切擇電路17在傳送效率P1 7的QPSK情況 .9 — ^選擇第一相位補償方式。另^6〇Q u. 7 5 (refer to Table 2). υ· ”=方9面2, in the second phase compensation mode, the transmission efficiency at this time, the phase compensation mode selection rate” is higher than the transmission efficiency Ρ2 (= 0 selects the power 9 way 2: for the transmission effect mode, selects 0.92) The multi-level adjustment of the two bits compensates the phase compensation of the square 1 . That is, the phase 7 check circuit 17 selects the first phase compensation mode in the QPSK case of the transmission efficiency P1 7 . another

方面’針對傳送效率Pltb傳送效 J 變方式&擇使用引示副載波的相位 =’相位補償方式選擇電路17在傳送效率ρι =·8 3的1 6QAM情況下，選擇第二相位補償 ▲方式。同様的，相位補償方式選擇電路” 效率ρ 1是0.75的64QAM情況下，選擇第_Aspect 'Transmission efficiency Pltb transmission efficiency J variable mode & use phase of the reference subcarrier = 'phase compensation mode selection circuit 17 in the case of transmission efficiency ρι = · 8 3 of 16 6AM, select the second phase compensation ▲ mode . The same, the phase compensation mode selection circuit" efficiency ρ 1 is 0.75 in the case of 64QAM, select the first _

相位補償方式。逻释第一總之，相位補償方式選擇電路i 7在發俨機2 〇所用之多層次調變方式（QPSK或i 6qctam =64QAM)的多值度比根據傳送效率p2的預 =值還低的情況下，使用引示副載波，進行相位補償。另一方面，相位補償方式選擇電路丄7在發信機2 0所用之多層次調變方式的多值度在根據效率P 2的預定值以上的情況下，使用引示符號， 26 201014291 進行相位補償。、此時，能端視發信機2 〇所用的多層次調變方式’進行最佳相位補償，且提高傳送效率。 (第二實施形態）本實施形態下的受信機4 〇被用於單載波通訊糸統。單載波通訊系統所用的發信機2 Q將發送至受 Φ 信機4 0的數據進行錯誤更正編碼。且發信機2 ^ 根據多層次調變方式決定的符號與位元列的對應關係，從錯誤更正編碼後的數據形成複數符號（符號對應）。發信機2 〇在複數符號進行適當波形成形處u 理，進行DA轉換後，利用去除Da轉換形成影像訊號的濾波器過濾的符號列而生成的基頻訊號乘上載波，藉此進行頻率轉換，往所需頻率帶遷移後，進行預定的訊號增幅’生成調❸皮。發信機2 〇將 ❿ 形成的調變波發送至傳送路徑3〇。這裡，發信機20具有多值度不同的多個多層次調變方式，如QPSK、16QAM。發信機2 0在進行4號對應之際，從多值度不同的多個多層 -人凋變方式中，端視傳送路徑3 〇的狀態，選擇傳 $速度變最快的多層次調變方式（意即，發信機2 〇進行可調適調變）。且’發#機2 0備有具石英振盪器的基準訊號來源（未以圖示）。發信機2 〇使用基準訊號來源所 27 201014291 發的基準頻率，進行前述頻率轉換。此外，受信機 4 0亦具備基準訊號來源。發信機2 0係發送視線路品質而從多個多層次凋變方式（QPSK、64QAM)選取的多層次 1 周變方式所生成的調變波。調變波具有顯示發送至文尨機4 0之數據的符號列。該符號根據發信機2 〇所選取的多層次調變方式’決定與位元列的對應關係。 ^ 本實施形態下的受信機4 〇如第七圖所示，備有A/D轉換電路4 1、F I R濾、波H4 2、縮減取樣電路4 3、相位補償電路4 4、決策電路4 5、多層次調變方式判別電路（多層次調變方式判別部） 4 6、相位補償方式選擇電路（相位補償方式選擇部）47。圖中將類比部的訊號增幅、去除干涉波濾波器等類比訊號處理電路予以省略。 A/D轉換電路4工以受信機‘〇的基準訊號來源（未以圖示）發出的基準頻率形成載波。A〆 D轉換電路4 1藉由透過傳送路徑3 〇接收的調變訊號乘上前述載波’將調變訊號下轉換後，形成基頻訊號。A/D轉換電路4 1將基頻訊號進行類比 /數位轉換後，輸出至F Z R濾波器4 2。縮減取樣電路4 3、M P T u i d將透過F I R濾波器4 2取得的基頻㈣進行料取樣。、_取樣電路4 3將已縮減取樣的基頻訊號輪出至相位補償電路4 4。 201014291 相位補償電路4 4 4 #有相位誤差去除部4 4 】，442、相位推定部⑷ ::4!擇性選用使用再調變的相位補償方= 一：付旒的相位補償方式等兩種相位補償方式’猎由頻率誤差進行相位旋轉補償。調變的相位補償方式是藉由相位誤差去 = 44!、調變器4 4 2、相位衫部443等 ^仃。使料調變進行相位補償方至決策雷路4 3的輸出直接發送決策L 調變器442係將決策電路45 行以藉2列轉換成複數平面上的1 Q訊號後，進付號化的再調變。相位推定部4 4 3 =:4 4 2輸出之再調變訊號與縮減取樣電路4 差輸出的積。相位推定部443藉此算出相位誤相位疾差去除部4 4 1將相位推定部4 4 出之相位誤差的逆相位（號。藉此，相位乘在各複數符純誤差去除部4 4 1就能補償頻率誤二致的相位旋轉。意即，相位補償電路“使用期被執行。冑用再調變的相位補償將定去除：=符號的相位補償方式是藉由相位誤差位的p 土狀。1進仃。迫裡的引示符號是具有已知相變的引：：：’並非如第—實施形態之經多載波調 — 使用此已知符號的相位補償方 29 201014291 式與實施形態相同，故省略其說明。 =次調變方式判別電路46根據接收自發信機2 0之調變訊號内含之調個封包的多層次調變方式是QPSK;;Phase compensation method. In the first embodiment, the multi-level modulation method (QPSK or i 6qctam = 64QAM) used by the phase compensation mode selection circuit i 7 in the hair dryer 2 is lower than the pre-value according to the transmission efficiency p2. In this case, phase compensation is performed using the pilot subcarrier. On the other hand, the phase compensation mode selection circuit 丄7 performs the phase using the pilot symbol, 26 201014291, in the case where the multi-value modulation of the multi-level modulation method used by the transmitter 20 is equal to or higher than the predetermined value of the efficiency P 2 . make up. At this time, it is possible to perform optimum phase compensation by looking at the multi-level modulation method used by the transmitter 2, and to improve the transmission efficiency. (Second Embodiment) The receiver 4 in the present embodiment is used for a single carrier communication system. The transmitter 2 Q used by the single carrier communication system transmits data to the Φ receiver 40 for error correction coding. And the transmitter 2 ^ forms a complex symbol (symbol correspondence) from the error corrected coding data according to the correspondence between the symbol determined by the multi-level modulation method and the bit column. The transmitter 2 进行 performs the appropriate waveform shaping at the complex symbol, and after DA conversion, the baseband signal generated by removing the symbol sequence filtered by the filter for forming the image signal by Da is multiplied by the carrier, thereby performing frequency conversion. After the migration to the desired frequency band, a predetermined signal increase is performed to generate a tune. The transmitter 2 transmits the modulated wave formed by ❿ to the transmission path 3〇. Here, the transmitter 20 has a plurality of multi-level modulation methods having different degrees of multi-value, such as QPSK and 16QAM. When the transmitter 20 performs the No. 4 correspondence, from the multi-layer-person fading mode with different ambiguities, the state of the transmission path 3 〇 is selected, and the multi-level modulation with the fastest transmission speed is selected. The way (ie, the transmitter 2 可调 can be adjusted). The 'Source#2' has a reference signal source with a quartz oscillator (not shown). Transmitter 2 进行 Use the reference frequency from the source of the source signal 27 201014291 to perform the above frequency conversion. In addition, the receiver 40 also has a reference signal source. The transmitter 20 transmits a modulated wave generated by a multi-level one-cycle variation method selected from a plurality of multi-level fade modes (QPSK, 64QAM). The modulated wave has a symbol column that displays the data sent to the file machine 40. The symbol determines the correspondence with the bit column according to the multi-level modulation method selected by the transmitter 2 。. The receiver 4 of the present embodiment, as shown in the seventh figure, is provided with an A/D conversion circuit 4 1 , an FIR filter, a wave H4 2 , a downsampling circuit 4 3 , a phase compensation circuit 4 4 , and a decision circuit 4 5 . Multi-level modulation method discrimination circuit (multi-level modulation method determination unit) 4 6. Phase compensation method selection circuit (phase compensation method selection unit) 47. In the figure, the analog signal processing circuit such as the signal amplification of the analog section and the removal of the interference wave filter are omitted. The A/D conversion circuit 4 forms a carrier with a reference frequency from the source of the reference signal (not shown) of the receiver. The A 〆 D conversion circuit 4 1 multiplies the modulated signal by multiplying the modulated signal received through the transmission path 3 ’ to convert the modulated signal to form a fundamental frequency signal. The A/D conversion circuit 4 1 performs analog/digital conversion on the fundamental frequency signal, and outputs it to the F Z R filter 42. The downsampling circuit 4 3, M P T u i d will sample the material through the fundamental frequency (4) obtained by the F I R filter 42. The _sampling circuit 43 rotates the downsampled baseband signal to the phase compensation circuit 44. 201014291 Phase compensation circuit 4 4 4 #With phase error removal unit 4 4 】, 442, phase estimation unit (4) ::4! Selectively use the phase compensation side using remodulation = one: the phase compensation method of the 旒等The phase compensation method 'hunting compensates for phase rotation by frequency error. The phase compensation method of modulation is by phase error = 44!, modulator 4 4 2, phase shirt 443, etc. ^仃. The material is modulated to phase compensation to the output of the decision channel 4 3. The direct transmission decision L 442 is to convert the decision circuit 45 row into 2 Q signals on the complex plane by borrowing 2 columns, and then paying the number. Change again. The phase estimation unit 4 4 3 =: 4 4 2 The product of the output of the remodulation signal and the difference output of the downsampling circuit 4. The phase estimating unit 443 calculates the inverse phase (number) of the phase error generated by the phase error phase difference removing unit 4 4 1 by the phase estimating unit 4 4 , whereby the phase is multiplied by each complex pure error removing unit 4 4 1 It can compensate the phase rotation of the frequency error. That is, the phase compensation circuit "use period is executed. The phase compensation with re-modulation will be removed: the phase compensation mode of the symbol is the phase of the phase error bit. The first sign is a reference with a known phase change:::' is not a multi-carrier modulation as in the first embodiment - the phase compensator using this known symbol 29 201014291 The description of the same is omitted. The multi-level modulation method of the sub-modulation mode discriminating circuit 46 according to the modulated packet contained in the modulated signal received from the transmitter 20 is QPSK;

的任一者。 a次：L 6 QAM 式判=償方式選擇電路47根據多層次調變方 =i電路46判別之多層次調變方式的多值度，、擇亡目位補償電路4 4執行的相位補償方式。廷裡若為1 6 QAM，則因複數平面上的間距較紐’故各符號點的容許誤差角度6> 1比q : sk小。因此，多層次調變方式若為16^撾，一旦使用與前述QPSK相同的相位補償方式復調後的位it列恐有許多錯誤’再調變後號未必正確。付因此，相位補償方式選擇電路47在多層次調變方式判別電路4 6的判別結果是q p s κ的情况下’選擇使用再調變的相位補償方式。相位補償方G 式選擇電路4 7在多層次調變方式判別電路4 6的判別結果是1 6 Q AM的情況下，選擇使用引示符號的相位補償方式。 ^ 決策電路4 5根據以多層次調變方式判別電路 4 6所判別的多層次調變方式，判定對應於相位補償電路4 4已補償相位後之符號的位元列。進行更詳盡的說明，決策電路4 5根據多層次調變方式判 30 201014291 別電路4 6判別的多層次調變方式，利用解映射將以相:立補償電路4 4補償相位的各複數符號轉換成軟決策值決策電路4 5藉此將接收自發信機2 〇之數據的位it列輸出至受信機4 Q内或受信機4 〇外未以圖示的數據處理電路。 ^如别述所言，本實施形態下的受信機4 0在發 k機2 0所用之多層次調變方式（QpsK或工6 • ?AM)的多值度小於預定值（對應的多值度）的情況下，使用再調變進行相位補償。且党信機4 0在發信機2 〇所用之多層次調變方式的多值度在預定值（對應1 6 QAM的多值度）以上的情況下，使用引示符號進行相位補償。如此，只要藉由受信機4 〇，即可根據發信機 ^◦所用之多層次調變方式的多值度，選擇相位補犒方式因此，就能端視發信機2 0所用的多層次 | 凋變方式，進行最佳相位補償。以上所述僅為本發明之較佳可行實施例，非因此侷限本發明之專利保護範圍，故舉凡運用本發明說明書及圖式内容所為之等效技術變化，均心於本發明之權利保護範圍内，合予陳明。【圖式簡單說明】第一圖是第一實施形態之受信機的概略圖。第二圖是具備同上受信機的通訊系統概略圖。 31 \ 201014291 第三圖是1 6 Q A M之調變號分佈星象圖的說明圖。第四圖是副載波與引示副載波的配置說明圖。第五圖是OFDM訊號結構的說明圖。第六圖是引示符號嵌入結構的說明圖。第七圖是第二實施形態之受信機的概略圖。【主要元件符號說明】 10 受信機 11 自動頻率補償電路 12 保護區間去除電路 13 快速傅利葉轉換電路 14 相位補償電路 15 決策電路 16 多層次調變方式判別電路 17 相位補償方式選擇電路 2 0 發信機 3 0 傳送路徑 4 0 受信機 4 1 A/D轉換電路 4 2 FIR濾波器 4 3 縮減取樣電路 4 4 相位補償電路 4 5 決策電路 4 6 多層次調變方式判別電路Any of them. a times: L 6 QAM type judgment = compensation mode selection circuit 47 according to the multi-level modulation method = multi-level modulation method of multi-level modulation method, the phase compensation method executed by the selection target position compensation circuit 44 . If Tingri is 1 6 QAM, the tolerance angle of each symbol point is smaller than q: sk because the spacing on the complex plane is relatively new. Therefore, if the multi-level modulation method is 16^, once the bit alignment column after the same phase compensation method as the QPSK is used, there may be many errors. The re-modulation number may not be correct. Therefore, the phase compensation mode selection circuit 47 selects the phase modulation method using the remodulation when the discrimination result of the multi-level modulation mode determination circuit 46 is q p s κ . The phase compensation side G type selection circuit 47 selects the phase compensation method using the index symbol when the discrimination result of the multi-level modulation method discrimination circuit 46 is 1 6 Q AM. The decision circuit 45 determines the bit sequence corresponding to the sign after the phase compensation circuit 44 has compensated the phase, based on the multi-level modulation method determined by the multi-level modulation method discrimination circuit 46. For a more detailed description, the decision circuit 45 judges the multi-level modulation method determined by the multi-level modulation method according to the multi-level modulation method, and uses the demapping to convert the complex symbol conversions of the phase compensation phase with the phase compensation circuit 44. The soft decision value decision circuit 45 thereby outputs the bit it column of the data received from the transmitter 2 to the data processing circuit not shown in the receiver 4 Q or the receiver 4 . As described above, the multi-level modulation method (QpsK or gong 6 • ?AM) used by the receiver 40 in the present embodiment is smaller than a predetermined value (corresponding multi-value). In the case of degree), phase compensation is performed using remodulation. Further, when the multi-level modulation method of the multi-level modulation method used by the party machine 40 is equal to or greater than a predetermined value (corresponding to the multi-value degree of 1 6 QAM), the phase compensation is performed using the pilot symbols. In this way, by means of the receiver 4, the phase compensation mode can be selected according to the multi-value modulation method of the multi-level modulation method used by the transmitter, so that the multi-level used by the transmitter 20 can be viewed. | Withstand mode for optimal phase compensation. The above description is only a preferred embodiment of the present invention, and is not intended to limit the scope of the present invention. Therefore, the equivalent technical changes of the present invention and the contents of the drawings are within the scope of the present invention. Within, combined with Chen Ming. BRIEF DESCRIPTION OF THE DRAWINGS The first figure is a schematic view of a receiver of the first embodiment. The second picture is an overview of the communication system with the same receiver. 31 \ 201014291 The third figure is an illustration of the astrological map of the 1 0 Q A M modulation number distribution. The fourth figure is a configuration explanatory diagram of subcarriers and pilot subcarriers. The fifth figure is an explanatory diagram of the structure of the OFDM signal. The sixth figure is an explanatory diagram of the symbol embedding structure. Fig. 7 is a schematic view showing a receiver of the second embodiment. [Main component symbol description] 10 Receiver 11 Automatic frequency compensation circuit 12 Protection interval removal circuit 13 Fast Fourier conversion circuit 14 Phase compensation circuit 15 Decision circuit 16 Multi-level modulation mode discrimination circuit 17 Phase compensation mode selection circuit 2 0 Transmitter 3 0 transmission path 4 0 receiver 4 1 A/D conversion circuit 4 2 FIR filter 4 3 downsampling circuit 4 4 phase compensation circuit 4 5 decision circuit 4 6 multi-level modulation mode discrimination circuit

32 201014291 47 相位補償方式選擇電路 141 推定部 142 等化部 143 相位誤差去除部 441 相位誤差去除部 442 調變器 443 相位推定部32 201014291 47 Phase compensation method selection circuit 141 Estimation unit 142 Isocratation unit 143 Phase error removal unit 441 Phase error removal unit 442 Modulator 443 Phase estimation unit

3333

Claims

201014291 七、申請專利範圍： n夺统去種又仏冑係與發信機—起用於可調適調變通讯糸統者，該發信機係用號係使用以預定基準從多數：二調變訊號，該調變訊多層次調變方式所生成者，其固中夕層二人調變方式選擇的述受=號具有符號列’其係用以顯示發送至前 e 刖述符號係根據前述發作施方式丄衫與位元列的對應關係；、擇的多層次調變方信機備有多層次調變方式判別部、相位補償方式f擇部、相位補償部、及決策部；前述::二::::式判別部的構成是，根據接收自前述多層二=變訊號，判別前述調變訊號所用的 -欠位補償方式選擇部的構成是，根據前述多層值部所判別的前述多層次調變方式的多值度〇數個預備的相位補償方式中，選擇二調變!1號之符號相位補償的相位補償方式；、“ 選擇陶成是’使用前述相位補償方式位；擇^述相位補償方式，補償前述符號的相部補償相位後之前::::位:;對應於已由相位補償 34 201014291 2 '如申請專利範圍第1項所述之受信機，其中，前述發信機具有一次調變方式與二次調變方式；剛述一次調變方式係從多值度不同的多數個多層次調變方式中以預定基準選擇的多層次調變方式，生成顯示前述符號的一次調變符號；述二次調變方式係多載波調變方式，根據前述一次調變符號，而將構成複數振幅的多個副载波重疊，生201014291 VII. The scope of application for patents: n. The system is used for adapting and adapting the communication system. The transmitter is used by the number system to determine the basis from the majority: the second modulation. Signal, the generator of the multi-level modulation method of the modulation, the selection of the solid-mode two-person modulation mode is controlled by the symbol number column, which is used to display the transmission to the former e-description symbol according to the foregoing Corresponding relationship between the application mode and the bit column; the multi-level modulation machine has a multi-level modulation method, a phase compensation method, a phase compensation unit, and a decision-making unit; The two-:::-type discriminating unit is configured to determine the offset-compensation method selection unit for determining the modulation signal based on the multi-layer two-signal signal received from the multi-valued unit. In the multi-level modulation method, the multi-valued method has a plurality of preparatory phase compensation modes, and the second phase change is selected: the phase compensation method of the symbol phase compensation of No. 1; "Selecting Tao Cheng is the use of the aforementioned phase compensation mode bit; ^Record The compensation method, after compensating for the phase compensation phase of the aforementioned symbol, before::::bit:; corresponds to the receiver that has been compensated by phase compensation 34 201014291 2 ', as described in claim 1, wherein the aforementioned transmitter has One-time modulation method and second-order modulation method; The first-time modulation method is a multi-level modulation method selected from a plurality of multi-level modulation methods with different multi-valued degrees, and a first-order modulation is displayed to display the aforementioned symbols. Variable symbol; the second modulation method is a multi-carrier modulation method, and a plurality of subcarriers constituting a complex amplitude are overlapped according to the first modulation symbol.

❹ 成一人調變捋號，構成由多數個前述二次調變符號所構成的前述調變訊號；引示符號； 5為已知的二次别述調變訊號在每預定時間具有則述引示符號係對前述受信機而調變符號；則述已知二次調變符號係副載波所構成；，、/、有已知複數振幅的 =述二次調變符號包含引示副載波；前述引示副载波係對前述受二載波；又乜機而δ為已知的副前述相位補償方式選擇部的振幅」，調變方式判別部所判別之筹成疋’在則述多層次多值度小於預定值的話，即選摆：：人調變方式的前述述符號之相位的第—相位補，述W示符號補償前到既定值以上的話，即選擇用=而在前述多值度達符號之相位的第二相位補償方式。丨不副載波補償前述 35 201014291 前述相位補利範圍第2項所述之受信機，其中，方式判别部所邦部Γ成是，在前述多層次謂變度小於預定值的話，僅變方式的前述多值在前述多值戶、素$，述第—相位補償方式，而補償方式與；述第二值以上的話’選擇前述第-相位月J迷第一相位補償方式兩者。 4、如申請專利範圍第3項所述刖述相位補償部又機，其中，選擇構成是，在前述相位補償方式選禮却選擇别述第一相位補償方式万式選擇部兩者的情況下，遵照前述第二相位補償方式號的相位後，再遵照前述第一相 =補J前述符號的相位。式補彳員前述符 5、如申請專利範圍第2項所則述預定值係根據前述相位補償部使^ 其中，方式選擇部所選擇的前述相位補 =述相位補償之相位補償之情況下的傳送效率而設行前述符❹ 一一调 , , , , , , , , 调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调调The symbol is a modulation symbol for the aforementioned receiver; the known second modulation symbol is composed of subcarriers; , /, having a known complex amplitude = the second modulation symbol includes a pilot subcarrier; The pilot subcarrier is paired with the second carrier; the amplitude is δ, and δ is the amplitude of the known phase compensation method selection unit. The modulation method determined by the modulation method determination unit is described in multiple levels. If the value is less than the predetermined value, that is, the pendulum: the first phase complement of the phase of the above-mentioned symbol of the human modulation mode, and the W sign is equal to or greater than the predetermined value before the symbol compensation, that is, the value of the multi-value is selected. The second phase compensation method of the phase of the symbol.丨副副副 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 2010 The multi-valued value is in the multi-valued household, the prime $, and the first-phase compensation method, and the compensation method and the second-order value or more 'select the first-phase first-phase first-phase compensation method. 4. The phase compensation unit is further described in the third paragraph of the patent application scope, wherein the selection configuration is such that, in the case where the phase compensation method is selected, the first phase compensation method is selected. After following the phase of the second phase compensation mode number, the phase of the first phase = complement J is followed. In the case of the second aspect of the patent application, the predetermined value is obtained by the phase compensation unit, wherein the phase selection unit selected by the mode selection unit compensates for the phase compensation of the phase compensation. Transmission efficiency