MXPA01006993A - Overload protection for a switch mode power supply - Google Patents

Abstract

When a main switching transistor (Q1) of a zero voltage switching power supply is conductive, a voltage (VR7) is developed in a current sensing resistor (R7) coupled in series with the transistor. The voltage in the current sensing resistor is coupled to a first input (base) of a comparator (Q3) of the control circuit (Q2, Q3). A second input (emitter) of the comparator is coupled to a capacitor (C6) that develops a voltage (VC6) that varies in accordance with an output voltage (IDOUT2) of the power supply. During a given conduction interval of the transistor, the comparator (Q3) is triggered in accordance with the difference between the current sensing resistor voltage and the capacitor voltage. An output of the comparator (Q3) is coupled to the base of the transistor for controlling the turn off instant of the transistor on a current pulse-by- current pulse basis. Under overload condition, when the transistor is turned off, a reverse collector current flows through the base-collector junction of the transistor. The reverse collector current produces a voltage (negative) in the current sensing resistor that is at opposite polarity with respect to the voltage there, during forward conduction of the transistor. The voltage produced by the sensed reverse current turns on a diode (D10) switch and charges the capacitor that is coupled to the comparator of the control circuit in a manner to reduce the peak forward collector current. Thereby, the peak forward current in the transistor decreases relative to a value that would have occurred without the protection.

Description

V 1 PROTECTION AGAINST OVERLOADS FOR AN ENERGY SUPPLY OF SWITCHING MODE FIELD OF THE INVENTION The invention relates to a protection circuit of a power supply.

BACKGROUND 10 An energy supply operating in zero voltage switching (ZVS) and advance modes described in the U.S.A. No. 5,877,946, issued March 2, 1999, entitled TO FORWARD CONVERTER WITH AN INDUCTOR COUPLED TO A 15 TRANSFORMER WINDING (An Advance Converter with an Inductor Coupled to a Transformer Winding), in the name of WV Fitzgerald (the patent from Fitzgerald), includes a main switching transistor coupled to a primary winding of a main power transformer. The supply voltages of outputs are developed from voltages developed in secondary windings of the transformer. When the transistor is conductive, a current pulse develops in the primary winding of the transformer and in the transistor. A voltage is also developed in a current sensing resistor coupled in series with the transistor. The voltage in the current sense resistor is coupled to a first input of a comparator of a control circuit. A second comparator input is coupled to a capacitor that develops a variable voltage according to an output voltage of the power supply to provide regulation. During a given conduction interval of the transistor, the comparator is driven when the current sensing resistor voltage exceeds a comparator threshold voltage established by the capacitor voltage. An output of the comparator is coupled to the base of the transistor to control the instantaneous shutdown of the transistor on a pulse current basis by pulse current. In normal operation, a voltage, present through the primary aspect of the main power transformer, reduces the voltage across the supply inductance. This voltage is proportional to the output voltage produced in a given secondary winding of the transformer. The output voltage produced by the secondary winding is stepped by the ratio of turns of the transformer. When the switching transistor is turned off at the end of each cycle, a negative voltage pulse, reflected from the secondary side of the transformer, reduces the collector voltage of the transistor. An excessive collector voltage may develop in the main switching transistor if an overload condition occurs in one of the secondary winding. The overvoltage is caused by an excessive current flowing in a resonant supply inductance that resonates with a resonant capacitor which is coupled to the collector of the main switching transistor to form the zero voltage switching, ZVS. If a severe overload occurs in one of the secondary windings, which causes the power supply to fall out of regulation, the voltage across the primary winding of the transformer also drops, since the voltage reflected by the ratio of turns of the transformer is reduces. The result is that the collector voltage of the transistor can become excessive. When the output voltage produced by the secondary winding falls outside the regulation, a maximum current limit is established by the control circuit. Under an overload condition, the transistor will still allow the maximum current to flow through the supply inductance. However, the energy stored in the supply inductance is not supplied to the load through the transistor. The stored energy produces resonant current in the resonant current in the resonant capacitor when the transistor is turned off at the end of the cycle and causes the collector voltage of the transistor to rise substantially above the normal operating voltage, possibly exceeding the voltage ratio of the transistor. interruption of the transistor. Since, under overload, the energy that is stored in the supply inductance during each cycle is not supplied to the load, the energy returns to the unregulated supply that energizes the transistor through a negative inverse current. It may be desirable to reduce the voltage Excessive collector resulting.

COMPENDIUM OF THE INVENTION In order to carry out an aspect of the invention, the inverse negative current is directed through the base-collector junction of the transistor, when the transistor is turned off, in a direction opposite to the forward current of the collector. The collector feed current occurs when the transistors are turned on. The reverse current of the collector produces a voltage in the aforementioned current perception resistor, at an opposite polarity with respect to its polarity, during the forward driving of the transistor. During an overload, the voltage produced by the perceived reverse current turns on a diode switch and changes a load on the capacitor that is coupled to the second input of the control circuit comparator in a manner that reduces the peak current of the collector . In this way, the peak forward current in the transistor is reduced relative to a value that could have occurred without protection. The result is that excessive collector voltage, advantageously, is avoided.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 illustrates a schematic diagram showing an illustrative embodiment of the circuit of the invention; and Figures 2a and 2b illustrate waveforms useful to explain the operation of the circuit of Figure 1.

DETAILED DESCRIPTION OF THE INVENTION Figure 1 illustrates a zero voltage commutation feed converter or power supply 300. A protection circuit 400, modeling one aspect of the invention, provides protection to the zero voltage switching power supply 300. The power supply of Zero voltage switching 300 operates in a manner similar in many respects to that described in the Fitzgerald patent. The energy, for example, of 200 watts, is supplied to a load 303 and a load 302 coupled to a secondary winding T1W2 and to a secondary winding T1W3, respectively, of a transformer T1, during the driving time of a switching transistor Q1 . The transistor Q1 operating as a switch is coupled in series as a primary winding T1W1 of the transformer T1 to conduct current from an input supply, directing the current voltage (DC) RAW B +. A current transformer T2, considered as an excitation transformer, supplies a base current iB to the switching transistor Q1. The RAW B + voltage can be derived from a bridge rectifier (not shown) that rectifies a main supply voltage, and is coupled to a filter capacitor (not shown). Also coupled to the emitter of the transistor Q1 in series with the transistor Q1 is a current sense resistor R7. A resonant capacitor C8 is coupled to a primary winding T1W1 and to the collector of transistor Q1. A resonant circuit 301 includes capacitor C8, a reflected capacitance CSEC, an inductor of limiting current supply Lres, a primary winding T1W1 and a primary winding T2W1 of transformer T2. The primary winding T1W1 is coupled in series with the primary winding T2W1 of the current transformer T2. The resonant circuit 301 produces a half-cycle resonant voltage VQ1 each cycle when the transistor switch Q1 is turned off. A collector voltage VQ1 through transistor Q1, (and capacitor C8) rises to a peak point and then falls to approximately zero at a substantially sinusoidal average wave. After the resonant voltage VQ1 reaches zero, a series array of the resistor R7, a diode D2, a zener diode D20, coupled in parallel with a capacitor C2, and the base-collector junction of the transistor Q1 form a low impedance which sets the voltage VQ1 to a voltage close to the terrestrial potential. The transistor Q1 is then switched back to approximately zero volts to provide zero voltage switching. A secondary winding T1W3 of transformer T1 is coupled to an anode of a rectifier diode DOUT3, the cathode of which is coupled to a CFILTER3 filter capacitor. The winding T1W3 is coupled through a low impedance current path, during the feed driving operation, to the filter capacitor CFILTER3 and to the load 302. Similarly, a secondary winding T1W2 is coupled through the rectifier diode DOUT2 to the CFILTER2 filter capacitor to provide the output voltage REG B +. The CSEC capacitor can be included in one or both of the secondary winding circuits T1W2 and T1W3 in parallel with the winding. The capacitor CSEC is coupled by the transformer to the winding T1W1 forming a part of the resonant circuit 301. Advantageously, each winding T1W2 and T1W3 are hermetically coupled to the primary winding T1W1 in the transformer T1, in such a way that it reduces the leakage inductance. The inductance Lres on the primary side of the transformer T1 is coupled to the transformer to limit the rate of change of each of the currents IDOUT3 and IDOUT2 in the current paths including the diodes DOUT3 and DOUT2, respectively, during the forward driving. Advantageously, the inductance Lres is shared in common with each of the windings T1W2 and T1W3. When the transistor Q1 is conductive, advantageously, a current produced in a secondary winding T2W2 is to supply the current in the primary winding T2W1 of the transformer T2. The winding T2W1 of the transformer T2 is coupled in series with the winding T1W1 of the transformer T1 and the switching transistor Q1. Therefore, a base current B varies approximately linearly with the collector current iQ1. Advantageously, overexcitation of the base of transistor Q1 is avoided by a proportional excitation technique. The control of a workstation of the transistor switch Q1 is based, for example, on the perception output voltage REG B +, directly, instead of the output voltage U. An error amplifier A is sensitive to the voltage REG B + and may include, for example, a comparator having inputs coupled to the output voltage REG B + and a voltage divider providing a predetermined threshold. The error amplifier A is optically coupled through an opto-coupler μ1 to control an activation level or threshold of a comparison transistor Q3. The voltage on the emitter of transistor Q3 is developed from the load on a C6 capacitor. The emitter voltage in capacitor C6 is limited to a forward diode drop by a diode D7, coupled to ground. The load on the capacitor C6 is regenerated, while the transistor Q3 is conducting and is drained by the octo-coupler μ1 when conducting in response to an output signal from the error amplifier A. When the transistor Q1 is conductive, a voltage VR7 a through the resistor R7, which is to provide the current level of the transistor Q1, is coupled to the base of a comparator transistor Q3. The voltage representing current VR7 is the resistor R7 which is coupled to a filter capacitor C7 through a resistor R8. A voltage developed in capacitor C7 of voltage VR7 is coupled to the base of transistor Q3. In a given driving cycle of transistor Q1 when the base voltage of transistor Q3 exceeds a threshold voltage of transistor Q3 which is determined by a control voltage VC6, developed in capacitor C6 in the emitter of transistor Q3, by an amount sufficient for the forward deviation of the base-emitter junction, transistor Q3 starts to drive. In this way, transistor Q3 begins to conduct, when a current i Q 1 in transistor Q1 develops a voltage VR7 in resistor R7 that exceeds the threshold voltage of transistor Q3. When the transistor Q3 conducts, it forms a regenerative closure with a transistor Q2. The collector of transistor NPN, Q3, is coupled to the base of transistor Q2 PNP and the collector of transistor Q2 is coupled to the base of transistor Q3, forming a regenerative switch. The emitter of the transistor Q2 is coupled to the base of the switching transistor Q1 through the diode D20 and a capacitor C2, coupled in parallel. When the closure formed by transistors Q2 and Q3 is activated or activated, transistor Q2 draws current away from the base of switching transistor Q1. A control voltage coupled to the base of the switching transistor Q1 is developed in the emitter of the transistor Q2. The emitter voltage of transistor Q2 forms an output of the regenerative switching array and is coupled to the base of transistor Q1 to turn off transistor Q1 when the lock formed by transistors Q2 and Q3 is activated. The secondary winding T2W2 of current transformer T2 provide the base current iB of the switching transistor Q1 The voltage across the winding T2W2 is an alternating current (AC) voltage, produced when the switching transistor Q1 alternately conducts and shuts off Advantageously, when the transistor Q1 is turned on, the transformer T2 provides a proportional excitation current iB to transistor Q12 to keep transistor Q1 in saturation without over-excitation of transistor Q1 On the other hand, immediately after transistor Q1 is turned off by the operation of transistors Q2 and Q3, the resonant voltage of Q1 in the collector of transistor Q1 is coupled to the base of transistor Q1 through the evacuated T2W2 in a way to keep transistor Q1 non-conductive A collector of an on / off transistor Q4 is coupled through a diode D11 to the emitter of transistor Q2 When transistor Q4 is conductive, according to a signal of ON / OFF, a base current in transistor Q2 occurs which causes transistor Q1 to remain non-conductive The emitter current of transistor Q4 produces a lead in Zener diode D13 Diode D13 is coupled in parallel with a slow-start capacitor C11 which are coupled to the emitter of transistor Q4 The start of the operation cycles occurs when transistor Q4 is turned off and causes transistor Q2 to turn off. Then, a current begins to flow through a resistor R4 and through the parallel arrangement of the Zener diode D20 and the capacitor C2 and produces a starting base current B in the switching transistor Q 1. The resistor R4 is large, and provides only a small amount of the starting base current that activates transistor Q1. As transistor Q1 begins to conduct, a current transformer T2 causes a current to flow in the secondary winding T2W2. The current in the secondary winding T2W2 is proportional to the current in the primary winding T2W1, as a function of its ratio of turns. The diode D1 and a parallel capacitor C10 are coupled in series with the secondary winding T2W2 and with the parallel arrangement of the Zener diode D20 and the capacitor C2 to produce the base current iB of the transistor Q1. The aggregate base excitation current obtains the saturation for the aggregate collector current in a regenerative form, causing the base current iB to increase in proportion to the increase in collector current Q1. When the Q1 transistor becomes saturated, the current of collector iQ1 continues to increase by a rate determined by the total supply inductance coupled in series with the collector of transistor Q1. When the voltage across the current sense resistor R7 is sufficient to cause the transistor Q3 to conduct, an activation current is provided at the base of the transistor Q2. The transistor Q2 conducts and causes an increase in the voltage at the base of the transistor Q3 by producing an additional excitation current in the capacitor C7 and also operating in a regenerative manner to close. The low impedance developed in the emitter of the closed excitation transistor Q2 quickly removes the base load from the base of the switching transistor Q1. The result is that transistor Q1 is quickly turned off. During the time that transistor Q1 is driving, the positive current flows to the base through diode D20 and capacitor C2, which causes capacitor C2 to charge several volts. The capacitor voltage C2 is more positive at the capacitor terminal C2 which is far from the base of the transistor Q1 and less positive than at the base of the transistor Q1. Therefore, when transistors Q2 and Q3 close, they provide a low impedance path to ground, causing the voltage on capacitor C2 to apply a negative offset to the base of transistor Q1. A diode D6 and a resistor R6, coupled in series between the collector of transistor Q2 and the current sense resistor R7, derive some of the reverse base current to resistor R7, which has low impedance, for example, a fraction of one ohm This derivation reduces the tendency to over-excite the base of the transistor Q3, which could otherwise result in excessive storage time and poor switching operation.

After the transistor Q1 is turned off, the winding T2W2 of the transformer T2 produces a negative voltage through a diode D2, having an anode which is coupled to the emitter of the transistor Q1. The excitation control transistors Q2 and Q3 remain closed until the current flowing through them falls below a threshold necessary to keep them regeneratively closed. Then, the negative voltage across diode D2 keeps transistor Q1 from conducting. In addition, a diode D3 and a capacitor C3 are coupled to rectify and filter the negative voltage produced by the transformer 72 to produce a negative supply voltage VMINUS. The resonant action of the resonant circuit 301 causes the base-emitter voltage to reverse the polarity through the winding T2W2. When the voltage at the base of switching transistor Q12 increases to a sufficient magnitude, the current begins to flow at the base of transistor Q1, producing collector current that grows regeneratively, as explained above, forming the beginning of the next cycle. The collector current iQ1 in transistor Q1 starts to flow when the collector voltage VQ1 is at zero volts. In this way, zero voltage switching is obtained. Advantageously, the current transformer T2 provides self-oscillations. In the circuit coupled to the secondary winding T2W2 of the transformer T2, the diode D2 limits the negative voltage developed during the time out of the transistor Q1. Since diode D1 and capacitor C10 form a low impedance, transformer T2 operates as a current transformer during the shutdown interval. The diode D1 provides a current path for the forward driving current and also limits the voltage on the capacitor C10 in parallel with the diode D1, to the forward voltage developed through the diode D1 when it drives. The diode D1, the capacitor C2 and the base-emitter junction of the transistor Q1 form a low impedance that operates as a current transformation. Advantageously, by operating as a current transformer, transformer T2 does not need to store a large magnetic energy and may have a small core. During the start interval, the VMINUS voltage produces a load current in a resistor R11 which is coupled to the capacitor C11, causing a negative start ramp voltage in the capacitor C11. The ramp voltage in the capacitor C11 is coupled through a resistor R13 to the resistor R8. Consequently, the threshold voltage of the comparator transistor Q3 varies in a ramp form to provide a slow start operation. The VMINUS voltage is also coupled to the emitter of the phototransistor in the opto-coupler μ1. The load on the capacitor C6 is adjusted through the conduct of the opto-coupler phototransistor μ 1, sensitive to signals from the error amplifier A. In this way, the voltage is tightly regulated on a pulse current basis. During normal operation, a voltage is present through the primary winding T1W1 of the main power transformer T1, which reduces the voltage across the limiting current inductance Lres- This voltage is to provide the output voltage REG B + approximately multiplied by the ratio of turns of windings T1W2 and T1W1. When the transistor Q1 is turned off at the end of each cycle, a negative voltage pulse, reflected from the secondary side of the transformer T1, counteracts the positive pulse that appears in a terminal of the current limiting inductance Lres, near the collector of the transistor Q1. In this way, advantageously, the collector voltage VQ1 of the transistor Q1 is reduced. A severe overload can occur, for example, in the secondary winding T1W2. Consequently, the power supply can stop the regulation in a negative feedback loop form. Therefore, the REG B + and U voltages will be reduced. Consequently, the voltage across the primary winding T1W1 of the main power transformer T1 that is reflected by the ratio of rotations of the transformer also falls. The result is that the aforementioned negative reflected pulse is greatly reduced, causing the collector voltage VQ1 through the primary winding T1W1 of the main power transformer T1 to increase substantially much more under normal, non-overload conditions. During a severe overload, the REG B + and U voltages are reduced, due to the loss of regulation. A maximum current limit is established on a current basis, pulse per pulse of current through the control circuit that includes transistors Q2 and Q3. Under an overload condition, transistor Q1 will continue to allow the maximum current? Q1 to flow through the supply inductance that includes the current limiting inductance Lres. However, the energy stored there, for example, the limiting inductance of Lre current is not supplied to the load through the transformer T1 The stored energy is developed in a resonant circuit 301, when the transistor Q1 is turned off at the end of the cycle The increased stored energy causes the collector voltage VQ1 of the transistor Q1 to increase substantially above the permissible normal operating voltage, possibly exceeding the interruption voltage ratio of transistor Q1 Since the energy stored there, for example, the current limiting inductance Lres during each cycle is not supplied to the load, the energy returns to the voltage supply RAW B + This energy back, produces a run reverse or negative element flowing through a series arrangement of the resistor R7, diode D2, Zener diode D20, coupled in parallel with the capacitor C2, the base-collector junction of the transistor Q1 and the windings T1W1 and Lres and develops the voltage VR7 through resistor R7 in a negative polarity FIG. 2b illustrates the waveform of a negative current? R7 in resistor R7 which produces a negative voltage VR7 of FIG. 1 The positive portion of current? R7 of FIG. 2b occurs during the forward driving of transistor Q1 of Figure 1 Figures 2a and 2b illustrate the effect of negative current iR7 of Figure 2b on the conirol voltage VC6 of Figures 1 and 2a. Similar symbols and numbers in Figures 1, 2a and 2b indicate similar items and functions. To carry out an aspect of the invention, a switching diode D10 of Figure 1 is coupled in series with a current limiting resistor R10 between the capacitor C6 and the resistor R7 to reduce the control voltage VC6 of Figure 2a developed in the capacitor C6 of Figure 1, when the negative current IR7 of Figure 2b in the resistor R7 of Figure 1 near the time tO of Figures 2a and 2b, is excessive. The reverse or negative current in Figure 2b occurs when the collector current iQ1 of transistor Q1 of Figure 1 flows to the supply terminal, where the RAW B + voltage develops. As explained above, the inverse current iQ1 flows in a path that includes diode D2, Zener diode D20, coupled in parallel with capacitor D2 and the base-collector junction of transistor Q1. The level of the control voltage VC6 of Figure 2a in the capacitor C6 of Figure 1 determines the maximum forward current i Q 1, when the transistor Q1 is conductive. During the overload, the negative current iR7 of Figure 2b and the negative voltage VR7 through the resistor R7 of Figure 1, turn on the diode 10 and cause the voltage VC6 of Figure 2a in the capacitor C6 in Figure 1 to be reduce. Consequently, the peak advance current in transistor Q1 is advantageously reduced. Advantageously, by forcing the reduction in the peak of each forward current pulse iQ1 in the transistor Q1 during overload conditions, the possibility of exceeding the voltage classification of the transistor Q1 is decreased and the reliability is increased. The addition of the resistor R10 in series with the diode D10 allows a small time constant of the capacitor C6 and the resistor R10 and minimizes the peak current through the diode D10.

Claims (9)

1. - A power supply of switching mode, comprising: a supply inductance (Lres, T1W1) coupled to a source of an input supply source (RAW B +) and a switching transistor (Q1) to produce pulses of current (i Q 1) in the inductance that is coupled to an output (302, 303) of the power supply, a given current pulse having a first (positive) portion to a first polarity, flowing in the transistor, during a first portion of a cycle, when the advancing conduction occurs in the transistor, and a second portion, in an opposite polarity, during the second portion of the cycle, after the advancing conduction ceases in the transistor; a source (C6) of a first control signal (VC6) having a value that is indicative of a current limit; a control circuit (Q2, Q3) coupled to a transistor and responsive to the first (positive) portion of a given current pulse and to the first control signal to limit amplitudes of the current pulses, in a current mode in a base of pulse current control per current pulse, according to the current limit value; wherein a protection circuit (400) coupled to the control circuit and responsive to the second portion (negative polarity) of the current pulse to vary the first control signal, according to the second portion of the current pulse, to vary the Current limit value, when a magnitude of the second portion is within a first scale of values, the variation of the first control signal being disabled, when said magnitude of the secondary portion is within a second scale of values.

2. A switching mode power supply according to claim 1, wherein the first control signal source comprises a capacitor (C6) and a switch (D10) responsive to the second (negative) portion of the pulse of given current to store a signal (VC6) indicative of the magnitude of the second portion in the capacitor to develop the first control signal there (VC6), during the second portion of the cycle, and to apply said first stored control signal to the control circuit, during the first portion of the cycle.

3. A switching mode power supply according to claim 1, further comprising a capacitor (C8) coupled to the inductance (T1W1, T2W1) to form a resonant circuit (301), when the transistor (Q1) it is non-conductive, wherein the second (negative) portion of the current pulse is produced in the resonant circuit.

4. A switch mode power supply according to claim 1, further comprising a transformer (T1) having a first winding (T1W1) that is included in the supply inductance and a second winding (T1W2) that is coupled to a rectifier (DOUT2), where the power supply operates in a forward mode.

5. A switch mode power supply according to claim 1, further comprising a current sensor (R7) coupled to a current path of the transistor (Q1) to apply said first (positive) and second (negative) ) portions of the current pulse given to the control (Q2, Q3) and protection (400) circuits, respectively.

6. A switching mode power supply according to claim 5, wherein the current sensor (R7) comprises a current sensing resistor that is coupled to a main current (emitter) conducting terminal of the transistor (Q1) away from the supply inductance.

7. A switch mode power supply according to claim 1, wherein the control circuit (Q2, Q3) comprises a comparator (Q3) having a first input (base of Q2) sensitive to the first portion ( positive) and a second input (transmitter) sensitive to the second (negative) portion of the given current pulse.

8. A power supply switching mode according to claim 1, wherein the first source of control signal (C6) is sensitive to an output of the power supply (REG B +) to vary said first control signal (VC6) according to a difference between the supply output and a signal of reference, (error amplifier A). 9.- A power supply of switching mode, comprising: a supply inductance (Lres, T1W1) coupled to a source of an input supply voltage (REF B +) and a switching resistor (Q1) to produce current pulses (iQ 1) in the inductance that are coupled to an output (302, 303) of the power supply; means (R7) coupled to the transistor to sense the current there; a control circuit (Q2, Q3) coupled to the transistor and responsive to a first (positive) portion of an output signal (VR7) of the current sensor to establish a current limit of a given current pulse, in a current on a pulse current control basis per current pulse, according to the first portion of the output signal of the current sensor; wherein a protection circuit (400) coupled to the control circuit and responsive to the second (negative) portion of the output signal of the current sensor for varying the current limit, according to the second portion of the current pulse, said second and first portions of the output signal occurring at mutually exclusive intervals. SUMMARY When a main switching transistor (Q1) of a zero voltage switching power supply is conductive, a voltage (VR7) is developed in a current sense resistor (R7) coupled in series with the transistor. The voltage in the current sense resistor is coupled to a first input (base) of a comparator (Q3) of the control circuit (Q2, Q3). A second input (transmitter) of the comparator is coupled to a capacitor (C6) that develops a voltage (VC6) that varies according to an output voltage (IDOUT2) of the power supply. During a given conduction interval of the transistor, the comparator (Q3) is activated according to the difference between the current sensing resistor voltage and the capacitor voltage. An output of the comparator (Q3) is coupled to the base of the transistor to control the instantaneous shutdown of the transistor on a pulse current basis by pulse current. Under an overload condition, when the transistor is turned off, a reverse collector current flows through the base-collector junction of the transistor. The reverse collector current produces a (negative) voltage in the current sense resistor which is at opposite polarity with respect to the voltage therein, during the forward driving of the transistor. The voltage produced by the perceived reverse current turns on a diode switch (D10) and charges the capacitor which is coupled to the control circuit comparator in a manner that reduces the peak advance collector current. In this way, the peak forward current in the transistor is reduced relative to a value that could have occurred without the protection.