MX2010005399A - Code enhanched staggercasting. - Google Patents

Abstract

A method and architecture for processing signal communications between an encoder and decoder operating according to the ATSC standard adapted for mobile handheld transmission is disclosed. The method and apparatus comprises transmitting a packet and a redundant packet according to spatial, time and frequency diversity to enhance the redundancy error processing.

Description

ALTERNATE MODELING IMPROVED CODE Cross Reference with Related Requests This application claims the benefit of United States Provisional Patent Application No. 61 / 003,041, entitled "Code Enhanced Staggercasting" and No. 61 / 002,977, entitled "Mobile Broadcasting ATSC M / H for Portable Services", which they are incorporated here as a reference in their entirety.

Field of the Invention The present invention relates to the transmission of data in a transmission system of multiple modes. In particular, the present invention relates to a transmission system, wherein multiple code rates may be used in the transmission of data within a single standard transmission protocol, such as ATSC.

Background of the Invention In previous decades, video transmission systems have migrated from analog to digital formats. In the United States, radio-broadcasters are in the final stages to complete the switching from the analog television system of the National Television System Committee (NTSC) to the Advanced Television Systems Committee (ATSC) digital television system A / 53. Standard A / 53 provides "the specification of system parameters, including formats video encoder input scan and the pre-processing and compression parameters of the video encoder, the format of the audio encoder input signal and the audio encoder pre-processing and compression parameters , the service multiplexing and the characteristics of the transport layer and of the normative specifications, and the VS B RFT / sub-transmission system. Standard A / 53 defines the way in which source data should be processed and modulated (for example, digital audio and video data) in a signal to be transmitted to the air. This processing adds redundant information to the source data so that a receiver can recover the source data even when the channel adds noise and interference of multiple paths in the transmitted signal. The reduced information added to the source data reduces the effective speed at which the source data is transmitted, but increases the potency for a successful recovery of the source data of the received signal.

The development process of the ATSC A / 53 standard focuses on H DTV and fixed reception. The system was designed to optimize the bit rate of the video for the high resolution television screens that are already entering the market. However, the diffusion of the transmissions under the ATSC A / 53 standard presents difficulties for mobile receivers. Improvements are required in the standard for the receipt of robu sta digital television signals by mobile devices.

Recognizing this fact, in 2007, the ATSC launched the launch of a process to develop a standard to allow broadcasters to deliver television content and data to third parties. mobile and portable devices through its digital broadcast signal. In response, multiple proposals were received. The resulting standard, called ATSC-M / H, is intended to be regressively compatible with the ATSC A / 53 standard, which allows the operation of the existing ATSC services in the same RF channel without an adverse impact on the equipment. existing receiver.

Many systems for transmission to mobile devices, such as some proposed ATSC-M / H systems, carry out the periodic transmission. Such systems may include a preamble in their transmissions in order to assist with the operation of the receiving system. Typically, the preambles include known information that portions of the receiver system can use for training to improve reception, which can be particularly useful in difficult environments, such as those found in the mobile operation. Such systems can also encode data at different code rates. The code rate or information rate of an error correction code without return channel (FEC), for example, a convolutional code, establishes the portion of the total amount of information that is not redundant. The code rate is typically a fractional number. When the code rate is k / n, for each k bits of useful information, the encoder generates n total data bits, of which n-k are redundant.

The existing ATSC M / H proposal includes the use of separable block codes to allow for improved code time and frequency diversity. In the example of a coded transmission of I saw speed, the mobile data is entered inside the FEC encoder that emits 2 bytes for each input byte. The two bytes represent the original data and the redundant data. A receiver can receive either the original data or the redundant data at the receiver's threshold of the original data. When both streams are received, there is an advantage of coding gain for the receiver to retrieve the data to a threshold below the original data. The mobile and portable operation of the communications equipment have some of the greatest challenges, with extreme differences in the transmission channel due to the constructions and moving vehicles, as well as other differences. A system can be used to provide data in a redundant manner. It would be desirable to take advantage of redundant information through frequency, time and spatial diversity to improve reception on mobile devices.

Brief Description of the Invention In accordance with one aspect of the present invention, a method is provided.

In accordance with another aspect of the invention, another method is provided.

Brief Description of the Drawings Figure 1 is a diagram of one modality of a terrestrial broadcast transmitter for mobile / portable reception of the present invention.

Figure 2 is a block diagram of one embodiment of a portion of an exemplary mobile / portable data stream of the present invention.

Figure 3 is a block diagram of an embodiment of an exemplary data chart of the present invention.

Figure 4 is a block diagram of a modality of a terrestrial broadcast receiver for mobile / portable reception of the present invention.

Figure 5 is a block diagram of a mode of a decoder of the present invention.

Figure 6 is a block diagram of another embodiment of a decoder of the present invention.

Figure 7 is a block diagram of a terrestrial broadcast environment, in accordance with the present invention.

Figure 8 is a block diagram of one embodiment of a portion of a transmitter, in accordance with the present invention.

The examples set forth herein illustrate the preferred embodiments of the invention, and such examples should not be construed as limiting the scope of the invention, in any sense.

Detailed description of the invention As described above, the present invention provides a method and apparatus for transmitting data in a mobile broadcasting system that utilizes data diversity and redundancy, such as the system ATSC-M / H proposed, while allowing regressive compatibility with legacy transmission and receiver paths, such as ATSC A / 53. While this invention has been described with a preferred design, the present invention can also be modified within the spirit and scope of the invention. Therefore, this application is intended to encompass any variation, use or adaptation of the invention with the use of its general principles. Furthermore, this application is intended to encompass the sections of the present invention that are within the practice known or accustomed in the art to which this invention pertains and which fall within the limits of the appended claims. For example, the described technique will be applicable to transmission systems designed for other types of data or that use different schemes of coding, error correction, redundancy, interleaving or modulation.

With reference now to the drawings and more particularly to the Figure 1, a block diagram of an embodiment of a terrestrial broadcast transmitter for mobile / portable reception of the present invention is illustrated. The embodiment 100 of Figure 1 comprises a plurality of signal transmission means, such as an MPEG transport stream source 110, a legacy ATSC A / 53 preprocessing path 115. The elements within the pre-processing ATSC-M / H 115 comprises a packet interleaver 120, a serial concatenated block encoder 125, a packet deinterleaver 130, a header modifier 135 of the MPEG transport stream, an inserter 140 of preamble packet. The ATSC A / 53 pre-processing path 145 bequeathed it comprises a data scrambler 150, a Reed-Solomon encoder 155, a byte interleaver 160, a convolutional encoder 165, a synchronization inserter 170, a pilot inserter 175 and a modulator 180.

In the ATSC-M / H preprocessing stream, the incoming MPEG transport data 112 from an MPEG transport stream source 110 are received in the packet interleaver 120. In packet interleaver 120 re-arranges the number in sequence of bytes within a different sequence to improve the bit error rate and the operation of the frame error rate. In this exemplary embodiment, the packet interleaver 120 takes the bytes of a fixed number of consecutive packets in a row order per row, and outputs the bytes column by column. In this way, all the first bytes of the packets are grouped together, all the second bytes of the packets are grouped together and so consecutively, until the last bytes of the packets. Each source packet is an MPEG transport stream packet with the synchronization byte removed, so that the packet length is 187 bytes. The number of packets in each code box is the same as the number of source symbols required for the block code concatenated in series GF (256).

The interleaved data is then linked with the concatenated block encoder (SCBC) 125 in series of the Galois field (256). The concatenated block code decoder (SCBC) 125 in series GF (256) will take different forms depending on the speed mode for the current symbols. In general, it consists of a decoder component that iteratively decode the smooth information in a turbo decoding manner. The SCBC 125 encodes the interleaved data of the packet in one of a plurality of ways depending on the desired data rate and the length of the code word. The SCBC 125 consists of one or more GF (256) codes cascaded in a serial form, linked by GF (256) code optimized block interleavers to improve overall code performance. This can optionally be followed by a GF (256) jump to reach the desired code word.

Specifically, the field Galois GF (pn) is a mathematical group that comprises a finite number of elements pn, where the values of p and n are integers. A particular Galois field is defined with the use of a generator polynomial g (x). Each element of the Galois field can be represented by a unique pattern of bits that has n bits. In addition, a single polynomial of degree p "can be associated with each element, where each polynomial coefficient is between 0 and p-1. In addition, mathematical operations in the Galois field have important properties." The addition of two elements of the field Galois GF (pn) is defined as an element associated with a polynomial that has coefficients that are the sum of a p-modulus of the coefficients of the polynomials associated with the two elements to be added in. Similarly, the multiplication of two elements is defined as the multiplication of the polynomials associated with the two element modules of the generator polynomial g (x) associated with the Galois field The addition and multiplication operators are defined on the Galois field, so that the sum and the product of any of the two elements of the Galois field are elements of the Galois field. One property of the Reed-Solomon code word is that it multiplies each byte of the code word by an element of the Galois field, which results in another valid Reed-Solomon code word. In addition, the byte-by-byte addition of two Reed-Solomon code words produces another Reed-Solomon code word. The standard A53 legacy defines a Galois field of 256 GF elements (28) and the polynomial g (x) associated generator for use in the Reed-Solomon algorithm. The properties of the Galois field also creates the ability to generate syndromes for the code words in order to determine errors. Another important property of the code word.

In an exemplary embodiment, the two code words or packets generated by the ½ velocity byte code coder include a duplicate of the original code word and a new code word that provides redundancy to the original code word. The two code words can also be described as systematic data and non-systematic data. It is important to note that code words representing systematic and non-systematic data can be arranged to form larger data structures. In a preferred embodiment, the code words can be organized to form a reinforced data stream of data packets. The reinforced data stream includes systematic packets, which are duplicates of data packets in a portion A, of stream and unsystematic packets generated by the processing of a byte code encoder in a stream portion A '. The non-systematic packages they also include packages that can be derived from other systematic and unsystematic packages of the reinforced data stream. In addition, packets in the reinforced data stream can also be composed of systematic bytes and non-systematic bytes. In such modalities, a systematic byte is a duplicate of the byte of the content data and the non-systematic byte is one that is derived from other systematic and non-systematic bytes.

The word code or non-systematic or redundant packet issued by a byte code encoder is the result of multiplying each byte of the code word or incoming packet by a b element of the Galois field GF (256). In a modality, when the MPEG transmission source 1 1 0 generates an M message, which is composed of the bytes M (1), M (2), ... M (187), where M (1) is the first byte of the message, M (2) is the second byte of the message, etc. , thus consecutively, the byte code encoder 104 produces the codewords A and A 'from the code word M, as follows: A (i) = M (i) ¡= 1, 2, ..., 187 (1) A '(i) = b * M (i) ¡= 1, 2 187 (2) The value b is a predetermined element (not zero) of the same Galois field GF (256) that can be used by the 1 55 Reed Solomon coder. In an illustrative mode, the value of element b is 2. It should be evident that with the use of the same Galois field for both, the byte code encoder and the Reed Solomon encoder allow operations between the two encoders based on the properties of the Galois field. The byte code encoder 125 encodes all bytes of the data packet, including the bytes forming the header containing the PID, to generate one or more non-systematic packets of the reinforced data stream. In this way, the PID of each non-systematic packet is a coded byte code and can no longer represent a PID value that can be recognized by a receiving device.

It should be evident that any packets encoded by the transmitter mode illustrated by the encoder 100 can be decoded by a mode of a decoder used in the legacy receiver that is compatible with the A53 standard. The decoder in a legacy receiver provides packets of the enhanced data stream to a data decoder. The reinforced data stream includes non-systematic packets that are encoded with the use of the byte code encoder that will be decoded correctly by a decoder in a legacy receiver, but will result in a data content that can not be recognized by the legacy receiver. However, because such packets have a PID that is not associated in the Program Map Table (PMT) with a legacy or existing data format, the content decoder in a legacy receiver ignores these unsystematic packets of the current of reinforced data.

The byte code encoder 125 uses equation (2) above to generate a non-systematic packet for each systematic packet and provides both packets for the legacy 8-VSB encoder for transmission, to produce a coded stream with a speed effective data of 72 (that is, 1 input byte, 2 output bytes). As mentioned above, the byte code encoder 125 may have the ability to use other coding rates to produce other effective data rates. In some embodiments, the byte code encoder can produce a byte coded packet for every two source packets, MA and MB, received from the MPEG TS source 110 to generate a reinforced data stream of 2/3 speed comprising two systematic packages and a non-systematic package calculated as follows: MAB (Í) = MA (i) * bi + MB (i) * b2 i = 1, 2 187 (3) wherein MA and MB are consecutive systematic packets produced by the data generator 102 and b, and b2 are predetermined elements of the Galois field, such as the Galois field used by the 155 Reed Solomon encoder. In an illustrative mode, the value of the elements b, and b2 is 2. In some embodiments, the values of bi and b2 may not be identical. The byte code encoder 125 provides the MA, MB and MAB packets for the legacy 8-VSB 130 encoder for other encoding and transmission.

The byte code encoder 125 may use different coding rates to produce enhanced data streams (ie ones having lower data rates) by including additional input data packets to generate redundant packets. Another mode of the byte code encoder 125 produces a data stream of 4/9 speed by employing four systematic MA, MB, Mc and MD packages from source 110 MPEG TS and 5 non-systematic packages calculated as follows: AB (Í) = A (i) * bt + s (i b2, 187 (4) CD (Í) = Mc (i) + MD (i) * b i 1"'1 Üd!, 187 (5) MAC (Í) = MA (¡rb5 + Mc (go 6 i = 1 2, 187 (6) MBD (Í) = Me (í) * b7 + Mo (i) * b8, 187 (7) MABCD (Í) = MAB (i) * b9 + MOD (i) * b10 ¡= 1,2F ..., 187 (8) The values b1f b2l ... b10 are default elements selected from the Galois field. In an illustrative mode, the values of bt b2l ... b 0 are 2. In addition, as shown in equation (8), the MABCD packet is a redundant packet generated from other redundant packets only, specifically the MAB and Meo packets. - It must be evident that the redundant MABCD packet can be generated in an alternative way with the use of the elements of the redundant MAC and MBc packets. In some embodiments of the generator 110 of the MPEG transmission source, the removal of one or more non-systematic packets can be carried out in an operation known as a hop. For example, a 4/8 skip velocity may occur when not generating one of the packets that used only redundant packets (ie, MABCD in this case) since this packet contains the smallest amount of intrinsic data. Any package or code word can be removed. However, the removal of a package or code word that contains the least amount of intrinsic data can be optimal. Code skipping can be used to change the number of transmitted packets in order to meet certain limitations in the number of packets or code words transmitted.

In addition, the byte code encoder 125 can also produce a reinforced data stream having a data rate of 8/27 by employing 8 data packets, MA, MB MH to produce 19 non-systematic packets, as follows: MAB (Í) = MA (i) * bi + M8 (i) * b2 i = 12, 187 (9) CD (i) = Mc (i) * b3 + D (i) * b4 i = 1,2, 187 (10) MAC (¡) = MA (i) * b5 + Mc (i) * b6 1 = 1.2, .., 187 (11) MBDO) = MB (¡) * b7 + D (¡) * b8 i, 2, .., 187 (12) MASCD (Í) = AB (i) * b9 + Mco (i) * bio ¡= 1,2, .., 187 (13) MEF (Í) = Me (irb + F (i) * b12 = 1,2, .., 187 (14) GH (Í) = G (i) * bi3 + MH (i) * b14 j = 12, 187 (15) EG (Í) = ME (¡) * b15 + MG (i) * b16 1,2, .., 187 (16) MFH (Í) ~ MF (i) * bi7 +? (?) ¾? ß i = 1,2,, 187 (17) MEFGH (Í) = MEF (i) * b19 + MGH (i) * b20 i = 1,2, ", 187 (18) MAe (i) = MA (i) * b2i + ME (l) * b22 1 = 1,2,, 187 (19) MBF (Í) = MBC¡) * b23 + Mp (i) * b24 Í = 1 -, 187 (20) Mco (i) = Mc (i) * b25 + G (j) * b26 = 1.2,., 187 (21) MOH (¡) = M0 (¡) * b27 + MH (írb28 ¡= 1,2, ..., 187 (22) MACEG (Í) = MAC (¡) * b29 + EG (i) * b3o ¡= 1,2, ..., 187 (23) MBDFH (Í) = MBo (i) * b3i + MFH (i) * b32 = 1.2, .., 187 (24) MABEF (Í) = AB (í) * b33 + MEF. { i) * 34 i = 1.2, .., 187 (25) CDGH (Í) = Mco. { ¡) * B35 + GH (i) * 36 i = = 1,2 ..., 187 (26) MABCDEFGH (Í) 1 * 1,2, ..., 187 (27) In addition, a skipped code with a data rate of 8/26 can be generated by the bytecode encoder 125 by not generating a packet MABCDEFGH of lower intrinsic data value or another packet generated from only the redundant packets.

As described above, the byte code encoder can be configured to produce certain coding code rates based on the number of code words or packets used and the number of code words or packets formed through a single coding process . In addition, the more complicated code rates can be constructed with the use of particular arrangements of the code-rate encoders described above as building blocks or code-rate coders formers. In addition, additional processing blocks can be included to form a concatenated byte code encoder. For example, the concatenated byte code encoder can use additional interleaving blocks between the byte code coders for trainers in addition to the redundancy to improve the reinforcement of the data stream produced. Various embodiments of the encoder enhanced and redundant alternate modeling transmission methods will be described below.

After coding, the data is coupled with a data de-interleaver 130. The packet deinterleaver 130 takes the bytes of the resulting SCBC code words for the original group of packets in one column order per column, and outputs the bytes in a row order per row. The original packages are reconstituted and create new packages from the parity bytes of the SCBC code words. Each packet corresponds to a common GF (256) symbol location in all SCBC code words created. The number of packets created in each codebox is nSCBC, where the first kSCBC packets are the original packets of data and the last packets (nSCBC - kSCBC) are parity packets.

The data is then coupled with the MPEG TS header modifier 135 where the MPEG headers are modified. The MPEG TS header modifier can modify the packet identifier (PID) of the transport stream headers to indicate the code rate used by the error correction scheme. The code rates are expressed as a fraction of the original number of bytes of data on the total number of bytes of data used. For example, a speed mode of 12/52 that complements the 12 data bytes with 40 parity bytes, each group of 12 bytes used an encoder R = 1/2, and two encoders R = 12/26, with each encoder 12/26 uses two encoders R = 2/3 and one jump 27/26, resulting in a speed mode of 12/52. The jump R = 27/26 is carried out in such a way that the last byte of the 27 bytes is dropped. The data blocks are used to transmit 12 MPEG TS packets under the 12/52 speed mode. The 12/26 speed mode complements the 12 bytes of data with 14 parity bytes, each group of 12 bytes of data uses two R = 2/3 encoders, and a jump R = 27/26, resulting in one mode of speed 12/26. The jump R = 26/27 must be done in such a way that the last byte of the 27 bytes is dropped. A Data block is used to transmit 12 MPEG TS packets under the 12/26 speed mode. The 17/26 speed mode complements the 17 speed bytes with 9 parity bytes, each group of 17 data bytes is grouped with the use of an R02 / 3 encoder to complement the 16 bytes of data with 8 parity bytes, and an R = 1/2 encoder complements 1 bytes of data with 1 parity byte, resulting in a speed mode of 17/26. A block of data is used to transmit 17 MPEG TS packets under the speed mode 17/26. The 24/208 speed mode complements 24 bytes of data with 184 parity bytes, each group of 24 bytes of data uses 24 R = 1/4 encoders, and eight 12/26 encoders, resulting in a 24-speed mode / 208 The jump R = 27/26 must be done in such a way that the last byte of the 27 bytes is dropped. Eight data blocks are used to transmit 24 MPEG TS packets under the 24/208 speed mode.

Each packet that uses the MPEG protocol typically contains a packet identification or PID portion. The current system allows approximately 8,000 possible unique identifier elements and currently only 50 are used. The PID is typically one or more bytes of information used to identify the type of data in the packet. Currently, many of the PID portions of the bits remain reserved and unused. These PIDs can be used to identify a specific error code rate that will be imposed on the packet. Certain rules based on the MPEG protocol must be maintained in order to ensure that the PID is properly identified by any receiving system. A heading 440 of three bytes contains a packet identifier (PID) of 13 bits that identifies each packet as part of the mobile / portable transmission. The 440 headers of the MPEG packets from the ATSC-M / H stream are modified after the packet deinterleaving to contain the packet identifiers (PID) that are not recognized by the legacy ATSC-M / H receivers. In this way, a legacy receiver must ignore the specific ATSC-M / H packets, which offers backward compatibility.

The data is then coupled with the preamble pack inserter 140, wherein the preamble packets, consisting of consecutive MPEG packets, are formed within a preamble block. MPEG packets are formed with a valid MPEG header with the data bytes generated from the PN generator (not shown). The number of bytes of data generated from the PN generator varies with the code rate used, for example, 184 bytes of data are generated in a 12/52 rate mode, to result in a total of 2208 bytes of PN data. In accordance with an exemplary embodiment, the PN generator is a 16-bit shift register with 9 feedback sockets, 8 of the displacement register outputs are selected as the output byte. The ATSC-M / H packets are placed between the preamble blocks in data blocks. Each data block contains 26 ATSC M / H encoded packets that have the same encoding or 26 ATSC A / 53 encoded packets. Once the preamble packets have been inserted 140, the ATSC M / H stream has been formed.

The ATSC-M / H datastream is then processed by the legacy ATSC A / 53 path 146, including the data scrambler 150, the Reed-Solomon encoder 155, the byte interleaver 160, the encoder 165 12-1, the synchronization insert 170, pilot insertion 175 and modulation 180. In data scrambler 150, each byte value is changed in accordance with a known pattern of pseudo-random number generation. This process is reversed in the receiver in order to recover the appropriate data values. With the exception of the segment and the field synchronizations, it is convenient that the bit stream 8 VSB has a noise nature, completely random, to reach the frequency response of the transmitted signal having a flat spectrum type noise in order to use the allocated channel space with maximum effncy.

The data is then coupled with the Reed-Solomon encoder 155, where the Reed-Solomon (RS) coding provides the additional error correction potential in the receiver through the addition of additional data for the transmitted stream. In an exemplary embodiment, the RS code used in the VSB transmission system is at the code t = 10 (207, 187). The size of the RS data block is 187 bytes, with 20 parity bytes added for error correction. A total RS block size of 207 bytes is transmitted by the RS code word. When creating bytes of the serial bit stream, the MSB must be the first bit in series and the 20 parity bytes RS are sent to the end of the data block or the RS code word.

The byte interleaver 160 then processes the output of the 155 Reed-Solomon encoder. Interlacing is a common technique for handling burst errors that may occur during transmission. Without interleaving, the burst error can have a large impact on a particular segment of data, which produces that incorrigible segment. When data is interlaced before transmission, however, the effect of a burst error can be effectively distributed across multiple data segments. Better than large errors to be introduced in a localized segment that can not be corrected, similar errors can be introduced in multiple segments, with each separated within the correction capabilities of the error correction without return channel, the parity bit or other data integrity schemes. For example, a common Reed-Solomon encoder (255, 223) will allow the correction of up to 16 symbol errors in each code word. When Reed-Solomon encoded data is intertwined before transmission, it is more likely that a large burst of error will be distributed across multiple code words after deinterlacing, reducing the chance that more than 16 correctable symbols are present in any particular code word.

The interleaver used in a VSB transmission system is a 52 data segment (intersegment) of the convolutional byte interleaver. The interlacing is provided at a depth of about 1/6 of the data field (4 ms depth). Only data bytes are intertwined. The interleaver is synchronized with the first data byte of the data field. The interlocking of the segment is also done for the benefit of the process of convolutional coding.

The signal is then coupled with the convolutional encoder 165. Convolutional coding is another form of error correction without a return channel. Unlike the Reed-Solomon encoding, which treats the complete MPEG-2 packet simultaneously as a block, convolutional coding is an involute code that tracks the stream in progress of bits as it develops over time. In accordance with this, Reed-Solomon coding is known with a form of block coding, while convolutional coding is a convolutional code.

In an ATSC convolutional encoding, each 8-bit byte is divided into a stream of four, 2 2-bit words. In the convolutional encoder, each arriving 2-bit word is compared to the past history of the previous 2-bit words. A 3-bit binary code is generated mathematically to describe the transition from previous 2-bit words to the current one. These 3-bit codes are replaced for the original 2-bit words and are transmitted to the air as eight level symbols of 8-VSB (3 bits = 8 combinations or levels). For every two bits that go inside the convolutional encoder, three bits come out. For this reason, the convolutional encoder in the 8-VSB system is said to be a 2/3 speed encoder. The waveform of the signaling used with the convolutional code is a constellation of an 8-level (3-bit) dimension. The transmitted signal is referred to as 8-VSB. A 4-state convolutional encoder should be used.

In an exemplary embodiment, intra-interlacing is used segments of convolutional code. This uses twelve identical convolutional coders and precoders that operate on interlaced data symbols. The code interleaving is achieved by coding the symbols (0, 12, 24, 36 ...) as a group, the symbols (1, 13, 25, 37, ...) as a second group and the symbols (2) , 14, 26, 38, ...) as a third group and so on, consecutively for a total of 12 groups.

Once the data has been encoded convolutionally, it is coupled with the inserter 170 of synchronization. The synchronization inserter 170 is a multiplexer that inserts the different synchronization signals (data segment synchronization and data field synchronization). A data segment synchronization of 4 two-level symbols (binary) is inserted into the 8-level digital data stream at the start of each data segment. The MPEG sync byte is replaced by the data segment synchronization. In an exemplary mode that uses ATSC transmission standards, a complete segment would consist of 832 symbols: 4 symbols for data segment synchronization and 828 data plus parity symbols. The same synchronization pattern occurs regularly at 77.3 intervals, and is the only signal that repeats at this speed. Unlike the data, the four symbols for the data segment synchronization are not Reed-Solomon or are not convolutionally encoded, nor are they intertwined. An ATSC segment synchronization is a repetitive pulse of four symbols (one byte) that is added to the front of the data segment and replaces the first missing byte (packet synchronization byte) of the original MPEG-2 data package. The correlation circuits in the 8-VSB receiver are based on the repetitive nature of the segment synchronization, which is easily contrasted against the background of the completely random data. The received synchronization signal is used to generate the receiver clock and recover the data.

Segment synchronizations can be easily retrieved by the receiver due to their receiving nature and their long duration. Accurate clock recovery can have noise and interference levels as well as exact data recovery is impossible, allowing rapid data recovery during channel changes and other transient conditions.

After the synchronization insertion, the signal is coupled with the pilot insert where a small DC offset is applied to the baseband signal of 8-VSB, which causes a small residual to appear at the zero point of frequency of the Modulated spectrum remaining. The ATSC pilot signal offers the RF PLL circuits in the 8-VSB receiver a signal to be fixed that is independent of the data to be transmitted. The frequency of the pilot is the same as the carrier-suppressed frequency. This can be generated by a small DC (digital) level (1.25) added to each symbol (data and synchronization) of the digital baseband data plus the synchronization signal (+1. +3. +5. +7) . The pilot power is typically 11.3dB below the average power of the data signal.

After the pilot signal is inserted, the data is coupled with a modulator 180. The amplitude of the modulator modulates the band signal 8VSB base in an intermediate frequency carrier (IF). With traditional amplitude modulation, a double sideband RF spectrum is generated approximately as our carrier frequency, with each RF sideband being a mirror image of the other. This represents redundant information and a sideband can be discarded without any loss of information. In the 8 VSB modulation, the VSB modulator receives the 10.76 Msymbols / as, the composite data signal encoded in convolutional form, of 8 levels (pilot and synchronization added). The performance of the ATV system is based on a linear phase high cosine Nyquist filter response in the concatenated transmitter and receiver, as shown in Figure 12. The filter response of the system is essentially flat across the entire band, except for the transition regions at each end of the band. In normal form, a detachment in the transmission must have the response of a root cosine filter, linear phase.

The transmission system includes the operation for mobile and portable devices in a transmission burst mode. Several key operating advantages in burst mode are described throughout the document and include the ability to be received by a new class of devices, while maintaining backward compatibility. These new classes of devices require a lower level of video resolution than is found in the existing broadcast standard and therefore, may allow for greater coding and compression, as well as other features that include working in the presence of higher noise levels. . An additional advantage of burst mode types of operation it focuses on the device's power savings by focusing device usage only when the signals that are for the device are going to be received.

The burst mode operations, such as those described, can take advantage of the periods of time during which high data transmission of a signal is not required, in order to maintain the total operation of the legacy system and the receiver. The burst mode operation may be based on the processing signals based on the so-called processing speed of new information, which may change depending on the characteristics of the current broadcast signal.

The backward compatibility with the legacy system is maintained by focusing the burst mode operations to the packed level of data when entering the information for the new program identifiers. The new program identifiers allow the new class of equipment to recognize the data, without affecting the operation of the existing equipment. There is another legacy support to include an overlay structure in order to maintain the legacy signal transmission operation during certain profiles of the burst mode.

By taking advantage of the complete redundancy of systematic data and non-systematic data and the ability of the signal to be retrieved from fully separable code words, adding frequency or special diversity, as well as time diversity can increase the probability of reception by the mobile device. This will be described later with respect to Figure 7.

Referring now to Figure 2, a block diagram of one embodiment of a portion of an exemplary mobile / portable data stream of the present invention is illustrated. 26 ATSC M / H coded packets are grouped in data block 1. In the legacy ATSC transmission, each data block typically has the same encoding, although this is not physically required. The preamble blocks are two long blocks and have 52 ATSC M / H encoded packets. The first MPEG packet after the preamble block is a control packet containing system information. After scrambling and error correction processing without return path, the data packets are formatted into data frames for transmission and data segment synchronization and data field synchronization are added.

The ATSC M / H data stream 200 is formed of bursts having the same preamble block 210 followed by a predetermined number of data blocks 230 appropriate for the selected data rate mode. In accordance with an exemplary embodiment, each data block 230 consists of 26 MPEG packets. Each data box consists of two data fields, each containing 313 data segments. The first data segment of each data field is a single synchronization signal (data field synchronization) and includes the training sequence used by the equalizer in the receiver. Each of the remaining 312 data segments carries the data equivalent of a packet and transport of 188 bytes plus its FEC header. The actual data in each data segment comes from several transport packages due to data interleaving. Each data segment consists of 832 symbols. The first 4 symbols are transmitted in binary form and provide segment synchronization. This data segment synchronization signal represents the synchronization byte of the MPEG compliant transport packet of 188 bytes. The remaining 828 symbols of each data segment carry data equivalent to the remaining 187 bytes of a transport packet and its associated FEC header. These 828 symbols are transmitted as 8-level signals and therefore carry three bits per symbol. In this way 828 x 3 = 2484 bits of data are carried in each data segment, which is the requirement to send a protected transport packet.

The ATSC M / H data stream consists of a sequence of blocks, each block consisting of 26 packets from a legacy A / 53 VSB system or from an ATSC M / H system. The ATSC M / H data stream is formed from bursts of blocks and each burst has a preamble block followed by the data blocks Nb, where Nb is a variable parameter of the system and a function of the ATSC data rate M / H general to be transmitted. Each data block is encoded in one of the defined ATSC M / H speed modes. This speed mode is applied to the complete data block. For each burst of blocks, the data blocks are delivered so that the highest coded FEC speeds (ie, the lowest fractional numbers) in the burst of blocks will be delivered earlier and the coded FEC speeds lower (ie say, the highest fractional numbers) will be delivered last, so starting from a block of preamble, any data block will have equal or less robustness than the current data block. Legacy data blocks, encoded ATSC A / 53 8 VSB of 26 packets can be placed in one more blocks for the legacy overlay operation.

Referring now to Figure 3, a data box 300 according to the present invention is shown. The displayed data box 300 is organized for transmission, where each data frame consists of two data fields, each containing 313 data segments. The first data segment of each data field is a single synchronization signal (data field synchronization) and includes the training sequence used by the equalizer in the receiver. The remaining 312 data segments each carry the data equivalent from a 188-byte transport packet plus its associated FEC header. The actual data in each data segment comes from several transport packets due to data interleaving. Each data segment consists of 832 symbols. The first 4 symbols are transmitted in binary form and provide segment synchronization. This segment synchronization signal also represents the synchronization byte of the MPEG-compatible transport packet of 188 bytes. The remaining 828 symbols of each data segment carry data equivalent to the remaining 187 bytes of a transport packet and its associated FEC header. These 828 symbols are transmitted as 8-level signals and therefore carry three bits per symbol. In this way, 828 x 3 = 2484 bits of data that are carried in each data segment, which coincides exactly with the requirement to send a transport package protected. 187 bytes + 20 bytes of parity = 207 bytes 207 bytes x 8 bits / byte = 1656 bits Convolutional encoding of 2/3 speed requires 3/2 x 1656 bits = 2484 bits.

The exact symbol speed is determined by the following equation 1: (1) Sr (MHZ) = 4.5 / 286 x 684 = 10.76 MHz The frequency of a data segment is determined by the following equation 2: (2) fseg = Sr / 832 = 12.94 ... x 103 data segments / s.

The speed of the data box is determined by the following equation 3: (3) fcuadro = fseg / 626 = 20.66 frames / s.

The symbol Sr speed and the transport speed Tr must be set to each other in frequency.

Symbols of 8 levels combined with binary data segment synchronization and data field synchronization signals are used to modulate a single carrier by suppressed carrier. Before transmission, however, most of the lower sideband must be removed. The resulting spectrum is flat, except for the band edges, where the high cosine response, nominal square root results in transition regions of 620 KHz. At the carrier frequency suppressed, 310 KHZ from the edge band lower, a small pilot is added to the signal, as described above.

Referring now to Figure 4, a mode of a terrestrial broadcast receiver 400 for mobile / portable reception of the present invention is shown. The receiver 400 comprises a signal receiving element 410, a first tuner 420, a second tuner 425, a first pre-equalizer demodulator 430, a second pre-equalizer demodulator 430, an equalizer controller 440, an equalizer 450, a processor 460 of post-equalizer correction, a transport decoder 470 and a tuner driver 480.

The signal receiving element 410 operates to receive signals including audio, video and / or signals (eg, television signals, etc.) from one or more signal sources, such as a terrestrial broadcast system and / or other type of signal broadcasting system. In accordance with an exemplary embodiment, the signal receiving element 410 is incorporated as an antenna, such as a periodic recording antenna, but can also be incorporated as any type of signal receiving element. The antenna 410, of this exemplary embodiment, operates to receive the audio, video and data signals transmitted in terrestrial ATSC M / H over a frequency bandwidth. In general, ATSC signals are transmitted over a frequency range of 54 to 870 MHz, with a bandwidth of approximately 6 MHz per channel. The sub-channels can also be multiplexed in time. The signals are coupled from the antenna through a transmission line, such as a coaxial cable or a printed circuit board trace.

The first and second tuners 420, 425 operate to carry perform a signal tuning function in response to a control signal from the tuner controller 480. In accordance with an exemplary embodiment, each tuner 420, 425 receives a different time, or an RF signal of varying frequency from one or a plurality of antennas 410 and performs the signal tuning function by filtering and converting the frequency into descending (that is, a single-stage or multi-stage downconversion) in the RF signal, in order to generate an intermediate frequency (IF) signal. The RF and IF signals may include audio, video and / or a data content (eg, television signals, etc.) and may be of a similar analogue standard (eg, NTSC, PAL, SECAM, etc.). and / or a digital signal standard (eg, ATSC, QAM, QPSK, etc.). Each tuner 420 operates to convert the ATSC M / H signal recited from the carrier frequency to an intermediate frequency. For example, the tuner can convert a 57 MHz signal to the antenna 410 in an IF signal of 43 MHz. The pre-equalizer demodulator 430 operates to demodulate the IF signal from the tuner 420 in a digital baseband stream. The demodulator 435 operates to demodulate the IF signal from the tuner 425. The baseband digital streams are then coupled with the equalizer.

The controller 480 of the tuner operates to receive instructions from the transport decoder 470 in response to the signal level and the frequency of the tuned channel or a desired tuned channel. The controller 480 of the tuner generates a control signal in response to the instructions received to control the operation of the tuner 420, 425.

The equalizer controller 440 operates to generate an error term in response to the decoded data received from the demodulators 430, 435. This offers the capability of a directed data equalizer. The equalizer controller 440 calculates the error between the received data and the decoded data and generates an error term. The error term is fed to the equalizer 450 to be minimized.

Equalizer 450 operates to receive the demodulated MPEG current and tuned from demodulators 430, 435 of the pre-equalizer and calculates the equalizer coefficients that are applied to the equalizer filter within the equalizer to produce an error-free signal. Equalizer 450 operates to compensate for transmission errors, such as attenuation and inter-symbol interference. The equalizer comprises a matched filter that performs envelope filtering that operates to cancel interference between symbols. During the training period of the equalizer, a training signal, previously selected, is transmitted through the channel and an appropriately delayed version of this signal, which is pre-stored in the receiver, is used as the reference signal. In general, the training signal is a pseudo-noise sequence long enough to allow the equalizer to compensate for channel distortions. The equalizer according to the exemplary embodiment of the present invention operates to store a plurality of pseudo-noise sequences, wherein each pseudo-noise sequence corresponds to a code speed. When the equalizer 450 receives the training signal from the pseudo-random sequence, the equalizer compares a portion of the received sequence with the plurality of stored sequences. When a match is found, the code rate associated with the received sequence is used by the decoder to decode the data received after the training sequence.

The post-equalizer correction processor 460 and the transport decoder 470 operate to perform the error correction and to decode the MPEG data stream. These elements are shown and described in detail with reference to Figures 5 and 6.

The receiver can be configured to operate with a single tuner and a single demodulator by sharing the time the tuner receives different frequency and at different times. Alternatively, the tuner can be configured with sufficient bandwidth to receive two signals simultaneously, so that both signals can be tuned to different IF frequencies and each of these IF frequencies can be processed simultaneously or multiplexed of time processed by the demodulator. In a signal tuner, with the use of time or frequency diversity, the packet combination is not carried out in the equalizer, rather it is carried out in the code since the equalizer must follow the transmitted signals. This offers three possibilities of receiving the packet correctly, the first packet correctly, the second packet correctly or the combination after the byte decoder. When the coding is used to combining the packets, as opposed to receiving a single packet, decreases the minimum amount of signal-to-noise ratio required to receive a virtually error-free signal. For example, at a rate of ½ code, the minimum threshold is decreased from 15dB for a single packet without coding to 7dB for 2 packets with the coding and to 3.5dB for 4 packets with coding.

Referring now to Figure 5, a block diagram of a decoder 500 used in a receiver system is shown. The decoder 500 includes circuitry that is adapted to use redundant packets, such as non-systematic packets in a data stream as described above, and to help decode the data received by the receiver. Also, the decoder 500 generally has the ability to decode data that has been encoded with the use of the A53 standard or existing.

In the decoder 500, after tuning, demodulation and processing by other circuits (Figure 4) a convolutional decoder 502 receives the incoming signal. The convolutional decoder 502 is connected to a convolutional deinterleaver 504. The output of the convolutional deinterleaver 504 is connected to a byte code decoder 506. The byte code decoder 506 has an output that connects to the Reed-Solomon decoder 508. The output of the Reed-Solomon decoder 508 is connected to a de-scrambler 510. The output of the de-scrambler 510 is connected to a data decoder 512. The data decoder 512 provides an output signal to be used in the remaining portion of the receiver system, such as the video display or audio playback.

In accordance with the existing or legacy A53 standard, the convolutional decoder 502 includes a signal demultiplexer, twelve 2/3 speed convolutional decoders, and a signal multiplexer. The demultiplexer distributes the digital samples between the twelve 2/3 speed convolutional decoders and the multiplexer multiplexes the estimates generated by each of the twelve 2/3 speed convolutional decoders. A de-interleaver 504, such as a convolutional interleaver, de-interleaves the current of the convolutionally decoded bit estimates, which produces sequences or packets arranged to include 207 bytes. The packet arrangement is carried out together with the determination and identification of the location of the synchronization signals, not shown. A Reed-Solomon error correction circuit 508 considers each 207-byte sequence produced by the de-interleaver 504 as one or more code words and determines if any byte in the codewords or packets were corrupted due to an error during transmission. The determination is often carried out when calculating and evaluating a group of syndromes or error patterns for the code words. When corruption is detected, the Reed-Solomon error correction circuit 508 attempts to recover the corrupted bytes with the use of the information encoded in the parity bytes. The resulting error corrected data stream is then descrambled by a de-scrambler 510 and then provided to a data decoder 512 that decodes the data stream in accordance with the type of data. content to be transmitted. Typically, the combination of the convolutional decoder 502, the deinterleaver 504, the decoder 508 Reed-Solomon and the descrambler 51 0 is identified as a decoder 8-VSB within a receiver. It is important to note, in general, that a typical receiver to receive signals compatible with the A53 standard bears the reception processing in the reverse order of the transmission process.

The data received, in the form of data bytes in the data packets, is decoded by the convolutional decoder 502 and deinterleaved by the de-interleaver 504. The data packets can contain 207 bytes of data and can also be de-interleaved. It can also be rupared in groups or in 24, 26 or 52 packages. The convolutional decoder 502 and the de-interleaver 504 have the ability to process the incoming legacy format data, as well as the encoded byte code data. Based on the predetermined packet transmission sequence that is known to the receiver, the byte code decoder 506 determines whether the packet is an embedded packet in a coded stream of robust data or byte code. When the received packet is not from the byte code encoded data stream then the received packet is provided to the Reed-Solomon decoder 508 without further processing in the byte code decoder 506. The byte code decoder 506 may also include a descrambler that removes the known sequence of constants multiplied or added to the data stream during coding. It is important to note that a reinforced data stream includes both systematic and bytes that are identical to the original data and non-systematic packets and bytes that contain redundant data.

When the byte code decoder 506 determines that the received one is a coded packet of byte code belonging to a reinforced or robust data stream, the packet can be decoded together with other packets comprising the same data stream. In one embodiment, the packets coded with byte code of the same data stream are decoded by multiplying each byte within the packet by the inverted value of the element that was used to develop the packet coded with bytes. The decoded values of the bytes of the non-systematic packet are compared with the values of the bytes of the systematic packets and the values of any byte in two packets that are not identical can be deleted (that is, set to zero) in the systematic packet and they can be replaced with the information in the non-systematic package. The systematic packet with erased error bytes can be decoded with the use of Reed-Solomon decoding carried out in the Reed-Solomon 508 decoder. A more detailed description of other modes of the byte code decoders will be found later.

The byte code decoder 506 may also be adapted to operate as a block coder to decode the coded signals, as shown in Figure 1. For example, the byte code decoder 506 may include a packet interleaver similar to the packet interleaver 120 and a packet deinterleaver similar to packet deinterleaver 130. In addition, the function of the encoder of Byte code can be adapted to decode a GF (256) series concatenated block encoded signal (SCBC). The byte code decoder 506 may also include a block identifier used to identify the encoded data for mobile reception or ATSC / H and / or the identification of an a-priori training package. In addition, the identifier block may include a packet identifier block to determine, for example, whether the headers in the incoming packets include a PID used for mobile reception.

It is important to note that in a preferred encoder, the byte code coding precedes the Reed-Solomon coding of the data packets. However, in the decoder 500 shown here, the incoming data is decoded with byte code before being decoded with Reed Solomon. Re-ordering is possible since the byte code operation and the Reed Solomon code operation are linear over the Galois field (256) used in the A53 standard, and the linear operators are switched in a Galois field. It is advantageous to do the block decoding first before the Reed-Solomon since there are soft decoding algorithms that make it practical to have an iterative decoding algorithm. The importance of re-ordering is that the coding with byte code provides the soft decoding algorithm, which then makes possible the iterative decoding or the turbo decoding, which has higher reliability to recover errors in the received signal. As a result, performing byte code decoding before Reed Solomon decoding results in improved receiver performance as measured in terms of rate of bit error and the signal to noise ratio.

Referring now to Figure 6, a block diagram of another mode of the decoder 600 used in a receiver is shown. The decoder 600 includes additional circuitry and processing to receive and decode signals that have been affected by the transmission of the signal on the transmission medium, such as electromagnetic waves in the air. The decoder 600 has the ability to decode a reinforced data stream and a legacy data stream.

In the decoder 600, the incoming signal, followed by an initial processing, is provided to the equalizer 606. The equalizer 606 is connected to the convolutional decoder 610, which provides two outputs. The first output from the convolutional decoder 610 provides the feedback and is connected as a feedback input to the equalizer 606. The second output of the convolutional decoder 610 is connected to a convolutional deinterleaver 614. The convolutional deinterleaver 614 is connected to a byte code decoder 616, which also provides two outputs. A first output from the byte code decoder 616 is connected from the rear as a feedback input for the convolutional decoder 610 through the convolutional interleaver 618. The second output of the byte code decoder 616 is connected to the Reed-Solomon decoder 620. The output of the Reed-Solomon decoder 620 is connected to the scrambler 624. The output of the descrambler 624 is connected to a data decoder 626. The Reed Solomon 620 decoder, the descrambler 624 and data decoder 626 are connected, and functionally operate, in a manner similar to Reed-Solomon, to descrambler and data decoder blocks described in Figure 5 and will not be described in more detail here.

An input signal from the main end processing (eg antenna, tuner, demodulator, A / D converter) of the receiver (not shown) is provided to the equalizer 606. Equalizer 606 processes the received signal to completely or partially remove the effect of the transmission channel in an attempt by recovering the received signal. The different methods of removal or equalization are well known to those skilled in the art and will not be described in more detail. Equalizer 506 may include multiple sections of the processing circuitry that include the direct feed equalizer (FFE) section and the decision feedback equalizer (DFE) section.

The equalized signal is provided to the convolutional decoder 610. The convolutional decoder 610 produces, as an output, a group of decision values that are provided to the DFE section of the equalizer 606. The convolutional decoder 610 'can also generate intermediate decision values that are also provided to the DFE section of the equalizer 606. The DFE section uses the decision values together with the intermediate decision values from the convolutional decoder 610 to adjust the values of the filter taps in the equalizer 606. The adjusted filter tap values cancel out the interference and signal reflections that are present in the signal received. The iterative process allows the equalizer 606, with the help of feedback from the convolutional decoder 610, to dynamically adjust the conditions of the potentially changing signal transmission environment over time. It is important to note that the iterative process can occur at a rate similar to the speed of incoming data of the signal, such as 19 Mb / s for a digital television broadcast signal. The iterative process can also occur at a higher speed than the incoming data rate.

The convolutional decoder 610 also provides a convolutionally decoded data stream to the convolutional deinterleaver 614. The convolutional deinterleaver 615 operates in a manner similar to the de-interleaver described in FIG. 5 and generates de-interleaved bytes organized within data packets. The data packets are provided to a byte code decoder 5616. As described above, packets that are not part of a reinforced data stream are simply passed through the byte code decoder 616 for the Reed-Solomon decoder 620. When the byte code decoder 616 identifies a group of packets as part of a reinforced data stream, the byte code decoder 616 uses the redundant information in the non-systematic packets to initially decode the bytes in the packets, as described before.

The byte code decoder 616 and the convolutional decoder 610 operate in an iterative manner, referred to as a turbo decoder to decode the enhanced data stream. Specifically, the Convolutional decoder 610 provides, after deinterlacing by a convolutional deinterleaver 614, a first soft decision vector to the byte code decoder 616 for each byte of the packets that are included in the reinforced data stream. Typically, the convolutional decoder 610 produces the soft decision as a vector of probability values. In some embodiments, each probability value in the vector is associated with a value that the byte associated with the vector may have. In other embodiments, the vector of the probability values is generated for each half-pair (ie, two bits) that are contained in the systematic packet, since the 2/3 speed convolutional decoder calculates two-bit symbols. In some embodiments, the convolutional decoder 610 combines four soft decisions associated with four half-pairs of a byte to produce a smooth decision that is a vector of the probabilities of values that the byte may have. In such embodiments, the soft decisions corresponding to the byte are provided to the decoder 616 of byte code. In other embodiments, the byte code decoder separates the soft decision with respect to one byte from the systematic packet into four soft decision vectors, wherein each of the four soft decisions is associated with a half-pair of the byte.

The byte code decoder 616 uses a soft decision vector associated with the bytes comprising packets of the reinforced data stream to produce a first estimate of the bytes comprising the packet. The byte code decoder 616 uses both systematic and non-systematic packets to generate a second soft decision vector for each byte of packets comprising the reinforced stream and providing the second soft decision vector to the convolutional decoder 610, after re-interleaving by the convolutional interleaver 618. The convolutional decoder 610 then uses the second soft decision vector to produce another iteration of the first decision vector, which is provided to the decoder 616 of byte code. The convolutional decoder 610 and the byte code decoder 616 iterate in this manner until the soft decision vector produced by the convolutional decoder and the byte code decoder converge or when a number of predetermined iterations are taken. After, the byte code decoder 616 uses the probability values in the soft decision vector for each byte of the systematic packets to generate a hard decision for each byte of the systematic packets. The hard decision values (ie, the decoded bytes) are output from the byte code encoder 616 to the Reed Solomon decoder 620. The convolutional decoder 610 can be implemented with the use of a maximum decoder a posteriori and can operate in soft decision byte or half-pair (symbol).

It is important to note that turbo decoding typically uses the iteration rates related to the passage of decision data between blocks that are higher than the incoming data rates. The number of possible iterations is limited to the rate of the data rate and the rate of iteration. As a result and for a practical purpose, the highest iteration rate in the turbo decoder Usually, it improves the error correction results. In one mode, the iteration rate is 8 times the speed of the incoming data that can be used.

A soft-entry soft-start byte code decoder, such as that described in Figure 6, may include vector decoding functions. Vector decoding involves grouping bytes of data including systematic and non-systematic bytes. For example, for a coded stream of ½ speed byte code, a systematic byte and an unsystematic byte will be grouped. The two bytes have about 64,000 possible values. The vector decoder determines or estimates a probability for each of the possible two-byte values and creates a probability map. A soft decision is made based on the probability weighting of some or all of the possibilities and the distance Euclidean for a possible code word. A hard decision can be made when the Euclidean distance error is below a threshold.

The byte code decoders, as described in FIGS. 5 and 6, can decode a reinforced data stream that has been encoded by the byte code encoders, as described above, including encoding by simple code encoders. byte or concatenated byte code encoders. The byte code decoders of Figures 5 and 6 describe the decoding of a reinforced data stream encoded by a single or composite byte code encoder that involves only a single coding step. The code decoding of Concatenated byte includes decoding incoming code words or bytes in more than one decoding step in addition to intermediate processing, such as deinterlacing, non-hopping and re-inserting.

With reference to Figure 7, an exemplary diffusion environment 700 in accordance with the present invention is shown. A first transmitter 720, a second transmitter 720 and a mobile receiver 730 are shown. The first transmitter 710 is located at a distance d1 from the mobile receiver 730 and the second transmitter 720 is located at a distance d2 from the mobile receiver 730.

By taking advantage of the redundant and separable codewords, a first codeword may be transmitted from the first transmitter 710 and a second codeword may be transmitted from a second transmitter 720. This reduces the occurrence of a total signal loss by varying the trajectories and propagation angles. These variations reduce the probability of total units of signal or of multiple trajectories of destruction. The signal reception can also be improved by transmitting each code word at a different frequency and / or at a different time.

This modality of spatial and frequency diversity can utilize the inherent "white space" between existing broadcast channels in a coverage area, while not increasing the complexity of the current receiver's equalizer. Such a proposed mode is particularly appropriate, although it is not limited to burst mode transmission as is now advised for advanced broadcast transmission systems. In the burst mode transmission, a single tuner in the receiver can receive a full transmission when receiving the non-simultaneous content at multiple frequencies. Full reception can be achieved by receiving only one of two or more bursts supplied from more than one source, including the main and secondary transmitters. The signal tuner can be maintained through a number of known techniques, including techniques already used for SFN.

With reference to Figure 8, there is shown an exemplary embodiment of a portion of a transmitter 800 in accordance with the present invention. The figure below illustrates an arrangement for a specific implementation of an improved code of an interleaved modeling structure interleaved with depth in the physical layer of a signal transmitting system. The implementation in a receiver can result in similar structures re-arranged to decode and demodulate as opposed to coding and modulation. During the operation, the process involves identifying and generating information for the alternate or redundant modeling operation. This content is received through channel output 805. The content is then provided to an encoder containing two or more parallel coding branches suitable for nominally generating a standard encoded burst mode signal. Then, each coding branch processes its supplied portion of the signal. Then, a branch is delayed by a RAM delay 815 by a predetermined amount. The amount of delay may represent a number of cycles of a signal, such as the tuner signal, and may also delay an equivalent time of a subsequent or future burst transmission time. Each signal in each branch is coded in its respective input stages 810, 820. The input stages may comprise legacy deinterlacing and packet deinterlacing. For time diversity signals, the non-delayed coding branch signal may be combined 825 with the previously encoded and delayed branch signal containing a portion of the previously processed data content and the combination provided to the remaining portions of the transmitter. The combined data is then decoded with block 830 and 840 is separated into different output stages. Each output stage 845, 850 may combine the legacy interleaving and the legacy encoding. The processes for the receiver can be essentially similar and mainly inverted from the processes for a transmitter.

For the transmission of frequency diversity and spatial diversity, the content is coupled from the channel input, before or after the RAM 815 and optionally, it is coupled with another RAM delay 855. The content can then be coupled with other channels for the transmission comprising the operations as shown with respect to Figure 1. An advantage of space diversity and frequency transmitters is that they use cooperative transmission. In this example, two television diffusers or a diffuser with two transmitters or frequencies, can transmit a first broadcast packet in a transmitter and the cooperative broadcast redundant packet. In this way, each diffuser transmits 2 packets, a burst and a redundant burst, but gains the advantage of diversity by having its package redundant transmitted on another frequency and / or on another transmitter simply by transmitting the redundant burst of the cooperative diffuser. Cooperative transmission provides the benefit of full frequency diversity and possibly spatial diversity without increasing each data output from cooperative transmitters or bandwidth.

In addition, from the inherent benefit associated with the time interleaving of the data content and the relationship with the operation in the burst mode transmissions, each branch of data enhanced with modified code can be transmitted the separate frequencies. In this way, frequency diversity, in addition to the diversity of time, can be achieved. For example, a first burst containing a portion of the alternating modeled content, after the code improvement, can be provided or transmitted to a particular broadcast channel by a first radio-diffuser. A second burst containing a remaining portion of the alternating modeled content, after the code improvement, can be provided or transmitted at a later point in time or in a second broadcast channel, possibly by a second broadcaster. Each diffusion channel represents a different frequency spectrum of the operation. The resulting operation also guarantees the retrieval of the content with original data by introducing the frequency diversity operation into the already inherent time diversity system.

While the present invention has been described in terms of a specific embodiment, it should be appreciated that modifications may be made that fall within the scope of the invention. By For example, several processing steps can be implemented separately or combined, and can be implemented in dedicated data processing or general purpose hardware. In addition, different coding and compression methods can be used for video, audio, images, text or other types of data. Also, packet sizes, speed modes, block coding and other parameters of information processing may vary in different embodiments of the invention.

Claims (14)

1. A method for encoding data, characterized in that it comprises the steps of: coding the data to generate a packet and a redundant packet, where each packet comprises the data; Y attach the package and the redundant package with the transmitter.

2. The method according to claim 1, characterized in that the packet and the redundant packet are coupled with different transmitters, where the different transmitters are spatially diverse.

3. The method according to claim 1, characterized in that the packet and the redundant packet are transmitted at different frequencies.

4. The method according to claim 1, characterized in that the packet and the redundant packet are transmitted at different times.

5. A method for receiving a signal characterized in that it comprises the steps of: receive a first package; receive a second packet, wherein the second packet is a redundant version of the first packet; combine the first and second packages; decode the first and second packets.

6. The method according to claim 5, characterized in that the first packet is received at a first frequency and the second packet is received at a second frequency.

7. The method according to claim 5, characterized in that the first packet is received at a first time, and the second packet is received at a second time.

8. An apparatus characterized because I understood: an encoder to encode data to generate a packet and a redundant packet; Y an interface for coupling the packet with a first transmitter and the redundant packet with a second transmitter.

9. The apparatus according to claim 8, characterized in that the packet is transmitted at a first frequency and the redundant packet is transmitted at a second frequency.

10. The apparatus according to claim 8, characterized in that the packet is transmitted to a first time and the redundant packet is transmitted to a second time.

11. An apparatus characterized in that it comprises: an interface for receiving a first packet and a second packet; a decoder for decoding the first packet and the second packet for generating a first decoded packet and a second decoded packet; a processor for combining the first decoded packet and the second decoded packet for generating a video signal.

12. The apparatus according to claim 11, characterized in that the second packet is a redundant version of the first packet.

13. The apparatus according to claim 11, characterized in that the first packet is received at a first frequency and the second packet is received at a second frequency.

14. The apparatus according to claim 11, characterized in that the first packet is received in a first time and the second packet is received in a second time.