JPS58120118A - Electromagnetic flowmeter - Google Patents

Abstract

PURPOSE:To improve the stability and response at a zero point, by sampling the voltages induced across electrodes for an exciting period and a rest period twice or more, and computing and compensating the noise components included in the induced voltages. CONSTITUTION:An electromagnetic flowmeter oscillator 2 is provided with an excitor coil 21, a pipe 22 wherein fluid flows, and the electrodes 23a and 23b. A constant current is supplied from a constant current power source 11 through switches 12a and 12b. The voltages ea induced across the electrodes 23a and 23b of the oscillator 2 are amplified 31, and its outputs eb are sampled 32. Its outputs eb are converted into digital signals 33. The voltages induced across the electrodes for the exciting period and for the rest period are sampled twice or more, respectively, and the noise components accompanied by the switching of the exciting current are computed. Then the noise components included in the voltages induced across the electrodes during the exciting period are compensated.

Description

【発明の詳細な説明】 本発明は、低周波励磁方式の電磁流量計の改良に関する
。DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an improvement in a low frequency excitation type electromagnetic flowmeter.

一般に電磁流量計は、流体の流れ方向に対して垂直に磁
界を与え、同時に流体流路中の電気的信号の変化を検出
し、これに基づいて流体の流量を計測する↓うに構成さ
れている。最近の電磁流量計は、父流励磁方式や直流励
磁方式に比して零点の安定性にすぐれている台形波励磁
や方形波励磁々どと呼ばれている低周波励磁方式のもの
が多く用いられている。低周波励磁方式の電磁流量計で
は、励磁コイルに供給する電流を２つの定常値間で周期
的に切換えて、励磁電流が一定になったとき電極間に発
生する誘起電圧をそれぞれ１回づつサンプリングした後
隣り合ったサンプリング信号の差をとることにより、電
気化学的な直流電圧や回路に基づくオフセット電圧によ
る影響を除去し、流体の流量に対応した信号を得ている
。このような低周波励磁方式の電磁流量計においても、
励磁電流が一定値に達してから十分な時間が経過した後
サンプリングしないと零点がドリフトする。これは電極
間に発生する誘起電圧に、流体の流量に比例した信号成
分と電気化学的な直流電圧や回路によるオフセット電圧
の外に、励磁電流の切換時に電極と電極リード間のルー
プで生ずる電磁結合ノイズと流体中を流れる渦電流が液
抵抗と電極の電磁結合ノイズと渦電流ノイズとは励磁電
流を切換えるたびに極性が反転するので、隣り合うサン
プリング信号の差をとっても消去できず、しかも電磁結
合ノイズは短時間で零になるが、渦電流ノイズは十分に
時間が経過しないと零にならないためである。よって、
零点の安定性の面から考えると励磁周波数は低いほど有
利であり、実用化されている電磁流量計には商用電源周
波数の＋／３２ｉＣ選ばれているものもある。ところが
励磁周波数をあまり低くすると応答性が遅くなったシ、
制御ループを組んだときハンチングを生じたシする。さ
らに励磁周波数を低くすると、電気化学的な直流電圧の
変化が問題となり、この変化を補償するだめの手段が新
たに必要となる。Generally, an electromagnetic flowmeter is configured to apply a magnetic field perpendicular to the direction of fluid flow, simultaneously detect changes in electrical signals in the fluid flow path, and measure the fluid flow rate based on this. . Most modern electromagnetic flowmeters use low-frequency excitation methods, such as trapezoidal wave excitation and square wave excitation, which have superior zero point stability compared to father-current excitation and DC excitation methods. It is being In a low-frequency excitation type electromagnetic flowmeter, the current supplied to the excitation coil is periodically switched between two steady-state values, and when the excitation current becomes constant, the induced voltage generated between the electrodes is sampled once each. After that, by taking the difference between adjacent sampling signals, the effects of electrochemical DC voltage and circuit-based offset voltage are removed, and a signal corresponding to the fluid flow rate is obtained. Even in such a low frequency excitation type electromagnetic flowmeter,
If sampling is not performed after a sufficient period of time has passed after the excitation current reaches a certain value, the zero point will drift. This is due to the induced voltage generated between the electrodes, a signal component proportional to the fluid flow rate, an electrochemical DC voltage, an offset voltage caused by the circuit, and an electromagnetic component generated in the loop between the electrode and electrode lead when the excitation current is switched. Coupling noise and eddy current flowing in the fluid Electromagnetic coupling noise and eddy current noise between liquid resistance and electrodes reverse polarity every time the excitation current is switched, so they cannot be canceled even if the difference between adjacent sampling signals is taken, and the electromagnetic This is because although coupled noise becomes zero in a short time, eddy current noise does not become zero until a sufficient amount of time has elapsed. Therefore,
Considering the stability of the zero point, the lower the excitation frequency is, the more advantageous it is, and some electromagnetic flowmeters in practical use use +/32iC of the commercial power frequency. However, if the excitation frequency was too low, the response became slow.
Hunting occurs when a control loop is constructed. Furthermore, when the excitation frequency is lowered, electrochemical changes in DC voltage become a problem, and new means are required to compensate for these changes.

本発明は、励磁コイルに励磁電流を流す励磁期間と励磁
電流を流さない休止期間を設け、励磁期間および休止期
間に電極間に誘起する電圧をそれぞれ２回以上サンプリ
ングして励磁電流の切換えに伴うノイズ成分を算出Ｌ、
励磁期間に電極間に誘起する電圧に含まれるノイズ成分
を補償することによって、零点の安定性および応答性に
すぐれた低周波励磁方式の電磁流量計を実現したもので
ある。The present invention provides an excitation period in which an excitation current is passed through the excitation coil and a rest period in which no excitation current is passed, and the voltage induced between the electrodes is sampled two or more times each during the excitation period and the rest period. Calculate the noise component L,
By compensating for the noise component included in the voltage induced between the electrodes during the excitation period, a low-frequency excitation type electromagnetic flowmeter with excellent zero point stability and responsiveness has been realized.

第１図は本発明電磁流振計の一実施例を示す接続図であ
る。図において、１は励磁回路で、直流定電流源１１と
、定電流源１１からの一定電流工を切換えるスイッチ１
２ａ、１２ｂとを有している。２は電磁流量計発信器で
、励磁コイル２１．流体が流れるパイプ２２および電極
２３ａ、２３ｂを備えている。３は信号処理回路で、電
磁流量計発信器２の電極２３゜２３ｂ間に誘起する電圧
ｅａを増幅する交流増幅器３１と、増幅器３１の出力ｅ
ｂをサンプリングするスイッチ３２と、スイッチ３２で
サンプリングされた増幅器出力ｅ、をディジタル信号に
変換するＡ／Ｄ変換器３３と、Ａ／Ｄ変換器３３からの
ディジタル信号に基づいて所望のディジタル演算を行う
マイクロプロセッサ３４と、マイクロプロセッサ３４の
出力をアナログ（３） 信号に変換するＤ／Ａ変換器３５と、Ｄ／Ａ変換器３５
の出力をサンプルホールドし出力電圧ｅ０を発生するサ
ンプルホールド回路３６とを有している。マイクロプロ
セッサ３４はディジタル演算を行うとともに、励磁回路
１のスイッチ１２ａ、１２ｂを駆動するパルスＰｌａ”
ｌｂ　’サンプリングスイッチ３２およびサンプルホー
ルド回路３６を制御するパルスｐ２１　Ｐ３を発生する
。FIG. 1 is a connection diagram showing an embodiment of the electromagnetic current vibration meter of the present invention. In the figure, 1 is an excitation circuit, and a switch 1 switches between a DC constant current source 11 and a constant current source from the constant current source 11.
2a and 12b. 2 is an electromagnetic flowmeter transmitter, which includes an exciting coil 21. It includes a pipe 22 through which fluid flows and electrodes 23a and 23b. 3 is a signal processing circuit, which includes an AC amplifier 31 that amplifies the voltage ea induced between the electrodes 23 and 23b of the electromagnetic flowmeter transmitter 2, and an output e of the amplifier 31.
a switch 32 for sampling the amplifier output e sampled by the switch 32, an A/D converter 33 for converting the amplifier output e sampled by the switch 32 into a digital signal, and a desired digital operation based on the digital signal from the A/D converter 33. a D/A converter 35 that converts the output of the microprocessor 34 into an analog (3) signal;
It has a sample and hold circuit 36 that samples and holds the output of and generates an output voltage e0. The microprocessor 34 performs digital calculations and also generates a pulse Pla'' that drives the switches 12a and 12b of the excitation circuit 1.
lb' Generates pulses p21 to P3 that control the sampling switch 32 and sample hold circuit 36.

このように構成した本発明の動作を第２図の波形図を参
照して以下に説明する。まずスイッチ１２ａ、１２ｂは
第２図（イ）、（ロ）に示す如き駆動パルスｐ　Ｐ　で
制御され、Ｐｌａがオンとなっている期ｌａ’　　　ｌ
ｂ 間Ｔには定電流源１１から電流を正方向に、Ｐｌｂがオ
ンとなっている期間Ｔ４には定電流源１１からの電流Ｉ
を逆方向に切換えて励磁コイル２１に流し、ｐｐ　　が
共にオフと々っている期間Ｔ□＋　Ｔ３にはｌａ’　　
　ｌｂ 励磁コイル２１に電流を流さない。よって励磁コイル２
１には第２図（ハ）に示すように定常値が零の休止期間
ＴＴと、正の期間Ｔおよび負の期間Ｔ４を有ＩＩ　　３
　　　　　　　　　　　　　２する励磁電流工が供給さ
れる。なお各期間Ｔｉ、Ｔ２゜（４） Ｔ３．Ｔ４はそれぞれ商用交流電源周期の整数倍に選ば
れており、また励磁電流工、はスイッチ１２ａ、１２ｂ
で切換えられたとき、励磁コイル２１のインダクタンス
と抵抗による時定数で実際には立上シ、立下り部分で遅
れを伴ったのち定常値となるが図では省略しである。電
磁流置引発信器２の電極２３ａ＋２３ｂ間には第２図に
）に示すように励磁電流工、に応じた誘起電圧ｅが発生
する。誘起電圧ｅａには、パイブ２２を流れる流体の流
量Ｆに比例した信号成分ｖ８の外に、励磁電流の切換え
に伴うノイズ成分ｖｎと、電気化学的な直流電位や回路
によるオフセット電圧成分Ｖ。、とが重畳されている。The operation of the present invention configured in this way will be explained below with reference to the waveform diagram of FIG. 2. First, the switches 12a and 12b are controlled by drive pulses p P as shown in FIGS.
During the interval T, the current flows from the constant current source 11 in the positive direction, and during the period T4 when Plb is on, the current I flows from the constant current source 11.
is switched in the opposite direction to flow through the excitation coil 21, and during the period T□+T3 when both pp are off, la'
lb No current is applied to the excitation coil 21. Therefore, excitation coil 2
1 has a rest period TT in which the steady-state value is zero, a positive period T and a negative period T4, as shown in FIG. 2 (c).
2. Exciting electric current is supplied. Note that each period Ti, T2゜(4) T3. T4 is selected to be an integer multiple of the commercial AC power supply period, and the excitation currents are selected as switches 12a and 12b.
When the current is switched, there is actually a delay in the rising and falling portions due to the time constant due to the inductance and resistance of the excitation coil 21, and then the steady value is reached, but this is not shown in the figure. As shown in FIG. 2, an induced voltage e is generated between the electrodes 23a and 23b of the electromagnetic field oscillator 2 in accordance with the excitation current. In addition to a signal component v8 proportional to the flow rate F of fluid flowing through the pipe 22, the induced voltage ea includes a noise component vn associated with switching of the excitation current, and an offset voltage component V due to electrochemical DC potential and circuits. , are superimposed.

ノイズ成分Ｖは、励磁電流の切換時に電極と電極リード
間のループで生ずる電磁結合ノイズと、流体中を流れる
渦電流が液抵抗Ｒと電極の界面電気二重層容量Ｃとで形
成される一次遅れ回路によって生ずる渦電流ノイズを含
んでいる１、その結果第２図に）に斜線で示すように誘
起電圧ｅを各期間で２回つつすンプリングしたときのサ
ンプリング電圧ｅａ１〜ｅ　は１サイクルの間流体の流
僅か変化しないと８ すると、それぞれ次式で与えられる。The noise component V is the electromagnetic coupling noise generated in the loop between the electrode and the electrode lead when switching the excitation current, and the first-order lag caused by the eddy current flowing in the fluid formed by the liquid resistance R and the interfacial electric double layer capacitance C of the electrode. 1, which includes eddy current noise generated by the circuit, and as a result, the sampling voltages ea1 to e when sampling the induced voltage e twice in each period are as shown by the diagonal lines in Figure 2) during one cycle. If the fluid flow does not change slightly, 8, then each is given by the following equation.

励磁電流が一定のときほぼ指数関数的（ｅｏＲ）に減少
していく。そこで各期間のサンプリング間隔Δｔを一定
とすれば、ｖｎｌ、ｖｎ２．ｖｎ３．ｖｎ４の間にはた
とえ励磁電流切換えに伴う立下がり、立上がり波形が異
っても次式の関係が成立する。When the excitation current is constant, it decreases almost exponentially (eoR). Therefore, if the sampling interval Δt of each period is constant, vnl, vn2. vn3. During vn4, even if the falling and rising waveforms due to excitation current switching are different, the following relationship holds true.

Δｔ よって、信号処理回路３で次式の演算を行えば、励磁電
流の切換えに伴うノイズ成分ｖｎＩＩｖｎ３．オフセッ
ト成分Ｖ　およびＫに相当する補償値ｅｎｌ　’ｆ ”ｎ３’　ｅｏｆ”を算出できる。Δt Therefore, if the signal processing circuit 3 calculates the following equation, the noise component vnIIvn3. A compensation value enl 'f ``n3'eof'' corresponding to the offset components V and K can be calculated.

しだがって、サンプリング電圧ｅａ３”ａ４’　”ａ７
’　ｅａ８をサンプリングする毎に次式の演算を行えは
、とな９、オフセット電圧成分ｖｏｆおよび励磁電流の
切換えに伴うノイズ成分ｖｎ３．Ｋｖｎ３の影響を除去
でき、流体の流量に比例した信号成分Ｖのみを得（７） ることかできる。彦おオフセット電圧成分Ｖ。、に相当
する補償値ｅ。ｆをあらかじめ算出しない場合には次式
の演算を行えばよい。Therefore, the sampling voltage ea3"a4'"a7
' Calculate the following equation every time ea8 is sampled.9, the offset voltage component vof and the noise component vn3 due to switching of the excitation current. The influence of Kvn3 can be removed, and only the signal component V proportional to the fluid flow rate can be obtained (7). Hikoo offset voltage component V. , the compensation value e corresponding to . If f is not calculated in advance, the following calculation may be performed.

図の信号処理回路５では、電磁流量計発信器２からの誘
起電圧ｅを増幅器３１で増幅した後、第２図（ホ）に示
す如き一定間隔Δｔで発生するサンプリングパルスｐ２
で駆動されるサンプリングスイッチ３２によって、第２
図に）に斜線で示すｅのサンプリングミ圧ｅａ工〜ｅＩ
ＩＬ８に相当する増幅器３１の出力ｅｂ□〜ｅｂ８が順
次Ａ／Ｄ変換器３３に与えられ、ディジタル信号に変換
されてマイクロプロセッサ３４に与えられる。マイクロ
プロセッサ３４はＡ／Ｄ変換器３３から与えられるディ
ジタル信号を用いて、まず（５）式に相当するディジタ
ル演算を行い補償値を算出しておき、この補償値を用い
て励磁電流が流れている期間Ｔ２Ｔ　Ｔ４に得られるサ
ンプリング電圧ｅａ３’（８） ｅａ４’　”ａ７’　ｅａ８に相当するディジタル信号
が入力される毎にディジタル演算によって（４）式およ
び（５）式に相当する補正演算を行い、オフセット電圧
成分ｖｏｆおよびノイズ成分ｖｎを除去し、流体の流量
のみに比例した信号成分Ｖに相当するディジタル値を順
次出力する。なおＡ／Ｄ変換器３３からマイクロプロセ
ッサ３４に入力されたサンプリング電圧ｅａ１〜ｅａ８
に相当するディジタル信号はそれぞれ専用のレジスタに
格納され、次のサイクルの信号が入力されるまでその値
がホールドされている。また算出した補償値も専用のレ
ジスタに格納されており、その値は休止期間Ｔ１．Ｔ３
に得られるサンプリング電圧ｅａ１．’　”ａ２”ａ５
’　”ａ６に相当するディジタル信号が入力される毎に
（３）式に相当する演算が行われ更新される。この場合
ｅｎ１とｅｎ３とｋの補償値として過去からの移動平均
値を用いると演算精度を上げることができる。マイクロ
プロセッサ３４の出力はＤ／Ａ変換器３５でアナログ信
号に変換され、第２図（へ）に示すタイミングで発生す
るパルスＰ３によってサンプルホールド回路３６に順次
与えられる。In the signal processing circuit 5 shown in the figure, after the induced voltage e from the electromagnetic flowmeter transmitter 2 is amplified by the amplifier 31, sampling pulses p2 are generated at regular intervals Δt as shown in FIG.
The sampling switch 32 driven by the second
Sampling pressure of e indicated by diagonal lines in the figure) ~ eI
The outputs eb□ to eb8 of the amplifier 31 corresponding to IL8 are sequentially applied to the A/D converter 33, converted into digital signals, and applied to the microprocessor 34. Using the digital signal given from the A/D converter 33, the microprocessor 34 first performs a digital operation corresponding to equation (5) to calculate a compensation value, and uses this compensation value to determine whether the excitation current flows. Each time a digital signal corresponding to the sampling voltage ea3' (8) ea4'``a7'' ea8 obtained during the period T2T T4 is input, correction calculations corresponding to equations (4) and (5) are performed by digital calculation. , the offset voltage component vof and the noise component vn are removed, and digital values corresponding to the signal component V proportional only to the fluid flow rate are sequentially output. ea1~ea8
The digital signals corresponding to are stored in dedicated registers, and their values are held until the next cycle's signal is input. The calculated compensation value is also stored in a dedicated register, and the value is stored in the pause period T1. T3
The sampling voltage ea1. '``a2'' a5
' ``Every time the digital signal corresponding to a6 is input, the calculation corresponding to equation (3) is performed and updated.In this case, if the moving average value from the past is used as the compensation value of en1, en3, and k, the calculation The output of the microprocessor 34 is converted into an analog signal by the D/A converter 35, and is sequentially applied to the sample and hold circuit 36 by the pulse P3 generated at the timing shown in FIG.

その結果サンプルホールド回路３６の出力には、流体の
流量のみに比例しだ信号成分Ｖに相当する出力電圧ｅが
得られる。As a result, an output voltage e corresponding to a signal component V which is proportional only to the flow rate of the fluid is obtained at the output of the sample and hold circuit 36.

このように本発明においては、零点のドリフトの原因と
ガるノイズ成分を有効に除去しているので、励磁周波数
を低くせずにすなわち応答性を犠牲にすることなく零点
の安定性が得られる。In this way, in the present invention, the cause of zero point drift and the noise component are effectively removed, so zero point stability can be obtained without lowering the excitation frequency, that is, without sacrificing response. .

なお上述では、オフセラ）！圧成分Ｖ。、が一定の場合
を例示したが、電気化学的直流電位の変動によってＶ　
が−次間数で変化し、各サンプリンｆ グ期間毎にΔｖｏｆづつ増加する場合には、誘起電圧ｅ
のサンプリング電圧ｅａ１〜ｅａ８は、”ａｌ　＝　ｖ
ｎｌ　＋”ｏｆ となるが、信号処理回路３で実質的に次式の演算を行え
ば、ノイズ成分ｖｎよ”ｎ３　’　オフセット電圧成分
Ｖ。０．オフセット電圧成分の変化分Δ”ｏｆおよびＫ
に相当する補償値”ｎｌ’　ｅｎ３’　ｅｏｆ’　Ｊ”
ｏｆ”を算出できる。In addition, in the above, Offsera)! Pressure component V. , is constant; however, due to fluctuations in the electrochemical DC potential, V
When the induced voltage e
The sampling voltages ea1 to ea8 are "al = v
However, if the signal processing circuit 3 essentially calculates the following equation, the noise component vn becomes "n3' offset voltage component V. 0. Change amount of offset voltage component Δ”of and K
Compensation value corresponding to "nl'en3'eof'J"
of” can be calculated.

したがって、これら補償値と（６）式のサンプリング値
ｅａ３’　”ａ４”ａ７’　ｅａ８との間で次式の演算
を行えは、オフセット電圧成分およびノイズ成分を有効
に除去でき、流体の流量に比例した信号成分ｖｓ□〜ｖ
ｓ４のみを得ることができる。Therefore, by calculating the following equation between these compensation values and the sampling values ea3'``a4''a7' ea8 of equation (6), the offset voltage component and noise component can be effectively removed, and the result is proportional to the fluid flow rate. signal component vs□~v
Only s4 can be obtained.

また上述では、増幅器３１の出力ｅｂをサンプリングス
イッチ３２を介してＡ／Ｄ変換器３３に与える場合を例
示したが、第３図に示すように増幅器出力ｅｂを゛積分
器３７を介してＡ／Ｄ変換器３３に与えるようにしても
よい。この場合積分時間Ｔ６を商用電源周期の整数倍に
選べば電源周波数ノイズの影響を除去できる。なお第３
図においては、積分器３７として、抵抗ＲＩと、演算増
幅器ｏｐと、ＯＰの帰還回路に接続された積分用コンデ
ンサＣ□と、入力積分時間を制御するタイミングスイッ
チＴＳおよび積分開始直前にそれ以前の積分値をリセッ
トするリセットスイッチＲ８とを伽えたものが例示され
ている。さらに上述では励磁期間と休止期間にそれぞれ
２回サンプリングしているが、６回以上サンプリングし
ても同様にできる。また上述では、励磁波形を零・正・
零・負の繰り返９の場合を例示したが、正・負・零の繰
り返しや負・正・零の繰シ返しであってもよい。また上
述では、サンプリング間隔を尋間隔として例示したが、
必ずしも等間隔でなくてもよい。Further, in the above description, the case where the output eb of the amplifier 31 is supplied to the A/D converter 33 via the sampling switch 32 has been exemplified, but as shown in FIG. It may also be applied to the D converter 33. In this case, if the integration time T6 is selected to be an integral multiple of the commercial power supply cycle, the influence of power supply frequency noise can be removed. Furthermore, the third
In the figure, the integrator 37 includes a resistor RI, an operational amplifier OP, an integrating capacitor C□ connected to the feedback circuit of OP, a timing switch TS that controls the input integration time, and a An example is shown in which a reset switch R8 for resetting the integral value is removed. Furthermore, although sampling is performed twice in each of the excitation period and rest period in the above description, the same result can be achieved even if sampling is performed six times or more. In addition, in the above, the excitation waveform is set to zero, positive,
Although the case of zero/negative repetition 9 has been illustrated, it may be positive/negative/zero repetition or negative/positive/zero repetition. In addition, in the above, the sampling interval was exemplified as the fathom interval, but
They do not necessarily have to be equally spaced.

以上説明したように本発明においては、励磁コイルに励
磁電流を流す励磁期間と励磁電流を流さない休止期間を
設け、励磁期間および休止期間に電極間１ｃ誘起する電
圧をそれぞれ２回以上サンプリングして誘起電圧に含ま
れるノイズ成分を算出し、励磁期間に電極間に誘起する
電圧に含まれるノイズ成分を補償するようにしているの
で、零点の安定性および応答性にすぐれた低周波励磁方
式の電磁流量計が得られる。As explained above, in the present invention, an excitation period in which an excitation current is passed through the excitation coil and a rest period in which no excitation current is passed are provided, and the voltage induced between the electrodes 1c is sampled two or more times each during the excitation period and the rest period. The noise component included in the induced voltage is calculated and the noise component included in the voltage induced between the electrodes during the excitation period is compensated for. A flow meter is obtained.

【図面の簡単な説明】[Brief explanation of the drawing]

第１図は本発明電磁流量計の一実施例を示す接続図、第
２図はその動作波形図、第３図は本発明電磁流量計の他
の笑施例を示す接続図である。 １・・・励磁回路、２・・・電磁流量計発信器、２１・
・・励磁コイル、２３ａ、　２３ｂ・・・電極、３・・
・信号処理回路、３１・・・増幅器、Ｓ２・・・サンプ
リングスイッチ、３３・・・Ａ／Ｄ変換器、３４・・・
マイクロプロセッサ、３５・・・Ｄ／Ａ変換器、３６・
・・サンプルホールド回路、３７・・・積分器。FIG. 1 is a connection diagram showing one embodiment of the electromagnetic flowmeter of the present invention, FIG. 2 is an operating waveform diagram thereof, and FIG. 3 is a connection diagram showing another embodiment of the electromagnetic flowmeter of the present invention. 1... Excitation circuit, 2... Electromagnetic flowmeter transmitter, 21.
...Exciting coil, 23a, 23b...Electrode, 3...
- Signal processing circuit, 31... amplifier, S2... sampling switch, 33... A/D converter, 34...
Microprocessor, 35...D/A converter, 36.
...sample hold circuit, 37...integrator.

Claims (1)

【特許請求の範囲】[Claims] 電磁流量計発信器の励磁コイルに正の励磁電流および負
の励磁電流を流す励磁期間と、励磁電流を流さない休止
期間を設けた低周波励磁方式の電磁流量計において、励
磁期間および休止期間に電磁流量計発信器の電極間に誘
起する電圧をそれぞれ２回以上サンプリングして励磁電
流の切換えに伴うノイズ成分を算出し、励磁期間に電磁
流量計発信器の電極間に誘起する電圧に含まれる励磁電
流の切換に伴うノイズ成分を補償する演算を行う信号処
理回路を設けたことを特徴とする電磁流量計。In an electromagnetic flowmeter using a low frequency excitation method, which has an excitation period in which a positive excitation current and a negative excitation current are passed through the excitation coil of the electromagnetic flowmeter transmitter, and a rest period in which no excitation current is passed, The voltage induced between the electrodes of the electromagnetic flowmeter transmitter is sampled two or more times each to calculate the noise component associated with switching the excitation current, and the noise component included in the voltage induced between the electrodes of the electromagnetic flowmeter transmitter during the excitation period is calculated. An electromagnetic flowmeter characterized by being equipped with a signal processing circuit that performs calculations to compensate for noise components associated with switching of excitation current.