JPH0721520B2 - Insulation resistance measurement method with phase compensation - Google Patents

Description

【発明の詳細な説明】 （産業上の利用分野） 本発明は活線状態で電路等の絶縁抵抗を測定する装置の
温度変化域は回路特性の変化等に対する補償方法に関す
る。TECHNICAL FIELD The present invention relates to a method of compensating for a temperature change range of a device for measuring an insulation resistance of an electric circuit or the like in a live state, against a change in circuit characteristics.

（従来技術） 従来，漏電等の電路に於けるトラブルの早期発見の為に
例えば第４図に示す如き電路の絶縁抵抗測定方法を用い
電路状態を監視するのが一般的であった。(Prior Art) Conventionally, in order to detect a trouble in an electric circuit such as an electric leakage at an early stage, it is general to monitor the electric circuit condition by using a method for measuring an insulation resistance of the electric circuit as shown in FIG. 4, for example.

これはＺなる負荷を有する受電変圧器Ｔの第２種接地線
L_Eを，商用電源周波数とは異なる周波数_１なる測定用
低周波信号発振器OSCに接続されたトランスOTに貫通せ
しめるか，或いは前記接地線L_Eに直列に前記発振器を挿
入接続する等して電路１及び電路２に測定用低周波信号
電圧を印加し，前記接地線L_Eを貫通せしめた交流器ZCT
によって電路と大地間に存在する絶縁抵抗Ro及び対地浮
遊容量Coを介して前記接地線に帰還する前記測定用低周
波信号の漏洩電流を検出しこれを増幅器AMPで増幅した
のち，フィルタFILによって周波数_１の成分のみを選
択し，これを例えば前記発振器OSCの出力信号を用いて
掛算器MULTで同期検波して漏洩電流中の有効分即ち，印
加低周波電圧と同相の成分を検出することにより電路の
絶縁抵抗を測定するよう構成したものであった。This is the second type ground wire of the power receiving transformer T having a load of Z.
L _E is penetrated through a transformer OT connected to a measurement low-frequency signal oscillator OSC having a frequency ₁ different from the commercial power supply frequency, or the oscillator is inserted and connected in series with the ground line L _E to connect the electric circuit. AC transformer ZCT in which a low-frequency signal voltage for measurement is applied to circuit 1 and circuit 2 to penetrate the ground line L _E
The leakage current of the low-frequency signal for measurement that returns to the ground line through the insulation resistance Ro existing between the electric line and the ground and the stray capacitance Co to the ground is detected by the amplifier AMP Only the component of ₁ is selected, and this is synchronously detected by the multiplier MULT using the output signal of the oscillator OSC, for example, and the effective component in the leakage current, that is, the component in phase with the applied low frequency voltage Was configured to measure the insulation resistance of the.

本発明の理解を助けるためにその測定理論を更に説明す
る。The theory of measurement will be further explained to help understanding of the present invention.

前記接地線L_Eに印加される測定用低周波信号電圧を例え
ば正弦波としてEsinω₁t（ω_１＝２π_１）とすれば，
接地点Ｅを介して接地線L_Eに帰還する周波数_１の漏洩
電流Ｉは と表わされ，印加する交流電圧と同相の成分，即ち上記
（１）式の右辺第１項の成分に比例した値を同期検波等
の手段で検出すればこの値は絶縁抵抗Roに逆比例したも
のとなるから，これによって電路を絶縁抵抗値を求める
ことができる。しかしこのように前記接地線に帰還する
漏洩電流を交流器ZCTで検出し，更に交流器出力に含ま
れる周波数_１の漏洩電流成分をフィルタFILで選択出
力する従来の方法では，通常変流器→増幅器→フィルタ
の系で周波数_１の漏洩電流の位相がずれるから，これ
らの同期検波出力からR₀に逆比例した値を得るためには
この位相ずれを補償する必要がある。このために同図に
示す如く同期検波器MULTの第１の入力端又は第２の入力
端に移相器PSを挿入し，これによって上記位相ずれを補
正して互いの同期をとっていた。即ちこの移相器PSを設
けることにより対地浮遊容量Coがない状態（Co＝０）に
て，同期検波器の第1,第２の入力端に印加される電圧の
位相差が零となるように前もって設定しておくものであ
った。If the measurement low-frequency signal voltage applied to the ground line L _E is a sinusoidal wave and is Esinω ₁ t (ω ₁ = 2π ₁ ), then
The leakage current I of frequency ₁ which returns to the ground line L _E via the ground point E is If a value proportional to the in-phase component of the applied AC voltage, that is, the component of the first term on the right side of equation (1) above is detected by means such as synchronous detection, this value is inversely proportional to the insulation resistance Ro. As a result, the insulation resistance value of the electric path can be obtained. However, in the conventional method in which the leakage current returning to the ground line is detected by the AC transformer ZCT and the leakage current component of frequency ₁ contained in the AC transformer output is selectively output by the filter FIL, the current transformer Since the phase of the leakage current at frequency ₁ shifts in the amplifier → filter system, it is necessary to compensate for this phase shift in order to obtain a value inversely proportional to R ₀ from these synchronous detection outputs. For this reason, as shown in the figure, a phase shifter PS is inserted into the first input terminal or the second input terminal of the synchronous detector MULT, and the phase shift is corrected by this to synchronize with each other. That is, by providing this phase shifter PS, the phase difference between the voltages applied to the first and second input terminals of the synchronous detector becomes zero when there is no stray capacitance Co to ground (Co = 0). Was set in advance.

しかしながら上述の如き従来の位相補償方法では変流器
ZCT,フィルタFIL,移相器PS等の位相特性は温度変化また
は使用部品特性の経年変化等によって変動するため，こ
の結果最初の調整値との位相誤差が発生し，正しい測定
結果を提供できなくなる欠点があった。また更に変流器
は一次電流が大きくなると位相特性が変動する場合があ
るためこの影響によっても誤差が生ずる欠点があった。
これらに対処するために従来は特性変動の少ない極めて
高品質な変流器或いはフィルタ等を採用することによっ
て位相誤差の影響を極力小さくすると共に変流器の一次
電流を極力小さくするよう配慮していたが，それでもそ
の影響を完全に除去することは困難であった。However, in the conventional phase compensation method as described above, the current transformer
Since the phase characteristics of ZCT, filter FIL, phase shifter PS, etc. fluctuate due to temperature changes or changes in the characteristics of the parts used over time, etc., a phase error with the first adjustment value will occur, and correct measurement results cannot be provided. There was a flaw. Further, the current transformer may have a drawback that the phase characteristic may fluctuate when the primary current becomes large, which causes an error due to this influence.
In order to deal with these problems, it has been attempted to minimize the influence of the phase error and the primary current of the current transformer by using an extremely high-quality current transformer or filter, etc., which has less characteristic fluctuations. However, it was still difficult to eliminate the effect completely.

（発明の目的） 本発明は以上説明したような従来の絶縁抵抗測定方法の
欠点を除去するためになされたものであって，高価な部
品を必要とせず安価に測定信号の位相ずれを常時補正
し，常に正確な測定結果をもたらしうる絶縁抵抗測定方
法を提供することを目的とする。(Object of the Invention) The present invention has been made in order to eliminate the drawbacks of the conventional insulation resistance measuring method as described above, and always corrects the phase shift of the measurement signal inexpensively without requiring expensive parts. However, it is an object of the present invention to provide an insulation resistance measuring method that can always provide accurate measurement results.

（発明の概要） 本発明はこの目的達成のため，前記被測定電路と接地線
との接地点間に所定値のリアクタンス素子例えばコンデ
ンサを挿入すると共に，この接続を繰返し周期Ｔの信号
で断接をくり返すか，又は前記所定用低周波電圧と90゜
位相の推移した所定値の大きさの電流を繰返し周期Ｔの
信号で変化させ，この電流の流れる導線を変流器に貫通
させる等により接地線から漏洩電流を導出する手段に関
与せしめる。一方、変流器出力中の周波数f1の電流を抽
出し、第１及び第２の同期検波器の一方の入力端に入力
すると共に、第１同期検波器の他の入力端には前記測定
用低周波電圧を入力し、また第２同期検波器の他の入力
端には前記測定用低周波電圧を90゜移相せしめた電圧を
入力することにより、第１及び２同期検波器出力を得
る。更に第１の同期検波器の出力中に含まれる周波数1/
Tの成分を抽出し、前記繰り返し周期Ｔの信号を用いて
第３の同期検波器により同期検波することによって第３
の同期検波器出力を得、該第３の同期検波器出力が零と
なるように前記第１及び第２の同期検波器に入力する信
号の位相を調整すると共に、前記第３の同期検波器出力
を定数倍した信号と前記第２の同期検波器出力とを掛け
算して得た信号と、前記第１の同期検波器出力とを引き
算した出力を用いることにより対地浮遊容量の影響を除
去した電路の絶縁抵抗を測定する。(Summary of the Invention) In order to achieve this object, the present invention inserts a reactance element such as a capacitor having a predetermined value between the ground points of the circuit to be measured and the ground wire, and connects and disconnects this connection with a signal having a cycle T. By repeating the above, or by changing the current of a predetermined value with a 90 ° phase shift with the predetermined low frequency voltage by a signal of a cycle T, and passing the current-carrying wire through a current transformer. Involve in the means for deriving the leakage current from the ground wire. On the other hand, the current of the frequency f1 in the output of the current transformer is extracted and input to one of the input ends of the first and second synchronous detectors, and the other input end of the first synchronous detector is used for the measurement. A low frequency voltage is input, and a voltage obtained by phase-shifting the measuring low frequency voltage by 90 ° is input to the other input terminal of the second synchronous detector to obtain the outputs of the first and second synchronous detectors. . Furthermore, the frequency 1 / included in the output of the first synchronous detector
The third component is extracted by extracting the T component and performing the synchronous detection by the third synchronous detector using the signal of the repetition period T.
Of the synchronous detector, the phase of the signal input to the first and second synchronous detectors is adjusted so that the output of the third synchronous detector becomes zero, and the third synchronous detector is The influence of the ground stray capacitance is removed by using the signal obtained by multiplying the signal obtained by multiplying the output by a constant and the output of the second synchronous detector, and the output obtained by subtracting the output of the first synchronous detector. Measure the insulation resistance of the circuit.

（実施例） 以下図示した実施に基づいて本発明を詳細に説明する
が，その前に本発明の理解を助ける為従来の方法及びそ
の欠点を少しく詳細に説明する。(Examples) The present invention will be described in detail below based on the illustrated embodiments, but before that, a conventional method and its drawbacks will be described in a little more detail in order to facilitate understanding of the present invention.

第４図に於いて第（１）式にて示される周波数_１の漏
洩電流成分Ｉが変流器ZCT,増幅器AMP,フィルタFILの系
を通過する際発生する位相ずれをθとすればフィルタFI
L出力I₁は となり，これは同期検波器MULTの第１の入力端に印加さ
れる。In Fig. 4, if the leakage current component I of frequency ₁ shown in the equation (1) passes through the system of the current transformer ZCT, the amplifier AMP, and the filter FIL, and the phase shift is θ, then the filter FI
L output I ₁ is And this is applied to the first input of the synchronous detector MULT.

また同期検波器の第２の入力端に印加される電圧を例え
ば一定振幅のaosin（ω₁t＋θ_１）とすれば，同期検波
器の出力Ｄは 従ってθ＝θ_１のときの出力Doは となり,V,aoは一定となるから絶縁抵抗R₀に逆比例した
値を測定することができる。したがって位相ずれθ−θ
_１が零でない時の上記Doに対するＤの誤差Ｅは となる。If the voltage applied to the second input terminal of the synchronous detector is, for example, aosin (ω ₁ t + θ ₁ ) with a constant amplitude, the output D of the synchronous detector is Therefore, the output Do when θ = θ ₁ is Since V and ao are constant, a value that is inversely proportional to the insulation resistance R ₀ can be measured. Therefore, the phase shift θ−θ
The error E of D with respect to the above Do when ₁ is not zero is Becomes

しかしながら今，例えば位相ずれをθ−θ_１＝１（度）
とすれば（６）式にて_１＝25Hzで,R₀＝20KΩ,Co＝５
μＦとするときω₁CoRo15.7となるから誤差εは27.4
％となり著しく測定誤差が大きくなることが分る。However, now, for example, the phase shift is θ−θ ₁ = 1 (degree)
Then, in Equation (6), ₁ = 25 Hz, R ₀ = 20 KΩ, Co = 5
When μF is set, ω ₁ CoRo is 15.7, so the error ε is 27.4.
It can be seen that the measurement error is significantly large because of%.

本発明は上述の如き位相ずれに伴う誤差の発生を極力抑
える方法を提案するものである。The present invention proposes a method for suppressing the occurrence of an error due to the phase shift as described above as much as possible.

第１図は本発明に係る絶縁抵抗測定方法の一実施例を示
す回路図である。FIG. 1 is a circuit diagram showing an embodiment of the insulation resistance measuring method according to the present invention.

同図に於いて接地線L_Eに，位相制御回路PCで発生された
周波数_１なる低周波電圧を位相特性変動の小さい電力
増幅器PAMPで増幅した後トランスOTを介してVsinω₁tな
る電圧を電路に印加する。この際接地線L_Eに直列挿入す
るトランスOTの出力インピーダンスは十分に低く選ぶ。
又接地線L_Eには変流器ZCTを結合しその出力を，周波数
_１を含む成分を通しかつ商用周波成分を除去するフィ
ルタFILに印加することにより前記（２）式に相当する
出力を得，これを２つの同期検波器MULT1,MULT2夫々の
第１の入力端に印加する。In the same figure, a low frequency voltage of frequency ₁ generated by the phase control circuit PC is amplified by the power amplifier PAMP with small phase characteristic fluctuation, and then a voltage of Vsinω ₁ t is passed through the transformer OT to the ground line L _E. Apply to. At this time, the output impedance of the transformer OT that is inserted in series with the ground line L _E is selected to be sufficiently low.
Moreover, a current transformer ZCT is connected to the ground line L _E , and its output is
By applying to the filter FIL that passes the component including ₁ and removes the commercial frequency component, the output corresponding to the above equation (2) is obtained, and this output is applied to the first input terminals of the two synchronous detectors MULT1 and MULT2 respectively. Apply.

又接地線L_Eに並列にコンデンサＣをスイッチSWを介して
接続し，このスイッチSWの開閉を位相制御回路PCによっ
て制御せしめる。Also, a capacitor C is connected in parallel to the ground line L _E via a switch SW, and the opening / closing of this switch SW is controlled by the phase control circuit PC.

更に，これら２つの同期検波器MULT1及びMULT2夫々の他
方入力端には前記位相制御回路PCに於いて発生する低周
波信号_１を入力するが，うち同期権波器MULT2には90
゜移相器PSOを介挿することによって90゜位相をシフト
する。Further, the low frequency signal ₁ generated in the phase control circuit PC is input to the other input terminal of each of these two synchronous detectors MULT1 and MULT2, of which 90 is input to the synchronous power detector MULT2.
A 90 ° phase shift is achieved by inserting a ° phase shifter PSO.

このようにして得た２つの同期検波出力のうち,MULT1の
出力を２分し，一方を減算器SUBの一入力端に又，他方
をフィルタBPを介して第３の同期検波器MULT3の比較信
号となす。Of the two synchronous detection outputs thus obtained, the output of MULT1 is divided into two, one of which is used as one input terminal of the subtractor SUB and the other of which is compared with the third synchronous detector MULT3 via the filter BP. Make a signal.

又第３の同期検波器MULT3の基準信号としては前記位相
制御回路PCが発生するスイッチング信号出力を入力す
る。The switching signal output generated by the phase control circuit PC is input as the reference signal of the third synchronous detector MULT3.

更に，該MULT3の出力は係数回路COFを経て掛算器MTに入
力し該部に於いて前記第２の同期検波器MULT2の出力と
乗じその結果を前記減算器SUBの他方端に入力して所望
信号OUTを得る。Further, the output of the MULT3 is input to the multiplier MT via the coefficient circuit COF, multiplied by the output of the second synchronous detector MULT2 in the section, and the result is input to the other end of the subtractor SUB to obtain the desired result. Get the signal OUT.

又，接地線L_Eと並列に接続したコンデンサＣとスイッチ
SWの直列回路に於けるスイッチSWは前記位相制御回路PC
によってそのON−OFFを制御するよう構成する。この構
成に於いて以下その動作を説明する。Also, a capacitor C and a switch connected in parallel with the ground line L _E
The switch SW in the series circuit of SW is the phase control circuit PC
Is configured to control its ON-OFF. The operation of this configuration will be described below.

今，前記スイッチSWをオンした場合を考えれば接地線L_E
にはω₁CVcosω₁tなる電流が追加されて流れることにな
り，接地線に流れる印加低周波成分の漏洩電流Ioは となる。したがってフィルタFILの出力I₂は（２）式の
関係から となり，このときの同期検波器MULT1の出力D₁は，
（４）式の関係から となる。Now, considering the case where the switch SW is turned on, the ground wire L _E
The will flow been added ω ₁ CVcosω ₁ t becomes current, leakage current Io of the applied low-frequency component flows to the ground line Becomes Therefore, the output I _{2 of the} filter FIL is And the output D ₁ of the synchronous detector MULT1 at this time is
From the relationship of equation (4) Becomes

ここで，前記スイッチSWを周期Ｔ でオン・オフすれば，（９）式の第２項に含まれるＣの
値が周期Ｔで変るため同期検波器MULT1の出力D₁には周
波数1/Tの成分が生ずることになるこの際（９）式から
も分るようにθ＝θ_１のときは，第２項は零となるから
周波数1/Tの成分は発生しない。更にこの第１の同期検
波器MULT1の出力を周波数1/Tの成分のみをとり出す前記
フィルタBPに印加すれば該フィルタBPの出力Ａは と表すことができる。ここでｋは定数，はフィルタBP
の特性等から定まる位相である。Here, the switch SW is set to cycle T If turned on and off with, the value of C included in the second term of the equation (9) changes with the period T, so that the output D ₁ of the synchronous detector MULT1 has a frequency 1 / T component. As can be seen from the equation (9), when θ = θ ₁ , the second term becomes zero, so that the frequency 1 / T component does not occur. Further, if the output of the first synchronous detector MULT1 is applied to the filter BP which extracts only the component of frequency 1 / T, the output A of the filter BP is It can be expressed as. Where k is a constant and is a filter BP
It is a phase determined from the characteristics of.

次にこのフィルタBPの出力を第３の同期検波器MULT3の
一方の入力端に印加し，他の入力端に位相制御回路PCに
て発生するスイッチSWをオン・オフする周期Ｔの繰返し
信号を印加すれば，該同期検波器MULT3の出力Soは と表す事ができ，これは又 So＝−Kocos・sin（θ−θ_１） ……（12） となる。Next, the output of this filter BP is applied to one input end of the third synchronous detector MULT3, and the repetitive signal of the cycle T for turning on / off the switch SW generated in the phase control circuit PC is applied to the other input end. If applied, the output So of the synchronous detector MULT3 is It can be expressed as follows, which also becomes So ＝ －Kocos ・ sin (θ−θ ₁ ) …… (12).

ここで であり定数である。したがって｜｜＜π/2,|θ−θ₁|
〈π/2であるならばθ〉θ_１のときS₀＜0,又はθθ_１
のときSoとなりSoの入力された位相制御回路PCでは位
相制御信号Soを用いて位相の調整方向を判定することが
でき，この判定結果を用いて位相θ_１を調整してSoが零
に近づくように制御すればθ−θ_１→０に近づけること
ができる。この位相制御回路PCは既存の技術で実現でき
るので詳述を省略する。here And is a constant. Therefore, || <π / 2, | θ−θ ₁ |
<Θ / 2 if θ> θ ₁ , S ₀ <0, or θ θ ₁
At this time, So becomes and the phase control circuit PC to which So is input can judge the phase adjustment direction using the phase control signal So, and adjusts the phase θ ₁ using this judgment result to bring So close to zero. If it is controlled in this way, it is possible to approach θ−θ ₁ → 0. Since this phase control circuit PC can be realized by the existing technology, its detailed description is omitted.

ところで，同期検波器MULT2の第２の入力端に印加する
電圧は同期検波器MULT1の第２の入力端に印加する電圧a
osin（ω₁t＋θ_１）が90゜移相したものであるからこれ
を今aocos（ω₁t＋θ_１）とすれば，同期検波器MULT2の
出力ＨはスイッチSWがオフのとき， となる。またスイッチSWがオンのときのMULT2の出力H₂
は となる。By the way, the voltage applied to the second input terminal of the synchronous detector MULT2 is the voltage a applied to the second input terminal of the synchronous detector MULT1.
Since osin (ω ₁ t + θ ₁ ) is 90 ° phase-shifted, if this is now aocos (ω ₁ t + θ ₁ ), the output H of the synchronous detector MULT2 is: Becomes MULT2 output H ₂ when switch SW is on
Is Becomes

また，同期検波器MULT3の出力を係数回路COFで例えば すれば係数回路COFの出力は−sin（θ−θ_１）となる。
（上述の如くKo,cosは定数と考えて差しつかえないた
め） したがって同期検波器MULT2の出力と係数回路出力との
積をかけ算器MTで演算し同期検波器MULT1の出力とを引
算器SUBにて引算すればスイッチSWがオフのときの引算
器SUBの出力OUT1は（４），（13）式から 一方，スイッチSWがオンのときの引算器の出力OUT2は
（９），（14）式から となる。In addition, the output of the synchronous detector MULT3 can be converted to Then, the output of the coefficient circuit COF becomes −sin (θ−θ ₁ ).
(Because Ko and cos can be considered as constants as described above) Therefore, the product of the output of the synchronous detector MULT2 and the coefficient circuit is calculated by the multiplier MT and the output of the synchronous detector MULT1 is subtracted by the subtractor SUB. If subtracted with, the output OUT1 of the subtractor SUB when the switch SW is off is calculated from equations (4) and (13). On the other hand, the output OUT2 of the subtractor when the switch SW is on is calculated from the equations (9) and (14). Becomes

｜θ−θ₁|≪１となるように位相θ_１が前記方法で調整
されているならば,sin（θ−θ_１）（θ−θ_１）,cos
（（θ−θ_１）1,sin²（θ−θ_１）０とおけるから
（15），（16）式で表わされる出力OUTは となる。If the phase θ ₁ is adjusted by the above method so that | θ−θ ₁ | << ₁ , sin (θ−θ ₁ ) (θ−θ ₁ ), cos
((Θ−θ ₁ ) 1, sin ² (θ−θ ₁ ) 0, so the output OUT expressed by equations (15) and (16) is Becomes

一方，従来のように同期検波器MULT1の出力を単に用い
た場合｜θ−θ₁|≪１ならばスイッチオフのとき（４）
式から 又スイッチオンのとき（９）式から となり，位相誤差（θ−θ_１）の影響を受け大地静電容
量Coが大きいときθ−θ_１を十分に小さくしないかぎり
（17）式の出力OUTより誤差が大きくなることが分る。On the other hand, when the output of the synchronous detector MULT1 is simply used as in the past, if | θ−θ ₁ | << _1, then it is switched off (4)
From the formula When the switch is turned on, from equation (9) Next, the phase error (theta-theta ₁₎ affected unless sufficiently small theta-theta ₁ when the earth capacitance Co is large (17) of the output OUT from it can be seen that the error becomes large.

上記の如く本発明の位相補償方法によれば従来の方法よ
り位相誤差の影響を受けにくいことが明らかであろう。As described above, it is clear that the phase compensation method of the present invention is less affected by the phase error than the conventional method.

以上の説明では，位相調整のため位相制御信号Soを用い
たがこれに代る信号として次の信号を用いることもでき
る。In the above description, the phase control signal So is used for phase adjustment, but the following signal can be used as a signal in place of this.

即ち，前記Soと同期検波器MULT2の出力中に含まれる周
波数1/Tの周波数成分を別途設けたフィルタで検出し前
記繰返し周期Ｔの信号で別途同期検波することにより得
た信号との積をとるときにより得られる信号であっても
よい。更に別の方法として，前記同期検波器MULT1とMUL
T2の出力中に含まれる周波数1/Tの成分を検出し，それ
ぞれの積を求め，その直流分を用いても同様な結果が得
られるが詳述は省略する。That is, the product of So and the signal obtained by separately detecting the frequency component of frequency 1 / T contained in the output of the synchronous detector MULT2 by a filter provided separately and separately performing synchronous detection on the signal of the repetition period T is obtained. It may be a signal obtained at the time of taking. As yet another method, the synchronous detectors MULT1 and MUL
A similar result can be obtained by detecting the frequency 1 / T component contained in the output of T2, obtaining the product of each, and using the DC component, but a detailed description is omitted.

更に，上述の説明ではコンデンサＣを電路と接地点間に
挿入する場合を述べたが，本発明はこれに限定する必要
はなく例えば非接地電路と接地点間に挿入してもよい。
ただし，この場合はコンデンサＣに商用電源が印加され
るためコンデンサＣ及びスイッチSWに流れる電流は著し
く大きくなるから耐え得るものを使用する必要がある。Furthermore, although the case where the capacitor C is inserted between the electric line and the ground point has been described in the above description, the present invention is not limited to this, and may be inserted between the non-grounded line and the ground point, for example.
However, in this case, since commercial power is applied to the capacitor C, the current flowing through the capacitor C and the switch SW becomes extremely large, so it is necessary to use one that can withstand.

第２図は本発明の変形実施例を示す主要部分のブロック
図であり，前記コンデンサＣならびにスイッチSWを介し
て測定用低周波信号電圧より90゜位相の推移した電流を
変流器ZCTの一次側に流すために印加トランスOTの代り
に第２の２次巻線を設けたOT′を用い，これより得られ
る測定用低周波信号電圧を印加する如くしたものであ
る。この例によれば第１図の実施例の如く，接地線L_Eに
コンデンサC,スイッチSWを接続する必要がないため設置
工事を簡易化することができる。FIG. 2 is a block diagram of a main part showing a modified embodiment of the present invention, in which a current having a phase shift of 90 ° from the low frequency signal voltage for measurement is transferred to the primary side of the current transformer ZCT through the capacitor C and the switch SW. In order to flow to the side, an OT 'provided with a second secondary winding is used instead of the applying transformer OT, and a low frequency signal voltage for measurement obtained from this is applied. As examples of FIG. 1 according to this embodiment, the capacitor C to the ground line L _E, the installation work is not necessary to connect the switch SW can be simplified.

なお動作については，第１図の説明で述べたものと全く
同じである。The operation is exactly the same as that described in FIG.

第３図は他の実施例を示しており、コンデンサC,スイッ
チSWを印加トランスOTの１次側に接続したものであり,1
次側の電圧が２次側の電圧より高い場合，その比率分だ
けコンデンサＣの容量を小さくすることにより第１図の
実施例で述べた動作を同じにしたものである。FIG. 3 shows another embodiment, in which the capacitor C and the switch SW are connected to the primary side of the applying transformer OT.
When the voltage on the secondary side is higher than the voltage on the secondary side, the operation described in the embodiment of FIG. 1 is made the same by reducing the capacity of the capacitor C by the ratio.

またコンデンサに接続された導線を変流器に貫通させる
のではなく，複数回巻線しても，その分コンデンサの容
量を小さくすることも可能である。Further, it is possible to reduce the capacitance of the capacitor by winding the conductor wire connected to the capacitor multiple times instead of passing through the current transformer.

なお，コンデンサに印加する電圧は（７）式で示した如
くＶとしたが，これに制約されないことは明らかであ
り，他の電圧であっても動作上は何ら問題ない。Although the voltage applied to the capacitor is set to V as shown in the equation (7), it is obvious that it is not limited to this, and other voltages will not cause any problem in operation.

また，位相制御回路から出力され位相同期回路の第２の
入力端に印加される位相は同位相となるように前もって
調整し，温度等による位相ずれのみを上述の自動位相制
御にて補償するようにすれば位相同期の時間を短縮する
ことができる。Also, the phase output from the phase control circuit and applied to the second input terminal of the phase synchronization circuit is adjusted in advance so as to be the same phase, and only the phase shift due to temperature or the like is compensated by the above-mentioned automatic phase control. By doing so, the phase synchronization time can be shortened.

さらに上記説明では測定用低周波信号電圧と90゜位相の
異なる電流を流すために，コンデンサ素子を用いたが，
必ずしもこれに限定されるものではなく他の回路網（例
えば，インダクタンスとコンデンサとを組合せた回路）
を用いてもよいしまた，他の電気回路で発生してもよい
ことは明らかである。Furthermore, in the above description, a capacitor element was used to pass a current with a 90 ° phase difference from the low frequency signal voltage for measurement.
It is not necessarily limited to this, and other circuit networks (for example, a circuit combining an inductance and a capacitor)
Obviously, it may be used or generated in other electric circuits.

また上記説明では測定用低周波信号電圧を正弦波として
説明したが，これに限定されるものではなく例えば矩形
波であってもよくその基本波成分或いは高調波成分を用
いてもよい。Further, in the above description, the measurement low-frequency signal voltage is described as a sine wave, but the present invention is not limited to this, and it may be, for example, a rectangular wave or its fundamental wave component or higher harmonic component.

なお上記説明においては位相調整をするに当り，同期検
波器MULT1,MULT2の第２の入力に印加される信号の位相
を調整する如くしたが，同期検波器MULT1,MULT2の第１
の入力に印加される信号の位相を調整しても同一の結果
が得られることは明らかである。In the above description, the phase of the signal applied to the second input of the synchronous detectors MULT1 and MULT2 is adjusted when the phase is adjusted.
Obviously, the same result can be obtained by adjusting the phase of the signal applied to the input of the.

また上記実施例では単相２線式電路の場合で示したが，
単相３線式電路,3相３線式電路であってもよい。In the above embodiment, the case of the single-phase two-wire type electric circuit is shown.
It may be a single-phase three-wire type electric line or a three-phase three-line type electric line.

（発明の効果） 以上説明したごとく，本発明は絶縁抵抗測定回路の位相
特性変動の自動的位相調整を可能にするものであるから
極めて安定な測定方法を実現するうえで著効を奏するも
のである。(Effects of the Invention) As described above, the present invention enables automatic phase adjustment of the phase characteristic fluctuation of the insulation resistance measuring circuit, and thus is extremely effective in realizing an extremely stable measurement method. is there.

【図面の簡単な説明】[Brief description of drawings]

第１図は本発明の一実施例を示すブロック図，第２図及
び第３図は本発明の他の実施例を示す部分的ブロック
図，第４図は従来の絶縁抵抗を測定する方法を示すブロ
ック図である。 Ｔ……トランス,1,2……電路,L_E……接地線,E……接地
点,ZCT……変流器,AMP……増幅器,FIL……フィルタ,MUL
T1,2,3……同期検波回路,OSC……発振器,OT,OT′……印
加トランス,PS……移相器,SW……スイッチ,MT……かけ
算器,PC……位相制御回路,BP……フィルタ,PSO……90゜
移相器,SUB……引算器,COF……係数回路,PAMP……電力
増幅器。FIG. 1 is a block diagram showing an embodiment of the present invention, FIGS. 2 and 3 are partial block diagrams showing another embodiment of the present invention, and FIG. 4 shows a conventional method for measuring insulation resistance. It is a block diagram shown. T …… transformer, 1,2 …… electric circuit, L _E …… ground wire, E …… ground point, ZCT …… current transformer, AMP …… amplifier, FIL …… filter, MUL
T1,2,3 …… Synchronous detection circuit, OSC …… Oscillator, OT, OT ′ …… Applying transformer, PS …… Phase shifter, SW …… Switch, MT …… Multiplier, PC …… Phase control circuit, BP: filter, PSO: 90 ° phase shifter, SUB: subtractor, COF: coefficient circuit, PAMP: power amplifier.

Claims (3)

【特許請求の範囲】[Claims] 【請求項１】電路に、商用周波数と異なる周波数f1なる
測定用低周波信号電圧を印加し、前記電路と大地との間
に挿入した所定のリアクタンス素子の値を繰り返し周期
Ｔなる信号で変化させ、電路の接地線に結合せしめた変
流器の出力中に含まれる前記周波数f1の漏洩電流成分を
抽出すると共に、この抽出した漏洩電流出力を前記測定
用低周波信号電圧で同期検波することにより第１の信号
を得、前記漏洩電流出力を前記測定用低周波信号電圧と
は位相が90゜推移した電圧で同期検波することにより第
２の信号を得ると共に、前記第１の信号中に含まれる周
波数1/Tの周波数成分を抽出して前記繰り返し周期Ｔの
信号で同期検波することにより第３の信号を得、該第３
の信号が零に近づくように前記第１及び第２の信号を得
るために同期検波器へ印加する前記測定用低周波信号電
圧ならびに前記測定用低周波信号電圧より90゜位相の推
移した電圧の位相を調整し、更に前記第３の信号を定数
倍した信号と前記第２の信号との積をとる掛算器出力と
前記第１の信号との差を用いて電路の絶縁抵抗を測定し
たことを特徴とする位相補償した電路の絶縁抵抗測定方
法。1. A low frequency signal voltage for measurement having a frequency f1 different from a commercial frequency is applied to an electric line, and a value of a predetermined reactance element inserted between the electric line and the ground is changed by a signal having a repetition period T. , By extracting the leakage current component of the frequency f1 included in the output of the current transformer coupled to the ground wire of the electric path, and by synchronously detecting the extracted leakage current output with the low frequency signal voltage for measurement. A first signal is obtained, and a second signal is obtained by synchronously detecting the leakage current output with a voltage whose phase shifts by 90 ° with respect to the low frequency signal voltage for measurement, and is included in the first signal. A frequency component of frequency 1 / T is extracted, and a third signal is obtained by performing synchronous detection with the signal of the repetition period T.
Of the low frequency signal voltage for measurement applied to the synchronous detector to obtain the first and second signals so that the signal of FIG. The insulation resistance of the electric circuit was measured using the difference between the first signal and the output of a multiplier that adjusts the phase and that further multiplies the third signal by a constant and the second signal. And a method for measuring insulation resistance of a phase-compensated electric circuit. 【請求項２】電路に、商用周波数と異なる周波数f1なる
測定用低周波信号電圧を印加し、該電路の接地線に結合
せしめた変流器の出力中に含まれる前記周波数f1の漏洩
電流成分を抽出すると共に、この抽出した出力を前記測
定用低周波信号電圧で同期検波することにより得られる
出力を用いて電路と大地間の絶縁抵抗を測定する方法に
おいて、 前記電路と大地との間に挿入した所定のリアクタンス素
子及び該リアクタンス素子の値を繰り返し周期Ｔなる信
号で変化させる電流印加手段と、 前記交流器出力中から商用周波数成分を除去して得た信
号と前記測定用低周波信号電圧とを入力し同期検波する
第１の同期検波手段と、 前記交流器出力中から商用周波数成分を除去して得た信
号と前記測定用低周波信号電圧とは位相が90゜推移した
電圧とを入力し同期検波する第２の同期検波手段と、 前記第１及び２の同期検波手段に入力する前記測定用低
周波信号若しくは商用周波数成分を除去した信号の位相
を調整する位相調整手段と、 前記第１の同期検波手段出力中に含まれる周波数1/Tの
成分を検出するフイルタ手段と、 前記フイルタ手段出力及び前記繰り返し周期Ｔの信号を
入力し同期検波する第３の同期検波手段と、 前記第３の同期検波手段出力を定数倍する係数手段と、 前記係数手段出力と前記第２の同期検波手段出力との積
をとる掛け算手段と、 前記掛け算手段出力と前記第１の同期検波手段出力との
差をとる引算手段とを備え、 前記第３の同期検波手段出力が零に近づくように前記位
相調整手段を制御することにより、前記引算手段の出力
信号を用いて電路の絶縁抵抗を測定したことを特徴とす
る特許請求の範囲第１項記載の絶縁抵抗測定方法。2. A leakage current component of the frequency f1 contained in the output of a current transformer connected to the ground wire of the electric line by applying a low frequency signal voltage for measurement having a frequency f1 different from the commercial frequency to the electric line. In the method of measuring the insulation resistance between the electric line and the ground using the output obtained by synchronously detecting the extracted output with the measuring low-frequency signal voltage, between the electric line and the ground. A predetermined reactance element inserted and current applying means for changing the value of the reactance element with a signal having a cycle T, a signal obtained by removing a commercial frequency component from the output of the AC device, and the low frequency signal voltage for measurement. And a signal obtained by removing the commercial frequency component from the output of the AC device and the low-frequency signal voltage for measurement, which are phase-shifted by 90 °. Second synchronous detection means for inputting and performing synchronous detection; phase adjusting means for adjusting the phase of the low frequency signal for measurement or the signal from which the commercial frequency component has been removed input to the first and second synchronous detection means; Filter means for detecting a frequency 1 / T component included in the output of the first synchronous detection means; third synchronous detection means for synchronously detecting the output of the filter means and the signal of the repetition period T; Coefficient means for multiplying the third synchronous detection means output by a constant, multiplication means for taking the product of the coefficient means output and the second synchronous detection means output, the multiplication means output and the first synchronous detection means output And subtracting means for calculating the difference between the subtraction means and the phase adjustment means so that the output of the third synchronous detection means approaches zero, thereby using the output signal of the subtraction means to isolate the insulation resistance of the electric path. Measure Insulation resistance measuring method of the claims paragraph 1, wherein it has. 【請求項３】前記電流印加手段は、前記測定用低周波信
号電圧と90゜位相の推移した所定の大きさの電流を繰り
返し周期Ｔの信号で変化させ、該電流の流れる導線を前
記交流器に貫通させ若しくは巻線したものであることを
特徴とする特許請求の範囲第２項記載の位相補償した電
路の絶縁抵抗測定方法。3. The current applying means changes a current of a predetermined magnitude having a 90 ° phase shift with the low frequency signal voltage for measurement with a signal having a repetition period T, and the conducting wire through which the current flows is the AC device. A method for measuring insulation resistance of a phase-compensated electric circuit according to claim 2, characterized in that it is penetrated through or wound.