JPH01321868A - Pwm electric power converting device and control thereof - Google Patents
Pwm electric power converting device and control thereofInfo
- Publication number
- JPH01321868A JPH01321868A JP63154748A JP15474888A JPH01321868A JP H01321868 A JPH01321868 A JP H01321868A JP 63154748 A JP63154748 A JP 63154748A JP 15474888 A JP15474888 A JP 15474888A JP H01321868 A JPH01321868 A JP H01321868A
- Authority
- JP
- Japan
- Prior art keywords
- circuit
- smoothing capacitor
- pwm
- conversion device
- frequency
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 239000003990 capacitor Substances 0.000 claims abstract description 66
- 238000009499 grossing Methods 0.000 claims abstract description 56
- 238000000034 method Methods 0.000 claims abstract description 11
- 238000006243 chemical reaction Methods 0.000 claims description 18
- 230000006698 induction Effects 0.000 abstract description 2
- 238000010586 diagram Methods 0.000 description 9
- 230000000694 effects Effects 0.000 description 7
- 238000001514 detection method Methods 0.000 description 3
- XEEYBQQBJWHFJM-UHFFFAOYSA-N Iron Chemical compound [Fe] XEEYBQQBJWHFJM-UHFFFAOYSA-N 0.000 description 2
- 238000010521 absorption reaction Methods 0.000 description 2
- 238000005516 engineering process Methods 0.000 description 2
- 239000013256 coordination polymer Substances 0.000 description 1
- 230000003247 decreasing effect Effects 0.000 description 1
- 239000000284 extract Substances 0.000 description 1
- 229910052742 iron Inorganic materials 0.000 description 1
- 230000010355 oscillation Effects 0.000 description 1
Landscapes
- Power Conversion In General (AREA)
- Inverter Devices (AREA)
Abstract
Description
【発明の詳細な説明】
〔産業上の利用分野〕
本発明はＰＷＭ電力変換装置に関し、特に平滑コンデン
サのりプル電流を低減してその容量を低減する方法に関
する。DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a PWM power converter, and more particularly to a method for reducing ripple current in a smoothing capacitor to reduce its capacitance.
単相交流を直流に変換するコンバータ装置において、整
流リプル電流の周波数に共振する直列共振回路を設けて
整流リプル電流を吸収し、平滑コンデンサの小形化を図
る方法が、エレクトリシエ・バーネン、４５ （１９７
４年）第１３５頁から１４２頁（Ｅｌｅｋｔｒｉｓｃｈ
ｅ Ｂａｈｎｅｎ ４５ （１９７４）ｐｐ１３５−１
４２）に記載されている。In a converter device that converts single-phase alternating current to direct current, there is a method in which a series resonant circuit that resonates with the frequency of the rectified ripple current is provided to absorb the rectified ripple current and thereby reduce the size of the smoothing capacitor. 197
4th year) pages 135 to 142 (Elektrisch
e Bahnen 45 (1974) pp135-1
42).
しかしながら、上記従来技術はＰＷＭ電力変換装置のＰ
ＷＭ制御に伴う直流電流のりプルの吸収に関しては触れ
られておらず、また直列共振回路の共振周波数が変動し
た場合のりプル電流の吸収効果の低下とその対策法につ
いても触れられていない。However, the above conventional technology
There is no mention of the absorption of DC current ripple associated with WM control, nor is there any mention of the reduction in the ripple current absorption effect when the resonant frequency of the series resonant circuit fluctuates, and how to deal with it.
本発明の目的は、ＰＷＭ電力変換装置の直流電流に含ま
れるリプル成分を効果的に吸収して平滑コンデンサに流
入するりプル電流を抑制し、平滑コンデン゛すを小形化
したＰＷＭ電力変換装置、並びにリプル吸収用直列共振
回路のコンデンサとリアクトルのキャパシタンス値及び
インダクタンス値が温度等により変動しても常にリプル
電流を効果的に吸収し、平滑コンデンサの容量を低減で
きるＰＷＭ電力変換装置の制御方法を提供することにあ
る。The object of the present invention is to provide a PWM power converter that effectively absorbs ripple components contained in the DC current of the PWM power converter, suppresses the pull current flowing into the smoothing capacitor, and downsizes the smoothing capacitor. In addition, a method for controlling a PWM power converter is provided that can effectively absorb ripple current at all times and reduce the capacitance of the smoothing capacitor even if the capacitance and inductance values of the capacitor and reactor of the ripple absorbing series resonant circuit vary due to temperature, etc. It is about providing.
上記目的は、ＰＷＭ電力変換装置の平滑コンデンサと並
列にＰＷＭ制御に伴うリプル電流を吸収する直列共振回
路を設け、平滑コンデンサに流れる電流リプルの大きさ
が最小となるようにＰＷＭ制御の搬送波周波数を制御す
ることにより達成される。The above purpose is to install a series resonant circuit in parallel with the smoothing capacitor of the PWM power converter to absorb the ripple current accompanying PWM control, and to adjust the carrier frequency of the PWM control so that the magnitude of the current ripple flowing through the smoothing capacitor is minimized. This is achieved through control.
ＰＷＭ電力変換装置の平滑コンデンサと並列にコンデン
サとリアクトルからなる直列共振回路を設け、さらに平
滑コンデンサに流れる電流のりプル成分の振幅が最小と
なるようにＰＷＭ制御の搬送波周波数を可変する。それ
によって、直列共振回路の共振周波数とりプル電流成分
の周波数が一致するようになるのでリプル成分は直列共
振回路を流れる共振電流によって吸収され、平滑コンデ
ンサにはりプル電流がほとんど流れない。また。A series resonant circuit consisting of a capacitor and a reactor is provided in parallel with the smoothing capacitor of the PWM power converter, and the carrier frequency of PWM control is varied so that the amplitude of the ripple component of the current flowing through the smoothing capacitor is minimized. As a result, the resonant frequency of the series resonant circuit and the frequency of the pull current component match, so that the ripple component is absorbed by the resonant current flowing through the series resonant circuit, and almost no pull current flows through the smoothing capacitor. Also.
直列共振回路のコンデンサとリアクトルの値が温度上昇
等によって変化した場合でも、常に共振条件が満足され
るように搬送波周波数を制御する。Even if the values of the capacitor and reactor of the series resonant circuit change due to temperature rise, etc., the carrier wave frequency is controlled so that the resonance condition is always satisfied.
すなわち、共振周波数変化を平滑コンデンサに流れるリ
プル成分から検出することができ、このリプル成分が最
小となるようにＰＷＭ制御の搬送波周波数を可変してい
るので共振回路により常時効果的にリプル成分を吸収す
ることができる。その結果、ＰＷＭ電力変換装置の平滑
コンデンサ容量を低減することができる。In other words, changes in the resonance frequency can be detected from the ripple component flowing through the smoothing capacitor, and since the carrier frequency of PWM control is varied so that this ripple component is minimized, the ripple component can be effectively absorbed by the resonant circuit at all times. can do. As a result, the smoothing capacitor capacity of the PWM power converter can be reduced.
〔実施例〕
第１図に本発明をＰＷＭインバータに適用した場合の一
実施例を示す。第１図において、ＰＷＭインバータ６は
直流電源１から加えられる直流電圧を交流電圧に変換す
る。インバータ６はグレーツ結線された自己消弧素子Ｔ
ＵＰ、ＴＶＰ、・・・・・・ＴＷＮと各自己消弧素子に
逆並列接続された帰還ダイオードＤＵＰ、ＤＶＰ、・・
・・・・ＤＷＮとから構成される。自己消弧素子として
はトランジスタやゲートターンオフサイリスタなどのス
イッチング素子が用いられる。インバータ６の各相Ｕ、
Ｖ。[Embodiment] FIG. 1 shows an embodiment in which the present invention is applied to a PWM inverter. In FIG. 1, a PWM inverter 6 converts a DC voltage applied from a DC power supply 1 into an AC voltage. The inverter 6 is a Graetz-connected self-extinguishing element T.
UP, TVP, ... TWN and feedback diodes DUP, DVP, ... connected in antiparallel to each self-extinguishing element.
...DWN. As the self-extinguishing element, a switching element such as a transistor or a gate turn-off thyristor is used. Each phase U of the inverter 6,
V.
Ｗの交流出力端に誘導電動機７が接続されている。An induction motor 7 is connected to the AC output terminal of W.
速度指令回路１５の速度指令信号は電圧−周波数変換器
１４に加えられる。電圧−周波数変換器１４は速度指令
信号に応じて周波数指令信号を出力する。周波数指令信
号でインバータ１の出力周波数（電動機７の１次周波数
）を決定する。電圧指令回回路１３は周波数指令信号の
大きさに比例した振幅で、周波数が周波数指令信号に比
例した１２０°位相差の電圧指令パターン信号Ｖｔ１１
゜Ｖｖ”ｙＶ−を比較器１２Ｕ、１２Ｖ、１２Ｗに出力
する。パルス幅変調制御のための搬送波信号を発生する
発振器１０の出力信号は比較器１２Ｕ。A speed command signal from speed command circuit 15 is applied to voltage-frequency converter 14 . The voltage-frequency converter 14 outputs a frequency command signal in response to the speed command signal. The output frequency of the inverter 1 (the primary frequency of the motor 7) is determined by the frequency command signal. The voltage command circuit 13 generates a voltage command pattern signal Vt11 with an amplitude proportional to the magnitude of the frequency command signal and a 120° phase difference whose frequency is proportional to the frequency command signal.
゜Vv''yV- is outputted to the comparators 12U, 12V, and 12W.The output signal of the oscillator 10, which generates a carrier wave signal for pulse width modulation control, is outputted to the comparator 12U.
１２Ｖ、１２Ｗ１ｍ加えられる。比較器１２Ｕ。12V, 12W is applied for 1m. Comparator 12U.
１２Ｖ、１２Ｗは電圧指令パターン信号Ｖｕ”。12V and 12W are voltage command pattern signals Vu''.
Ｖｖ＊ｐＶ−と搬送波信号とを比較し、ＰＷＭインバー
タ６を構成するスイッチング素子ＴＵＰ。A switching element TUP that compares Vv*pV- with a carrier signal and configures a PWM inverter 6.
ＴＶＰ、・・・・・・ＴＷＮをオン、オフするためのパ
ルス幅変調パ・ルスをゲート回路１１に出力する。ゲー
ト回路１１はパルス幅変調パルスに応じてスイッチング
素子ＴＵＰ、ＴＶＰ、・・・・・・ＴＷＮにゲート信号
を与える。TVP, . . . outputs pulse width modulated pulses for turning on and off TWN to the gate circuit 11. The gate circuit 11 provides gate signals to the switching elements TUP, TVP, . . . TWN in response to the pulse width modulation pulse.
直流電源１の直流電圧は、平滑コンデンサ２と直列共振
回路１８とインバータ６に与えられる。The DC voltage of the DC power supply 1 is applied to the smoothing capacitor 2, the series resonant circuit 18, and the inverter 6.
直列共振回路１８はインダクタンス４とコンデンサ５か
ら構成される。平滑コンデンサ２に流れる電流は電流検
出器３によって検出され、その検出電流信号がフィルタ
回路８に加えられる。フィルタ回路８は検出電流信号に
含まれるＰＷＭ制御に伴うリプル成分を取り出してトラ
ッキング制御回路９に加える。トラッキング制御回路９
はそのリプル成分の大きさが最小となる周波数指令を演
算して発振器１０に加える。発振器１０はトラッキング
制御回路９の出力信号に基づいて発振周波数を可変して
その出力信号を比較器１２Ｕ、１２Ｖ。The series resonant circuit 18 is composed of an inductance 4 and a capacitor 5. The current flowing through the smoothing capacitor 2 is detected by the current detector 3, and the detected current signal is applied to the filter circuit 8. The filter circuit 8 extracts a ripple component associated with PWM control included in the detected current signal and applies it to the tracking control circuit 9. Tracking control circuit 9
calculates a frequency command that minimizes the magnitude of the ripple component and applies it to the oscillator 10. The oscillator 10 varies the oscillation frequency based on the output signal of the tracking control circuit 9 and sends the output signal to the comparators 12U and 12V.
１２Ｗに加える。Add to 12W.
まず、その動作を簡単に説明する。第１図はＶ／ｆ制御
インバータ装置であり、インバータ６の出力電圧／出力
周波数の値が一定の交流出力を電動機７に供給する。First, its operation will be briefly explained. FIG. 1 shows a V/f control inverter device, which supplies an AC output with a constant value of output voltage/output frequency of an inverter 6 to an electric motor 7.
次に、本発明に関する動作を第２図、第３図を参照して
説明する。ＰＷＭインバータ６の直流入力電流にはＰＷ
Ｍ動作に伴うリプル電流が含まれるが、それは平滑コン
デンサ（通常、電解コンデンサが用いられる）に流入し
、その温度が上昇する。そのため、リプル電流に応じて
大容量の平滑コンデンサ２を直流回路に設けている。と
ころが、このリプル電流の大きさはインバータ６の出力
電流の大きさに比例して増加するため、大容量Ｐｗにイ
ンバータでは平滑コンデンサ２の大きさが大容量となる
。そこで、本発明では、リプル電流の周波数がＰＷＭ制
御の搬送波周波数ｆｃに関係するｆｃの整数倍であるこ
とに着目し、このりプル電流を吸収する直列共振回路１
８を平滑コンデンサ２に並列に設けている。直列共振回
路１８の共振周波数ｆ、はインダクタンス４の大きさＬ
ｘとコンデンサ５の大きさＣ２から決まる。第２図は直
流回路の等価回路である。第２図において、ｒｓ。Next, the operation of the present invention will be explained with reference to FIGS. 2 and 3. The DC input current of the PWM inverter 6 is
A ripple current associated with the M operation is included, which flows into a smoothing capacitor (usually an electrolytic capacitor), increasing its temperature. Therefore, a large capacity smoothing capacitor 2 is provided in the DC circuit depending on the ripple current. However, since the magnitude of this ripple current increases in proportion to the magnitude of the output current of the inverter 6, the smoothing capacitor 2 in the inverter has a large capacitance due to the large capacitance Pw. Therefore, in the present invention, focusing on the fact that the frequency of the ripple current is an integral multiple of fc related to the carrier frequency fc of PWM control, a series resonant circuit that absorbs this ripple current
8 is provided in parallel with the smoothing capacitor 2. The resonant frequency f of the series resonant circuit 18 is the size L of the inductance 4.
It is determined from x and the size C2 of the capacitor 5. FIG. 2 is an equivalent circuit of a DC circuit. In FIG. 2, rs.
ｒ’ｉ、 ｒｚは配線による抵抗、Ｑ８は電源側のイン
ダクタンスである。８ｄｅは平滑コンデンサ２の端子に
表わされるリプル電圧である。平滑コンデンサ２に流れ
る電流１１直列共振回路１８に流れる電流ｉｚ及び共振
周波数ｆｒ、共振の鋭さＱは次式で表わされる。但し、
ω＝２πｆｃである。r'i and rz are resistances caused by wiring, and Q8 is inductance on the power supply side. 8de is a ripple voltage expressed at the terminal of the smoothing capacitor 2. The current 11 flowing through the smoothing capacitor 2, the current iz flowing through the series resonant circuit 18, the resonance frequency fr, and the resonance sharpness Q are expressed by the following equations. however,
ω=2πfc.
ここで、直列共振回路１８の共振周波数！、とリプル電
流の周波数ｆｃを一致させると、直流電圧の変動による
電流成分は直列共振回路１８を流れ、平滑コンデンサ２
には流れない。Here, the resonant frequency of the series resonant circuit 18! , and the ripple current frequency fc, the current component due to the fluctuation of the DC voltage flows through the series resonant circuit 18, and the smoothing capacitor 2
It doesn't flow.
ところが、直列共振回路１８に電流が流れるのに伴い温
度上昇等によりコンデンサ５とリアクトル４の値が変化
し、共振周波数ｆ、がリプル電流の周波数ｆｃと一致し
なくなる。その結果、平滑コンデンサ２に流入するりプ
ル電流が増加する。However, as current flows through the series resonant circuit 18, the values of the capacitor 5 and the reactor 4 change due to temperature rise, etc., and the resonance frequency f no longer matches the frequency fc of the ripple current. As a result, the pull current flowing into the smoothing capacitor 2 increases.
そこで９本♀明ではさらに、直列共振回路１８の共振周
波数ｆ、とりプル電流の周波数ｆｃが常に一致するよう
に動作させる。次に、これを説明する。Therefore, in the ninth test, the series resonant circuit 18 is further operated so that the resonant frequency f and the frequency fc of the pull current always match. Next, this will be explained.
直列共振回路１８の電流ｉｚと平滑コンデンサ２の電流
１１には次式及び第３図に示す関係が（１）、 （２）
式より得られる。The current iz of the series resonant circuit 18 and the current 11 of the smoothing capacitor 2 have the relationship shown in the following equation and FIG. 3 as (1), (2)
It is obtained from Eq.
第３図は直列共振回路１８の共振の鋭さＱが１００と６
０の場合について、ｆｃ／ｆ、に対してｌｉｚ／ｉｚｌ
の大きさを計算したものである。第３図より、共振周波
数！、が直流電圧の変動周波数ｆＣからはずれると平滑
コンデンサ２に流れる電流が増加することを示している
。そこで、本発明では平滑コンデンサ２の電流を電流検
出器３により検出し、さらにリプル電流成分のみをフィ
ルタ回路８を介して検出し、その大きさが最小となる周
波数値を第３図に示す関係に基づいてトラッキング制御
回路９で演算し、それによりＰＷＭインバータ６の搬送
波周波数を共振周波数ｆ、と一致するように発振器１０
を制御する。トラッキング制御回路９は、リプル電流の
大きさ八が所定値以上であれば、以下の手順でリプル電
流の大きさを最小とする周波数値を決定する。現在の搬
送波周波数ｆをΔｆ＞Ｏだけ増加し、リプル電流の大き
さの変化ΔＡの極性を調べ、ΔＡ＞Ｏであれば搬送波周
波数ｆを下げ、逆にΔＡくＯであれば搬送波周波数ｆを
上げて行く。その結果、リプル電流の大きさＡが所定値
以下となった時点でその周波数値をホールドする。Figure 3 shows that the resonance sharpness Q of the series resonant circuit 18 is 100 and 6.
For the case 0, liz/izl for fc/f,
The size of is calculated. From Figure 3, the resonance frequency! deviates from the fluctuation frequency fC of the DC voltage, the current flowing through the smoothing capacitor 2 increases. Therefore, in the present invention, the current of the smoothing capacitor 2 is detected by the current detector 3, and only the ripple current component is detected via the filter circuit 8, and the frequency value at which the magnitude thereof is minimized is determined by the relationship shown in FIG. The tracking control circuit 9 calculates the carrier frequency of the PWM inverter 6 based on the resonant frequency f.
control. If the magnitude of the ripple current is greater than or equal to a predetermined value, the tracking control circuit 9 determines a frequency value that minimizes the magnitude of the ripple current using the following procedure. Increase the current carrier frequency f by Δf>O, check the polarity of the change in ripple current magnitude ΔA, and if ΔA>O, lower the carrier frequency f, and conversely, if ΔA×O, increase the carrier frequency f. I'm going to raise it. As a result, when the magnitude A of the ripple current becomes less than or equal to a predetermined value, the frequency value is held.
その結果、ＰＷＭ動作に伴うリプル電流を直列共振回路
１８により効果的に吸収し、平滑コンデンサ２の容量を
低減することができる。As a result, the ripple current accompanying PWM operation can be effectively absorbed by the series resonant circuit 18, and the capacitance of the smoothing capacitor 2 can be reduced.
第４図は本発明をＰＷＭコンバータに適用した場合の他
の実施例である。第１実施例と同一物には同じ番号を付
しているので説明を省略する。第４図において、ＰＷＭ
コンバータ１６は交流電源１７から加えられる交流電圧
を直流電圧に変換する。コンバータ１６は自己消弧素子
ＧＵＰ、ＧＶＰ。FIG. 4 shows another embodiment in which the present invention is applied to a PWM converter. Components that are the same as those in the first embodiment are given the same numbers, and therefore their description will be omitted. In Figure 4, PWM
Converter 16 converts AC voltage applied from AC power supply 17 into DC voltage. The converter 16 includes self-extinguishing elements GUP and GVP.
・・・・・・ＧＷＮと各自己消弧素子に逆並列接続され
た帰還ダイオードＤＵＰ、ＤＶＰ、・・・・・・ＤＷＮ
とから構成される。自己消弧素子としてはゲートターン
オフサイリスタやトランジスタなどのスイッチング素子
が用いられる。コンバータの制御構成は、直流電圧指令
器１９よりのＶｄｅ”と、平滑コンデンサ２の端子電圧
検出器２１の検出信号とが加算器２０に加えられ、加算
器２０の出力が電圧調節器２２に加えられる。電圧調節
器２２の出力信号と交流電源１７の電圧検出器２４の検
出信号とが力率調節器２３に加えられ、コンバータ１６
の入力電流の瞬時値指令パターン信号が演算されて加算
器２５に加えられる。加算器２５において、コンバータ
トロの入力電流の電流検出器２６の検出信号と力率調節
器２３の出力信号との偏差が演算され、その偏差が電流
調節器２７に加えられる。...Feedback diodes DUP, DVP, ...DWN connected in antiparallel to GWN and each self-extinguishing element
It consists of As the self-extinguishing element, a switching element such as a gate turn-off thyristor or a transistor is used. The control configuration of the converter is such that Vde'' from a DC voltage command 19 and a detection signal from a terminal voltage detector 21 of a smoothing capacitor 2 are added to an adder 20, and the output of the adder 20 is added to a voltage regulator 22. The output signal of voltage regulator 22 and the detection signal of voltage detector 24 of AC power supply 17 are applied to power factor regulator 23, and converter 16
The instantaneous value command pattern signal of the input current is calculated and added to the adder 25. The adder 25 calculates the deviation between the detection signal of the current detector 26 of the input current of the converter and the output signal of the power factor regulator 23, and adds the deviation to the current regulator 27.
電流調節器２７においてコンバータ１６の各相の出力電
圧指令Ｖｕ＊、Ｖｖ＊、Ｖ−が演算されて各々が比較器
１２Ｕ、１２Ｖ、１２Ｗに加えられる。Current regulator 27 calculates output voltage commands Vu*, Vv*, and V- for each phase of converter 16, and applies them to comparators 12U, 12V, and 12W, respectively.
次に動作を簡単に説明する。コンバータ１６の制御は、
直流出力電圧を制御するための電圧調節器２２と交流電
源電圧を基準にして所定の力率とするコンバータ１６の
入力電流瞬時値指令を出力する力率調節器２３と、その
入力電流瞬時値指令に一致するようにコンバータ入力電
流を制御する電流調節器２７および比較器１２Ｕ−１２
Ｗ等からなるＰＷＭ制御回路により行なわれる。Next, the operation will be briefly explained. Control of the converter 16 is as follows:
A voltage regulator 22 for controlling the DC output voltage, a power factor regulator 23 for outputting an instantaneous input current value command for the converter 16 to achieve a predetermined power factor based on the AC power supply voltage, and an instantaneous input current value command for the converter 16. current regulator 27 and comparator 12U-12 to control the converter input current to match
This is performed by a PWM control circuit composed of W or the like.
ＰＷＭ制御は、コンバータ１６の出力電圧の平滑コンデ
ンサ２に流れるリプル電流の振幅が最小となるようにト
ラッキング制御回路９で演算された周波数値に一致した
搬送波周波数で行なう。その結果、コンバータ１６のＰ
ＷＭ動作に伴う直流電流に含まれるリプル成分は、平滑
コンデンサ２と並列に設けられた直列共振回路１８によ
って吸収され、平滑コンデンサ２へのりプル電流の流入
が抑制できるので、平滑コンデンサ２の容量を低減する
ことができる。The PWM control is performed at a carrier frequency that matches the frequency value calculated by the tracking control circuit 9 so that the amplitude of the ripple current flowing through the smoothing capacitor 2 of the output voltage of the converter 16 is minimized. As a result, P of converter 16
The ripple component included in the DC current accompanying the WM operation is absorbed by the series resonant circuit 18 provided in parallel with the smoothing capacitor 2, and the flow of ripple current into the smoothing capacitor 2 can be suppressed, so the capacitance of the smoothing capacitor 2 can be reduced. can be reduced.
第５図は本発明をＰＭＷコンバータに適用した場合の他
の実施例である。第４図に示す第２実施例と同一物には
同じ番号を付しているので説明を省略する。第２実施例
と異なる点は、電流検出器３を直列共振回路１８に設け
たことである。（２）式に示すように直列共振回路１８
の電流は共振周波数ｆ、と直流電流のりプル周波数ｆｃ
が一致する時に最大となる。この関係は第３図に示す特
性とちょうど逆になる。そこで、本実施例ではトラッキ
ング制御回路９において、直列共振回路１８の電流を検
出してリプル成分が最大となように周波数値を決定し、
搬送波周波数を制御する。トラッキング制御回路９は、
リプル電流の大きさＡが最大値となるように以下の手順
で周波数値を決定する。現在の搬送波周波数ＪをΔｆ＞
Ｏだけ増加し、リプル電流の大きさの変化ΔＡの極性を
調べ、ΔＡ＞ Ｏ，であれば搬送波周波数ｆを上げ、逆
にΔＡ＜Ｏであれば搬送波周波数Ｊを下げて行く。FIG. 5 shows another embodiment in which the present invention is applied to a PMW converter. Components that are the same as those in the second embodiment shown in FIG. 4 are given the same numbers, and therefore their explanation will be omitted. The difference from the second embodiment is that the current detector 3 is provided in the series resonant circuit 18. As shown in equation (2), the series resonant circuit 18
The current has a resonant frequency f, and a direct current pull frequency fc
is maximum when they match. This relationship is exactly opposite to the characteristic shown in FIG. Therefore, in this embodiment, the tracking control circuit 9 detects the current of the series resonant circuit 18 and determines the frequency value so that the ripple component is maximized.
Control carrier frequency. The tracking control circuit 9 is
The frequency value is determined by the following procedure so that the magnitude A of the ripple current becomes the maximum value. The current carrier frequency J is Δf>
The polarity of the change ΔA in the magnitude of the ripple current is checked, and if ΔA>O, the carrier wave frequency f is increased, and conversely, if ΔA<O, the carrier wave frequency J is decreased.
その結果、リプル電流の変化ΔＡが所定値以下の幅の範
囲に入った時点でその周波数値をホールドする。As a result, the frequency value is held when the change ΔA of the ripple current falls within a range of width equal to or less than a predetermined value.
本実施例によっても第２実施例と同様の効果を得ること
ができる。This embodiment also provides the same effects as the second embodiment.
また、同様の考えで平滑コンデンサ２の端子電圧からり
プル成分の大きさを検出して、第２実施例と同様の構成
で本発明を実施することもできる。Furthermore, the present invention can also be implemented with the same configuration as the second embodiment by detecting the magnitude of the pull component from the terminal voltage of the smoothing capacitor 2 based on the same idea.
第６図は本発明をＰＷＭコンバータに適用した場合の他
の実施例である。第４図に示す第２実施例と同一物には
同じ番号を付しているので説明を省略する。第２実施例
と異なる点は、直列共振回路１８をＰＷＭコンバータ１
６に近い側に設け、さらに直列共振回路１８と平滑コン
デンサ２との間にリアクトル２８を設けたことである。FIG. 6 shows another embodiment in which the present invention is applied to a PWM converter. Components that are the same as those in the second embodiment shown in FIG. 4 are given the same numbers, and therefore their explanation will be omitted. The difference from the second embodiment is that the series resonant circuit 18 is connected to the PWM converter 1.
6, and further a reactor 28 is provided between the series resonant circuit 18 and the smoothing capacitor 2.
リアクトル２８はＰＷＭコンバータ１６のＰＷＭ動作に
伴うリプル電流成分が平滑コンデンサ２に流れ込むこと
を防止するように作用する。その結果、平滑コンデンサ
のりプル電流をさらに効果的に低減することができる。The reactor 28 acts to prevent ripple current components associated with the PWM operation of the PWM converter 16 from flowing into the smoothing capacitor 2 . As a result, the smoothing capacitor ripple current can be further effectively reduced.
また、同様の考えでリアクトル２８の代りに抵抗要素を
用いることも出来ると共に、平滑コンデンサ２と直列に
リアクトルあるいは抵抗を設けても同様の効果を得るこ
とができる。Further, based on the same idea, a resistor element can be used instead of the reactor 28, and even if a reactor or a resistor is provided in series with the smoothing capacitor 2, the same effect can be obtained.
第７図は本発明の他の実施例である。第５図に示す第３
実施例と同一物には同じ番号を付しているので説明を省
略する。第３実施例と異なる点は、ＰＷＭコンバータ１
６に対する直列共振回路１８Ａと、ＰＷＭコンバータ６
に対する直列共振回路１８Ｂを設け、各々の直列共振回
路１８Ａ、１８Ｂに流れる電流を電流検出器３Ａ、３Ｂ
で検出し。FIG. 7 shows another embodiment of the invention. 3 shown in Figure 5.
Components that are the same as those in the embodiment are given the same numbers, and therefore their description will be omitted. The difference from the third embodiment is that the PWM converter 1
Series resonant circuit 18A for 6 and PWM converter 6
A series resonant circuit 18B is provided for each of the series resonant circuits 18A and 18B, and currents flowing through the respective series resonant circuits 18A and 18B are detected by current detectors 3A and 3B.
Detected with.
その大きさが最大となるようにＰＷＭコンバータ１６と
ＰＷＭインバータ６の搬送波周波数を制御するようにし
た点である。The point is that the carrier wave frequencies of the PWM converter 16 and the PWM inverter 6 are controlled so that the magnitude thereof is maximized.
本実施例によっても第１実施例及び第２実施例と同様の
効果を得ることができる。This embodiment also provides the same effects as the first and second embodiments.
以上、本発明をＰＷＭインバータとＰＷＭコンバータに
適用した場合の実施例について説明したが、本発明はこ
の他のＰＷＭ動作を行う電力変換装置にも適用できると
共に、電力変換装置の制御構成に限定されるものでなく
、アナログ及びデジイタル制御でも実施することができ
る。The embodiments in which the present invention is applied to a PWM inverter and a PWM converter have been described above, but the present invention can also be applied to power converters that perform other PWM operations, and is not limited to the control configuration of power converters. It can also be implemented with analog and digital control.
第８図において、従来は平滑コンデンサに流れるリプル
電流を許容値以下とするためには、曲線ＡのＰ点より大
きくする必要があったが、本発明の実施例では曲線Ｂの
ように平滑コンデンサに流れ込むリプル電流を低減でき
るので、その容量を小さくすることができる。従って、
本発明によれば、ＰＷＭ電力変換装置の平滑コンデンサ
に流れ込むリプル電流を低減して平滑コンデンサの容量
を小さくすることができるので、装置の小形化ができ、
さらに平滑コンデンサに蓄積されるエネルギーが低減さ
れて電力変換装置の故障時の保護がしやすくなる効果が
ある。In FIG. 8, conventionally, in order to keep the ripple current flowing through the smoothing capacitor below the allowable value, it was necessary to make the ripple current larger than point P of curve A, but in the embodiment of the present invention, as shown in curve B, the smoothing capacitor Since the ripple current flowing into the capacitor can be reduced, its capacitance can be reduced. Therefore,
According to the present invention, it is possible to reduce the ripple current flowing into the smoothing capacitor of a PWM power conversion device and reduce the capacitance of the smoothing capacitor, so the device can be downsized.
Furthermore, the energy accumulated in the smoothing capacitor is reduced, making it easier to protect the power converter in the event of a failure.
第１図は本発明の一実施例を示す制御構成図。
第２図は本発明に用いる直列共振回路の特性を評価する
ための等価回路図、第３図は本発明の動作原理を説明す
る特性図、第４図は本発明の第２実施例を示す制御構成
図、第５図は本発明の第３実施例を示す制御構成図、第
６図は本発明の第４実施例を示す制御構成図、第７図は
本発明の第５実施例を示す制御構成図、第８図は本発明
による平滑コンデンサ容量に対する効果を説明する特性
図である。
２・・・平滑コンデンサ、３・・・電流検出器、６・・
・ＰＷＭインバータ、９・・・トラッキング制御回路、
１０・・・発振器、１６・・・ＰＭＷコンバータ、１８
・・・直列鉄則１図
第２−図
第３図
＋ｃ／ ｆ、ＣＰ、ｕ、Ｊ
第６図
第８図
−平Ａ↑コンｉトン−リース引ｔｃFIG. 1 is a control configuration diagram showing an embodiment of the present invention. Fig. 2 is an equivalent circuit diagram for evaluating the characteristics of the series resonant circuit used in the present invention, Fig. 3 is a characteristic diagram explaining the operating principle of the present invention, and Fig. 4 shows a second embodiment of the present invention. Control configuration diagram, FIG. 5 is a control configuration diagram showing a third embodiment of the present invention, FIG. 6 is a control configuration diagram showing a fourth embodiment of the present invention, and FIG. 7 is a control configuration diagram showing a fifth embodiment of the present invention. The control configuration diagram shown in FIG. 8 is a characteristic diagram illustrating the effect on the capacitance of the smoothing capacitor according to the present invention. 2... Smoothing capacitor, 3... Current detector, 6...
・PWM inverter, 9...tracking control circuit,
10... Oscillator, 16... PMW converter, 18
... Series iron rule 1 Figure 2 - Figure 3 + c/ f, CP, u, J Figure 6 Figure 8 - Flat A ↑ Kon i ton - Lease pull tc
Claims (1)
デンサと並列に直列共振回路を備え、ＰＷＭ制御に伴う
前記電力変換装置の直流電流に含まれるリプル成分の周
波数に前記直列共振回路の共振周波数を所定範囲内に一
致させることを特徴とするＰＷＭ電力変換装置。 ２、ＰＷＭ制御を行う電力変換装置において、平滑コン
デンサと並列に直列共振回路を備え、ＰＷＭ制御に伴う
前記電力変換装置の直流電流に含まれるリプル成分の周
波数に前記直列共振回路の共振周波数を所定範囲内に一
致させるように前記ＰＷＭ制御の搬送波周波数を可変す
るようにしたことを特徴とするＰＷＭ電力変換装置。 ３、ＰＷＭ制御を行う電力変換装置において、平滑コン
デンサと並列に直列共振回路を備え、また前記ＰＷＭ制
御の搬送波周波数を可変する手段を備え、前記搬送波周
波数を前記直列共振回路の共振周波数に一致させるよう
に制御するようにしたことを特徴とするＰＷＭ変換装置
の制御方法。 ４、ＰＷＭ制御を行う電力変換装置において、直流回路
の平滑コンデンサと直列共振回路並びに前記ＰＷＭ制御
の搬送波周波数を可変する手段と前記平滑コンデンサに
流れる電流を検出する手段を備え、該電流検出信号に基
づいて前記ＰＷＭ制御の搬送波周波数を制御するように
したことを特徴とするＰＷＭ電力変換装置の制御方法。 ５、ＰＷＭ制御を行う電力変換装置において、直流回路
の平滑コンデンサと直列共振回路並びに前記ＰＷＭ制御
の搬送波周波数を可変する手段と前記直列共振回路に流
れる電流を検出する手段を備え、該電流検出信号に基づ
いて前記ＰＷＭ制御の搬送波周波数を制御するようにし
たことを特徴とするＰＷＭ電力変換装置の制御方法。 ６、前記平滑コンデンサと並列に直列共振回路を複数組
備え、該各直列共振回路の共振周波数を異ならせたこと
を特徴とする請求項１または請求項２記載のＰＷＭ電力
変換装置。 ７、前記平滑コンデンサにリアクトルあるいは抵抗要素
を直列に接続したことを特徴とする請求項１または請求
項２記載のＰＷＭ電力変換装置。 ８、直流回路の平滑コンデンサより前記電力変換装置に
近い側に前記直列共振回路を接続したことを特徴とする
請求項１または請求項２記載のＰＷＭ電力変換装置。 ９、前記平滑コンデンサの直流回路接続点と前記直列共
振回路の同接続点の間にリアクトル又は抵抗要素を接続
したことを特徴とする請求項８記載のＰＷＭ電力変換装
置。 １０、前記直列共振回路をコンデンサとリアクトルの直
列回路としたことを特徴とする請求項１または請求項２
記載のＰＷＭ電力変換装置。[Scope of Claims] 1. A power conversion device that performs PWM control, which includes a series resonant circuit in parallel with a smoothing capacitor, and generates the series resonance at the frequency of a ripple component included in the DC current of the power conversion device accompanying PWM control. A PWM power conversion device characterized in that the resonant frequency of a circuit is matched within a predetermined range. 2. In a power conversion device that performs PWM control, a series resonant circuit is provided in parallel with a smoothing capacitor, and the resonant frequency of the series resonant circuit is set to a frequency of a ripple component included in a direct current of the power conversion device accompanying PWM control. A PWM power conversion device characterized in that the carrier wave frequency of the PWM control is varied so as to match within a range. 3. A power conversion device that performs PWM control, including a series resonant circuit in parallel with a smoothing capacitor, and means for varying the carrier frequency of the PWM control, so that the carrier frequency matches the resonant frequency of the series resonant circuit. 1. A method for controlling a PWM conversion device, characterized in that the control method is performed as follows. 4. A power conversion device that performs PWM control, comprising a smoothing capacitor of a DC circuit, a series resonant circuit, a means for varying the carrier frequency of the PWM control, and a means for detecting a current flowing through the smoothing capacitor, and a means for detecting a current flowing through the smoothing capacitor. A method for controlling a PWM power conversion device, characterized in that the carrier frequency of the PWM control is controlled based on the PWM control. 5. A power conversion device that performs PWM control, comprising a smoothing capacitor of a DC circuit, a series resonant circuit, a means for varying the carrier frequency of the PWM control, and a means for detecting a current flowing through the series resonant circuit, and a means for detecting a current flowing through the series resonant circuit, and a means for detecting a current flowing through the series resonant circuit. A method for controlling a PWM power conversion device, characterized in that the carrier frequency of the PWM control is controlled based on the following. 6. The PWM power conversion device according to claim 1 or 2, further comprising a plurality of series resonant circuits connected in parallel with the smoothing capacitor, and each series resonant circuit has a different resonance frequency. 7. The PWM power conversion device according to claim 1 or 2, characterized in that a reactor or a resistance element is connected in series to the smoothing capacitor. 8. The PWM power conversion device according to claim 1 or 2, wherein the series resonant circuit is connected to a side closer to the power conversion device than a smoothing capacitor of the DC circuit. 9. The PWM power converter according to claim 8, characterized in that a reactor or a resistance element is connected between the DC circuit connection point of the smoothing capacitor and the same connection point of the series resonant circuit. 10. Claim 1 or Claim 2, wherein the series resonant circuit is a series circuit of a capacitor and a reactor.
The described PWM power conversion device.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP63154748A JP2765857B2 (en) | 1988-06-24 | 1988-06-24 | PWM power converter |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP63154748A JP2765857B2 (en) | 1988-06-24 | 1988-06-24 | PWM power converter |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPH01321868A true JPH01321868A (en) | 1989-12-27 |
| JP2765857B2 JP2765857B2 (en) | 1998-06-18 |
Family
ID=15591049
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP63154748A Expired - Lifetime JP2765857B2 (en) | 1988-06-24 | 1988-06-24 | PWM power converter |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JP2765857B2 (en) |
Cited By (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR100671958B1 (en) * | 2006-07-20 | 2007-01-19 | 주식회사 삼현 | Minimizing torque pulsation type current limit controller for bldc motor |
| JP2009038961A (en) * | 2007-07-06 | 2009-02-19 | Nissan Motor Co Ltd | Power conversion apparatus |
| WO2009147985A1 (en) * | 2008-06-03 | 2009-12-10 | 株式会社村田製作所 | Capacitor circuit and power conversion circuit |
| JP2010252535A (en) * | 2009-04-16 | 2010-11-04 | Nissan Motor Co Ltd | Power conversion apparatus |
| JP2011125114A (en) * | 2009-12-09 | 2011-06-23 | Toyo Electric Mfg Co Ltd | Auxiliary power supply device |
| US8138712B2 (en) | 2007-06-26 | 2012-03-20 | Toyota Jidosha Kabushiki Kaisha | Motor drive system and its control method |
| JP2014030287A (en) * | 2012-07-31 | 2014-02-13 | Denso Corp | Motor drive |
| JP2014528229A (en) * | 2011-07-08 | 2014-10-23 | ローベルト ボッシュ ゲゼルシャフト ミット ベシュレンクテル ハフツング | How to control a multiphase machine |
| CN112072909A (en) * | 2020-09-07 | 2020-12-11 | 电子科技大学 | Drive signal modulation method for inhibiting electromagnetic interference of electric vehicle power module |
Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS5583471A (en) * | 1978-12-16 | 1980-06-23 | Fuji Electric Co Ltd | High frequency invertor |
| JPS63148859A (en) * | 1986-12-12 | 1988-06-21 | Toshiba Corp | Inverter |
-
1988
- 1988-06-24 JP JP63154748A patent/JP2765857B2/en not_active Expired - Lifetime
Patent Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS5583471A (en) * | 1978-12-16 | 1980-06-23 | Fuji Electric Co Ltd | High frequency invertor |
| JPS63148859A (en) * | 1986-12-12 | 1988-06-21 | Toshiba Corp | Inverter |
Cited By (11)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR100671958B1 (en) * | 2006-07-20 | 2007-01-19 | 주식회사 삼현 | Minimizing torque pulsation type current limit controller for bldc motor |
| US8138712B2 (en) | 2007-06-26 | 2012-03-20 | Toyota Jidosha Kabushiki Kaisha | Motor drive system and its control method |
| JP2009038961A (en) * | 2007-07-06 | 2009-02-19 | Nissan Motor Co Ltd | Power conversion apparatus |
| WO2009147985A1 (en) * | 2008-06-03 | 2009-12-10 | 株式会社村田製作所 | Capacitor circuit and power conversion circuit |
| US8531850B2 (en) | 2008-06-03 | 2013-09-10 | Murata Manufacturing Co., Ltd. | Capacitor circuit and power conversion circuit including a resonant frequency adjustment element |
| JP2010252535A (en) * | 2009-04-16 | 2010-11-04 | Nissan Motor Co Ltd | Power conversion apparatus |
| JP2011125114A (en) * | 2009-12-09 | 2011-06-23 | Toyo Electric Mfg Co Ltd | Auxiliary power supply device |
| JP2014528229A (en) * | 2011-07-08 | 2014-10-23 | ローベルト ボッシュ ゲゼルシャフト ミット ベシュレンクテル ハフツング | How to control a multiphase machine |
| JP2014030287A (en) * | 2012-07-31 | 2014-02-13 | Denso Corp | Motor drive |
| US9438160B2 (en) | 2012-07-31 | 2016-09-06 | Denso Corporation | Motor drive apparatus |
| CN112072909A (en) * | 2020-09-07 | 2020-12-11 | 电子科技大学 | Drive signal modulation method for inhibiting electromagnetic interference of electric vehicle power module |
Also Published As
| Publication number | Publication date |
|---|---|
| JP2765857B2 (en) | 1998-06-18 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US4935857A (en) | Transistor conduction-angle control for a series-parallel resonant converter | |
| CA1295672C (en) | Quasi-resonant current mode power conversion method and apparatus | |
| US9099934B2 (en) | Three phase active rectifier system | |
| US7990097B2 (en) | Power conversion system and method for active damping of common mode resonance | |
| US5099410A (en) | Single phase ac power conversion apparatus | |
| US6038152A (en) | Active rectifier utilizing a fixed switching pattern | |
| JP3311743B2 (en) | Control method of resonant inverter | |
| JPH02184267A (en) | Series resonance inverter with non-loss snabber-reset part | |
| Sun et al. | Automatic power decoupling controller of dependent power decoupling circuit for enhanced transient performance | |
| US7042194B1 (en) | Controller for a wound-rotor induction motor | |
| Jacobina et al. | Fault-tolerant reversible AC motor drive system | |
| Jin et al. | Reduction of dc-link capacitance for three-phase three-wire shunt active power filters | |
| JPH01321868A (en) | Pwm electric power converting device and control thereof | |
| Viriya et al. | Low-harmonic GTO converter for fundamental power factor compensation | |
| JP3758062B2 (en) | Active filter control method | |
| JP3509495B2 (en) | Rectifier circuit | |
| JPH0488884A (en) | Voltage type inverter | |
| JP3690558B2 (en) | Multiphase voltage source converter | |
| JP3274208B2 (en) | Deflection control method | |
| JPH0236762A (en) | Control method for pwm power converter | |
| US6225862B1 (en) | Series resonant circuit with inherent short circuit protection | |
| JP2858520B2 (en) | Power factor improving smoothing circuit | |
| JPH09312976A (en) | Controller of voltage type converter | |
| JPH08251938A (en) | Inverter | |
| JPH10164845A (en) | Pwm rectifier |