JP2020078194A - Motor driving method - Google Patents
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- 230000000295 complement effect Effects 0.000 claims description 26
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- 238000005259 measurement Methods 0.000 claims description 6
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- 230000001360 synchronised effect Effects 0.000 claims 1
- 230000005284 excitation Effects 0.000 description 14
- 238000010586 diagram Methods 0.000 description 13
- 238000006243 chemical reaction Methods 0.000 description 5
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- 230000008859 change Effects 0.000 description 3
- 230000008030 elimination Effects 0.000 description 3
- 238000003379 elimination reaction Methods 0.000 description 3
- 238000004804 winding Methods 0.000 description 3
- XEEYBQQBJWHFJM-UHFFFAOYSA-N Iron Chemical compound [Fe] XEEYBQQBJWHFJM-UHFFFAOYSA-N 0.000 description 2
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
- H02P6/16—Circuit arrangements for detecting position
- H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
- H02P6/182—Circuit arrangements for detecting position without separate position detecting elements using back-emf in windings
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
Description
本開示は、例えば三相ブラシレスＤＣモータ等の高効率二相通電を実現する電動機の駆動方法に関する。 The present disclosure relates to a driving method of an electric motor such as a three-phase brushless DC motor that realizes highly efficient two-phase energization.
従来、小型直流モータはブラシ付きＤＣモータが用いられてきたが、ブラシ音・電気ノイズ・耐久性等に問題がありホールセンサで位置検出するブラシレスＤＣモータが登場しさらに位置センサを省略したセンサレスモータも普及しはじめている。小型モータは今後ますます軽量化・低価格化・堅牢化が要求されセンサレスモータ市場が拡大すると思われる。またバッテリー機器においては特に高効率化が求められる。
一方、駆動回路は従来の矩形波駆動から高効率なサイン波駆動に移行しつつある。しかしセンサレスモータのサイン波駆動は始動性や高速性あるいは位相誤差等に難があり矩形波駆動が有利であり、矩形波駆動の高効率化が望まれる。
Conventionally, a brushless DC motor has been used as a small DC motor, but a brushless DC motor that detects a position with a hall sensor has appeared due to problems with brush noise, electric noise, durability, etc., and a sensorless motor without a position sensor. Is also becoming popular. Small motors are expected to expand in the sensorless motor market in the future due to the demand for lighter weight, lower price and robustness. Further, in battery equipment, high efficiency is particularly required.
On the other hand, the drive circuit is shifting from the conventional rectangular wave drive to highly efficient sine wave drive. However, the sine wave drive of a sensorless motor is difficult in starting property, high speed property, phase error, etc., and the rectangular wave drive is advantageous, and high efficiency of the rectangular wave drive is desired.
図８に位置センサを備えないセンサレスモータの一例として三相ブラシレス直流（ＤＣ）モータの構成を示す。回転子軸１を中心に回転する回転子２にはＳ極とＮ極で一対の永久磁石３が設けられている。永久磁石界磁の磁極構造（ＩＰＭ，ＳＰＭ）あるいは極数等は様々である。固定子４には１２０°位相差で設けられた極歯に電機子巻線（コイル）Ｕ，Ｖ，Ｗが配置され、中性点（コモン）Ｃを介してスター結線されている。 FIG. 8 shows the configuration of a three-phase brushless direct current (DC) motor as an example of a sensorless motor that does not include a position sensor. A rotor 2 that rotates around the rotor shaft 1 is provided with a pair of permanent magnets 3 having S and N poles. The magnetic pole structure (IPM, SPM) or the number of poles of the permanent magnet field is various. Armature windings (coils) U, V, and W are arranged on the pole teeth provided at a phase difference of 120 ° on the stator 4, and star-connected via a neutral point (common) C.
図９にセンサレス駆動回路例のブロックダイアグラムを示す。ＭＯＴＯＲは三相センサレスモータである。ＭＰＵ５１はＰＷＭ（Pulse Ｗidth Ｍodulation）制御回路５３やＡＤコンバータ（Analog-to-Digital Converter）５４を内蔵するマイクロコントローラ（制御回路）である。ＩＮＶ５２は、三相ハーフブリッジ型インバータ回路（出力回路）である。ＲＡはコイル電圧の分圧回路で誘起電圧から回転子２の位置を検出する。なお実際の回路にはこのほかに電源部、ホストインターフェース部等が必要であるが煩雑化を避けるため省略してある。 FIG. 9 shows a block diagram of an example of a sensorless drive circuit. MOTOR is a three-phase sensorless motor. The MPU 51 is a microcontroller (control circuit) that includes a PWM (Pulse Width Modulation) control circuit 53 and an AD converter (Analog-to-Digital Converter) 54. The INV 52 is a three-phase half-bridge type inverter circuit (output circuit). RA is a voltage dividing circuit for the coil voltage and detects the position of the rotor 2 from the induced voltage. In addition, the actual circuit requires a power supply unit, a host interface unit, and the like in addition to these, but these are omitted to avoid complication.
図１０に三相ブラシレスＤＣモータの二相通電の代表的な例として１２０°通電のタイミングチャートを示す。区間１はＵ相からＶ相に、区間２はＵ相からＷ相に、区間３はＶ相からＷ相に、区間４はＶ相からＵ相に、区間５はＷ相からＵ相に、区間６はＷ相からＶ相に、矩形波通電される。破線は誘起電圧波形である。ＨＵ〜ＨＷはモータに内蔵されるホールセンサの出力波形であり、位置センサ付きブラシレスＤＣモータはこの信号に基づいて励磁切り替えが行われる。
尚、三相ブラシレスＤＣモータの正弦波駆動を行う際の始動時の相切り替えのブレーキ動作を回避するものとして、以下の文献が存在する（特許文献１：特開２００４−２４２４３２号公報）。
FIG. 10 shows a timing chart of 120 ° energization as a typical example of two-phase energization of a three-phase brushless DC motor. Section 1 is from U phase to V phase, Section 2 is from U phase to W phase, Section 3 is from V phase to W phase, Section 4 is from V phase to U phase, Section 5 is from W phase to U phase, In the section 6, a rectangular wave is energized from the W phase to the V phase. The broken line is the induced voltage waveform. HU to HW are output waveforms of a hall sensor incorporated in the motor, and the brushless DC motor with a position sensor switches excitation based on this signal.
In addition, as a method for avoiding a brake operation for phase switching at the time of starting when performing a sine wave drive of a three-phase brushless DC motor, there is the following document (Patent Document 1: Japanese Patent Laid-Open No. 2004-242432).
サイン波駆動に対し二相矩形波駆動はＰＷＭ通電コイルが少なく効率面で有利であるが実際には効率が劣る。それは以下に述べる開放相ブレーキ電流による機械的損失及びクランプダイオード順方向電圧降下による電気的損失が大きな要因と考えられる。 In contrast to sine wave drive, two-phase rectangular wave drive is advantageous in terms of efficiency because it has few PWM energizing coils, but is actually inferior in efficiency. It is considered that this is largely due to the mechanical loss due to the open phase brake current and the electrical loss due to the forward voltage drop of the clamp diode.
（課題１）開放相ブレーキ電流の解消
図１上段に１２０°通電のコイル電圧及び図１下段にコイル電流の実測波形を示す。ハーフブリッジ構成の出力回路のローサイドアームをＧＮＤ電源電位（以後Ｌと表記する）に固定しハイサイドアームをＰＷＭ駆動した時のＵ相のコイル電圧波形（上図）とコイル電流波形（下図）である。Ｖ相及びＷ相は１２０°位相差で同様の波形となるので図示しない。
コイルを正極電源電位（以後Ｈと表記する）に接続するＨ通電区間ではＰＷＭ周期内の通電期間（以後「ＰＷＭオンサイクル」と言う）は正極電源に接続し、ＰＷＭ周期内の遮断期間（以後「ＰＷＭオフサイクル」という）ではハイインピーダンス状態（以後「Ｚ」あるいは「開放」と言う）とし、スパイク電圧によりＬとなり断続波形が表れている。コイルを負極電源電位に接続するＬ通電区間はＬに固定されている。
Ｈ通電区間とＬ通電区間に挟まれた非通電区間には２本の傾斜した誘起電圧波形が表れており、高電位側の波形はＰＷＭオンサイクルの誘起電圧波形、低電位側の波形はＰＷＭオフサイクルの誘起電圧波形である。
(Problem 1) Elimination of open-phase brake current The upper part of FIG. 1 shows the coil voltage of 120 ° conduction, and the lower part of FIG. 1 shows the measured waveform of the coil current. The coil voltage waveform (upper diagram) and coil current waveform (lower diagram) of the U phase when the low side arm of the output circuit of the half bridge configuration is fixed to the GND power supply potential (hereinafter referred to as L) and the high side arm is PWM driven is there. The V phase and the W phase have the same waveform with a phase difference of 120 °, and are not shown.
In the H energization section in which the coil is connected to the positive electrode power supply potential (hereinafter referred to as H), the energization period in the PWM cycle (hereinafter referred to as “PWM on cycle”) is connected to the positive electrode power supply and the cutoff period in the PWM cycle (hereinafter referred to as “PWM on cycle”). In the "PWM off cycle", a high impedance state (hereinafter referred to as "Z" or "open") is set, and the spike voltage changes to L and an intermittent waveform appears. The L conducting section connecting the coil to the negative power source potential is fixed to L.
Two inclined induced voltage waveforms appear in the non-energized section sandwiched between the H-energized section and the L-energized section. The waveform on the high potential side is the induced voltage waveform of the PWM on-cycle, and the waveform on the low potential side is PWM. It is an off-cycle induced voltage waveform.
ここでコイル電流波形を注意深く観察すると本来は電流がゼロであるべき非通電区間においてパルス状に電流が流れていることが判る。この想定外の電流を矢印で示す。パルス電流は誘起電圧にほぼ比例し非通電区間の始点あるいは終点で最大で非通電区間の中点でゼロとなり、電流極性は駆動電流とは反対となっていることからブレーキとして作用していることが判る。このパルス電流（以後「開放相ブレーキ電流」と言う）が流れるタイミングはコイル電圧波形をみると判るように非通電時のＰＷＭオフサイクルの誘起電圧がＧＮＤ電源電位以下の時である。
このように三相ＢＬＤＣモータのＰＷＭ二相通電において、ＰＷＭオフサイクルの開放相誘起電圧は電源レールを超える期間があり出力素子に並置されたクランプダイオード（ボディーダイオード）により電源にクランプされ、誘起電圧により駆動時とは逆方向の開放相ブレーキ電流が流れる現象が発生している。このブレーキ電流によりモータは瞬間的に制動しながら回転しており無駄な機械的損失が発生しているため効率が低下し振動や騒音が発生している。上述した特許文献１は、矩形波駆動にて回転時の開放相ブレーキ電流を解消するものではない。ＰＷＭオフサイクルの開放相ブレーキ電流の解消に関して、開放相ブレーキ電流を解消する二相通電手法は未だに知見されていない。
Here, careful observation of the coil current waveform reveals that the current flows in a pulsed manner in the non-energized section where the current should originally be zero. This unexpected current is indicated by an arrow. Since the pulse current is almost proportional to the induced voltage and is maximum at the start point or end point of the non-energized section and zero at the midpoint of the non-energized section, and the current polarity is opposite to the drive current, it acts as a brake. I understand. As can be seen from the coil voltage waveform, the timing at which this pulse current (hereinafter referred to as the "open phase brake current") flows is when the induced voltage of the PWM off cycle during non-conduction is below the GND power supply potential.
As described above, in the PWM two-phase conduction of the three-phase BLDC motor, the open-phase induced voltage of the PWM off cycle has a period exceeding the power supply rail and is clamped in the power supply by the clamp diode (body diode) juxtaposed to the output element. Due to this, a phenomenon occurs in which an open phase brake current flows in the opposite direction to that during driving. The brake current causes the motor to rotate while being instantaneously braked, resulting in unnecessary mechanical loss, resulting in reduced efficiency and vibration and noise. The above-mentioned Patent Document 1 does not eliminate the open phase brake current during rotation by the rectangular wave drive. Regarding elimination of the open-phase brake current in the PWM off cycle, a two-phase energization method for eliminating the open-phase brake current has not yet been found.
（課題２）クランプダイオード損失の低減
ＰＷＭ制御のオフサイクル時はコイル蓄積エネルギーによりスパイク電流が流れ、出力素子に並置されたクランプダイオード（ボディーダイオード）を経由して電流が流れるため、クランプダイオードの順方向電圧降下ＶＦの損失が発生する。
図１にクランプダイオードの順方向電圧降ＶＦを図示する。ＶＦは０．６Ｖ以上にもなりスパイク電流が流れる際の損失は大きく、特に小型モータでは電源電圧として１２Ｖが多く用いられ１相あたりの印可電圧は６Ｖと低くＶＦはコイル電圧の１０％にも相当し、この電気的損失により効率が低下するという課題がある。
(Problem 2) Reduction of clamp diode loss During the off cycle of PWM control, a spike current flows due to the energy stored in the coil, and the current flows through the clamp diode (body diode) juxtaposed to the output element. A loss of the directional voltage drop VF occurs.
FIG. 1 illustrates the forward voltage drop VF of the clamp diode. VF becomes 0.6V or more and loss when spike current flows is large. Especially in small motors, 12V is often used as the power supply voltage, and the applied voltage per phase is as low as 6V and VF is 10% of the coil voltage. Correspondingly, there is a problem that efficiency is reduced due to this electrical loss.
以下に述べるいくつかの実施形態に適用される開示は、上記課題を解決すべくなされたものであり、矩形波駆動にてサイン波駆動以上の高効率をめざすものである。
第一の目的は、ＰＷＭオフサイクルにおける通電二相の接続を最適化し開放相ブレーキ電流を解消する二相通電方法（誘起電圧クランプレス通電方法）を用いたモータ駆動方法を提供することにある。
また、第二の目的は、誘起電圧クランプレス通電を行いつつＰＷＭオフサイクルに発生するスパイク電流によるクランプダイオード損失を解消して効率を改善するモータ駆動方法を提供することにある。
The disclosure applied to some embodiments described below has been made in order to solve the above problems, and aims at higher efficiency than that of sine wave driving by rectangular wave driving.
A first object of the present invention is to provide a motor driving method using a two-phase energization method (induced voltage clampless energization method) that optimizes connection of energized two-phases in a PWM off cycle and eliminates open-phase brake current.
A second object of the present invention is to provide a motor driving method for improving the efficiency by eliminating the clamp diode loss due to the spike current generated in the PWM off cycle while performing the induced voltage clampless energization.
永久磁石界磁を有する回転子と三相コイルを有する固定子を備える電動機を、パルス幅変調（ＰＷＭ）方式にて二相通電する電動機の駆動方法であって、前記永久磁石界磁位置を検出あるいは推定する位置検出回路と、三相コイル電圧をＡＤ変換して制御回路に送出する測定回路と、ハーフブリッジ型インバータ回路を介して前記三相コイルに双方向通電する出力回路と、上位コントローラからのトルク指令に基づいてＰＷＭ方式にてコイル出力を制御し、連続回転が可能な通電角度情報と通電パターン情報とを記憶し、前記位置検出回路の出力に基づいて前記出力回路を制御して通電状態を切り替える制御回路と、を備え、ＰＷＭ周期内の通電期間において正極電源に接続する相を電源相、接地電源に接続する相を接地相、ハイインピーダンス（開放）状態とする相を開放相とし、三相の共通接続点電位を中性点電位として、前記制御回路は前記測定回路の出力に応じて当該ＰＷＭ周期内の遮断期間の電源相及び接地相の出力状態を切り替え、開放相の誘起電圧が中性点電位に対して正の時は遮断期間において接地相を接地電源に接続し電源相も接地電源に接続するかまたはハイインピーダンス状態とし、開放相の誘起電圧が中性点電位に対して負の時は遮断期間において電源相を正極電源に接続し接地相も正極電源に接続するかまたはハイインピーダンス状態とすることを特徴とする。
これにより非通電区間において、誘起電圧ゼロクロス点を境界としてＰＷＭオフサイクルの中性点電位がＨとＬとに切り替わり開放相誘起電圧が電源電圧を超えることがなくなり開放相ブレーキ電流を完全に防止することができる。
A method of driving a motor having a rotor having a permanent magnet field and a stator having a three-phase coil, in which two-phase current is applied by a pulse width modulation (PWM) method, and the permanent magnet field position is detected. Alternatively, a position detection circuit for estimating, a measurement circuit for AD-converting the three-phase coil voltage and sending it to the control circuit, an output circuit for bidirectionally energizing the three-phase coil via a half-bridge type inverter circuit, and a host controller The coil output is controlled by the PWM method based on the torque command of No. 2, the energization angle information and the energization pattern information capable of continuous rotation are stored, and the output circuit is controlled based on the output of the position detection circuit to energize. And a control circuit for switching the state, and a phase connected to the positive power source during the energization period within the PWM cycle is a power phase, a phase connected to the ground power source is a ground phase, and a phase in a high impedance (open) state is an open phase. , The common connection point potential of the three phases is set as the neutral point potential, and the control circuit switches the output states of the power supply phase and the ground phase in the cutoff period in the PWM cycle according to the output of the measurement circuit to induce the open phase. When the voltage is positive with respect to the neutral point potential, the ground phase is connected to the ground power supply and the power supply phase is also connected to the ground power supply in the cutoff period, or the high-impedance state is set, and the induced voltage of the open phase becomes the neutral point potential. On the other hand, when it is negative, the power supply phase is connected to the positive power supply and the ground phase is also connected to the positive power supply in the cutoff period, or the high impedance state is set.
As a result, in the non-energized section, the neutral point potential of the PWM off cycle is switched to H and L with the induced voltage zero crossing point as a boundary, and the open phase induced voltage does not exceed the power supply voltage, and the open phase brake current is completely prevented. be able to.
開放相誘起電圧ゼロクロス点を検出するゼロクロス検出回路を設け、１２０°通電における通電区間を前記ゼロクロス点で前方区間と後方区間に分けて電気角を１２区間とし、ＰＷＭ周期の遮断期間において接地相を接地電源に接続し電源相も接地電源に接続またはハイインピーダンス状態とするか、あるいは電源相を正極電源に接続し接地相も正極電源に接続またはハイインピーダンス状態とするかを、前記１２区間に応じて選択するようにしてもよい。
開放相誘起電圧ゼロクロス点を検出するゼロクロス検出回路としては、前述の測定回路（ＡＤコンバータ）にてコイル電圧を測定する方法以外に、ゼロクロスコンパレータ（ゼロクロス検出回路）を用いる方法、あるいは位置センサで検出する方法、あるいは励磁切り替え点から３０°遅延タイマーにより検出する方法など様々ある。これらの方法を用いてゼロクロスを検出すればＡＤコンバータを省略でき回路及び制御ソフトの簡略化が図れる。
A zero-cross detection circuit for detecting the zero-cross point of the open-phase induced voltage is provided, and the energization section at 120 ° energization is divided into the front section and the rear section at the zero-cross point to set the electrical angle to 12 sections, and the ground phase is set in the interruption period of the PWM cycle. Whether to connect to the ground power supply and connect the power supply phase to the ground power supply or be in the high impedance state, or to connect the power supply phase to the positive power supply and also connect the ground phase to the positive power supply or be in the high impedance state, depending on the 12 sections. You may make it select by selecting.
As a zero-crossing detection circuit for detecting the open-phase induced voltage zero-crossing point, a method using a zero-crossing comparator (zero-crossing detection circuit) or a position sensor is used in addition to the method for measuring the coil voltage by the above-mentioned measurement circuit (AD converter). There are various methods, such as a method for detecting the difference, and a method for detecting with a 30 ° delay timer from the excitation switching point. If zero crossing is detected by using these methods, the AD converter can be omitted and the circuit and control software can be simplified.
所謂、相補モードでＰＷＭ制御することにより、クランプダイオードによる損失を低減して効率を改善できる。前記出力回路はスイッチング素子として電界効果トランジスタ（ＦＥＴ）を備え、ＰＷＭ制御回路は、相ごとに前記ハーフブリッジ型インバータ回路のハイサイドアームとローサイドアームを対で制御し、ＰＷＭオフサイクル中はＰＷＭ周期内の通電期間（ＰＷＭオンサイクル）とは逆サイドのアームをオンとする相補モードでＰＷＭ制御を行い、ＰＷＭオンサイクル中はＨ（ハイサイドアームオン）としＰＷＭオフサイクル中はＬ（ローサイドアームオン）とするＨＬ通電と、ＰＷＭオンサイクル中はＬ（ローサイドアームオン）としＰＷＭオフサイクル中はＨ（ハイサイドアームオン）とするＬＨ通電の双方の通電モードを備え、開放相の誘起電圧と中性点電位の大小関係に応じて通電モードを切り換え、開放相の誘起電圧が中性点電位に対して負の期間はＬＨ通電し、正の期間はＨＬ通電することで開放相ブレーキ電流を阻止するようにしてもよい。
上述のようにＨＬ通電とＬＨ通電の二つの通電モードを使う相補ＰＷＭ駆動を行うことで、ＰＷＭオフサイクルのスパイク電流をＦＥＴにより電源レールにクランプすることができ、全期間を通じてクランプダイオードによる損失を解消しつつ開放相ブレーキ電流を阻止する通電を行うことで効率を向上することができる。
The PWM control in the so-called complementary mode can reduce the loss due to the clamp diode and improve the efficiency. The output circuit includes a field effect transistor (FET) as a switching element, and the PWM control circuit controls the high-side arm and the low-side arm of the half-bridge type inverter circuit in pairs for each phase, and a PWM cycle during a PWM off cycle. The PWM control is performed in a complementary mode in which the arm on the side opposite to the energization period (PWM on cycle) is turned on, and H (high side arm on) is set during the PWM on cycle and L (low side arm on) during the PWM off cycle. ) And LH energization mode of L (low side arm on) during PWM on-cycle and H (high side arm on) during PWM off cycle. The energization mode is switched according to the magnitude relationship of the sex point potential, and LH energization is performed when the induced voltage of the open phase is negative with respect to the neutral point potential, and HL energization is performed during the positive period to prevent the open phase brake current. You may do so.
By performing the complementary PWM drive using the two conduction modes of HL conduction and LH conduction as described above, the spike off current of the PWM off cycle can be clamped to the power supply rail by the FET, and the loss due to the clamp diode can be reduced throughout the entire period. The efficiency can be improved by performing the energization for blocking the open-phase brake current while canceling it.
マイクロコントローラに内蔵されるＰＷＭ制御回路からＦＥＴプリドライバに送出される６個のＦＥＴゲート信号のそれぞれに論理を反転する反転回路を設け、あるいは相ごとにハイサイドアームとローサイドアームを入れ替える反転回路を設け、前記マイクロコントローラはＬＨ通電が必要な相に対して、いずれかの前記反転回路へＰＷＭキャリアに同期して反転指令を出力し、当該通電相のＨＬ通電モード状態の２個のＦＥＴゲート信号を論理反転あるいは入れ替えることでＬＨ通電モード状態の信号に変換してＬＨ通電を行うようにしてもよい。
これにより、ＬＨ通電モードを備えていないマイクロコントローラを用いても外付けでハードウェアを追加しＦＥＴゲート信号を反転させることでＬＨ通電を実現し、クランプダイオード損失を解消しなおかつ開放相ブレーキ電流を阻止することができる。
An inversion circuit that inverts the logic for each of the six FET gate signals sent from the PWM control circuit built in the microcontroller to the FET pre-driver is provided, or an inversion circuit that switches the high side arm and the low side arm for each phase. The micro controller outputs an inversion command to one of the inversion circuits in synchronization with a PWM carrier for a phase requiring LH energization, and two FET gate signals in the HL energization mode state of the energized phase. May be converted to a signal in the LH energization mode state by performing logical inversion or replacement, and LH energization may be performed.
As a result, even if a microcontroller not equipped with the LH energization mode is used, LH energization is realized by externally adding hardware and inverting the FET gate signal, eliminating the clamp diode loss and releasing the open phase brake current. Can be stopped.
前記ＰＷＭ制御回路は、開放相の誘起電圧が中性点電位に対して負の時はＰＷＭキャリアに同期して、ＰＷＭ制御モードをＰＷＭオンサイクル中はＬ（ローサイドアームオン）、ＰＷＭオフサイクル中はＺ（ハイインピーダンス状態）とする独立モードに切り替えてＬＺ通電にてＰＷＭ制御し、開放相の誘起電圧が中性点電位に対して正の時はＰＷＭキャリアに同期して、ＰＷＭ制御モードをＰＷＭオンサイクル中はＨ（ハイサイドアームオン）、ＰＷＭオフサイクル中はＬ（ローサイドアームオン）とする相補モードに切り替えてＨＬ通電にてＰＷＭ制御するようにしてもよい。
これによりスパイク電流はＦＥＴを経由して流れクランプダイオードを経由する期間を半分にすることができ、ソフトウェアの変更のみでクランプダイオード損失を半減しつつ開放相ブレーキ電流を阻止することができる。
The PWM control circuit synchronizes with the PWM carrier when the induced voltage of the open phase is negative with respect to the neutral point potential, and sets the PWM control mode to L (low side arm on) during the PWM on cycle and during the PWM off cycle. Switches to the independent mode for Z (high impedance state) and performs PWM control by LZ energization. When the induced voltage in the open phase is positive with respect to the neutral point potential, the PWM control mode is switched in synchronization with the PWM carrier. The PWM control may be performed by switching to a complementary mode of H (high side arm on) during the PWM on cycle and L (low side arm on) during the PWM off cycle, and performing HL energization for PWM control.
As a result, the spike current flows through the FET and the period through the clamp diode can be halved, and the open-phase brake current can be blocked while halving the clamp diode loss by only changing the software.
上述した電動機の駆動方法を用いれば、効率を損なうＰＷＭオフサイクルの開放相ブレーキ電流を完全に解消でき効率が向上する。また上述したＰＷＭ制御方法を用いれば、ＰＷＭオフサイクルのクランプダイオード損失を完全に解消または半減でき効率が向上する。これらの制御には複雑な演算を必要とせず演算時間が短くて済むことからＰＷＭキャリア周波数を上げることも可能で、磁気回路の鉄損を減らすことで効率を向上できる。さらに通電開始位相角を前方に進める進角制御あるいは三相通電期間を挿入して通電角度を拡大するオーバーラップ通電なども可能で効率を向上しあるいは低振動化・静音化できる。
以上によりモータ効率が向上する結果、消費電力が削減でき、同じ電源電圧でも最高回転数が高くなる。
また、開放相の誘起電圧ゼロクロス点及び区間終点は誘起電圧及びインダクタンス変化から検出可能であることからセンサレス駆動に適し、位置検出のためのリーケージフラックスが不要となり磁気回路損失を減らすことができること、また位置センサ自体の消費電力の削減や位置センサでは取り除けない着磁誤差による励磁切り替えタイミング誤差によるトルク発生効率の低下を低減できることなどからもモータ効率が改善される。また、ゼロクロスコンパレータ（ゼロクロス検出回路）を用いＡＤコンバータを省略することも可能で回路を簡素化できる。
あるいは三相サイン波通電のＰＷＭ制御回路ではセンターアライメント方式のデューティコントローラ及びデッドタイムコントローラが３チャンネル必要であったが、本案は二相矩形波通電であり３チャンネルを１チャンネルに減らすことができ、しかもエッジアライメント方式のため簡素な構成のデューティコントローラで済み回路を簡略化できる。
By using the above-described method of driving the electric motor, the open phase brake current in the PWM off cycle, which impairs efficiency, can be completely eliminated, and the efficiency is improved. Further, if the PWM control method described above is used, the clamp diode loss in the PWM off cycle can be completely eliminated or halved, and the efficiency can be improved. Since these controls do not require complicated calculations and the calculation time is short, it is possible to increase the PWM carrier frequency, and the efficiency can be improved by reducing the iron loss of the magnetic circuit. Further, advance control for advancing the energization start phase angle forward or overlap energization for enlarging the energization angle by inserting a three-phase energization period can be performed to improve efficiency or reduce vibration and noise.
As a result of improving the motor efficiency as described above, the power consumption can be reduced and the maximum rotation speed can be increased even with the same power supply voltage.
In addition, the zero-cross point and the end point of the induced voltage in the open phase can be detected from the induced voltage and the change in the inductance, so that they are suitable for sensorless driving, and the leakage flux for position detection is unnecessary, and the magnetic circuit loss can be reduced. The motor efficiency is also improved by reducing the power consumption of the position sensor itself and reducing the decrease in the torque generation efficiency due to the excitation switching timing error due to the magnetization error that cannot be removed by the position sensor. Further, the AD converter can be omitted by using a zero-cross comparator (zero-cross detection circuit), and the circuit can be simplified.
Alternatively, the PWM control circuit for three-phase sine wave energization required three channels for the center alignment type duty controller and dead time controller, but the present invention uses two-phase rectangular wave energization and can reduce three channels to one channel. Moreover, since the edge alignment method is used, the duty controller having a simple structure can simplify the circuit.
以下、本発明に係る電動機の高効率駆動方法の実施形態について、添付図面を参照しながら説明する。本願発明は、電動機の一例として、回転子に永久磁石界磁を備え、固定子に巻き線を１２０°位相差で配置してスター結線し、相端がモータ出力回路に接続されたＢＬＤＣモータがあげられ、ここでは近年利用が拡大している位置センサレスモータを用いて説明する。 An embodiment of a high-efficiency driving method for an electric motor according to the present invention will be described below with reference to the accompanying drawings. As an example of an electric motor, the present invention provides a BLDC motor in which a rotor is provided with a permanent magnet field, windings are arranged on a stator at a phase difference of 120 ° and star-connected, and phase ends are connected to a motor output circuit. The position sensorless motor, which has been widely used in recent years, will be described below.
図８を参照して３相ＢＬＤＣセンサレスモータの一実施例を示す。一例として２極永久磁石ローターと３スロットを設けた固定子４を備えた３相ブラシレスＤＣモータを例示する。モータはインナーローター型でもアウターローター型でもいずれでもよい。また、永久磁石型界磁としては永久磁石埋め込み型（ＩＰＭ）モータや表面永久磁石型（ＳＰＭ）モータのいずれであってもよい。 An example of a three-phase BLDC sensorless motor will be described with reference to FIG. As an example, a three-phase brushless DC motor including a two-pole permanent magnet rotor and a stator 4 provided with three slots will be illustrated. The motor may be either an inner rotor type or an outer rotor type. The permanent magnet type field may be either a permanent magnet embedded type (IPM) motor or a surface permanent magnet type (SPM) motor.
図８において、回転子軸１には回転子２が一体に設けられ、界磁として２極の永久磁石３が設けられている。固定子４には１２０°位相差で極歯Ｕ，Ｖ，Ｗが永久磁石３に対向して配置されている。固定子４の各極歯Ｕ，Ｖ，Ｗに巻線ｕ，ｖ，ｗを設けて相間をコモンＣでスター結線して後述するモータ駆動装置に配線された３相ブラシレスＤＣモータとなっている。尚、コモン線は、不要であるので省略されている。 In FIG. 8, a rotor 2 is integrally provided on a rotor shaft 1, and a two-pole permanent magnet 3 is provided as a field magnet. On the stator 4, pole teeth U, V, W are arranged facing the permanent magnet 3 with a phase difference of 120 °. A winding u, v, w is provided on each pole tooth U, V, W of the stator 4 and the phases are star-connected with a common C to form a three-phase brushless DC motor wired in a motor drive device described later. .. The common line is omitted because it is unnecessary.
次に、図９に示す三相センサレスモータ駆動回路ブロック図を参照して説明する。本案はＡＤコンバータを用いたセンサレスモータ駆動回路でも実現できる。ＭＯＴＯＲは三相センサレスモータである。ＭＰＵ５１はマイクロコントローラ（制御回路）である。ＭＰＵ５１は、三相コイル（Ｕ，Ｖ，Ｗ）に対する６通りの通電パターンと各通電パターンに対応する１２０°通電の励磁切り替え区間（区間１〜区間６）を指定する界磁位置情報を記憶し、上位コントローラ５０からのトルク指令に応じて後述する出力回路（ハーフブリッジ型インバータ回路５２）をスイッチング制御して励磁状態を任意に切り替える。また、ＭＰＵ５１は、誘起電圧クランプレス通電が可能な反転ＰＷＭ制御回路５３及びＡＤ変換回路５４（ＡＤＣ：ＡＤコンコバータ（測定回路））を内蔵する。 Next, description will be made with reference to the block diagram of the three-phase sensorless motor drive circuit shown in FIG. The present invention can also be realized by a sensorless motor drive circuit using an AD converter. MOTOR is a three-phase sensorless motor. The MPU 51 is a microcontroller (control circuit). The MPU 51 stores six types of energization patterns for the three-phase coils (U, V, W) and field position information that specifies the 120 ° energization excitation switching sections (section 1 to section 6) corresponding to each energization pattern. In accordance with a torque command from the host controller 50, the output circuit (half bridge type inverter circuit 52) described later is switching-controlled to arbitrarily switch the excitation state. Further, the MPU 51 incorporates an inversion PWM control circuit 53 and an AD conversion circuit 54 (ADC: AD converter (measurement circuit)) capable of energizing the induced voltage clampless energization.
ハーフブリッジ型インバータ回路５２（ＩＮＶ：出力回路）は、三相コイルに通電し、モータトルクを制御するために励磁相切り替えあるいはＰＷＭ制御などのスイッチング動作を行う。上記ハーフブリッジ型インバータ回路５２は、スイッチング素子として電界効果トランジスタ（ＦＥＴ）及びこれに逆並列に接続されるダイオードを備え、正極電源ライン及び接地電源ラインに任意に接続可能なハーフブリッジ型スイッチング回路が３相分設けられている。 The half-bridge inverter circuit 52 (INV: output circuit) energizes the three-phase coil and performs switching operation such as excitation phase switching or PWM control to control the motor torque. The half-bridge type inverter circuit 52 includes a field-effect transistor (FET) as a switching element and a diode connected in antiparallel to the field-effect transistor (FET), and a half-bridge type switching circuit that can be arbitrarily connected to a positive power source line and a ground power source line is provided. Three phases are provided.
ＡＤコンバータ５４（ＡＤＣ）は、分圧回路ＲＡ（位置検出回路）を介してコイル出力端子Ｕ，Ｖ，Ｗが接続され、制御回路（ＭＰＵ５１）からの変換開始信号により三相それぞれのコイル電圧を同時サンプリングし、順次アナログ・デジタル変換し、変換結果を制御回路（ＭＰＵ５１）に送出する。通常ＡＤコンバータ５４はＭＰＵ５１に内蔵されており、内蔵ＡＤコンバータ５４を利用する場合は最大入力電圧が低いため抵抗による分圧回路ＲＡを設けることが望ましい。 The AD converter 54 (ADC) is connected to the coil output terminals U, V, W via a voltage dividing circuit RA (position detection circuit), and outputs a coil voltage for each of the three phases in response to a conversion start signal from the control circuit (MPU51). Simultaneous sampling is performed, analog / digital conversion is sequentially performed, and the conversion result is sent to the control circuit (MPU51). Normally, the AD converter 54 is built in the MPU 51. When the built-in AD converter 54 is used, the maximum input voltage is low, so that it is desirable to provide the voltage dividing circuit RA by resistance.
上記センサレスモータを駆動する通電方式はセンサ付きモータと同様であり、図１０を参照しながら代表的な通電方式である１２０°通電について説明する。
１２０°通電では相ごとに６０°の非通電区間を挟んで正負１２０°の通電区間があり、相ごとに１２０°の位相差を持っている。従って１電気角は６０°単位の６ステップで区切られ、区間１から６へＵ−Ｖ、Ｕ−Ｗ、Ｖ−Ｗ，Ｖ−Ｕ，Ｗ−Ｕ，Ｗ−Ｖと励磁される。Ｕ−Ｖ励磁とはＵ相が正極電源にＶ相がＧＮＤ電源に接続されることを表す。
The energization method for driving the sensorless motor is the same as that for the sensor-equipped motor, and 120 ° energization, which is a typical energization method, will be described with reference to FIG.
In the 120 ° energization, there is a positive / negative 120 ° energization section across a 60 ° non-energization section for each phase, and there is a phase difference of 120 ° for each phase. Therefore, one electrical angle is divided into 6 steps of 60 °, and sections 1 to 6 are excited as UV, UW, VW, VU, WU, and WV. U-V excitation means that the U phase is connected to the positive power source and the V phase is connected to the GND power source.
また、非通電区間の中間で誘起電圧は正負が切り替わるいわゆるゼロクロス点が発生する。センサレスモータではこのゼロクロス点を検出してタイマーを用いて３０°ディレーを設けゼロクロスコンパレータ５５（ゼロクロス検出回路）により励磁切り替えを行う位置センサレス駆動が多用されており（ゼロクロスコンパレータ方式：図１１参照）、位置センサ付きブラシレスＤＣモータはホールセンサ出力ＨＵ〜ＨＷにより励磁切り替え点を検出して励磁切り替えが行われる方式が主流である。
尚、位置センサレス駆動でも励磁切り替え点を直接検出或いは推定しセンサ付きと同等の閉ループ制御が可能な手法も提案されている。それによれば進角を設ける場合も進み位相の励磁切り替え点を直接検出或いは推定することが可能であり、ゼロクロス点から３０°遅延タイマーにより励磁切り替え点を検出するよりも位置誤差が少なく速度変動に対しても有利であり制御プログラムも簡略化できる。
In addition, a so-called zero-cross point at which the positive and negative of the induced voltage are switched occurs in the middle of the non-energized section. In the sensorless motor, the position sensorless drive in which the zero cross point is detected and a 30 ° delay is provided using a timer to switch the excitation by the zero cross comparator 55 (zero cross detection circuit) is often used (zero cross comparator method: see FIG. 11). The mainstream of the brushless DC motor with a position sensor is a method in which the excitation switching point is detected by the Hall sensor outputs HU to HW and the excitation switching is performed.
A method has also been proposed in which the excitation switching point is directly detected or estimated even in the position sensorless drive, and closed loop control equivalent to that with a sensor is possible. According to this, it is possible to directly detect or estimate the excitation switching point of the lead phase even when the advance angle is provided, and there is less position error than when the excitation switching point is detected by the 30 ° delay timer from the zero cross point, resulting in speed fluctuation. It is also advantageous to the control program and can be simplified.
図１１に一例としてゼロクロスコンパレータ５５（位置検出回路）を用いる駆動回路のブロック構成図を示す。なお区間終点はゼロクロス点から３０°遅延タイマーにて検出でき、あるいはホールセンサ等の位置センサを用いてもよい（図示せず）。
図９と同一部材には同一符号を付して説明を援用するものとし、異なる点のみを説明する。ＺＥＲＯは相ごとのゼロクロスコンパレータ５５である。中性点として三相を抵抗で合成したＣＯＭ（ダミーコモン）を用い、各相のコイル電圧と比較される。ゼロクロスコンパレータ出力はＭＰＵ５１へ送出され、ＭＰＵ５１内にＡＤコンバータは不要である。
FIG. 11 shows a block diagram of a drive circuit using the zero-cross comparator 55 (position detection circuit) as an example. The end point of the section can be detected by a 30 ° delay timer from the zero cross point, or a position sensor such as a hall sensor may be used (not shown).
The same members as those in FIG. 9 are designated by the same reference numerals and the description thereof is cited, and only different points will be described. ZERO is a zero cross comparator 55 for each phase. COM (dummy common) in which three phases are combined by resistance is used as a neutral point, and is compared with the coil voltage of each phase. The output of the zero-cross comparator is sent to the MPU 51, and no AD converter is required in the MPU 51.
開放相誘起電圧ゼロクロス点と区間終点が判れば、１２０°通電における通電区間をゼロクロス点で分割して１２個の区間に分けることができる。上述のごとく通電パターンはゼロクロス点を境界として切り替わることから、区間をゼロクロス点で分けることで区間ごとに通電パターンを決定することができる。従って１２分割された区間ごとにあらかじめ通電パターンを記憶しておき、区間に応じて通電パターンを切り換えるだけで誘起電圧クランプレス通電を実現できる。 If the open-phase induced voltage zero cross point and the section end point are known, the energization section at 120 ° energization can be divided at the zero cross point into 12 sections. As described above, the energization pattern is switched with the zero-cross point as the boundary, and thus the energization pattern can be determined for each section by dividing the section at the zero-cross point. Therefore, the induced voltage clampless energization can be realized by storing the energization pattern in advance for each of the 12 divided sections and switching the energization pattern according to the section.
表１に１２区間の通電パターンを示す。
（表１）１２区間の通電パターン
注１:通電欄
丸付き数字１〜６は１２０°通電の６区間の区間番号に対応している。
「ＵＶ」はＵ相をＨ通電しＶ相をＬ通電することを表す。ＵＷ〜ＷＶも同様である。「前」及び「後」は、ゼロクロス点の前方区間及び後方区間を指す。
注２：Ｕ相〜Ｗ相欄
Ｈは、Ｈ固定通電を表しＰＷＭ周期を通じて出力される。
Ｌは、Ｌ固定通電を表しＰＷＭ周期を通じて出力される。
Ｚは、非通電状態を表しＰＷＭ周期を通じて出力される。
ＨＬは、ＰＷＭオンサイクル中はＨとしＰＷＭオフサイクル中はＬとするＨＬ通電を表しＰＷＭオフサイクル中はＺとするＨＺ通電も含む。
ＬＨは、ＰＷＭオンサイクル中はＬとしＰＷＭオフサイクル中はＨとするＬＨ通電を表しＰＷＭオフサイクル中はＺとするＬＺ通電も含む。
Table 1 shows the energization patterns for the 12 sections.
(Table 1) 12 sections of energization pattern
Note 1: Energization column Circled numbers 1 to 6 correspond to the section numbers of the 6 sections of 120 ° energization.
“UV” means that the U phase is energized with H and the V phase is energized with L. The same applies to UW to WV. "Front" and "rear" refer to the front section and the rear section of the zero-cross point.
Note 2: U-phase to W-phase column H represents H fixed energization and is output through the PWM cycle.
L represents L fixed energization and is output through the PWM cycle.
Z represents a non-energized state and is output through the PWM cycle.
HL represents HL energization that is H during the PWM on cycle and L during the PWM off cycle, and also includes HZ energization that is Z during the PWM off cycle.
LH represents LH energization that is L during the PWM on cycle and H during the PWM off cycle, and also includes LZ energization that is Z during the PWM off cycle.
（誘起電圧クランプレス通電の実施例）
誘起電圧クランプレス通電は、ＰＷＭオフサイクルの開放相ブレーキ電流の解消方法である。具体的には、ＰＷＭオンサイクル時に三相のコイル電圧を測定し、通電２相の平均電圧（＝中性点電位）と開放相電圧（＝誘起電圧）の大小比較をする。中性点電位より開放相電圧が低いときはＬＨ通電（相補モード）かＬＺ通電（独立モード）とし、高いときはＨＬ通電（相補モード）かＨＺ通電（独立モード）とする。これによりＰＷＭオフサイクルの中性点電位がゼロクロス点を境にＬとＨとで切り替わり、開放相誘起電圧は電源レール内に収まり開放相ブレーキ電流は流れなくなる。
(Example of energizing induced voltage clampless)
Induction voltage clampless energization is a method of eliminating the open phase brake current in the PWM off cycle. Specifically, three-phase coil voltages are measured during the PWM on-cycle, and the magnitude comparison between the average voltage (= neutral point potential) and the open-phase voltage (= induced voltage) of the two energized phases is performed. When the open phase voltage is lower than the neutral point potential, it is LH conduction (complementary mode) or LZ conduction (independent mode), and when it is high, it is HL conduction (complementary mode) or HZ conduction (independent mode). As a result, the neutral point potential of the PWM off cycle switches between L and H at the zero cross point, the open phase induced voltage is contained in the power supply rail, and the open phase brake current does not flow.
図２に非通電区間に現れる開放相誘起電圧波形の模式図を示す。図中央の太い破線はＰＷＭオンサイクル時の誘起電圧波形である。その上下の破線はＰＷＭオフサイクル時の誘起電圧波形で、ゼロクロスより前方期間ではＰＷＭオフサイクル時に通電二相をＬとしたときの誘起電圧でありＬを基準に発生し、ゼロクロスより後方期間ではＰＷＭオフサイクル時に通電二相をＨとしたときの誘起電圧でありＨを基準に発生するものとした。図２からＰＷＭオンサイクル及びＰＷＭオフサイクルとも誘起電圧は＋Ｖ電位及びＧＮＤ電位からなる電源レールを超えないことは明白であり従って開放相にブレーキ電流が流れることはない。 FIG. 2 shows a schematic diagram of an open phase induced voltage waveform that appears in the non-energized section. The thick broken line in the center of the figure is the induced voltage waveform during the PWM on-cycle. The dashed lines above and below that are the induced voltage waveforms during the PWM off cycle. In the period before the zero cross, the induced voltage is when L is the two energized phases during the PWM off cycle. It is an induced voltage when the energized two phases are set to H during the off cycle, and is generated based on H. From FIG. 2 it is clear that in both the PWM on-cycle and the PWM off-cycle the induced voltage does not exceed the power supply rail consisting of the + V potential and the GND potential, so no brake current flows in the open phase.
なお上述のとおりＰＷＭオフサイクル時に通電相をハイインピーダンス状態とする動作も含むことでいわゆる独立モードも許容する。その理由はＰＷＭオフサイクルになって通電相が開放されるとスパイク電圧が発生し正極電源近傍あるいはＧＮＤ電源近傍にクランプされてＨあるいはＬを出力した時とほぼ同じ電位となり、開放相誘起電圧が電源電圧を大きく超えることがなくなり開放相ブレーキ電流を防止できるからである。ただし、この場合はスパイク電流によりクランプダイオード損失が発生する。 As described above, the so-called independent mode is allowed by including the operation of putting the energized phase in the high impedance state during the PWM off cycle. The reason is that when the PWM off cycle occurs and the energized phase is released, a spike voltage is generated and the potential is almost the same as when H or L is output by being clamped near the positive power source or near the GND power source, and the open phase induced voltage is This is because the power supply voltage is not greatly exceeded and the open phase brake current can be prevented. However, in this case, the clamp diode loss occurs due to the spike current.
図３に相補モードＰＷＭ制御にて実際に動作させたときの実施波形例を示す。三相のうちのＵ相について図示してあり、上段はコイル電圧波形、下段はコイル電流波形である。残る二相も１２０°位相差で同様の波形となるので省略する。
コイル電流ゼロの水平な直線部分に着目すると、この区間は非通電区間であり従来は図１で示したとおりパルス状のブレーキ電流が流れていたが図から明らかなように全く流れていないことが判る。それは誘起電圧クランプレス通電が行われているからでありこれにより開放相ブレーキ電流が完全に解消されていることが検証できる。
FIG. 3 shows an example of an implementation waveform when actually operating in complementary mode PWM control. The U phase of the three phases is shown in the figure. The upper stage shows the coil voltage waveform and the lower stage shows the coil current waveform. The remaining two phases have the same waveform with a phase difference of 120 °, and are omitted.
Focusing on the horizontal straight line portion where the coil current is zero, this section is a non-energized section, and in the past, a pulsed brake current was flowing as shown in FIG. 1, but as is clear from the figure, it does not flow at all. I understand. This is because the induced voltage clampless energization is performed, and it can be verified that the open phase brake current is completely eliminated by this.
ここで開放相ブレーキ電流について詳しく説明する。二相通電では１相をＰＷＭ制御しもう１相を正極電源あるいはＧＮＤ電源に固定して通電が行われる。従って、ＰＷＭオンサイクルの中性点電位はほぼ電源電圧の半分となる。一方、ＰＷＭオフサイクルもコイル蓄積エネルギーによりＰＷＭオンサイクルと同じ方向に電流が流れており、ＰＷＭオフサイクルの通電二相は同極電源に接続されるため中性点電位はほぼ正極電源電圧あるいはほぼＧＮＤ電源電圧となる。 Here, the open phase brake current will be described in detail. In the two-phase energization, one phase is PWM-controlled and the other phase is fixed to a positive power source or a GND power source and energized. Therefore, the neutral point potential of the PWM on-cycle becomes almost half of the power supply voltage. On the other hand, in the PWM off cycle, the current flows in the same direction as the PWM on cycle due to the energy stored in the coil, and since the energized two phases of the PWM off cycle are connected to the same polarity power supply, the neutral point potential is almost the positive power supply voltage or almost the same. It becomes the GND power supply voltage.
開放相端には中性点電位を基準として正負に誘起電圧が発生し、ＰＷＭオフサイクル時の誘起電圧は正極電源電圧あるいはＧＮＤ電源電圧を基準に振れる。そのため開放相誘起電圧のゼロクロス点より前方期間あるいは後方期間で開放相電圧は電源電圧を超えることとなり、電源電圧±クランプダイオード順方向電圧降下ＶＦを超えるとクランプダイオードを経由して開放相に電流が流れる。この電流は励磁電流とは逆方向に流れるため制動作用となることから本案では開放相ブレーキ電流と呼んでいる。 Positive and negative induced voltages are generated at the open phase ends with reference to the neutral point potential, and the induced voltage during the PWM off cycle fluctuates with the positive power supply voltage or the GND power supply voltage as a reference. Therefore, the open-phase voltage exceeds the power supply voltage in the period before or after the zero-cross point of the open-phase induced voltage, and when the power supply voltage ± clamp diode forward voltage drop VF is exceeded, the current flows to the open phase via the clamp diode. Flowing. Since this current flows in the opposite direction to the exciting current and has a braking action, it is called an open-phase braking current in this proposal.
図１２にＰＷＭオフサイクルに通電相を正極電源に接続した時の電流経路図を示す。
Ｌ１〜Ｌ３はコイル、ＣＯＭは中性点、＋Ｖは正極電源、Ｑ１及びＱ２はハイサイドアーム出力素子、Ｄ１〜Ｄ３はクランプダイオードである。通電する二相コイルＬ１及びＬ２を正極電源に接続するＨＨ接続は、ハイサイドアーム出力素子Ｑ１及びＱ２を通じて行われる。コイルＬ３を開放相とするＨＺ接続は、クランプダイオードＤ１又はＤ２を通じて正極電源＋Ｖに接続されスパイク電圧の発生により中性点ＣＯＭもＨとなり、開放相コイルＬ３の誘起電圧がＨを超えるとクランプダイオードＤ３を経由して開放相コイルＬ３から正極電源＋Ｖにスパイク電流が流れる。
FIG. 12 shows a current path diagram when the energized phase is connected to the positive electrode power source in the PWM off cycle.
L1 to L3 are coils, COM is a neutral point, + V is a positive power source, Q1 and Q2 are high side arm output elements, and D1 to D3 are clamp diodes. The HH connection for connecting the energized two-phase coils L1 and L2 to the positive power source is performed through the high side arm output elements Q1 and Q2. The HZ connection in which the coil L3 is in the open phase is connected to the positive power source + V through the clamp diode D1 or D2, the neutral point COM also becomes H due to the generation of the spike voltage, and the clamp diode is generated when the induced voltage of the open phase coil L3 exceeds H. A spike current flows from the open phase coil L3 to the positive power source + V via D3.
図１３にＰＷＭオフサイクルに通電相をＧＮＤ電源に接続した時の電流経路図を示す。符号は図１２を援用する。Ｑ１１及びＱ１２はローサイドアーム出力素子、Ｄ１１〜Ｄ１３はクランプダイオードである。通電する二相コイルＬ１及びＬ２をＧＮＤ電源に接続するＬＬ接続は、ローサイドアーム出力素子Ｑ１１及びＱ１２を通じて行われる。コイルＬ３を開放相とするＬＺ接続は、クランプダイオードＤ１１又はＤ１２を通じてＧＮＤに接続されスパイク電圧の発生によりＣＯＭもＬとなり、開放相コイルＬ３の誘起電圧がＬを超えるとクランプダイオードＤ１３を経由してＧＮＤ電源から開放相コイルＬ３にスパイク電流が流れる。 FIG. 13 shows a current path diagram when the energized phase is connected to the GND power supply in the PWM off cycle. For the reference numerals, FIG. 12 is used. Q11 and Q12 are low side arm output elements, and D11 to D13 are clamp diodes. The LL connection for connecting the energized two-phase coils L1 and L2 to the GND power source is performed through the low side arm output elements Q11 and Q12. The LZ connection in which the coil L3 is in the open phase is connected to the GND through the clamp diode D11 or D12, and COM also becomes L due to the generation of the spike voltage. A spike current flows from the GND power supply to the open phase coil L3.
以上から開放相ブレーキ電流が流れる原因は中性点電位が電源電位になりそのため開放相誘起電圧が電源電圧を超えてしまうことにあることが判った。従って開放相ブレーキ電流を阻止するためにはＰＷＭオフサイクルの中性点電位を制御すればよい。即ち、開放相の誘起電圧が中性点電位に対して正の時は中性点電位をＧＮＤ電源電圧に、負の時は正極電源電圧にすれば誘起電圧が電源電圧を超えることを回避でき開放相ブレーキ電流は流れない。そこで開放相誘起電圧が中性点電位に対して正の時はＰＷＭオフサイクル時に通電二相をＧＮＤ電源に接続（ＬＬ接続）あるいは一相はＬに接続し他相はハイインピーダンス状態（ＬＺ接続）とし、負の時は通電二相を正極電源に接続（ＨＨ接続）あるいは一相はＨに接続し他相はハイインピーダンス状態（ＨＺ接続）とする。これにより非通電区間において、誘起電圧ゼロクロス点を境界として中性点電位が正極電源とＧＮＤ電源とに切り替わり、誘起電圧が電源電圧を超えることはなくなり開放相ブレーキ電流を完全に防止することができる。 From the above, it was found that the cause of the open phase brake current is that the neutral point potential becomes the power supply potential, and therefore the open phase induced voltage exceeds the power supply voltage. Therefore, in order to prevent the open phase brake current, the neutral point potential of the PWM off cycle may be controlled. That is, when the induced voltage in the open phase is positive with respect to the neutral point potential, the neutral point potential is set to the GND power supply voltage, and when the negative voltage is set to the positive power supply voltage, the induced voltage can be prevented from exceeding the power supply voltage. Open phase brake current does not flow. Therefore, when the open-phase induced voltage is positive with respect to the neutral point potential, the energized two phases are connected to the GND power supply (LL connection) during the PWM off cycle, or one phase is connected to L and the other phase is in a high impedance state (LZ connection). ), When negative, the two energized phases are connected to the positive power source (HH connection), or one phase is connected to H and the other phase is in a high impedance state (HZ connection). Thus, in the non-energized section, the neutral point potential is switched between the positive power source and the GND power source with the induced voltage zero crossing point as a boundary, the induced voltage does not exceed the power source voltage, and the open phase brake current can be completely prevented. ..
（相補モードＰＷＭ制御によるクランプダイオード損失の解消）
ハーフブリッジ型インバータ回路５２のインバータ出力素子を電界効果トランジスタ（ＦＥＴ）にて構成し相補ＰＷＭモードで通電すると、ＦＥＴは逆方向にも電流が流せることからスパイク電流はクランプダイオードを経由せずＦＥＴを通って電源レールにクランプされる。
ＦＥＴのオン抵抗は通常数ｍΩ〜数十ｍΩと小さいのでスパイク電流による電圧降下は非常に小さくそのためクランプダイオード損失に比べてＦＥＴ損失は小さくなり損失を抑えることができる。
(Elimination of clamp diode loss by complementary mode PWM control)
When the inverter output element of the half-bridge type inverter circuit 52 is composed of a field effect transistor (FET) and energized in the complementary PWM mode, a current can flow through the FET in the opposite direction, so that the spike current does not pass through the clamp diode and the FET Through it is clamped to the power rail.
Since the ON resistance of the FET is usually as small as several mΩ to several tens of mΩ, the voltage drop due to the spike current is very small. Therefore, the FET loss is smaller than the clamp diode loss and the loss can be suppressed.
図３に誘起電圧クランプレス通電による実施波形例が示されており、開放相ブレーキ電流が完全に解消されている。ここで図３の上段コイル電圧波形の電源レール部に着目すると、ノイズを除くすべての波形が電源電圧内に収まっている。これは即ち出力素子ＦＥＴによりスパイク電圧がクランプされていることを意味しておりクランプダイオード損失も完全に解消されていることが判る。以下にその通電制御方法を詳しく説明する。 FIG. 3 shows an example of the waveform implemented by the induced voltage clampless energization, in which the open phase brake current is completely eliminated. Here, focusing on the power supply rail portion of the upper coil voltage waveform in FIG. 3, all waveforms except noise are within the power supply voltage. This means that the spike voltage is clamped by the output element FET, and the clamp diode loss is completely eliminated. The energization control method will be described in detail below.
相補モードＰＷＭ制御時はＰＷＭオンサイクル中をＨとしＰＷＭオフサイクル中をＬとするＨＬ通電と、ＰＷＭオンサイクル中をＬとしＰＷＭオフサイクル中をＨとするＬＨ通電の二通りがある。図３のコイル電流が＋側に流れているＨ通電区間をみると、コイル電圧波形はまず断続通電する３０°期間があり続いて連続通電する６０°期間となり再び３０°断続通電期間がある。断続通電期間が上述の相補モードＨＬ通電に相当し、ＰＷＭオンサイクル中はＨとなりＰＷＭオフサイクル中はＬに落ちている。また、連続通電期間は出力をＨに固定している期間であり両者を合わせて１２０°位相角がＨ通電されている。
同様にコイル電流が−側に流れているＬ通電区間をみると、コイル電圧波形は断続通電する３０°期間が２か所と連続通電する６０°期間が１か所あり、断続通電期間が上述の相補モードＬＨ通電に相当し、ＰＷＭオンサイクル中はＬとなりＰＷＭオフサイクル中はＨに上昇している。連続通電期間は出力をＬに固定している期間であり両者を合わせて１２０°位相角がＬ通電されている。このようにＨＬ通電とＬＨ通電を使い分けることでＨ通電区間とＬ通電区間が実現される。
In the complementary mode PWM control, there are two types of HL energization in which the PWM on-cycle is H and the PWM off-cycle is L, and the LH energization is L during the PWM on-cycle and H during the PWM off-cycle. Looking at the H energization section in which the coil current flows to the + side in FIG. 3, the coil voltage waveform has a 30 ° period during which intermittent energization is first performed, and then becomes a 60 ° period during which continuous energization occurs, and there is another 30 ° intermittent energization period. The intermittent energization period corresponds to the complementary mode HL energization described above, and is H during the PWM on-cycle and falls to L during the PWM off-cycle. Further, the continuous energization period is a period in which the output is fixed at H, and a total of 120 ° phase angle is H energized.
Similarly, looking at the L energization section in which the coil current flows to the-side, the coil voltage waveform has two 30 ° periods for intermittent energization and one 60 ° period for continuous energization, and the intermittent energization period is as described above. This corresponds to the complementary mode LH energization, and is L during the PWM on cycle and rises to H during the PWM off cycle. The continuous energization period is a period in which the output is fixed at L, and a total of 120 ° phase angle L is energized. As described above, by selectively using the HL energization and the LH energization, the H energization section and the L energization section are realized.
引き続きコイル電流波形がゼロで直線状となっている２か所の６０°非通電区間についてみると、コイル電圧波形は傾斜した開放相誘起電圧となっており、中央部の波形がＰＷＭオンサイクルの誘起電圧波形であり、上下に分かれている波形がＰＷＭオフサイクルの誘起電圧波形である。 Next, looking at two 60 ° non-energized sections where the coil current waveform is zero and linear, the coil voltage waveform is an inclined open phase induced voltage, and the waveform in the center is the PWM on-cycle. It is an induced voltage waveform, and the waveform divided into the upper and lower parts is the induced voltage waveform of the PWM off cycle.
ＰＷＭオンサイクルの誘起電圧より高電位側の波形は、通電二相をともにＨとすることで中性点電位をＨとした時の開放相誘起電圧波形である。通電二相の内の一相についてはＨに固定することでＰＷＭオフサイクルもＨとすることができる。他相については相補モードＰＷＭ制御とし、ＰＷＭオンサイクルはＬ、ＰＷＭオフサイクルはＨとするＬＨ通電によりＰＷＭオフサイクルをＨとすることができる。これにより通電二相ともＨレベルとなる。
同様に低電位側の波形は、通電二相をともにＬとすることで中性点電位をＬとした時の開放相誘起電圧波形である。通電二相の内の１相はＬに固定することでＰＷＭオフサイクルもＬとすることができる。他相は相補モードＰＷＭ制御とし、ＰＷＭオンサイクルはＨ、ＰＷＭオフサイクルはＬとするＨＬ通電によりＰＷＭオフサイクルをＬとすることができる。これにより通電二相ともＬレベルとなる。
The waveform on the higher potential side than the induced voltage of the PWM on-cycle is an open phase induced voltage waveform when the neutral point potential is set to H by setting both energized two phases to H. By fixing the phase of one of the two energized phases to H, the PWM off cycle can also be set to H. The complementary mode PWM control is applied to the other phase, and the PWM on-cycle is set to L, and the PWM off-cycle is set to H. The PWM off-cycle can be set to H by LH energization. As a result, both energized two phases become H level.
Similarly, the waveform on the low potential side is an open phase induced voltage waveform when the neutral point potential is set to L by setting both energized two phases to L. By fixing one of the two energized phases to L, the PWM off cycle can also be set to L. The other phase is complementary mode PWM control, the PWM on-cycle is H, and the PWM off-cycle is L. The PWM off-cycle can be set to L by HL energization. As a result, both energized two phases become L level.
このように相補モードＰＷＭ制御しＨＬ通電とＬＨ通電を使い分けることでＰＷＭオフサイクルの誘起電圧波形もコントロールできる。通電区間及び非通電区間を通じてスパイク電流は出力素子ＦＥＴを経由して電源レールにクランプされる為、クランプダイオード損失が解消される。なおかつ開放相誘起電圧が電源レールを超えることがないので開放相ブレーキ電流も流れず高効率化される。 In this way, complementary mode PWM control is performed to selectively use HL energization and LH energization to control the induced voltage waveform of the PWM off cycle. Since the spike current is clamped to the power supply rail via the output element FET through the energized section and the non-energized section, the clamp diode loss is eliminated. Further, since the open phase induced voltage does not exceed the power supply rail, no open phase brake current flows and the efficiency is improved.
また本案は進角制御あるいはオーバーラップ通電など各種の通電手法にも容易に適用できる。図４に誘起電圧クランプレス通電を１５°進角制御した実測波形例を示す。図４は上段から下段に向かって、ホールセンサ波形ＨＵ，Ｕ相のコイル電圧波形Ｖｕ、Ｕ相のコイル電流波形Ｉｕ、三相全体のコイル電流波形Ｉｓ（レンジはＩｕとは異なる）である。なおハードディスク用のＳＰＭモータを位置センサレス駆動しており、ホールセンサ波形ＨＵはエンコーダ信号から生成したもので位相角を示すためだけに用いている。 Further, the present invention can be easily applied to various energization methods such as advance angle control or overlap energization. FIG. 4 shows an example of an actually measured waveform in which the induced voltage clampless energization is advanced by 15 °. FIG. 4 shows the Hall sensor waveform HU, the U-phase coil voltage waveform Vu, the U-phase coil current waveform Iu, and the three-phase overall coil current waveform Is (range is different from Iu) from the top to the bottom. The SPM motor for the hard disk is driven without the position sensor, and the Hall sensor waveform HU is generated from the encoder signal and is used only for indicating the phase angle.
図５に効率比較のため従来の１２０°通電の実測波形例を示す。図４は誘起電圧クランプレス通電及び相補モードＰＷＭ等を行っており、図５と比較することでその効果を評価できる。コイル電流波形Ｉｓを比較すると明らかに電流値が少なく高効率化されていることが判り、本案適用時は消費電力が約１６％減少し効率は約１０％向上した。これはサイン波駆動をほぼ５％上回る効率である。また電流リップルが見受けられるがこれは定トルク制御時の波形に類似しており、特段の制御をせずとも定トルク性が発揮されていることを意味しており好ましい特性と言える。 FIG. 5 shows an example of a measured waveform of conventional 120 ° energization for efficiency comparison. In FIG. 4, induced voltage clampless energization and complementary mode PWM are performed, and the effect can be evaluated by comparing with FIG. When the coil current waveform Is is compared, it is clear that the current value is small and the efficiency is improved, and when the present invention is applied, the power consumption is reduced by about 16% and the efficiency is improved by about 10%. This is almost 5% more efficient than sine wave drive. A current ripple can be seen, which is similar to the waveform during constant torque control, which means that constant torque is exhibited without any special control, which is a preferable characteristic.
（ハードウェア追加によるＬＨ通電の実施例）
前述した相補モードにてＬＨ通電を行えばクランプダイオード損失を完全に解消できるがＬＨ通電モードを備えていないＭＰＵ５１（マイクロコントローラ）も多い。以下は、ＭＰＵ５１に外付けでハードウェアを追加し、ＦＥＴゲート信号を反転することでＬＨ通電を実現し、クランプダイオード損失を解消しなおかつ開放相ブレーキ電流を阻止できる場合を説明する実施例である。
(Example of LH energization by adding hardware)
Clamp diode loss can be completely eliminated by conducting the LH conduction in the complementary mode described above, but there are many MPU 51 (microcontrollers) that do not have the LH conduction mode. The following is an example for explaining a case where external hardware is added to the MPU 51, LH conduction is realized by inverting the FET gate signal, the clamp diode loss can be eliminated and the open phase brake current can be blocked. ..
図６にＬＨ通電を可能とするＭＰＵ外付け回路例を示す。一例として反転回路５６としてエクスクルーシブオワゲート（ＸＯＲ回路：排他的論理和回路）を６個用いてＦＥＴゲート信号を相ごとに反転させることでＬＨ通電を行う。
ＭＰＵ５１はマイクロコントローラで６本のＦＥＴゲート信号をＦＥＴプリドライバに出力する。ＵＨ〜ＷＨはハイサイドアームのＦＥＴゲート信号、ＵＬ〜ＷＬはローサイドアームのＦＥＴゲート信号である。反転回路ＵＨ〜ＷＬは６個のＸＯＲゲートである。反転指令Ｕ〜Ｗは反転回路ＵＨ・ＵＬ、ＶＨ・ＶＬ，ＷＨ・ＷＬを制御するＭＰＵ５１のデジタル出力である。ＵＨ′〜ＷＬ′は反転回路を経由しＦＥＴプリドライバに出力される６個のＦＥＴゲート信号である。なおＦＥＴプリドライバ（図示せず）は６個のＦＥＴゲート信号ＵＨ′〜ＷＬ′を電力増幅してハーフブリッジ型インバータ回路５２のＦＥＴを駆動する回路である。またＦＥＴゲート信号の反転動作はＰＷＭ周期に同期して行う必要があり、ＰＷＭキャリア割り込みが発生したら直ちに反転指令を出力することとする。
ここではＦＥＴゲート信号の論理を反転させる方法としてＸＯＲゲートを用いる方法を例示したが、その他マルチプレクサにてハイサイドアーム側とローサイドアーム側の信号を入れ替える方法など様々考えられ例示した回路に限定するものではない。
FIG. 6 shows an example of an MPU external circuit that enables LH energization. As an example, six exclusive OR gates (XOR circuits: exclusive OR circuits) are used as the inverting circuit 56 to invert the FET gate signal for each phase to perform LH energization.
The MPU 51 is a microcontroller and outputs six FET gate signals to the FET pre-driver. UH to WH are high-side arm FET gate signals, and UL to WL are low-side arm FET gate signals. The inverting circuits UH to WL are 6 XOR gates. The inversion commands U to W are digital outputs of the MPU 51 that controls the inversion circuits UH · UL, VH · VL, and WH · WL. UH 'to WL' are six FET gate signals output to the FET predriver via the inverting circuit. The FET pre-driver (not shown) is a circuit that power-amplifies the six FET gate signals UH 'to WL' to drive the FETs of the half-bridge type inverter circuit 52. Further, the inversion operation of the FET gate signal needs to be performed in synchronization with the PWM cycle, and the inversion command is output immediately when the PWM carrier interrupt occurs.
Here, the method of using the XOR gate is exemplified as the method of inverting the logic of the FET gate signal, but other various methods such as the method of exchanging the signals on the high side arm side and the low side arm side by the multiplexer are limited to the exemplified circuits. is not.
また反転指令Ｕ〜Ｗの出力タイミングと対象相は１２個の通電区間に応じて決定されており下記に一覧表を示す。表２は前出表１に最下段の反転指令欄を追加したもので、符号は表１を援用する。
（表２）反転指令を用いる通電パターン
注１：Ｕ相〜Ｗ相欄のＬＨは、相補モードＬＨ通電を表す。この区間ではＭＰＵはＨＬ通電を出力しているが反転指令により外部回路にて反転されＬＨ通電となる。
注２：反転指令欄のＵ〜Ｗは、反転指令を出力する対象相を表す。これにより該当相はＬＨ通電モードとなる。
The output timings of the reversal commands U to W and the target phase are determined according to the twelve energized sections, and the list is shown below. Table 2 is a table in which the reversal command column at the bottom is added to the above-mentioned Table 1, and reference numerals are used in Table 1.
(Table 2) Energization pattern using inversion command
Note 1: LH in the U-phase to W-phase columns indicates complementary mode LH energization. In this section, the MPU outputs HL energization, but it is inverted in an external circuit by the inversion command and becomes LH energization.
Note 2: U to W in the reversal command column indicate the target phase that outputs the reversal command. As a result, the corresponding phase becomes the LH energization mode.
実施波形例は図３に示されているので、図３を参照しながら表２のＵ相を例として説明する。表の区間７及び区間１０がＬＨ通電期間であり、図３ではＬ通電期間の開始時と終了時の３０°断続通電期間に該当する。この時ＭＰＵはＨＬ通電を出力し反転指令により反転回路が動作しＬＨ通電となる。これ以外の区間はＨＬ通電なので反転せずそのまま出力する。
このようにＬＨ通電モードを備えないＭＰＵ５１では、ＨＬ通電パターンを出力しておいて該当相の反転指令をＩＯポートから出力してＬＨ通電とすればよい。反転指令の対象相と出力タイミングは表２の反転指令欄に記載されている通りである。Ｖ相及びＷ相についても１２０°位相差で同様に通電される。
Since an example of the implemented waveform is shown in FIG. 3, the U phase in Table 2 will be described as an example with reference to FIG. Sections 7 and 10 in the table are LH energization periods, which correspond to the 30 ° intermittent energization period at the start and end of the L energization period in FIG. At this time, the MPU outputs HL energization, and the inversion circuit operates according to the inversion command to become LH energization. The other sections are HL-energized, so they are output as they are without being inverted.
In this way, in the MPU 51 that does not have the LH energization mode, the HL energization pattern may be output and the inversion command for the corresponding phase may be output from the IO port for LH energization. The target phase of the reversal command and the output timing are as described in the reversal command column of Table 2. The V-phase and the W-phase are similarly energized with a phase difference of 120 °.
（ソフトウェアによるＬＨ通電の実施例）
ＬＨ通電モードを備えていないＭＰＵ５１（マイクロコントローラ）を使用して、ＰＷＭ制御モード（独立モードと相補モード）をソフトウェアで切り替えることでＬＨ通電のかわりにＬＺ通電を行い、クランプダイオード損失を半減し尚かつ開放相ブレーキ電流を阻止できる実施例について説明する。
(Example of LH energization by software)
Using the MPU51 (micro controller) that does not have the LH energization mode, by switching the PWM control mode (independent mode and complementary mode) by software, LZ energization is performed instead of LH energization, and the clamp diode loss is halved. An embodiment capable of blocking the open phase brake current will be described.
ＬＨ通電のかわりにＰＷＭオフサイクル時にＺ（ハイインピーダンス）とするＬＺ通電としてもよい。なぜならば、ＰＷＭオンサイクル時にＬに通電した相をオフサイクル時にＺ（開放）とするとスパイク電圧が発生しコイル電圧はＨとなりＬＨ通電と同じ効果が得られるからである。ただしクランプダイオード損失が発生する。
一方、ＬＺ通電は独立モードにて実現できる。従ってＬＨ通電が必要となる期間は相補モードから独立モードへと替えてＬＺ通電とすれば、相補モードＬＨ通電と同様の動作となりＬＨ通電モードを備えていないＭＰＵが使用可能となる。モード切り換え機能は一般的なＰＷＭ制御回路でも標準的ファンクションとして備えており、ＰＷＭキャリア割り込みに同期してモード切り換えを実行すればタイミングも問題ない。この方法はクランプダイオード損失が全期間の半分で発生するがソフトウェアのみで簡易的に開放相ブレーキ電流を阻止できるメリットがある。
Instead of LH energization, LZ energization in which Z (high impedance) is set during the PWM off cycle may be used. This is because if a phase that has been energized to L during the PWM on-cycle is set to Z (open) during the off-cycle, a spike voltage is generated, the coil voltage becomes H, and the same effect as LH energization can be obtained. However, clamp diode loss occurs.
On the other hand, LZ energization can be realized in the independent mode. Therefore, if the LZ energization is performed by changing the complementary mode to the independent mode during the period in which the LH energization is required, the same operation as the complementary mode LH energization is performed, and the MPU not equipped with the LH energization mode can be used. The mode switching function is also provided as a standard function in a general PWM control circuit, and there is no problem in timing if the mode switching is executed in synchronization with the PWM carrier interrupt. With this method, the clamp diode loss occurs in half of the entire period, but it has the advantage that the open phase brake current can be blocked simply by software.
図７に区間終点検出後の制御プログラムフローチャート例を示す。以下ステップごとに説明する。
区間終点検出すると制御フローチャートが開始する（ＳＴＡＲＴ）。
ＰＷＭ制御回路は、通電区間番号を歩進する。励磁切り替えは次のＰＷＭ周期にて実行する（ＳＴＥＰ１）。通電区間前半３０°の通電パターンを出力レジスタにセットする（ＳＴＥＰ２）。上位コントローラからのトルク指令に基づきＰＷＭデューティ比をレジスタにセットする（ＳＴＥＰ３）。
FIG. 7 shows an example of a control program flow chart after detection of the section end point. Each step will be described below.
When the section end point is detected, the control flow chart starts (START).
The PWM control circuit increments the energization section number. Excitation switching is executed in the next PWM cycle (STEP 1). The energization pattern of 30 ° in the first half of the energization section is set in the output register (STEP 2). The PWM duty ratio is set in the register based on the torque command from the host controller (STEP 3).
次に、ＰＷＭキャリアの割り込み待ちを行う。割り込みが無ければＬＯＯＰ２へ戻る（ＳＴＥＰ４）。ＰＷＭキャリア割り込みがあれば、ＰＷＭモード設定（独立モードまたは相補モード）を行う（ＳＴＥＰ５）。ＰＷＭオンサイクル時にＡＤＣにより三相のコイル電圧測定する（ＳＴＥＰ６）。 Next, the PWM carrier interrupt is waited for. If there is no interrupt, it returns to LOOP2 (STEP4). If there is a PWM carrier interrupt, the PWM mode setting (independent mode or complementary mode) is performed (STEP 5). During the PWM on-cycle, the three-phase coil voltage is measured by the ADC (STEP6).
次に、三相コイルのうち開放相にスパイク電圧が発生したか否かを判定する（スパイク判定）。スパイク電圧を検出したらＬＯＯＰ２へ戻る（ＳＴＥＰ７）。スパイク電圧でなければ、誘起電圧のゼロクロス判定を行う。ゼロクロス点でなければＬＯＯＰ２へ戻る（ＳＴＥＰ８）。セロクロス点を検出すると、区間後半３０°の通電パターン（ＬＺ通電）を出力レジスタにセットする（ＳＴＥＰ９）。 Next, it is determined whether or not a spike voltage is generated in the open phase of the three-phase coil (spike determination). When the spike voltage is detected, the process returns to LOOP2 (STEP7). If it is not the spike voltage, the zero cross determination of the induced voltage is performed. If it is not the zero-cross point, the process returns to LOOP2 (STEP8). When the cello cross point is detected, the energization pattern (LZ energization) in the latter half 30 ° of the section is set in the output register (STEP 9).
次に、ＰＷＭキャリアの割り込み待ちを行う。割り込みが無ければＬＯＯＰ３へ戻る（ＳＴＥＰ１０）。ＰＷＭキャリア割り込みがあれば、ＰＷＭモード設定（独立モードまたは相補モード）を行う（ＳＴＥＰ１１）。ＰＷＭオンサイクル時にＡＤＣにより三相のコイル電圧測定する（ＳＴＥＰ１２）。ＰＷＭ制御回路は、通電区間終点か否かを判定し、通電区間終点でなければＬＯＯＰ３へ戻る。また、通電区間終点ならＬＯＯＰ１へ戻る（ＳＴＥＰ１３）。
以上の手順で期間の半分を相補モードとしてクランプダイオード損失を半減し、残りの半分を独立モードとしてＬＺ通電を行い誘起電圧クランプレス通電を実現できる。
Next, the PWM carrier interrupt is waited for. If there is no interrupt, the process returns to LOOP3 (STEP 10). If there is a PWM carrier interrupt, the PWM mode setting (independent mode or complementary mode) is performed (STEP 11). During the PWM on-cycle, the three-phase coil voltage is measured by the ADC (STEP 12). The PWM control circuit determines whether it is the end point of the energization section, and if it is not the end point of the energization section, returns to LOOP3. If it is the end point of the energization section, the process returns to LOOP1 (STEP 13).
By the above procedure, half of the period is set as the complementary mode to reduce the clamp diode loss by half, and the other half is set as the independent mode to perform the LZ energization, thereby realizing the induced voltage clampless energization.
なお、モータ駆動回路の構成や制御プログラム構成は様々考えられ、本実施例に開示された態様に限定されるものではなく、本案主旨を逸脱しない範囲で電子回路技術者あるいはプログラマー（当業者）であれば当然なし得る回路構成の変更やプログラム構成の変更も含まれる。 It should be noted that various configurations of the motor drive circuit and control program configurations are conceivable, and the present invention is not limited to the mode disclosed in the present embodiment, and an electronic circuit engineer or a programmer (a person skilled in the art) can use the present invention without departing from the gist of the present invention. It also includes a change in the circuit configuration and a change in the program configuration, which can be done naturally.
１ 回転子軸 ２ 回転子 ３ 永久磁石 ４ 固定子 ５０ 上位コントローラ ５１ ＭＰＵ ５２ ハーフブリッジ型インバータ回路（ＩＮＶ） ５３ 反転ＰＷＭ制御回路 ５４ ＡＤ変換回路（ＡＤコンバータ：ＡＤＣ） ５５ ゼロクロスコンパレータ ５６ 反転回路 1 Rotor Axis 2 Rotor 3 Permanent Magnet 4 Stator 50 Upper Controller 51 MPU 52 Half Bridge Inverter Circuit (INV) 53 Inversion PWM Control Circuit 54 AD Conversion Circuit (AD Converter: ADC) 55 Zero Cross Comparator 56 Inversion Circuit
Claims (5)
前記永久磁石界磁位置を検出あるいは推定する位置検出回路と、
三相コイル電圧をＡＤ変換して制御回路に送出する測定回路と、
ハーフブリッジ型インバータ回路を介して前記三相コイルに双方向通電する出力回路と、
上位コントローラからのトルク指令に基づいてＰＷＭ方式にてコイル出力を制御し、連続回転が可能な通電角度情報と通電パターン情報とを記憶し、前記位置検出回路の出力に基づいて前記出力回路を制御して通電状態を切り替える制御回路と、を備え、
ＰＷＭ周期内の通電期間において正極電源に接続する相を電源相、接地電源に接続する相を接地相、ハイインピーダンス（開放）状態とする相を開放相とし、三相の共通接続点電位を中性点電位として、前記制御回路は前記測定回路の出力に応じて当該ＰＷＭ周期内の遮断期間の電源相及び接地相の出力状態を切り替え、開放相の誘起電圧が中性点電位に対して正の時は遮断期間において接地相を接地電源に接続し電源相も接地電源に接続するかまたはハイインピーダンス状態とし、開放相の誘起電圧が中性点電位に対して負の時は遮断期間において電源相を正極電源に接続し接地相も正極電源に接続するかまたはハイインピーダンス状態とすることを特徴とする電動機の駆動方法。 A method of driving a motor having a rotor having a permanent magnet field and a stator having a three-phase coil, in which two-phase current is applied by a pulse width modulation (PWM) method,
A position detection circuit for detecting or estimating the permanent magnet field position;
A measuring circuit for AD-converting the three-phase coil voltage and sending it to the control circuit;
An output circuit for bidirectionally energizing the three-phase coil via a half-bridge type inverter circuit,
The coil output is controlled by the PWM method based on the torque command from the host controller, the energization angle information and the energization pattern information capable of continuous rotation are stored, and the output circuit is controlled based on the output of the position detection circuit. And a control circuit for switching the energized state,
During the energization period in the PWM cycle, the phase connected to the positive power supply is the power supply phase, the phase connected to the ground power supply is the ground phase, and the phase in the high impedance (open) state is the open phase, and the common connection point potential of the three phases is medium. As the neutral point potential, the control circuit switches the output states of the power supply phase and the ground phase in the cutoff period within the PWM cycle according to the output of the measurement circuit, and the induced voltage of the open phase is positive with respect to the neutral point potential. In the cutoff period, the ground phase is connected to the ground power supply and the power supply phase is also connected to the ground power supply or in a high impedance state.When the induced voltage in the open phase is negative with respect to the neutral point potential, the power supply is cut in the cutoff period. A method for driving an electric motor, characterized in that the phase is connected to a positive electrode power source and the ground phase is also connected to a positive electrode power source or is in a high impedance state.
ＰＷＭ周期の遮断期間において接地相を接地電源に接続し電源相も接地電源に接続またはハイインピーダンス状態とするか、あるいは電源相を正極電源に接続し接地相も正極電源に接続またはハイインピーダンス状態とするかを、前記１２区間に応じて選択する請求項１記載の電動機の駆動方法。 A zero-cross detection circuit for detecting an open-phase induced voltage zero-cross point is provided, and the energization section at 120 ° energization is divided into a front section and a rear section at the zero-cross point, and the electrical angle is 12 sections.
In the cutoff period of the PWM cycle, the ground phase is connected to the ground power supply and the power supply phase is also connected to the ground power supply or in the high impedance state, or the power supply phase is connected to the positive power supply and the ground phase is also connected to the positive power supply or in the high impedance state. The driving method for the electric motor according to claim 1, wherein whether to perform the selection is selected according to the 12 sections.
ＰＷＭ制御回路は、相ごとに前記ハーフブリッジ型インバータ回路のハイサイドアームとローサイドアームを対で制御し、ＰＷＭオフサイクル中はＰＷＭ周期内の通電期間（ＰＷＭオンサイクル）とは逆サイドのアームをオンとする相補モードでＰＷＭ制御を行い、
ＰＷＭオンサイクル中はＨ（ハイサイドアームオン）としＰＷＭオフサイクル中はＬ（ローサイドアームオン）とするＨＬ通電と、ＰＷＭオンサイクル中はＬ（ローサイドアームオン）としＰＷＭオフサイクル中はＨ（ハイサイドアームオン）とするＬＨ通電の双方の通電モードを備え、
開放相の誘起電圧と中性点電位の大小関係に応じて通電モードを切り換え、開放相の誘起電圧が中性点電位に対して負の期間はＬＨ通電し、正の期間はＨＬ通電することで開放相ブレーキ電流を阻止する請求項１記載の電動機の駆動方法。 The output circuit includes a field effect transistor (FET) as a switching element,
The PWM control circuit controls the high side arm and the low side arm of the half bridge type inverter circuit in pairs for each phase, and controls the arm on the side opposite to the energization period (PWM on cycle) in the PWM cycle during the PWM off cycle. PWM control is performed in the complementary mode to turn on,
H (high side arm on) during the PWM on cycle and L (low side arm on) during the PWM off cycle, and L (low side arm on) during the PWM on cycle and H (high side during the PWM off cycle). It has both energization modes of LH energization (side arm on),
The energization mode should be switched according to the magnitude relationship between the open-phase induced voltage and the neutral point potential, and LH should be energized when the open-phase induced voltage is negative with respect to the neutral point potential, and HL should be energized during the positive period. The method of driving an electric motor according to claim 1, wherein the open-phase braking current is blocked by.
前記マイクロコントローラはＬＨ通電が必要な相に対して、いずれかの前記反転回路へＰＷＭキャリアに同期して反転指令を出力し、当該通電相のＨＬ通電モード状態の２個のＦＥＴゲート信号を論理反転あるいは入れ替えることでＬＨ通電モード状態の信号に変換してＬＨ通電を行う請求項３記載の電動機の駆動方法。 An inversion circuit that inverts the logic for each of the six FET gate signals sent from the PWM control circuit built in the microcontroller to the FET pre-driver is provided, or an inversion circuit that switches the high side arm and the low side arm for each phase. Provided,
The microcontroller outputs an inversion command to one of the inversion circuits in synchronization with the PWM carrier for a phase requiring LH energization, and logically outputs two FET gate signals in the HL energization mode state of the energized phase. 4. The method of driving an electric motor according to claim 3, wherein the LH energization is performed by converting the signal into a signal in the LH energization mode state by reversing or replacing.
開放相の誘起電圧が中性点電位に対して正の時はＰＷＭキャリアに同期して、ＰＷＭ制御モードをＰＷＭオンサイクル中はＨ（ハイサイドアームオン）、ＰＷＭオフサイクル中はＬ（ローサイドアームオン）とする相補モードに切り替えてＨＬ通電にてＰＷＭ制御する請求項３記載の電動機の駆動方法。 The PWM control circuit synchronizes with the PWM carrier when the induced voltage of the open phase is negative with respect to the neutral point potential, and sets the PWM control mode to L (low side arm on) during the PWM on cycle and during the PWM off cycle. Is switched to the independent mode of Z (high impedance state) and PWM controlled by LZ energization,
When the induced voltage in the open phase is positive with respect to the neutral point potential, the PWM control mode is synchronized with the PWM carrier, and the PWM control mode is H (high side arm on) during the PWM on cycle and L (low side arm) during the PWM off cycle. The method of driving an electric motor according to claim 3, wherein PWM control is performed by switching to a complementary mode of turning on and energizing HL.
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