EP2503846B1 - Lighting device and illumination apparatus using the same - Google Patents
Lighting device and illumination apparatus using the same Download PDFInfo
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- EP2503846B1 EP2503846B1 EP12001522.7A EP12001522A EP2503846B1 EP 2503846 B1 EP2503846 B1 EP 2503846B1 EP 12001522 A EP12001522 A EP 12001522A EP 2503846 B1 EP2503846 B1 EP 2503846B1
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- Prior art keywords
- switching element
- voltage
- unit
- lighting device
- pwm signal
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/30—Driver circuits
- H05B45/37—Converter circuits
- H05B45/3725—Switched mode power supply [SMPS]
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/30—Driver circuits
- H05B45/37—Converter circuits
- H05B45/3725—Switched mode power supply [SMPS]
- H05B45/375—Switched mode power supply [SMPS] using buck topology
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/30—Driver circuits
- H05B45/37—Converter circuits
- H05B45/3725—Switched mode power supply [SMPS]
- H05B45/38—Switched mode power supply [SMPS] using boost topology
Definitions
- JP2006-511078A a power feeding assembly (lighting device) for feeding a power to a-light emitting diode (LED) illumination module
- JP2006-511078A Japanese Patent Application Publication No. 2006-511078
- the prior art example described in JP2006-511078A includes a series circuit of a diode D10 and a control switch 101 configured with a MOSFET which are connected to both ends of a DC power supply 100.
- an inductor L10 and an LED illumination module 102 are connected to both ends of the diode D10.
- a controller 103 generates a dual-PWM (Pulse-Width Modulation) switching signal supplied to a control input unit of a control switch 101 through an amplifier 104.
- the dual-PWM switching signal is a combination of a high-frequency PWM switching signal component and pulse bursts of a low-frequency, i.e., a low-frequency PWM switching signal component.
- the controller 103 includes a current mode pulse width modulator 105, which receives an LED current reference signal, a detection current, and a high-frequency sawtooth wave signal from a current supply 106.
- the current mode pulse width modulator 105 generates a high-frequency PWM switching signal component supplied as one input of an AND gate 107, and the other input of the AND gate 107 is a low-frequency PWM switching signal component.
- An output from the AND gate 107 is supplied to a gate of the control switch 101 through the amplifier 104.
- an average current flowing through the LED illumination module 102 can be changed by changing the low-frequency component of the dual-PWM switching signal, and thus, the intensity of light output from the LED illumination module 102 is changed.
- the frequency of the PWM signal is required to be increased to have a certain value or higher to prevent blinking due to an interference with a frequency of the imaging device from being observed.
- the frequency of the PWM signal is increased, the ratio of one period of the driving signal of control switch 101 to one period of the PWM signal is increased. Then, the light output is increased by one period of the driving signal of the control switch 101 and it is more conspicuously seen such that the light output from the LED illumination module 102 is changed by one step at a time.
- the frequency of the driving signal of the control switch 101 is required to be increased, but considering an increase in a switching loss or an upper limit of the frequency of the driving signal in case of driving with a low-priced part such as a general IC, and the like, a desirable high-frequency is hardly guaranteed.
- US 2010/0127672 A1 discloses a power supply device including a DC-DC boost converter and a charge recycling circuit.
- the DC-DC boost converter is utilized for boosting an input voltage to generate an output voltage, and adjusting a voltage level switching signal.
- WO 2007/049198 A1 refers to a system for driving a constant current load comprising the constant current load and a down-converter.
- the down-converter comprises a series arrangement of a switch and an inductor. This series arrangement is arranged in series with the constant current load.
- the present invention provides a lighting device capable of smoothly changing a light output from a light source unit in sweeping a PWM signal without making a driving signal of a switching element have a high-frequency, and an illumination apparatus using the same.
- a lighting device including: a lighting unit for supplying a lighting power to a light source unit including one or more solid-state light emitting elements by using a DC voltage from a power supply unit as an input; and a controller for controlling the lighting unit.
- the lighting unit has a series circuit of an inductor and a switching element, and a diode for recovering stored energy of the inductor for the light source unit during an OFF period of the switching element
- the controller has a unit for intermittently driving an ON/OFF operation of the switching element by a PWM signal and a unit for driving the switching element by a frequency higher than that of the PWM signal during an ON period of the PWM signal.
- the controller reduces a peak value of a load current flowing through the light source unit during a certain period.
- the lighting unit may further has a detection circuit for detecting the load current flowing through the light source unit, and the controller may further has: a threshold value adjusting unit for setting and outputting the peak value of the load current; a comparator for comparing an output from the detection circuit with an output from the threshold value adjusting unit: and a driving controller for controlling an ON period of the switching element based on an output from the comparator.
- the comparator compares a superimposed voltage obtained by superimposing the output from the detection circuit and that from the threshold value adjusting unit, with a certain reference voltage.
- the certain period during which the peak value of the load current is reduced is preferably longer than the OFF period of the switching element during the ON period of the PWM signal.
- the controller When the PWM signal rises, the controller preferably controls the ON period the switching element to increase the peak value of the load current during a certain period.
- the controller may control the switching element in a current critical mode.
- the controller may control the switching element in a current discontinuous mode.
- the controller may control the switching element in a current continuous mode.
- the DC voltage from the power supply unit may be obtained from an AC/DC converter and a frequency of the PWM signal is 600 Hz or a multiple of 600 Hz.
- the lighting unit 1 includes a series circuit of a switching element Q1, an inductor L1, and a resistor R1 connected to both ends of the DC power supply DC1.
- the lighting unit 1 includes a diode D1 for allowing a flyback current of the inductor L1 to flow during an OFF period of the switching element Q1, and is configured as a buck chopper circuit as a whole.
- the switching element Q1 is configured with, e.g., an n-channel type MOSFET and performs ON/OFF switching depending on a driving signal applied from a driving circuit 20C (to be described later).
- the resistor R1 detects a current flowing through the inductor L1 through the switching element Q1, whereby a load current flowing through the light source unit 3 can be detected.
- the resistor R1 serves as a detection circuit which detects a voltage between the two ends thereof to thereby detect a load current flowing through the light source unit 3 through the switching element Q1.
- the driving controller 20 includes an RS type flipflop FF1, and an output signal from the OR circuit OR1 is inputted to an S terminal of the flipflop FF1. Also, the driving controller 20 includes a driving circuit 20C for providing a driving signal to the switching element Q1, and an output signal from a Q terminal of the flipflop FF1 is inputted to the driving circuit 20C.
- the threshold value adjusting unit 21 includes a parallel circuit of a constant current supply CS1 and a capacitor C1, and a constant voltage supply VS1 connected to one end of a high pressure side of the capacitor C1 through a switching element Q2. An ON/OFF operation of the switching element Q2 is switched by a low-frequency PWM signal. Also, the one end of the high pressure side of the capacitor C1 is connected to the inverting input terminal of the comparator COM1.
- the constant voltage VRef1 of the constant voltage supply VS1 is applied as the reference voltage Vth1 to the inverting input terminal of the comparator COM1 and the capacitor C1 is charged. Further, when the switching element Q2 is turned off, the charge voltage of the capacitor C1 is applied as the reference voltage Vth1 to the inverting input terminal of the comparator COM1 and the capacitor C1 is discharged by the constant current supply CS1. That is, in the threshold value adjusting unit 21, the constant voltage supply VS1, the switching element Q2, and the constant current supply CS1 constitute a charging/discharging circuit of the capacitor C1. Namely, an output voltage from the threshold value adjusting unit 21 is a charge/discharge voltage of the capacitor C1.
- a startup signal is inputted to the OR circuit OR1 from the starting circuit 20B, and a high level set signal is inputted to the S terminal of the flipflop FF1 from the OR circuit OR1. Accordingly, an output signal from the Q terminal of the flipflop FF1 becomes a high level and a driving signal is applied to the switching element Q1 from the driving circuit 20C, whereby the switching element Q1 is changed to be turned on. Then, a current flows through the light source unit 3, the inductor L1, the switching element Q1, and the resistor R1, thus increasing the load current (see Fig. 2A ).
- the detection voltage VR1 Since the load current is increased, the voltage between two ends of the resistor R1, i.e., the detection voltage VR1, is increased. And, when the detection voltage VR1 reaches the reference voltage Vth1, the output signal from the comparator COM1 is inverted and a high level reset signal is inputted to the R terminal of the flipflop FF1. Accordingly, the output signal from the Q terminal of the flipflop FF1 becomes a low level and the supply of driving signal to the switching element Q1 from the driving circuit 20C is stopped, whereby the switching element Q1 is changed to be turned off.
- the zero current detection circuit 20A detects inversion of the voltage applied to the inductor L1, namely, a zero cross of the current flowing through the inductor L1, it inputs a high signal to the OR circuit OR1. Accordingly, a high level set signal is inputted to the S terminal of the flipflop FF1 from the OR circuit OR1. Thus, an output signal from the Q terminal of the flipflop FF1 becomes a high level, and a driving signal is applied to the switching element Q1 from the driving circuit 20C, whereby the switching element Q1 is changed to be turned on.
- the driving controller 20 of the controller 2 controls the switching element Q1 in a current critical mode. Also, while the load current flows through the light source unit 3, the respective LEDs 30 of the light source unit 3 are turned on.
- the peak value Ith1 of the load current is also linearly reduced as indicated by the dashed single-dotted line in the same drawing. Namely, when the solid line and the dashed single-dotted line in the same drawing are compared, it can be seen that the peak value Ith1 of the load current in the threshold value down period TD1 is continuously changed depending on a continuous change in the ON duty ratio of the PWM signal.
- the load current i.e., the light output from the light source unit 3
- the change in the light output from the light source unit 3 when the PWM signal is swept can be smoothly made.
- the change ratio of the light output from the light source unit 3 is increased, the change is notably seen.
- the change in the light output from the light source unit 3 can be smoothly made.
- the light source unit 3 is viewed through a different imaging device such as a video camera or the like, even when the frequency of the PWM signal is increased to be a certain value or higher to prevent blinking due to the interference with a frequency of the imaging device from being observed, the change of the light output from the light source unit 3 can be smoothly made. Thus, it is not required to make the driving signal of the switching element Q1 have a high-frequency.
- the threshold value down period TD1 is set to be about 3 times the OFF time T1 to smoothly change the light output from the light source unit 3 in comparison to the case in which the threshold value down period TD1 is about 1.5 times the OFF time T1 (see Fig. 3 ). This is because, as shown in Fig. 2B , since the number of triangular wave pulses of the load current is increased during the threshold value down period TD1, the change in the load current when the ON duty ratio of the PWM signal is swept is close to be linear.
- the DC power supply DC1 is used as a power supply unit, but as shown in Fig. 4A , the power supply unit may be configured with the AC power supply AC1, an AC/DC converter unit 4 for converting an AC voltage from the AC power supply AC1 into a DC voltage and outputting the same, and a smoothing capacitor C0. Meanwhile, the power supply unit may be configured with the DC power supply DC1 and the DC/DC converter unit for converting a DC voltage from the DC power supply DC1 into a desired DC voltage and outputting the same, as shown Fig. 4c . In either case, the same effect can be obtained.
- the smoothing capacitor C2 may be provided to be connected in parallel to the light source unit 3. In this case, since the ripples of the load current flowing through the light source unit 3 can be reduced to be small, the light output from the light source unit 3 can be smoothly changed.
- the switching element Q1 is disposed at a lower pressure side of the DC power supply DC1, but the switching element Q1 may also be disposed at a high pressure side of the DC power supply DC1 to configure the lighting unit 1.
- 'Id' is a current flowing through the switching element Q1
- 'E' is a DC voltage from the DC power supply DC1
- 'V' is a load voltage of the light source unit 3
- 'L' is inductance of the inductor L1
- 't' is a lapse time.
- the current i.e., the load current
- the current flowing through the inductor L1 when the switching element Q1 is turned on is the same as the current flowing through the switching element Q1 expressed by Eq. 1.
- 'IL' is a current flowing through the inductor L1 when the switching element Q1 is turned off
- 'T2' is an ON time in one cycle of the switching element Q1 during the ON period of the PWM signal.
- T 1 L V Ith 1
- T 2 L E - V Ith 1
- the amount of change of the ON time T2 of the switching element Q2 is larger than that of the ON time of the PWM signal. Further, since the last triangular wave pulse of the load current generated during the threshold value down period TD1 is equivalent to minimum resolution of the load current, i.e., the light output from the light source unit 3, the light output from the light source unit 3 can be smoothly changed as the corresponding triangular wave pulse is smaller.
- the corresponding triangular wave pulse is smaller as the ON duty ratio of the switching element Q1 is larger.
- the light output from the light source unit 3 can be more smoothly changed by increasing the ON duty ratio of the switching element Q1.
- the DC voltage of the DC power supply DC1 is preferably equal to or less than five times the load voltage of the light source unit 3. Further, a lower limit of the DC voltage of the DC power supply DC1 is required to be at least larger than the load voltage of the light source unit 3, i.e., K is greater than one (K>1), to ensure the chopper operation by the lighting unit 1. More preferably, considering the change in the load voltage depending on temperature characteristics of the respective LEDs 30 of the light source unit 3, K needs to be equal to or greater than 1.2 (K ⁇ 1.2).
- the present embodiment features that a constant current supply CS2 is provided instead of the constant voltage supply VS1 in the threshold value adjusting unit 21, thus linearly increasing the peak value Ith1 of the load current when the PWM signal rises.
- the constant voltage VRef1 of the constant voltage supply VS1 is inputted as the reference voltage Vth1 to the inverting input terminal of the comparator COM1, but in the present embodiment, a charge voltage of the capacitor C1 is inputted instead.
- the switching element Q2 is changed to be turned on, and the capacitor C1 is charged by the difference between a constant current flowing from the constant current supply CS2 and a constant current flowing from the constant current supply CS1. Accordingly, since the charge voltage of the capacitor C1 is linearly increased, the reference voltage Vth1 is also linearly increased as indicated by the dotted line in Fig. 6B .
- a time duration until the reference voltage Vth1 reaches the constant voltage VRef1 is called a "threshold value up period TU1'.
- the ON/OFF operation of the switching element Q1 is controlled by using the gradually increased reference voltage Vth1 as a threshold value.
- a tilt of the reference voltage Vth1 during the threshold value up period TU1 is determined by the charge current of the capacitor C1, namely, by the difference between the constant current flowing from the constant current supply CS2 and the constant current flowing from the constant current supply CS1.
- the ON duty ratio of the PWM signal when the ON duty ratio of the PWM signal is small (close to 0 %), the reference voltage Vth1 does not reach the constant voltage Vref1 during the threshold value up period TU1 as indicated by the dotted line in Fig. 7A .
- the peak value Ith1 of the load current during the threshold value up period TU1 is continuously changed depending on a continuous change in the ON duty ratio of the PWM signal. For this reason, as the ON duty ratio of the PWM signal is close to 0 %, the peak value Ith1 of the load current is continuously reduced to zero.
- a change in the light output from the light source unit 3 when the ON duty ratio of the switching element Q1 is changed will be described with reference to Fig. 8 .
- a correlation between the ON duty ratio of the PWM signal and the light output in the case where the threshold value up period TU1 is considered (i.e., in case of employing the present embodiment) under the foregoing condition is indicated by a dotted line.
- the threshold value up period TU1 since the threshold value up period TU1 is provided, the light output from the light source unit 3 can be smoothly changed from an almost zero to a maximum output.
- the threshold value up period TU1 and the threshold value down period TD1 such that they are almost equal, the ON duty ratio of the PWM signal and the light output from the light source unit 3 have an almost proportional relationship, which is preferable.
- the present embodiment features that the constant voltage VRef1 is inputted to the inverting input terminal of the comparator COM1 of the driving controller 20 and an superimposed voltage V1 (to be described later) is increased during the OFF period of the PWM signal, thereby reducing the peak value Ith1 of the load current.
- the constant current supply CS1 and the capacitor C1 are connected in series and the capacitor C1 and the switching element Q2 are connected in parallel.
- the capacitor C1 is discharged during the ON period of the PWM signal and it is charged by the constant current from the constant current supply CS1 during the OFF period of the PWM signal.
- the resistor R3 is connected in series to the capacitor C1 and the resistor R2 is connected in series to the resistor R1 of the lighting unit 1. Further, a connection point of the resistors R2 and R3 is connected to the non-inverting input terminal of the comparator COM1.
- the charge voltage V1 which is the sum of the voltages obtained by respectively multiplying coefficients determined in the resistors R2 and R3 to the detection voltage VR1, as the voltage between two ends of the resistor R1, and the charge voltage of the capacitor C1, is inputted to the non-inverting input terminal of the comparator COM1.
- the operation of the present embodiment will be described with reference to Fig. 9B .
- the switching element Q2 is in an ON state, and thus, the capacitor C1 is not charged. Therefore, since the superimposed voltage V1 based only on the detection voltage VR1 is inputted to the non-inverting input terminal of the comparator COM1, the switching element Q1 is repeatedly turned on and off periodically, and the peak value Ith1 of the load current becomes uniform.
- the switching element Q2 is changed to be turned off, and thus, the capacitor C12 starts to be charged.
- the superimposed voltage V1 based on the detection voltage VR1 and the charge voltage of the capacitor C1 is inputted to the non-inverting input terminal of the comparator COM1.
- the charge voltage of the capacitor C1 is linearly increased with the lapse of time, and finally, is higher than the reference voltage VRef1.
- the threshold value down period TD1 can be provided.
- the threshold value down period TD1 can be provided, and therefore, the same effect as that of the first embodiment can be obtained.
- the controller 2 is configured by using a general PFC (Power Factor Correction) control IC such as MC33262 of ON Semiconductor or L6562 of ST Micro Electronics in order to eliminate harmonics.
- a general PFC (Power Factor Correction) control IC such as MC33262 of ON Semiconductor or L6562 of ST Micro Electronics in order to eliminate harmonics.
- the general PFC control IC has a reference voltage supply therein, in the configuration of the first embodiment, the reference voltage Vth1 cannot be variably controlled, and thus, the peak value Ith1 of the load current cannot be variably controlled. Meanwhile, in the configuration of this embodiment, the peak value Ith1 of the load current can be variably controlled even when the global PFC control IC is utilized, whereby the number of components constituting the controller 2 can be reduced.
- the present embodiment features that a series circuit of the constant voltage supply VS1 and a resistor R4 instead of the constant current supply CS1 is provided in the threshold value adjusting unit 21.
- the charge voltage of the capacitor C1 is linearly increased by the constant current of the constant current supply CS1.
- the resistor R4 and the capacitor C1 constitute an integrator circuit, the charge voltage of the capacitor C1 is exponentially increased as shown in Fig. 10B .
- the peak value Ith1 of the load current is also exponentially reduced.
- this embodiment features that a resistor R5 is connected in series to the switching element Q2 in the threshold value adjusting unit 21.
- the switching element Q2 when the PWM signal is shifted from the OFF period to the ON period, the switching element Q2 is changed to be turned ON and shorted, the superimposed voltage V1 becomes zero almost in a moment.
- the resistor R5 and the capacitor C1 constitute an integrator circuit, the capacitor C1 is discharged and the charge voltage is exponentially reduced, and thus, the superimposed voltage V1 is also exponentially reduced when the PWM signal is shifted from the OFF period to the ON period, as shown in Fig. 11B .
- the peak value Ith1 of the load current is linearly increased. Namely, during the ON period of the PWM signal, likewise as in the third embodiment, the threshold value up period TU1 can be provided.
- the present embodiment features that an oscillator 20D for outputting an oscillation signal having a certain cycle, instead of the secondary coil of the inductor L1, is connected to the zero current detection circuit 20A of the driving controller 20.
- the zero current detection circuit 20A inputs a high signal to the OR circuit OR1 periodically based on the cycle of the oscillation signal applied from the oscillator 20D. Namely, in the present embodiment, only the ON time of the switching element Q1 is variably controlled, and the switching element Q1 is driven periodically, without detecting a zero cross of the load current. Accordingly, in the present embodiment, as shown in Fig. 12B , the switching element Q1 is controlled in a so-called current discontinuous mode in which the load current intermittently flows.
- the switching element Q1 is controlled in the current discontinuous mode, but the same effect as that of the first embodiment can be obtained unlike the first embodiment.
- the oscillation signal of the oscillator 20D is inputted to the zero current detection circuit 20A, but the zero current detection circuit 20A is not necessarily required and, e.g., a universal PWM control IC may be configured instead. Namely, a configuration, in which a high signal is inputted to the OR circuit OR1 periodically, is desirable.
- the present embodiment features that a mono-stable multivibrator 20E, instead of the secondary coil of the inductor L1, is connected to the zero current detection circuit 20A of the driving controller 20.
- the mono-stable multivibrator 20E is connected to the driving circuit 20C, and after the driving signal from the driving circuit 20C is changed to be a low level, the signal is inputted to the zero current detection circuit 20A after the lapse of a certain period of time.
- the zero current detection circuit 20A inputs a high signal to the OR circuit OR1.
- the OFF time of the switching element Q1 is made constant and only the ON time of the switching element Q1 is variably controlled without detecting a zero-cross of the load current. Accordingly, in the present embodiment, as shown in Fig. 13B , the switching element Q1 is controlled in a so-called current continuous mode in which the load current continuously flows without being cut midway.
- the switching element Q1 is controlled in the current continuous mode but the same effect as that of the first embodiment can be obtained. Also, in the present invention, although a signal from the mono-stable multivibrator 20E is inputted to the zero current detection circuit 20A, the zero current detection circuit 20A is not necessarily required. Namely, a configuration, in which, after the switching element Q1 is changed to be turned off, a high signal is inputted to the OR circuit OR1 after the lapse of certain time, is desirable.
- the present embodiment features that, in the zero current detection circuit 20A, the switching element Q1 is controlled based on the first peak value Ith1 and the second peak value Ith2 of the load current, instead of detecting a zero cross of the load current.
- the driving controller 20 includes a comparator COM2 in which the detection voltage VR1 is inputted to an inverting input terminal and the reference voltage Vth1 is inputted to a non-inverting input terminal through an attenuator 20F. Further, the attenuator 20F attenuates the reference voltage Vth1 by K1 times (K1 ⁇ 1). An output terminal of the comparator COM2 is connected to the zero current detection circuit 20A.
- the first peak value Ith1 and the second peak value Ith2 of the load current are set by the comparators COM1 and COM2, respectively. That is, with regard to the comparator COM1, as in the first embodiment, a constant voltage from the constant voltage supply VS1 or the charge voltage from the capacitor C1 in the threshold value adjusting unit 21 is inputted as the reference voltage Vth1 to the inverting input terminal. Accordingly, the driving controller 20 controls the switching element Q1 by using the first peak value Ith1 of the load current as the upper limit value.
- the constant voltage from the constant voltage supply VS1 or the charge voltage from the capacitor C1 in the threshold value adjusting unit 21 is attenuated by the attenuator 20F and then inputted to the non-inverting input terminal.
- the comparator COM2 when the detection voltage VR1 is lower than the input voltage of the non-inverting input terminal, a high signal is outputted to the zero current detection circuit 20A.
- the zero current detection circuit 20A inputs the high signal to the OR circuit OR1. Accordingly, the driving controller 20 controls the switching element Q1 by using the second peak value Ith2 of the load current as a lower limit value.
- a signal is inputted from the zero current detection circuit 20 to the OR circuit OR1, but the zero current detection circuit 20A is not necessarily required. Namely, it may be configured such that when an output signal from the comparator COM2 is changed to be high level, the high signal may be inputted to the OR circuit OR1.
- the present embodiment features that the lighting unit 1 is configured as a boost chopper circuit. Also, in order to reduce ripple of the load current, the smoothing capacitor C2 is connected to the light source unit 3 in parallel.
- a current which is equivalent to a load current, flows through the diode D1 during an OFF period of the switching element Q1, as shown in Fig. 15C .
- the threshold value down period TD1 is provided as in the first embodiment, the same effect as that of the first embodiment can be obtained.
- the lighting unit 1 may be configured as a buck-boost chopper circuit.
- the smoothing capacitor C2 is connected to the light source unit 3 in parallel.
- a current flows through the diode D1 during the OFF period of the switching element Q1 to obtain the same effect as that of in the first embodiment.
- a lighting device in accordance with any of the foregoing embodiments may be used as a lighting device A1.
- this embodiment is a power source separation type illumination apparatus in which a power supply unit and the lighting device A1 are disposed to be separated from the light source unit 3, and a main body 5 for accommodating the light source unit 3 is disposed to be buried in a ceiling 8.
- the main body 5 is made of a metallic material such as, e.g., an aluminum dicast or the like, and has a cylindrical shape with a bottom portion having an opening.
- the light source unit 3 including multiple (three in the drawing) of LEDs 30 and a substrate 31 mounting the respective LEDs 30 thereon is disposed bellow a ceiling portion within the main body 5. Further, the respective LEDs 30 are disposed such that a light irradiation direction faces downward to irradiate a light to an external space through the bottom portion of the main body 5. Further, a light-transmitting plate 6 is provided on the opening of the bottom portion of the main body 5 in order to diffuse light from the respective LEDs 30.
- the lighting device A1 is disposed at a different position from that of the main body 5 on a rear surface of the ceiling 8, and the lighting device A1 and the light source unit 3 are connected by a lead wire 7 through a connector 70.
- the present embodiment as described above, which uses the lighting device A1 of any of the foregoing embodiments, can obtain the same effect as that of any of the foregoing embodiments.
- the present embodiment may be provided with an illumination apparatus of power supply integration type in which the lighting device A1 is installed along with the light source unit 3 in the main body 5.
- a heat dissipation plate 50 formed of an aluminum plate or a copper plate may be disposed to be in contact with the main body 5 on the rear surface of the substrate 31. Accordingly, a heat generated from the respective LEDs 30 can be released to the outside through the heat dissipation plate 50 and the main body 5.
- the foregoing first to tenth embodiments and the circuits of the respective drawings may be appropriately combined to be used.
- the AC-DC converter in Fig. 4A may be applied to the lighting device of the first embodiment
- the boost chopper circuit or the buck-boost chopper circuit of the tenth embodiment may be applied to the lighting device of the first embodiment.
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Description
- The present invention relates to a lighting device for lighting a solid-state light emitting element such as an LED (Light-Emitting Diode), an OLED (Organic Light-Emitting Diode) or the like, and an illumination apparatus using the same.
- Conventionally, a power feeding assembly (lighting device) for feeding a power to a-light emitting diode (LED) illumination module has been provided, which is disclosed, e.g., in
Japanese Patent Application Publication No. 2006-511078 JP2006-511078A Fig. 17 , the prior art example described inJP2006-511078A control switch 101 configured with a MOSFET which are connected to both ends of aDC power supply 100. In addition, an inductor L10 and anLED illumination module 102 are connected to both ends of the diode D10. A controller 103 generates a dual-PWM (Pulse-Width Modulation) switching signal supplied to a control input unit of acontrol switch 101 through anamplifier 104. The dual-PWM switching signal is a combination of a high-frequency PWM switching signal component and pulse bursts of a low-frequency, i.e., a low-frequency PWM switching signal component. - The controller 103 includes a current mode
pulse width modulator 105, which receives an LED current reference signal, a detection current, and a high-frequency sawtooth wave signal from acurrent supply 106. The current modepulse width modulator 105 generates a high-frequency PWM switching signal component supplied as one input of anAND gate 107, and the other input of theAND gate 107 is a low-frequency PWM switching signal component. An output from theAND gate 107 is supplied to a gate of thecontrol switch 101 through theamplifier 104. - Thus, in the prior art example, an average current flowing through the
LED illumination module 102 can be changed by changing the low-frequency component of the dual-PWM switching signal, and thus, the intensity of light output from theLED illumination module 102 is changed. - However, in the prior art example disclosed in
JP2006-511078A Fig. 18A , when the PWM signal falls during an ON period of thecontrol switch 101, the driving signal from thecontrol switch 101 becomes a low level. In this manner, the ON period of thecontrol switch 101 is changed depending on the change in the ON duty ratio of the PWM signal, and accordingly, a load current flowing through the LED illumination module 102 (light source unit), i.e., a light output from theLED illumination module 102, is changed. Thus, dimming of theLED illumination module 102 is performed by changing the ON duty ratio of the PWM signal. Also, the waveform shown inFig. 18A is an example when thecontrol switch 101 is operated in a critical current mode. - Meanwhile, during an OFF period of the
control switch 101, since a flyback current of the inductor L10 flows to theLED illumination module 102 through the diode D10, although the PWM signal falls during the corresponding period, a light output from theLED illumination module 102 is not changed. That is, as shown inFig. 18A , within the range indicated by the dashed single-dotted line in the same drawing, although the ON duty ratio of the PWM signal is swept, a subsequent ON pulse of the driving signal of thecontrol switch 101 is not generated. For this reason, during the interval indicated by the arrow inFig. 18A , although the ON duty ratio of the PWM signal is swept, the light output from theLED illumination module 102 is not changed. Thus, as shown inFig. 18B , with respect to the ON duty ratio of the PWM signal, the light output from theLED illumination module 102 is changed stepwise. A light output difference by one step is equivalent to a light output of one cycle of the driving signal of thecontrol switch 101. - Thus, in the prior art example described in
JP2006-511078A LED illumination module 102 is changed by one step at a time, causing problems in which the light output is not changed smoothly so that a user can see the notable change. In particular, in the prior art, when theLED illumination module 102 is dimmed at a low luminous flux, the change ratio of the light output from theLED illumination module 102 is increased, and thus, the change is seen further notable. - Further, when the
LED illumination module 102 is imaged through various imaging devices such as a video camera or the like, the frequency of the PWM signal is required to be increased to have a certain value or higher to prevent blinking due to an interference with a frequency of the imaging device from being observed. In this case, however, when the frequency of the PWM signal is increased, the ratio of one period of the driving signal ofcontrol switch 101 to one period of the PWM signal is increased. Then, the light output is increased by one period of the driving signal of thecontrol switch 101 and it is more conspicuously seen such that the light output from theLED illumination module 102 is changed by one step at a time. - In order to avoid this, the frequency of the driving signal of the
control switch 101 is required to be increased, but considering an increase in a switching loss or an upper limit of the frequency of the driving signal in case of driving with a low-priced part such as a general IC, and the like, a desirable high-frequency is hardly guaranteed. -
US 2010/0127672 A1 discloses a power supply device including a DC-DC boost converter and a charge recycling circuit. The DC-DC boost converter is utilized for boosting an input voltage to generate an output voltage, and adjusting a voltage level switching signal. -
US 20030085 749 A1 disclosed a lighting device according to the preamble portion ofclaim 1. -
WO 2007/049198 A1 refers to a system for driving a constant current load comprising the constant current load and a down-converter. The down-converter comprises a series arrangement of a switch and an inductor. This series arrangement is arranged in series with the constant current load. - Therefore, the present invention provides a lighting device capable of smoothly changing a light output from a light source unit in sweeping a PWM signal without making a driving signal of a switching element have a high-frequency, and an illumination apparatus using the same.
- While the invention is defined in the independent claims, further aspects of the invention are set forth in the dependent claims, the drawings and the following description.
- In accordance with an aspect of the present invention, there is provided a lighting device including: a lighting unit for supplying a lighting power to a light source unit including one or more solid-state light emitting elements by using a DC voltage from a power supply unit as an input; and a controller for controlling the lighting unit.
- The lighting unit has a series circuit of an inductor and a switching element, and a diode for recovering stored energy of the inductor for the light source unit during an OFF period of the switching element, and the controller has a unit for intermittently driving an ON/OFF operation of the switching element by a PWM signal and a unit for driving the switching element by a frequency higher than that of the PWM signal during an ON period of the PWM signal. When the PWM signal falls, the controller reduces a peak value of a load current flowing through the light source unit during a certain period.
- The lighting unit may further has a detection circuit for detecting the load current flowing through the light source unit, and the controller may further has: a threshold value adjusting unit for setting and outputting the peak value of the load current; a comparator for comparing an output from the detection circuit with an output from the threshold value adjusting unit: and a driving controller for controlling an ON period of the switching element based on an output from the comparator.
- The threshold value adjusting unit may have a capacitor and a charging/discharging circuit for charging or discharging the capacitor based on the PWM signal, and output a charge/discharge voltage of the capacitor as the output.
- Preferably, the comparator compares a superimposed voltage obtained by superimposing the output from the detection circuit and that from the threshold value adjusting unit, with a certain reference voltage.
- The certain period during which the peak value of the load current is reduced is preferably longer than the OFF period of the switching element during the ON period of the PWM signal.
- When the PWM signal rises, the controller preferably controls the ON period the switching element to increase the peak value of the load current during a certain period.
- Preferably, the lighting unit is a buck chopper circuit.
- The controller may control the switching element in a current critical mode.
- The controller may control the switching element in a current discontinuous mode.
- The controller may control the switching element in a current continuous mode.
- The power supply unit preferably includes an, AC/DC converter unit for converting an AC voltage into a desired DC voltage and outputting the converted DC voltage, or a DC/DC converter unit for converting a DC voltage into a desired DC voltage and outputting the converted DC voltage.
- The DC voltage from the power supply unit may be obtained from an AC/DC converter and a frequency of the PWM signal is 600 Hz or a multiple of 600 Hz.
- In accordance with another aspect of the present invention, there is provide an illumination apparatus including the lighting device described-above and a main body for accommodating at least the light source unit.
- The objects and features of the present invention will become apparent from the following description of embodiments, given in conjunction with the accompanying drawings, in which:
-
Fig. 1 is a schematic circuit diagram showing a first embodiment of a lighting device in accordance with the present invention; -
Figs. 2A and2B are views describing a dimming operation of the lighting device, in whichFig. 2A shows a case in which a threshold value down period is about 1.5 times an off time in one cycle of a switching element during an ON period of a PWM signal, andFig. 2B shows a case in which the threshold value down period is about 3 times the off time in one cycle of the switching element during the ON period of the PWM signal; -
Fig. 3 is a view showing a correlation between an ON duty ratio of the PWM signal and a light output in the lighting device; -
Figs. 4A to 4C are views showing different configurations of the lighting device, in whichFig. 4A is a schematic circuit diagram when an AC/DC converter unit is applied to a power supply unit,Fig. 4B is a schematic circuit diagram when a smoothing capacitor is connected in parallel to a light source unit, andFig. 4C is a schematic circuit diagram when a DC/DC converter unit is applied to the power supply unit; -
Figs. 5A and5B are views illustrating a second embodiment of a lighting device in accordance with the present invention, in whichFig. 5A is a waveform view in case of dimming, andFig. 5B is a view showing a correlation between the ON duty ratio of the PWM signal and a light output; -
Figs. 6A and6B views illustrating a third embodiment of a lighting device in accordance with the present invention, in whichFig. 6A is a schematic circuit diagram, andFig. 6B is a waveform view in case of dimming; -
Figs. 7A and7B are views for explaining an operation of the lighting device, in whichFig. 7A is a waveform view when the ON duty ratio of the PWM signal is small, andFig. 7B is a waveform view when the ON duty ratio of the PWM signal is large; -
Fig. 8 is a view showing a correlation between the ON duty ratio of the PWM signal and a light output in the lighting device; -
Figs. 9A and9B are views showing a fourth embodiment of a lighting device in accordance with the present invention, in whichFig. 9A is a schematic circuit diagram, andFig. 9B is a waveform view in case of dimming; -
Figs. 10A and10B are views showing a fifth embodiment of a lighting device in accordance with the present invention, in whichFig. 10A is a schematic circuit diagram, andFig. 10B is a waveform view in case of dimming; -
Figs. 11A and11B are views showing a sixth embodiment of a lighting device in accordance with the present invention, in whichFig. 11A is a schematic circuit diagram, andFig. 11B is a waveform view in case of dimming; -
Figs. 12A and12B are views showing a seventh embodiment of a lighting device in accordance with the present invention, in whichFig. 12A is a schematic circuit diagram, andFig. 12B is a waveform view in case of dimming; -
Figs. 13A and13B are views showing an eighth embodiment of a lighting device in accordance with the present invention, in whichFig. 13A is a schematic circuit diagram, andFig. 13B is a waveform view in case of dimming; -
Figs. 14A and14B are views showing a ninth embodiment of a lighting device in accordance with the present invention, in whichFig. 14A is a schematic circuit diagram, andFig. 14B is a waveform view in case of dimming; -
Figs. 15A, 15B , and15C are views showing a tenth embodiment of a lighting device in accordance with the present invention, in whichFig. 15A is a schematic circuit diagram when a lighting unit is configured as a boost chopper circuit,Fig. 15B is a schematic circuit diagram when a lighting unit is configured as a buck-boost chopper circuit, andFig. 15C is a waveform view in case of dimming; -
Figs. 16A and 16B are views showing an embodiment of an illumination apparatus in accordance with the present invention, in whichFig. 16A is a schematic view of an illumination apparatus of a power source-separation type, andFig. 16B is an illumination apparatus of a power source-integration type; -
Fig. 17 is a schematic circuit diagram of a conventional power feeding assembly for an LED illumination module; and -
Figs. 18A and 18B are views for explaining the problems of the conventional power feeding assembly for the LED illumination module, in whichFig. 18A is a waveform view in case of dimming, andFig. 18B is a view showing a correlation between an ON duty ratio of a PWM signal and a light output. - Hereinafter, a first embodiment of a lighting device in accordance with the present invention will be described with reference to the accompanying drawings. As shown in
Fig. 1 , the present embodiment includes alighting unit 1 for supplying a lighting power to alight source unit 3 by stepping down a DC voltage from a DC power supply (power supply unit) DC1, and acontroller 2 for controlling an output from thelighting unit 1. - The
lighting unit 1 includes a series circuit of a switching element Q1, an inductor L1, and a resistor R1 connected to both ends of the DC power supply DC1. In addition, thelighting unit 1 includes a diode D1 for allowing a flyback current of the inductor L1 to flow during an OFF period of the switching element Q1, and is configured as a buck chopper circuit as a whole. The switching element Q1 is configured with, e.g., an n-channel type MOSFET and performs ON/OFF switching depending on a driving signal applied from a drivingcircuit 20C (to be described later). The resistor R1 detects a current flowing through the inductor L1 through the switching element Q1, whereby a load current flowing through thelight source unit 3 can be detected. One end of a high pressure side of the resistor R1 is connected to a non-inverting input terminal of a comparator COM1 (to be described later). That is, the resistor R1 serves as a detection circuit which detects a voltage between the two ends thereof to thereby detect a load current flowing through thelight source unit 3 through the switching element Q1. - The
controller 2 includes a drivingcontroller 20 for controlling driving of the switching element Q1 of thelighting unit 1 and a thresholdvalue adjusting unit 21 for adjusting a peak value of the load current. The thresholdvalue adjusting unit 21 also serves as a unit for intermittently driving ON/OFF operation of the switching element Q1 by a PWM signal. The drivingcontroller 20 includes a zerocurrent detection circuit 20A for detecting a zero-cross of the load current with a voltage induced to a secondary coil of the inductor L1, a startingcircuit 20B for generating a startup signal, and an OR circuit OR1 to which output signals from the zero-current detection circuit 20A and the startingcircuit 20B are inputted. - Additionally, the driving
controller 20 includes an RS type flipflop FF1, and an output signal from the OR circuit OR1 is inputted to an S terminal of the flipflop FF1. Also, the drivingcontroller 20 includes a drivingcircuit 20C for providing a driving signal to the switching element Q1, and an output signal from a Q terminal of the flipflop FF1 is inputted to thedriving circuit 20C. - Further, the driving
controller 20 includes a comparator COM1 having a non-inverting input terminal to which a detection voltage VR1, which is the voltage between two ends of the resistor R1, is inputted, and an inverting input terminal to which a reference voltage Vth1 (to be described later) is inputted. An output signal from the comparator COM1 is inputted to an R terminal of the flipflop FF1. - The threshold
value adjusting unit 21 includes a parallel circuit of a constant current supply CS1 and a capacitor C1, and a constant voltage supply VS1 connected to one end of a high pressure side of the capacitor C1 through a switching element Q2. An ON/OFF operation of the switching element Q2 is switched by a low-frequency PWM signal. Also, the one end of the high pressure side of the capacitor C1 is connected to the inverting input terminal of the comparator COM1. - Accordingly, when the switching element Q2 is turned on, the constant voltage VRef1 of the constant voltage supply VS1 is applied as the reference voltage Vth1 to the inverting input terminal of the comparator COM1 and the capacitor C1 is charged. Further, when the switching element Q2 is turned off, the charge voltage of the capacitor C1 is applied as the reference voltage Vth1 to the inverting input terminal of the comparator COM1 and the capacitor C1 is discharged by the constant current supply CS1. That is, in the threshold
value adjusting unit 21, the constant voltage supply VS1, the switching element Q2, and the constant current supply CS1 constitute a charging/discharging circuit of the capacitor C1. Namely, an output voltage from the thresholdvalue adjusting unit 21 is a charge/discharge voltage of the capacitor C1. - The
light source unit 3 is configured by connecting multiple (three in the drawing) light emitting diodes (LEDs) 30 in series. Further, in this embodiment, the threeLEDs 30 are used, but one ormore LEDs 30 may be configured. Further, therespective LEDs 30 may be configured to be connected in parallel, rather than in series. Furthermore, in the present embodiment, theLEDs 30 are used in thelight source unit 3, but thelight source unit 3 may also be configured with any other solid-state light emitting element (e.g., organic EL device). - Hereinafter, the operation of the present embodiment will be described with reference to the accompanying drawings. First, when a PWM signal becomes a high level for entering an ON period, a startup signal is inputted to the OR circuit OR1 from the starting
circuit 20B, and a high level set signal is inputted to the S terminal of the flipflop FF1 from the OR circuit OR1. Accordingly, an output signal from the Q terminal of the flipflop FF1 becomes a high level and a driving signal is applied to the switching element Q1 from the drivingcircuit 20C, whereby the switching element Q1 is changed to be turned on. Then, a current flows through thelight source unit 3, the inductor L1, the switching element Q1, and the resistor R1, thus increasing the load current (seeFig. 2A ). At this time, the PWM signal has the ON period, the switching element Q2 of the thresholdvalue adjusting unit 21 is turned on, and the constant voltage VRef1 of the constant voltage supply VS1 is inputted as the reference voltage Vth1 to the inverting input terminal of the comparator COM1. - Since the load current is increased, the voltage between two ends of the resistor R1, i.e., the detection voltage VR1, is increased. And, when the detection voltage VR1 reaches the reference voltage Vth1, the output signal from the comparator COM1 is inverted and a high level reset signal is inputted to the R terminal of the flipflop FF1. Accordingly, the output signal from the Q terminal of the flipflop FF1 becomes a low level and the supply of driving signal to the switching element Q1 from the driving
circuit 20C is stopped, whereby the switching element Q1 is changed to be turned off. - When the switching element Q1 is turned off, a flyback current flows along the closed path of the diode D1, the
light source unit 3, and the inductor L1 by stored energy of the inductor L1. The load current, i.e., the current flowing through the inductor L1 is gradually reduced to be finally zero (seeFig. 2A ). When the current flowing through the inductor L1 reaches zero and the current is inverted by the action of the inductor L1, charges charged in the switching element Q1 are discharged through parasitic capacitance of an element such as the diode D1 or the like, and a voltage between a drain and a source of the switching element Q1 is lowered. Accordingly, the voltage applied to the inductor L1 is inverted, and thus the corresponding inversion is detected by the zerocurrent detection circuit 20A with a voltage induced to the secondary coil of the inductor L1. - When the zero
current detection circuit 20A detects inversion of the voltage applied to the inductor L1, namely, a zero cross of the current flowing through the inductor L1, it inputs a high signal to the OR circuit OR1. Accordingly, a high level set signal is inputted to the S terminal of the flipflop FF1 from the OR circuit OR1. Thus, an output signal from the Q terminal of the flipflop FF1 becomes a high level, and a driving signal is applied to the switching element Q1 from the drivingcircuit 20C, whereby the switching element Q1 is changed to be turned on. By repeatedly performing these sequential operations, the drivingcontroller 20 of thecontroller 2 controls the switching element Q1 in a current critical mode. Also, while the load current flows through thelight source unit 3, therespective LEDs 30 of thelight source unit 3 are turned on. - Next, when the PWM signal has a low level to be shifted to an OFF period, the switching element Q2 is changed to be turned off, and thus, the charge voltage of the capacitor C1 is applied as the reference voltage Vth1 to the inverting input terminal of the comparator COM1. At this time, the capacitor C1 is discharged by the constant current supply CS1, the charge voltage is linearly reduced. Thus, as indicated by the dotted line in
Fig. 2A , the reference voltage Vth1 is also linearly reduced. Herein, after, a time period during which the reference voltage Vth1 reaches zero will be referred to as a 'threshold value down period TD1'. - During the threshold value down period TD1, the ON/OFF operation of the switching element Q1 is controlled by using the reference voltage Vth1 which is gradually reduced as a threshold value. Namely, as indicated by the dotted line in
Fig. 2A , during the threshold value down period TD1, a peak value Ith1 of the load current is linearly reduced and the ON period of one cycle of the switching element Q1 is also reduced depending on the reduction in the peak value Ith1. In other words, when the PWM signal falls, thecontroller 2 controls the peak value Ith1 of the load current to be reduced in a certain time period, the load current flowing through thelight source unit 3. Accordingly, as shown inFig. 2A , the cycle of the driving signal is reduced in comparison to the ON period of the PWM signal during the threshold value down period TD1. - Further, when the reference voltage Vth1 reaches zero, since a high level reset signal is consistently inputted to the R terminal of the flipflop FF1, supply of the driving signal to the switching element Q1 from the driving
circuit 20C is stopped and the switching element Q1 is maintained in an OFF state. Accordingly, until the PWM signal is shifted to be ON period, the load current does not flow to thelight source unit 3, and thus, therespective LEDs 30 of thelight source unit 30 are turned off. - In the present embodiment, by repeatedly performing the foregoing sequential operations, the
light source unit 3 is dimmed by so-called burst dimming that ON/OFF operation of the switching element Q1 is changed by the low-frequency PWM signal. Namely, thecontroller 2 intermittently drives ON/OFF operation of the switching element Q1 to control dimming of thelight source unit 3 and drives the switching element Q1 by a frequency higher than that of the PWM signal, as shown inFig. 2A . Accordingly, in the present embodiment, by changing the ON duty ratio of the PWM signal, the ratio between a turn-on time and a turn-off time of therespective LEDs 30 of thelight source unit 3 can be changed, and dimming of thelight source unit 3 can be executed. - Here, as illustrated by the dashed line in
Fig. 2A , when the ON duty ratio of the PWM signal is swept, the reference voltage Vth1 is linearly reduced as indicated by the dashed single-dotted line. Accordingly, the peak value Ith1 of the load current is also linearly reduced as indicated by the dashed single-dotted line in the same drawing. Namely, when the solid line and the dashed single-dotted line in the same drawing are compared, it can be seen that the peak value Ith1 of the load current in the threshold value down period TD1 is continuously changed depending on a continuous change in the ON duty ratio of the PWM signal. - As described above, in the present embodiment, since the load current, i.e., the light output from the
light source unit 3, is continuously changed depending on the continuous change in the ON duty ratio of the PWM signal, the change in the light output from thelight source unit 3 when the PWM signal is swept can be smoothly made. In particular, in the prior art, whenlight source unit 3 is dimmed at a low luminous flux, since the change ratio of the light output from thelight source unit 3 is increased, the change is notably seen. However, in the present embodiment, even when thelight source unit 3 is dimmed at a low luminous flux, the change in the light output from thelight source unit 3 can be smoothly made. - Further, in case where the
light source unit 3 is viewed through a different imaging device such as a video camera or the like, even when the frequency of the PWM signal is increased to be a certain value or higher to prevent blinking due to the interference with a frequency of the imaging device from being observed, the change of the light output from thelight source unit 3 can be smoothly made. Thus, it is not required to make the driving signal of the switching element Q1 have a high-frequency. - Further, in the dimming shown in
Fig. 2A , the threshold value down period TD1 is about 1.5 times the OFF time T1 in one cycle of the switching element Q1 during the ON period of the PWM signal. This is because, if the threshold value down period TD1 is shorter than the OFF time T1, a triangular wave pulse of the load current is not generated during the threshold value down period TD1 and the light output from thelight source unit 3 is not changed. Thus, in the present embodiment, the threshold value down period TD1 is set to be longer than the OFF time T1. Also, the threshold value down period TD1 can be changed by changing a capacitance value of the capacitor C1 or changing a current value of the constant current supply CS1 in the thresholdvalue adjusting unit 21. - Further, as shown in
Fig. 2B , the threshold value down period TD1 is set to be about 3 times the OFF time T1 to smoothly change the light output from thelight source unit 3 in comparison to the case in which the threshold value down period TD1 is about 1.5 times the OFF time T1 (seeFig. 3 ). This is because, as shown inFig. 2B , since the number of triangular wave pulses of the load current is increased during the threshold value down period TD1, the change in the load current when the ON duty ratio of the PWM signal is swept is close to be linear. - Moreover, in the present embodiment, the DC power supply DC1 is used as a power supply unit, but as shown in
Fig. 4A , the power supply unit may be configured with the AC power supply AC1, an AC/DC converter unit 4 for converting an AC voltage from the AC power supply AC1 into a DC voltage and outputting the same, and a smoothing capacitor C0. Meanwhile, the power supply unit may be configured with the DC power supply DC1 and the DC/DC converter unit for converting a DC voltage from the DC power supply DC1 into a desired DC voltage and outputting the same, as shownFig. 4c . In either case, the same effect can be obtained. - Herein, when a commercial power supply having a power frequency of 50 Hz or 60 Hz is used as the AC power supply AC1, ripples of 100 Hz or 120 Hz are generated at the voltage between two ends of the smoothing capacitor C0 due to the design of the AC/
DC converter unit 4 or the capacity of the smoothing capacitor C0. Then, there is a possibility in which, depending on the frequency of the PWM signal, the low-frequency of the load current is changed due to an interference of the corresponding ripples and the light output from thelight source unit 3 blinks. In order to avoid this, when the power supply unit is configured by using the commercial power supply and the AC/DC converter unit 4, it is preferable to set the frequency of the PWM signal by 600 Hz or a multiple of 600 Hz. Accordingly, the light output from thelight source unit 3 is substantially uniform and can be restrained from blinking due to the interference of ripples. - Further, as shown in
Fig. 4B , in thelighting unit 1, the smoothing capacitor C2 may be provided to be connected in parallel to thelight source unit 3. In this case, since the ripples of the load current flowing through thelight source unit 3 can be reduced to be small, the light output from thelight source unit 3 can be smoothly changed. - In the
lighting unit 1 in accordance with the present embodiment, the switching element Q1 is disposed at a lower pressure side of the DC power supply DC1, but the switching element Q1 may also be disposed at a high pressure side of the DC power supply DC1 to configure thelighting unit 1. - Hereinafter, a second embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 5A , in comparison to the first embodiment, the present embodiment features that the ON duty ratio of the switching element Q1 is large. The reason will be described hereinafter. -
- In the above Eq. 1, 'Id' is a current flowing through the switching element Q1, 'E' is a DC voltage from the DC power supply DC1, 'V' is a load voltage of the
light source unit 3, 'L' is inductance of the inductor L1, and 't' is a lapse time. Also, a turn-on start time of the switching element Q1 is set to be 't=0'. - Herein, the current, i.e., the load current, flowing through the inductor L1 when the switching element Q1 is turned on is the same as the current flowing through the switching element Q1 expressed by Eq. 1. Meanwhile, a change in time of the current, i.e., the load current, flowing through the inductor L1 when the switching element Q1 is turned off is expressed by Eq. 2 shown below:
- In the above Eq. 2, 'IL' is a current flowing through the inductor L1 when the switching element Q1 is turned off, and 'T2' is an ON time in one cycle of the switching element Q1 during the ON period of the PWM signal.
-
-
- In the above Eq. 5, 'Don' denotes the ON duty ratio of the switching element Q1. Thus, it can be seen that the ON duty ratio of the switching element Q1 is determined by the DC voltage from the DC power supply DC1 and the load voltage of the
light source unit 3. - Herein, in consideration of stability of the dimming operation or the accuracy of dimming of the light output from the
light source unit 3, it is preferable that the amount of change of the ON time T2 of the switching element Q2 is larger than that of the ON time of the PWM signal. Further, since the last triangular wave pulse of the load current generated during the threshold value down period TD1 is equivalent to minimum resolution of the load current, i.e., the light output from thelight source unit 3, the light output from thelight source unit 3 can be smoothly changed as the corresponding triangular wave pulse is smaller. When the peak value Ith1 of the load current and the driving frequency of the switching element Q1 during the ON period of the PWM signal are uniform, the corresponding triangular wave pulse is smaller as the ON duty ratio of the switching element Q1 is larger. Thus, the light output from thelight source unit 3 can be more smoothly changed by increasing the ON duty ratio of the switching element Q1. - Hereinafter, a change in the light output from the
light source unit 3 when the ON duty ratio of the switching element Q1 is changed will be described with reference toFig. 5B . - In
Fig. 5B , 'K' is an integer represented as 'K=1/Don'. InFig. 5B , a correlation between the ON duty ratio of the PWM signal and the light output in the prior art example is indicated by the solid line, and in this case, K is assumed to be 10 (K=10). Further, a correlation between the ON duty ratio of the PWM signal and the light output in the case of 'TD1/T1=1.5' in the first embodiment is indicated by the dotted line, and in the corresponding case, K=10 as in the prior art example. - Furthermore, the correlation between the ON duty ratio of the PWM signal and the light output in the case of 'TD1/T1=1.5' in the present embodiment is indicated by the dashed line, and in the corresponding case, K is assumed to be two (K=2). Thus, as noted in
Fig. 5B , as 'K' is smaller, namely, as the ON duty ratio of the switching element Q1 is larger, the light output from thelight source unit 3 can be more smoothly (linearly) changed. - With this regard, in an actual operation, considering the stability of dimming operation and accuracy of dimming of the light output from the
light source unit 3, the DC voltage of the DC power supply DC1 is preferably equal to or less than five times the load voltage of thelight source unit 3. Further, a lower limit of the DC voltage of the DC power supply DC1 is required to be at least larger than the load voltage of thelight source unit 3, i.e., K is greater than one (K>1), to ensure the chopper operation by thelighting unit 1. More preferably, considering the change in the load voltage depending on temperature characteristics of therespective LEDs 30 of thelight source unit 3, K needs to be equal to or greater than 1.2 (K≥1.2). - Hereinafter, a third embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As illustrated in
Figs. 6A and6B , the present embodiment features that a constant current supply CS2 is provided instead of the constant voltage supply VS1 in the thresholdvalue adjusting unit 21, thus linearly increasing the peak value Ith1 of the load current when the PWM signal rises. - Hereinafter, the operation when the PWM signal rises will be described with reference to the accompanying drawings. In the first embodiment, during the ON period of the PWM signal, the constant voltage VRef1 of the constant voltage supply VS1 is inputted as the reference voltage Vth1 to the inverting input terminal of the comparator COM1, but in the present embodiment, a charge voltage of the capacitor C1 is inputted instead.
- First, when the PWM signal rises, the switching element Q2 is changed to be turned on, and the capacitor C1 is charged by the difference between a constant current flowing from the constant current supply CS2 and a constant current flowing from the constant current supply CS1. Accordingly, since the charge voltage of the capacitor C1 is linearly increased, the reference voltage Vth1 is also linearly increased as indicated by the dotted line in
Fig. 6B . A time duration until the reference voltage Vth1 reaches the constant voltage VRef1 is called a "threshold value up period TU1'. During the threshold value up period TU1, the ON/OFF operation of the switching element Q1 is controlled by using the gradually increased reference voltage Vth1 as a threshold value. - An operation after the reference voltage Vth1 reaches the constant voltage VRef1 is the same as that of the first embodiment. Also, a tilt of the reference voltage Vth1 during the threshold value up period TU1 is determined by the charge current of the capacitor C1, namely, by the difference between the constant current flowing from the constant current supply CS2 and the constant current flowing from the constant current supply CS1.
- Herein, when the ON duty ratio of the PWM signal is small (close to 0 %), the reference voltage Vth1 does not reach the constant voltage Vref1 during the threshold value up period TU1 as indicated by the dotted line in
Fig. 7A . Thus, the peak value Ith1 of the load current during the threshold value up period TU1 is continuously changed depending on a continuous change in the ON duty ratio of the PWM signal. For this reason, as the ON duty ratio of the PWM signal is close to 0 %, the peak value Ith1 of the load current is continuously reduced to zero. - Further, when the ON duty ratio of the PWM signal is large (close to 100 %), the reference voltage Vth1 does not reach zero during the threshold value down period TD1 and the threshold value up period TU1 as indicated by the dotted line in
Fig. 7B . Thus, as the ON duty ratio of the PWM signal is close to 100 %, the peak value Ith1 of the load current is continuously increased until the light output from thelight source 3 is maximized. - Hereinafter, a change in the light output from the
light source unit 3 when the ON duty ratio of the switching element Q1 is changed will be described with reference toFig. 8 . InFig. 8 , a correlation between the ON duty ratio of the PWM signal and the light output in case of 'K=2' in the second embodiment is indicated by a dashed line. Also, inFig. 8 , a correlation between the ON duty ratio of the PWM signal and the light output in the case where the threshold value up period TU1 is considered (i.e., in case of employing the present embodiment) under the foregoing condition is indicated by a dotted line. - As can be seen from
Fig. 8 , since the threshold value up period TU1 is provided, the light output from thelight source unit 3 can be smoothly changed from an almost zero to a maximum output. In particular, by setting the threshold value up period TU1 and the threshold value down period TD1 such that they are almost equal, the ON duty ratio of the PWM signal and the light output from thelight source unit 3 have an almost proportional relationship, which is preferable. - Hereinafter, a fourth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since, a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Figs. 9A and9B , the present embodiment features that the constant voltage VRef1 is inputted to the inverting input terminal of the comparator COM1 of the drivingcontroller 20 and an superimposed voltage V1 (to be described later) is increased during the OFF period of the PWM signal, thereby reducing the peak value Ith1 of the load current. - In the threshold
value adjusting unit 21, the constant current supply CS1 and the capacitor C1 are connected in series and the capacitor C1 and the switching element Q2 are connected in parallel. Thus, the capacitor C1 is discharged during the ON period of the PWM signal and it is charged by the constant current from the constant current supply CS1 during the OFF period of the PWM signal. In addition, the resistor R3 is connected in series to the capacitor C1 and the resistor R2 is connected in series to the resistor R1 of thelighting unit 1. Further, a connection point of the resistors R2 and R3 is connected to the non-inverting input terminal of the comparator COM1. - Thus, the charge voltage V1, which is the sum of the voltages obtained by respectively multiplying coefficients determined in the resistors R2 and R3 to the detection voltage VR1, as the voltage between two ends of the resistor R1, and the charge voltage of the capacitor C1, is inputted to the non-inverting input terminal of the comparator COM1.
- Hereinafter, the operation of the present embodiment will be described with reference to
Fig. 9B . During the ON period of the PWM signal, the switching element Q2 is in an ON state, and thus, the capacitor C1 is not charged. Therefore, since the superimposed voltage V1 based only on the detection voltage VR1 is inputted to the non-inverting input terminal of the comparator COM1, the switching element Q1 is repeatedly turned on and off periodically, and the peak value Ith1 of the load current becomes uniform. - Further, when the PWM signal is shifted into the OFF period, the switching element Q2 is changed to be turned off, and thus, the capacitor C12 starts to be charged. Thus, the superimposed voltage V1 based on the detection voltage VR1 and the charge voltage of the capacitor C1 is inputted to the non-inverting input terminal of the comparator COM1. Herein, as indicated by the dashed single-dotted line in
Fig. 9B , the charge voltage of the capacitor C1 is linearly increased with the lapse of time, and finally, is higher than the reference voltage VRef1. For this reason, during the OFF period of the PWM signal, since the superimposed voltage V1 is gradually increased, the cycle of the switching element Q1 is gradually reduced and the peak value Ith1 of the load current is linearly reduced. Namely, during the OFF period of the PWM signal, as in the first embodiment, the threshold value down period TD1 can be provided. - As described above, in the present embodiment, as in the first embodiment, the threshold value down period TD1 can be provided, and therefore, the same effect as that of the first embodiment can be obtained.
- Here, it may be considered that the
controller 2 is configured by using a general PFC (Power Factor Correction) control IC such as MC33262 of ON Semiconductor or L6562 of ST Micro Electronics in order to eliminate harmonics. Since the general PFC control IC has a reference voltage supply therein, in the configuration of the first embodiment, the reference voltage Vth1 cannot be variably controlled, and thus, the peak value Ith1 of the load current cannot be variably controlled. Meanwhile, in the configuration of this embodiment, the peak value Ith1 of the load current can be variably controlled even when the global PFC control IC is utilized, whereby the number of components constituting thecontroller 2 can be reduced. - Hereinafter, a fifth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the fourth embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 10A , the present embodiment features that a series circuit of the constant voltage supply VS1 and a resistor R4 instead of the constant current supply CS1 is provided in the thresholdvalue adjusting unit 21. - In the fourth embodiment, during the OFF period of the PWM signal, the charge voltage of the capacitor C1 is linearly increased by the constant current of the constant current supply CS1. Meanwhile, in the present embodiment, since the resistor R4 and the capacitor C1 constitute an integrator circuit, the charge voltage of the capacitor C1 is exponentially increased as shown in
Fig. 10B . Thus, during the threshold value down period TD1, the peak value Ith1 of the load current is also exponentially reduced. - As described above, in the present embodiment, since the constant voltage supply VS1 and the resistor R4 are used without the constant current CS1, the same effect as that of the fourth embodiment can be obtained.
- Hereinafter, a sixth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Here, a basic configuration of the present embodiment is common to that of the fifth embodiment, so the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 11A , this embodiment features that a resistor R5 is connected in series to the switching element Q2 in the thresholdvalue adjusting unit 21. - In the fifth embodiment, when the PWM signal is shifted from the OFF period to the ON period, the switching element Q2 is changed to be turned ON and shorted, the superimposed voltage V1 becomes zero almost in a moment. Meanwhile, in the present embodiment, since the resistor R5 and the capacitor C1 constitute an integrator circuit, the capacitor C1 is discharged and the charge voltage is exponentially reduced, and thus, the superimposed voltage V1 is also exponentially reduced when the PWM signal is shifted from the OFF period to the ON period, as shown in
Fig. 11B . Thus, when the PWM signal is shifted from the OFF period to the ON period, the peak value Ith1 of the load current is linearly increased. Namely, during the ON period of the PWM signal, likewise as in the third embodiment, the threshold value up period TU1 can be provided. - As described above, in the present embodiment, since the constant voltage supply VS1 and the resistors R4 and R5 are used without the constant current CS1, the same effect as that of the third and fourth embodiments can be obtained.
- Hereinafter, a seventh embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 12A , the present embodiment features that anoscillator 20D for outputting an oscillation signal having a certain cycle, instead of the secondary coil of the inductor L1, is connected to the zerocurrent detection circuit 20A of the drivingcontroller 20. - The zero
current detection circuit 20A inputs a high signal to the OR circuit OR1 periodically based on the cycle of the oscillation signal applied from theoscillator 20D. Namely, in the present embodiment, only the ON time of the switching element Q1 is variably controlled, and the switching element Q1 is driven periodically, without detecting a zero cross of the load current. Accordingly, in the present embodiment, as shown inFig. 12B , the switching element Q1 is controlled in a so-called current discontinuous mode in which the load current intermittently flows. - As described above, in the present embodiment, the switching element Q1 is controlled in the current discontinuous mode, but the same effect as that of the first embodiment can be obtained unlike the first embodiment. Further, in the present embodiment, the oscillation signal of the
oscillator 20D is inputted to the zerocurrent detection circuit 20A, but the zerocurrent detection circuit 20A is not necessarily required and, e.g., a universal PWM control IC may be configured instead. Namely, a configuration, in which a high signal is inputted to the OR circuit OR1 periodically, is desirable. - Hereinafter, an eighth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 13A , the present embodiment features that a mono-stable multivibrator 20E, instead of the secondary coil of the inductor L1, is connected to the zerocurrent detection circuit 20A of the drivingcontroller 20. - The mono-
stable multivibrator 20E is connected to thedriving circuit 20C, and after the driving signal from the drivingcircuit 20C is changed to be a low level, the signal is inputted to the zerocurrent detection circuit 20A after the lapse of a certain period of time. When the signal is inputted from the mono-stable multivibrator 20E, the zerocurrent detection circuit 20A inputs a high signal to the OR circuit OR1. Namely, in the present embodiment, the OFF time of the switching element Q1 is made constant and only the ON time of the switching element Q1 is variably controlled without detecting a zero-cross of the load current. Accordingly, in the present embodiment, as shown inFig. 13B , the switching element Q1 is controlled in a so-called current continuous mode in which the load current continuously flows without being cut midway. - As described above, in the present embodiment, different from the first embodiment, the switching element Q1 is controlled in the current continuous mode but the same effect as that of the first embodiment can be obtained. Also, in the present invention, although a signal from the mono-
stable multivibrator 20E is inputted to the zerocurrent detection circuit 20A, the zerocurrent detection circuit 20A is not necessarily required. Namely, a configuration, in which, after the switching element Q1 is changed to be turned off, a high signal is inputted to the OR circuit OR1 after the lapse of certain time, is desirable. - Hereinafter, a ninth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Figs. 14A and14B , the present embodiment features that, in the zerocurrent detection circuit 20A, the switching element Q1 is controlled based on the first peak value Ith1 and the second peak value Ith2 of the load current, instead of detecting a zero cross of the load current. - The driving
controller 20 includes a comparator COM2 in which the detection voltage VR1 is inputted to an inverting input terminal and the reference voltage Vth1 is inputted to a non-inverting input terminal through anattenuator 20F. Further, theattenuator 20F attenuates the reference voltage Vth1 by K1 times (K1<1). An output terminal of the comparator COM2 is connected to the zerocurrent detection circuit 20A. - In the present embodiment, the first peak value Ith1 and the second peak value Ith2 of the load current are set by the comparators COM1 and COM2, respectively. That is, with regard to the comparator COM1, as in the first embodiment, a constant voltage from the constant voltage supply VS1 or the charge voltage from the capacitor C1 in the threshold
value adjusting unit 21 is inputted as the reference voltage Vth1 to the inverting input terminal. Accordingly, the drivingcontroller 20 controls the switching element Q1 by using the first peak value Ith1 of the load current as the upper limit value. - Meanwhile, with regard to the comparator COM2, as mentioned above, the constant voltage from the constant voltage supply VS1 or the charge voltage from the capacitor C1 in the threshold
value adjusting unit 21 is attenuated by theattenuator 20F and then inputted to the non-inverting input terminal. Thus, in the comparator COM2, when the detection voltage VR1 is lower than the input voltage of the non-inverting input terminal, a high signal is outputted to the zerocurrent detection circuit 20A. When the high signal is inputted from the comparator COM2, the zerocurrent detection circuit 20A inputs the high signal to the OR circuit OR1. Accordingly, the drivingcontroller 20 controls the switching element Q1 by using the second peak value Ith2 of the load current as a lower limit value. - As described above, in the present embodiment, the switching element Q1 is controlled based on the first peak value Ith1 and the second peak value Ith2 of the load current, thus being controlled in the current continuous mode as in the eighth embodiment. Accordingly, in the present embodiment, the same effect as that of the first embodiment can also be obtained. In addition, in the present embodiment, the switching element Q1 can be controlled in the critical current mode by increasing the attenuation factor of the
attenuator 20F to bring the second peak value Ith2 of the load current close to zero. - Further, in the present embodiment, a signal is inputted from the zero
current detection circuit 20 to the OR circuit OR1, but the zerocurrent detection circuit 20A is not necessarily required. Namely, it may be configured such that when an output signal from the comparator COM2 is changed to be high level, the high signal may be inputted to the OR circuit OR1. - Hereinafter, a tenth embodiment of the lighting device in accordance with the present invention will be described with reference to the accompanying drawings. Since a basic configuration of the present embodiment is common to that of the first embodiment, the same reference numerals are used for the common parts and a description thereof will be omitted. As shown in
Fig. 15A , the present embodiment features that thelighting unit 1 is configured as a boost chopper circuit. Also, in order to reduce ripple of the load current, the smoothing capacitor C2 is connected to thelight source unit 3 in parallel. - When the
lighting unit 1 is configured as the boost chopper circuit, a current, which is equivalent to a load current, flows through the diode D1 during an OFF period of the switching element Q1, as shown inFig. 15C . Further, in the present embodiment, since the threshold value down period TD1 is provided as in the first embodiment, the same effect as that of the first embodiment can be obtained. - Further, as shown in
Fig. 15B , thelighting unit 1 may be configured as a buck-boost chopper circuit. In order to reduce ripple of the load current, the smoothing capacitor C2 is connected to thelight source unit 3 in parallel. Also, in this case, as shown inFig. 15C , a current flows through the diode D1 during the OFF period of the switching element Q1 to obtain the same effect as that of in the first embodiment. - Hereinafter, an embodiment of an illumination apparatus in accordance with the present invention will be described with reference to the accompanying drawings. Also, the up and down direction in
Fig. 16A is referred to as a vertical direction in the following description. Further, in the present embodiment, a lighting device in accordance with any of the foregoing embodiments may be used as a lighting device A1. As illustrated inFig. 16A , this embodiment is a power source separation type illumination apparatus in which a power supply unit and the lighting device A1 are disposed to be separated from thelight source unit 3, and amain body 5 for accommodating thelight source unit 3 is disposed to be buried in aceiling 8. - The
main body 5 is made of a metallic material such as, e.g., an aluminum dicast or the like, and has a cylindrical shape with a bottom portion having an opening. Thelight source unit 3 including multiple (three in the drawing) ofLEDs 30 and asubstrate 31 mounting therespective LEDs 30 thereon is disposed bellow a ceiling portion within themain body 5. Further, therespective LEDs 30 are disposed such that a light irradiation direction faces downward to irradiate a light to an external space through the bottom portion of themain body 5. Further, a light-transmittingplate 6 is provided on the opening of the bottom portion of themain body 5 in order to diffuse light from therespective LEDs 30. The lighting device A1 is disposed at a different position from that of themain body 5 on a rear surface of theceiling 8, and the lighting device A1 and thelight source unit 3 are connected by alead wire 7 through aconnector 70. - The present embodiment as described above, which uses the lighting device A1 of any of the foregoing embodiments, can obtain the same effect as that of any of the foregoing embodiments. In addition, as shown in
Fig. 16B , the present embodiment may be provided with an illumination apparatus of power supply integration type in which the lighting device A1 is installed along with thelight source unit 3 in themain body 5. In this configuration, aheat dissipation plate 50 formed of an aluminum plate or a copper plate may be disposed to be in contact with themain body 5 on the rear surface of thesubstrate 31. Accordingly, a heat generated from therespective LEDs 30 can be released to the outside through theheat dissipation plate 50 and themain body 5. - Further, the foregoing first to tenth embodiments and the circuits of the respective drawings may be appropriately combined to be used. For example, the AC-DC converter in
Fig. 4A may be applied to the lighting device of the first embodiment, and the boost chopper circuit or the buck-boost chopper circuit of the tenth embodiment may be applied to the lighting device of the first embodiment. - In accordance with the present invention, it is possible smoothly change a light output from a light source unit in sweeping a PWM signal without making a driving signal of a switching element have a high-frequency.
- While the invention has been shown and described with respect to the embodiments, the present invention is not limited thereto. It will be understood by those skilled in the art that various changes and modifications may be made without departing from the scope of the invention as defined in the following claims.
Claims (13)
- A lighting device comprising:a lighting unit(1) for supplying a lighting power to a light source unit(3) including one or more solid-state light emitting elements by using a DC voltage from a power supply unit(DC1, AC1) as an input; anda controller(2) for controlling the lighting unit(1),wherein the lighting unit(1) includes, a series circuit of an inductor(L1) and a switching element(Q1), and a diode(D1) for recovering a stored energy of the inductor(L1) for the light source unit(3) during an OFF period of the switching element(Q1), characterized in thatthe controller(2) includes a unit(21, COM1, FF1) for intermittently driving an ON/OFF operation of the switching element(Q1) by a PWM signal and a unit(20C) for driving the switching element(Q1) by a frequency higher than that of the PWM signal during an ON period of the PWM signal, and when the PWM signal falls, the controller(2) reduces a peak value of a load current flowing through the light source unit(3) during a certain period,wherein the certain period is set to be longer than an OFF time(T1) in one cycle of the switching element(Q1) during the ON period of the PWM signal.
- The lighting device of claim 1, wherein the lighting unit(1) further includes a detection circuit(R1) for detecting the load current flowing through the light source unit(3), and
the controller(2) further includes: a threshold value adjusting unit(21) for setting and outputting the peak value of the load current; a comparator(COM1) for comparing an output from the detection circuit(R1) with an output from the threshold value adjusting unit(21): and a driving controller(20) for controlling an ON period of the switching element(Q1) based on an output from the comparator(COM1). - The lighting device of claim 2, wherein the threshold value adjusting unit(21) has a capacitor(C1) and a charging/discharging circuit(21) for charging or discharging the capacitor(C1) based on the PWM signal, and outputs a charge/discharge voltage of the capacitor(C1) as the output.
- The lighting device of claim 2 or 3, wherein the comparator(COM1) compares a superimposed voltage obtained by superimposing the output from the detection circuit(R1) and that from the threshold value adjusting unit(21), with a certain reference voltage.
- The lighting device of any one of claims 1 to 4, wherein the certain period during which the peak value of the load current is reduced is longer than the OFF period of the switching element(Q1) during the ON period of the PWM signal.
- The lighting device of any one of claims 1 to 5, wherein when the PWM signal rises, the controller(2) controls the ON period the switching element(Q1) to increase the peak value of the load current during a certain period.
- The lighting device of any one of claims 1 to 6, wherein the lighting unit(1) is a buck chopper circuit.
- The lighting device of any one of claims 1 to 7, wherein the controller(2) controls the switching element(Q1) in a current critical mode.
- The lighting device of any one of claims 1 to 7, wherein the controller(2) controls the switching element(Q1) in a current discontinuous mode.
- The lighting device of any one of claims 1 to 7, wherein the controller(2) controls the switching element(Q1) in a current continuous mode.
- The lighting device of any one of claims 1 to 10, wherein the power supply unit(DC1, AC1) includes an AC/DC converter unit(4) for converting an AC voltage into a desired DC voltage and outputting the converted DC voltage, or a DC/DC converter unit(5) for converting a DC voltage into a desired DC voltage and outputting the converted DC voltage.
- The lighting device of any one of claims 1 to 10, wherein the DC voltage from the power supply unit(DC1, AC1) is obtained from an AC/DC converter(4) and a frequency of the PWM signal is 600 Hz or a multiple of 600 Hz.
- An illumination apparatus comprising the lighting device described in any one of claims 1 to 12 and a main body for accommodating at least the light source unit(3).
Applications Claiming Priority (1)
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JP2011062611A JP5576818B2 (en) | 2011-03-22 | 2011-03-22 | Lighting device and lighting fixture using the same |
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- 2012-03-06 EP EP12001522.7A patent/EP2503846B1/en not_active Not-in-force
- 2012-03-08 CN CN201210059662.7A patent/CN102695327B/en not_active Expired - Fee Related
- 2012-03-09 US US13/416,037 patent/US8749149B2/en not_active Expired - Fee Related
Also Published As
Publication number | Publication date |
---|---|
US20120242235A1 (en) | 2012-09-27 |
JP2012199392A (en) | 2012-10-18 |
US8749149B2 (en) | 2014-06-10 |
EP2503846A1 (en) | 2012-09-26 |
CN102695327B (en) | 2014-08-06 |
CN102695327A (en) | 2012-09-26 |
JP5576818B2 (en) | 2014-08-20 |
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