EP1872634A1 - Ballast electronique a reduction des oscillations de courant reactif - Google Patents

Ballast electronique a reduction des oscillations de courant reactif

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Publication number
EP1872634A1
EP1872634A1 EP06722663A EP06722663A EP1872634A1 EP 1872634 A1 EP1872634 A1 EP 1872634A1 EP 06722663 A EP06722663 A EP 06722663A EP 06722663 A EP06722663 A EP 06722663A EP 1872634 A1 EP1872634 A1 EP 1872634A1
Authority
EP
European Patent Office
Prior art keywords
inductance
voltage
electronic ballast
phase
input capacitance
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
EP06722663A
Other languages
German (de)
English (en)
Other versions
EP1872634B1 (fr
Inventor
Klaus Fischer
Josef Kreittmayr
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Osram GmbH
Original Assignee
Patent Treuhand Gesellschaft fuer Elektrische Gluehlampen mbH
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Patent Treuhand Gesellschaft fuer Elektrische Gluehlampen mbH filed Critical Patent Treuhand Gesellschaft fuer Elektrische Gluehlampen mbH
Publication of EP1872634A1 publication Critical patent/EP1872634A1/fr
Application granted granted Critical
Publication of EP1872634B1 publication Critical patent/EP1872634B1/fr
Expired - Fee Related legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Classifications

    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/36—Controlling
    • H05B41/38—Controlling the intensity of light
    • H05B41/39—Controlling the intensity of light continuously
    • H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor
    • H05B41/3921—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations
    • H05B41/3924—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations by phase control, e.g. using a triac
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
    • H05B41/285—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2851—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2856—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against internal abnormal circuit conditions
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y10—TECHNICAL SUBJECTS COVERED BY FORMER USPC
    • Y10S—TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y10S315/00—Electric lamp and discharge devices: systems
    • Y10S315/04—Dimming circuit for fluorescent lamps

Definitions

  • the present invention relates to an input capacitance having electronic ballast with a step-up converter for operating a discharge lamp, such as a low-pressure discharge lamp on a built-in or parasitic inductance having phase gating dimmer.
  • Electronic ballasts for the operation of discharge lamps are known in many designs. l.d.R. They include a rectifier circuit for rectifying an AC power supply and charging a capacitor often referred to as a DC link capacitor. The voltage applied to this capacitor DC voltage is used to supply an inverter or inverter (hereinafter inverter), which operates the discharge lamp.
  • inverter inverter
  • an inverter generates from a rectified AC power supply or a DC voltage supply a supply voltage for the lamp to be operated with high-frequency current.
  • Similar devices are also known for other lamp types, for example in the form of electronic transformers for halogen lamps.
  • Step-up converter circuits can be used for reducing the mains current harmonic of discharge lamps.
  • Step-up converters have a storage inductor, a switching element, a diode and an intermediate circuit capacitor.
  • the DC link capacitor supplies, for example, a low-pressure discharge lamp via an inverter circuit.
  • Such a boost converter operates as follows: The AC line voltage is converted in a rectifier into a pulsating DC voltage. Between the supply potential of this pulsating DC voltage and the DC link capacitor, the storage inductor and the diode is connected. When switched on, the switching element ensures an increasing current flow in the storage choke up to an adjustable value, the switch-off current threshold. After switching off the switching element, the diode conducts the current impressed in the storage inductor into the intermediate circuit capacitor.
  • Phase control dimmers for power control are also known. Phase gating dimmers provide a periodic mains supply to the load. In each half-cycle, however, the mains supply is only delivered to the load after a settable time.
  • phase gating dimmers as a switching element controlling the current flow from a supply network to a load, include a triac. With such a switching element, it is possible to allow a current flow from the network to the load from an adjustable time within a network half-wave.
  • a voltage available At the output of the phase gating dimmer is a voltage available, which is zero in a first time interval, namely in the phase gating, and in a second time interval substantially equal to the input voltage of the dimmer.
  • phase gating dimmers include an inductor connected in series with the switching element.
  • a parasitic inductance can occur between the phase gating dimmer and the capacitive load, for example caused by line inductances. Jegli- rather reference to an "inductance in the phase gating dimmer" is to be understood in the following text in this sense.
  • the invention is based on the technical problem of providing an improved electronic ballast for dimmable discharge lamps with regard to the operating behavior.
  • the invention relates to an electronic ballast with a switching element and an input capacitance having boost converter for operation on a serially acting for serially acting inductance phase gating dimmer, characterized in that within a half-wave, the operating parameters of the boost converter during the Ab- magnetization of the inductance in the phase gating dimmer, in time after completion of the phase control, be set so that, compared with the operation of the boost converter after the demagnetization of the inductance of the dimmer, a temporarily increased current flows through the boost converter.
  • Electronic ballasts for operating discharge lamps often have an effective input capacitance.
  • the invention is based on the consideration that the effective input capacitance of the electronic ballast together with the serially acting inductance of the phase gating dimmer forms a resonant circuit and an overshoot of the voltage across the input capacitance can occur.
  • Such voltage oscillations can impair the operating behavior of electronic ballasts for discharge lamps when operating on a phase-angle dimmer.
  • the switching element is brought into a conductive state in the phase gating dimmer; then the input capacitance of the ballast is charged to the instantaneous value of the supply voltage.
  • This charging of the input capacitance takes place via the inductance of the phase gating dimmer, which determines the increase of the current.
  • the voltage across the input capacitance initially reaches the instantaneous value of the supply voltage, but then goes beyond that. This is done because the inductance in the phase gating dimmer now demagnetizes and maintains a current flow in the original current direction. If the inductance in the phase gating dimmer is demagnetized and the voltage across the input capacitance is greater than the applied supply voltage, then no mains current flows through the ballast until the overvoltage at the input capacitor has been dissipated by discharging.
  • the triacs often used as switching elements in phase gated dimmers require a certain holding current, i. if they are brought into a conductive state, a minimum current is needed to maintain the conductivity. If this is missing, the triac locks again. If there is no current flowing through the phase gating dimmer for a short time, the triac may transition from the conducting to the blocking state.
  • the reactive current oscillations described above can cause such mains power interruptions.
  • the interruption of the mains current can be prevented.
  • a temporarily increased current is passed through the step-up converter, ie within a time interval defined by the demagnetization.
  • the word "during” is to be understood throughout this text as meaning that this current discharges the input capacitance and the voltage above it decreases again to the level of the instantaneous value of the supply voltage be to reduce the voltage over the input capacitance before the inductance in the phase gating dimmer is completely demagnetized.
  • a step-up converter can be operated in various operating modes, with a distinction in particular between the discontinuous operation and the continuous operation. Often boost converters are operated continuously in discontinuous mode. This means that the switching element in
  • Step-up converter is only switched on when the storage reactor of the boost converter is completely demagnetized and no current flows through it. Switching losses are minimal in this mode of operation.
  • the switch-on element in the boost converter is not maintained until the storage choke is completely demagnetized, then this is called continuous operation.
  • the switching element is therefore switched on when falling below a threshold for the current through the storage inductor - the switch-on threshold -.
  • This inrush current threshold can be of different magnitude and take on a different value in each cycle of the boost converter.
  • the step-up converter is operated during the demagnetization of the inductance in Phasenanroughdim- mer with temporarily increased inrush current compared to the operation of the boost converter following the demagnetization of the inductance in the phase gating dimmer.
  • the current flow through the boost converter during this period can be significantly increased.
  • the switching losses in the boost converter temporarily increase as a result of this measure, these losses are not high, averaged over the network half-wave.
  • this may mean that during the demagnetization of the inductance in the phase gating dimmer, the boost converter in the continuous operation and, after this period, either immediately or after a delay, goes into discontinuous operation.
  • the above embodiment of the invention also includes, in particular, the case that following the demagnetization of the inductance in the phase gating dimmer, it is not switched to the discontinuous operation of the boost regulator, but remains in the boost converter in a continuous operation with lower inrush current thresholds of the switching element.
  • the Abschaltstromschwelle the switching element of the boost converter during the demagnetization of the inductance of the phase gating dimmer increases. Even with this measure, as an alternative or in addition to continuous operation, the current flow through the boost converter can be significantly increased.
  • the current flowing through the step-up converter is reduced or even prevented.
  • no or only a small current discharging the input capacitance can flow.
  • the magnetization of the inductance in the phase gating dimmer and thus the energy stored in it can be reduced to a minimum.
  • the current flowing through the boost converter during the magnetization of the inductance in the phase gating dimmer is reduced by the fact that the Abschaltstromschwelle the boost converter is chosen small compared to the operation of the boost converter following the magnetization of the inductor in the dimmer.
  • the high set a smaller amplitude current The average current flowing through the inductance of the phase gating dimmer can thus be set very small, or even negligible.
  • the Abschaltstrom- threshold starting from no or a small current through the switching element during the magnetization of the inductance in Phasenan cut d always, however, so that also absorbed by the boost converter current from switching cycle to switching cycle of the boost converter increases.
  • the switch-off threshold of the boost converter can reach the increased switch-off threshold for the time following the magnetization of the inductance.
  • the switching element of the boost converter is permanently disabled during the magnetization of the inductance. Thus no current discharging the input capacitance can flow.
  • a preferred embodiment of the invention has a circuit arrangement for the metrological detection of the termination of the phase gating, the beginning of the demagnetization of the inductance in the phase gating dimmer and the completion of the demagnetization of the same inductance. These three times determine the two relevant time intervals within which embodiments of the invention work to reduce the voltage overshoot across the input capacitance. Between the conclusion of the phase control and the time at which the voltage across the input capacitance reaches the instantaneous value of the supply voltage, the inductance is magnetized in the phase gating dimmer; From this point on it will be demagnetized.
  • the circuit arrangement preferably consists of a series connection of two differentiators, which may for example be connected in parallel to the input capacitance.
  • the output voltage of the second differentiator corresponds to the second derivative of the voltage across the input capacitance and has the property that it has a different sign during the magnetization of the inductance in the phase gating dimmer than during the demagnetization of the same inductance.
  • the two relevant time intervals are determined and the output signal of the second differentiator can be used to set the operating parameters of the boost converter.
  • the output voltage of the second differentiator can be converted via two threshold elements into signals corresponding to the logic states zero and one. If the output voltage of the second differentiator has the first sign, then the output signal of the first threshold element is logically one, and the same applies to the second threshold element and the time in which a voltage of the other sign is present at the output of the second differentiator.
  • these threshold elements are Schmitt trigger.
  • the voltage across the input capacitance is superimposed by the boost setting function with a high-frequency, comparatively low alternating voltage. These high-frequency vibrations are decoupled from the first differentiator, a second differentiation may not provide a meaningful result.
  • a preferred embodiment of the invention therefore provides a peak detection circuit. By means of peak value detection, the first derivative of the voltage across the input capacitance is smoothed. The quality of a subsequent differentiation increases with it.
  • circuit arrangements just described for determining the relevant time intervals perform a metrological detection of the waveform of the voltage across the input capacitance. Therefore, you can These circuits reliably determine the relevant time intervals under different circumstances. These circuits can be used for various combinations of input capacitance and inductance of the phase gating dimmer, since they measure the course of the voltage across the input capacitance and therefore no assumptions about the input capacitance, the inductance of the phase gating dimmer or the course of the voltage across the input capacitance following include the phase angle.
  • the inventors have found that the values of the inductances of the dimmers available on the market are in a comparatively small range. Furthermore, the voltage overshoot across the input capacitance can also be limited by supplementary measures (see below and claims 14 ff.). If the voltage overshoot above the input capacitance is not so pronounced, the relevant time intervals after the phase gating also do not vary so much.
  • the electronic ballast on a threshold value for detecting the completion of the phase gating and a first timer, which provides a first fixed time and is set by the threshold value element following the phase gating.
  • the threshold value element can be supplied with approximately the rectified AC voltage supply, optionally via a voltage divider.
  • the output signal of the threshold element can correspond to logic zero, while it jumps to logic one directly after the phase control.
  • the first timer can be set, which is for a constant predetermined time, the holding time, remains set. After that it can reset itself independently.
  • the predetermined hold time of the timer can be determined by averaging over one of the relevant time intervals, which takes into account as many dimmers as possible available on the market.
  • the input capacitance of the electronic ballast containing this circuit is known to the manufacturer and can be considered accordingly in the hold time of the first timer.
  • the first timer may, for example, specify a time corresponding to the duration of the magnetization of the inductance in the phase gating dimmer at the end of the phase gating. This corresponds to the time when the supply voltage is greater than the voltage across the input capacitance. If this time has expired, according to the invention, a temporarily increased current can be passed through the step-up converter.
  • the operating parameters of the boost converter can of course also be set so that during the magnetization of the inductance in the phase gating dimmer very little current is passed through the boost converter.
  • the electronic ballast on two timers can be set by the same threshold element following the phase gating, but have different hold times.
  • Both timers can be set by the same threshold element following the phase gating, but have different hold times.
  • an AND link three relevant for the operation of the electronic ballast time intervals can be specified.
  • the first fixed time interval of the first timer begins with the end of the phase gating and lasts at most until the end of the magnetization of the inductance in the phase gating dimmer.
  • the second fixed time interval of the second timer also begins with the end of the phase gating and lasts at least until complete reduction of the voltage overshoot across the input capacitance.
  • the threshold elements are Schmitt triggers.
  • the transition to the subsequent operation with lower inrush current thresholds is preferably carried out slowly. This means that the inrush current thresholds of the switching element distributed in the boost converter become smaller over some current consumption cycles of the boost converter. As a result, further load current oscillations can be reduced.
  • reactive current oscillations can be reduced by means of an adjustment of the time profile of the current through the boost converter.
  • reactive current oscillations can be reduced by suitably charging or discharging the input capacitance before the end of the phase gating, and thus additionally the current charging the inductance in the phase gating dimmer can be reduced quantitatively.
  • a ballast which realizes both possibilities of reactive current reduction, attenuates reactive current oscillations more effectively.
  • the voltage overshoots are particularly pronounced when the voltage across the input capacitance following the phase gating is significantly less than the instantaneous value of the supply voltage.
  • the "instantaneous value of the supply following the phase control" is to be understood as meaning that the supply voltage across the ballast or above the phase gating dimmer has already fully built up.
  • the current flowing during the magnetization of the inductance of the dimmer increases as long as the voltage across the input capacitance is less than the supply voltage at the load.
  • the input capacitance is charged to a value which corresponds at most to the instantaneous value of the supply voltage at the end of the phase control by a charging process (charging or discharging) before the end of the phase control of a mains half-wave.
  • the voltage across the input capacitance should not, however, at that time exceed supply voltage, otherwise no continuous mains current can be guaranteed.
  • the instantaneous value of the supply voltage at the end of the phase control within a system half-cycle is not known in advance.
  • the invention in this embodiment has a memory device for storing a prognosis value for the supply voltage at the time of the end of the phase gating, which was obtained from one or more preceding network half-waves.
  • a storage device for storing a prognosis value for the supply voltage at the time of the end of the phase gating, which was obtained from one or more preceding network half-waves.
  • the invention has a device for storing an instantaneous value of the supply voltage at the end of the phase control of one or more preceding network half-waves.
  • the instantaneous value of the supply voltage at the end of the phase gating of a preceding mains half-wave does not have to be identical to the instantaneous value of the supply voltage at the end of the phase gating of a following mains half-wave - it is rather a forecast for the supply voltage value, as explained in the previous paragraph.
  • the stored value for the current network half-wave is very similar. This is the case because changes in the phase angle between successive line half-waves usually take place slowly. Reactive current oscillations are reduced most effectively when the input capacitance is charged exactly to the value of the supply voltage at the end of the phase gating. However, to be sure that the voltage across the input capacitance is not greater than the supply voltage at the end of the phase gating, the input capacitance is charged to a value slightly less than the stored predictive value.
  • the prognosis value of the supply voltage following the phase gating is newly stored in each network half-wave and used in the respectively following network half-wave.
  • the memory device preferably stores the instantaneous value of the supply voltage within a time window after the completion of the phase control.
  • a peak detection circuit is used for this purpose.
  • the time window can be used for charging a capacitor, for example, but is very short compared to the sinusoidal supply voltage.
  • the time window is preferably set to open and close within a time interval beginning with the gating of the phase gating dimmer and ending with the voltage across the input capacitance reaching the value of the instantaneous supply voltage. This excludes in particular the case that a value is stored which is greater than the supply voltage when switching on the dimmer.
  • a reactive current oscillation can not be ruled out, as no prognosis value has yet been stored. After a few half waves, however, a stable state is reached.
  • the length of the time window is determined by a monoflop. This is set by a signal from a control circuit of the electronic ballast and resets after a given time back.
  • the incipient current flow through the storage choke of the boost converter can trigger the setting of the monoflop.
  • the monoflop defines the time window for the storage of the instantaneous value of the supply voltage at the end of the phase control, for example by means of a switch controlled by the monoflop.
  • the time window is specified in a further preferred embodiment by means of a differentiator of a capacitor and a resistor.
  • the differentiator is addressed by an edge of a signal from a control circuit of the ballast. As a result of the flank occurs across the resistance of the differentiator on a voltage jump followed by an exponential decay.
  • the time constant of the exponential decay is determined by the size of the resistor and the capacitor in the differentiator.
  • the exponential decay defines the time window for the storage of the instantaneous value of the supply voltage.
  • a further preferred embodiment for the determination of a time window and for storing a prognosis value of the supply voltage at the end of the phase gating is based on the following relationship: At the time of the end of the magnetization of the inductance in the dimmer, the instantaneous value of the voltage across the input capacitance corresponds to the instantaneous value of the supply voltage. Since the end of the phase control, the supply voltage has barely changed, then corresponds to the voltage across the input capacitance about the instantaneous value of the supply voltage at the end of the phase control.
  • the time of the end of the magnetization of the inductance in the dimmer corresponds to the zero crossing of the second derivative of the voltage across the input capacitance of the ballast and is easy to determine, if necessary also estimate (as described in the embodiments in connection with FIG. 12). In this case, one can store as a prediction value, the voltage across the input capacitance of the ballast at this time.
  • an embodiment comprises a comparison device. This compares the value from the memory device with the current value of the voltage across the input capacitance. Before the end of the phase control, the comparison device controls the control circuit of the boost converter, which then discharges the input capacitor accordingly. For example, if the voltage across the input capacitance is greater than the stored value, the input capacitance is discharged. In the exemplary embodiment, it is described more concretely how the output signal of the comparison device can contribute to the control of the charging process of the input capacitance.
  • the input capacitance is discharged via activation of the boost regulator before termination of the phase gating.
  • the input capacitance is charged by the DC link capacitor.
  • a diode connected between the supply potential-side terminals of the intermediate circuit capacitor and the input capacitance can be bridged with a resistor.
  • boost converters that have several between the supply potential-side terminals of the DC link capacitor and the input capacitance. having switched diodes; here one or more diodes can be bridged.
  • the boost converter can then be activated in order to discharge the input capacitance to the desired value (at most the forecast value).
  • the invention thus also relates in principle to a method for operating an electronic ballast having a step-up converter having a switching element and an input capacitance on a phase-gating dimmer having a series-acting inductance, characterized in that the operating parameters of the step-up converter during a half-wave Demagnetization of the inductance in the phase gating dimmer, temporally after completion of the phase gating, be set so that, compared with the operation of the boost converter after the demagnetization of the inductance, a temporarily increased current flows through the boost converter.
  • FIG. 1 shows schematically a boost converter as part of an electronic ballast with an upstream phase gating dimmer.
  • FIG. 2 schematically shows the supply voltage UIN for an electronic ballast according to the prior art, the voltage across an input capacitance of a load UC, the line current IN and the average current ILH flowing through the step-up converter. There are three relevant time intervals T1, T2, T3 entered.
  • FIG. 3 shows schematically for an electronic ballast with a first device for reactive current oscillation reduction, the supply voltage UIN, the voltage across the input capacitance of the load UC, the Netzstrom IN and the middle flowing through the boost converter current ILH. There are two relevant time intervals T1, T2 entered.
  • FIG. 4 shows a first circuit arrangement for a reactive current oscillation reduction corresponding to FIG.
  • FIG. 5 shows relevant voltage profiles of the circuit arrangement from FIG. 4.
  • FIG. 6 shows a second circuit arrangement for a reactive current oscillation reduction corresponding to FIG.
  • FIG. 7 shows a third circuit arrangement for reactive-current reduction corresponding to FIG.
  • FIG. 8 shows relevant signal curves of the circuit arrangement from FIG. 7.
  • FIG. 9 schematically shows, for a prior art electronic ballast, the supply voltage UIN, the voltage UC across an input capacitance C of the load, the voltage UL across an inductance of the load
  • FIGS. 10 a, b show schematically a profile of the voltage UC over the input capacitance C during discharging or charging of the input capacitance C and the supply voltage
  • FIG. 11 schematically shows, for an electronic ballast with a second device for reactive current oscillation reduction, the supply voltage UIN, the voltage UC across the input capacitance C of the load, the voltage UL across the inductance of the dimmer and the mains current IN. Again, three relevant time intervals T1, T2, T3 are entered.
  • FIG. 12 a shows a circuit arrangement for storing prognosis values and for comparing a prognosis value with the voltage UC over the input capacitance C.
  • FIG. 12 b shows a variation of the circuit arrangement from FIG. 12 b
  • FIG. 13 shows a variation of the circuit arrangement from FIG
  • Figure 1 shows schematically a boost converter as part of an electronic ballast of a compact fluorescent lamp CFL.
  • the electronic ballast upstream of a phase gating dimmer DIM This has a series circuit of a triac TR and an inductance LP. This series circuit is connected in series in an AC supply line of the electronic ballast. Another AC supply line is passed through the phase gating dimmer DIM. If the triac TR is switched through, then a supply voltage UIN is present between a node between the triac TR and the inductance LP and the other AC voltage supply line. The outputs of the phase gating dimmer DIM are connected to the inputs of a rectifier GL of the electronic ballast.
  • the boost converter is formed by a capacitor C, a DC link capacitor CH, a diode DH, a storage inductor LH and a switching element SH, here a MOSFET.
  • boost converters also include a control circuit not shown here for driving the switching element SH.
  • a control circuit as described in EP 1 465 330 A2 can be used.
  • the DC link capacitor CH is charged via the rectifier GL via the storage inductor LH and the diode DH.
  • the DC link capacitor For example, tor supplies a compact fluorescent lamp CFL via an inverter circuit INV.
  • the circuit operates as follows: The AC line voltage is converted in the rectifier GL in a pulsating DC voltage. Parallel to the rectifier GL, the capacitor C is connected on the DC side for radio interference suppression. In the positive supply line a storage inductor LH is connected. The switching element SH ensures in the on state for a rising to an adjustable value current flow in the storage inductor LH. After switching off the switching element SH, the diode DH conducts the current impressed in the storage inductor LH into the intermediate circuit capacitor CH.
  • the supply voltage UIN 1 shows the voltage across the input capacitance of the load UC, the line current IN and the average current ILH flowing through the step-up converter.
  • the end of the phase angle defines the beginning of the first interval TL
  • a current flow IN from the supply network through the dimmer begins.
  • the increase of the current IN is determined by the inductance of the dimmer.
  • the voltage UC across the input capacitance C increases.
  • the interval T1 ends as soon as the voltage UC across the input capacitance C corresponds to the instantaneous value of the supply voltage UIN.
  • the input capacitance C is further charged by the serial inductance L of the phase gating dimmer.
  • the complete demagnetization of the inductance L defines the end of the interval T2.
  • the voltage across the input capacitance C increases. Her is than the supply voltage UIN, a mains current continues to flow IN, because the inductance demagnetizes in the phase gating dimmer and maintains the flow of IN in the same direction.
  • a small current IN flows from the input capacitance C back to the supply, since the rectifier diodes commutate in the reverse direction.
  • the voltage across the input capacitance C decreases and then reaches the instantaneous value of the supply voltage. This time corresponds to the end of the interval T3.
  • FIG. 3 shows, for an electronic ballast according to the invention, the supply voltage UIN, the voltage across the input capacitance of the load UC, the line current IN and the average current ILH flowing through the step-up converter. Two relevant intervals T1, T2 are entered.
  • a current ILH flows through the step-up converter.
  • This current ILH must be so large that the temporary voltage overshoot above the input capacitance C is not as pronounced as in FIG Within the interval T2, the energy transmitted by ILH must be greater than the energy stored in the serial inductance of the phase gating dimmer L at the beginning of the interval T2.
  • An increased current flow within the interval T2 can be achieved by operating the boost converter temporarily in continuous operation mode, in contrast to the otherwise used discontinuous operating mode.
  • the above result can also be achieved by increasing the Abschaltstromschwelle. If the boost converter operates with an increased turn-off current threshold, a larger average current flows through the storage inductor during the power consumption cycles. So that the storage throttle does not saturate, it may need to be re-dimensioned.
  • FIG. 4 shows a circuit arrangement according to the invention for detecting the limits of the intervals T1 and T2.
  • Threshold components specifically two Schmitt triggers ST1 and ST2, whose outputs mark the intervals T1 and T2, are connected to the connection node between R2 and C3.
  • FIG. 5 shows relevant voltage profiles of the circuit arrangement from FIG. 4.
  • a step function is assumed as the supply voltage UIN.
  • This assumption about the supply voltage UIN is a good approximation for the actual time course of the phase-cut supply voltage on the interesting time scale.
  • the current ILH is neglected by the boost converter. This is only of minor importance for the consideration of the vibration processes when switching through the phase gating dimmer.
  • FIG. 5 shows in the uppermost diagram the profile of the supply voltage UIN and the voltage UC across the capacitive input load. Notwithstanding Figures 2, 3, 9 and 11, the voltage UC is not shown schematically as a linear function, but drawn somewhat more realistic.
  • the voltage UR1 across R1 is proportional to the current loading the input capacitance C.
  • R1 and C2 are dimensioned such that UR1 corresponds to the first derivation of the time course of UC.
  • these are dimensioned so that a voltage drops across the resistor R2, which corresponds to the second derivative of the time course of the voltage UC.
  • the resistor R1 in series with the input capacitance C and to dispense with the capacitor C2.
  • a first Schmitt trigger ST1 generates an output voltage USTA1, which assumes a positive value in the interval T1.
  • the second derivative of UC is positive.
  • USTA1 corresponds to the reference potential.
  • a second Schmitt trigger ST2 generates an output voltage USTA2, which assumes a positive value in the interval T2.
  • During the interval T2 is the second derivative of UC negative. Outside T2, USTA2 is equal to the reference potential.
  • the voltage UC across the input capacitance may be superimposed by high-frequency alternating voltages.
  • the differentiation by the series connection of the capacitor C2 and the resistor R1 primarily decouples the high-frequency AC voltage components.
  • the voltage UR1 may then no longer be meaningfully evaluable for the following differentiator.
  • FIG. 6 shows a correspondingly improved circuit arrangement according to the invention.
  • the capacitor C3 of the second differentiator is no longer connected directly to the connection node of R1 and C2, but via a parallel connection of a diode D1 and a resistor R3.
  • the diode is poled so that a current from C2 to C3 can flow through it, but no current from C3 to C2.
  • another capacitor C4 is used, which is parallel to the series circuit of C3 and R2. With this peak detection circuit, the first derivative of the voltage across the input capacitance UC is smoothed.
  • the peak value of the voltage is stored via R1 via the diode D1.
  • R3 allows slow unloading of C4.
  • FIG. 7 shows a circuit arrangement according to the invention for specifying an estimation of the intervals T1 and T2.
  • a voltage divider of a resistor R4 and a resistor R5 is connected between the DC voltage outputs of the rectifier GL. Above this voltage divider R4, R5, a boost converter input voltage UINL drops. Parallel to the voltage divider R4, R5, a series circuit of a diode DC and the input capacitance C is connected. The diode DC blocks if there is a voltage UC across the input capacitance which is greater than the boost converter input voltage UINL.
  • the center tap of the voltage divider is connected to an input of a threshold value element ST3.
  • the threshold element ST3 is here a Schmitt Trigger and generates an output signal ST3A, which may be logically one or zero.
  • the circuit arrangement has two timers TIM1 and TIM2, to which the output signal ST3A of the threshold value element ST3 is supplied.
  • the timers TIM1 and TIM2 each provide an output signal TIM1 A and TIM2A, which may also be logically one or zero.
  • Figure 8 shows the relevant waveforms of the circuit of Figure 7 together with the supply voltage UIN and the voltage UC across the input capacitance C following the phase gating.
  • the threshold value element ST3 is set via the voltage divider R4, R5.
  • the output signal ST3A of the threshold element jumps from logic zero to logic one. It thus simultaneously sets the two timers TIM1 and TIM2.
  • the timer TIM1 is designed so that its output signal TIM1A jumps back to zero at the latest following the magnetization of the inductance in the phase gating dimmer.
  • the timer TIM2 is designed such that its output signal TIM2A returns to logical zero at the earliest after the complete reduction of the voltage overshoot above the input capacitance.
  • the time in which the output signal TIM1A of the timer TIM1 is logically one thus corresponds to the time interval T1.
  • the output signal TIM2A of the timer TIM2 is logically one during the time intervals T1 and T2.
  • the hold times of the timers TIM1, TIM2 are already specified here by the manufacturer.
  • the two variables which essentially determine the time intervals T1 and T2 are the inductance of the phase gating dimmer and the input capacitance C.
  • the input capacitance C is known to the manufacturer and can easily be taken into account by the latter.
  • the inductance of the The phase-angle dimmer that is actually used can not be considered in advance; instead, an average over the time intervals T1, T2 of the dimmers available on the market is established for determining the holding times of the timers TIM1, TIM2.
  • the difference between permanently set time intervals T1 and T2 and the actual intervals dependent on the inductance in the dimmer is greater the greater the voltage overshoot across the input capacitance C following the phase gating. If further measures are taken to reduce the voltage overshoot across the input capacitance C, then the inductance-related deviations of the actual intervals from the predefined time intervals T1 and T2 have less of an effect on the reduction of the voltage overshoot than without such further measures.
  • One of these further measures will be explained with reference to FIGS. 9 et seq.
  • the circuit arrangements according to the invention described in FIGS. 4, 6 and 7 can advantageously be used with the electronic ballast from EP 1 465 330 A2 by being connected there in parallel to the input capacitance C (C1 in EP 1 465 330 A2).
  • the circuit arrangements control the boost converter so that in the interval T1 the current through LH and thus the current discharging the input capacitance is minimal. This can be achieved by permanently locking the switch SH by controlling the switch SH via the control device of the boost converter of EP 1 465 330 A2 by the voltage signal STA1 from one of the circuit arrangements according to the invention.
  • a temporarily increased average current ILH is to flow through the boost converter in the interval T2.
  • the mode of operation of the boost converter can be varied via the control device from EP 1 465 330 A2 (in EP 1 465 330 A2 the control circuit is designated BCC).
  • the boost converter is operated in the so-called discontinuous mode.
  • the switch SH is always turned on only when in the storage inductor of the boost converter no current flows, so if the storage inductor LH is just completely demagnetized. Switching losses are minimal in this mode of operation.
  • the boost converter is operated in the interval T2 in the continuous mode.
  • the continuous mode is characterized by the fact that it is not as long to wait for the switching element SH to be switched on as in the discontinuous case, so that a current flows continuously through the storage inductor LH.
  • the average current flow is increased by the boost converter in the interval T2 compared to normal operation. Since the interval T2 is short compared to a whole half-wave, the increased switching losses caused are small, negligible.
  • FIG. 9 as in FIG. 2, the supply voltage UIN, the voltage UC across the input capacitance C of the load and the line current IN are first shown in order to understand an electronic ballast according to the prior art.
  • the voltage UL is shown above the inductance in the phase gating dimmer.
  • the three equal intervals " Pl 1 T2, T3 are entered as in FIG.
  • the increase of the current IN is determined by the inductance of the dimmer, the size of the input capacitance C and the voltage UL across the inductance of the dimmer. Note the large peak values of the voltage UL across the inductance in the phase gating dimmer, the voltage UC across the input capacitance C and the mains current IN.
  • the reactive current superimposed on the active current required to supply the discharge lamp is to be reduced in size.
  • This reactive current is caused by the magnetization and demagnetization of the inductance in the phase gating dimmer, continues to charge the input capacitance C during demagnetization T2 of the inductance in the dimmer and causes the voltage overshoot.
  • the current IN through the inductance of the phase gating dimmer increases as long as the voltage UC across the input capacitance C is lower than the supply voltage UIN. This is the case in interval T1.
  • the voltage UC at the end of the phase control corresponds to the instantaneous value of the supply voltage UIN. It will be shown below that it is technically sensible to choose the value of the voltage UC slightly smaller.
  • the instantaneous value of the supply voltage is stored in each line half-wave of the supply network at the end of the phase control; at a cleverly selected time of storage, the stored value corresponds to the instantaneous value of the supply voltage UIN at the end of the phase control.
  • a corresponding circuit will be described below.
  • the input capacitance C is then charged to just before switching on the switching element in the dimmer in the next half-wave (90%) of the value stored in the previous half-wave. It can be assumed that the change made by an operator to the phase section of the dimmer in successive mains half-waves is only slight.
  • FIGS. 10 a and b schematically show the profile of the voltage UC during the discharging or charging of the input capacitance C to the value of the supply voltage UIN stored in the previous half-cycle. At the times when the input capacitance C is charged or discharged, the course of the voltage UC is shown in dashed lines, because the exact course is not relevant.
  • FIG. 10 a shows the case that the input capacitance C is discharged before the end of the phase gating
  • FIG. 10 b shows the case that the input capacitance C is charged in the dimmer before the switching element is switched on. How this happens is described below. In both cases, the difference between the voltage UC across the input capacitance C and the instantaneous value of the supply voltage UIN at the end of the phase control becomes small or almost disappears.
  • FIG. 11 shows, for the further features of the exemplary embodiments, the supply voltage UIN, the voltage UC across the input capacitance C, the line current IN and the voltage UL across the inductance of the dimmer.
  • the supply voltage UIN the voltage UC across the input capacitance C
  • the line current IN the voltage UL across the inductance of the dimmer.
  • the voltage UC across the input capacitance C is slightly smaller than the value of the instantaneous voltage UIN at the end of the phase control. It can be seen that the peak value of the mains current IN is significantly smaller in comparison with FIG. 9. The peak value of the voltage UL applied across the inductance is also smaller. The mains current IN oscillates much less. After the demagnetization T3 of the inductance in the dimmer, unlike in FIG. 9, a continuous mains current IN flows. The invention prevents the holding current of the switching element is fallen below in the dimmer.
  • FIG. 11 shows that the voltage UC across the input capacitance C is set to a value at the end of the phase gating which is below the corresponding instantaneous value of the supply voltage. This can ensure that in each case a current can flow to the load following the phase control.
  • Another way of predicting the instantaneous value of the supply voltage UIN is as follows: A further component, for example an inductance, can be connected in series with the input of the electronic ballast. At the end of the phase-cut, approximately one voltage proportional to the difference UIN-UC drops across this component, which voltage can then be used in a subsequent half-wave to adjust the voltage across the input capacitance.
  • FIG. 12 a describes a less expensive and more reliable circuit arrangement.
  • the task of the circuit is to measure the instantaneous value of the voltage UIN following the phase control. Furthermore, the circuit should address a control device of the boost converter for the described charge of the input capacitance C.
  • the circuit includes a monoflop MF, which is activated via a signal input A at the end of the phase control.
  • a monoflop MF At one output B of the monoflop MF is one of two states. One of them indicates that the monoflop MF is set, the other state takes the monoflop MF in the remaining time.
  • the output B of the monoflop MF is applied to a control input C of a switch AS.
  • the switch AS forwards a signal AVIN from a second input D to an output E when it is activated via the control input C.
  • the signal AVIN is proportional to the supply voltage UIN of the load.
  • a diode DS and a capacitor CS for peak detection is connected to the output E of the switch AS.
  • a resistor RS is connected in parallel with the capacitor CS. Over this, the capacitor CS can be discharged slowly as the peak values to be detected become smaller.
  • the discharge time of the capacitor CS is determined only by the size of the capacitance CS and the resistance RS. The speaking time scale is chosen so that it is appropriate to the change of the phase control by an operator.
  • the voltage across the capacitor CS is supplied to a first input COM2 of a comparator COM.
  • a second input COM1 of the comparator COM is supplied with a signal UC proportional to the voltage UC.
  • An output COMA of the comparator assumes a first state when the signal AVC at input COM1 is smaller than the signal at the other input COM2, and a second state when the signal at COM1 is greater than the signal at COM2.
  • the output COMA of the comparator COM can be connected, for example, to the control device of the boost converter.
  • the length of the time window in which the monoflop MF is set is very small compared to the period of the supply voltage UIN. In the longest case, the monoflop MF remains set during the entire magnetization of the inductance in the dimmer (in the interval T1).
  • FIG. 12b shows how the length of the time window can also be predetermined by means of a differentiator comprising a capacitor CT and a resistor RT.
  • the differentiator is addressed via a signal input at the end of the phase control.
  • a voltage jump occurs across the resistor RT, which decays exponentially.
  • the time constant of the exponential decay is the product of the size of the resistor RT and the capacitance CT.
  • the duration of the decay of the voltage jump across the resistor RT specifies the time window in which the switch AS remains switched on.
  • a suitable time window for the storage of a prognosis value for the supply voltage UIN can also be detected or predetermined by means of one of the circuit arrangements from FIGS. 4, 6 or 7.
  • the detected time of the end of the magnetization T1 of the inductance in the dimmer corresponds to the zero crossing of the second derivative of Voltage UC over the input capacitance C. This time is indicated by the signal outputs STA1 and STA2 or TIM1A and TIM2A and determines the end of the time window.
  • the peak voltage UC over the input capacitance C can be stored up to this time. Since the supply voltage has scarcely changed since the end of the phase control, the voltage UC across the input capacitance C corresponds at this time to the instantaneous value of the supply voltage UIN at the end of the phase control.
  • the circuit arrangements from FIGS. 12 a and 12 b can be well installed in the step-up converter described in EP 1 465 330 A2.
  • This has a control circuit BCC, which can be controlled among other things by the circuit arrangement of Figure 12 a and b. Further measures for charging or discharging the input capacitance C can be concretely described for this boost converter.
  • the time of switching on of the switching element in the dimmer can be detected in the boost converter from EP 1 465 330 A2 by the incipient current flow through, for example, the storage inductor LH (L1 in EP 1 465 330 A2) of the boost converter.
  • This incipient current flow triggers the monoflop MF via the input A.
  • the monoflop MF switches on at the end of the phase control to the end of a predetermined time interval (the time window) via the input C, the switch AS. While the switch AS is turned on, the capacitance CS via the diode DS detects the voltage applied to the input AVIN peak voltage.
  • the boost converter from EP 1 465 330 A2 can be activated as long as the voltage UC across the input capacitance C is greater than the stored value. Characterized the input capacitance C is discharged to a value which is slightly below the value of the supply voltage UIN at the end of the phase control. Specifically, the signal line COMA with an element of the control circuit BCC of the high linked. In EP 1 465 330 A2 in its figure 5a is a flip-flop FF2 described, which can be set by means of the output COMA of the comparator COM, so that the boost converter is activated.
  • the input capacitance C can also be discharged by a switching element connected in parallel, for example a series connection of a transistor and a resistor. This is controlled via the signal line COMA so that it turns on and the input capacitance C discharges.
  • Figure 13 shows a variation of the boost converter circuit of Figure 1; There is an additional resistor RH connected in parallel to the diode DH.
  • the diode DH can be bridged with a resistor RH.
  • the input capacitance C can be charged via the intermediate circuit capacitor before the end of the phase control.
  • control is required. If you do not want to add such a special, so the input capacitance C can be charged by the DC link capacitor so strong that the voltage UC on the input capacitance C is too high. Thereafter, the boost converter can be activated to discharge the input capacitance C to the desired value.
  • boost converters which have a plurality of diodes connected between the supply potential of the intermediate circuit capacitor CH and the supply potential of the input capacitance C; here one or more diodes can be bridged.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Circuit Arrangements For Discharge Lamps (AREA)

Abstract

L'invention concerne un ballast électronique présentant un condensateur d'entrée (C) et comprenant un hacheur élévateur (LH, DH, SH, CH) pour le fonctionnement d'une lampe à décharge (CFL) sur un gradateur à commande de phase (DIM) présentant une inductance intégrée ou parasite (L). Selon l'invention, les surélévations de tension suivant le découpage en début de phase sont réduites par réglage du courant au moyen du hacheur élévateur (LH, DH, SH, CH).
EP06722663A 2005-04-22 2006-03-22 Ballast electronique a reduction des oscillations de courant reactif Expired - Fee Related EP1872634B1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
DE102005018795A DE102005018795A1 (de) 2005-04-22 2005-04-22 Elektronisches Vorschaltgerät mit Blindstromschwingungsreduzierung
PCT/DE2006/000510 WO2006111123A1 (fr) 2005-04-22 2006-03-22 Ballast electronique a reduction des oscillations de courant reactif

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EP1872634A1 true EP1872634A1 (fr) 2008-01-02
EP1872634B1 EP1872634B1 (fr) 2009-06-24

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US (1) US7777423B2 (fr)
EP (1) EP1872634B1 (fr)
CN (1) CN101204121B (fr)
CA (1) CA2605217A1 (fr)
DE (2) DE102005018795A1 (fr)
WO (1) WO2006111123A1 (fr)

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WO2008071232A1 (fr) * 2006-12-13 2008-06-19 Osram Gesellschaft mit beschränkter Haftung Installation de circuit pour la commande de lampes à décharge et procédé de commande de lampes à décharge
WO2010133990A1 (fr) * 2009-05-20 2010-11-25 Koninklijke Philips Electronics N.V. Circuit d'alimentation électrique pour alimenter une lampe en énergie
US8264165B2 (en) * 2009-06-30 2012-09-11 Linear Technology Corporation Method and system for dimming an offline LED driver
DE102009035371B4 (de) * 2009-07-30 2017-10-26 Osram Gmbh Elektronisches Vorschaltgerät zum Betreiben mindestens einer Entladungslampe
US8466631B1 (en) * 2010-05-24 2013-06-18 Cooper Technologies Company Lamp driver with triac dimmer compensation
US20120229052A1 (en) * 2011-03-08 2012-09-13 Ching-Nan Yang Automatic lighting system
US8847498B2 (en) 2012-05-16 2014-09-30 Technical Consumer Products, Inc. Resonant damping circuit for triac dimmable
JP6056227B2 (ja) * 2012-07-10 2017-01-11 富士ゼロックス株式会社 画像形成装置およびバイアス電源装置
EP2701461A1 (fr) 2012-08-24 2014-02-26 Dialog Semiconductor GmbH Éclaircissement pilote à l'état solide d'un commutateur de puissance de commande mixte
US9112649B2 (en) * 2012-10-11 2015-08-18 Qualcomm Incorporated Method and apparatus for predicting signal characteristics for a nonlinear power amplifier
DE102013216877A1 (de) * 2013-08-23 2015-02-26 Osram Gmbh Getakteter elektronischer Energiewandler
CN107347222B (zh) * 2016-05-04 2019-02-12 台达电子企业管理（上海）有限公司 调光驱动电路及其控制方法

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US5656891A (en) * 1994-10-13 1997-08-12 Tridonic Bauelemente Gmbh Gas discharge lamp ballast with heating control circuit and method of operating same
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CN1653863A (zh) * 2002-09-12 2005-08-10 松下电器产业株式会社 无电极放电灯点灯装置、灯泡形无电极荧光灯及放电灯点灯装置
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Publication number Publication date
DE102005018795A1 (de) 2006-10-26
US20090302773A1 (en) 2009-12-10
CA2605217A1 (fr) 2006-10-26
US7777423B2 (en) 2010-08-17
CN101204121B (zh) 2011-09-07
DE502006004065D1 (de) 2009-08-06
EP1872634B1 (fr) 2009-06-24
CN101204121A (zh) 2008-06-18
WO2006111123A1 (fr) 2006-10-26

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