CN115622421A - Control method of battery simulator based on multi-sampling technology - Google Patents
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Abstract
The invention provides a control method of a battery simulator based on a multi-sampling control technology, and belongs to the field of control delay of a current loop of a PWM (pulse-width modulation) rectification circuit. The control method comprises the steps of sampling voltage once and sampling current for multiple times in a switching period, and converting alternating current voltage and alternating current obtained by sampling by using CLARK-PARK to obtain a voltage component of a dq axis and a current component of the dq axis; reducing errors of a voltage component of a dq axis and a current component of the dq axis by using a PI controller to obtain a dq component of a phase voltage reference value; and finally, obtaining a modulation wave signal through CLARK-PARK inverse transformation. The invention aims at the dynamic characteristic in the high-order battery equivalent circuit and reduces the control delay of the current loop, thereby improving the rapidity of the dynamic response of the current and realizing more accurate tracking and simulation of the high-order dynamic characteristic of the battery equivalent circuit.
Description
Technical Field
The invention relates to the field of high-order dynamic characteristic simulation of a battery equivalent circuit, in particular to a control method of a battery simulator based on a multi-sampling technology.
Background
With the rapid development of new energy technology and the rapid transformation of clean energy, the external working characteristics of an actual battery are simulated through a battery simulator, the development of new energy and the improvement of energy storage technology are facilitated, time and capital cost are saved, and the safety performance of experiments is improved.
However, in the battery simulator, the external characteristics of the battery are generally simulated by directly utilizing the internal resistance model of the equivalent circuit of the battery, the real-time change of the circuit current cannot be tracked in time, and the high-order dynamic characteristics of the equivalent circuit model of the battery are highlighted; meanwhile, the delay of a current control loop is introduced into the control method of the battery simulator, so that the rapidity of current response is reduced, and the tracking and simulation of the high-order dynamic characteristics of the circuit cannot be accurately realized.
Currently, there are a plurality of academic papers for analyzing and reporting the control method of the battery simulator, for example:
1. the article entitled "sliding mode control-based battery simulator simulation and research", measurement and control technology and instrument, and 2022, volume 48, page 47-50, 5, proposes that sliding mode control is compared with traditional PI control, verifies that the regulation time of sliding mode control is superior to that of PI control when load fluctuates, and has better robustness and dynamic performance.
2. The invention provides a method for realizing the function of a battery simulator on a three-level PWM rectifier, which is provided with good dynamic regulation speed and smaller output voltage error, and is entitled ' universal energy storage battery simulation research based on the three-level PWM rectifier ', an electrical and energy efficiency management technology ', 2018, pages 21-27.
However, the following problems exist for the output delay and stability of the current loop of the battery simulator:
1) The sliding mode control method has the advantages that the uncertainty of the nonlinear system is overcome, the control effect on the nonlinear system is good, the algorithm is simple, the response speed is high, and the realization is easy. However, when the state trajectory reaches the sliding mode surface, it is difficult to slide strictly along the sliding mode surface toward the equilibrium point, but the state trajectory approaches the equilibrium point while passing back and forth on both sides thereof, thereby generating buffeting and causing unstable output.
2) Although the dynamic voltage regulation range of the battery simulator is improved, the current loop control still has larger control delay, the current loop bandwidth is limited to a certain extent, and the dynamic response of a high-order equivalent circuit model cannot be improved well.
Disclosure of Invention
The invention aims to solve the problems of dynamic characteristic simulation and current control loop time delay of an equivalent circuit of a battery simulator, and provides a control method of the battery simulator based on a multi-sampling technology, which can improve the sampling frequency of current, reduce the tracking time delay of the equivalent circuit of the battery, reduce the time delay of the current control loop, and improve the rapidity of current response under the condition of not changing the switching frequency so as to realize more accurate tracking and simulation of high-order dynamic characteristics of the equivalent circuit of the battery.
The object of the invention is thus achieved. The invention provides a control method of a battery simulator based on a multi-sampling technology, wherein a circuit of the battery simulator is a PWM (pulse-width modulation) rectifying circuit with energy flowing in two directions, and specifically, a topological structure of the battery simulator comprises a three-phase alternating-current power supply Q, a three-phase alternating-current filter inductor L, T type three-phase three-level rectifying circuit and an upper direct-current side bus capacitor C bus1 Lower DC side bus capacitor C bus2 And DC side load R load (ii) a The upper direct current side bus capacitor C bus1 And a lower DC side bus capacitor C bus2 A capacitor branch circuit is connected in series and then is loaded with a direct current side load R load The direct current output end of the T-shaped three-phase three-level rectifying circuit is connected with the capacitor branch in parallel, the alternating current input end of the T-shaped three-phase three-level rectifying circuit is connected with a three-phase alternating current filter inductor L, and the other end of the three-phase alternating current filter inductor L is connected with a three-phase alternating current power supply Q; load the DC side with R load The voltage at is recorded as the DC side voltage U dc And the current is recorded as direct currentSide current I dc ；
The control method comprises the following steps:
Given switching frequency f c ，
Wherein T is c For a switching period T c ；
At a switching frequency f c Lower to DC side voltage U dc Sampling three-phase input voltage of the T-shaped three-phase three-level rectifying circuit for one time, and recording the sampling result as direct-current side voltage U dc And three-phase input voltage U a ，U b ，U c Inputting three-phase voltage U a ，U b ，U c Obtaining a feedforward voltage d-axis component E through CLARK-PARK conversion d And a feedforward voltage q-axis component E q The CLARK-PARK transformation formula is as follows:
wherein, theta u Is the phase angle integral value of the current time voltage sample, theta u ＝θ u * + ω and ω are increment values of phase angle at any time of voltage sampling, ω =2 π f, f is fundamental frequency, θ u * The phase angle integral value of the voltage sampling at the last moment;
In synchronism with step 1, during a switching period T c Internal to direct side current I dc Three-phase input current I of T-type three-phase three-level rectification circuit a ，I b ，I c Sampling for N times, and recording the obtained N direct current side currents and N three-phase input currents as kth direct current side current I dck And kth time three-phase input current I ak ，I bk ，I ck N is in a switching period T c The total sampling times of the current in the current sampling device are N > 2,N which is a positive integer; k represents adoptSample number, k =1,2.. N;
recording the current sampling frequency as f s ，f s ＝Nf c (ii) a Recording the current sampling period as T s ，
For the kth three-phase input current I ak ，I bk ，I ck CLARK-PARK conversion is carried out to obtain the kth three-phase input current d-axis component I dk And the kth three-phase input current q-axis component I qk N, whose CLARK-PARK transformation formula is:
wherein, theta i The integral value of the phase angle, theta, sampled for the current at the present moment i ＝θ i * +∫ω，θ i * The phase angle integral value of current sampling at the last moment;
Establishing a second-order RC battery equivalent circuit model, and solving the terminal voltage E of the lithium battery:
E＝E m -I dcN G z (s)
wherein G is z (s) is the impedance transfer function in the battery model, s is the Laplace operator;
in the formula, K pu Is the voltage loop proportionality coefficient, K u Is a voltage loop integral coefficient;
order the q-axis component I of the command current q_ref ＝0；
e dik ＝I d_ref -I dk
e qik ＝I q_ref -I qk
d-axis error signal e of the k-th current loop dik And a k-th current loop q-axis error signal e qik As the input signal of the current loop PI controller, the output is the d-axis component reference value I of the k-th current loop PI controller abc_ref_dk And a k-th current loop PI controller q-axis component reference value I abc_ref_qk Transfer function G of said current loop PI controller pi The expression of(s) is:
in the formula, K pi Is the current loop proportionality coefficient, K i Is the current loop integral coefficient;
respectively recording a d-axis component of a k-th sub-phase voltage reference value and a q-axis component of the k-th sub-phase voltage reference value as U abc_ref_dk And U abc_ref_qk K =1,2.. N, calculated as:
U abc_ref_dk ＝I abc_ref_dk +E d
U abc_ref_qk ＝I abc_ref_qk +E q
recording the A phase signal of the k modulation wave as U ak_pu_ref The B phase signal of the k-th modulation wave is U bk_pu_ref The k-th modulation wave C phase signal is U ck_pu_ref K =1,2.. N, which are calculated as:
therefore, the control method of the battery simulator based on the multi-sampling technology is constructed.
Compared with the control method of the existing battery simulator, the invention has the following beneficial effects:
1. the invention adopts a method of sampling for many times to realize the rapid tracking and simulation of the dynamic characteristics of the equivalent circuit of the battery;
2. according to the invention, the current is sampled for multiple times, so that the current loop delay is reduced, the loop current is quickly tracked, and the dynamic response speed of a current loop is increased;
3. the multi-sampling technology can be used for various PWM (pulse-width modulation) rectifying circuits and inverter circuits and has universality.
Drawings
FIG. 1 is a topology diagram of a battery simulator in accordance with the present invention.
Fig. 2 is a control structure diagram of the battery simulator based on the multi-sampling technology according to the present invention.
FIG. 3 shows the sampling mode of the battery simulator based on the conventional battery simulator and the sampling mode of the battery simulator of the present invention at K pi And =20 time-constant current waveform.
FIG. 4 shows a sampling mode of a conventional battery simulator and a sampling mode of the battery simulator of the present invention at K pi And =60 time steady-state current waveform.
Fig. 5 is a dynamic waveform of an output current in a sampling mode based on a conventional battery simulator and a sampling mode based on a battery simulator according to the present invention.
Fig. 6 shows the delay characteristics of the current loop in the sampling mode of the conventional battery simulator and the sampling mode of the battery simulator according to the present invention.
Detailed Description
The invention is further described below with reference to the accompanying drawings and examples.
FIG. 1 is a topological diagram of the battery simulator of the present invention, as can be seen from FIG. 1, the circuit of the battery simulator is a PWM rectification circuit with bidirectional energy flow, specifically, the topological structure comprises a three-phase AC power supply Q, a three-phase AC filter inductor L, T type three-phase three-level rectification circuit, and an upper DC side bus capacitor C bus1 Lower DC side bus capacitor C bus2 And a DC side load R load . The upper direct current side bus capacitor C busl And a lower DC side bus capacitor C bus2 A capacitor branch circuit is connected in series and then is loaded with a direct current side load R load And the direct current output end of the T-shaped three-phase three-level rectifying circuit is connected with the capacitor branch in parallel, the alternating current input end of the T-shaped three-phase three-level rectifying circuit is connected with a three-phase alternating current filter inductor L, and the other end of the three-phase alternating current filter inductor L is connected with a three-phase alternating current power supply Q. Load the DC side with R load The voltage at is recorded as the DC side voltage U dc The current is recorded as the direct current I dc 。
In FIG. 1, Q a ，Q b ，Q c Three components, L, of a three-phase ac power supply Q a ，L b ，L c Three components of a three-phase AC filter inductor L, and T is a T-shaped three-phase three-level rectification circuit。
Fig. 2 is a control structure diagram of the battery simulator based on the multi-sampling technique according to the present invention, and as can be seen from fig. 2, the control method according to the present invention includes the following steps:
Given switching frequency f c ，
Wherein T is c For a switching period T c ；
At a switching frequency f c Lower DC side voltage U dc Sampling three-phase input voltage of the T-shaped three-phase three-level rectifying circuit for one time, and recording the sampling result as direct-current side voltage U dc And three-phase input voltage U a ，U b ，U c Inputting three-phase voltage U a ，U b ，U c Obtaining a feedforward voltage d-axis component E through CLARK-PARK conversion d And a feedforward voltage q-axis component E q The CLARK-PARK transformation formula is as follows:
wherein, theta u Is the integral value of the phase angle of the voltage sample at the present moment, theta u ＝θ u * ω +ω, ω is the incremental value of the phase angle at any time when the voltage is sampled, ω =2 π f, f is the fundamental frequency, θ u * The phase angle integral value of the voltage sampling at the last moment is obtained.
In the present embodiment, f =50Hz, θ u * Is 0,f c ＝20KHz，
In synchronism with step 1, during a switching period T c Internal to direct side current I dc Three-phase input power of T-type three-phase three-level rectification circuitStream I a ，I b ，I c Sampling for N times, and recording the obtained N direct current side currents and N three-phase input currents as a kth direct current side current I dck And kth time three-phase input current I ak ，I bk ，I ck N is in a switching period T c The total sampling times of the current in the current sampling device are N > 2,N which is a positive integer; k denotes the number of sampling times, k =1,2,. N;
recording the current sampling frequency as f s ，f s ＝Nf c (ii) a Recording the current sampling period as T s ，
For the kth three-phase input current I ak ，I bk ，I ck CLARK-PARK conversion is carried out to obtain the kth three-phase input current d-axis component I dk And the kth three-phase input current q-axis component I qk N, whose CLARK-PARK transformation formula is:
wherein, theta i The integral value of the phase angle, theta, sampled for the current at the present moment i ＝θ i * +∫ω，θ i * The phase angle integral value of the current sampling at the last moment.
In this embodiment, N =4,f s ＝80KHz，
θ i * Is 0.
Establishing a second-order RC battery equivalent circuit model, and solving the terminal voltage E of the lithium battery:
E＝E m -I dcN G z (s)
wherein, G z (s) is the impedance transfer function in the cell model, and s is the Laplace operator.
In this embodiment, E m ＝900V，R 1 ＝1.8Ω，R p1 ＝0.9Ω，C p1 ＝10F，R p2 ＝0.6Ω，C p2 ＝3.2F。
in the formula, K pu Is the voltage loop scaling factor, K u Is a voltage loop integral coefficient;
make the q-axis component I of the command current q_ref ＝0。
In this embodiment, K pu ＝0.3，K u ＝15。
e dik ＝I d_ref -I dk
e qik ＝I q_ref -I qk
d-axis error signal e of the kth current loop dik And a k-th current loop q-axis error signal e qik As the input signal of the current loop PI controller, the output is the d-axis component reference value I of the kth current loop PI controller abc_ref_dk And a k-th current loop PI controller q-axis component reference value I abc_ref_qk Transfer function G of said current loop PI controller pi The expression of(s) is:
in the formula, K pi Is the current loop proportionality coefficient, K i Is the current loop integral coefficient;
respectively recording a d-axis component of a k-th sub-phase voltage reference value and a q-axis component of the k-th sub-phase voltage reference value as U abc_ref_dk And U abc_ref_qk K =1,2.. N, calculated as:
U abc_ref_dk ＝I abc_ref_dk +E d
U abc_ref_qk ＝I abc_ref_qk +E q
in this embodiment, K pi ＝60，K i ＝0.005。
recording the phase A signal of the k-th modulation wave as U ak_pu_ref The B phase signal of the k-th modulation wave is U bk_pu_ref The k-th modulation wave C-phase signal is U ck_pu_ref K =1,2.. N, which are calculated as:
so far, the control method of the battery simulator based on the multi-sampling technology is constructed.
In order to prove the technical effect of the invention, the control method of the invention is simulated. In the simulation, the current loop proportionality coefficients K are respectively set pi ＝20，K pi ＝60。
FIG. 3 shows the sampling mode of the battery simulator based on the conventional battery simulator and the sampling mode of the battery simulator of the present invention at K pi And =20 time-constant current waveform. As indicated in FIG. 3, the abscissa is time in units of s and the ordinate is the DC side current I dc The unit is A. As can be seen from fig. 3, under the condition of low bandwidth, the time for recovering to the steady state by single sampling and four times of sampling is approximately consistent, but the four times of sampling does not overshoot, and the regulation effect of the current loop is relatively stable before the steady state is reached. Namely, the stable time of the traditional battery simulator sampling mode and the battery simulator multi-sampling mode of the invention is approximately equal under low bandwidth, but the battery simulator multi-sampling mode of the invention has no overshoot, and the dynamic regulation is more stable.
FIG. 4 shows a sampling mode of a conventional battery simulator and a sampling mode of the battery simulator of the present invention at K pi A steady-state current waveform of =60, as indicated in fig. 4, with time on the abscissa and s on the ordinate, and direct-side current I on the ordinate dc The unit is A. As can be seen from fig. 4, fig. 4 is a waveform in which the single sample and the four samples return to a steady state under the condition of a higher bandwidth, and as can be seen from fig. 4, the four samples reach a steady state before the single sample, and there is no overshoot, the output is more stable, and it also indicates that the four samples have a higher bandwidth. As can be seen from fig. 4, the multi-sampling control technique of the present invention has a higher bandwidth, improves the dynamic response of the current loop, and can reach a stable state more quickly.
FIG. 5 is a dynamic waveform of output current in a multi-sampling mode of a battery simulator based on a conventional battery simulator and the battery simulator of the present invention, respectively, in which a user sits on a bedMarked as time in units of s, ordinate DC side current I dc The unit is A, the graph shows that the current value suddenly changes at 5ms, single sampling and multi-sampling restore to a steady waveform when the current value suddenly changes from 20A to 30A, and the dynamic response of four sampling is faster than that of single sampling, and the regulation is more stable without high overshoot. Therefore, the sampling mode of the invention can improve the dynamic response of the current loop, can quickly track when the output current of the battery simulator changes, and reduces the influence of the control delay of the current loop. As shown in fig. 5, the multi-sampling control technique of the present invention can improve the dynamic response speed of the current loop and simulate the dynamic characteristics of the battery high-order equivalent circuit.
Fig. 6 is a time delay characteristic of a current loop in a sampling mode based on a conventional battery simulator and a sampling mode of a battery simulator according to the present invention, where the abscissa is adjustment time in units of s, the ordinate is an error of a d-axis output by the current loop, and the unit is a. As can be seen from fig. 6, the multi-sampling control technique of the present invention can effectively reduce the control delay of the current loop and improve the rapidity of the dynamic response of the current loop.
Claims (1)
1. A control method of a battery simulator based on a multi-sampling technology is characterized in that a circuit of the battery simulator is a PWM (pulse-width modulation) rectification circuit with energy flowing in two directions, and specifically, a topological structure of the battery simulator comprises a three-phase alternating-current power supply Q, a three-phase alternating-current filter inductor L, T type three-phase three-level rectification circuit and an upper direct-current side bus capacitor C bus1 Lower DC side bus capacitor C bus2 And a DC side load R load (ii) a The upper direct current side bus capacitor C bus1 And a lower DC side bus capacitor C bus2 A capacitor branch circuit is connected in series and then is loaded with a direct current side load R load The direct current output end of the T-shaped three-phase three-level rectifying circuit is connected with the capacitor branch in parallel, the alternating current input end of the T-shaped three-phase three-level rectifying circuit is connected with a three-phase alternating current filter inductor L, and the other end of the three-phase alternating current filter inductor L is connected with a three-phase alternating current power supply Q(ii) a Loading a DC side with a load R load The voltage at is recorded as the DC side voltage U dc The current is recorded as the direct current I dc ；
The control method is characterized by comprising the following steps:
step 1, single sampling and coordinate transformation of voltage
Given switching frequency f c ，
Wherein T is c For a switching period T c ；
At a switching frequency f c Lower to DC side voltage U dc Sampling three-phase input voltage of the T-shaped three-phase three-level rectifying circuit for one time, and recording the sampling result as direct-current side voltage U dc And three-phase input voltage U a ，U b ，U c Inputting three-phase voltage U a ，U b ，U c Obtaining a feedforward voltage d-axis component E through CLARK-PARK conversion d And a feedforward voltage q-axis component E q The CLARK-PARK transformation formula is as follows:
wherein, theta u Is the integral value of the phase angle of the voltage sample at the present moment, theta u ＝θ u * ω +ω, ω is the incremental value of the phase angle at any time when the voltage is sampled, ω =2 π f, f is the fundamental frequency, θ u * The phase angle integral value of the voltage sampling at the last moment;
step 2, multiple sampling and coordinate transformation of current
In synchronism with step 1, during a switching period T c Internal to direct current side current I dc Three-phase input current I of T-type three-phase three-level rectification circuit a ，I b ，I c Sampling for N times, and recording the obtained N direct current side currents and N three-phase input currents as a kth direct current side current I dck And the kth three-phase input current I ak ，I bk ，I ck N is in a switching period T c The total sampling times of the current in the current sampling device are N > 2,N which is a positive integer; k denotes the number of sampling times, k =1,2,. N;
recording the current sampling frequency as f s ，f s ＝Nf c (ii) a Recording the current sampling period as T s ，
For the kth three-phase input current I ak ，I bk ，I ck CLARK-PARK conversion is carried out to obtain the kth three-phase input current d-axis component I dk And the kth three-phase input current q-axis component I qk N, whose CLARK-PARK transformation formula is:
wherein, theta i The integral value of the phase angle, theta, sampled for the current at the present moment i ＝θ i * +∫ω，θ i * The phase angle integral value of the current sampling at the last moment;
step 3, acquiring experimental data of lithium battery charging and discharging from a known database as a database of the battery simulator, wherein the database comprises open-circuit voltage E of the lithium battery m Ohmic internal resistance R of lithium battery 1 Electrochemical polarization internal resistance R of lithium battery p1 And electrochemical polarization capacitance C of lithium battery p1 Internal resistance of lithium battery p2 Concentration difference capacitor C of lithium battery p2 ；
Establishing a second-order RC battery equivalent circuit model, and solving the terminal voltage E of the lithium battery:
E＝E m -I dcN G z (s)
wherein G is z (s) is the impedance transfer function in the battery model, s is the Laplace operator;
step 4, recording the d-axis error signal as e du ，e du ＝E-U dc The d-axis error signal e du The output signal of the voltage loop PI controller is an instruction current d-axis component I d_ref A transfer function G of the voltage loop PI controller pu The expression of(s) is:
in the formula, K pu Is the voltage loop proportionality coefficient, K u Is a voltage loop integral coefficient;
order the q-axis component I of the command current q_ref ＝0；
Step 5, recording the d-axis error signal of the k-th current loop as e dik And the q-axis error signal of the kth current loop is recorded as e qik K =1,2.. N, calculated as:
e dik ＝I d_ref -I dk
e qik ＝I q_ref -I qk
d-axis error signal e of the kth current loop dik And a k-th current loop q-axis error signal e qik As the input signal of the current loop PI controller, the output is the d-axis component reference value I of the k-th current loop PI controller abc_ref_dk And a k-th current loop PI controller q-axis component reference value I abc_ref_qk Transfer function G of said current loop PI controller pi The expression of(s) is:
in the formula, K pi Is the current loop proportionality coefficient, K i Is the current loop integral coefficient;
d-axis component of reference value of kth phase voltage is addedThe q-axis components of the reference value of the kth phase voltage are respectively marked as U abc_ref_dk And U abc_ref_k K =1,2.. N, calculated as:
U abc_ref_dk ＝I abc_ref_dk +E d
U abc_ref_qk ＝I abc_ref_qk +E q
step 6, d-axis component U of reference value of kth phase voltage obtained in step 5 abc_ref_dk And the kth phase voltage reference q-axis component U abc_ref_qk Performing coordinate inverse transformation to obtain the kth A-phase voltage reference value U ak_ref Reference value U of phase-B voltage at kth time bk_ref And the kth C-phase voltage reference value U ck_ref N, whose inverse CLARK-PARK transform has the formula:
recording the A phase signal of the k modulation wave as U ak_pu_ref The B phase signal of the k-th modulation wave is U bk_pu_ref The k-th modulation wave C phase signal is U ck_pu_ref K =1,2.. N, which are calculated as:
so far, the control method of the battery simulator based on the multi-sampling technology is constructed.
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| CN116992693A (en) * | 2023-09-01 | 2023-11-03 | 湖南恩智测控技术有限公司 | Construction method and device of battery simulator, electronic equipment and storage medium |
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2022
- 2022-09-26 CN CN202211179017.9A patent/CN115622421A/en active Pending
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN116992693A (en) * | 2023-09-01 | 2023-11-03 | 湖南恩智测控技术有限公司 | Construction method and device of battery simulator, electronic equipment and storage medium |
| CN116992693B (en) * | 2023-09-01 | 2024-04-05 | 湖南恩智测控技术有限公司 | Construction method and device of battery simulator, electronic equipment and storage medium |
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