CN115296441A - Multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation - Google Patents

Multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation Download PDF

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CN115296441A
CN115296441A CN202210972894.5A CN202210972894A CN115296441A CN 115296441 A CN115296441 A CN 115296441A CN 202210972894 A CN202210972894 A CN 202210972894A CN 115296441 A CN115296441 A CN 115296441A
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frequency
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mcr
wpt
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CN115296441B (en
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沈艳霞
夏***
瞿俞楠
赵芝璞
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Jiangnan University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/40—Circuit arrangements or systems for wireless supply or distribution of electric power using two or more transmitting or receiving devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration

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Abstract

The invention relates to a multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation. The method comprises the steps of constructing a required multi-frequency multi-load MCR-WPT system, determining zero-phase working frequency, system basic parameters and target working frequency of the multi-frequency multi-load MCR-WPT system, and constructing a fitness function according to the determined zero-phase working frequency, system basic parameters and target working frequency; based on the constructed fitness function, optimizing and determining compensation network parameters of an LCL compensation topological network in a transmitting loop so as to configure the multi-frequency multi-load MCR-WPT system to be in a ZCS state when the transmitting loop based on the determined compensation network parameters works. The multi-frequency multi-load MCR-WPT designed by the invention can realize multi-frequency multi-channel power independent transmission on the basis of a single inverter, works in a ZCS state, and can effectively reduce cost and loss.

Description

Multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation
Technical Field
The invention relates to a design method, in particular to a multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation.
Background
Compared with the traditional wired power transmission mode, the wireless power transmission technology has the advantages of convenience and safety, and people hope to get rid of the dependence on the wired power transmission mode. Therefore, research on wireless power transmission technology is increasingly being conducted and is increasingly being applied to practical applications in modern society, such as the fields of medical equipment, electronic equipment, vehicles, and the like.
The current WPT (Wireless Power Transmission) technology is increasingly popularized, and the device requirements of the user side also present complex and diverse characteristics such as different frequencies, power and the like. The tightly-coupled induction technology based on the wireless charging alliance (WPC) for providing the Qi charging standard and the loosely-coupled magnetic resonance technology based on the AirFuel organization are gradually applied to a multi-receiver WPT system, and can meet the power supply requirements of load equipment with different charging standards.
According to whether the respective resonance frequencies of the receivers are the same or not, the multi-receiver WPT system can be divided into a single-frequency multi-reception WPT system and a multi-frequency multi-reception WPT system. A multi-frequency multi-receiving WPT system has gradually become the focus of the current multi-load WPT system research, wherein the system has receivers with different resonant frequencies; meanwhile, the transmitter outputs the frequency mixture with the same resonance frequency as the receiver, so that the effect of establishing independent power transmission channels between each receiver and the power supply at the transmitting side is realized.
Through the multi-frequency multi-receiving WPT system, power distribution on different loads can be achieved on the transmitting side, namely, the power of the corresponding frequency of the transmitting end is adjusted, and the cross coupling influence among receiving coils is reduced. Most of the existing multi-frequency multi-receiving MCR-WPT (Magnetically Coupled Resonant Power Transfer) system designs adopt a multi-inverter to be Coupled to a transmitting side, so that the volume and cost are high.
Therefore, the design of the multi-frequency multi-load MCR-WPT with small volume, low cost and no increase of loss is a technical problem which needs to be solved urgently at present.
Disclosure of Invention
The designed multi-frequency multi-load MCR-WPT can realize independent power transmission of multiple frequencies and multiple channels on the basis of a single inverter, works in a ZCS state, and can effectively reduce cost and loss.
According to the technical scheme provided by the invention, the multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation,
constructing a required multi-frequency multi-load MCR-WPT system, wherein the multi-frequency multi-load MCR-WPT system comprises a transmitting side and a receiving side, the transmitting side transmits mixed signals based on MFMA superposition modulation, the receiving side is used for receiving the mixed signals transmitted by the transmitting side, the transmitting side comprises a single inverter and a transmitting loop which is adaptively connected with the single inverter, and the transmitting loop comprises an LCL compensation topological network; the receiving side comprises n paths of mutually independent receivers, and any receiver in the receiving side is coupled with a fundamental frequency signal matched with the resonance frequency of the receiver in a received mixed signal so as to provide direct current power for a load connected to the receiver;
determining zero-phase working frequency, system basic parameters and target working frequency of the multi-frequency multi-load MCR-WPT system to construct a fitness function according to the determined zero-phase working frequency, system basic parameters and target working frequency;
and optimizing and determining compensation network parameters of an LCL compensation topological network in the transmitting loop based on the constructed fitness function so as to configure the multi-frequency multi-load MCR-WPT system to be in a ZCS state when the transmitting loop based on the determined compensation network parameters works.
The single inverter is an inverter constructed based on a gallium nitride field effect transistor;
the transmitting loop comprises n-2 LC resonant loops and an LCL basic compensation network, the LC resonant loops in the transmitting loop are sequentially connected in an adaptive manner, and the basic compensation network comprises a transmitting coil L p ；
When n fundamental frequency signals to be modulated are emitted and output based on mixed signals subjected to MFMA superposition modulation, the resonant frequencies of n receivers correspond to the frequencies of the n fundamental frequency signals subjected to MFMA superposition modulation one by one.
The frequencies of the n fundamental frequency signals are completely different, and the amplitude of any fundamental frequency signal is independently adjustable.
Determining a circuit model matrix of the multi-frequency multi-load MCR-WPT system by adopting a fundamental wave analysis method, and obtaining a total reflection impedance frequency characteristic curve according to the determined circuit model matrix, wherein the zero-phase working frequency of the multi-frequency multi-load MCR-WPT system is obtained by using a graphical method for the obtained total reflection impedance frequency characteristic curve.
And any receiver comprises a coupling inductance coil and an uncontrollable rectifier, wherein two ends of the coupling inductance coil are respectively in adaptive connection with the uncontrollable rectifier through a receiver loop lead resistor and a receiving tuning capacitor so as to be in adaptive connection with a load through the uncontrollable rectifier to provide required direct current power for the load.
The determined basic parameters of the system include the desired resonance frequency f of the receiver ei (i =1, 2.. N), a transmitting coil L p Inductor and receiving coil L of si Inductance (i =1, 2.. Multidot.n), transmitting coil L p With respective coupling inductors L si Mutual inductance M psi (i =1, 2.. Multidot.n), receiver load R Li (i =1, 2.. N) and a receive tuning capacitance C si (i＝1,2,...,n)。
Constructing a fitness function according to the zero-phase working frequency, the basic system parameters and the target working frequency as follows:
Figure BDA0003797555280000021
wherein Fitness is the constructed Fitness function, f ei Indicating the desired resonance frequency, f, of the ith receiver si Is the ith actual resonance frequency of the system, i.e. the ith zero-phase working frequency of the system, I si For the load current of the ith receiver, I sj For the jth receiver load current, k si The derivative, k, of the angular frequency characteristic curve of the input impedance of the system at the ith actual resonance frequency point sj For the angular frequency characteristic curve of the input impedance of the system at the jth practicalDerivative at resonance frequency point, W 1 Optimizing weight coefficient, W, for resonant frequency 2 Optimizing weight coefficient and W for load current 3 And the weight coefficient of the derivative of the angular frequency characteristic curve of the input impedance of the system at the resonance point.
And for the constructed fitness function, optimizing and determining a compensation network parameter of an LCL compensation topological network in a transmitting loop based on a PSO (particle swarm optimization) method, wherein the process of determining the compensation network parameter comprises the following steps:
step 1, configuring a transmitting resonance inductor in a transmitting loop into inductive particles, configuring a transmitting resonance capacitor in the transmitting loop into capacitive particles, and determining the ith inductive particle L fi And the ith capacitive particle C fi Corresponding value range and motion speed range, wherein i =1, 2.. And n-1;
step 2, randomly initializing inductive particles L fi And a capacitor particle C fi ；
Step 3, according to the inductive particles L after random initialization fi And a capacitor particle C fi Determining the value of the fitness function, and updating the inductance particles L fi Capacitor particle C fi Corresponding individual optimal fitness and population optimal fitness;
step 4, updating the inductive particles L according to the above fi Capacitor particle C fi Updating the inductive particles L according to the individual best fitness and the group best fitness fi Capacitor particle C fi The corresponding speed and position, wherein,
Figure BDA0003797555280000031
in the formula, L mi Is an inductive particle L fi Individual optimum particle, Y Lmi Is an inductive particle L fi Of population-optimal particles of, L i Is an inductive particle L fi All particles of the contemporary population; c mi Is a capacitive particle C fi Individually optimized particles, Y Cmi Is a capacitive particle C fi Population-optimal particles, C i Is a capacitive particle C fi All particles of the contemporary population; beta is the inertial weight, lambda 1 And λ 2 Self-learning factors and group learning factors are respectively; rand is a random number generator;
step 5, iteration frequency control is carried out until the inductive particles L with the optimal fitness are found fi And capacitor particles C fi To determine the compensation network parameters of the LCL compensation topology network in the transmission loop.
In step 1, inductive particles L fi And a capacitor particle C fi The corresponding value range and the movement speed range are as follows:
Figure BDA0003797555280000032
Figure BDA0003797555280000033
wherein, vL fi Is the ith inductive particle L fi Velocity, vC fi Is the ith capacitive particle C fi The corresponding speed.
The base frequency signal based on MFMA superposition modulation is a sine wave signal;
based on the fact that the amplitude of the fundamental frequency signal in the MFMA superposition modulation is a normalized amplitude, for a plurality of receivers on the receiving side, the direct current power of the load adaptive to the receivers is in one-to-one correspondence with the amplitude of the fundamental frequency signal coupled by the receivers.
The invention has the advantages that: when a multi-frequency multi-load MCR-WPT system is constructed, a transmitting side transmits a mixed frequency signal based on MFMA superposition modulation, the transmitting side comprises a single inverter and a transmitting loop which is adaptively connected with the single inverter, and the transmitting loop comprises an LCL compensation topology network;
the MFMA superposition modulation method can solve the problems of high cost and high loss caused by the coupling of the existing multi-inverter to the transmitting side through a transformer; according to the number of receivers on the receiving side, the transmitting side adopts a proper LCL topology compensation network, so that the same number of system natural resonant frequencies can be brought to a multi-frequency multi-load MCR-WPT system, and independent power channels can be established for transmission; and optimizing a Fitness function Fitness constructed by the derivative of the angular frequency characteristic curve of the resonance frequency, the load current and the input impedance of the resonance point of the base pair system by adopting a PSO algorithm, so that the transmission frequency can be ensured to be close to the natural resonance frequency point of each receiver, and the received power of each load is controlled to keep balance.
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Fig. 1 is a schematic circuit diagram of a multi-frequency multi-load MCR-WPT system constructed according to the invention.
FIG. 2 is a schematic diagram of a specific implementation circuit for constructing a dual-frequency dual-load MCR-WPT system according to the invention.
FIG. 3 is a schematic diagram of the double-frequency double-load MCR-WPT system in FIG. 2, wherein an equivalent circuit model is established on the basis of a fundamental wave analysis method.
FIG. 4 is a graph of input impedance angle versus input frequency for the embodiment of FIG. 2.
FIG. 5 is a graph of input impedance versus input frequency for the embodiment of FIG. 3.
Fig. 6 is a schematic diagram of an implementation of the load voltage of the dual load adjusted by the input normalized voltage amplitude Ai in the embodiment of fig. 2.
Fig. 7 is a schematic diagram of another embodiment of the load voltage of the dual load of fig. 2 after the input-side normalized voltage amplitude Ai is adjusted.
Fig. 8 is a schematic diagram of a third implementation of the load voltage of the dual load of the embodiment of fig. 2 after the input-side normalized voltage amplitude Ai is adjusted.
Fig. 9 is a diagram of FFT analysis of an embodiment of the output voltage of the dual receiver adjusted by the input-side normalized voltage amplitude Ai in the embodiment of fig. 2.
Fig. 10 is an FFT analysis diagram of another implementation of the output voltage of the dual receiver after the input end normalized voltage amplitude Ai is adjusted in the embodiment of fig. 2.
Fig. 11 is an FFT analysis diagram of a third embodiment of the output voltage of the dual receiver after the input normalized voltage amplitude Ai is adjusted.
Detailed Description
The invention is further illustrated by the following specific figures and examples.
As shown in fig. 1: for the designed multi-frequency multi-load MCR-WPT, in order to realize independent multi-frequency multi-channel power transmission on the basis of a single inverter, work in a ZCS state and effectively reduce cost and loss, the invention discloses a multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation, which specifically comprises the following steps:
constructing a required multi-frequency multi-load MCR-WPT system, wherein the multi-frequency multi-load MCR-WPT system comprises a transmitting side and a receiving side, the transmitting side transmits mixed signals based on MFMA superposition modulation, the receiving side is used for receiving the mixed signals transmitted by the transmitting side, the transmitting side comprises a single inverter and a transmitting loop which is adaptively connected with the single inverter, and the transmitting loop comprises an LCL compensation topological network; the receiving side comprises n paths of mutually independent receivers, and any receiver in the receiving side is coupled with a fundamental frequency signal matched with the resonance frequency of the receiver in a received mixed signal so as to provide direct current power for a load connected to the receiver;
determining zero-phase working frequency, system basic parameters and target working frequency of the multi-frequency multi-load MCR-WPT system to construct a fitness function according to the determined zero-phase working frequency, system basic parameters and target working frequency;
based on the constructed fitness function, optimizing and determining compensation network parameters of an LCL compensation topological network in a transmitting loop so as to configure the multi-frequency multi-load MCR-WPT system to be in a ZCS state when the transmitting loop based on the determined compensation network parameters works.
Specifically, a Multi-Frequency Multi-load MCR-WPT system is constructed according to actual working scene requirements and the like, and similar to the existing MCR-WPT system, the constructed MCR-WPT system also comprises a transmitting side and a receiving side corresponding to the transmitting side, but when the MCR-WPT system is different from the existing MCR-WPT system, the transmitting side of the MCR-WPT system constructed by the present invention can transmit a mixed signal based on MFMA (Multi-Frequency Multi-Amplitude) superposition modulation, wherein the transmitting side comprises a single inverter and a transmitting loop adapted to the single inverter, and the single inverter is an inverter, that is, in the Multi-Frequency Multi-load MCR-WPT system constructed by the present invention, the transmitting side comprises an inverter and a transmitting loop adapted to the inverter. In order to solve the problem that no compensation network is arranged in a transmitting loop and loss is increased, in the embodiment of the invention, an LCL compensation topological network is arranged in the transmitting loop, namely the loss can be reduced through the LCL compensation topological network.
For the constructed MCR-WPT system, a receiving side can receive the mixing signals transmitted by a transmitting side, the receiving side generally comprises n paths of mutually independent receivers, and the number n of the receivers in the receiving side can be selected according to actual needs; the n-path receivers are independent from each other, specifically, the resonance frequencies of the receivers are different from each other, and when any one of the n-path receivers receives the fundamental frequency signal matched with the resonance frequency, the reception of the corresponding fundamental frequency signal by other receivers is not influenced. The resonance frequency of the receiver is matched with the fundamental frequency signal, specifically, the resonance frequency of the receiver is consistent with the frequency of the fundamental frequency signal. When the receiver works specifically, any receiver is connected with a load in a matching mode, so that the direct current power is provided for the load connected to the receiver according to the received fundamental frequency signal, the specific condition that the receiver provides the direct current power for the load is consistent with the existing condition, and the specific condition that the direct current power is provided is subject to meeting the actual application requirement.
During specific implementation, for the constructed multi-frequency multi-load MCR-WPT system, working parameters of a receiver on a receiving side and the like can be determined according to actual needs, that is, for the constructed multi-frequency multi-load MCR-WPT system, working parameters of an LCL compensation topology network in a transmitting side internal transmitting loop need to be determined, so that the multi-frequency multi-load MCR-WPT system can meet required working state requirements, for example, the multi-frequency multi-load MCR-WPT system is configured to be in a ZCS (zero-current switching) state, when the multi-frequency multi-load MCR-WPT system is in the ZCS state, switching loss can be reduced, and by a resonance method, current at two ends of an electronic switch is in a zero state when the switch is turned off, and the switching loss is close to zero.
The working parameters of the LCL compensation topological network are compensation network parameters of the LCL compensation topological network. In order to determine the compensation network parameters of the LCL compensation topology network in the transmission loop, a fitness function needs to be established, wherein when the fitness function is established, the zero-phase working frequency, the system basic parameters and the target working frequency of the multi-frequency multi-load MCR-WPT system need to be determined. The details of the zero-phase operating frequency, the basic system parameters, and the target operating frequency of the multi-frequency multi-load MCR-WPT system are specifically explained below.
After the fitness function is established, the compensation network parameters of the LCL compensation topological network in the transmitting loop are determined in an optimization mode according to the established fitness function, the working parameters of the transmitting loop can be determined after the compensation network parameters of the LCL compensation topological network in the transmitting loop are determined, and the construction and design of the multi-frequency multi-load MCR-WPT system can be realized after the working parameters of the transmitting loop are determined.
Further, the single inverter is an inverter constructed based on a gallium nitride field effect transistor;
the transmitting loop comprises n-2 LC resonant loops and an LCL basic compensation network, the LC resonant loops in the transmitting loop are sequentially connected in an adaptive manner, and the basic compensation network comprises a transmitting coil L p ；
When n fundamental frequency signals to be modulated are transmitted and output based on mixed signals modulated by MFMA superposition, the resonant frequencies of n receivers correspond to the frequencies of n fundamental frequency signals modulated by MFMA superposition one by one.
In the embodiment of the invention, the single inverter adopts the inverter constructed based on the gallium nitride field effect tube, namely, the bridge arm in the inverter is formed by connecting the gallium nitride field effect tubes, and when the bridge arm in the inverter adopts the inverter based on the gallium nitride field effect tube, the formed single inverter has higher frequency working performance and smaller conduction loss.
Fig. 1 shows a specific implementation of the single inverter, specifically, the single inverter includes an NMOS transistor Q1, an NMOS transistor Q2, an NMOS transistor Q3, and an NMOS transistor Q4, that is, the NMOS transistor Q1, the NMOS transistor Q2, the NMOS transistor Q3, and the NMOS transistor Q4 are all gallium nitride field effect transistors, and a drain end of the NMOS transistor Q1, a drain end of the NMOS transistor Q2, and a signal source V in Is connected with the positive terminal of the NMOS tube Q1The drain terminal of the NMOS tube Q3, the source terminal of the NMOS tube Q2 and the drain terminal of the NMOS tube Q4, the source terminal of the NMOS tube Q3 and the source terminal of the NMOS tube Q4 are connected with the signal source V in The connecting end of (1).
For the transmitting loop, one end of the transmitting loop is connected with the source end of the NMOS tube Q1 and the drain end of the NMOS tube Q3, and the other end of the transmitting loop is connected with the source end of the NMOS tube Q2 and the drain end of the NMOS tube Q4.
For the transmitting loop, the transmitting loop comprises n-2 LC resonant loops and one LCL basic compensation loop, the LCL basic compensation network can provide two resonant frequencies for the system, and each time one LC resonant loop is added, one resonant frequency can be added for the system. Therefore, when n-2 LC resonant circuits are sequentially adaptively connected with the basic LCL compensation network, n resonant frequencies can be provided for the system, namely, power transmission of n different frequency channels is realized.
In FIG. 1, L p Being transmitting coils, R p The capacitance in the LC resonance circuit is the transmission resonance capacitance, and the inductance in the LC resonance circuit is the transmission resonance inductance, wherein, the transmission resonance capacitance C p1 And a transmission resonance inductor L f1 Form a resonant LC resonant circuit and a transmitting resonant inductor L f1 First terminal of (1), transmission resonance capacitor C p1 First end of and transmitting coil L p Is connected to one end of a transmitting coil L p Another end of the resistor and a transmission loop resistor R p Is connected with a transmitting loop resistor R p The other end of the second resistor is connected with the second ends of all the transmitting resonant capacitors and a bridge arm where the NMOS tube Q2 is located.
In specific implementation, the second end of the transmitting resonant capacitor in the LC resonant circuit and the resistor R of the transmitting circuit p And the bridge arm where the NMOS tube Q2 is positioned is in adaptive connection, the first end of the transmitting resonant capacitor is in adaptive connection with the end part of the transmitting resonant inductor forming the LCL structure, such as the transmitting resonant capacitor C p1 First terminal of (2) and transmission resonance inductor L f1 And a transmitting coil L p Connected when there is a transmission resonant capacitance C p2 While, the transmission resonance capacitor C p2 First terminal of (2) and transmission resonance inductor L f1 And a transmission resonance inductance L f2 And (6) adapting connection.
Specifically, when a plurality of LC resonant circuits exist in the transmitting circuit, reference may be made to fig. 1 and the above description, and details are not described here.
When the transmitting side transmits the mixed signal based on the MFMA superposition modulation, that is, the MFMA superposition modulation is adopted to superpose n kinds of sine wave signals with different frequencies and different amplitudes on a time domain, so as to obtain the mixed signal containing n kinds of sine wave information, that is, the fundamental frequency signal based on the MFMA superposition modulation is a sine wave signal, and specifically, the method comprises the following steps:
Figure BDA0003797555280000071
wherein, ω is 1 ≠ω 2 ≠…≠ω n ∈[ω min ,ω max ]，max|y(t)|≤y max ，y 1 (t)、y 2 (t)、…y n And (t) is n sine wave signals with different frequencies and different amplitudes. For convenience of subsequent analysis and experiments, A 1 ，A 2 ，…，A n Respectively representing the normalized voltage amplitudes of the n sine wave signals.
In specific implementation, the mixing signal can be generated by an external signal source, and the voltage amplitude of the fundamental frequency signal in the mixing signal generated by the external signal source can be adjusted according to needs. In order to realize the transmission of the mixing signals, the mixing signals containing n kinds of fundamental frequency signals are compared with the high-frequency triangular carrier, so that high level and low level are respectively output according to the comparison result, and the corresponding high level and low level are utilized to control the conduction state of corresponding NMOS tubes in two pairs of single inverters, namely, the required mixing signals can be transmitted through a transmitting loop according to the conduction state of the single inverters. Omega min 、ω max The method can be selected and determined according to actual needs, so that the requirements of actual application can be met.
In fig. 1 and fig. 2, the MFMA Modulation circuit is a circuit module for modulating the mixed signal based on the MFMA superposition to obtain the complementary square wave control signal. In particular high frequency triangular wavesThe frequency needs to be greater than twice the maximum input voltage signal frequency; in the embodiment of the invention, an external signal source generates a mixing signal, a voltage comparator in the circuit module is adopted to realize the comparison of the mixing signal and the high-frequency triangular wave, the two signals are respectively input into a non-inverting input end and an inverting input end of the voltage comparator, when the voltage of the mixing signal is greater than the voltage of the triangular wave, a high level is output, otherwise, a low level is output. In specific implementation, the obtained square wave signals are inverted, so that a set of complementary square wave control signals is obtained. The square wave signal contains the amplitude and frequency information of the fundamental frequency signal in the mixing signal, and is the control signal at this time. The control signal is used for controlling the conduction of a corresponding NMOS switch tube in the single inverter and converting the control signal into a power signal, the power signal also contains the amplitude and frequency information of a fundamental frequency signal, and the voltage amplitude of the power signal is input into a direct current power supply V of the single inverter in And (6) determining.
In conclusion, a mixed signal based on MFMA superposition modulation can be generated, and the mixed signal is transmitted to the receiving side of the multi-frequency multi-load MCR-WPT system through a transmitting loop.
In particular, the amplitude of any fundamental frequency signal is independently adjustable, such as the normalized voltage amplitude A of the ith fundamental frequency signal i The power distribution circuit can be independently adjusted to adjust the output voltage of the ith receiver corresponding to the resonant frequency, so that the power distribution among loads is independent. In specific implementation, the fundamental frequency signal can adopt the existing common technical means to realize the adjustment of the voltage amplitude, and the specific voltage amplitude adjustment mode can be selected according to the requirement so as to meet the requirement of the adjustment of the voltage amplitude.
Figure BDA0003797555280000081
The effective value of the voltage of the corresponding frequency actually output by the single inverter is as follows:
Figure BDA0003797555280000082
wherein, A 1 ，A 2 ，…，A n Respectively for n fundamental frequency signalsNormalized voltage amplitude, V in The supply voltage amplitude of the dc power supply.
As can be seen from the above description, when n fundamental frequency signals to be modulated are transmitted and output based on the mixed signal modulated by the MFMA superposition, the resonant frequencies of the n receivers are in one-to-one correspondence with the frequencies of the n fundamental frequency signals modulated by the MFMA superposition, that is, one receiver only receives the fundamental frequency signal corresponding to the resonant frequency.
When the transmission frequency of the base frequency signal in the transmitting side internal transmission mixing signal is near the natural resonance frequency point of each receiver, when the amplitude of the base frequency signal in the MFMA superposition modulation is a normalized amplitude, the direct current power of the load adaptive to the receivers and connected with the receivers on the receiving side corresponds to the amplitude of the base frequency signal coupled by the receivers one to one, and thus the independent adjustment of the receiving power of each load can be realized; the method includes the steps of realizing independent adjustment of received power of each load, specifically, enabling direct current power received by a load at a receiving side to correspond to amplitude values of fundamental frequency signals of a connected receiver, and enabling the ratio of the direct current power of two loads which are just corresponding to the amplitude values to be consistent with the ratio of the adjusted amplitude values when the amplitude values of the two fundamental frequency signals are adjusted independently.
Furthermore, for any receiver, the receiver comprises a coupling inductance coil and an uncontrollable rectifier, wherein two ends of the coupling inductance coil are respectively connected with the uncontrollable rectifier in an adaptive manner through a receiver loop wire resistor and a receiving tuning capacitor so as to be connected with a load in an adaptive manner through the uncontrollable rectifier, so that the required direct current power is provided for the load.
Fig. 1 shows a case where the receiving side includes n receivers, which may generally take the same form, and the following description is made in the case of the first receiver in fig. 1, where the first receiver includes a receiver loop wire resistor R in fig. 1 s1 Reception tuning capacitor C s1 And a coupling inductance coil L s1 Coupled inductor L s1 The dotted terminal and the loop conductor resistance R of the receiver s1 Is connected with one end of the coupling inductance coil L s1 Non-homonymous terminal of and receiving tuning capacitor C s1 Is connected to the receiver loop wire resistance R s1 And a receiving tuning capacitor C s1 The other ends of the two are connected with an uncontrollable rectifier (Rectifler 1) which is used as a load R L1 The uncontrollable rectifier can adopt the existing common form and can be selected according to the requirement.
The natural resonant frequency of the first receiver is adjusted by a receiving tuning capacitor C in the receiver s1 And a coupling inductance coil L s1 And determining that the natural resonant frequency of the other receivers is consistent with the natural resonant frequency of the first receiver, which is not illustrated here. In specific implementation, reference may be made to fig. 1 and the above description for implementation of other receivers, and a description thereof is not repeated.
In particular, the determined system essential parameter comprises the expected resonance frequency f of the receiver ei (i =1, 2.. N), a transmitting coil L p Inductor and receiving coil L of si I =1,2, n, the transmitting coil L p With respective coupling inductors L si Mutual inductance between M psi (i =1,2.. Multidot., n), receiver load R Li (i =1, 2.. Multidot., n) and a reception tuning capacitance C si (i＝1,2,...,n)。
In the embodiment of the invention, basic parameters of the constructed multi-frequency multi-load MCR-WPT system can be specifically determined according to conditions such as practical application scenes, and in addition, the resistance R of the transmitting loop in the figure 1 p The resistance of the receiver loop conductor is determined according to the length of the line, and the load resistance R is determined according to the length of the line L1 The resistance of the receiver is determined based on the resistance of the device to which the receiver outputs. Therefore, the parameters to be determined are the parameters corresponding to the transmission resonant inductance and the transmission resonant capacitance in the LCL compensation topology network.
Further, a circuit model matrix of the multi-frequency multi-load MCR-WPT system is determined by adopting a fundamental wave analysis method, and a total reflection impedance frequency characteristic curve is obtained according to the determined circuit model matrix, wherein the zero-phase working frequency of the multi-frequency multi-load MCR-WPT system is obtained by using a graphical method for the obtained total reflection impedance frequency characteristic curve.
Specifically, the steady-state Analysis is performed by a Fundamental Analysis (FHA) method, which is performed according to KCL (kirchhoff's current law) and KVL (kirchhoff's voltage law), and only the Fundamental component is considered, and the direct-current component or higher harmonics are ignored. And taking signals with different frequencies in the mixed signals as fundamental waves respectively, regarding the multi-frequency multi-load MCR-WPT system as circuits with the same number, wherein each circuit only has one fundamental wave signal with corresponding frequency.
In the embodiment of the invention, an equivalent circuit of a pair of multi-channel MCR-WPT systems is established by adopting a fundamental wave analysis method, the original multi-frequency multi-load MCR-WPT system is divided into a plurality of single-frequency multi-load MCR-WPT systems according to power channels for analysis, each single system is driven by an alternating current voltage source, and all the single systems have the same circuit structure.
Performing system impedance analysis on a circuit model matrix of a multi-frequency multi-load MCR-WPT system generated based on a fundamental wave analysis method to calculate the total input impedance Z of the system in(i) And the system operating frequency f i The relationship between the input impedance and the frequency of the input impedance of the system can be drawn. The reflected impedance of each receiver at the transmitting side is Z rn(i) The calculation formula is as follows:
Figure BDA0003797555280000091
wherein, ω is i Representing the actual operating angular frequency of the system, M ps1 ，M ps2 ，...，M psn Are respectively a transmitting coil L p Mutual inductance, C, between coupled inductors s1 ，C s2 ，...，C sn Respectively, the capacitance value, L, of the receiving tuning capacitor in each receiver s1 ，L s2 ，...，L sn Coil self-inductance, R, of the coupled inductors in each receiver s1 ，R s2 ，...，R sn Respectively, the resistance value, R, of the loop conductor resistor of the receiver in each receiver eq1 ，R eq2 ，...，R eqn Respectively uncontrollable in each receiverThe equivalent resistance of the current device and the connected load.
Z r1(i) Is the reflected impedance of the first receiver on the transmitting side, Z r2(i) For the reflected impedance of the second receiver at the transmitting side, and so on, e.g. Z rn(i) The reflected impedance of the nth receiver on the transmit side. After obtaining the corresponding reflection impedance of any receiver at the transmitting side, the working frequency f of the system can be calculated according to the circuit theory i Input impedance of lower is Z in(i) 。
When the working frequency is f i Time, system input impedance angle θ i The calculation formula is as follows: theta.theta. i ＝arctan(Im(Z in(i) )/Re(Z in(i) ) Based on the characteristic curve, the angular frequency characteristic curve of the input impedance of the system can be drawn, the frequency of the system working at a Zero Phase Angle (ZPA) can be obtained through the analysis of the angular frequency characteristic curve of the input impedance of the system, the actual resonance frequency of the multi-frequency multi-system MCR-WPT system is obtained, and therefore the Zero Phase working frequency of the multi-frequency multi-load MCR-WPT system is obtained through a graphical method. The operating frequency is a frequency at a point where the input impedance angle is 0 ° and the system input impedance angular frequency characteristic curve intersects, as shown in fig. 4.
Further, constructing a fitness function according to the zero-phase working frequency, the system basic parameters and the target working frequency as follows:
Figure BDA0003797555280000101
wherein Fitness is the constructed Fitness function, f ei Indicating the desired resonance frequency, f, of the ith receiver si Is the ith actual resonance frequency of the system, i.e. the ith zero-phase working frequency of the system, I si For the load current of the ith receiver, I sj For the jth receiver load current, k si The derivative, k, of the angular frequency characteristic curve of the input impedance of the system at the ith actual resonance frequency point sj The derivative, W, of the angular frequency characteristic of the input impedance of the system at the j-th actual resonance frequency point 1 Is at resonanceFrequency optimizing weight coefficient, W 2 Optimizing weight coefficient and W for load current 3 And the weight coefficient of the derivative of the angular frequency characteristic curve of the input impedance of the system at the resonance point.
In the embodiment of the invention, after the Fitness function Fitness is established, optimization can be carried out to determine the compensation network parameters of the LCL compensation topological network in the transmitting loop. In specific implementation, the expected resonant frequency f of the ith receiver ei The specific frequency range may be 100kHz-300kHz, wherein the desired resonance frequency f of the i-th receiver ei I.e., the target operating frequency of the ith receiver, the desired resonant frequency point may be assigned according to the number of receivers.
Ith actual resonance frequency f of system si Obtaining a zero-phase working frequency point of the system by a graphical method, wherein generally, when n receivers exist on a receiving side, and an LCL basic compensation network is sequentially adaptively connected with an n-2 LC resonant circuits on a transmitting side, n actual resonant frequencies of the system exist in the multi-frequency multi-load MCR-WPT system. The expected resonance frequency of the receiver is preset and determined according to requirements, the total input impedance of a system (the system specifically refers to a constructed multi-frequency multi-load MCR-WPT system, namely a transmitting side and a receiving side) can be obtained through an LCL compensation topological network, and the actual resonance frequency of the multi-frequency multi-load MCR-WPT system in a ZCS state is obtained through a graphical method, wherein the actual resonance frequency is close to the designed expected resonance frequency as much as possible so as to guarantee power transmission.
In specific implementation, the optimization weight coefficient W1 of the resonant frequency is greater than the optimization weight coefficient W2 of the load current and is greater than or equal to the weight coefficient W3 of the derivative of the angular frequency characteristic curve of the input impedance at the resonant point, for example, the optimization weight coefficient W1 of the resonant frequency can be respectively set to be 0.6,0.2 and 0.2.
Load current I of ith receiver si According to a theoretical matrix of a system circuit, the load current of the ith receiver is obtained by solving an equation by matlab; the load currents of the other receivers can be determined in the same manner. According to the system input impedance frequency characteristic curve, after a system zero-phase working frequency working point is obtained, the derivative of the point is calculated, and the derivative k at the ith actual resonance frequency point is obtained si ，k sj With the derivative k at the ith actual resonance frequency point si Similarly, no further description is provided herein.
Therefore, the Fitness function Fitness can be specifically constructed based on the system basic parameters, the target working frequency, the load current and the derivative of the angular frequency characteristic curve of the input impedance of the resonance point.
Further, for the constructed fitness function, optimizing and determining a compensation network parameter of an LCL compensation topological network in a transmitting loop based on a PSO method, wherein the process of determining the compensation network parameter comprises the following steps:
step 1, configuring a transmitting resonance inductor in a transmitting loop into inductive particles, configuring a transmitting resonance capacitor in the transmitting loop into capacitive particles, and determining the ith inductive particle L fi And the ith capacitive particle C fi Corresponding value range and motion speed range, wherein i =1, 2.. And n-1;
specifically, the inductive particle L fi And a capacitor particle C fi The corresponding value range and the movement speed range are as follows:
Figure BDA0003797555280000111
Figure BDA0003797555280000112
wherein, vL fi For the ith inductive particle L fi Velocity, vC fi Is the ith capacitive particle C fi The corresponding speed.
Step 2, randomly initializing inductive particles L fi And a capacitor particle C fi ；
Specifically, at the time of random initialization, the inductive particles L fi And a capacitor particle C fi The corresponding value range and the movement speed range need to be within the value range.
Step 3, according to the inductance particles L after random initialization fi And electricityVolume particle C fi Determining the value of fitness function, and updating inductance particles L fi Capacitive particle C fi Corresponding individual optimal fitness and population optimal fitness;
step 4, updating the inductive particles L according to the above fi Capacitive particle C fi Updating the inductive particles L according to the individual best fitness and the group best fitness fi Capacitive particle C fi The corresponding speed and position, wherein,
Figure BDA0003797555280000121
in the formula, L mi Is an inductive particle L fi Individual optimum particle, Y Lmi Is an inductive particle L fi Population-optimal particles of, L i Is an inductive particle L fi All particles of the contemporary population; c mi Is a capacitive particle C fi Individually optimized particles, Y Cmi Is a capacitive particle C fi Population-optimal particles, C i Is a capacitive particle C fi All particles of the contemporary population; beta is the inertial weight, lambda 1 And λ 2 Self-learning factors and group learning factors are respectively; rand is a random number generator;
specifically, the next generation of inductive particles and capacitive particles can be obtained through the updating of step 4. The inertia weight beta determines the magnitude of the position updating amplitude of each generation of inductive particles and capacitive particles, and determines the convergence capability of a fitness function; self-learning factor lambda 1 The learning ability of inductance particles and capacitance particles to self optimal particles can be adjusted; population learning factor lambda 2 The ability of learning from inductive particles and capacitive particles to the group of optimal particles can be adjusted. Typically, the inertial weight β is set to 0.9, the self-learning factor λ 1 And group learning factor lambda 2 And may both be set to 1.2 in general.
During specific implementation, when optimization is carried out based on PSO, parameters of the transmitting resonant inductor and the transmitting resonant capacitor and basic system parameters are substituted into a system input impedance formula, an angular frequency characteristic curve of the system input impedance can be drawn, and the actual resonant frequency and the derivative of the corresponding actual resonant frequency point of the multi-frequency multi-load MCR-WPT system can be obtained through the angular frequency characteristic curve of the system input impedance.
Substituting the parameters of the transmitting resonant inductor and the transmitting resonant capacitor and the basic system parameters into a circuit matrix model to calculate the load current; therefore, the value of the constructed fitness function can be calculated, and the conversion process from the transmission resonance inductance and the transmission resonance capacitance to the fitness is completed.
For example, 100 groups of inductive particles and capacitive particles are initialized randomly, each inductive particle and each capacitive particle is an individual, the fitness corresponding to each group of inductive particles and capacitive particles is calculated and recorded as the individual optimal fitness, the lowest fitness obtained by transverse comparison is the group optimal fitness, and the corresponding group of inductive capacitors is recorded as the historical group optimal particle.
And (3) according to a speed and position updating formula, obtaining values of the next 100 groups of inductive capacitors, then carrying out fitness calculation, carrying out transverse comparison on the fitness of the generation to obtain the current best fitness, and then carrying out longitudinal comparison on the current best fitness and the population best fitness to update the population best fitness.
Step 5, iteration frequency control is carried out until the inductive particles L with the optimal fitness are found fi And a capacitor particle C fi To determine compensation network parameters of the LCL compensation topology network in the transmission loop.
In specific implementation, the iteration times can be generally set to 500 times, and if the optimal fitness is still high after the iteration is finished, the iteration times are continuously added until the optimal fitness of the system is obtained. In specific implementation, the optimal fitness is 0, and in specific implementation, the closer the calculated fitness value is to 0, the better, and the fitness value can be determined according to actual needs so as to meet the actual application requirements.
In the embodiment of the invention, in the range of iteration times, the values corresponding to the inductive particles and the capacitive particles and corresponding to the optimal fitness value are used as the values corresponding to the transmitting resonant inductance and the transmitting resonant capacitance, namely, the compensation network parameters of the LCL compensation topological network in the transmitting loop are determined by utilizing a PSO optimization mode. Of course, in specific implementation, other manners of determining the compensation network parameter may also be adopted, and the method may be specifically selected according to needs.
As shown in fig. 2, the multi-frequency multi-load MCR-WPT system based on the MFMA superposition modulation method and the parameter design process thereof are specifically described below by taking a one-to-two dual-frequency dual-load MCR-WPT system as an example, in this case, the receiving side includes two receivers.
Specifically, a dual-frequency dual-load MCR-WPT system is constructed, and specific cases of the constructed dual-frequency dual-load MCR-WPT system may refer to fig. 2 and the above description, which are not described herein again.
After the double-frequency double-load MCR-WPT is constructed, configuring the expected transmission frequency: f. of e1 =180kHz and f e2 =220kHz, i.e. the desired resonance frequency of a receiver is f e1 =180kHz, the desired resonance frequency of the other receiver is f e2 ＝220kHz。
By adopting an MFMA superposition modulation method, a sine wave signal of a system desired transmission frequency is superposed on a time domain to obtain a mixed signal containing information of the two sine waves, and the method includes:
Figure BDA0003797555280000131
wherein, ω is 1 ＝2πf e1 ，ω 2 ＝2πf e2 ，max|y(t)|≤y max 。
I in(i) ，I p(i) ，I s1(i) ，I s2(i) Respectively, the system input current, the transmitting side current and the receiver loop current, wherein i in brackets represents the switching frequency f i (i＝1,2)。
Figure BDA0003797555280000132
And
Figure BDA0003797555280000133
if the effective value of the voltage of the corresponding frequency actually output by the inverter is the following value:
Figure BDA0003797555280000134
equivalent load R of uncontrollable rectifier and load eq1 And R eq2 Can be expressed as:
Figure BDA0003797555280000135
a circuit model matrix of a double-frequency double-load MCR-WPT system is established by using KCL and KVL theorem, and the matrix comprises the following components:
Figure BDA0003797555280000136
from the above description, it can be seen that the receiver loop current I is based on the basic system parameters s1(i) ，I s2(i) The loop current I of the receiver can be obtained by solving matlab and utilizing the matlab s1(i) ，I s2(i) The specific processes are consistent with the prior art and are well known to those skilled in the art, and will not be described herein. In specific implementation, for a multi-frequency multi-load MCR-WPT system, the specific situation of the receiver loop current may refer to the solution determination process of the dual-frequency dual-load MCR-WPT system, which is well known to those skilled in the art.
Establishing an equivalent circuit of a double-frequency double-load MCR-WPT system by adopting a fundamental wave analysis method, and dividing the original double-frequency MCR-WPT system into two single-frequency multi-load MCR-WPT systems according to a power channel to perform impedance analysis, as shown in figure 3; at this time, the reflection impedance of the receiver at the transmission side is Z r1(i) And Z r2(i) The formula is as follows:
Figure BDA0003797555280000141
therefore, the double-frequency double-load MCR-WPT system is at a certain working frequency f i Input impedance of time Z in(i) Can be expressed as:
Figure BDA0003797555280000142
and (5) drawing and analyzing an angular frequency characteristic curve of the input impedance of the system by utilizing matlab to obtain the derivative of the actual resonant frequency of the system and the actual resonant frequency point.
The above gives the impedance Z according to the reflection at the transmitting side r1(i) And the reflection impedance Z of the transmitting side r2(i) Determining the input impedance Z of the corresponding dual-frequency dual-load in(i) For multi-frequency multi-load, determining the operating frequency f i Input impedance of time Z in(i) Reference may be made to the description herein, which is well known to those skilled in the art, and the description is not repeated here.
Fig. 4 is a characteristic curve of angular frequency of input impedance of the system, and in fig. 4, the ordinate is the angular frequency of the input impedance, and the abscissa is the input frequency; fig. 5 is a frequency characteristic curve of the input impedance of the system, and the ordinate of fig. 5 is the input impedance and the abscissa is the input frequency. And according to the graphs in the figures 4 and 5, the corresponding actual resonant frequency of the double-frequency double-load MCR-WPT system can be obtained.
The basic system parameters of the double-frequency double-load MCR-WPT system are set as shown in the following table:
electrical parameter of
V in 100V
L p 100μH
L s1 100μH
L s2 100μH
C s1 8nF
C s2 5nF
R L1 10Ω
R L2 10Ω
M ps1 50μH
M ps2 50μH
After basic parameters of the system are set, the transmission resonant inductor L in the LCL compensation topological network of the remaining transmission side is selected f And a transmission resonance capacitor C p As a quantity to be optimized of the PSO algorithm, the expected resonant frequency, the receiver loop current and the derivative of the angular frequency characteristic curve of the input impedance of the resonant point are confirmed as an optimization target.
Setting W based on the expected resonant frequency as the main optimizing target 1 Optimizing weight coefficient, W, for resonant frequency 2 Optimizing weight coefficient and W for load current 3 The weight coefficients of the derivatives of the input impedance angular frequency characteristic curves for the resonance points are 0.6,0.2, respectively. Constructed fitness function as：
Fitness＝W 1 (|f s1 -f e1 | 2 +|f s2 -f e2 | 2 )+W 2 |I s1 -I s2 | 2 +W 3 |k s1 -k s2 | 2
When a PSO optimizing mode is adopted, the final optimizing result is specifically as follows:
L f ＝72μH，C p ＝10nF，f s1 ＝180.7kHz,f s2 ＝211.5kHz。
in summary, by the above design process of the dual-frequency dual-load MCR-WPT system and the specific design method, the above description can be referred to for other design situations.
By adjusting the normalized voltage amplitude of each frequency at the input terminal, the two load voltages are as shown in fig. 6-8. In FIG. 6, a load voltage U of the receiver output L1 16.08V, and the load voltage U output by another receiver L2 Is 15.95V, and the corresponding amplitude A of the fundamental frequency signal 1 ＝A 2 =1. In FIG. 7, is A 1 ＝1.5，A 2 =0.5, at which a load voltage U is output from a receiver L1 22.57V, the load voltage U output by another receiver L2 8.375V; in FIG. 8, is A 1 ＝1，A 2 =0.8, at this time, a load voltage U of a receiver output L1 15.55V, and the output of another receiver L2 And 12.44V.
FFT (fast Fourier transform) analysis was performed on the voltages output from the two receivers, as shown in fig. 9 to 11. As can be seen from the figure, the variation of the system output voltage is consistent with the amplitude of the normalized voltage corresponding to the frequency, i.e., the amplitude of the fundamental frequency signal determines the magnitude of the frequency signal received by the corresponding receiver, which can determine the magnitude of the power voltage output to the load.
In matlab, FFT analysis is carried out on the output voltage of the receiver by using an FFT analysis tool, wherein the abscissa of the FFT analysis tool is frequency, and the ordinate of the FFT analysis tool is the component of each frequency in the signal; the mode and the process of the FFT analysis tool in matlab specifically carrying out the FFT analysis on the voltage are consistent with those of the prior art. In FIG. 9, the receiver outputs powerIncluding the actual resonant frequency f of the system s1 And the actual resonance frequency f of the system s2 Frequency component, ratio thereof to fundamental frequency amplitude A 1 ＝A 2 The ratio of =1 is the same. In fig. 10, the receiver output voltage contains f above s1 And f s2 Frequency component, ratio thereof to fundamental frequency amplitude A 1 ＝1.5，A 2 The ratio corresponding to 0.5 is the same. In FIG. 11, the receiver output voltage contains f as described above s1 And f s2 Frequency component, ratio thereof to fundamental frequency amplitude A 1 ＝1，A 2 The ratio is the same for 0.8.
The above embodiments are merely preferred embodiments for fully illustrating the present invention, and the scope of the present invention is not limited thereto. The equivalent substitution or change made by the technical personnel in the technical field on the basis of the invention is all within the protection scope of the invention. The protection scope of the invention is subject to the claims.

Claims (10)

1. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation is characterized in that:
constructing a required multi-frequency multi-load MCR-WPT system, wherein the multi-frequency multi-load MCR-WPT system comprises a transmitting side and a receiving side, the transmitting side transmits mixed signals based on MFMA superposition modulation, the receiving side is used for receiving the mixed signals transmitted by the transmitting side, the transmitting side comprises a single inverter and a transmitting loop which is adaptively connected with the single inverter, and the transmitting loop comprises an LCL compensation topological network; the receiving side comprises n paths of mutually independent receivers, and any receiver in the receiving side is coupled with a fundamental frequency signal matched with the resonance frequency of the receiver in a received mixed signal so as to provide direct current power for a load connected to the receiver;
determining zero-phase working frequency, system basic parameters and target working frequency of the multi-frequency multi-load MCR-WPT system, and constructing a fitness function according to the determined zero-phase working frequency, system basic parameters and target working frequency;
based on the constructed fitness function, optimizing and determining compensation network parameters of an LCL compensation topological network in a transmitting loop so as to configure the multi-frequency multi-load MCR-WPT system to be in a ZCS state when the transmitting loop based on the determined compensation network parameters works.
2. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation as claimed in claim 1, wherein: the single inverter is an inverter constructed based on a gallium nitride field effect transistor;
the transmitting loop comprises n-2 LC resonant loops and an LCL basic compensation network, the LC resonant loops in the transmitting loop are sequentially connected in an adaptive manner, and the basic compensation network comprises a transmitting coil L p ；
When n fundamental frequency signals to be modulated are transmitted and output based on mixed signals modulated by MFMA superposition, the resonant frequencies of n receivers correspond to the frequencies of n fundamental frequency signals modulated by MFMA superposition one by one.
3. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation as claimed in claim 1, wherein: the frequencies of the n fundamental frequency signals are completely different, and the amplitude of any fundamental frequency signal is independently adjustable.
4. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation as claimed in claim 1, wherein: determining a circuit model matrix of the multi-frequency multi-load MCR-WPT system by adopting a fundamental wave analysis method, and obtaining a total reflection impedance frequency characteristic curve according to the determined circuit model matrix, wherein the zero-phase working frequency of the multi-frequency multi-load MCR-WPT system is obtained by using a graphical method for the obtained total reflection impedance frequency characteristic curve.
5. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation as claimed in claim 4, characterized by: and for any receiver, the receiver comprises a coupling inductance coil and an uncontrollable rectifier, wherein two ends of the coupling inductance coil are respectively in adaptive connection with the uncontrollable rectifier through a receiver loop wire resistor and a receiving tuning capacitor so as to be in adaptive connection with a load through the uncontrollable rectifier to provide required direct current power for the load.
6. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation according to claim 5, characterized in that,
the determined basic parameters of the system include the desired resonance frequency f of the receiver ei (i =1,2.. Multidot., n), a transmitting coil L p Inductance, receiving coil L si Inductance (i =1, 2.. Multidot.n), transmitting coil L p With respective coupling inductors L si Mutual inductance M psi (i =1, 2.. Multidot.n), receiver load R Li (i =1, 2.. N) and a receive tuning capacitance C si (i＝1,2,...,n)。
7. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation according to claim 6, wherein the fitness function is constructed according to the zero-phase working frequency, the system basic parameters and the target working frequency as follows:
Figure FDA0003797555270000021
wherein Fitness is the constructed Fitness function, f ei Indicating the desired resonance frequency, f, of the ith receiver si Is the ith actual resonance frequency of the system, i.e. the ith zero-phase working frequency of the system, I si Is the load current of the ith receiver, I sj For the jth receiver load current, k si Is the derivative, k, of the angular frequency characteristic of the input impedance of the system at the ith actual resonance frequency point sj The derivative of the angular frequency characteristic curve of the input impedance of the system at the j-th actual resonance frequency point, W 1 Optimizing weight coefficient, W, for resonant frequency 2 Optimizing weight coefficient and W for load current 3 And the weight coefficient of the derivative of the angular frequency characteristic curve of the input impedance of the system at the resonance point.
8. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation according to any one of claims 2 to 7, wherein the constructed fitness function is optimized to determine the compensation network parameters of the LCL compensation topological network in the transmission loop based on the PSO method, wherein the process of determining the compensation network parameters comprises the following steps:
step 1, configuring a transmitting resonance inductor in a transmitting loop as an inductive particle, configuring a transmitting resonance capacitor in the transmitting loop as a capacitive particle, and determining the ith inductive particle L fi And the ith capacitive particle C fi Corresponding value range and motion speed range, wherein i =1, 2.. And n-1;
step 2, initializing inductive particles L randomly fi And a capacitor particle C fi ；
Step 3, according to the inductive particles L after random initialization fi And a capacitor particle C fi Determining the value of fitness function, and updating inductance particles L fi Capacitor particle C fi Corresponding individual optimal fitness and group optimal fitness;
step 4, updating the inductive particles L according to the above fi Capacitor particle C fi Updating the inductive particles L according to the individual best fitness and the group best fitness fi Capacitor particle C fi The corresponding speed and position, wherein,
Figure FDA0003797555270000022
in the formula, L mi Is an inductive particle L fi Individually optimized particles, Y Lmi Is an inductive particle L fi Of population-optimal particles of, L i Is an inductive particle L fi All particles of the contemporary population; c mi Is a capacitive particle C fi Individually optimized particles, Y Cmi Is a capacitive particle C fi Population-optimal particles, C i Is a capacitive particle C fi All particles of the contemporary population; beta is the inertial weight, lambda 1 And λ 2 Self-learning factors and group learning factors are respectively; rand is a random number generator;
step 5, iteration frequency control is carried out until the inductive particles L with the optimal fitness are found fi And a capacitor particle C fi To determine compensation network parameters of the LCL compensation topology network in the transmission loop.
9. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation as claimed in claim 8, wherein in step 1, the inductive particle L fi And a capacitor particle C fi The corresponding value range and the movement speed range are as follows:
Figure FDA0003797555270000031
Figure FDA0003797555270000032
wherein, vL fi Is the ith inductive particle L fi Velocity, vC fi Is the ith capacitance particle C fi The corresponding speed.
10. A multi-frequency multi-load MCR-WPT design method based on MFMA superposition modulation according to any one of claims 1 to 7, wherein the fundamental frequency signal based on MFMA superposition modulation is a sine wave signal;
based on the fact that the amplitude of the fundamental frequency signal in the MFMA superposition modulation is a normalized amplitude, for a plurality of receivers on the receiving side, the direct current power of the load adaptive to the receivers is in one-to-one correspondence with the amplitude of the fundamental frequency signal coupled by the receivers.
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