CN115065299A - Midpoint voltage balancing method applied to control of three-level permanent magnet synchronous motor - Google Patents
Midpoint voltage balancing method applied to control of three-level permanent magnet synchronous motor Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
- H02P21/0007—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using sliding mode control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
- H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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Abstract
A midpoint voltage balancing method applied to control of a three-level permanent magnet synchronous motor belongs to the technical field of motor control. The problem of midpoint potential drift caused by unbalanced upper and lower capacitors on the direct current side in practical application usually exists in the use process of the three-level inverter, and the practicability of the three-level inverter in a permanent magnet synchronous motor control system is reduced. The invention avoids the increase of the cost of the inverter caused by the change of the hardware structure, can also adapt to the requirements of different working conditions by changing the PI parameter value of the loop, and has wide applicability. The addition of the neutral point voltage balance ring can reduce the generation of low-order harmonics on the alternating current output side and increase the practicability of the three-level inverter in the PMSM control system. In addition, the invention combines and uses a vector control method based on the sliding mode speed controller, and improves the dynamic quality and the disturbance resistance of the permanent magnet synchronous motor speed regulating system by utilizing the advantages of insensitivity of the sliding mode control to disturbance and parameters, high response speed and the like.
Description
Technical Field
The invention belongs to the technical field of motor control, and particularly relates to a midpoint voltage balancing method applied to control of a three-level permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the advantages of simple mechanism, small rotational inertia, high power density, high torque current ratio and the like, and can meet the requirements of high-performance speed regulating systems for realizing quick response, enlarging the speed regulating range and the like. With the increase of the capacity and the voltage of the frequency conversion device and the limit of the capacity of a single tube, the requirement cannot be met by adopting the traditional two-level control mode, the voltage drop born by the single tube of the switching device can be reduced by adopting the three-level inverter, the quality of an output waveform is improved, and a better effect can be obtained in application. However, the problem of midpoint potential drift caused by unbalanced upper and lower capacitances at the direct current side in practical application usually exists in the use process of the three-level inverter, and the practicability of the three-level inverter in the permanent magnet synchronous motor control system is reduced.
The imbalance of the midpoint potential firstly causes the output voltage of the inverter to contain certain low-order harmonic waves, the output waveform is distorted, and even the number of output levels is reduced when the output waveform is serious. Secondly, the voltages at the two ends of the upper capacitor and the lower capacitor are different, and the total voltage is constant, so that the voltage at the two ends of one capacitor is lower, the voltage at the two ends of the other capacitor is higher, and the service life of the capacitors is influenced. Therefore, in order to ensure the normal operation of the inverter and better output performance, the midpoint potential should be maintained in a substantially balanced state as much as possible.
Disclosure of Invention
The invention aims to provide a midpoint voltage balancing method applied to control of a three-level permanent magnet synchronous motor, and solves the problem of midpoint potential drift caused by unbalance of upper and lower capacitors on a direct current side in a control system of the three-level permanent magnet synchronous motor.
The problem of midpoint potential drift caused by unbalanced upper and lower capacitors on the direct current side in practical application usually exists in the use process of the three-level inverter, and the practicability of the three-level inverter in a permanent magnet synchronous motor control system is reduced. The invention avoids the increase of the cost of the inverter caused by the change of the hardware structure, can also adapt to the requirements of different working conditions by changing the PI parameter value of the loop, and has wide applicability. The addition of the neutral point voltage balance ring can reduce the generation of low-order harmonics on the alternating current output side and increase the practicability of the three-level inverter in a PMSM control system. In addition, the invention combines and uses a vector control method based on the sliding mode speed controller, and improves the dynamic quality and the disturbance resistance of the permanent magnet synchronous motor speed regulating system by utilizing the advantages of insensitivity of the sliding mode control to disturbance and parameters, high response speed and the like. The technical scheme adopted by the invention for solving the technical problem specifically comprises the following steps:
the method comprises the following steps: measuring angular speed omega of motor by current detection circuit or encoder m Three-phase stator current, electromagnetic torque T e And a rotor position θ. Clark and Park coordinate transformation is carried out on the measured three-phase stator current value and the rotor position theta, and the fed-back d-axis current i is calculated d And q-axis current i q (ii) a Comparing the measured motor speed with the given speed to obtain an input value E of the sliding mode speed controller Nr 。
Step two: will be provided withE Nr Sending into sliding mode speed controller, and outputting as i qref With i fed back q Taking the difference as the input of a q-axis current controller, and taking 0 and i fed back d Taking the difference as the input of d-axis current controller, and then the output is respectively compensated with feedforward
And-omega e L q i q Add to obtain u q And u d 。
Step three: will u q And u d Inverse Park transformation is carried out to obtain u α And u β The voltage difference between the upper and lower capacitors on the dc side is measured simultaneously as two inputs to the three-level SVPWM module, and then enters the midpoint voltage balancing loop of fig. 11 provided by the present invention.
Step four: and reasonably distributing the redundant small vectors to realize the function of midpoint potential control. The rational allocation of the redundant small vectors requires two factors to be considered: first, the neutral point current I flows out 0 The direction of (a); second, midpoint potential error U diff A change in situation. Now will U diff Is defined as U diff ＝Bus n -Bus p In which Bus n Representing the capacitor voltage, Bus, on the DC side p Representing the capacitor voltage on the dc side.
There are 3 output states, P, O, N respectively, for each phase of the three-level inverter. In each sector, the invention uniformly selects a negative small vector as an initial small vector, and takes the 1 st small sector of the first large sector as an example, and the symmetrical space vector sequence of the corresponding basic voltage vector is as follows: ONN-OON-OOO-POO-OOO-OON-ONN.
Definition when the first vector is ONN, there is I O ＝I A When I is O If the voltage is greater than 0, the lower capacitor is considered to be discharged, and the upper capacitor is considered to be charged. When I O When the voltage is less than 0, the lower capacitor is charged, and the upper capacitor is discharged. Table 1 shows I corresponding to the starting small vectors of all sectors O ：
TABLE 1 all sectors with their starting small vector corresponding to I O
Step five: the invention obtains I O And then, designing a middle point voltage balance PI loop to adjust the action time of the initial small vector according to the charging relation of the corresponding upper capacitor and the lower capacitor. The input of the PI loop is U diff ＝Bus n -Bus p And the output is BusBalancEPIout, the BusBalancEPIout is added with a default value of 0.5 when the PI loop is not added, and the upper and lower limits of the output of the PI controller are designed to be plus or minus 0.5.
The process of positive and negative BUS regulation is discussed in two cases:
1) when the upper capacitor voltage is greater than the lower capacitor voltage, i.e. Bus n ＜Bus p ，U diff Is less than 0. At this point the PI output tends to be negative and the effect of PI is to adjust the default value 0.5 in the smaller direction.
At this time, it is judged that O In the direction of (b), if I O And > 0, the action time of the initial vector needs to be reduced, namely the time for discharging the lower capacitor is reduced. If I O If the value is less than 0, the action time of the first-generation vector needs to be increased, namely the time for charging the lower capacitor is increased.
2) When the upper capacitor voltage is less than the lower capacitor voltage, i.e. Bus n ＞Bus p ，U diff Is greater than 0. The PI output tends to a positive value, i.e., the default value of 0.5 is adjusted in the large direction.
At this time, it is judged that O In the direction of (b), if I O If the voltage is more than 0, the action time of the initial vector is shortened, namely the time for discharging the lower capacitor is prolonged. If I O <0, the action time of the initial vector is reduced, namely the time for charging the lower capacitor is reduced.
And recalculating and distributing the action time of the basic vector in each sector through the correctly judged direction and the PI loop designed by the invention, and finally outputting the PWM wave.
Step six: and controlling the switching state of each switching device of the three-level inverter by using the PWM wave generated in the step, thereby realizing the control of the permanent magnet synchronous motor.
The invention has the beneficial effects that:
the invention designs a neutral point voltage balance ring, and the addition of the neutral point voltage balance ring can also reduce the generation of low-order harmonics at the AC output side and increase the practicability of the three-level inverter in a PMSM control system.
The invention combines and uses a vector control method based on the sliding mode speed controller, and improves the dynamic quality and the disturbance resistance of the permanent magnet synchronous motor speed regulating system by utilizing the advantages of insensitivity of the sliding mode control to disturbance and parameters, high response speed and the like.
Drawings
Fig. 1 is a block diagram of a sliding-mode vector control structure of a three-level permanent magnet synchronous motor.
Fig. 2 is a power supply circuit.
Fig. 3 shows a U-phase driving circuit.
Fig. 4 is a phase current sampling circuit.
Fig. 5 shows a rotation speed and position detection circuit.
Fig. 6 is a dc bus overvoltage protection circuit.
FIG. 7 is a flowchart of a main program of the control system
FIG. 8 is a flowchart of an interrupt service routine.
Fig. 9 is a topology structure diagram of a three-level inverter.
Fig. 10 is a space vector distribution and size sector division diagram of a three-level inverter.
FIG. 11 is a diagram of a midpoint voltage balancing ring simulation design.
FIG. 12 is a plot of midpoint potential before addition of the PI loop.
FIG. 13 is a plot of midpoint potential after addition of the PI loop.
Detailed Description
In order to make the technical solutions better understood by those skilled in the art, the technical solutions in the embodiments of the present invention will be described below with reference to the drawings in the embodiments of the present invention.
Example 1
This embodiment is mainly constituted by a drive circuit and a control circuit. The driving circuit adopts three-phase full-control rectification, and obtains a motor driving signal through a three-phase inversion part formed by a power module after capacitance filtering. The driving circuit mainly comprises a three-phase current detection circuit, an optical coupling isolation circuit, an inverter, a filter capacitor, a 20kW alternating current permanent magnet synchronous motor set and the like; the control circuit adopts DSP as a main control chip, mainly completes the functions of generation of 6 paths of SVPWM control signals, AD sampling, rotor position and rotating speed detection and the like, and mainly comprises an encoder signal processing circuit, a power supply circuit, an overvoltage and overcurrent detection circuit and the like.
The method comprises the following steps: TMS320F28335 of TI company C2000 series is adopted as a main control chip. TMS320F28335 adopts low power consumption design, and the dominant frequency can reach 150MHz at most; by adopting a Harvard bus architecture, programs and data can be operated simultaneously; the high-performance 32-bit central processing unit is arranged, so that the computing capability is strong, and the response to an interrupt signal is quick; an ADC sampling module with 12-bit sampling precision; abundant on-chip memory cells and external memory interfaces; a variety of communication means may be selected.
Step two: the core voltage of TMS320F28335 is 3.3V, the I/O voltage is 1.9V, the input voltage of the current, voltage and temperature detection module is required to be 5V, the operational amplifier is required to be connected with a power supply of 15V, the input voltage of the encoder is 5.3V, and the direct current side bus voltage of the system is 24V. As shown in FIG. 2, a direct current 24V power supply is introduced from a terminal, filtered by a power supply filter LZJB11-3A, enters a DC-DC power supply module URB2405YMD-10WR3, outputs a 5V power supply, and outputs 3.3V and 1.9V power supply voltages for the DSP by LT1085CM-3.3 and AS1117-ADJ chips respectively after being filtered by capacitors.
Step three: as shown in fig. 3, the driving circuit uses IGBT modules as power switching devices, each module includes two single-phase full bridges, and a temperature sensor is designed inside each module to detect the module temperature and prevent the device from overheating. The circuit uses 3 integrated modules in total to form a three-phase full-bridge inverter. And the driving circuits of the upper and lower bridge arms use the IRS21867 to drive the chip to output two paths of complementary signals so as to control the on-off of the IGBT. The IRS21867 chip is powered by 15V, an upper bridge arm bootstrap circuit and an output protection circuit are integrated inside the chip, bridge arms are effectively prevented from being directly connected, the switching time of an output end is only 170ns at most, the requirement of high-speed control is completely met, and input signals come from PWM pulses of DSP pins.
Step four: as shown in fig. 4, the current sampling circuit uses a hall current sensor, model HC5FW, the output being a linear voltage signal. The signal is filtered, amplified, and then the voltage is output by a pin 6 of an operational amplifier OP27, in order to avoid the phenomenon that the output voltage is too large and the DSP pin is burnt out, the output end uses R16 to limit the current, and a diode D3 and a voltage regulator tube D4 further clamp the output voltage.
Step five: in this embodiment, an incremental encoder is used to detect the rotation speed of the motor and the position of the rotor, as shown in fig. 5, the differential pulse signal enters the drive board through a DB15 interface, is converted into a single-ended A, B, Z encoder signal through an AM26LS32 chip, and is finally processed by an eQEP unit of the DSP to obtain the motor position and rotation speed information.
Step six: as shown in fig. 6, in the dc bus overvoltage protection circuit, an overvoltage protection value UDREFH is set by a sliding rheostat RP3, when a sampling voltage UD exceeds a set voltage, a comparator outputs a low level, an optical coupler is turned on, an overcurrent signal UdOI is output to a high level, and a DSP responds to an external interrupt and executes a corresponding processing procedure.
Example 2
The main functions of the software part of the embodiment are realized by a main program and a timer interrupt service program.
The method comprises the following steps: main program and initialization program
As shown in fig. 7, the main program initializes variables such as a system clock, a watchdog, a GPIO port, a system interrupt, a timer, an ePWM module, an eQEP module, a correlation register, a PI controller, and SVPWM, and sets an enable of the GPIO port control system. After the system is enabled, when the timer interrupt occurs, the program is transferred to execute the interrupt service program, and after the interrupt program is executed, the program returns to the dead cycle to continue the GPIO port scanning and waits for the next interrupt.
Step two: interrupt service routine
As shown in fig. 8, in this example, a sector where the rotor is located is first determined by hall signals, and the intermediate position information corresponding to the sector is used as an estimated initial position, and then vector calculation is performed to output a PWM wave to drive the motor to move. And when the eQEP module receives the z signal, determining the accurate rotor position and finishing position detection. Each carrier cycle timer generates an interrupt, and the frequency of the timer interrupt is determined according to the switching frequency of the inverter. This example sets the timer interrupt frequency to 10kHz, i.e., the system PWM period to 100 mus. And the timer interrupts and calls an SMC (sheet Molding Compound) calculation function, a PID (proportion integration differentiation) calculation function, an SVPWM (space vector pulse width modulation) calculation function and the like to complete the sliding mode vector control process.
Claims (1)
1. A midpoint voltage balancing method applied to control of a three-level permanent magnet synchronous motor is characterized in that the adopted technical scheme comprises the following steps:
the method comprises the following steps: measuring angular speed omega of motor by current detection circuit or encoder m Three-phase stator current, electromagnetic torque T e Rotor position θ; clark and Park coordinate transformation is carried out on the measured three-phase stator current value and the rotor position theta, and the fed-back d-axis current i is calculated d And q-axis current i q (ii) a Comparing the measured motor speed with the given speed to obtain an input value E of the sliding mode speed controller Nr ；
Step two: will E Nr Sending into a sliding mode speed controller, and outputting as i qref And i is fed back q Taking the difference as the input of a q-axis current controller, and taking 0 and i fed back d Taking the difference as the input of d-axis current controller, and then the output is respectively compensated with feedforward
And-omega e L q i q Add to obtain u q And u d ；
Step three: will u q And u d Inverse Park transformation is carried out to obtain u α And u β The three-level SVPWM module is used as two inputs of the three-level SVPWM module, and simultaneously measures the potential difference of upper and lower capacitors at the direct current side, and then enters a midpoint voltage balance ring;
step four: reasonably distributing redundant small vectors to realize midpoint potential controlManufacturing a function; the rational allocation of the small redundant vectors requires two factors to be considered: one is to flow out neutral point current I 0 The direction of (a); second, midpoint potential error U diff (ii) a change in condition; now will U diff Is defined as U diff ＝Bus n -Bus p In which Bus n Representing the capacitor voltage, Bus, on the DC side p Representing the capacitor voltage on the dc side;
there are three output states, P, O, N for each phase of the three-level inverter; in each sector, uniformly selecting an I small sector of a first large sector by taking a negative small vector as a starting small vector: the symmetrical space vector sequence of the corresponding basic voltage vector is as follows: ONN-OON-OOO-POO-OOO-OON-ONN;
definition when the first vector is ONN, there is I O ＝I A When I is O When the voltage is larger than 0, the lower capacitor is considered to be discharged, and the upper capacitor is considered to be charged; when I is O When the voltage is less than 0, the lower capacitor is charged, and the upper capacitor is discharged;
step five: in the acquisition of I O Then, according to the charging relation of the corresponding upper and lower capacitors, designing a middle point voltage balance PI loop to adjust the action time of the initial small vector; the input of the PI loop is U diff ＝Bus n -Bus p The output is BusBalancEPUO, the BusBalancEPUO is added with a default value of 0.5 when the PI loop is not added, and the upper and lower limits of the output of the PI controller are designed to be plus or minus 0.5;
the process of positive and negative BUS regulation is discussed in two cases:
1) when the upper capacitor voltage is greater than the lower capacitor voltage, i.e. Bus n ＜Bus p ，U diff Less than 0; at the moment, the PI output tends to be a negative value, and the action of the PI adjusts the default value of 0.5 to a small direction;
at this time, it is judged that O In the direction of (I) O If the voltage is more than 0, the action time of the first-generation vector needs to be reduced, namely the time for discharging the lower capacitor is reduced; when I is O If the value is less than 0, the action time of the first-generation vector needs to be increased, namely the time for charging the lower capacitor is increased;
2) when the upper capacitor voltage is less than the lower capacitor voltage, i.e. Bus n ＞Bus p ，U diff Is greater than 0; at the moment, the PI output tends to be a positive value, namely the default value of 0.5 is adjusted towards a large direction;
at this time, judgment of I O In the direction of (I) O If the current is more than 0, the action time of the first-generation vector is shortened, namely the time for discharging the lower capacitor is prolonged; if I O If the value is less than 0, the action time of the first-generation vector is reduced, namely the time for charging the lower capacitor is reduced;
recalculating and distributing the action time of the basic vector in each sector through the correctly judged direction and the designed PI loop, and finally outputting a PWM wave;
step six: and controlling the switching state of each switching device of the three-level inverter by using the PWM wave generated in the step so as to realize the control of the permanent magnet synchronous motor.
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Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO2021044325A1 (en) * | 2019-09-05 | 2021-03-11 | Politecnico Di Torino | Method for sensorless estimating rotor position and rotor angular speed of a synchronous reluctance machine |
| CN112564567A (en) * | 2020-12-09 | 2021-03-26 | 天津工业大学 | Three-level inverter driving permanent magnet synchronous motor system finite set prediction control method |
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Patent Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO2021044325A1 (en) * | 2019-09-05 | 2021-03-11 | Politecnico Di Torino | Method for sensorless estimating rotor position and rotor angular speed of a synchronous reluctance machine |
| CN112564567A (en) * | 2020-12-09 | 2021-03-26 | 天津工业大学 | Three-level inverter driving permanent magnet synchronous motor system finite set prediction control method |
Non-Patent Citations (1)
| Title |
|---|
| 郑诗程;胡青松;彭勃;: "T型三电平拓扑及其中点电位平衡控制策略", 电力***及其自动化学报, no. 12, 15 December 2017 (2017-12-15) * |
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