CN112039409B - Low switching frequency control method and system for direct current bias type sine current motor system - Google Patents

Low switching frequency control method and system for direct current bias type sine current motor system Download PDF

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CN112039409B
CN112039409B CN202010967007.6A CN202010967007A CN112039409B CN 112039409 B CN112039409 B CN 112039409B CN 202010967007 A CN202010967007 A CN 202010967007A CN 112039409 B CN112039409 B CN 112039409B
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inverter
voltage
voltage vector
value
action time
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CN112039409A (en
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孔武斌
于子翔
曲荣海
李大伟
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Huazhong University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H02M7/53876—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output based on synthesising a desired voltage vector via the selection of appropriate fundamental voltage vectors, and corresponding dwelling times
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention discloses a low switching frequency control method and system for a direct current bias type sine current motor system, and belongs to the field of alternating current motor driving and controlling. In the output voltage vector of one inverter in the application, u is adopted in turn0(000) Or u7(111) One of the zero voltage vectors; correspondingly, in another inverter output voltage vector, u is adopted in turn0(000) Or u7(111) On the basis of one of the zero voltage vectors, the insertion part u7(111) Or u0(111) Vector, which realizes the positive output zero sequence voltage of the whole open winding inverter, thereby reducing the action times of a switching tube in each switching period, reducing the switching loss of the inverter and improving the operation efficiency of the whole driving system on the premise of ensuring the normal speed regulation operation of the motor; the distribution mode of complementation is set for the action time of two zero voltage vectors in an odd sector and an even sector, so that the condition that the upper and lower switch tubes of the same phase bridge arm bear voltage stress in turn is ensured, and the service life of the whole inverter is prolonged.

Description

Low switching frequency control method and system for direct current bias type sine current motor system
Technical Field
The invention belongs to the field of alternating current motor driving and control, and particularly relates to a low switching frequency control method and system for a direct current bias type sine current motor system.
Background
The direct current bias type sine current motor adopts sine current with direct current bias to drive the motor to operate, as shown in figure 1. The rotor magnetic flux in the motor can be flexibly changed by changing the magnitude of the direct current bias current, so that the speed regulation range can be widened, the torque density can be improved, and the motor is particularly suitable for working occasions requiring frequent speed regulation.
The topology of such a motor system is shown in fig. 2. Compare in traditional three-phase motor, the neutral point is opened with DC offset type sinusoidal current motor, and two three-phase inverters are connected respectively to three-phase winding's both ends. In order to provide a path for direct current bias current, the two three-phase inverters use a common direct current bus connection mode. The inverter structure has the characteristics of high voltage utilization rate, strong fault-tolerant capability and the like, and has wide prospects in application occasions such as aviation starting/power generation, automobile turbocharging, mining machinery and the like.
However, compared with the traditional three-phase motor system, the number of switching tubes of the driving system is doubled, the total switching times in each switching period is 12 times when the traditional control strategy is used, the switching loss is inevitably increased, and the operation efficiency of the whole system is influenced. Therefore, a proper control strategy needs to be selected, and the switching frequency of the inverter is reduced on the basis of ensuring the sine current with direct current bias output by the motor, so that the practical process of the inverter is promoted.
Disclosure of Invention
Aiming at the defects or improvement requirements of the prior art, the invention provides a low switching frequency control method and system for a direct current bias type sine current motor system, and aims to reduce the switching frequency of an inverter and improve the operation efficiency of the whole system on the basis of ensuring the sine current with direct current bias output by a motor.
To achieve the above object, according to one aspect of the present invention, there is provided a low switching frequency control method for a dc-biased sinusoidal current motor system, comprising:
s1, according to a voltage vector given value u under a three-phase static coordinate systema *、ub *、uc *Located sector VSectorAnd 0 axis voltage set value u0 *To obtain the zero-voltage vector action time T of the inverter 1z1Zero voltage vector action time T of inverter 2z2；VSector＝{1,2,3,4,5,6}；
If the given voltage vector value is in the odd sector {1,3,5}, Tz1And Tz2Comprises the following steps:
Figure BDA0002682712790000021
if the voltage vector setpoint is located in even sector {2,4,6}, Tz1And Tz2Comprises the following steps:
Figure BDA0002682712790000022
the inverter 1 and the inverter 2 share a direct current bus and are respectively connected to two ends of a three-phase winding of the direct current bias type sinusoidal current motor; u. ofdcRepresents the dc bus voltage;
s2, utilizing a voltage vector given value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure BDA0002682712790000023
Figure BDA0002682712790000024
obtaining the independent voltage vector given value u of the inverter 1α1*、uβ1Independent voltage vector given value u of sum inverter 2α2*、uβ2*；
S3, according to the independent voltage vector given value u of the inverter iαi*、uβiSector VSectoriAnd zero voltage vector action time T of inverter iziDistributing the action time T of two adjacent voltage vectors of the inverter i in each sector according to the average equivalent principle1i，T2iAnd the action time T of 000 voltage vectors0i111 action time T of the Voltage vector7i(ii) a Wherein i is 1,2, VSectori＝{I,II,III,IV,V,VI}；T0i、T7iThe action time of (A) is respectively as follows:
Figure BDA0002682712790000031
Figure BDA0002682712790000032
wherein, TsRepresents a switching cycle;
s4, according to T1i、T2i、T0i、T7iObtaining the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverter iAi、TBi、TCiThe current is acted on a phase winding of a direct current bias type sine current motor to control A, B, C phase current of the direct current bias type sine current motor winding, and the current of d, q and 0 axes is tracked without difference.
Further, the given value u of the voltage vector under the three-phase static coordinate systema *、ub *、uc *Comprises the following steps:
Figure BDA0002682712790000033
wherein u isα *、uβ *And setting a voltage vector under a two-phase static coordinate system.
Further, uα *、uβ *The acquisition process comprises the following steps:
01. collecting rotor mechanical angle signal thetarWhen n is 9.55. d.theta.rThe feedback value n of the motor speed is obtained by the calculation of/dt according to thetae＝θr/npCalculating to obtain the electric angle theta of the motor rotoreWherein n ispThe number of pole pairs of a direct current bias type sine current motor is the motor;
02. using electric angle signals theta of motor rotoreFor three-phase current signal ia、ib、icPerforming stationary-rotating coordinate transformation to obtain d-axis, q-axis and 0-axis current feedback values i in a synchronous rotating coordinate systemd、iq、i0；
03. Regulating given value n of motor speed*Difference value with motor speed feedback value n to make n always follow n*Is changed;
04. given value i of q-axis currentq *Performing multiplication to obtain a given value of 0-axis current
Figure BDA0002682712790000041
Let d-axis current set value id *＝0；
05. According to d-axis current given value id *And d-axis current feedback value idTo obtain a d-axis voltage given value ud *By adjusting ud *So that idAlways follow id *Is changed; according to the given value i of q-axis currentq *And q-axis current feedback value iqTo obtain a given value u of the q-axis voltageq *By adjusting uq *So that iqAlways follow iq *Is changed; according to the given value i of the 0-axis current0 *And 0 axis current feedback value i0To obtain a given value u of the 0-axis voltage0 *By adjusting u0 *So that i0Always follow i0 *Is changed;
06. d-axis voltage given value ud *Q-axis voltage set value uq *Rotor position electrical angle signal thetaeRotating-static coordinate transformation is carried out to obtain a voltage vector given value u under a two-phase static coordinate systemα *、uβ *。
Furthermore, in the inverter i, the action time T of two adjacent voltage vectors in different sectors1i，T2iComprises the following steps:
Figure BDA0002682712790000042
furthermore, the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverterAi、TBi、TCiComprises the following steps:
Figure BDA0002682712790000043
Figure BDA0002682712790000051
Figure BDA0002682712790000052
according to another aspect of the present invention, there is provided a low switching frequency control system for a dc biased sinusoidal current motor system, comprising: the device comprises a zero sequence voltage action time setting module, a first voltage distribution module, a second voltage distribution module, a first pulse width modulation module and a second pulse width modulation module;
the zero sequence voltage action time setting module is used for setting a value u according to a voltage vector under a three-phase static coordinate systema *、ub *、uc *Located sector VSectorAnd 0 axis voltage set value u0 *To obtain the zero-voltage vector action time T of the inverter 1z1Zero voltage vector action time T of inverter 2z2；VSector＝{1,2,3,4,5,6}；
If the given voltage vector value is in the odd sector {1,3,5}, Tz1And Tz2Comprises the following steps:
Figure BDA0002682712790000053
if the voltage vector setpoint is located in even sector {2,4,6}, Tz1And Tz2Comprises the following steps:
Figure BDA0002682712790000054
the inverter 1 and the inverter 2 share a direct current bus and are respectively connected to two ends of a three-phase winding of the direct current bias type sinusoidal current motor; u. ofdcRepresents the dc bus voltage;
the first voltage distribution module is used for utilizing a given voltage vector value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure BDA0002682712790000061
obtaining the independent voltage vector given value u of the inverter 1α1*、uβ1*；
The second voltage distribution module is used for utilizing a given voltage vector value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure BDA0002682712790000062
obtaining the independent voltage vector given value u of the inverter 2α2*、uβ2*；
The first pulse width modulation module and the second pulse width modulation module are used for setting a value u according to the independent voltage vector of the inverter iαi*、uβiSector VSectoriAnd zero voltage vector action time T of inverterziDistributing the action time T of two adjacent voltage vectors in each sector corresponding to the inverter i according to the average value equivalence principle1i，T2iAnd the action time T of 000 voltage vectors0i111 action time T of the Voltage vector7iAnd according to T1i、T2i、T0i、T7iObtain inverter correspondenceThe action time T of PWM signals of A phase, B phase and C phaseAi、TBi、TCiThe current is acted on a phase winding of a direct current bias type sine current motor to control A, B, C phase current of the direct current bias type sine current motor winding, and the current of d, q and 0 axes is tracked without difference;
wherein i is 1,2, VSectori＝{I,II,III,IV,V,VI}；T0i、T7iThe action time of (A) is respectively as follows:
Figure BDA0002682712790000063
Figure BDA0002682712790000064
wherein, TsIndicating the switching period.
Furthermore, the action time T of PWM signals of A phase, B phase and C phase corresponding to the inverter iAi、TBi、TCiComprises the following steps:
Figure BDA0002682712790000071
Figure BDA0002682712790000072
Figure BDA0002682712790000073
in general, the above technical solutions contemplated by the present invention can achieve the following advantageous effects compared to the prior art.
(1) Compared with the existing control strategy, in the output voltage vector of one inverter, u is adopted in turn0(000) Or u7(111) One of the zero voltage vectors; correspondingly, in another inverter output voltage vector, u is adopted in turn0(000) Or u7(111) On the basis of one of the zero voltage vectors, the insertion part u7(111) Or u0(111) Vector, the output zero sequence voltage of the whole open winding inverter is positive, so that the action times of a switching tube in each switching period are reduced, the switching loss of the inverter is reduced, and the operation efficiency of the whole driving system is improved on the premise of ensuring the normal speed regulation operation of the motor.
(2) The distribution mode of complementation is set for the action time of two zero voltage vectors in an odd sector and an even sector, so that the condition that the upper and lower switch tubes of the same phase bridge arm bear voltage stress in turn is ensured, and the service life of the whole inverter is prolonged.
Drawings
FIG. 1 is a phase current waveform for normal operation of an open-winding DC-biased sinusoidal current motor system;
FIG. 2 is a topological structure diagram of an open-winding DC-biased sinusoidal current motor system;
FIG. 3 is a schematic block diagram of a control strategy implemented in accordance with the present invention;
fig. 4(a) shows the resultant voltage vector sector distribution of the entire open-winding inverter, and fig. 4(b) shows the independent voltage vector sector distribution of the inverters 1 and 2.
Fig. 5(a) shows a case where the voltage vector set value is located in an odd-numbered sector, and fig. 5(b) shows a switching sequence where the voltage vector set value is located in an even-numbered sector.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
In order to implement the control method of the present invention, the control method for controlling the dc offset type sinusoidal current motor system according to the control block diagram shown in fig. 3 includes the following steps:
(1) rotor mechanical angle signal theta obtained by motor position sensorrCalculating the rotating speed feedback value and the rotor electrical angle to obtain a motor rotating speed feedback value n and a motor rotor electrical angle thetae. The motor rotating speed feedback value calculation process is represented as follows: n is 9.55 d thetarAnd/dt. The process of calculating the electrical angle of the motor rotor is represented as follows: thetae＝θr/npWherein n ispIs the pole pair number of the DC offset sine current motor.
(2) The electric angle signal theta of the motor rotor is measuredeThree-phase current signal i obtained by sampling three-phase current Hall sensora、ib、icSending the current feedback values to a static/rotating coordinate transformation module to obtain d-axis, q-axis and 0-axis current feedback values i under a synchronous rotating coordinate systemd、iq、i0The transformation process is as follows:
Figure BDA0002682712790000081
(3) setting the motor speed to a given value n*The difference value of the feedback value n and the motor speed is sent to a motor speed regulator, and the motor speed regulator outputs a q-axis current given value iq *. If n is*If the difference from n is positive, i is increasedq *Until the difference is 0; if n is*If the difference from n is negative, i is decreasedq *Until the difference is 0. By adjusting iq *So that n always follows n*May vary.
(4) Given value i of q-axis currentq *Performing multiplication to obtain a given value of 0-axis current
Figure BDA0002682712790000091
d-axis current set value id *＝0。
(5) D-axis current is set to a given value id *And d-axis current feedback value idIs fed into a d-axis current regulator (2a), said d-axisD-axis voltage set value u output by current regulatord *. If id *And idIf the difference of (d) is positive, u is increasedd *Until the difference is 0; if id *And idIf the difference is negative, u is decreasedd *Until the difference is 0. By adjusting ud *So that idAlways follow id *Is changed;
setting q-axis current to be a given value iq *And q-axis current feedback value iqIs fed into a q-axis current regulator (2b) which outputs a q-axis voltage setpoint value uq *. If iq *And iqIf the difference of (d) is positive, u is increasedq *Until the difference is 0; if iq *And iqIf the difference is negative, u is decreasedq *Until the difference is 0. By adjusting uq *So that iqAlways follow iq *Is changed;
setting the shaft current to 0 to a given value i0 *And 0 axis current feedback value i0Is fed to a 0-axis current regulator (2c), which outputs a 0-axis voltage setpoint u0 *. If i0 *And i0If the difference of (d) is positive, u is increased0 *Until the difference is 0; if i0 *And i0If the difference is negative, u is decreased0 *Until the difference is 0. By adjusting u0 *So that i0Always follow i0 *Is changed;
(6) d-axis voltage given value ud *Q-axis voltage set value uq *Rotor position electrical angle signal thetaeSending the voltage vector to a rotating-static coordinate transformation module to obtain a voltage vector given value u under a two-phase static coordinate systemα *、uβ *The process is as follows:
Figure BDA0002682712790000092
(7) setting the voltage vector under the two-phase static coordinate system to be a given value uα *、uβ *Sending the voltage to a zero sequence voltage action time setting module which outputs zero voltage vector action time T of the inverter 1 and the inverter 2z1And Tz2The method comprises the following steps:
judgment is made byα *、uβ *Sector V in which the given value of the formed composite voltage vector is locatedSectorAnd the sector V in which the given value of the independent voltage vector of each of the inverter 1 and the inverter 2 is locatedSector1、VSector2. As shown in FIG. 4(a), the resultant voltage vector of the entire open-winding inverter can be spatially divided into 6 sectors V Sector1,2,3,4,5, 6. As shown in fig. 4(b), the given value of the independent voltage vector of the inverter i can be divided into 6 sectors V in spaceSectori＝{I,II,III,IV,V,VI}。
Converting the given voltage vector value under the two-phase static coordinate system according to the following equation to obtain the given voltage vector value u under the three-phase static coordinate systema *、ub *、uc *：
Figure BDA0002682712790000101
The formula has simple calculation process and accurate calculation result.
Further, u is judged from Table Ia *、ub *、uc *Obtaining the sector V where the given value of the synthetic voltage vector is locatedSectorAnd the sector V where the independent voltage vector given value of the inverter 1 and the inverter 2 is locatedSector1、VSector2。
TABLE I sector decision rule for composite and independent voltage vectors
V Sector 1 2 3 4 5 6
VSector1 I II III IV V VI
VSector2 III IV V VI I II
ua * ≥0 ≥0 <0 <0 <0 ≥0
ub * <0 ≥0 ≥0 ≥0 <0 <0
uc * <0 <0 <0 ≥0 ≥0 ≥0
According to the sector V where the given value of the synthesized voltage vector is locatedSectorGiven value u of axis voltage 00 *Respectively calculating the zero voltage vector action time T of the inverter 1 and the inverter 2z1、Tz2The method comprises the following steps:
if the given value of the resultant voltage vector is in an odd sector (V)Sector1,3,5) according to u0 *Positive and negative of (1), output Tz1And Tz2：
Figure BDA0002682712790000102
Wherein, TsIndicating the switching period. If the given value of the resultant voltage vector is in an even number of sectors (V)Sector2,4,6) according to u0 *Positive and negative of (1), output Tz1And Tz2：
Figure BDA0002682712790000111
By inserting the action time T of the zero-voltage vectorz1And Tz2The method can realize flexible adjustment of zero sequence voltage, further realize closed-loop control of zero sequence current and ensure stable operation of the direct current offset type sine motor.
(8) The given value u of the synthetic voltage vector under the two-phase static coordinate systemα *、uβ *Respectively sent to a first voltage distribution module and a second voltage distribution module, wherein the first voltage distribution module outputs a voltage vector given value u of the inverter 1α1*、uβ1The second voltage distribution module outputs a given voltage vector value u of the inverter 2α2*、uβ2*. The process is represented as:
Figure BDA0002682712790000112
Figure BDA0002682712790000113
by the above operation, the phase lead u can be obtainedαSum of uβGiven value u of output voltage vector of inverter 1 with composite voltage vector of 30 degrees and size of composite voltage vector √ 3/3 timesα1*、uβ1A first step of; phase lead of uαSum of uβGiven value u of output voltage vector of inverter 2 with composite voltage vector of 150 degrees and size of √ 3/3 times of composite voltage vectoriα2*、uiβ2As shown in fig. 4(a) -4 (b). And further avoid the influence of 3-time voltage harmonic waves generated by PWM modulation on the operation of the motor.
(9) Setting the voltage vector of the inverter 1 to a given value uα1*、uβ1Sector V where independent voltage vector given value of inverter 1 is locatedSector1And an inverter1 zero voltage vector action time Tz1Sending the voltage vector to a first pulse width modulation module to set the voltage vector of the inverter 2 to a given value uα2*、uβ2Sector V where independent voltage vector given value of inverter 2 is locatedSector2And zero voltage vector action time T of inverter 2z2And sending the pulse signals to a second pulse width modulation module. The first and second PWM modules respectively output the action time T of the A-phase, B-phase and C-phase PWM signals corresponding to the inverter 1A1、TB1、TC1And the operating time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverter 2A2、TB2、TC2The method comprises the following steps:
distributing the action time T of two adjacent voltage vectors in each sector of the inverter i according to the average value equivalence principle1i，T2iAnd (000) the action time T of the voltage vector0iAction time T of (111) Voltage vector7i. The procedure is shown in Table II. Wherein, Xi,Yi,ZiIs an intermediate variable, which is defined by the formula:
Figure BDA0002682712790000121
TABLE II calculation of the action time of each voltage vector in different sectors
VSectori I II III IV V VI
T1i -Zi Zi Xi -Xi -Yi Yi
T2i Xi Yi -Yi Zi -Zi -Xi
T0i Tzi Ts-T1i-T2i-Tzi Tzi Ts-T1i-T2i-Tzi Tzi Ts-T1i-T2i-Tzi
T7i Ts-T1i-T2i-Tzi Tzi Ts-T1i-T2i-Tzi Tzi Ts-T1i-T2i-Tzi Tzi
In the same sector, the sum of action time of two non-zero voltage vectors and two zero voltage vectors is equal to Ts. V corresponding to the inverter i in the motor operation processSectori(II) duration of action T of the (000) voltage vector obtained according to Table II, continuously switched between { I to VI }0iTime of action T with (111) voltage vector7iIn the sector switching process, the action time of two zero voltage vectors in the odd-numbered sector and the action time of two zero voltage vectors in the even-numbered sector both show a complementary conversion relation. The conversion relation ensures that the upper and lower switching tubes of the same phase bridge arm share complementary voltage stress, so that the service life of the whole inverter is prolonged.
According to the table III, the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverter 1 is finally obtainedA1、TB1、TC1(ii) a And the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverter 2A2、TB2、TC2. The converter generates output voltage proportional to the action time of the PWM signal, acts on a phase winding of the direct current bias type sinusoidal current motor, controls the current of A, B, C phases of the direct current bias type sinusoidal current motor winding, and realizes the non-difference tracking of d, q and 0 axis currents.
TABLE III PWM Signal action time calculation for inverter 1, inverter 2
VSectori I II III IV V VI
TAi T1i+T2i+T7i T2i+T7i T7i T7i T2i+T7i T1i+T2i+T7i
TBi T2i+T7i T1i+T2i+T7i T1i+T2i+T7i T2i+T7i T7i T7i
TCi T7i T7i T2i+T7i T1i+T2i+T7i T1i+T2i+T7i T2i+T7i
FIG. 5(a) shows that the given value of the voltage vector is located in the odd sector (V)Sector1,3, 5). Eight basic voltage vectors output by the inverter 1 and the inverter 2 are defined as follows: u. of0(000)，u1(100)，u2(110)，u3(010)，u4(011)，u5(001)，u6(101)，u7(111). Wherein the two zero voltage vectors are respectively u0(000) And u7(111) And the remaining six are non-zero voltage vectors. The sequence of the voltage vector action of the inverter 1 in the figure is as follows: u. of1-u2-u7-u7-u2-u1. The sequence of the voltage vector action of the inverter 2 is as follows: u. of0-u5-u6-u7-u7-u6-u5-u0. Of the output voltage vectors of the inverter 1, only u is present in the zero voltage vector7(111) No use of u0(000). And the inverter 2 introduces a part u0And vector, so that the output zero sequence voltage of the whole open-winding inverter is positive. In each switching period, the action times of the switching tube are 10 times, and are reduced by 2 times compared with the traditional control strategy.
FIG. 5(b) shows that the given voltage vector value is located in the even sector (V)Sector2,4, 6). The sequence of the voltage vector action of the inverter 1 in the figure is as follows: u. of0-u3-u2-u7-u7-u2-u3-u0. The sequence of the voltage vector action of the inverter 2 is as follows: u. of0-u1-u6-u6-u1-u0. Of the output voltage vectors of the inverter 2, only u is present in the zero voltage vector0(000) No use of u7(111). While the inverter 1 introduces the portion u7(111) And vector, so that the output zero sequence voltage of the whole open-winding inverter is positive. Also in each switching period, the number of switching tube actions is 10, which is reduced by 2 compared with the traditional control strategy.
Because the odd sectors and the even sectors adopt a complementary zero voltage vector distribution mode, the voltage stress borne by the upper and lower switching tubes of the same phase bridge arm is symmetrical. The service life of the inverter switch tube cannot be influenced by changing the distribution mode of the zero voltage vector.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.

Claims (7)

1. A low switching frequency control method for a direct current bias type sine current motor system is characterized by comprising the following steps:
s1, according to a voltage vector given value u under a three-phase static coordinate systema *、ub *、uc *Located sector VSectorAnd 0 axis voltage set value u0 *To obtain the zero-voltage vector action time T of the inverter 1z1Zero voltage vector action time T of inverter 2z2；VSector＝{1,2,3,4,5,6}；
If the given voltage vector value is in the odd sector {1,3,5}, Tz1And Tz2Comprises the following steps:
Figure FDA0003334516290000011
if the voltage vector setpoint is located in even sector {2,4,6}, Tz1And Tz2Comprises the following steps:
Figure FDA0003334516290000012
the inverter 1 and the inverter 2 share a direct current bus and are respectively connected to two ends of a three-phase winding of the direct current bias type sinusoidal current motor; u. ofdcRepresents the dc bus voltage;
s2, utilizing a voltage vector given value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure FDA0003334516290000013
Figure FDA0003334516290000014
obtaining the independent voltage vector given value u of the inverter 1α1*、uβ1Independent voltage vector given value u of sum inverter 2α2*、uβ2*；
S3, according to the independent voltage vector given value u of the inverter iαi*、uβiSector VSectoriAnd zero voltage vector action time T of inverter iziDistributing the action time T of two adjacent voltage vectors of the inverter i in each sector according to the average equivalent principle1i，T2iAnd the action time T of 000 voltage vectors0i111 action time T of the Voltage vector7i(ii) a Wherein i is 1,2, VSectori＝{I,II,III,IV,V,VI}；T0i、T7iThe action time of (A) is respectively as follows:
Figure FDA0003334516290000021
Figure FDA0003334516290000022
wherein, TsRepresents a switching cycle;
s4, according to T1i、T2i、T0i、T7iObtaining the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverter iAi、TBi、TCiThe current is acted on a phase winding of a direct current bias type sine current motor to control A, B, C phase current of the direct current bias type sine current motor winding, and the current of d, q and 0 axes is tracked without difference.
2. The method as claimed in claim 1, wherein the voltage vector set value u is determined by a three-phase stationary coordinate systema *、ub *、uc *Comprises the following steps:
Figure FDA0003334516290000023
wherein u isα *、uβ *And setting a voltage vector under a two-phase static coordinate system.
3. A method according to claim 1 or 2, wherein u is the low switching frequency control of a dc-biased sinusoidal current motor systemα *、uβ *The acquisition process comprises the following steps:
01. collecting rotor mechanical angle signal thetarWhen n is 9.55. d.theta.rThe feedback value n of the motor speed is obtained by the calculation of/dt according to thetae＝θr/npCalculating to obtain the electric angle theta of the motor rotoreWherein n ispThe number of pole pairs of a direct current bias type sine current motor is the motor;
02. using electric angle signals theta of motor rotoreFor three-phase current signal ia、ib、icAnd then the static-rotation coordinate transformation is carried out,obtaining d-axis, q-axis and 0-axis current feedback values i under a synchronous rotating coordinate systemd、iq、i0；
03. Regulating given value n of motor speed*Difference value with motor speed feedback value n to make n always follow n*Is changed;
04. given value i of q-axis currentq *Performing multiplication to obtain a given value of 0-axis current
Figure FDA0003334516290000031
Let d-axis current set value id *＝0；
05. According to d-axis current given value id *And d-axis current feedback value idTo obtain a d-axis voltage given value ud *By adjusting ud *So that idAlways follow id *Is changed; according to the given value i of q-axis currentq *And q-axis current feedback value iqTo obtain a given value u of the q-axis voltageq *By adjusting uq *So that iqAlways follow iq *Is changed; according to the given value i of the 0-axis current0 *And 0 axis current feedback value i0To obtain a given value u of the 0-axis voltage0 *By adjusting u0 *So that i0Always follow i0 *Is changed;
06. d-axis voltage given value ud *Q-axis voltage set value uq *Rotor position electrical angle signal thetaeRotating-static coordinate transformation is carried out to obtain a voltage vector given value u under a two-phase static coordinate systemα *、uβ *。
4. The method as claimed in claim 1, wherein the action time T of two adjacent voltage vectors in different sectors in inverter i is determined by the control of the switching frequency of the DC offset sinusoidal current motor system1i，T2iComprises the following steps:
Figure FDA0003334516290000032
Xi,Yi,Ziis an intermediate variable, which is defined by the formula:
Figure FDA0003334516290000041
5. the method as claimed in claim 1, wherein the inverter has PWM signal action time T of A, B and C phasesAi、TBi、TCiComprises the following steps:
Figure FDA0003334516290000042
Figure FDA0003334516290000043
Figure FDA0003334516290000044
6. a low switching frequency control system for a dc-biased sinusoidal current motor system, comprising: the device comprises a zero sequence voltage action time setting module, a first voltage distribution module, a second voltage distribution module, a first pulse width modulation module and a second pulse width modulation module;
the zero sequence voltage action time setting module is used for setting a value u according to a voltage vector under a three-phase static coordinate systema *、ub *、uc *Located sector VSectorAnd 0 axis voltage set value u0 *To obtain the zero-voltage vector action time T of the inverter 1z1Zero voltage vector action time T of inverter 2z2；VSector＝{1,2,3,4,5,6}；
If the given voltage vector value is in the odd sector {1,3,5}, Tz1And Tz2Comprises the following steps:
Figure FDA0003334516290000051
if the voltage vector setpoint is located in even sector {2,4,6}, Tz1And Tz2Comprises the following steps:
Figure FDA0003334516290000052
the inverter 1 and the inverter 2 share a direct current bus and are respectively connected to two ends of a three-phase winding of the direct current bias type sinusoidal current motor; u. ofdcRepresents the dc bus voltage;
the first voltage distribution module is used for utilizing a given voltage vector value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure FDA0003334516290000053
obtaining the independent voltage vector given value u of the inverter 1α1*、uβ1*；
The second voltage distribution module is used for utilizing a given voltage vector value u under a two-phase static coordinate systemα *、uβ *According to the formula:
Figure FDA0003334516290000054
obtaining the independent voltage vector given value u of the inverter 2α2*、uβ2*；
The first pulse width modulation module and the second pulse width modulation module are used for setting a value u according to the independent voltage vector of the inverter iαi*、uβiSector VSectoriAnd zero voltage vector action time T of inverterziDistributing the action time T of two adjacent voltage vectors in each sector corresponding to the inverter i according to the average value equivalence principle1i，T2iAnd the action time T of 000 voltage vectors0i111 action time T of the Voltage vector7iAnd according to T1i、T2i、T0i、T7iObtaining the action time T of the PWM signals of the A phase, the B phase and the C phase corresponding to the inverterAi、TBi、TCiThe current is acted on a phase winding of a direct current bias type sine current motor to control A, B, C phase current of the direct current bias type sine current motor winding, and the current of d, q and 0 axes is tracked without difference;
wherein i is 1,2, VSectori＝{I,II,III,IV,V,VI}；T0i、T7iThe action time of (A) is respectively as follows:
Figure FDA0003334516290000061
Figure FDA0003334516290000062
wherein, TsIndicating the switching period.
7. The system of claim 6, wherein the inverter i has PWM signal action time T corresponding to phase A, B and CAi、TBi、TCiComprises the following steps:
Figure FDA0003334516290000063
Figure FDA0003334516290000064
Figure FDA0003334516290000065
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