CN113241981A - 一种多相容错型磁通切换永磁电机反步滑模控制方法 - Google Patents
一种多相容错型磁通切换永磁电机反步滑模控制方法 Download PDFInfo
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Abstract
本发明公开了一种多相容错型磁通切换永磁电机反步滑模控制方法,反步控制器根据自适应滑模控制器的输出控制量、电机角速度差值、输入的d轴参考电流、电机实际解耦合后的d‑q轴电流、电压将d‑q轴电压ud、uq经过静止坐标系转换后,得到参考电压矢量分量;根据参考电压矢量分量驱动电机运行。本发明设计了根据反步控制率计算的自适应滑模控制器的输出控制量和虚拟控制变量,以跟踪给定的速度,最小化电流静态误差,提高了***的动态响应和抑制外界干扰的能力;把电机自身的参数变化和外界干扰看作是***的未知项,采用滑模自适应率逼近未知项,提高了电机***的鲁棒性和控制性能。
Description
技术领域
本发明涉及同步电机技术领域,具体涉及一种同步电机的励磁控制装置及使用方法。
背景技术
随着稀土永磁材料技术的进步以及在电机制造中的应用,电机制造越来越先进,各种电机层出不穷,永磁电机由于其性能优越,被广泛应用于各行各业。近年来发展起来的磁通切换永磁(Flux-switching permanent–magnet,FSPM)电机,由于其永磁体安装于定子上,易于冷却,降低永磁体过热所引起的退磁风险,FSPM电机不仅具有转子永磁电机那样的高效率和高功率密度,而且转子结构简单,机械完整性好,易于磁体散热,克服了传统无刷电机转子内有磁体的缺点,适合高速运行,具有广阔的发展前景。在某些特殊场合,电机的高可靠性显得越来越重要。目前,三相电动机在各个行业都有着积极的作用。但仍存在一些不足,三相永磁电机在故障情况下无法继续运行,整个电机***的可靠性无法保证。相数的增加使多相电机可以提供比三相电机更多的控制自由度,采用全桥驱动,可以提高控制性能,减小转矩脉动的幅值,实现低压大功率。多相永磁电机具有效率高、密度高、相数冗余等优点,广泛应用于船舶推进、风力发电、电动汽车、航空航天和军事装备等领域。同时,逆变技术的发展也使多相逆变***的实现成为可能。对于多相电机,通过增加容错齿,相数冗余保证了电机在发生一相或两相故障的情况下,实现无故障运行和高可靠性。
为了使FSPM电机获得理想的性能,大多数控制策略都是基于***的精确数学模型,但由于永磁电机参数不确定、负载变化、摩擦非线性、外界干扰、强耦合等原因,是一个强非线性***,建模困难,控制性能得不到保证。因此,线性控制方案很难获得高性能。随着制造技术和计算机处理速度的进步,各种先进的智能算法被引入调节电机***。与其他方法相比,滑模方法对***干扰和参数变化具有较高的鲁棒性,但滑模面存在抖振,这是需要解决的问题,在永磁电机的滑模控制中,一类是针对***参数变化的不确定性,采用低通滤波器和转子位置的补偿来减少抖振,切换函数通常为符号函数、饱和函数或者S函数,采用非线性、高阶终端滑模、非奇异滑模等,并进行干扰补偿,提高***的动态性能和追踪精度,并对干扰观测器进行前馈补偿。另一类是采用无位置传感器技术,通过高速滑模观测器,由反电动势来估计转子位置、角速度,或者采用迭代滑模算法估计转子的速度和位置。为了提高鲁棒性和跟踪精度,提出了自适应反步观测器和积分器,但也存在静态误差和较大的速度超调,反步法与滑模控制相结合已成为不确定非线性***研究的热点之一。
发明内容
本发明提供了一种多相容错型磁通切换永磁电机反步滑模控制方法,以解决现有技术中采用传统矢量控制中存在的速度响应慢、跟随性能低的技术问题。
本发明提供了一种多相容错型磁通切换永磁电机反步滑模控制方法,包括如下步骤:
步骤1:采集多相定子电流,根据多相定子电流获取电机实际解耦合后的d-q轴电流,作为反步控制器的输入量;
步骤2:采集永磁电机的电机实时位置角度,根据电机实时位置角度计算电机实时角速度,将电机给定角速度与电机实时角速度的电机角速度差值作为反步控制器的输入量;
将电机实时角速度、电机实时位置角度、电机期望位置角度、永磁电机的扰动量作为自适应滑模控制器的输入量,自适应滑模控制器根据输入量形成输出控制量,作为反步控制器的输入;
步骤3:反步控制器根据自适应滑模控制器的输出控制量、电机角速度差值、输入的d轴参考电流、电机实际解耦合后的d-q轴电流,产生d-q轴电压ud、uq;
步骤4:将d-q轴电压ud、uq经过静止坐标系转换后,得到参考电压矢量分量;
步骤5:根据参考电压矢量分量驱动电机运行。
进一步地,所述步骤1中根据多相定子电流获取电机实际耦合后的d-q轴电流,具体为:
多相定子电流经多相/两相静止坐标变换后得到α-β轴电流分量iα、iβ;再经两相静止/两相旋转坐标系变换,得到电机实际的解耦后d-q轴电流iq和id;
所述步骤4中将d-q轴电压ud、uq经过两相旋转/两相静止坐标系转换后,得到参考电压矢量分量uα、uβ。
进一步地,所述永磁电机的扰动量包括:电感、磁场以及电流变化引起的扰动A的变化量ΔA,
其中,J为转动惯量、ψm为永磁链、Lmd、Lmq分别为d和q定子电感;
摩擦系数变化量ΔB;电机负载干扰引起转矩的变化量ΔTl。
进一步地,所述步骤2中自适应滑模控制器根据输入量形成的输出控制量,具体公式如下:
其中,ω为转子角速度;Tl为负载转矩;J为转动惯量。
进一步地,所述虚拟控制量x2为:
进一步地,所述步骤5中将参考电压矢量分量经过2/5变换后输入SVPWM调制模块,产生调制波发送给多相逆变器,驱动电机运行。
进一步地,所述步骤3中输入的d轴参考电流id *=0。
本发明的有益效果:
针对多相容错型磁通切换永磁电机,提出了一种新的基于反步方法的滑模控制策略(B-SMC),设计了根据反步控制率计算的自适应滑模控制器的输出控制量u和虚拟控制变量x2,以跟踪给定的速度,最小化电流静态误差,提高了***的动态响应和抑制外界干扰的能力。把电机自身的参数变化和外界干扰看作是***的未知项,采用滑模自适应率逼近未知项,提高了电机***的鲁棒性和控制性能。
附图说明
通过参考附图会更加清楚的理解本发明的特征和优点,附图是示意性的而不应理解为对本发明进行任何限制,在附图中:
图1为针对FT-FSPM的五相容错型磁通切换永磁电机反步滑模控制方法***框图;
图2为FT-FSPM电机绕组结构图;
图3为负载从5N.m升为7.5N.m时的波形图;
图4为负载从5N.m降为2.5N.m时的波形图;
图5为速度突增和突降时的跟踪波形图;
图6为速度突增和突降时的转子位置跟踪波形图;
图7为电流id和iq测量波形图;
图8为转子位置角测量波形图。
具体实施方式
为使本发明实施例的目的、技术方案和优点更加清楚,下面将结合本发明实施例中的附图,对本发明实施例中的技术方案进行清楚、完整地描述,显然,所描述的实施例是本发明一部分实施例,而不是全部的实施例。基于本发明中的实施例,本领域技术人员在没有作出创造性劳动前提下所获得的所有其他实施例,都属于本发明保护的范围。
本发明具体实施例采用如图2所示的五相容错式磁通切换永磁电动机(FT-FSPM)为例,如图1所示,针对FT-FSPM的五相容错型磁通切换永磁电机反步滑模控制方法,包括如下步骤:
步骤S1:FT-FSPM驱动采用的是id=0的磁场定向策略,根据空间矢量控制原理,五相逆变器的五相定子电流ia、ib、ic、id、ie由电流霍尔传感器采集后,经五相/两相静止坐标变换后得到α-β轴电流分量iα、iβ;α-β轴电流分量iα、iβ经两相静止/两相旋转坐标系变换,得到电机实际的解耦后的d-q轴电流iq和id,将d-q轴电流iq和id作为反步控制器的两路输入;
FT-FSPM五相电流表示为:
其中,Im为电流幅值,
通过坐标变换得到简化的FT-FSPM数学模型,实现电机***的降阶、解耦和线性化。这里只考虑电机绕组的基波磁链。通过五相/两相静止坐标变换和两相静止/两相旋转坐标系变换,d-q轴电流id和iq表示为:
步骤S2:运用光电编码器获取FT-FSPM电动机的电机实时位置角度θ,电机实时位置角度θ微分后可以计算出电机实时角速度ω,将电机给定角速度与电机实时角速度的电机角速度差值作为反步控制器的输入量;
在设计反步控制器时,根据反步控制的思想,首先是速度子***,设计第一个Lyapunov函数:并定义虚拟控制项为速度反馈增益,保证速度子***稳定的虚拟控制律。其次,针对电流子***,设计Lyapunov函数的电和压控制律,通过虚拟控制项实现子***的电流跟踪。反步控制律如下
针对电机本身参数的变化和外界干扰,设计了自适应滑模控制器,用于逼近不确定项,在进行自适应反步滑模控制律u设计时,引入第三个Lyapunov函数:并定义x1=θ-θ*为位置角误差,为虚拟控制量,用于设计不确定项的逼近算法。
将电机实时角速度、电机实时位置角度、电机期望位置角度、永磁电机的扰动量作为自适应滑模控制器的输入量,其中永磁电机的扰动量包括:电感、磁场以及电流变化引起的扰动A的变化量ΔA,
其中,J为转动惯量、ψm为永磁链、Lmd、Lmq分别为d和q定子电感;摩擦系数变化量ΔB;电机负载干扰引起转矩的变化量ΔTl,自适应滑模控制器根据输入量形成输出控制量,根据反步滑模控制率,输出控制量计算公式如下:
其中,ω为转子角速度;Tl为负载转矩;J为转动惯量;
为减少滑模抖振,引入一阶线性滑动开关函数η,形式如下:
η=kx1+x2
总不确定度项F自适应律设计为:
自适应滑模控制器的输出u作为反步控制器的输入;
步骤S3:反步控制器根据自适应滑模控制器的输出控制量、输入的d轴参考电流id *=0、步骤1中获取的d-q轴电流,产生d-q轴电压ud、uq;
步骤S4:将d-q轴电压ud、uq经过静止坐标系转换后,得到参考电压矢量分量;
步骤S5:根据参考电压矢量分量驱动电机运行。
所述步骤4中将d-q轴电压ud、uq经过两相旋转/两相静止坐标系转换后,得到参考电压矢量分量uα、uβ。
为了验证发明闭环***的稳定性,设计了三个Lyapunov函数。
式(2)中除了速度项,还有当前项,所以当前控制项的期望值定义为:
为了消除控制中q轴电流误差导数项,使***趋于稳定,定义第二个Lyapunov函数为:
导数为:
为了保证电流环的全局渐近稳定性,将d-q轴控制电压设计为:
式(6)代入式(5),得到
本发明反步控制率u的设计考虑了外界干扰给电机本身参数带来的变化,设计了第三个Lyapunov函数来验证,第三个Lyapunov函数为:
微分得到:
为了验证该算法的有效性,在matlab/simulink中建立了***仿真模型。五相10/19FT-FSPM电动机样机参数如下:额定功率P=1.8KW,相电压U=200V,额定转速n=600rpm,额定转矩Te=22.8N.m,定子电阻rs=2.56Ω,绕组电感Lmd=36mH,Lmq=32mH,转动惯量J=0.00062kg.m2,摩擦系数B=0.00031N.m.s,永磁磁通ψm=0.183Wb。
建议的BSMC参数选择如下:
η=0.3,β=1.5,γ=12,μ=0.1,k1=15,c1=10,h=20。
为了测试控制器对负载扰动变化的鲁棒性,参考速度为ω*=600rmp。在两种不同的情况下进行了仿真。第一种情况,在t=0.2s时,负载转矩Tl从5n.m变为7.5n.m,结果如图3所示,其中图3(a)为速度响应波形、图3(b)为采用SMC方法的电流id和iq波形、图3(c)为采用B-SMC方法的电流id和iq波形、图3(d)五相电流波形、图3(e)电磁转矩响应波形。第二种情况,在t=0.3s时,负载转矩Tl从5n.m变为2.5n.m,结果如图4所示,其中图4(a)为速度响应波形、图4(b)为采用SMC方法的电流id和iq波形、图4(c)为采用B-SMC方法的电流id和iq波形、图4(d)五相电流波形、图4(e)电磁转矩响应波形。
从图3和图4可以看出,在起动瞬间,仿真结果表明,在t=0.01s的时间段内,速度响应可以很快地跟踪参考速度,但SMC滞后0.05s。与SMC相比,B-SMC方法的速度超调可以从10%降到7.5%。滑模控制使电流id和iq在初始阶段产生较大的抖振,但由于电压ud、uq控制律和反步算法的有效性,B-SMC可以减小瞬时抖振。可以看出,d轴电流id与电机转速很好地解耦,并且可以很好地调节到零。Te与q轴电流成正比,与iq变化几乎同步。在电动机起动时,由于电动机位置转矩大,所以存在较大的电磁转矩和电流尖峰。但在正常运行时,电磁转矩脉动较小。五相电流响应轨迹与理论分析一致,呈正弦波,幅值发生相应变化,在0.01s内趋于稳定,在负载转矩突变瞬间,B-SMC的转速、电流和转矩波动均小于SMC,反应速度几乎相同。由此可见,B-SMC方法比SMC方法能更好地实现FT-FSPM电机的运行。
为了测试该算法在变速情况下的速度跟踪性能,在0.25s内,速度分别从600rpm增加到800rpm、600rpm下降到400rpm。如图5和图6所示,图5(a)、图6(a)为速度突增曲线;图5(b)、图6(b)为速度突降曲线。采用本发明方法,响应速度快,跟踪误差小。从转子位置角仿真可以看出,位置角可以反映转速的变化。
为了进一步验证B-SMC的可行性,搭建了FT-FSPM控制实验平台,实验平台包括DSP2812芯片、三菱IPM、2048线光电编码器、霍尔传感器、传递转矩传感器,并以直流电机作为负载。试验结果如下:
电流id和iq的实验波形如图7所示,其中图7(a)为SMC方法下负载增加的波形、图7(b)为B-SMC方法下负载增加的波形、图7(c)为SMC方法下负载降低的波形、图7(d)为B-SMC方法下负载降低的波形。从中可以看出,由于采用id=0的控制策略,id几乎是0的直线,而电流iq也随着负载的变化而变化。由于采用了B-SMC方法,提高了电流iq响应的快速性,速度响应时间缩短了0.1s左右,在负载变化的瞬间,波动很小,几乎没有超调。在正常情况下,电流纹波也会显著降低,约为10-15%。图8所示为转子位置角的测量波形,其中图8(a)为正常运行状态、图8(b)为负荷变化状态。由于B-SMC方法的鲁棒性,负载在负载变化瞬间发生变化,转子位置角变化不显著。与理论波形一致。结果表明,无论是快速性还是鲁棒性,该方法都优于普通滑模控制。
虽然结合附图描述了本发明的实施例,但是本领域技术人员可以在不脱离本发明的精神和范围的情况下作出各种修改和变型,这样的修改和变型均落入由所附权利要求所限定的范围之内。
Claims (7)
1.一种多相容错型磁通切换永磁电机反步滑模控制方法,其特征在于,包括如下步骤:
步骤1:采集多相定子电流,根据多相定子电流获取电机实际解耦合后的d-q轴电流,作为反步控制器的输入量;
步骤2:采集永磁电机的电机实时位置角度,根据电机实时位置角度计算电机实时角速度,将电机给定角速度与电机实时角速度的电机角速度差值作为反步控制器的输入量;
将电机实时角速度、电机实时位置角度、电机期望位置角度、永磁电机的扰动量作为自适应滑模控制器的输入量,自适应滑模控制器根据输入量形成输出控制量,作为反步控制器的输入;
步骤3:反步控制器根据自适应滑模控制器的输出控制量、电机角速度差值、输入的d轴参考电流、电机实际解耦合后的d-q轴电流,产生d-q轴电压ud、uq;
步骤4:将d-q轴电压ud、uq经过静止坐标系转换后,得到参考电压矢量分量;
步骤5:根据参考电压矢量分量驱动电机运行。
2.如权利要求1所述的多相容错型磁通切换永磁电机反步滑模控制方法,其特征在于,所述步骤1中根据多相定子电流获取电机实际耦合后的d-q轴电流,具体为:
多相定子电流经多相/两相静止坐标变换后得到α-β轴电流分量iα、iβ;再经两相静止/两相旋转坐标系变换,得到电机实际的解耦后d-q轴电流iq和id;
所述步骤4中将d-q轴电压ud、uq经过两相旋转/两相静止坐标系转换后,得到参考电压矢量分量uα、uβ。
6.如权利要求1所述的多相容错型磁通切换永磁电机反步滑模控制方法,其特征在于,所述步骤5中将参考电压矢量分量经过2/5变换后输入SVPWM调制模块,产生调制波发送给多相逆变器,驱动电机运行。
7.如权利要求1所述的多相容错型磁通切换永磁电机反步滑模控制方法,其特征在于,所述步骤3中输入的d轴参考电流id *=0。
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