CN113141121B - Current source type high-frequency isolation matrix type cascade converter and control method - Google Patents
Current source type high-frequency isolation matrix type cascade converter and control method Download PDFInfo
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- CN113141121B CN113141121B CN202110437779.3A CN202110437779A CN113141121B CN 113141121 B CN113141121 B CN 113141121B CN 202110437779 A CN202110437779 A CN 202110437779A CN 113141121 B CN113141121 B CN 113141121B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
- H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
- H02M5/22—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M5/275—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M5/293—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
- H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
- H02M5/22—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M5/225—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode comprising two stages of AC-AC conversion, e.g. having a high frequency intermediate link
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
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Abstract
The invention discloses a current source type high-frequency isolation matrix type cascade converter and a control method, comprising the following steps: after being filtered by LC, the three-phase AC power grid is respectively connected with two current source type matrix converters to transmit electric energy; the input end of the primary side of the high-frequency isolation transformer is connected with the matrix converter in series, the output end of the secondary side of the high-frequency isolation transformer is connected with the uncontrolled rectifier bridge, the alternating current sides of the two sets of current source type high-frequency isolation matrix type cascade converters are connected in parallel and share the LC filter, and the direct current output side of the high-frequency isolation matrix type cascade converter is connected with the resistance-inductance load in series. In the system, power frequency alternating current is converted into a high-frequency alternating current form through a three-phase/single-phase direct matrix converter, a direct current bus capacitor is removed, the system fault rate is reduced, the power conversion stage number is reduced, and the system volume is greatly reduced. The control technical scheme of the invention enables the switch of the matrix converter device to be flexible, inhibits the problem of electromagnetic interference caused by high-frequency operation of the switch device, meets the industrial requirements of offshore wind power generation, electric vehicle charging and the like, and has wide application prospect.
Description
Technical Field
The invention relates to the field of wind power generation, in particular to a current source type high-frequency isolation matrix type cascade converter and a control method.
Background
Wind energy (onshore and offshore) is one of the most important energy sources in renewable energy power generation systems, and at present, with the continuous development of offshore wind power generation, the power of an electric energy transmission system is continuously increased, and the size of a corresponding converter is also continuously enlarged. A power frequency transformer in a wind energy system of a traditional permanent magnet synchronous generator is large in size, and the traditional generator in the wind energy system mainly comprises a fan impeller, a three-stage gear box, a synchronous generator, a bidirectional PWM rectifier and the like. However, the conventional transformer in such a system has a large weight and volume, a small power capacity, and poor power transmission controllability, and because the transformer has a solid iron core and copper windings, the maintenance cost is high, and the manufacturing cost is expensive, so that the modular production and assembly of the wind power system are not facilitated. The transmission power of the wind power system is continuously increased, and the requirements on the power electronic converter system are also higher and higher. When a certain phase in the power circuit breaks down, the transmission of other bridge arms is influenced, and the normal operation of the whole system is further influenced. For the above reasons, research on power electronic converters has been a hot spot in offshore wind power generation high voltage direct current transmission systems.
The cascade structure is a promising offshore wind farm configuration and can replace expensive and huge offshore substations. The cascade structure based on the modular direct matrix converter is very suitable for a high-voltage direct-current transmission system with a current source converter due to the characteristics of high power output and high dynamic response. A high-frequency isolation transformer is adopted to replace a low-frequency input transformer, and an electrolytic capacitor which is large in size and easy to damage in a wind energy conversion system is eliminated. Therefore, the power density, reliability and transmission efficiency of the system can be improved. In addition, the direct matrix converter can also realize single-stage power conversion and soft switching, and further reduce the loss of the system. The modular design of the direct matrix converter facilitates the installation and maintenance of the system, which is of great importance to offshore wind farms, and the cost reduction is beneficial to the large-scale application of offshore wind farms. Finally, the series configuration eliminates offshore substations and facilitates the pooling of multiple low voltage wind power generation systems into a medium voltage power grid. And a proper power device can be selected, and the advantages of high reliability, controllable power factor, high waveform quality and the like are realized.
A large part of loss of the high-frequency isolation matrix type cascade converter comes from switching loss of a switching device, and the switching loss of the device can be greatly reduced and the transmission efficiency of the converter can be improved by applying the soft switching technology of the bidirectional switch of the matrix converter. The traditional soft switching technology of the matrix converter mainly aims at a voltage source type converter, the current source type soft switching technology is not researched much, and the defect of large loss of the current source type matrix converter can be effectively overcome by adopting the soft switching technology. An active damping harmonic suppression scheme is adopted in a control strategy, so that low-order harmonics in a cascade system are effectively suppressed, and harmonic loss of the system is reduced.
Disclosure of Invention
In order to solve the defects mentioned in the background technology, the invention aims to provide a current source type high-frequency isolation matrix type cascade converter and a control method, the input sides of two current source type high-frequency isolation matrix converters are connected in parallel and the output sides of the two current source type high-frequency isolation matrix converters are connected in series, the power conversion capacity of a system is improved, the reliability of the system is enhanced, and the volume of the system is reduced; by applying the soft switching technology, the switching loss of the current source type high-frequency isolation matrix converter is reduced, and the efficiency of the system is improved.
The purpose of the invention can be realized by the following technical scheme:
a current source type high-frequency isolation matrix type cascade converter comprises a three-phase power grid, an LC filter, a first current source type matrix converter, a second current source type matrix converter, a first leakage inductor, a second leakage inductor, a first high-frequency isolation transformer, a second high-frequency isolation transformer, a first uncontrolled rectifier bridge, a second uncontrolled rectifier bridge, a first inductance resistance load and a second inductance resistance load, wherein the three-phase power grid is connected with the LC filter; the LC filter is respectively connected with the first current source type matrix converter and the second current source type matrix converter in parallel; the alternating current output side of the first current source type matrix converter is connected with the first high-frequency isolation transformer in series, and the alternating current output side of the second current source type matrix converter is connected with the second high-frequency isolation transformer in series; the secondary output side of the first high-frequency isolation transformer is connected with the first uncontrolled rectifier bridge in series, and the secondary output side of the second high-frequency isolation transformer is connected with the second uncontrolled rectifier bridge in series, so that energy is transferred through a magnetic field; the first uncontrolled rectifier bridge output end and the second uncontrolled rectifier bridge output end are connected in series through a first resistance-inductance load and a second resistance-inductance load to form a closed loop.
Further, the power direction and the power magnitude of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the first current source type matrix converter and the second current source type matrix converter; the first current source type matrix converter and the second current source type matrix converter are respectively connected with ports of the LC filter and receive electric energy transmitted by the three-phase power grid; the current of the first resistance-inductance load and the second resistance-inductance load output by the direct current side is controlled by a current loop.
Further, the control method adopted by the control module of the first current source type matrix converter comprises the following processes:
1) three capacitor voltages U passing through three-phase LC filter abc And the network voltage V g Electric angle theta obtained by phase-locked loop e Obtaining a capacitance voltage d-axis component U of the LC filter through coordinate transformation d And q-axis component U q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q Obtaining the low-frequency component of the capacitor voltage through a low-pass filter, and obtaining the electrical angle theta e Obtaining the electrical angular velocity omega after differentiation e The low-frequency capacitance current of the filter capacitor is obtained after calculation of the capacitance current compensation module I
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two is obtained by a D-axis direct current component through a PI controller
To realize the transmission of unit power factor, let the reference value Q of reactive power ref Is 0, a q-axis current component is obtained
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltage hd And U hq Multiplying the high-frequency components of the capacitor voltage by the virtual resistance coefficient k pv1 (2.8) and k pv2 (2.9) obtaining the dq axis component value of the virtual current, wherein the virtual resistance coefficient k pv1 (2.8) and k pv2 (2.9) the values are the same, and the fifth harmonic and the seventh harmonic of the system current are eliminated through an active damping scheme of the virtual resistance suppression circuit harmonic;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion 1dc * And phase angle theta α1 ；
6) Phase current fundamental wave peak value I 1dc * Divided by a given value of DC current I dc * Obtaining a modulation ratio m of the first current source type matrix converter 1i Phase angle theta α1 Plus the electrical angle theta measured by the phase-locked loop e Obtaining the switching pulse phase angle theta of the matrix converter 1ω Using the modulation ratio m 1i And angle theta 1ω And the switching period Ts generates twelve switching pulses of the first current source type matrix converter.
Further, the control method adopted by the control module of the second current source type matrix converter comprises the following processes:
1) three capacitor voltages U through three-phase LC filter abc And the network voltage V g Electric angle theta obtained by phase-locked loop e Obtaining a capacitance voltage d-axis component U of the LC filter through coordinate transformation d And q-axis component U q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q Obtaining the low-frequency component of the capacitor voltage through a low-pass filter, the electrical angle theta e Obtaining the electrical angular velocity omega after differentiation e The low-frequency capacitance current of the filter capacitor can be obtained after calculation of the capacitance current compensation module II
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two obtains a d-axis direct current component through a PI controller
To realize a sheetTransmission of bit power factor, making reference value Q of reactive power ref Is 0, and a q-axis current component is obtained
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltage hd And U hq Multiplying the high-frequency components of the capacitor voltage by the virtual resistance coefficient k pv3 (2.15) and k pv4 (2.16) obtaining the dq-axis component value of the virtual current, the virtual resistivity k pv3 (2.15)、k pv4 (2.16) and k pv1 (2.8)、k pv2 (2.9) the values are the same, and the fifth harmonic and the seventh harmonic of the system current are eliminated through an active damping scheme of the virtual resistance suppression circuit harmonic;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion 2dc * And phase angle theta α2 ；
6) Phase current fundamental wave peak value I 2dc * Divided by a given value of DC current I dc * Obtaining a modulation ratio m of the second current source type matrix converter 2i Phase angle θ α2 Plus the electrical angle theta measured by the phase-locked loop e Obtaining the switching pulse phase angle theta of matrix converter 2ω Using the modulation ratio m 2i And an angle theta 2ω And the switching period Ts generates twelve switching pulses of the second current source type matrix converter.
Further, the modulation method adopted by the first current source type matrix converter and the second current source type matrix converter includes the steps of:
three current vectors acting on the first current source type matrix converter in one switching period are respectively I 11 ，I 12 ，I 10 The output voltage of the corresponding first current source type matrix converter is U 11 ，U 12 ，U 10 By changing the order of action of the current vectors, so that U 11 >U 12 >U 10 (ii) a Similarly, the three current vectors acting on the second current source type matrix converter in one switching period are respectively I 21 ，I 22 ，I 20 The output voltage of the matrix converter corresponding to the second current source is U 21 ，U 22 ，U 20 By changing the order of action of the current vectors, so that U 21 >U 22 >U 20 (ii) a Because the first current source matrix converter and the second current source matrix converter are connected in parallel, the output voltage waveforms of the matrix converters in a switching period are consistent, and the working states are consistent, taking the working state of the first current source matrix converter as an example, the specific working process of the soft switch in a switching period is as follows,
1) state 0: primary side commutation
The current vectors corresponding to the first current source type matrix converter and the second current source type matrix converter are zero vectors I respectively just after the start of a switching period 10 And zero vector I 20 The matrix converter output voltage drops to 0;
2) state 1: conduction time of switch tube
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer are equal, and the primary side current of the first high-frequency isolation transformer flows through a switch tube S of the first current source type matrix converter 11 、S 21 And S 14 、S 24 The current of the secondary side rectifier bridge of the first high-frequency isolation transformer flows through D 1 、D 4 The three-phase capacitor provides a current path for the three-phase inductor, and the state has no energy transmission, S 11 And S 14 Conducting at zero voltage;
3) state 2: conduction time of switch tube
Zero vector I of first current source type matrix converter 10 End of action, current vector I 11 In operation, the inductor output current of the first uncontrolled rectifier bridge flows through the diode D 1 、D 4 According to four-step current conversion, the switching tube S of the first current source type matrix converter 14 And S 24 Off since the voltage u is now ab >0, current to S 16 And S 26 The output capacitor of the first current source type matrix converter 16 And S 26 Zero voltage conduction is carried out, and the voltage of the input side of the first high-frequency isolation transformer is equal to u ab Energy flows from the grid;
4) state 3: conduction time of switch tube
Similar to the operating state of state 2, the effective current vector I of the first current source matrix converter 11 End of action, current vector I 12 In operation, the inductor output current of the first uncontrolled rectifier bridge flows through the diode D 1 、D 4 According to the four-step commutation, the power switch tube S of the first current source type matrix converter 16 And S 26 Closing, S 12 And S 22 Zero voltage conduction is carried out, and the input side voltage of the first high-frequency isolation transformer is equal to u ac Energy flows from the grid;
5) and 4, state 4: commutation of uncontrolled rectifier bridge
The rectification of the first uncontrolled rectifier bridge is completed with the aid of the first current source matrix converter, at t 3 At the moment, the diode D of the first uncontrolled rectifier bridge 2 、D 3 Zero current conduction, the current of the first uncontrolled rectifier bridge flowing through the inductor linearly decreases through D 2 、D 3 Through D 1 、D 4 Is linearly decreased at the same rate, the commutation overlap time T d Selecting 100ns, and commutating current on the first uncontrolled rectifier bridge inductor before the mode is finished;
6) and state 5: conduction time of switch tube
Commutation overlap time T d After finishingPower switch tube S of a first current source matrix converter 12 And S 22 Off, S 14 And S 24 Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer is 0, and no direct current energy is transmitted under the mode;
7) and 6, state: conduction time of switch tube
First current source type matrix converter current vector I 12 End of action, zero vector I 10 In operation, a direct current flows through the diode D of the first uncontrolled rectifier bridge on the load side 2 And D 3 While the input side current of the first high-frequency isolation transformer flows through the power switch tube S of the first current source type matrix converter 11 、S 21 And S 24 、S 14 And the three-phase capacitor in the LC filter provides a current channel for the three-phase inductor.
The invention has the beneficial effects that:
(1) the direct conversion from power frequency alternating current to high frequency alternating current is realized through the current source type matrix converter, the direct current capacitor is removed, the failure rate of the converter is reduced, the power conversion stage number is reduced, and the reliability and the power density of system operation are greatly improved;
(2) a high-frequency isolation transformer is adopted to replace a low-frequency input transformer, so that the system volume is reduced;
(3) the capacity of the power converter is increased through a cascade structure of a plurality of direct matrix converters, the output voltage of a system is improved, and the collection of medium-high voltage direct-current power transmission is facilitated;
(4) the control strategy of the matrix converter can realize unit power factor operation, the soft switching technology reduces the loss of the system, the modularized matrix converter is easy to install and maintain, and the cost of the offshore wind power generation system is reduced.
Drawings
The invention is further described below with reference to the accompanying drawings.
FIG. 1 is an overall architecture diagram of the present invention;
FIG. 2 is a schematic block diagram of a control method for two current source matrix converters according to the present invention;
FIG. 3(a) is a first current source type moment of the present inventionSchematic diagram (V) of current vector corresponding to output voltage of array converter ab ＞V ac )；
FIG. 3(b) is a schematic diagram (V) of the output voltage corresponding to the current vector of the second current source type matrix converter according to the present invention ac ＞V ab )；
Fig. 4 is a schematic diagram of current flow paths of two current source matrix converters in an operating mode based on a space current vector modulation method according to the present invention, wherein fig. 4(a) -fig. 4(f) are schematic diagrams of current flow paths in operating modes of states 1-6, respectively;
FIG. 5.1(a) is a diagram showing an input voltage V of the current source matrix converter of the present invention p And an output voltage V s An experimental oscillogram;
FIG. 5.1(b) is a diagram showing an input voltage V of the current source matrix converter of the present invention p And current i p An experimental oscillogram;
FIG. 5.2 is a simulation waveform diagram of the current source type high frequency isolation matrix type cascaded converter of the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
As shown in fig. 1, a current source type high frequency isolation matrix type cascade converter includes a three-phase power grid 1.1, an LC filter 1.2, two sets of matrix converters (1.3, 1.4), leakage inductors (1.5, 1.6), high frequency isolation transformers (1.7, 1.8), two sets of uncontrolled rectifier bridges (1.9, 1.10), and resistive-inductive loads (1.11, 1.12).
The three-phase power grid 1.1 is connected with the LC filter 1.2;
the LC filter 1.2 is respectively connected with the first matrix converter 1.3 and the second matrix converter 1.4 in parallel; the matrix converters (1.3 and 1.4) are respectively connected with high-frequency isolation transformers (1.7 and 1.8); the secondary sides of the high-frequency isolation transformers (1.7 and 1.8) are connected with uncontrolled rectifier bridges (1.9 and 1.10) and transmit energy through magnetic fields; the uncontrolled rectifier bridges (1.9, 1.10) are respectively connected with a resistance-inductance load 1.11 and a resistance-inductance load 1.12; the direct current output side resistance-inductance load 1.11 and the resistance-inductance load 1.12 are mutually connected in series to form a closed loop;
the matrix converters 1.3 and 1.4 at the input side of the high-frequency isolation transformer are connected in parallel;
direct current bus inductors 1.11 and 1.12 at the output side of the high-frequency isolation transformer are mutually connected in series;
the power direction and the power size of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the matrix converters 1.3 and 1.4;
the current of the DC side output resistive-inductive loads 1.11 and 1.12 is controlled by a current loop.
The current source type matrix converter comprises a first current source type matrix converter 1.3 and a second current source type matrix converter 1.4, the high-frequency isolation transformer comprises a first high-frequency isolation transformer 1.7 and a second high-frequency isolation transformer 1.8, the rectifying circuit comprises a first uncontrolled rectifier bridge 1.9 and a second uncontrolled rectifier bridge 1.10, wherein:
the alternating current output side of the first current source type matrix converter 1.3 is connected with a first high-frequency isolation transformer 1.7 in series;
the alternating current output side of the second current source type matrix converter 1.4 is connected with a second high-frequency isolation transformer 1.8 in series;
the alternating current input side of the first current source type matrix converter 1.3 is connected with the alternating current input side of the second current source type matrix converter 1.4 in parallel;
the output end of the secondary side of the first high-frequency isolation transformer 1.7 is connected with a first uncontrolled rectifier bridge 1.9 in series;
the output end of the secondary side of the second high-frequency isolation transformer 1.9 is connected with a second uncontrolled rectifier bridge 1.10 in series;
the output side of the first uncontrolled rectifier bridge 1.9 and the output side of the second uncontrolled rectifier bridge 1.10 are connected in series through two sets of inductance-resistance loads 1.11 and 1.12 to form a closed loop;
the first current source type matrix converter 1.3 and the second current source type matrix converter 1.4 are respectively connected with the port of the three-phase LC filter 1.2 and receive electric energy transmitted by a three-phase power grid.
As shown in fig. 2, fig. 5.1(a), fig. 5.1(b) and fig. 5.2, the current closed-loop control method adopted by the control strategy of the first current source type matrix converter 1.3 comprises the following steps:
1) three capacitor voltages U through three-phase LC filter 1.2 abc And the network voltage V g Electrical angle theta obtained by phase-locked loop e Obtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.1 1d And q-axis component U 1q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q The low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter 2.2 e Obtaining the electrical angular velocity omega after differentiation of 2.3 e The low-frequency capacitance current of the filter capacitor is obtained after calculation by the capacitance current compensation module I2.6
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two obtains a d-axis direct current component through a PI controller 2.5
To realize the transmission of unit power factor, let the reference value Q of reactive power ref Is 0, and a q-axis current component is obtained
Is zero;
4) the dq axis component of the capacitor voltage passes through a high-pass filter 2.7 to obtain a high-frequency component U of the capacitor voltage hd And U hq Multiplying the high frequency components of the capacitor voltage by the virtual resistance systemNumber k pv1 (2.8)、k pv2 (2.9), the virtual resistivity k pv1 (2.8) and k pv2 (2.9) obtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate 2.17 conversion 1dc * And phase angle theta α1 ；
6) Phase current fundamental wave peak value I 1dc * Divided by a given value of DC current I dc * 2.18 obtaining modulation ratio m of the first current source type matrix converter 1i Phase angle θ α1 Plus the electrical angle theta measured by the phase-locked loop e 2.19 obtaining the switching pulse phase angle theta of matrix converter 1ω Using the modulation ratio m 1i And an angle theta 1ω And the switching period Ts generates twelve switching pulses of the first current source matrix converter 1.3.
As shown in fig. 2, the dc current control method adopted by the control strategy of the second current source type matrix converter 1.4 includes the following steps:
1) three capacitor voltages U through the three-phase LC filter 1.2 abc And the network voltage V g Electric angle theta obtained by phase-locked loop e Obtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.10 d And q-axis component U q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q Through a low passThe filter 2.12 obtains the low frequency component of the capacitor voltage, the electrical angle theta e Obtaining the electrical angular velocity omega after differentiating by 2.11 e The low-frequency capacitance current of the filter capacitor can be obtained after calculation of the second capacitance current compensation module 2.13
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two obtains a d-axis direct current component through a PI controller 2.5
To realize the transmission of unit power factor, let the reference value Q of reactive power ref Is 0, and a q-axis current component is obtained
Is zero;
4) the dq axis component of the capacitor voltage passes through a high pass filter 2.14 to obtain a high frequency component U of the capacitor voltage hd And U hq Multiplying the high-frequency components of the capacitor voltage by the virtual resistance coefficient k pv3 (2.15)、k pv4 (2.16) obtaining the dq-axis component value of the virtual current, wherein the virtual resistance coefficient k pv3 (2.15)、k pv4 (2.16) and k pv1 (2.8)、k pv2 (2.9) the values are the same, and the fifth harmonic and the seventh harmonic of the system current are eliminated through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after conversion of polar coordinates 2.20 2dc * And phase angle theta α2 ；
6) Phase current fundamental wave peak value I 2dc * Divided by a given value of DC current I dc * 2.21 obtaining modulation ratio m of second current source type matrix converter 2i Phase angle theta α2 Plus the electrical angle theta measured by the phase-locked loop e 2.22 obtaining the switching pulse phase angle theta of the matrix converter 2ω Using the modulation ratio m 2i And an angle theta 2ω And the switching period Ts generates twelve switching pulses of the second current source matrix converter 1.4.
The space vector modulation method of the matrix converters 1.3 and 1.4 comprises the following steps:
for simplicity of analysis, the three current vectors acting on the first current source matrix converter 1.3 in a switching cycle are each I 11 ，I 12 ，I 10 Corresponding to the first current source type matrix converter 1.3, the output voltage is U 11 ，U 12 ，U 10 By changing the order of action of the current vectors, U is made 11 >U 12 >U 10 (ii) a Similarly, the three current vectors acting on the second current source matrix converter 1.4 in one switching cycle are respectively I 21 ，I 22 ，I 20 Corresponding to the output voltage of the second current source type matrix converter 1.4 as U 21 ，U 22 ，U 20 By changing the order of action of the current vectors, so that U 21 >U 22 >U 20 (ii) a Since the first current source matrix converter 1.3 and the second current source matrix converter 1.4 are connected in parallel, the output voltage waveforms of the matrix converters in one switching period are consistent, and the working states are consistent, taking the working state of the first current source matrix converter 1.3 as an example, the specific working process of the soft switch in one switching period is as follows, without assuming that the first current source matrix converter 1.3 works in the first sector, and at this time, the second current source matrix converter 1.4 works in the second sectorThe device 1.4 also operates in the first sector.
As shown in fig. 3(a), (b) and fig. 4 (light color for off and dark color for on):
1) state 0: primary side commutation (t) 0 )
When a switching cycle begins, the current vectors corresponding to the first current source type matrix converter 1.3 and the second current source type matrix converter 1.4 are zero vectors I 10 And zero vector I 20 The output voltage of the matrix converter is reduced to 0;
2) state 1 (3.1): switch tube on-time (t) 0 -t 1 )
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer 1.7 are equal, and the primary side current of the first high-frequency isolation transformer 1.7 flows through the switching tube S of the first current source type matrix converter 1.3 11 、S 21 And S 14 、S 24 The current of the secondary side rectifier bridge of the first high-frequency isolation transformer 1.7 flows through D 1 、D 4 Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S 11 And S 14 Conducting at zero voltage;
3) state 2 (3.2): conduction time (t) of switching tube 1 -t 2 )
Zero vector I of the first current source matrix converter 1.3 10 End of action, current vector I 11 In operation, the inductive output current of the first uncontrolled rectifier bridge 1.9 flows through the diode D 1 、D 4 According to a four-step commutation, the switching tube S of the first current source matrix converter 1.3 14 And S 24 Is turned off because the voltage u is now ab >0, current to S 16 And S 26 Output capacitor charging, matrix converter power switch tube S 16 And S 26 Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to u ab Energy flows from the grid;
4) state 3 (3.3): switch tube on-time (t) 2 -t 3 )
Similar to the operating state of State 2, firstEffective current vector I of current source matrix converter 1.3 11 End of action, current vector I 12 In operation, the inductive output current of the first uncontrolled rectifier bridge 1.9 flows through the diode D 1 、D 4 According to a four-step commutation, the switching tube S of the first current source matrix converter 1.3 16 And S 26 Closing, S 12 And S 22 Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to u ac Energy flows from the grid;
5) state 4 (3.4): commutation of an uncontrolled rectifier bridge (t) 3 -t 4 )
The rectification of the first uncontrolled rectifier bridge 1.9 is completed with the aid of the first current source matrix converter 1.3, at t 3 At the moment, the diode D of the first uncontrolled rectifier bridge 1.9 2 、D 3 Zero current conduction. The current flowing through the inductor of the first uncontrolled rectifier bridge 1.9 decreases linearly through D 2 、D 3 Through D 1 、D 4 Is linearly decreased at the same rate, commutation overlap time T in this context d Selecting 100ns, and commutating current on an inductor of a first uncontrolled rectifier bridge 1.9 before the mode is finished;
6) state 5 (3.5): conduction time (t) of switching tube 4 -t 5 )
Commutation overlap time T d After that, the power switch tube S of the first current source type matrix converter 1.3 12 And S 22 Off, S 14 And S 24 Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer 1.7 is 0, and no direct current energy is transmitted in the mode;
7) state 6 (3.6): switch tube on-time (t) 5 -t 6 )
First current source matrix converter 1.3 current vector I 12 End of action, zero vector I 10 In operation, a direct current flows on the load side through the diode D of the first uncontrolled rectifier bridge 1.9 2 And D 3 While the input side current of the first high-frequency isolation transformer 1.7 flows through the power switch tube S of the first current source type matrix converter 1.3 11 、S 21 And S 24 、S 14 And the three-phase capacitor in the LC filter 1.2 provides a current channel for the three-phase inductor.
The foregoing shows and describes the general principles, principal features, and advantages of the invention. It will be understood by those skilled in the art that the present invention is not limited to the embodiments described above, which are described in the specification and illustrated only to illustrate the principle of the present invention, but that various changes and modifications may be made therein without departing from the spirit and scope of the present invention, which fall within the scope of the invention as claimed.
Claims (4)
1. A current source type high-frequency isolation matrix type cascade converter comprises a three-phase power grid (1.1), an LC filter (1.2), a first current source type matrix converter (1.3) and a second current source type matrix converter (1.4), a first leakage inductor (1.5) and a second leakage inductor (1.6), a first high-frequency isolation transformer (1.7) and a second high-frequency isolation transformer (1.8), a first uncontrolled rectifier bridge (1.9) and a second uncontrolled rectifier bridge (1.10), a first inductance resistance load (1.11) and a second inductance resistance load (1.12), and is characterized in that the three-phase power grid (1.1) is connected with the input end of the LC filter (1.2); the input ends of the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) are connected to the output end of the LC filter (1.2) in parallel; the alternating current output side of the first current source type matrix converter (1.3) is connected with the primary side of a first high-frequency isolation transformer (1.7) in series through a first leakage inductor (1.5), and the alternating current output side of the second current source type matrix converter (1.4) is connected with the primary side of a second high-frequency isolation transformer (1.8) in series through a second leakage inductor (1.6); the secondary output side of the first high-frequency isolation transformer (1.7) is connected with the input end of the first uncontrolled rectifier bridge (1.9) in series, the secondary output side of the second high-frequency isolation transformer (1.8) is connected with the input end of the second uncontrolled rectifier bridge (1.10) in series, and energy is transferred through a magnetic field; the output end of the first uncontrolled rectifier bridge (1.9) and the output end of the second uncontrolled rectifier bridge (1.10) are connected in series through a first resistance-inductance load (1.11) and a second resistance-inductance load (1.12) to form a closed loop;
the modulation method adopted by the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) comprises the following steps:
three current vectors acting on the first current source matrix converter (1.3) in one switching cycle are respectively I 11 ，I 12 ，I 10 Corresponding to the output voltage of the first current source type matrix converter (1.3) being U 11 ，U 12 ，U 10 By changing the order of action of the current vectors, so that U 11 >U 12 >U 10 (ii) a Similarly, three current vectors acting on the second current source type matrix converter (1.4) in one switching period are respectively I 21 ，I 22 ，I 20 Corresponding to the output voltage of the second current source type matrix converter (1.4) being U 21 ，U 22 ，U 20 By changing the order of action of the current vectors, so that U 21 >U 22 >U 20 (ii) a Because the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) are connected in parallel, the output voltage waveforms of the matrix converters in a switching period are consistent, the working state is consistent, taking the working state of the first current source type matrix converter (1.3) as an example, the specific working process of the soft switch in a switching period is as follows,
1) state 0: primary side commutation: t is t 0
The current vectors corresponding to the first current source matrix converter (1.3) and the second current source matrix converter (1.4) are zero vectors I respectively just after the beginning of one switching period 10 And zero vector I 20 The matrix converter output voltage drops to 0;
2) state 1 (3.1): conduction time of the switching tube: t is t 0 -t 1
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer (1.7) are equal, and the primary side current of the first high-frequency isolation transformer (1.7) flows through a switch tube S of the first current source type matrix converter (1.3) 11 、S 21 And S 14 、S 24 The current of the secondary side rectifier bridge of the first high-frequency isolation transformer (1.7) flows through D 1 、D 4 Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S 11 And S 14 Conducting at zero voltage;
3) state 2 (3.2): conducting time of the switching tube: t is t 1 -t 2
Zero vector I of a first current source matrix converter (1.3) 10 End of action, current vector I 11 In the beginning of the operation, the inductive output current of the first uncontrolled rectifier bridge (1.9) flows through the diode D 1 、D 4 According to a four-step commutation, the switching tube S of the first current source matrix converter (1.3) 14 And S 24 Off since the voltage u is now ab >0, current to S 16 And S 26 Is charged, the power switch tube S of the first current source type matrix converter (1.3) 16 And S 26 Zero voltage conduction is carried out when the input side voltage of the first high-frequency isolation transformer (1.7) is equal to u ab Energy flows from the grid;
4) state 3 (3.3): conducting time of the switching tube: t is t 2 -t 3
The effective current vector I of the first current source matrix converter (1.3) is similar to the operating state of state 2 11 End of action, current vector I 12 In the beginning of the operation, the inductive output current of the first uncontrolled rectifier bridge (1.9) flows through the diode D 1 、D 4 According to a four-step commutation, the power switch S of the first current source matrix converter (1.3) 16 And S 26 Closing, S 12 And S 22 Zero voltage conduction is carried out, and the input side voltage of the first high-frequency isolation transformer (1.7) is equal to u ac Energy flows from the grid;
5) state 4 (3.4): rectification and commutation of an uncontrolled rectifier bridge: t is t 3 -t 4
The rectification of the first uncontrolled rectifier bridge (1.9) is completed with the aid of the first current source matrix converter (1.3), at t 3 At the moment, the diode D of the first uncontrolled rectifier bridge (1.9) 2 、D 3 Zero current conduction, the current flowing through the inductor of the first uncontrolled rectifier bridge (1.9) decreases linearly through D 2 、D 3 Through D 1 、D 4 Is linearly decreased at the same rate, the commutation overlap time T d The selection of 100ns is made for which,before the mode is finished, the current on the inductor of the first uncontrolled rectifier bridge (1.9) is commutated;
6) state 5 (3.5): conduction time of the switching tube: t is t 4 -t 5
Commutation overlap time T d After finishing, the power switch tube S of the first current source type matrix converter (1.3) 12 And S 22 Off, S 14 And S 24 Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer (1.7) is 0, and no direct current energy is transmitted under the mode;
7) state 6 (3.6): conducting time of the switching tube: t is t 5 -t 6
First current source type matrix converter (1.3) current vector I 12 End of action, zero vector I 10 In operation, a load-side direct current flows through the diode D of the first uncontrolled rectifier bridge (1.9) 2 And D 3 Meanwhile, the current at the input side of the first high-frequency isolation transformer (1.7) flows through the power switch tube S of the first current source type matrix converter (1.3) 11 、S 21 And S 24 、S 14 And the three-phase capacitor in the LC filter (1.2) provides a current channel for the three-phase inductor.
2. A current source high frequency isolated matrix type cascaded converter as claimed in claim 1, wherein the power direction and power magnitude of the current source high frequency isolated matrix type cascaded converter is determined by the control module of the first current source matrix converter (1.3) and the second current source matrix converter (1.4); the first current source type matrix converter (1.3) and the second current source type matrix converter (1.4) are respectively connected with the port of the LC filter (1.2) and receive electric energy transmitted by a three-phase power grid; the current of the first resistance-inductance load (1.11) and the second resistance-inductance load (1.12) output by the direct current side is controlled by a current loop.
3. A current source high frequency isolated matrix type cascaded converter as claimed in claim 2, characterized in that said control module of said first current source matrix converter (1.3) adopts a control method comprising the following steps:
1) three capacitor voltages U through a three-phase LC filter (1.2) abc And the network voltage V g Electric angle theta obtained by phase-locked loop e Obtaining a capacitance voltage d-axis component U of the LC filter (1.2) through coordinate transformation (2.1) d And q-axis component U q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q Low-frequency component of capacitor voltage, electrical angle theta, is obtained through a low-pass filter (2.2) e Obtaining the electrical angular velocity omega after differentiation (2.3) e The low-frequency capacitance current of the filter capacitor is obtained after calculation of the capacitance current compensation module I (2.6)
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two obtains a d-axis direct current component through a PI controller (2.5)
To realize the transmission of unit power factor, let the reference value Q of reactive power ref Is 0, a q-axis current component is obtained
Is zero;
4) the dq axis component of the capacitor voltage is subjected to a high-pass filter (2.7) to obtain a high-frequency component U of the capacitor voltage hd And U hq Multiplying the high-frequency components of the capacitor voltage by the virtual resistance coefficient k pv1 (2.8) and k pv2 (2.9) obtaining the dq-axis component value of the virtual current, wherein the virtual resistivity k is pv1 (2.8) and k pv2 (2.9) the values are the same, and the fifth harmonic and the seventh harmonic of the system current are eliminated through an active damping scheme of the virtual resistance suppression circuit harmonic;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after conversion of polar coordinates (2.17) 1dc * And phase angle theta α1 ；
6) Phase current fundamental wave peak value I 1dc * Divided by a given value of DC current I dc * (2.18) obtaining a modulation ratio m of the first current source type matrix converter (1.3) 1i Phase angle θ α1 Plus the electrical angle theta measured by the phase-locked loop e (2.19) obtaining the switching pulse phase angle theta of the matrix converter 1ω Using the modulation ratio m 1i And angle theta 1ω And a switching period Ts generates twelve switching pulses of the first current source matrix converter (1.3).
4. A current source high frequency isolated matrix type cascaded converter as claimed in claim 2, wherein the control method adopted by the control module of said second current source matrix converter (1.4) comprises the following processes:
1) three capacitor voltages U passing through a three-phase LC filter (1.2) abc And the network voltage V g Electrical angle theta obtained by phase-locked loop e Obtaining a capacitance voltage d-axis component U of the LC filter (1.2) through coordinate transformation (2.10) d And q-axis component U q ；
2) Component U under filter capacitor voltage dq coordinate system d And U q The low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter (2.12) e The electrical angular velocity omega is obtained after differentiation (2.11) e Passing through a capacitor current compensation module II(2.13) calculating to obtain the low-frequency capacitance current of the filter capacitor
And
3) given load side DC bus current I dc * And the actual DC bus current I dc The error value between the two obtains a d-axis direct current component through a PI controller (2.5)
To realize the transmission of unit power factor, let the reference value Q of reactive power ref Is 0, a q-axis current component is obtained
Is zero;
4) the dq component of the capacitor voltage is processed by a high-pass filter (2.14) to obtain a high-frequency component U of the capacitor voltage hd And U hq Multiplying the high-frequency components of the capacitor voltage by the virtual resistance coefficient k pv3 (2.15) and k pv4 (2.16) obtaining the dq-axis component value of the virtual current, the virtual resistivity k pv3 (2.15) and k pv4 (2.16) the values are the same, and the fifth harmonic and the seventh harmonic of the system current are eliminated through an active damping scheme of the virtual resistance suppression circuit harmonic;
5) d-axis DC component
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
And
obtaining matrix transformation by operationThe given value of the final current at the input side of the device is converted by polar coordinates (2.20) to obtain a phase current fundamental wave peak value I 2dc * And phase angle theta α2 ；
6) Phase current fundamental wave peak value I 2dc * Divided by a given value of DC current I dc * (2.21) obtaining a modulation ratio m of the second current source type matrix converter (1.4) 2i Phase angle θ α2 Plus the electrical angle theta measured by the phase-locked loop e (2.22) obtaining the switching pulse phase angle theta of the matrix converter 2ω Using the modulation ratio m 2i And angle theta 2ω And a switching period Ts generates twelve switching pulses of the second current source matrix converter (1.4).
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