CN112865646A - Dead-beat prediction control method for single current sensor of permanent magnet synchronous motor - Google Patents

Abstract

The invention provides a dead-beat prediction control method for a single current sensor of a permanent magnet synchronous motor, which is characterized in that based on three-phase current reconstructed by a bus current sensor, an Extended State Observer (ESO) is adopted, total disturbance of the motor, namely disturbance caused by factors such as parameter mismatch, dead time, non-whole period prediction and the like is used as an extended state quantity, and the extended state quantity can change along with voltage disturbance in real time to compensate output dq axis voltage. The method reduces the cost of the control system, reduces the volume of the control system, and has an inhibiting effect on the disturbance of the motor such as parameter change and the like.

Description

Dead-beat prediction control method for single current sensor of permanent magnet synchronous motor

Technical Field

The invention relates to the technical field of permanent magnet synchronous motor control, in particular to single current sensor dead-beat prediction control realized based on a bus current sensor aiming at the condition of disturbance caused by parameter mismatch and the like of a permanent magnet synchronous motor.

Background

In the prior art of permanent magnet synchronous motor control, by means of traditional control strategies such as PI control, dead beat model predictive control and the like, the dependence degree on system models and motor parameters is very high, the inherent problem of poor parameter robustness exists, when the parameters change, the control effect is rapidly reduced, so that the control strategies depending on models and parameter accuracy cannot meet the requirement of high-performance control, and the application range of the control strategies is limited.

At present, the main methods for controlling the permanent magnet synchronous motor comprise vector control and direct torque control, and most of the methods are based on closed-loop control and require the acquisition of three-phase current of the motor for feedback to form a closed loop. Accurate acquisition of three-phase current is particularly important for the control strategies, the simplest mode is to detect currents of two-phase windings respectively, but the increase of torque fluctuation of the whole system is brought by direct current bias difference or current gain difference between different sensors, and the improvement of system performance is limited. Meanwhile, the disturbance problem caused by the phenomena of parameter mismatch and the like of the permanent magnet synchronous motor in the control process also has adverse effect on the motor control

Disclosure of Invention

In view of this, the invention provides a dead-beat prediction control method for a single current sensor of a permanent magnet synchronous motor, which specifically includes the following steps:

step one, collecting the bus current i of the inverter in real time on line_dcThe rotor rotating speed w and the rotor position angle theta, and the real-time switching state of the inverter and the collected bus current i of the inverter are utilized_dcThree-phase current i is reconstructed_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a traditional deadbeat current prediction control model, and predicting d and q axis voltages of the motor at the next moment;

introducing a nonlinear active disturbance rejection control fal function to establish an extended state observer, observing the total disturbance of the motor as the extended state quantity, and compensating the d-q axis voltage at the next moment obtained by dead-beat current prediction control;

and step four, calculating the reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the step one, the reconstructed three-phase current and the compensated d-axis and q-axis voltages.

Further, the mathematical model of the permanent magnet synchronous motor established in the second step under the α - β coordinate system is as follows:

u_α＝R_si_α+L_spi_α-w_eψ_rsinθ

u_β＝R_si_β+L_spi_β+w_eψ_rcosθ

ψ_α＝L_si_α+ψ_rcosθ

ψ_β＝L_si_β+ψ_rsinθ

T_e＝1.5p_mψ_r(i_βcosθ-i_αsinθ)

in the formula u_α、u_βIs the stator voltage under an alpha-beta coordinate system; i.e. i_α、i_βIs the stator current under an alpha-beta coordinate system; Ψ_rIs a rotor flux linkage; r_sIs a stator resistor; l is_sIs a stator inductance; w is a_e、w_mThe electrical angular velocity of the rotor and the mechanical angular velocity of the rotor, respectively; theta is a rotor position angle; p is a differential operator; t is_eIs an electromagnetic torque;

T_Lis the load torque; b is a viscosity coefficient; p is a radical of_mThe number of pole pairs of the motor is shown; Ψ_α、Ψ_βIs a stator flux linkage under an alpha-beta coordinate system; t is a time variable; j is load moment of inertia;

the mathematical model of the permanent magnet synchronous motor under a d-q coordinate system is as follows:

u_d＝R_si_d+pψ_d-w_eψ_q

u_q＝R_si_q+pψ_q+w_eψ_d

ψ_d＝L_di_d+ψ_r

ψ_q＝L_qi_q

T_e＝1.5p_m(ψ_ri_q+(L_d-L_q)i_di_q)

in the formula u_d、u_qIs the stator voltage under a d-q coordinate system; i.e. i_d、i_qIs the stator current under a d-q coordinate system; Ψ_d、Ψ_qIs a stator flux linkage under a d-q coordinate system; l is_d、L_qArmature inductances of d and q axes, respectively;

the traditional deadbeat current prediction control model is established as follows:

i_d-p(k+1)＝i_d(k)×(1-T_s×R_s/L_s)+i_q(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q(k)×(1-T_s×R_s/L_s)-i_d(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_r/L_s

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×ψ_r)

in the formula i_d-p(k +1) d-axis predicted current, i, at time k +1_q-p(k +1) is the predicted current of the q-axis at the time k +1, i_d-refFor d-axis reference current, i_q-refFor q-axis reference current, u_d-p(k +1) d-axis predicted voltage at time k +1, u_q-p(k +1) is the predicted voltage of the q-axis at the time k +1,T_sis a switching cycle.

Further, the third step is specifically:

introducing a nonlinear active disturbance rejection control fal function to establish an extended state observer to estimate real-time disturbance, and specifically comprising the following steps:

for the second rising edge in a certain k-1 moment, the corresponding d-q axis current predicted value is as follows:

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_e×Ψ_f/L_s

e_d(k-1)＝i_d-p(t₂)-i_d

e_q(k-1)＝i_q-p(t₂)-i_q

wherein, t₂For the moment at said second rising edge, i.e. the moment at which the three-phase current is reconstructed from the bus current, i_d-p(t₂) And i_q-p(t₂) Is said t₂Predicted values of d and q axis currents, i_d-p(k-1) and i_q-p(k-1) is the predicted value of d and q axis currents at the initial moment of the k-1 period, u_d(k-1) and u_q(k-1) voltages applied to d and q axes at the time k-1, i_dAnd i_qAre each t₂D-and q-axis currents, i.e. t, reconstructed from the bus current at that moment₂Actual values of the d and q axis currents at the time;

the method comprises the following steps of considering factors such as parameter mismatch, dead time, non-whole period prediction and the like for the total disturbance of the motor, and determining according to motor parameters and a switching period: alpha is alpha₁、α₂、β₁、β₂、β₃、β₄、β₅、β₆Δ several observer parameters; thend. The total perturbation of the q-axis voltage can be expressed as:

the introduced fal function has the following relationship:

d and q axis currents at the k moment and the k +1 moment and disturbance values at the k +1 moment are obtained through prediction:

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)/L_s×f_d(k-1)-fal _d1

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)×w_e×Ψ_f/L_s-T_s-t₂)/L_s×f_q(k-1)-fal _q1

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_d(k)+i_q-p(k)×T_s×w_e-T_s/L_s×f_d(k)-fal _d2

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_q(k)-i_d-p(k)×T_s×w_e-T_s×w_e×Ψ_f/L_s-T_s/L_s×f_q(k)-fal _q2

predicting d and q voltages of the (k +1) th switching period and compensating by using a disturbance value:

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×Ψ_f)

u_d(k+1)＝u_d-p(k+1)+f_d(k+1)

u_q(k+1)＝u_q-p(k+1)+f_q(k+1)

the dead-beat predictive control method for the single current sensor of the permanent magnet synchronous motor has the characteristic of good parameter disturbance inhibition, and can completely replace the three-phase current of the motor to carry out predictive control.

Compared with the prior art, the method of the invention at least has the following beneficial effects:

(1) the method provides improved dead-beat predictive control, effectively inhibits the influence caused by the mismatching of the motor parameters, and enables the system to have good control characteristics.

(2) The method combines improved dead-beat predictive control with phase current reconstruction based on the bus current sensor, reduces the cost of a control system, reduces the volume of the control system, and avoids the limitation of the difference between different sensors on the performance improvement of the whole system.

Drawings

FIG. 1 is a block diagram of a system model corresponding to the method of the present invention.

Fig. 2 is a comparison of d and q-axis currents and reference currents obtained based on the present invention in the case of a resistance mismatch (Rs ═ 10 Rs');

FIG. 3 shows flux linkage mismatch (Ψ)_f＝2Ψ_f') the d and q-axis currents obtained based on the invention are compared with a reference current.

Fig. 4 shows a comparison of the d-axis and q-axis currents obtained according to the invention with a reference current in the case of an inductance mismatch (Ls ═ 1.5 Ls').

Detailed Description

The technical solutions of the present invention will be described clearly and completely with reference to the accompanying drawings, and it should be understood that the described embodiments are some, but not all embodiments of the present invention. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.

The invention provides a dead-beat prediction control method for a permanent magnet synchronous motor single current sensor, which specifically comprises the following steps as shown in figure 1:

step one, collecting the bus current i of the inverter in real time on line_dcThe rotor rotating speed w and the rotor position angle theta, and the real-time switching state of the inverter and the collected bus current i of the inverter are utilized_dcThree-phase current i is reconstructed_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a traditional deadbeat current prediction control model, and predicting d and q axis voltages of the motor at the next moment;

introducing a nonlinear active disturbance rejection control fal function to establish an extended state observer, observing the total disturbance of the motor as the extended state quantity, and compensating the d-q axis voltage at the next moment obtained by dead-beat current prediction control;

and step four, calculating the reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the step one, the reconstructed three-phase current and the compensated d-axis and q-axis voltages.

Conventional deadbeat current predictive control is based on a current reference i_refAnd the voltage vector u (k) applied to the motor at the current moment and the motor parameters, and outputting the motor reference voltage u (k +1) at the next moment. At the k-th instant u (k +1) is calculated and applied at the next instant, so that the motor current reaches the current reference value at the instant k + 2. Thus, u (k +1) can be calculated at the kth time according to the following relation:

u_d(k+1)＝(2T_sR_s-2L_s)wi_q(k)-(L_s/T_s+T_sR_sR_s/L_s-2R_s)i_d(k)+T_sL_swwi_d(k)-T_swu_q(k)-(1-T_sR_s/L_s)u_d(k)+wwT_sψ_f

u_q(k+1)＝L_s/T_s×i_q-ref-(L_s/T_s+T_sR_sR_s/L_s-2R_s)i_q(k)-(2T_sR_s-2L_s)i_d(k)i_q(k)+T_sL_swwi_q(k)+T_swu_d(k)-(1-T_sR_s/L_s)u_q(k)+w(2ψ_f-T_sR_sψ_f/L_s)

in the formula T_sIs a control period; i.e. i_q-refIs a q-axis reference current.

When the calculated reference voltage at the k +1 moment exceeds the maximum output voltage limit of the SVPWM, the output reference voltage needs to be adjusted to obtain the reference voltage within the SVPWM output range:

in the formula u_d-j、u_q-jThe calculated stator reference voltage under the d-q coordinate system; u. of_d-x、u_q-xThe reference voltage within the corrected SVPWM output voltage range under the d-q coordinate system is obtained; u. of_dcIs the dc bus voltage.

It can be seen that the traditional dead-beat predictive control has high dependence on the motor model and parameter accuracy, and once the motor has parameter mismatch in the running process, the control effect is rapidly reduced. In addition, because a single current sensor is adopted, bus current is collected twice at different voltage vector action moments (namely the first two rising edges) in the first half period of a switching period according to an SVPWM seven-segment type modulation mode, the moment is not at the initial moment of each switching period, the fact means that for the dead-beat prediction control of each period, only current values reconstructed at the rising edge moment of the last period can be utilized, and the time span from the rising edge moment of the last period to the initial moment of the period is not the whole period.

The invention changes the traditional dead beat prediction control, changes the traditional dead beat control from two-beat control to three-beat control aiming at the characteristics of a single current sensor, adopts an extended state observer, takes the total disturbance of a motor, namely the disturbance caused by factors such as parameter mismatch, dead time, non-whole period prediction and the like as an extended state quantity, and the extended state quantity can change along with the voltage disturbance in real time and compensate the output d-axis voltage and q-axis voltage.

Taking a d-axis current equation as an example, the building process of the extended observer introducing the fal function is specifically as follows:

the d-axis current equation of the permanent magnet synchronous motor is as follows:

considering the d-axis disturbance, let it be f, and let

The d-axis current equation of the permanent magnet synchronous motor can be rewritten as follows:

writing the disturbance into a state equation form, expanding the unknown disturbance f into a new state variable according to the theory of the extended observer, and setting

x₁＝i_d

x₂＝f

u＝u_dIs an input to the system

The state equation can be written as:

for a system equation of state in the form of the above, its extended state observer can be designed as:

wherein,

e＝z₁-x₁

the expression of the fal function is:

expanded state z₂A good estimate of the real-time contribution of the unknown disturbance can be made.

State z to be expanded₂Is discretized by the expression of

Then z is put₁Is discretized by the expression of

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_d(k)+i_q-p(k)×T_s×w_e-T_s/L_s×f_d(k)-fal _d2

Then according to the principle of deadbeat control, the voltage output by the d-axis at the time k +1 should be:

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

estimation f of real-time effect of the extended observer on unknown disturbance_d(k +1) to the d-axis voltage of the above formula,

u_d(k+1)＝u_d-p(k+1)+f_d(k+1)

the voltage which should be output at the moment k +1 is obtained under the consideration of factors such as parameter disturbance and the like.

In the same way, an extended observer expression of the q-axis current equation and a k +1 time output voltage expression can also be obtained.

It should be noted that, because a single current sensor is adopted, for the dead-beat prediction control of each period, only the current value reconstructed at the rising edge time of the previous period can be used, which means that the traditional dead-beat control needs to be changed from two-beat control to three-beat control, taking d-axis current as an example, namely the current prediction value i at the aforementioned k time_d-p(k) The current actual value and the current predicted value at the current reconstruction moment in the previous period need to be obtained according to the difference between the actual value and the predicted value, and the current actual value and the current predicted value are obtained by using an extended observer:

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

e_d(k-1)＝i_d-p(t₂)-i_d

since the reconstruction of the k-1 cycle current is at t₂Is done at a time, which means that the current error calculated for that cycle is also t₂Current error at time, so next at pair f_d(k) When the expression of (a) is discretized, it should be divided into 0-t₂And t₂-T_sTwo periods of time, i.e. 0 to t₂Within time f_d(k) The current error e that should be actually calculated from the k-2 period_d(k-2) determination, t₂To T_sCurrent error e calculated from k-1 period only in time_d(k-1). Thus f_d(k) The discretization expression of (a) is:

then i_d-p(k) The expression of (a) is:

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)/L_s×f_d(k-1)-fal _d1

the derivation processes are arranged and summarized according to the time sequence, so that a new deadbeat current prediction control model can be obtained:

calculating d-axis and q-axis current errors of a last period:

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_eΨ×_f/L_s

e_d(k-1)＝i_d-p(t₂)-i_d

e_q(k-1)＝i_q-p(t₂)-i_q

secondly, taking the total disturbance of the motor, namely the disturbance caused by factors such as parameter mismatch, dead time, non-whole period prediction and the like as an expansion state quantity, introducing a nonlinear fal function, and discretizing an expression of the motor disturbance:

respectively establishing an extended state observer for d and q axes of the motor, and performing first two-beat operation in three-beat dead-beat control:

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)/L_s×f_d(k-1)-fal _d1

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)×w_e×Ψ_f/L_s-(T_s-t₂)/L_s×f_q(k-1)-fal _q1

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_d(k)+i_q-p(k)×T_s×w_e-T_s/L_s×f_d(k)-fal _d2

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_q(k)-i_d-p(k)×T_s×w_e-T_s×w_e×Ψ_f/L_s-T_s/L_s×f_q(k)-fal _q2

fourthly, performing third beat calculation of dead beat control, and compensating the d-axis voltage and the q-axis voltage at the moment of k +1 according to the disturbance calculated in the third beat:

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×Ψ_f)

u_d(k+1)＝u_d-p(k+1)+f_d(k+1)

u_q(k+1)＝u_q-p(k+1)+f_q(k+1)

fig. 2-4 show the resistance mismatch (Rs 10 Rs'), flux mismatch (ψ), in a preferred embodiment of the present invention_f＝2ψ_f') and inductance mismatch (Ls ═ 1.5 Ls') are different, the d and q axis actual currents obtained based on the present invention are compared with the reference current.

It should be understood that, the sequence numbers of the steps in the embodiments of the present invention do not mean the execution sequence, and the execution sequence of each process should be determined by the function and the inherent logic of the process, and should not constitute any limitation on the implementation process of the embodiments of the present invention.

Although embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that changes, modifications, substitutions and alterations can be made in these embodiments without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.

Claims (3)

1. A dead-beat prediction control method for a permanent magnet synchronous motor single current sensor is characterized by comprising the following steps: the method specifically comprises the following steps:

step one, collecting the bus current i of the inverter in real time on line_dcThe rotor rotating speed w and the rotor position angle theta, and the real-time switching state of the inverter and the collected bus current i of the inverter are utilized_dcThree-phase current i is reconstructed_a、i_b、i_c；

Establishing mathematical models of the permanent magnet synchronous motor in an alpha-beta coordinate system and a d-q coordinate system, establishing a traditional deadbeat current prediction control model, and predicting d and q axis voltages of the motor at the next moment;

introducing a nonlinear active disturbance rejection control fal function to establish an extended state observer, observing the total disturbance of the motor as the extended state quantity, and compensating the d-q axis voltage at the next moment obtained by dead-beat current prediction control;

and step four, calculating the reference voltage within the SVPWM output voltage range at the next moment in real time by combining the data collected in the step one, the reconstructed three-phase current and the compensated d-axis and q-axis voltages.

2. The method of claim 1, wherein: the mathematical model of the permanent magnet synchronous motor established in the second step under the alpha-beta coordinate system is as follows:

u_α＝R_si_α+L_spi_α-w_eψ_rsinθ

u_β＝R_si_β+L_spi_p+w_eψ_rcosθ

ψ_α＝L_si_α+ψ_rcosθ

ψ_β＝L_si_β+ψ_rsinθ

T_e＝1.5p_mψ_r(i_βcosθ-i_αsinθ)

in the formula u_α、u_βIs the stator voltage under an alpha-beta coordinate system; i.e. i_α、i_βIs the stator current under an alpha-beta coordinate system; Ψ_rIs a rotor flux linkage; r_sIs a stator resistor; l is_sIs a stator inductance; w is a_e、w_mThe electrical angular velocity of the rotor and the mechanical angular velocity of the rotor, respectively; theta is a rotor position angle; p is a differential operator; t is_eIs an electromagnetic torque; t is_LIs the load torque; b is a viscosity coefficient; p is a radical of_mThe number of pole pairs of the motor is shown; Ψ_α、Ψ_βIs a stator flux linkage under an alpha-beta coordinate system; t is a time variable; j is load moment of inertia;

the mathematical model of the permanent magnet synchronous motor under the d-q coordinate system is as follows:

u_d＝R_si_d+pψ_d-w_eψ_q

u_q＝R_si_q+pψ_q+w_eψ_d

ψ_d＝L_di_d+ψ_r

ψ_q＝L_qi_q

T_e＝1.5p_m(ψ_ri_q+(L_d-L_q)i_di_q)

in the formula u_d、u_qIs the stator voltage under a d-q coordinate system; i.e. i_d、i_qIs the stator current under a d-q coordinate system; Ψ_d、Ψ_qIs a stator flux linkage under a d-q coordinate system; l is_d、L_qArmature inductances of d and q axes, respectively;

the traditional deadbeat current prediction control model is as follows:

i_d-p(k+1)＝i_d(k)×(1-T_s×R_s/L_s)+i_q(k)×T_s×w_e+T_s/L_s×u_d(k)

i_q-p(k+1)＝i_q(k)×(1-T_s×R_s/L_s)-i_d(k)×T_s×w_e+T_s/L_s×u_q(k)-T_s×w_e×ψ_r/L_s

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×ψ_r)

in the formula i_d-p(k +1) d-axis predicted current, i, at time k +1_q-p(k +1) is the predicted current of the q-axis at the time k +1, i_d-refFor d-axis reference current, i_q-refFor q-axis reference current, u_d-p(k +1) d-axis predicted voltage at time k +1, u_q-p(k +1) is the predicted voltage of the q-axis at time k +1, T_sIs a switching cycle.

3. The method of claim 2, wherein: the third step is specifically as follows:

introducing a nonlinear active disturbance rejection control fal function to establish an extended state observer to estimate real-time disturbance, and specifically comprising the following steps:

for the second rising edge in a certain k-1 moment, the corresponding d-q axis current predicted value is as follows:

i_d-p(t₂)＝i_d-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_d(k-1)+i_q-p(k-1)×t₂×w_e

i_q-p(t₂)＝i_q-p(k-1)×(1-t₂×R_s/L_s)+t₂/L_s×u_q(k-1)-i_d-p(k-1)×t₂×w_e-t₂×w_e×Ψ_f/L_s

e_d(k-1)＝i_d-p(t₂)-i_d

e_q(k-1)＝i_q-p(t₂)-i_q

wherein, t₂For the moment at said second rising edge, i.e. the moment at which the three-phase current is reconstructed from the bus current, i_d-p(t₂) And i_q-p(t₂) Is said t₂Predicted values of d and q axis currents, i_d-p(k-1) and i_q-p(k-1) is the predicted value of d and q axis currents at the initial moment of the k-1 period, u_d(k-1) and u_q(k-1) voltages applied to d and q axes at the time k-1, i_dAnd i_qAre each t₂D-and q-axis currents, i.e. t, reconstructed from the bus current at that moment₂Actual values of the d and q axis currents at the time;

the method comprises the following steps of considering factors such as parameter mismatch, dead time, non-whole period prediction and the like for the total disturbance of the motor, and determining according to motor parameters and a switching period: alpha is alpha₁、α₂、β₁、β₂、β₃、β₄、β₅、β₆Δ several observer parameters; the total perturbation of the d, q-axis voltages can be expressed as:

the introduced fal function has the following relationship:

d and q axis currents at the k moment and the k +1 moment and disturbance values at the k +1 moment are obtained through prediction:

i_d-p(k)＝i_d-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_d(k-1)+i_q-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)/L_s×f_d(k-1)-fal_d1

i_q-p(k)＝i_q-p(t₂)×(1-(T_s-t₂)×R_s/L_s)+(T_s-t₂)/L_s×u_q(k-1)-i_d-p(t₂)×(T_s-t₂)×w_e-(T_s-t₂)×w_e×Ψ_f/L_s-(T_s-t₂)/L_s×f_q(k-1)-fal_q1

i_d-p(k+1)＝i_d-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_d(k)+i_q-p(k)×T_s×w_e-T_s/L_s×f_d(k)-fal_d2

i_q-p(k+1)＝i_q-p(k)×(1-T_s×R_s/L_s)+T_s/L_s×u_q(k)-i_d-p(k)×T_s×w_e-T_s×w_e×Ψ_f/L_s-T_s/L_s×f_q(k)-fal_q2

predicting d and q voltages of the (k +1) th switching period and compensating by using a disturbance value:

u_d-p(k+1)＝L_s/T_s×(i_d-ref-(1-T_s×R_s/L_s)×i_d-p(k+1)-T_s×i_q-p(k+1)×w_e)

u_q-p(k+1)＝L_s/T_s×(i_q-ref-(1-T_s×R_s/L_s)×i_q-p(k+1)+T_s×i_d-p(k+1)×w_e+T_s×w_e/L_s×Ψ_f)

u_d(k+1)＝u_d-p(k+1)+f_d(k+1)

u_q(k+1)＝u_q-p(k+1)+f_q(k+1)。

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