CN111525828A - Control method of bidirectional isolation type resonant power converter based on virtual synchronous motor - Google Patents

Control method of bidirectional isolation type resonant power converter based on virtual synchronous motor Download PDF

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CN111525828A
CN111525828A CN202010431653.0A CN202010431653A CN111525828A CN 111525828 A CN111525828 A CN 111525828A CN 202010431653 A CN202010431653 A CN 202010431653A CN 111525828 A CN111525828 A CN 111525828A
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converter
current
voltage
power
mode
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CN111525828B (en
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任春光
贾燕冰
徐浩祥
孟祥齐
张佰富
韩肖清
秦文萍
郭东鑫
孔健生
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Taiyuan University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/66—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal
    • H02M7/68—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters
    • H02M7/72—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/79—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/797—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/60—Monitoring or controlling charging stations
    • B60L53/63—Monitoring or controlling charging stations in response to network capacity
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L55/00—Arrangements for supplying energy stored within a vehicle to a power network, i.e. vehicle-to-grid [V2G] arrangements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/80—Technologies aiming to reduce greenhouse gasses emissions common to all road transportation technologies
    • Y02T10/92—Energy efficient charging or discharging systems for batteries, ultracapacitors, supercapacitors or double-layer capacitors specially adapted for vehicles
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02T90/10—Technologies relating to charging of electric vehicles
    • Y02T90/12—Electric charging stations

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)

Abstract

The invention provides a control method of a bidirectional isolation type resonant power converter based on a virtual synchronous motor, which aims to solve the problems of lack of rotational inertia in the charging and discharging processes of an electric automobile, low voltage stability of a power electronic converter, efficiency reduction caused by large reactive power in the operation process and the like. The bidirectional power converter is composed of a DC/DC level and a DC/AC level, the DC/AC level three-phase converter can be equivalent to a synchronous motor according to the structural similarity of a three-phase synchronous motor model and the three-phase converter, the whole electric vehicle charging pile is equivalent to a synchronous motor at the grid connection point of the electric vehicle charging pile, and the synchronous motor can adaptively respond to voltage and frequency disturbance of a power grid and provide necessary inertia and damping for the power grid. In order to overcome the defect that the power loss is caused by large reactive current of the traditional DAB converter, the zero voltage conduction and the zero current turn-off of a switching device of the interface converter are realized by adding the resonance module, and the integral operation efficiency of the converter is improved.

Description

Control method of bidirectional isolation type resonant power converter based on virtual synchronous motor
Technical Field
The invention relates to the field of bidirectional control of interaction between a large power grid and a power battery of an electric vehicle, in particular to a bidirectional isolation type resonant power converter control method based on a virtual synchronous motor, which is suitable for realizing friendly and efficient interaction and bidirectional flow of energy between the electric vehicle and the power grid.
Background
The shortage of fossil energy and concern over air pollution have accelerated vehicle electrification. A large number of electric automobiles are interconnected with a power grid, so that the impact of intermittent renewable energy on the power grid is stabilized, and the electric automobile is also one of effective emergency power supply substitution solutions and is generally accepted and popular all over the world. The performance of the charging and discharging device of the electric automobile is a key part for ensuring the charging and discharging efficiency and speed of the electric automobile and friendly interaction with a power grid.
The bidirectional interface converter capable of regulating bidirectional power flow is an important component of an electric automobile charger. For the bidirectional interface converter, on one hand, the electric vehicle charging/discharging equipment and the power distribution network are required to have good interaction characteristics, and the power distribution network has high stability and steady-state precision when a transient fault occurs. The researchers proposed a bidirectional droop control method using frequency and voltage at both sides of ac/dc to control power flow direction, so that both sides of ac/dc can bear load in a balanced manner, but when the permeability of the electric vehicle is gradually increased, droop control may impact the stability of both the battery and the power grid of the electric vehicle. In order to increase the stability of the system, in recent years, a virtual synchronous motor control theory and a method of the inverter have been proposed, and a three-phase converter is equivalent to a virtual synchronous motor. The scholars have studied the implementation of a virtual synchronous generator with virtual inertia and the control strategy as an inverter power supply, but the interface converter in the scheme can only be applied to a single power flow direction and can only be applied to a power electronic converter in a specific occasion.
On the other hand, the charging and discharging process of the charging equipment of the electric automobile is required to be as rapid and efficient as possible, so that the service life and the operation safety of the power battery of the electric automobile are improved, and the energy loss in the charging and discharging process is reduced. To improve the efficiency of a bidirectional interface converter, it should meet a variety of requirements, such as wide output voltage regulation, low electrical stress, bufferless circuits, low circulating current and good switching conditions, and buck/boost operation. The isolation/bidirectional PWM converter structure is adopted, the buck/boost operation is met, the bidirectional power flow is realized, but a high-frequency inverter on a current feed side is subjected to strong voltage stress caused by the leakage inductance of the converter, and the isolation/bidirectional PWM converter structure is a main obstacle for improving the efficiency of a bidirectional isolation type converter. Therefore, for a bidirectional converter operating at high voltage and high power, a DAB (dual-active-bridge) structure suitable for various voltages and high power is adopted, which can significantly improve the transmission power of the interface converter, however, the conventional DAB converter has large reactive current, which generates electrical stress on its switching elements and increases power loss, so that the overall efficiency of the interface converter is reduced. Therefore, there are still many defects in the related control technology of the existing converter in the charging and discharging of the electric vehicle, and a novel control method capable of improving the operation efficiency and friendly interacting with the power grid is needed for the bidirectional power converter.
Disclosure of Invention
The invention provides a control method of a bidirectional isolation type resonant power converter based on a virtual synchronous motor, which aims to solve the problems of lack of rotational inertia in the charging and discharging processes of an electric automobile, low voltage stability of a power electronic converter, efficiency reduction caused by large reactive power in the operation process and the like.
The invention is realized by the following technical scheme: a control method of a bidirectional isolation type resonance power converter based on a virtual synchronous motor is characterized in that a power grid alternating current bus passes through line impedance ZacFilter resistor RacAnd the LC filter is connected to the AC side of the AC interface converter; the DC side of the AC interface converter passes through a DC capacitor CdcConnecting a DC/DC converter; the DC/DC converter passes through a voltage stabilizing capacitor CfAnd a filter inductor LfFinally, connected to a power battery; the control method virtualizes the AC interface converter as a synchronous motor according to the structural similarity of a three-phase synchronous motor model and the AC interface converter, the control method comprises three parts of active power control, virtual excitation control and voltage and current double closed-loop control, and the control method of each part is as follows:
(1) active power control: setting the pole pair number of the virtual synchronous motor to be 1, the torque equation can be expressed as:
Figure BDA0002500814240000021
wherein J represents the moment of inertia of the synchronous machine in kg.m2，ωNExpressing the AC rated angular speed of the power grid in unit rad/s; peAnd PmElectromagnetic and mechanical power of the synchronous motor respectively; is the power angle of the generator, unit rad; omega is the virtual rotor angular frequency of the synchronous motor, in units rad/s; k is a radical ofωIs an AC primary frequency modulation droop coefficient; the active power control part is mainly used for realizing active power closed loop and generating mechanical torque; the active power is calculated from the ac side voltage and current and is expressed as:
P＝uaia+ubib+ucic
in the formula ua、ub、ucIs the terminal voltage of the synchronous machine ia、ib、icIs the terminal current of the synchronous motor;
(2) virtual excitation control: in the virtual excitation control part, the excitation control of the generator is simulated, the alternating voltage and the reactive power are controlled and adjusted, and the virtual potential effective value E of the virtual synchronous motor model is adjusted to generate reactive power; the virtual potential effective value E of the virtual synchronous motor is composed of 3 parts in total:
one, part Δ E of reactive power regulationQThe second is the part of the voltage regulation at the end of the reactor, Delta EUThe automatic excitation regulator can be equivalent to a synchronous motor, and the third is the effective value E of the no-load potential of the synchronous motor0(ii) a The virtual potential effective value of the motor is as follows:
E＝E0+ΔEQ+ΔEU
the vector value of the motor virtual potential is expressed as:
Figure BDA0002500814240000031
(3) voltage current double closed loop control: based on KVL's law, the electromagnetic equation for a synchronous machine can be expressed as:
Figure BDA0002500814240000032
wherein
Figure BDA0002500814240000033
Is an alternating side voltage; l and R are respectively stator inductance and resistance of synchronous motor, and the values are respectively taken as filter inductance L of LC filter of AC interfaceacAnd a filter resistor RacThe value of (a) is,
Figure BDA0002500814240000034
is the current of the alternating-current interface,
Figure BDA0002500814240000035
is an alternating bus side current; obtaining a signal e through voltage-current double closed-loop control and carrying out the controlThe modulation wave is input into SPWM for modulation, a control signal of the AC interface converter is generated, and the on and off of each IGBT tube of the AC interface converter are controlled.
The bidirectional power converter is composed of a DC/DC level and a DC/AC level, the DC/AC level three-phase converter can be equivalent to a synchronous motor according to the structural similarity of a three-phase synchronous motor model and the three-phase converter, the whole electric vehicle charging pile is equivalent to a synchronous motor at the grid-connected point of the electric vehicle charging pile, and the motor can adaptively respond to voltage and frequency disturbance of a power grid and provide necessary inertia and damping for the power grid.
Further, a primary side and a secondary side of a DC/DC converter on a DC side are connected by a CLC resonance module including a capacitor C for voltage multiplication operation on the primary sidevAnd a resonant capacitor C on the secondary side for resonant PWM operationrAnd a resonant inductor LrA resonant structure of composition; the DC/DC converter has 8 switches for charge or discharge operation, including M on the primary side1、M2、M3、M4Four IGBT tubes and M on secondary side5、M6、M7、M8And four IGBT tubes.
The DC/DC level of the interface converter adopts the structure of a DAB converter to meet the requirements of high power and wide output voltage range. In order to solve the defect that the traditional DAB converter has power loss caused by large reactive current, the CLC resonance module is added to realize zero voltage conduction and zero current turn-off of a switching device of the interface converter, so that the overall operation efficiency of the converter is improved.
The direct current control unit of the bidirectional power converter is provided with 8 switches for charging or discharging operation, and different from the traditional resonant structure, the DC/DC converter is only controlled by PWM, so that the problems of efficiency reduction caused by overhigh switching frequency or audible noise or no-load regulation caused by overlow switching frequency can be avoided. The bidirectional charger proposed herein maintains the structural advantages similar to those of a DAB-converter, while simultaneously using a voltage-doubler rectification structure, i.e. M is added during the discharge operation3Kept in a conducting state to make the interface converter direct currentThe secondary side voltage is increased to twice the original voltage, so that the converter realizes bidirectional power flow.
Compared with the prior art, the invention has the following beneficial effects:
(1) meanwhile, the requirements of the voltage of a power grid and the voltage stability of a battery of the electric automobile in the charging and discharging processes of the electric automobile are considered, and the voltage control of an alternating current/direct current bus can be realized simultaneously in the control process;
(2) the converter can realize stable control of power, carry out stable charging and discharging operation on the electric automobile, and enable the whole system to have higher stability and inertia when the power grid generates larger fluctuation;
(3) meanwhile, the direct current side has the characteristics of wide output voltage range and high-power transmission capability, zero-voltage conduction and zero-current turn-off can be well realized through the design of the resonant circuit, the conversion efficiency of the converter is effectively improved, and the requirements of efficient charging and discharging of the electric automobile are met;
(4) in addition, the charge-discharge state change can be effectively controlled through the change of the single switch, the operation can be simple and convenient in practical engineering application, and the safety is improved.
Drawings
Fig. 1 is a topology diagram of a bidirectional power converter.
FIG. 2 is a control block diagram of an AC-side virtual synchronous machine control-based bidirectional power converter
Fig. 3 is a control block diagram of a dc-side bidirectional power converter.
Fig. 4 is a schematic diagram of the dc-side bi-directional power converter in a charging operation.
Fig. 5 is a schematic diagram of the dc-side bi-directional power converter in discharging operation.
Fig. 6 is a waveform diagram of a dc-side bi-directional power converter during charge and discharge operations.
Detailed Description
The following describes an embodiment of the present invention with reference to the drawings.
The embodiment is mainly used for a bidirectional power converter serving as a charging and discharging machine of an electric automobile, and the structural topology of a bidirectional interface converter is as followsAs shown in fig. 1. Line impedance Z of AC bus of power gridacAnd LC filter (L)acIs a filter inductor, CacIs a filter capacitor, RacFilter resistance) to the ac side of the interface converter; the DC side of the interface converter passes through a DC capacitor CdcThe DC/DC converter is connected, a CLC resonance module is connected between the primary side and the secondary side of the DC side of the interface converter, and the CLC resonance module are connected through a voltage stabilizing capacitor CfAnd a filter inductor LfAnd finally connected to a power battery electric vehicle battery.
The AC side adopts a power control method based on a virtual synchronous motor, the control block diagram of the power control method is shown in figure 2, an AC control unit of the power control method comprises active power control, virtual excitation control and voltage and current double closed-loop control, and the control method of each part is as follows:
(1) active power control: setting the pole pair number of the virtual synchronous motor to be 1, the torque equation can be expressed as:
Figure BDA0002500814240000051
wherein J represents the moment of inertia of the synchronous machine in kg.m2，ωNExpressing the AC rated angular speed of the power grid in unit rad/s; t ise、TmAnd TdElectromagnetic torque, mechanical torque and damping torque of the synchronous motor respectively; d is damping coefficient, and when the action of the damping winding is not considered, the damping coefficient is matched with an AC primary frequency modulation droop coefficient kωEqual; is the power angle of the generator, unit rad; ω is the virtual rotor angular frequency of the synchronous machine, in units rad/s. Due to the moment of inertia J, the charge/discharge machine exhibits the capacity for mechanical inertia in the event of fluctuations in the grid voltage. When the alternating current droop coefficient is selected, the inertia of the alternating current frequency can be increased by increasing the inertia moment J, but the system stability is reduced by increasing the inertia moment J, so that the inertia moment J is not suitable for being increased excessively. In addition, the increase of the damping coefficient D can increase the inertia of the whole system, and the same excessive D can affect the stability of the system. And the electromagnetic torque of the motor can be controlled by the virtual synchronous motor potential eabcAnd an output current iabcGet electromagnetic torque and virtual identityElectromagnetic power P output by step generatoreThe relationship between can be expressed as:
Te＝Pe/ω＝(eaia+ebib+ecic)/ω
rated mechanical torque T of synchronous motor under constant power load condition with power P0With the frequency omega of the power supply networkNThe frequency and the rotating speed are in positive correlation, and therefore the mechanical damping is increased or decreased with the same amplitude, namely the frequency of the response power grid is changed. In view of this, we can adjust the virtual synchronous machine mechanical torque TmTo regulate the active command, mechanical torque T, in the network-side converter interfacemIs a fixed torque T0The difference from the frequency deviation feedback command △ T, therefore, the mechanical torque can be expressed as:
Tm＝T0+ΔT＝(P-Pref)/ωN
wherein, PrefThe active power control section is primarily used to implement an active power closed loop and generate mechanical torque for a selected reference power. The active power is calculated from the ac side voltage and current and can be expressed as:
P＝uaia+ubib+ucic
and (3) integrating the value obtained by subtracting the electromagnetic torque and the damping torque from the mechanical torque, and dividing the value by the moment of inertia J to further obtain the virtual rotor angular frequency omega of the synchronous motor, and continuously integrating the rotor angular frequency to obtain the power angle of the virtual synchronous generator.
(2) Virtual excitation control: in the virtual excitation control part, the excitation control of the generator is simulated, the alternating voltage and the reactive power are controlled and adjusted, and the virtual potential effective value E of the virtual synchronous motor model is adjusted to generate reactive power. The virtual potential effective value E of the virtual synchronous machine is composed of 3 parts in total.
First, is a reactionPart of the reactive power regulation Δ EQThe second is the part of the voltage regulation at the end of the reactor, Delta EUThe automatic excitation regulator can be equivalent to a synchronous motor, and the third is the effective value E of the no-load potential of the motor0。
Figure BDA0002500814240000061
Wherein k isqIs a reactive-voltage droop coefficient, QrefAnd Q is the instantaneous reactive power reference value and the actual value output by the AC interface terminal respectively. k is a radical ofvIs a voltage regulation factor; u shaperefAnd U is a reference value and a real value of the effective value of the grid-connected inverter terminal line voltage respectively. The virtual potential effective value of the motor is as follows:
E＝E0+ΔEQ+ΔEU
the vector value of the motor virtual potential is expressed as:
Figure BDA0002500814240000062
(3) voltage current double closed loop control: based on KVL law and KCL law, the electromagnetic equation of a synchronous machine can be expressed as:
Figure BDA0002500814240000063
wherein
Figure BDA0002500814240000064
Is an alternating side voltage; l and R are the stator inductance and resistance of the synchronous machine, respectively. It should be noted that the stator inductance L and the resistor R are connected to the filter inductance L of the ac interfaceacAnd a filter resistor RacIn response to this, the mobile terminal is able to,
Figure BDA0002500814240000065
is the current of the alternating-current interface,
Figure BDA0002500814240000066
is an alternating bus side current. The active power controls the mechanical motion equation of the analog synchronous motor, the active power is adjusted, and the control link outputs virtual potential
Figure BDA0002500814240000067
Frequency and phase of; the virtual excitation control simulates the excitation regulation of the synchronous motor, controls the reactive power and outputs
Figure BDA0002500814240000071
Is determined.
To simplify the control, the mathematical model of the interface converter is converted from an abc three-phase stationary coordinate system to a two-phase rotating d-q coordinate system. The transformation matrix is as follows:
Figure BDA0002500814240000072
in the formula, θ is ω t + θ0Denotes the angle between the d-axis and the a-axis, theta0The included angle when t is 0.
And obtaining a mathematical model of the interface converter under a d-q coordinate system through coordinate conversion, so that a three-phase alternating current component in the interface converter is changed into a two-phase direct current component. Then there are:
Figure BDA0002500814240000073
in the formula ud、uqRespectively representing alternating voltages
Figure BDA0002500814240000074
D, q axis components of (1); e.g. of the typed、eqRespectively representing alternating voltages
Figure BDA0002500814240000075
D, q axis components of (1); i.e. idAnd iqRespectively representing alternating currents
Figure BDA0002500814240000076
D, q axis components of (1); i.e. ildAnd ilqRespectively representing alternating currents
Figure BDA0002500814240000077
D, q-axis components of (1).
According to the above formula, as shown in FIG. 2(b), the coupling term in the voltage equation is ω LiqAnd ω LidThe coupled term of the current equation is ω CuqAnd ω Cud. The coupling influence between d and q axes is eliminated by introducing a negative coupling term into the control; the interface converter is enabled to track the reference signal without a static error by utilizing PI control. In the double closed loop shown in fig. 2(b), the controlled variables of the voltage loop and the current loop are respectively voltage and current, and the transfer functions when PI control is adopted are respectively as follows:
Figure BDA0002500814240000078
Figure BDA0002500814240000079
wherein u isd-ref、uq-refD, q-axis components, i, of a given AC voltage reference value, respectivelyd-ref、iq-refD, q-axis components, e, respectively, of a given AC voltage referenced-ref、eq-refThe d and q-axis components of a given ac voltage reference, respectively. Last item id、iq、udAnd uqThe feedforward terms are d-axis components and q-axis components of measured values of voltage and current, and can accelerate the response of the controller. Gu(s) and Gi(s) are all PI regulators, and the transfer functions are respectively as follows:
Figure BDA0002500814240000081
Figure BDA0002500814240000082
in the formula, Ku-P、Ku-IThe proportional and integral coefficients of the voltage loop are provided. Ki-P、Ki-IRespectively, the proportional and integral coefficients of the current loop.
Finally, a signal e is obtained through voltage-current double closed-loop control, and is input into SPWM for modulation as a modulation wave to generate a control signal of the interface converter to control the VT of the IGBT tube1～VT6On and off.
The DC control unit of the bidirectional power converter has 8 switches for charging or discharging operation, such as M in FIG. 31～M8. By a resonant capacitor CrAnd a resonant inductor LrA capacitor C formed on the primary side and having a resonant structure on the secondary side for resonant PWM operationvFor voltage multiplication operations. Unlike conventional resonant structures, where the converter is controlled solely by PWM, efficiency degradation due to too high switching frequency, or audible noise or no-load regulation problems due to too low switching frequency can be avoided. The bidirectional charger proposed in the present application maintains the structural advantages similar to those of the DAB-converter while employing a voltage-doubler rectification structure, i.e. M is applied during the discharge operation3And keeping the converter in a conducting state, so that the voltage of the direct current secondary side of the interface converter is increased to twice of the original voltage, and the converter realizes bidirectional power flow. Wherein, VrefRepresenting the reference value of the voltage at the secondary side of the DC interface, IrefRepresenting the secondary side current reference value.
The pattern analysis plots of fig. 4 and 5 were obtained during the charging and discharging phases, respectively.
FIG. 4(a, b, and c in FIG. 4 represent modes 1, 2, and 3, respectively) is a diagram of DC-side bi-directional power converter modes in a charging operation where power flows from M1～M4Control, M5～M8The diodes of (a) are used for full bridge rectification. Due to CvHas a value of several tens of microfarads, and thus has a sufficiently large value in this mode, which can be regarded as a dc coupling capacitor, without affecting the charging operation. The mode diagram and equivalent circuit during charging are shown in fig. 4, and the drain-source capacitance C of the switch can be ignored because the drain-source capacitance of the switch device is usually smallds. While assuming a resonant capacitor voltage vcrNot exceeding the battery voltage Vbatt，M3Initially in a conducting state. The waveforms of the switching tube and the soft switch during the whole charging process are shown in fig. 6 (a).
Mode 1 (t)0≤t<t1): when M is1At t0Is on and the primary current ipFlows through M1，M3And a transformer primary side. Secondary side current isIncreases from zero and flows through Cr，Lr，M5、M7And a converter secondary side. V can be derived from the equivalent circuit of mode 1 of fig. 4(a)crAnd is:
Figure BDA0002500814240000091
Figure BDA0002500814240000092
Wherein:
Figure BDA0002500814240000093
Vcrfis vcrPeak voltage, primary current i during charging operationpIs a primary side current isAnd a magnetizing current imAnd (4) summing. When M is1When turned off, the primary current is used to control M1And M2C of (A)dsAnd charging and discharging are carried out. If M is2C of (A)dsComplete discharge before the end of mode 1, M can be achieved2Zero Voltage Switching (ZVS). And M2The ZVS condition of (1) is easily satisfied because the ZVS uses ipPeak value of (a). This operation is indicated by a light line in fig. 4 (a).
Mode 2 (t)1≤t<t2)：M2On, mode 2 starts. Primary current ipFlows through M3、M2And a transformer primary side. Secondary current isMaintain the same current path as the previous state until isUntil it is reduced to zero. FIG. 4(b) the equivalent circuit of mode 2, vcrAnd isCan be expressed as:
vcr(t)＝-Vbatt-(Vcr(t1)+Vbatt)cosωr(t-t1)+ZriLr(t1)sinωr(t-t1)
Figure BDA0002500814240000094
assuming that the dead time between the up and down gate signals is sufficiently short to be negligible, i can be obtained from the above equationLr(t1). The following were used:
Figure BDA0002500814240000095
Figure BDA0002500814240000096
in the formula DfWhen it is charging M1(or M)4) Duty cycle. From the above formula, T can be derived2MfDuration of (2)
Figure BDA0002500814240000097
Mode 3 (t)2≤t<t3): after mode 2, only the magnetizing current imBy M3And M2And (6) circulating. In mode 3, the resonant capacitor voltage vcrIs maintained as vcr(t2) Is a number VcrfThe peak of the defined resonance capacitance. Can obtain Vcrf：
Figure BDA0002500814240000101
VcrfShould not exceed VbattTo prevent M5And M7The body diode is not normally conductive, and normal operation is guaranteed. Only M3And M4C of (A)dsThe peak magnetizing current is adopted for charging and discharging. If it is notM4C of (A)dsComplete discharge before the end of mode 3, M can be achieved4ZVS of (1). And M2Different ZVS conditions of4ZVS is not easy because ZVS uses only the peak value of the magnetizing current and is affected by the load condition. The operation of the next half cycle consisting of modes 4-6 is the same as the previous half cycle.
Fig. 5 is a schematic diagram of the dc-side bi-directional power converter in discharging operation. Power flow from M during discharge5～M8And (5) controlling. In order to increase the voltage gain, a diode M is used1And M2As a voltage doubler rectifier, to make M3Is in a conducting state. Capacitor CvAs a supply capacitor for supplying power. Suppose the impedance of the secondary side magnetizing inductor of the converter is omegasLm/n2ω，(LmThe magnetizing inductance of the transformer of the resonant module in fig. 1) and this impedance is related to the resonant tank impedance ωsLm+1/(ωsCr) Is sufficiently large. Where the switching frequency omegas(means M1-M8) In units of rad/sec. The switching tube and soft switching waveform conditions during the entire charging process are shown in fig. 6 (b).
Mode 1 (t)0≤t<t1): when M8 is at t0When the switch is on, the secondary current passes through Lr、Cr、M8、M6And a converter secondary side. Primary current ipStarting from zero and increasing by M2、M3、CvAnd a transformer primary side. From FIG. 5(a), vcrAnd primary current and secondary side primary current nipCan be approximately derived as:
Figure BDA0002500814240000102
Figure BDA0002500814240000103
vcrris vcrThe peak voltage during discharge is expressed asThe last formula is shown. Secondary current equal to nipAnd nimAnd (4) summing. When M is8When closed, isFor making M7The ZVS state is reached.
Mode 2 (t)1≤t<t2): when M is7On, mode 2 starts. Power flow is shown in FIG. 5(b), ipAnd begins to fall. V is obtained from the equivalent circuit of mode 2crAnd nipComprises the following steps:
Figure BDA0002500814240000104
Figure BDA0002500814240000111
using similar assumptions as for mode 2 in the charging operation, ni can be derivedp(t1) And vcr(t1). Their derivation is as follows:
Figure BDA0002500814240000112
Figure BDA0002500814240000113
in the formula DrFor M in discharging operation5(or M)8) The duty cycle of (c). Mode 2 continues until ipReduced to zero, duration T of mode 2 in discharge mode2MrIs composed of
Figure BDA0002500814240000114
Mode 3 (t)2≤t<t3): after mode 2, only the secondary side nimPassing a magnetizing current of M7And M6. At the same time, v can be deducedcr(t2) The following were used:
Figure BDA0002500814240000115
vcrris vcrPeak voltage during discharge. When M is6Peak magnetizing current pair M on secondary side when turned off5And M6Drain-source capacitance CdsCharging and discharging are performed. The first half cycle consisting of modes 1-3 is to charge the capacitor CvBy M2And M3The charging period of the channel, the next half-cycle consisting of modes 4-6 is the charging capacitor CvAnd M1、M3The charging cycle of (a). Except for the rectifying operation part, the two half cycle operations of the modes 1 to 3 and the modes 4 to 6 are basically the same.

Claims (3)

1. A control method of a bidirectional isolation type resonance power converter based on a virtual synchronous motor is characterized in that a power grid alternating current bus passes through line impedance ZacFilter resistor RacAnd the LC filter is connected to the AC side of the AC interface converter; the DC side of the AC interface converter passes through a DC capacitor CdcConnecting a DC/DC converter; the DC/DC converter passes through a voltage stabilizing capacitor CfAnd a filter inductor LfFinally, connected to a power battery; the control method is characterized in that the AC interface converter is virtualized to be a synchronous motor according to the structural similarity of a three-phase synchronous motor model and the AC interface converter, the control method comprises three parts of active power control, virtual excitation control and voltage and current double closed-loop control, and the control method of each part is as follows:
(1) active power control: setting the pole pair number of the virtual synchronous motor to be 1, the torque equation can be expressed as:
Figure FDA0002500814230000011
wherein J represents the moment of inertia of the synchronous machine in kg.m2，ωNExpressing the AC rated angular speed of the power grid in unit rad/s; peAnd PmElectromagnetic and mechanical power of the synchronous motor respectively; is the power angle of the generator, unit rad; omega is the virtual rotor angular frequency of the synchronous machine,unit rad/s; k is a radical ofωIs an AC primary frequency modulation droop coefficient; the active power control part is mainly used for realizing active power closed loop and generating mechanical torque; the active power is calculated from the ac side voltage and current and is expressed as:
P＝uaia+ubib+ucic
in the formula ua、ub、ucIs the terminal voltage of the synchronous machine ia、ib、icIs the terminal current of the synchronous motor;
(2) virtual excitation control: in the virtual excitation control part, the excitation control of the generator is simulated, the alternating voltage and the reactive power are controlled and adjusted, and the virtual potential effective value E of the virtual synchronous motor model is adjusted to generate reactive power; the virtual potential effective value E of the virtual synchronous motor is composed of 3 parts in total:
one, part Δ E of reactive power regulationQThe second is the part of the voltage regulation at the end of the reactor, Delta EUThe automatic excitation regulator can be equivalent to a synchronous motor, and the third is the effective value E of the no-load potential of the synchronous motor0(ii) a The virtual potential effective value of the motor is as follows:
E＝E0+ΔEQ+ΔEU
the vector value of the motor virtual potential is expressed as:
Figure FDA0002500814230000021
(3) voltage current double closed loop control: based on KVL's law, the electromagnetic equation for a synchronous machine can be expressed as:
Figure FDA0002500814230000022
wherein
Figure FDA0002500814230000027
Is an alternating side voltage; l and R are stator inductance of synchronous motor respectivelyAnd resistors, the values of which are respectively the filter inductances L of the LC filters of the AC interfaceacAnd a filter resistor RacThe value of (a) is,
Figure FDA0002500814230000026
is the current of the alternating-current interface,
Figure FDA0002500814230000028
is AC bus side current, and C is C in filter capacitor of LC filteracA value of (d); and obtaining a signal e through voltage-current double closed-loop control, inputting the signal e as a modulation wave into SPWM for modulation, generating a control signal of the AC interface converter, and controlling the on and off of each IGBT tube of the AC interface converter.
2. The virtual synchronous machine-based bidirectional isolated resonant power converter control method of claim 1, wherein a primary side and a secondary side of a DC/DC converter direct current side are connected by a CLC resonant module, the CLC resonant module including a capacitor C for voltage multiplication operation on the primary sidevAnd a resonant capacitor C on the secondary side for resonant PWM operationrAnd a resonant inductor LrA resonant structure of composition; the DC/DC converter has 8 switches for charge or discharge operation, including M on the primary side1、M2、M3、M4Four IGBT tubes and M on secondary side5、M6、M7、M8And four IGBT tubes.
3. The virtual synchronous machine-based bidirectional isolated resonant power converter control method of claim 2, wherein (one) in the charging operation, power flow is from M1～M4Control, M5～M8The diode of (2) is used for full-bridge rectification; suppose a resonant capacitor voltage vcrNot exceeding the battery voltage Vbatt，M3Initially in a conducting state; a complete charging process is divided into 6 modes according to the time sequence:
mode 1 (t)0≤t<t1): when M is1At t0Is on and the primary current ipFlows through M1，M3And a transformer primary side; secondary side current isIncreases from zero and flows through Cr，Lr，M5、M7And a converter secondary side; v can be derived from the equivalent circuit of mode 1crAnd is:
Figure FDA0002500814230000023
Figure FDA0002500814230000024
Wherein:
Figure FDA0002500814230000025
Vcrfis vcrPeak voltage, V, during charging operationdcIs a DC capacitor CdcVoltage at two ends; primary current ipIs a primary side current isAnd a magnetizing current imSumming; n is the turn ratio of a secondary side of the DC/DC converter; when M1 is turned off, the primary current is used to control M1And M2Drain-source capacitance CdsCarrying out charge and discharge; if the drain-source capacitance C of M2dsComplete discharge before the end of mode 1, M can be achieved2The zero voltage switch of (2);
mode 2 (t)1≤t<t2)：M2On, mode 2 starts; primary current ipFlows through M3、M2And a transformer primary side; secondary current isMaintain the same current path as mode 1 until isReducing to zero; from the equivalent circuit of mode 2, vcrAnd isExpressed as:
vcr(t)＝-Vbatt-(Vcr(t1)+Vbatt)cosωr(t-t1)+ZriLr(t1)sinωr(t-t1)
Figure FDA0002500814230000031
from the above formula, i can be obtainedLr(t1) And vcr(t1) The following are:
Figure FDA0002500814230000032
Figure FDA0002500814230000033
in the formula DfWhen it is charging M1(or M)4) Duty ratio, TsFor driving the signal switching period, the duration T of mode 2 can be derived from the above equation2Mf：
Figure FDA0002500814230000034
Mode 3 (t)2≤t<t3): after mode 2, only the magnetizing current imBy M3And M2Circulating; in mode 3, the resonant capacitor voltage vcrIs maintained as vcr(t2) Is a number VcrfA peak value of the defined resonance capacitance; can obtain Vcrf：
Figure FDA0002500814230000035
VcrfShould not exceed VbattTo prevent M5And M7The body diode is abnormally conductive, so that normal work is ensured; only M3And M4C of (A)dsPeak magnetizing current is adopted for charging and discharging; if M is4Drain-source capacitance CdsComplete discharge before the end of mode 3, M can be achieved4The zero voltage switch of (2); next by mode 4 ~ 6 compositionThe operation of the half period is the same as that of the previous half period;
power flow from M in discharge process5～M8Controlling; in order to increase the voltage gain, a diode M is used1And M2As a voltage doubler rectifier, to make M3In a conducting state; capacitor CvAs a power supply capacitor for supplying power; a complete discharge process is divided into 6 modes in time sequence:
mode 1 (t)0≤t<t1): when M8 is at t0When the switch is on, the secondary current passes through Lr、Cr、M8、M6And a DC/DC converter secondary side; primary current ipStarting from zero and increasing by M2、M3、CvAnd a DC/DC converter primary side; v. ofcrPrimary current ipAnd a primary current ni of the secondary sidepCan be approximately derived as:
Figure FDA0002500814230000041
Figure FDA0002500814230000042
vcrris vcrPeak voltage during discharge; secondary current equal to nipAnd a magnetizing current nimSumming; when M is8When closed, isFor making M7Reaching a zero voltage switch state;
mode 2 (t)1≤t<t2): when M is7On, mode 2 starts, ipBeginning to descend; v is obtained from the equivalent circuit of mode 2crAnd nipComprises the following steps:
Figure FDA0002500814230000043
Figure FDA0002500814230000044
using a similar assumption here as for charging mode of operation 2, ni can be derivedp(t1) And vcr(t1):
Figure FDA0002500814230000045
Figure FDA0002500814230000046
In the formula DrFor M in discharging operation5Or M8Until i, mode 2 continuespReduced to zero, duration T of mode 2 in discharge mode2MrIs composed of
Figure FDA0002500814230000047
Mode 3 (t)2≤t<t3): after mode 2, only the secondary side magnetizing current nimBy M7And M6At the same time, v can be deducedcr(t2) The following were used:
Figure FDA0002500814230000051
vcrris vcrPeak voltage during discharge; when M is6Peak magnetizing current pair M on secondary side when turned off5And M6C of (A)dsCharging and discharging; the first half cycle consisting of modes 1-3 is to charge the capacitor CvBy M2And M3The charging period of the channel, the next half-cycle consisting of modes 4-6 is the charging capacitor CvAnd M1、M3A charging cycle of (a); except for the rectifying operation part, the two half cycle operations of the discharge process modes 1 to 3 and 4 to 6 are the same.
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