CN111262486B - Permanent magnet synchronous motor control method - Google Patents

Permanent magnet synchronous motor control method Download PDF

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CN111262486B
CN111262486B CN201811453300.XA CN201811453300A CN111262486B CN 111262486 B CN111262486 B CN 111262486B CN 201811453300 A CN201811453300 A CN 201811453300A CN 111262486 B CN111262486 B CN 111262486B
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reference value
value
current
axis
voltage
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CN111262486A (en
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石高峰
杨洪波
彭再武
凌岳伦
陈慧民
蔡磊
姚超
夏一帆
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CRRC Electric Vehicle Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]

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  • Control Of Ac Motors In General (AREA)

Abstract

A permanent magnet synchronous motor control method, comprising: acquiring motor voltage of a permanent magnet synchronous motor; and judging whether the motor voltage reaches a preset inverter limit voltage, wherein if the motor voltage reaches the preset inverter limit voltage, a preset weak magnetic control model is adopted to control the permanent magnet synchronous motor so that the voltage applied to the permanent magnet synchronous motor is kept at the preset inverter limit voltage. After the back electromotive force of the permanent magnet synchronous motor reaches the limit voltage of the inverter, the method is changed to adopt a preset weak magnetic control model to control the permanent magnet synchronous motor. When the permanent magnet synchronous motor is controlled by adopting the preset weak magnetic control model, the method does not use a PI regulator to carry out closed-loop control on the current, but switches to regulate a load angle to control the T-axis current and simultaneously leads the M-axis current to operate in an open loop mode, thereby ensuring the advantages of the high-speed operation stability and rapidity of the permanent magnet synchronous motor and the like.

Description

Permanent magnet synchronous motor control method
Technical Field
The invention relates to the technical field of motor control, in particular to a permanent magnet synchronous motor control method.
Background
Permanent magnet synchronous machines are widely used in various electric drive systems due to their advantages of high power density and high efficiency. The control technology of the high-performance permanent magnet synchronous motor mainly comprises three types of rotor magnetic field directional vector control, stator magnetic field directional vector control and direct torque control. In practical engineering application, a rotor magnetic field orientation vector control technology based on a dq synchronous coordinate system is generally adopted.
Although the control of the rotor field orientation vector is simple and easy to implement, one obvious disadvantage of the control of the rotor field orientation vector is that the control is sensitive to the variation of the motor parameters. In addition, flux linkage and torque are not completely decoupled based on the rotor field orientation vector control, and flux linkage and back electromotive force are cross-coupled, namely d-axis flux linkage generates back electromotive force on a q-axis, and q-axis flux linkage generates back electromotive force on a d-axis. This coupling increases the difficulty of vector control of the motor. Because the control of the directional vector based on the magnetic field of the rotor usually needs to calculate the voltage feedforward values for compensating the back electromotive forces of the d-axis and the q-axis, and under the influence of the parameter change of the motor and the dynamic process, the feedforward voltages of the two axes are difficult to accurately given.
Under the working condition of high-speed flux weakening, if the current trajectory is unreasonable, the actual current can not track the given current easily, so that the current regulator is quickly saturated, and the current is out of control. Once the current is out of control, the motor and the controller thereof have the possibility of faults such as overspeed, overcurrent, direct current bus voltage rise and the like, which can not only damage equipment, but also endanger the personal safety of field personnel.
The stator magnetic field orientation vector control technology based on the M-T synchronous coordinate system has relatively good high-speed weak magnetic control capability because the excitation current and the torque current are completely decoupled and the stator flux linkage is closed-loop controlled.
However, if the stator field orientation is inaccurate, the voltage required by the PI regulator increases with the weakening depth. In the high-speed weak magnetic region, the output voltage of the inverter reaches the maximum modulatable voltage, so the actual voltage for PI regulation is almost zero, and at this time, the current is controlled by continuously adopting a PI voltage regulation mode no longer properly, so that the actual current cannot track the given current easily, the current regulator is saturated quickly, and the current is out of control.
The direct torque control technology based on space voltage vector modulation solves the problem of torque pulsation generated based on Bang-Bang control on one hand, and omits a current closed loop to directly control a stator flux linkage on the other hand, thereby retaining the advantage of high dynamic response of direct torque control. The core of direct torque control is based on accurate stator flux linkage and torque observations, and in practice, effective stator flux linkage observation is generally a complex calculation process.
Disclosure of Invention
In order to solve the above problems, the present invention provides a method for controlling a permanent magnet synchronous motor, the method comprising:
step one, obtaining motor voltage of a permanent magnet synchronous motor;
and step two, judging whether the motor voltage reaches a preset inverter limit voltage, wherein if the motor voltage reaches the preset inverter limit voltage, a preset weak magnetic control model is adopted to control the permanent magnet synchronous motor so that the voltage applied to the permanent magnet synchronous motor is kept at the preset inverter limit voltage.
According to an embodiment of the invention, when the permanent magnet synchronous motor is controlled by adopting a preset weak magnetic control model:
acquiring a stator flux linkage reference value and a torque reference value of a current control period;
determining a T-axis stator current reference value according to the stator flux linkage reference value and the torque reference value of the current control period
Figure BDA0001887128650000021
Alpha axis voltage feedforward reference value
Figure BDA0001887128650000022
And beta axis voltage feedforward reference value
Figure BDA0001887128650000023
According to the alpha axis voltage feedforward reference value
Figure BDA0001887128650000024
And beta axis voltage feedforward reference value
Figure BDA0001887128650000025
A corresponding inverter control signal is generated.
According to one embodiment of the invention, the alpha axis voltage feedforward reference value is generated according to the following expression
Figure BDA0001887128650000026
And beta axis voltage feedforward reference value
Figure BDA0001887128650000027
Figure BDA0001887128650000028
Figure BDA0001887128650000029
Wherein R issThe resistance of the stator phase is represented,
Figure BDA0001887128650000031
and
Figure BDA0001887128650000032
representing the components of the stator current reference values on the alpha and beta axes respectively,
Figure BDA0001887128650000033
and
Figure BDA0001887128650000034
respectively representing the amplitude reference value and the phase reference value of the stator flux linkage vector of the previous control period,
Figure BDA0001887128650000035
and
Figure BDA0001887128650000036
respectively representing the amplitude reference value and the phase reference value, T, of the stator flux linkage vector of the current control periodswIndicating a control period.
According to an embodiment of the invention, when the permanent magnet synchronous motor is controlled by adopting a preset weak magnetic control model, a phase reference value of a stator flux linkage vector is determined according to the following expression:
θ*＝δ*+θm+δ′
wherein, theta*Phase reference value, delta, representing stator flux linkage vector*Representing the load angle feed-forward value, thetamIndicating the rotor field position and delta' the first load angle adjustment value.
According to one embodiment of the invention, for a control period, the load angle feedforward value for the control period is determined according to the following steps:
step a, calculating a difference value between the torque reference value and a torque value obtained by the previous iteration to obtain a torque deviation value of the current iteration;
step b, obtaining a second load angle adjustment value of the current iteration according to the torque deviation value of the current iteration, and calculating the sum of the second load angle adjustment value of the current iteration and the load angle reference value obtained by the last iteration to obtain a load angle reference value of the current iteration;
step c, calculating the torque value of the iteration according to the load angle reference value of the iteration;
and d, repeating the steps a to c to obtain a load angle feedforward value of the control period.
According to an embodiment of the present invention, in the step b, a product of the torque deviation value of the current iteration and a preset adjustment coefficient is calculated to obtain a second load angle adjustment value of the current iteration.
According to an embodiment of the present invention, in step c, the torque value of the current iteration is determined according to the following expression:
Figure BDA0001887128650000037
wherein, TeRepresenting the torque value of the current iteration, P representing the number of motor poles,. phisIndicating stator flux linkage, #fDenotes the main pole flux linkage, LdsAnd LqsAnd d-axis inductance and q-axis inductance of the stator are respectively represented, and delta represents the load angle of the iteration.
According to one embodiment of the invention, the load angle adjustment value δ' is determined according to the following steps:
calculating the reference value of the stator current of the T shaft
Figure BDA0001887128650000038
And the actual value I of the stator current of the T shaftTA difference of (d);
and generating the load angle adjusting value delta' according to the difference value by utilizing a preset PID controller.
According to one embodiment of the invention, the reference value of the T-axis stator current is determined according to the following expression
Figure BDA0001887128650000041
Figure BDA0001887128650000042
Wherein, Te *Representing the reference value of torque, P the number of poles of the motor, psis *Representing the stator flux linkage reference value.
According to one embodiment of the present invention, if the motor voltage does not reach the preset inverter limit voltage, the permanent magnet synchronous motor is controlled using a preset vector control model,
when the permanent magnet synchronous motor is controlled by adopting a preset limit voltage control model:
acquiring a stator flux linkage reference value and a torque reference value of a current control period;
generating a M-axis stator current reference value according to the stator flux linkage reference value and the torque reference value of the current control period
Figure BDA0001887128650000043
Reference value of T-axis stator current
Figure BDA0001887128650000044
Alpha axis voltage feedforward reference value
Figure BDA0001887128650000045
And beta axis voltage feedforward reference value
Figure BDA0001887128650000046
According to the M-axis stator current reference value
Figure BDA0001887128650000047
And actual value of M-axis stator current IMGenerating an alpha-axis current control regulation voltage value UαAnd according to the T-axis stator current reference value
Figure BDA0001887128650000048
And actual value of stator current I of T axisTGenerating a beta axis current control regulated voltage value Uβ；
According to the alpha axis voltage feedforward reference value
Figure BDA0001887128650000049
Feed-forward reference value of beta axis voltage
Figure BDA00018871286500000410
Alpha axis current control regulating voltage value UαAnd beta axis current control regulating voltage value UβRespectively generate alpha axesReference value of voltage
Figure BDA00018871286500000411
And beta axis voltage reference
Figure BDA00018871286500000412
According to the alpha axis voltage reference value
Figure BDA00018871286500000413
And beta axis voltage reference
Figure BDA00018871286500000414
A corresponding inverter control signal is generated.
According to one embodiment of the invention, the M-axis stator current reference value is determined according to the following expression
Figure BDA00018871286500000415
Figure BDA00018871286500000416
Wherein the content of the first and second substances,
Figure BDA00018871286500000417
denotes the reference value of the stator current of the T axis, Is *Representing the stator current reference value.
According to an embodiment of the present invention, when the permanent magnet synchronous motor is controlled by using a preset limit voltage control model, a phase reference value of a stator flux linkage vector is determined according to the following expression:
θ*＝δ*+θm
wherein, theta*Phase reference value, delta, representing stator flux linkage vector*Representing the load angle feed-forward value, thetamIndicating the rotor field position.
Aiming at the problems of directional deviation and high-speed weak magnetic area current control, the permanent magnet synchronous motor control method provided by the invention changes to control the permanent magnet synchronous motor by adopting a preset weak magnetic control model after the back electromotive force of the permanent magnet synchronous motor reaches the limit voltage of an inverter. When the permanent magnet synchronous motor is controlled by adopting the preset weak magnetic control model, the method does not use a PI regulator to carry out closed-loop control on the current, but switches to regulate a load angle to control T-axis current, and simultaneously, the M-axis current is operated in an open-loop mode, so that the voltage applied to the motor is fixed to the maximum voltage which can be modulated by the inverter, and the advantages of high-speed operation stability, rapidity and the like of the permanent magnet synchronous motor are also ensured.
Meanwhile, the method provided by the invention adopts a test method to obtain the expected torque meeting the MTPA
Figure BDA0001887128650000051
Reference flux linkage
Figure BDA0001887128650000052
And current distribution in M-T coordinates (including M-axis stator current reference values)
Figure BDA0001887128650000053
And T-axis stator current reference
Figure BDA0001887128650000054
) Therefore, the complex stator flux linkage observation algorithm can be avoided, so that the influence of parameter change of the permanent magnet synchronous motor can be effectively avoided.
In addition, the method provided by the invention calculates the feedforward voltage by calculating the differential of the given stator flux linkage in a switching period, and can effectively avoid the influence of other parameters except the resistance of the permanent magnet synchronous motor, thereby improving the accuracy of the feedforward voltage and obtaining faster dynamic response.
Additional features and advantages of the invention will be set forth in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The objectives and other advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.
Drawings
In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the following briefly introduces the drawings required in the description of the embodiments or the prior art:
FIG. 1 is a schematic view of a d-q coordinate system of a permanent magnet synchronous machine according to one embodiment of the present invention;
FIG. 2 is a phasor diagram of flux linkage, voltage and current vectors in a d-q coordinate system and an M-T coordinate system for an interior permanent magnet synchronous motor under certain operating conditions according to one embodiment of the invention;
fig. 3 is a schematic flow chart of an implementation of a permanent magnet synchronous motor control method according to an embodiment of the invention;
fig. 4 is a schematic flow chart of an implementation of a control of a permanent magnet synchronous motor using a preset vector control model according to an embodiment of the present invention;
FIG. 5 is a control block diagram for controlling a permanent magnet synchronous machine using a preset vector control model, according to one embodiment of the present invention;
FIG. 6 is a graph of determining a load angle feedforward value δ according to one embodiment of the invention*The implementation flow diagram of (1);
FIG. 7 is a graph of determining a load angle feedforward value δ according to one embodiment of the invention*A logic block diagram of (a);
fig. 8 is a schematic flow chart of an implementation of controlling a permanent magnet synchronous motor by using a preset flux weakening control model according to an embodiment of the invention;
fig. 9 is a control logic block diagram for controlling a permanent magnet synchronous motor using a preset field weakening control model according to an embodiment of the present invention.
Detailed Description
The following detailed description of the embodiments of the present invention will be provided with reference to the drawings and examples, so that how to apply the technical means to solve the technical problems and achieve the technical effects can be fully understood and implemented. It should be noted that, as long as there is no conflict, the embodiments and the features of the embodiments of the present invention may be combined with each other, and the technical solutions formed are within the scope of the present invention.
In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of the embodiments of the invention. It will be apparent, however, to one skilled in the art that the present invention may be practiced without some of these specific details or with other methods described herein.
Additionally, the steps illustrated in the flow charts of the figures may be performed in a computer system such as a set of computer-executable instructions and, although a logical order is illustrated in the flow charts, in some cases, the steps illustrated or described may be performed in an order different than here.
Under the synchronous rotating coordinate system, the coupling between phases in the mathematical model of the permanent magnet synchronous motor is eliminated, and the mathematical model is obviously simplified. The d-q coordinate system shown in fig. 1 can be established by taking the axis of the composite magnetic field of the rotor permanent magnets as the d axis (the positive direction of the composite magnetic field is consistent with the direction of the magnetic lines of force), and taking the axis which rotates 90 degrees counterclockwise as the q axis.
Under the d-q synchronous rotation coordinate system of the permanent magnet synchronous motor, the voltage equation of the permanent magnet synchronous motor can be expressed as follows:
Figure BDA0001887128650000061
wherein, VdsAnd VqsRespectively representing stator d-axis voltage and stator q-axis voltage, RsDenotes stator phase resistance, IdsAnd IqsRespectively representing stator d-axis current and stator q-axis current, #dsAnd psiqsRespectively represent stator d-axis flux linkage and stator q-axis flux linkage, omegaeIndicating the synchronous frequency and p the number of motor poles.
The flux linkage equation can be expressed as:
Figure BDA0001887128650000071
wherein psidrAnd psiqrRespectively representing rotor d-axis flux linkage and rotor q-axis flux linkage, LdsAnd LqsRespectively representing stator d-axis inductance and stator q-axis inductance, LdmAnd LqmRepresenting d-axis stator-rotor mutual inductance and q-axis stator-rotor mutual inductance, psifWhich represents the flux linkage (main pole flux linkage) that the permanent magnet can generate at the stator.
The torque equation can be expressed as:
Figure BDA0001887128650000072
wherein, TeRepresenting the motor electromagnetic torque.
The stator magnetic field orientation is to use the stator flux linkage vector phase as the reference zero for analysis and control. The axis of the stator magnetic field is used as an M axis (the positive direction of the stator magnetic field is consistent with the direction of magnetic lines of force), the axis rotating by 90 degrees in the counterclockwise direction is used as a T axis, and an M-T coordinate system is established. When stator magnetic field orientation is adopted, motor stator magnetic linkage psisAll located on the M-axis and the flux linkage component on the T-axis is zero. In the stator field orientation, the permanent magnet synchronous motor equation is simplified, wherein the stator voltage equation is simplified as:
Figure BDA0001887128650000073
the torque equation at the stator field coordinates is simplified to:
Figure BDA0001887128650000074
wherein, VMRepresenting stator M-axis voltage, VTRepresenting stator T-axis voltage, IMRepresenting stator M-axis current, ITRepresenting stator T-axis current.
From the voltage equation, it can be seen that the voltage component on the M-axis at steady state is simply the voltage drop of the M-axis current component on the stator resistance.
When the permanent magnet synchronous motor is in no load, the phases of the stator magnetic field and the main pole magnetic field are the same; when loaded, there is a phase angle difference (also called load angle, indicated by the symbol δ) between the stator magnetic field and the main pole magnetic field. Fig. 2 shows phasor diagrams of flux linkage, voltage and current vectors of the embedded permanent magnet synchronous motor under a d-q coordinate system and an M-T coordinate system under a specific working condition. As can be seen from fig. 2, the phase difference of the two coordinate systems is equal to the load angle δ. In FIG. 2, IsRepresenting stator current vectors having components I in the M, T coordinate axesMAnd IT. M-axis component I of stator current vectorMThe stator flux linkage is in phase with the stator flux linkage and is a current component of reactive power generated by the motor; t-axis component I of stator current vectorTIn phase with the stator back emf, which is the current component of the motor that delivers active power.
In particular, IM、IT、IdsAnd IqsHas the following relationship:
IM＝Iqs sinδ+Idscosδ (6)
IT＝Iqs cosδ-Ids sinδ (7)
permanent magnet synchronous motors for variable speed operation usually have position sensors mounted on the rotor, so that the d-axis phase can be detected directly by the position sensors. The invention determines the phase of the stator magnetic field by adding the load angle delta to the d-axis phase detected by the position sensor, so that the key of orientation is to accurately calculate the value of the load angle delta.
Since the inductance of the stator of the permanent magnet synchronous motor is affected by the position of the rotor, the motor design usually only gives the inductance values on the d axis and the q axis in the d-q coordinate system, and therefore, the calculation in the M-T coordinate system must be performed by means of the conversion of the load angle delta into the calculation in the d-q coordinate system.
The flux linkage equation of the permanent magnet synchronous motor can be obtained as follows:
Figure BDA0001887128650000081
Figure BDA0001887128650000082
by substituting formula (3) with formula (8) and formula (9), it is possible to obtain:
Figure BDA0001887128650000083
expression (10) describes the electromagnetic torque TeMagnetic linkage psi with statorsAnd the relationship between the load angle δ, is the basic formula used in the present invention to calculate the load angle δ.
Fig. 3 shows a schematic implementation flow diagram of the permanent magnet synchronous motor control method provided by this embodiment.
As shown in fig. 3, the method for controlling a permanent magnet synchronous motor according to this embodiment obtains a motor voltage of the permanent magnet synchronous motor in step S301, and determines whether the motor voltage of the permanent magnet synchronous motor reaches a preset inverter limit voltage in step S302.
If the motor voltage of the permanent magnet synchronous motor does not reach the preset inverter limit voltage, the method adopts a preset vector control model to control the permanent magnet synchronous motor in step S303; if the motor voltage of the permanent magnet synchronous motor reaches the preset inverter limit voltage, the method controls the permanent magnet synchronous motor by using a preset weak magnetic control model in step S304, so that the voltage applied to the permanent magnet synchronous motor is kept at the preset inverter limit voltage.
It should be noted that, in different embodiments of the present invention, the specific value of the preset inverter limit voltage may be configured to be different reasonable values according to actual situations, and the present invention does not limit the specific value of the preset inverter limit voltage.
In this embodiment, when the method uses a preset vector control model to control the permanent magnet synchronous motor, the method is preferably configured to implement control of the permanent magnet synchronous motor by using a PI regulator to perform closed-loop control on the M-axis current and the T-axis current.
Specifically, fig. 4 shows a schematic flow chart of implementing the control of the permanent magnet synchronous motor by using the preset vector control model in this embodiment, and fig. 5 shows a control block diagram of controlling the permanent magnet synchronous motor by using the preset vector control model in this embodiment.
As shown in fig. 4 and 5, in this embodiment, the method obtains the stator flux linkage reference value ψ of the current control cycle (i.e., the current control late period) in step S401s *And a torque reference value Te *. The method preferably can acquire the torque reference value T of the permanent magnet synchronous motor through a bench teste *Stator flux linkage reference value psi under condition of meeting MTPAs *. Meanwhile, the method can also acquire the torque reference value T of the permanent magnet synchronous motor in the same waye *Stator reference current I under condition of meeting MTPA conditions *。
Of course, in other embodiments of the present invention, the method may also use other reasonable ways to obtain the stator flux linkage reference value ψ of the current control cycles *And a stator reference current Is *The present invention is not limited thereto.
As shown in fig. 4 and 5, in the present embodiment, the stator flux linkage reference value ψ is obtained in the current round of the control periods *And a torque reference value Te *Thereafter, the method will proceed to step S402 according to the reference value ψ of the stator flux linkage of the present round of control periods *And a torque reference value, generating an M-axis stator current reference value
Figure BDA0001887128650000091
Reference value of T-axis stator current
Figure BDA0001887128650000092
Alpha axis voltage feedforward reference value
Figure BDA0001887128650000093
And betaShaft voltage feedforward reference value
Figure BDA0001887128650000094
Specifically, in the present embodiment, the method preferably determines the current distribution scheme in the M-T coordinate satisfying the MTPA, that is, the M-axis stator current reference value, according to the following expression
Figure BDA0001887128650000095
Reference value of T-axis stator current
Figure BDA0001887128650000096
Figure BDA0001887128650000097
Figure BDA0001887128650000101
Wherein P represents the number of motor poles.
The following relationship exists between the stator flux linkage and the stator voltage of the permanent magnet synchronous motor:
Figure BDA0001887128650000102
wherein the content of the first and second substances,
Figure BDA0001887128650000103
and
Figure BDA0001887128650000104
respectively representing the stator flux and stator voltage, R, of a PMSMsThe resistance of the stator phase is represented,
Figure BDA0001887128650000105
representing the stator current.
As can be seen from expression (13), the motor stator flux linkage is equal to the integral of the stator voltage (excluding the voltage drop across the stator resistor) over time, and as long as the actual voltage applied to the permanent magnet synchronous motor is equal to the given voltage, the motor can establish a corresponding flux linkage.
By compensating factors such as dead zone of the inverter, tube voltage drop, line voltage drop, digital control delay and the like, the amplitude and the phase of the output voltage of the controller are accurate as much as possible, so that the accuracy of the amplitude and the phase of the stator flux linkage is indirectly ensured. Under the premise of accurate flux linkage, the calculated active current given value
Figure BDA0001887128650000106
Is also accurate.
To be provided with
Figure BDA0001887128650000107
For reference, the load angle feed-forward error is compensated by the regulator, and the magnitude of the load angle is completely determined by the active current to be generated. In summary, the controller is enabled to accurately output a given voltage and adjust the load angle with the active current component as a reference, and the two points together ensure the accuracy of orientation and the accuracy of torque control.
In actual operation, in order to increase the control response speed, the control amount needs to be given by feedforward. The feed forward voltage is used to compensate for stator resistive voltage drops and back emf. The amplitude and the phase of the stator flux linkage can be changed in the running process of the motor, and flux linkage vector expected values at the initial time and the ending time of a switching period are decomposed to an alpha-beta two-phase static coordinate system to obtain flux linkage variable quantity on each coordinate axis. This amount of change reflects both the change in flux linkage amplitude and the advance in flux linkage phase. The instantaneous value of back emf is equal to the differential of the flux linkage with respect to time, and for discrete control, the average value of back emf in a switching period is approximately equal to the amount of change in flux linkage divided by the switching period.
Therefore, based on the above principle, the present embodiment preferably generates the α -axis voltage feedforward reference value using the following expression
Figure BDA0001887128650000108
And beta axis voltage feed forwardReference value
Figure BDA0001887128650000109
Figure BDA00018871286500001010
Figure BDA00018871286500001011
Wherein R issThe resistance of the stator phase is represented,
Figure BDA0001887128650000111
and
Figure BDA0001887128650000112
representing the components of the stator current reference values on the alpha and beta axes respectively,
Figure BDA0001887128650000113
and
Figure BDA0001887128650000114
respectively representing the amplitude reference value and the phase reference value of the stator flux linkage vector of the previous control period,
Figure BDA0001887128650000115
and
Figure BDA0001887128650000116
respectively representing the amplitude reference value and the phase reference value, T, of the stator flux linkage vector of the current control periodswIndicating a control period.
It should be noted that in other embodiments of the present invention, the method may also use other reasonable ways to generate the M-axis stator current reference value according to practical situations
Figure BDA0001887128650000117
Reference value of T-axis stator current
Figure BDA0001887128650000118
Alpha axis voltage feedforward reference value
Figure BDA0001887128650000119
And beta axis voltage feedforward reference value
Figure BDA00018871286500001110
The present aspect is not limited thereto.
As shown in fig. 4 and 5, in the present embodiment, the M-axis stator current reference value is obtained
Figure BDA00018871286500001111
Reference value of T-axis stator current
Figure BDA00018871286500001112
Alpha axis voltage feedforward reference value
Figure BDA00018871286500001113
And beta axis voltage feedforward reference value
Figure BDA00018871286500001114
Then, the method will follow the M-axis stator current reference value in step S403
Figure BDA00018871286500001115
And actual value of M-axis stator current IMGenerating an alpha-axis current control regulation voltage value UαAlso, in step S404, the reference value is determined according to the T-axis stator current
Figure BDA00018871286500001116
And actual value of stator current I of T axisTGenerating a beta axis current control regulated voltage value Uβ。
Specifically, in the present embodiment, the method preferably calculates the M-axis stator current reference value in step S403
Figure BDA00018871286500001117
With the actual value of M-axis stator current IMAnd obtaining the M-axis current control regulation voltage value U by utilizing a PID regulator according to the difference valueM. Similarly, the method calculates a T-axis stator current reference value in step S404
Figure BDA00018871286500001118
And the actual value I of the stator current of the T shaftTAnd obtaining a T-axis current control regulation voltage value U by utilizing a PID regulator according to the difference valueT. Regulating the voltage value U by controlling the M-axis currentMAnd T-axis current control regulating voltage value UTCoordinate transformation is carried out, and then the alpha-axis current control regulation voltage value U can be obtainedαAnd beta axis current control regulating voltage value Uβ。
As shown in fig. 4 and 5, the α -axis current control adjustment voltage value U is obtainedαAnd beta axis current control regulating voltage value UβThen, the method will feed forward the reference value according to the α -axis voltage in step S405
Figure BDA00018871286500001119
Feed-forward reference value of beta axis voltage
Figure BDA00018871286500001120
Alpha axis current control regulating voltage value UαAnd beta axis current control regulating voltage value UβRespectively generating alpha axis voltage reference values
Figure BDA00018871286500001121
And beta axis voltage reference
Figure BDA00018871286500001122
Namely, the existence of:
Figure BDA00018871286500001123
Figure BDA00018871286500001124
subsequently, the method may also include step S406 of referring to the α -axis voltage reference value
Figure BDA00018871286500001125
And beta axis voltage reference
Figure BDA00018871286500001126
And generating corresponding inverter control signals so as to control the operation state of the inverter.
Of course, in other embodiments of the present invention, according to practical situations, the method may also use other reasonable manners to control the permanent magnet synchronous motor by using the preset vector control model, and the present invention is not limited thereto.
In the embodiment, the feedforward reference value of the alpha-axis voltage is determined
Figure BDA0001887128650000121
And beta axis voltage feedforward reference value
Figure BDA0001887128650000122
And controlling and regulating voltage value U for M-axis currentMAnd T-axis current control regulating voltage value UTCoordinate transformation is carried out to obtain an alpha-axis current control regulating voltage value UαAnd beta axis current control regulating voltage value UβPhase reference theta of stator flux linkage vector used in the process*Preferably determined according to the expression:
θ*＝δ*+θm (18)
wherein, delta*Representing the load angle feed-forward value, thetamIndicating the rotor magnetic field position (which may preferably be acquired by a position sensor).
The expression (10) is a unitary equation about the load angle, but the equation simultaneously contains sine and cosine of the load angle and a product of the sine and cosine, and analysis shows that the analytical solution of the load angle is difficult to directly obtain according to the expression (10). To solve this problem, in this embodiment, the method preferably obtains the load angle by using a numerical calculation methodSufficiently accurate numerical solution to obtain the feed-forward value delta of the load angle*。
FIG. 6 shows the determination of the load angle feedforward value δ in this embodiment*Fig. 7 shows the determination of the load angle feedforward value δ in this embodiment*A logic block diagram of (a).
As shown in fig. 6 and 7, in the present embodiment, the method first calculates a difference between the torque reference value and the torque value obtained in the previous iteration in step S601, so as to obtain the torque deviation value in the current iteration. Namely, the existence of:
Figure BDA0001887128650000123
wherein, Delta TeIndicating the torque deviation value, TepRepresenting the torque values resulting from the iteration.
Then, the method will proceed to step S602 according to the torque deviation Δ T of the current iterationeTo obtain a second load angle adjustment value delta for the current iteration. Specifically, in the present embodiment, the method preferably calculates the torque deviation value Δ T of the current iterationeAnd a preset adjustment coefficient KeTo obtain the second load angle adjustment value delta of the current iteration.
Namely, there are:
Δδ＝Ke·ΔTe (20)
after obtaining the second load angle adjustment value Δ δ of the current iteration, the method calculates the second load angle adjustment value Δ δ of the current iteration and the load angle reference value T obtained from the previous iteration in step S603epAnd summing to obtain the load angle reference value of the iteration.
Of course, in various embodiments of the present invention, the adjustment coefficient K is preseteThe specific value can be configured into different reasonable values according to the actual situation, and the preset adjustment coefficient K is not adjusted in the inventioneThe specific value of (a) is defined.
Subsequently, the method calculates a torque value of the current iteration according to the load angle reference value of the current iteration in step S604. Namely, the load angle reference value of the current iteration is substituted into the expression (10) to obtain the torque value of the current iteration.
Thus, an iterative process is completed, and the load angle feedforward value δ can be obtained by repeating the steps S601 to S604 in a control cycle*To the exact numerical solution of.
It should be noted that in other embodiments of the present invention, the method may also use other reasonable ways to determine the load angle feedforward value δ according to actual needs*The present invention is not limited thereto. Meanwhile, it should be noted that, in different embodiments of the present invention, the iteration number performed in one control period may be configured to different reasonable values according to actual needs, and the value of the iteration number is not limited in the present invention.
The PID closed-loop control shown in the figure 4 and the figure 5 can ensure that the motor can stably output the torque in the locked-rotor and starting processes. However, due to parameter errors and inaccurate orientation, the voltage required by the PID regulator will increase with increasing motor speed. However, in the high speed region, in order to make full use of the dc voltage, the voltage reserved for the regulator is usually very small or even zero, and it is no longer suitable to continue to control the current by means of PI regulation of the voltage.
In view of this situation, as shown in fig. 3 again, in this embodiment, if the motor voltage of the permanent magnet synchronous motor reaches the preset inverter limit voltage, the method uses a preset weak magnetic control model to control the permanent magnet synchronous motor in step S304.
Fig. 8 is a schematic diagram illustrating an implementation flow of controlling a permanent magnet synchronous motor by using a preset weak magnetic control model in this embodiment, and fig. 9 is a control logic block diagram of controlling a permanent magnet synchronous motor by using a preset weak magnetic control model in this embodiment.
As shown in fig. 8 and 9, in this embodiment, the method first obtains the stator flux linkage reference value ψ of the current control cycle (i.e., the current control late stage) in step S801s *And a torque reference value Te *. Then againThe stator flux linkage reference value ψ according to the present round of the control period in step S802s *And a torque reference value Te *Determining a reference value of the stator current of the T-axis
Figure BDA0001887128650000131
Alpha axis voltage feedforward reference value
Figure BDA0001887128650000132
And beta axis voltage feedforward reference value
Figure BDA0001887128650000133
In the embodiment, the method determines the alpha-axis voltage feedforward reference value
Figure BDA0001887128650000134
And beta axis voltage feedforward reference value
Figure BDA0001887128650000135
Is preferably the same as the determination of the α -axis voltage feedforward reference value in the above-described step S402
Figure BDA0001887128650000136
And beta axis voltage feedforward reference value
Figure BDA0001887128650000137
The principle and the process are the same, and the details of the part are not described herein again.
In the embodiment, the method determines the feedforward reference value of the alpha-axis voltage
Figure BDA0001887128650000141
And beta axis voltage feedforward reference value
Figure BDA0001887128650000142
Preferably, the PI regulator is used for performing closed-loop control on the T-axis stator current, while the PI regulator is not used for performing closed-loop control on the M-axis stator current, but the M-axis stator current is used in an open-loop modeAnd (6) rows.
Specifically, in the present embodiment, the method preferably determines the phase reference value θ of the stator flux linkage vector according to the following expression*：
θ*＝δ*+θm+δ′ (21)
Wherein, delta*Representing the load angle feed-forward value, thetamIndicating the rotor field position and delta' the first load angle adjustment value.
In the present embodiment, as shown in fig. 9, the method preferably first calculates the T-axis stator current reference value when determining the load angle adjustment value δ
Figure BDA0001887128650000143
And the actual value I of the stator current of the T shaftTThen a preset PID controller is used to generate the load angle adjustment value δ' according to the difference.
Of course, in other embodiments of the present invention, the method may also determine the load angle adjustment value δ' in other reasonable manners according to actual situations, and the present invention is not limited thereto.
As shown in fig. 8, in the present embodiment, in step S803, the method may also feed forward the reference value according to the α -axis voltage
Figure BDA0001887128650000144
And beta axis voltage feedforward reference value
Figure BDA0001887128650000145
And generating corresponding inverter control signals so as to control the operation state of the inverter.
As can be seen from the above description, the control method of the permanent magnet synchronous motor provided by the present invention is directed to the problems of directional deviation and current control in the high-speed weak magnetic region, and after the back electromotive force of the permanent magnet synchronous motor reaches the limit voltage of the inverter, the control method is switched to control the permanent magnet synchronous motor by using the preset weak magnetic control model. When the permanent magnet synchronous motor is controlled by adopting the preset weak magnetic control model, the method does not use a PI regulator to carry out closed-loop control on the current, but switches to regulate a load angle to control T-axis current, and simultaneously, the M-axis current is operated in an open-loop mode, so that the voltage applied to the motor is fixed to the maximum voltage which can be modulated by the inverter, and the advantages of high-speed operation stability, rapidity and the like of the permanent magnet synchronous motor are also ensured.
Meanwhile, the method provided by the invention adopts a test method to obtain the expected torque meeting the MTPA
Figure BDA0001887128650000146
Reference flux linkage
Figure BDA0001887128650000147
And current distribution in M-T coordinates (including M-axis stator current reference values)
Figure BDA0001887128650000148
And T-axis stator current reference
Figure BDA0001887128650000149
) Therefore, the complex stator flux linkage observation algorithm can be avoided, so that the influence of parameter change of the permanent magnet synchronous motor can be effectively avoided.
In addition, the method provided by the invention calculates the feedforward voltage by calculating the differential of the given stator flux linkage in a switching period, and can effectively avoid the influence of other parameters except the resistance of the permanent magnet synchronous motor, thereby improving the accuracy of the feedforward voltage and obtaining faster dynamic response.
It is to be understood that the disclosed embodiments of the invention are not limited to the particular structures or process steps disclosed herein, but extend to equivalents thereof as would be understood by those skilled in the relevant art. It is also to be understood that the terminology used herein is for the purpose of describing particular embodiments only, and is not intended to be limiting.
Reference in the specification to "one embodiment" or "an embodiment" means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment of the invention. Thus, the appearances of the phrase "one embodiment" or "an embodiment" in various places throughout this specification are not necessarily all referring to the same embodiment.
While the above examples are illustrative of the principles of the present invention in one or more applications, it will be apparent to those of ordinary skill in the art that various changes in form, usage and details of implementation can be made without departing from the principles and concepts of the invention. Accordingly, the invention is defined by the appended claims.

Claims (9)

1. A method of controlling a permanent magnet synchronous motor, the method comprising:
step one, obtaining motor voltage of a permanent magnet synchronous motor;
step two, judging whether the motor voltage reaches a preset inverter limit voltage, wherein if the motor voltage reaches the preset inverter limit voltage, a preset weak magnetic control model is adopted to control the permanent magnet synchronous motor so that the voltage applied to the permanent magnet synchronous motor is kept at the preset inverter limit voltage;
when the permanent magnet synchronous motor is controlled by adopting a preset weak magnetic control model:
acquiring a stator flux linkage reference value and a torque reference value of a current control period;
determining a T-axis stator current reference value according to the stator flux linkage reference value and the torque reference value of the current control period
Figure FDA0003197646840000011
Alpha axis voltage feedforward reference value
Figure FDA0003197646840000012
And beta axis voltage feedforward reference value
Figure FDA0003197646840000013
According to the alpha axis voltage feedforward reference value
Figure FDA0003197646840000014
And beta axis voltage feedforward reference value
Figure FDA0003197646840000015
Generating a corresponding inverter control signal;
determining a T-axis stator current reference value according to the following expression
Figure FDA0003197646840000016
Figure FDA0003197646840000017
Wherein, Te *Representing the reference value of torque, P the number of poles of the motor, psis *Representing a stator flux linkage reference value;
generating the alpha axis voltage feedforward reference value according to the following expression
Figure FDA0003197646840000018
And beta axis voltage feedforward reference value
Figure FDA0003197646840000019
Figure FDA00031976468400000110
Figure FDA00031976468400000111
Wherein R issThe resistance of the stator phase is represented,
Figure FDA00031976468400000112
and
Figure FDA00031976468400000113
representing the components of the stator current reference values on the alpha and beta axes respectively,
Figure FDA00031976468400000114
and
Figure FDA00031976468400000115
respectively representing the amplitude reference value and the phase reference value of the stator flux linkage vector of the previous control period,
Figure FDA00031976468400000116
and
Figure FDA00031976468400000117
amplitude reference value and phase reference value, T, of stator flux linkage vector representing the control period of the current round respectivelyswIndicating a control period.
2. The method of claim 1, wherein when the permanent magnet synchronous motor is controlled by using a preset weak magnetic control model, the phase reference value of the stator flux linkage vector is determined according to the following expression:
θ*＝δ*+θm+δ′
wherein, theta*Phase reference value, delta, representing stator flux linkage vector*Representing the load angle feed-forward value, thetamIndicating the rotor field position and delta' the first load angle adjustment value.
3. A method as claimed in claim 2, wherein for a control period, the load angle feed forward value for the control period is determined in accordance with the steps of:
step a, calculating a difference value between the torque reference value and a torque value obtained by the previous iteration to obtain a torque deviation value of the current iteration;
step b, obtaining a second load angle adjustment value of the current iteration according to the torque deviation value of the current iteration, and calculating the sum of the second load angle adjustment value of the current iteration and the load angle reference value obtained by the last iteration to obtain a load angle reference value of the current iteration;
step c, calculating the torque value of the iteration according to the load angle reference value of the iteration;
and d, repeating the steps a to c to obtain a load angle feedforward value of the control period.
4. The method of claim 3, wherein in the step b, the product of the torque deviation value of the current iteration and a preset adjustment coefficient is calculated to obtain a second load angle adjustment value of the current iteration.
5. A method according to claim 3, wherein in step c, the torque value for the current iteration is determined according to the expression:
Figure FDA0003197646840000021
wherein, TeRepresenting the torque value of the current iteration, P representing the number of motor poles,. phisIndicating stator flux linkage, #fDenotes the main pole flux linkage, LdsAnd LqsAnd d-axis inductance and q-axis inductance of the stator are respectively represented, and delta represents the load angle of the iteration.
6. The method according to claim 2, characterized in that the load angle adjustment value δ' is determined according to the following steps:
calculating the reference value of the stator current of the T shaft
Figure FDA0003197646840000022
And the actual value I of the stator current of the T shaftTA difference of (d);
and generating the load angle adjusting value delta' according to the difference value by utilizing a preset PID controller.
7. A method according to any of claims 1-6, characterized in that if the motor voltage does not reach the preset inverter limit voltage, the permanent magnet synchronous motor is controlled using a preset vector control model,
when the permanent magnet synchronous motor is controlled by adopting a preset limit voltage control model:
acquiring a stator flux linkage reference value and a torque reference value of a current control period;
generating a M-axis stator current reference value according to the stator flux linkage reference value and the torque reference value of the current control period
Figure FDA0003197646840000031
Reference value of T-axis stator current
Figure FDA0003197646840000032
Alpha axis voltage feedforward reference value
Figure FDA0003197646840000033
And beta axis voltage feedforward reference value
Figure FDA0003197646840000034
According to the M-axis stator current reference value
Figure FDA0003197646840000035
And actual value of M-axis stator current IMGenerating an alpha-axis current control regulation voltage value UαAnd according to the T-axis stator current reference value
Figure FDA0003197646840000036
And actual value of stator current I of T axisTGenerating a beta axis current control regulated voltage value Uβ；
According to the alpha axis voltage feedforward reference value
Figure FDA0003197646840000037
Feed-forward reference value of beta axis voltage
Figure FDA0003197646840000038
Alpha axis current control regulating voltage value UαAnd beta axis current control regulating voltage value UβRespectively generating alpha axis voltage reference values
Figure FDA0003197646840000039
And beta axis voltage reference
Figure FDA00031976468400000310
According to the alpha axis voltage reference value
Figure FDA00031976468400000311
And beta axis voltage reference
Figure FDA00031976468400000312
A corresponding inverter control signal is generated.
8. The method of claim 7, wherein the M-axis stator current reference value is determined according to the following expression
Figure FDA00031976468400000313
Figure FDA00031976468400000314
Wherein the content of the first and second substances,
Figure FDA00031976468400000315
denotes the reference value of the stator current of the T axis, Is *Representing the stator current reference value.
9. The method of claim 7, wherein the phase reference value of the stator flux linkage vector is determined according to the following expression when the permanent magnet synchronous motor is controlled using a preset limit voltage control model:
θ*＝δ*+θm
wherein, theta*Phase reference value, delta, representing stator flux linkage vector*Representing the load angle feed-forward value, thetamIndicating the rotor field position.
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