CN110362890B - Method for calculating iron loss resistance of variable frequency motor under PWM harmonic condition - Google Patents

Method for calculating iron loss resistance of variable frequency motor under PWM harmonic condition Download PDF

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CN110362890B
CN110362890B CN201910572359.9A CN201910572359A CN110362890B CN 110362890 B CN110362890 B CN 110362890B CN 201910572359 A CN201910572359 A CN 201910572359A CN 110362890 B CN110362890 B CN 110362890B
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张冬冬
武新章
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Xian Jiaotong University
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Abstract

The invention discloses a method for calculating iron loss resistance of a variable frequency motor under a PWM harmonic condition, which belongs to the field of alternating current motor loss analysis and calculation. The method accounts for the additional iron losses generated by the PWM converter harmonics and the surface and pulse losses generated by the motor space harmonics. By utilizing the method, the accurate variable frequency motor equivalent circuit under the condition of iron loss can be obtained. Finally, taking a 5.5kW and a 30kW variable frequency induction motor as an example, the iron loss resistance change rules are obtained. And the effectiveness of the invention is verified by using the two induction motors.

Description

Method for calculating iron loss resistance of variable frequency motor under PWM harmonic condition
Technical Field
The invention belongs to the field of alternating current motor loss analysis and calculation, and particularly relates to a method for calculating iron loss resistance of a variable frequency motor under a PWM harmonic condition.
Background
At present, the PWM variable frequency driving system is highly integrated with the induction motor. PWM variable frequency drives contain a large number of harmonic components that have a significant effect on induction motor losses. In order to reduce the iron loss of the motor, the iron loss of the PWM variable-frequency power supply induction motor can be quickly and accurately calculated. The iron loss of the induction motor can be generally calculated by a road calculation (an analytical calculation method) or a field calculation (a finite element method). The iron loss calculated based on the finite element method is more accurate due to consideration of the complex geometry and material characteristics of the motor. However, the iron loss model calculation amount based on the finite element method is very large, which brings great difficulty to the real-time acquisition of the iron loss. In addition, all the parameters required for motor control are parameters (calculation parameters) in an induction motor equivalent circuit. Therefore, in order to optimize the efficiency of the induction motor from the viewpoint of motor control, it is necessary to accurately account for the equivalent resistance of the iron loss in the equivalent circuit of the induction motor. The equivalent iron loss equivalent resistance needs to be calculated by using parameters in an equivalent circuit of the induction motor (obtained by using an analytical calculation method).
In order to obtain more accurate iron loss of the motor at the initial stage of the design of the induction motor, many scholars propose a plurality of iron loss calculation models from different angles. Wherein, the widely applied classical iron loss model is obtained by optimizing the Steinmetz equation; the classical iron loss model divides the iron loss into magnetic hysteresis loss and eddy current loss according to different factors generated by the iron loss, and the coefficients of the model are all constants. Due to the complex nonlinear characteristics of ferromagnetic materials, when the induction motor works in a large variation range of rotating speed and voltage amplitude, a classical iron loss model is not applicable any more. In order to consider the influence of harmonic magnetic field on the loss of stator and rotor iron cores of induction motor, the document "Zhang Dong, Zhao Hai Sen et al. 16-24, Zhang Dong, Guo Xin Zhi, etc. A DFT-based efficient separation method for the harmonic flux density of an induction motor rotor and the iron loss characteristics of a down-conversion motor rotor under a load condition [ J ] the report of electrotechnology, 2019,34(01):75-83 ] provides a segmented variable coefficient iron loss model which can accurately calculate the iron loss of the down-conversion motor. But the model is based on a finite element method and the computation speed is still very slow.
Disclosure of Invention
In order to overcome the defects of the prior art, the invention aims to provide a method for calculating the iron loss resistance of the variable frequency motor under the condition of PWM harmonic wave, which is based on a piecewise variable coefficient iron loss model, expresses the iron loss resistance of the induction motor as a function of induction potential and rotating speed, takes the additional iron loss generated by the harmonic wave of the PWM frequency converter and the surface and pulse vibration loss generated by the space harmonic wave of the motor into account, and has high processing precision.
In order to achieve the purpose, the invention adopts the following technical scheme to realize the purpose:
the invention discloses a method for calculating the iron loss resistance of a down-conversion motor under a PWM harmonic condition,
on the basis of the sectional variable coefficient model, replacing the flux density and the frequency variable in the sectional variable coefficient model with an induced potential and a rotating speed variable respectively; and calculating the change of the basic iron loss of the motor along with the power frequency in a sectional variable coefficient mode, so as to solve and obtain the basic iron loss of the down-conversion motor under the PWM harmonic condition.
Preferably, the specific method for establishing the basic iron loss of the down-conversion motor under the condition of considering the PWM harmonic wave is as follows:
hysteresis loss P expressed in terms of rotation speed and induced potentialH_sinAs shown in formula (8):
Figure BDA0002111238570000021
in the formula (I), the compound is shown in the specification,
Figure BDA0002111238570000022
Bmthe fundamental wave flux density amplitude value; k is a radical ofhAnd alpha is a classical hysteresis loss term coefficient; f is the fundamental frequency of the supply voltage; k is a radical of1And beta1To add a coefficient of magnetic density term, k, of hysteresis loss1And beta1Changes with magnetic density and frequency; n is a radical of*Each phase of the stator is connected with equivalent turns in series, and S is the cross section area of an equivalent iron core of the motor; em1Is the amplitude of the fundamental induced potential; synchronous speed omega1(ii) a p is the number of pole pairs of the motor;
eddy current loss P in terms of rotation speed and induced potentialE_sinAs shown in formula (9)
Figure BDA0002111238570000031
In the formula (I), the compound is shown in the specification,
Figure BDA0002111238570000032
keis a classical eddy current loss term coefficient; k is a radical of2And beta2To add the eddy current loss magnetic density term coefficient, k2And beta2Varying with magnetic density and frequency.
Preferably, the magnetic density and frequency variables of the piecewise variable coefficient model are replaced by induced potential and rotating speed variables, and meanwhile, the influence of PWM frequency converter harmonic waves on the basic iron loss of the motor is also taken into account by introducing coefficients related to the output voltage of the PWM frequency converter.
Further preferably, hysteresis and eddy current losses due to supply voltage harmonics are compensated by a factor related to the supply voltage of the induction machine, the compensated hysteresis loss PH_PWMAnd eddy current loss PE_PWMRespectively as follows:
Figure BDA0002111238570000033
PE_PWM＝χ2PE_sin (11)
Figure BDA0002111238570000034
Figure BDA0002111238570000035
in the formula, EavIs the average value of the induced potential; eav1Is the average value of the fundamental induced potential; ermsIs an effective value of the induced potential; erms1Is an effective value of the fundamental induced potential; e (t) is a function of the induced potential over time; and T is the fundamental period of the induced electromotive force.
Preferably, the influence of the motor tooth grooves on the iron loss of the variable frequency motor is also considered while the magnetic density and the frequency variable of the segmented variable coefficient model are replaced by the induction potential and the rotating speed variable.
Further preferably, the specific solution is as follows:
surface additional iron loss P generated by motor slottingsurfLAs shown in formula (16):
PsurfL＝CsurfLEm1 2Ω1.5 (16)
Figure BDA0002111238570000041
CsurfL＝KL1Csurf0 (18)
in the formula (I), the compound is shown in the specification,
Figure BDA0002111238570000042
Z1the number of stator teeth; alpha's'pTo calculate the pole arc coefficient, the coefficient is related to the core saturation level; lmThe motor shaft length; d2Is the outer diameter of the rotor; lδIs the motor air gap width; omega is the motor rotating speed; csurf0Adding a loss coefficient to the no-load surface; k is a radical of0Is a coefficient related to the material quality and the processing factor of the silicon steel sheet; kδ1For slotting the stator, rotorsAir gap coefficient when the surface is smooth; beta is a01The specific numerical value of the function of the width of the stator slot can be obtained by looking up a table; kL1The harmonic load coefficient of the stator teeth is the coefficient of which the value is related to the motor load, the size of the tooth grooves of the stator and the rotor and the like; t is t1The pitch of the stator teeth; t is t2Is the rotor pitch; b02Is the width b of the rotor slot02；
Pulse vibration loss P of induction motor stator caused by motor slottingpsLAnd rotor pulsation loss PprLRespectively as follows:
PpsL＝CpsLEm1 2Ω2 (19)
PprL＝CprLEm1 2Ω2 (20)
wherein the content of the first and second substances,
Figure BDA0002111238570000043
Figure BDA0002111238570000044
in the formula, Z2The number of rotor teeth; gt1And Gt2The weights of the motor stator and rotor teeth are respectively; kL1And KL2Harmonic load coefficients of the stator and rotor teeth are respectively; gamma ray1And gamma2The coefficients related to the stator and rotor slot widths, respectively, are given by:
Figure BDA0002111238570000051
total pulse vibration loss P of stator and rotor of variable frequency motorpLComprises the following steps:
Figure BDA0002111238570000052
the model makes the iron loss of the motor equal to the loss generated by the iron loss resistor on the induced potential, and the total iron loss of the variable frequency motor is shown as the formula (25):
Figure BDA0002111238570000053
iron loss equivalent resistance RFeAs shown in formula (26), formula (26) gives a feedback that the iron loss resistance changes with the change of the induced potential and the rotation speed:
Figure BDA0002111238570000061
compared with the prior art, the invention has the following beneficial effects:
1) the invention discloses a variable frequency motor iron loss resistance obtaining method, which replaces the flux density and frequency variable of a sectional variable coefficient model with an induced potential and a rotating speed variable, calculates the change of the basic iron loss of a motor along with the power frequency in a sectional variable coefficient mode, and introduces a coefficient related to the output voltage of a PWM frequency converter to calculate the influence of the harmonic wave of the PWM frequency converter on the basic iron loss of the motor. Therefore, great convenience is brought to the basic iron loss of the motor under the harmonic condition of the PWM frequency converter and the angle suppression of motor control;
2) the invention discloses a method for obtaining the iron loss resistance of a variable frequency motor, which considers the influence of a motor tooth space on the iron loss of the variable frequency motor and uses an equation of the amplitude value and the rotating speed of the partial loss induced potential fundamental wave. Therefore, great convenience is brought to the control angle of the motor for restraining the loss generated by the motor tooth grooves;
3) the method for obtaining the iron loss resistance is not only suitable for solving the iron loss resistance of the common variable frequency induction motor, but also can be used for a permanent magnet motor, a switched reluctance motor and other types of motors;
4) the invention can obtain the accurate variable frequency motor equivalent circuit under the condition of considering the iron loss. Specifically, taking a 5.5kW and a 30kW variable frequency induction motor as an example, the iron loss resistance change rules are obtained. And the two induction motors are used for verifying the effectiveness of the method.
Drawings
FIG. 1 is an equivalent circuit diagram of an induction motor in consideration of the iron loss condition of the motor;
FIG. 2 is a diagram of measured line voltage and current of a 5.5kW variable frequency motor during rated power supply;
FIG. 3 is a diagram of the measured line voltage and current of a 30kW variable frequency motor during rated power supply;
FIG. 4 is a change rule of iron loss resistance of the 5.5kW induction motor;
FIG. 5 is a change rule of iron loss resistance of the 30kW induction motor;
FIG. 6a is comparison between calculation and actual measurement of iron loss of a 5.5kW induction motor during sinusoidal power supply;
FIG. 6b is a comparison between the calculation and actual measurement of the iron loss of the 5.5kW induction motor during variable frequency power supply;
FIG. 7a is comparison between calculation and actual measurement of iron loss of a 30kW induction motor during sinusoidal power supply;
and FIG. 7b is a comparison between the calculation and actual measurement of the iron loss of the 30kW induction motor during variable frequency power supply.
Detailed Description
In order to make the technical solutions of the present invention better understood, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
It should be noted that the terms "first," "second," and the like in the description and claims of the present invention and in the drawings described above are used for distinguishing between similar elements and not necessarily for describing a particular sequential or chronological order. It is to be understood that the data so used is interchangeable under appropriate circumstances such that the embodiments of the invention described herein are capable of operation in sequences other than those illustrated or described herein. Furthermore, the terms "comprises," "comprising," and "having," and any variations thereof, are intended to cover a non-exclusive inclusion, such that a process, method, system, article, or apparatus that comprises a list of steps or elements is not necessarily limited to those steps or elements expressly listed, but may include other steps or elements not expressly listed or inherent to such process, method, article, or apparatus.
The invention is described in further detail below with reference to the accompanying drawings:
the invention provides a calculation method for the iron loss resistance of a down-conversion motor under the condition of considering PWM harmonic waves, which is based on a segmented variable coefficient iron loss model and expresses the iron loss resistance of an induction motor as a function of induction potential and rotating speed. The method accounts for the additional iron losses generated by the PWM converter harmonics and the surface and pulse losses generated by the motor space harmonics. The method for establishing the down-conversion motor iron loss resistance under the PWM harmonic condition comprises the following steps:
the classic iron loss model is limited in that when the magnetic flux density is greater than 1.2T or the frequency exceeds 400Hz, the iron loss value calculated by the classic model is smaller than the measured value; all coefficients in the classical model are constant coefficients, so that the method cannot be applied to the condition that the amplitude or the frequency variation range of the magnetic density of the motor core is large.
In order to obtain a more accurate predicted value when the variation of the supply voltage and the rotation speed of the induction motor is large, the hysteresis loss of the induction motor can be obtained by the following formula:
Figure BDA0002111238570000081
in the formula, k1And beta1To add hysteresis loss flux density term coefficients, they vary with flux density and frequency.
The eddy current loss of an induction motor can be obtained by the following formula:
Figure BDA0002111238570000082
in the formula, k2And beta2To add to the eddy current loss flux density term coefficients, they vary with flux density and frequency.
The magnetic density distribution of the iron core tooth part and the yoke part of the induction motor is not uniform, so that the difficulty is brought to solving the equivalent sectional areas of the stator tooth part and the yoke part of the motor and the rotor tooth part and the yoke part of the motor. When the equivalent area of the motor core is solved, the nonuniformity of the magnetic density distribution of the tooth part and the yoke part of the induction motor core is ignored. Under the condition of neglecting space harmonic and leakage magnetic flux generated by self structural factors due to tooth grooves of the induction motor and the like, the magnetic density of a yoke part of the motor is considered to only run in the tangential direction, and the magnetic density of a tooth part of the motor only runs in the radial direction. And since the induction machine slip frequency is typically very low, the basic iron losses on the rotor side are ignored. At this time, the relationship between the induced potential e (t) and the magnetic flux density change rate is
Figure BDA0002111238570000083
In the formula, N*The equivalent number of turns of each phase of the stator in series connection, S is the cross section area of an equivalent iron core of the motor, t is time, and B (t) magnetic flux density.
From formula (3):
Figure BDA0002111238570000084
the relationship between the fundamental potential and the fundamental flux density is:
Figure BDA0002111238570000091
wherein E ism1Is the amplitude of the fundamental induced potential.
Following synchronous speed omega of induction machine1Replacing the supply voltage fundamental frequency f. The relationship between them is:
Figure BDA0002111238570000092
in the formula, p is the pole pair number of the motor.
By bringing formula (6) into formula (5), it is possible to obtain:
Figure BDA0002111238570000093
the hysteresis loss expressed by the rotation speed and the induced potential can be obtained by bringing formula (7) into formula (1), as shown in formula (8):
Figure BDA0002111238570000094
similarly, taking equation (7) into equation (2) can find the eddy current loss expressed by the rotation speed and the induced potential, as shown in equation (9):
Figure BDA0002111238570000101
because the induction motor generally applies variable frequency speed regulation at present, the power supply voltage of the induction motor is not a standard sine wave. Hysteresis and eddy current losses caused by supply voltage harmonics can be compensated by coefficients related to the supply voltage of the induction machine, the compensated hysteresis and eddy current losses being respectively:
Figure BDA0002111238570000102
PE_PWM＝χ2PE_sin (11)
in the formula (I), the compound is shown in the specification,
Figure BDA0002111238570000103
Figure BDA0002111238570000104
at present, induction motors generally adopt a distributed winding structure to eliminate air gap magnetic fields generated by sub-space harmonic magnetomotive forces of 5, 7, 11 and 13, and even though a motor stator adopts a whole-pitch winding, the air gap magnetic fields are compared with those generated by a stator of the motorThe iron loss generated by the first-order tooth harmonic magnetic field is negligible, and the iron loss generated by the 5, 7, 11, 13 and other sub-space harmonic magnetomotive force is negligible. Thus, the present invention accounts for iron losses generated by first-order tooth harmonics only. Air gap flux density B when voltage harmonics are ignoredδThis can be found by the following equation:
Figure BDA0002111238570000105
in the formula, BδAir gap flux density; alpha's'pFor calculating the pole arc coefficient, the saturation degree of the iron core is related; lδIs the motor air gap width.
The tooth harmonic magnetic field caused by stator slotting generates surface loss and stator tooth pitch t on the surface of the rotor1Rotor pitch t2Motor speed omega and rotor slot width b02And the like. At no load, the surface parasitic loss can be obtained from equation (15)
Psurf0＝Csurf0Em1 2Ω1.5 (15)
In the formula, Csurf0A loss factor is added to the unloaded surface.
The surface loss under load can be determined by equation (16):
PsurfL＝CsurfLEm1 2Ω1.5 (16)
Figure BDA0002111238570000111
CsurfL＝KL1Csurf0 (18)
in the formula, k0Is a coefficient related to the material quality and the processing factor of the silicon steel sheet; kδ1Air gap coefficient when the stator is grooved and the surface of the rotor is smooth; beta is a01The specific numerical value of the function of the width of the stator slot can be obtained by looking up a table; kL1The harmonic load coefficient of the stator teeth is a coefficient of which the value is related to the motor load, the size of the tooth grooves of the stator and the rotor and the like.
The corresponding relation of the fixed rotation tooth grooves of the motor is changed continuously due to the tooth groove effect of the motor. The permanent rotation tooth slot corresponding relation will change constantly and will cause the change of the iron core magnetic conductance, and then lead to the stator rotor tooth portion magnetic density to fluctuate. The pulse vibration loss of the stator and the rotor of the induction motor caused by the fluctuation of the magnetic density of the teeth parts of the stator and the rotor is respectively as follows:
PpsL＝CpsLEm1 2Ω2 (19)
PprL＝CprLEm1 2Ω2 (20)
in the formula (I), the compound is shown in the specification,
Figure BDA0002111238570000112
Figure BDA0002111238570000121
in the formula, KL2Is the rotor tooth harmonic load coefficient; gamma ray1And gamma2The coefficients related to the stator and rotor slot widths, respectively, are given by:
Figure BDA0002111238570000122
therefore, the total pulsating loss of the stator and the rotor of the induction motor is as follows:
Figure BDA0002111238570000123
the model equates the motor iron losses to losses due to iron loss resistance at the induced potential. The total iron loss of the variable frequency motor is shown as the formula (25).
Figure BDA0002111238570000124
Wherein, the iron loss equivalent resistance RFeAs shown in equation (26):
Figure BDA0002111238570000131
as can be seen from equation (26), the iron loss resistance varies with changes in the induced potential and the rotation speed. And the expression expresses the iron loss resistance of the variable frequency motor as a function of the induced potential fundamental wave potential and the rotating speed, and is very easy to obtain.
Example 1
The parameters of a 5.5kW variable frequency induction motor are shown in Table 1. Under nominal conditions, the voltage and current waveforms of the two motors are shown in figure 2. The change of the iron loss resistance along with the rotating speed and the modulation factor of the PWM frequency converter is calculated by using the solving method of the iron loss resistance provided by the invention as shown in figure 4. The switching frequency of the frequency converter is 5kHz, and when the rotating speed is lower than the rated synchronous rotating speed, the ratio of the induced potential to the frequency is ensured to be unchanged; and when the synchronous rotating speed exceeds the rated synchronous rotating speed, taking the induction potential as a constant value. It can be seen that the higher the modulation factor of the PWM inverter, the higher the rotation speed of the induction motor, the larger the iron loss equivalent resistance.
Example 2
The parameters of a 30kW variable frequency induction motor are shown in Table 2. Under nominal conditions, the voltage and current waveforms for the two motors are shown in figure 3. The change of the iron loss resistance along with the rotating speed and the modulation factor of the PWM frequency converter is calculated by using the solving method of the iron loss resistance provided by the invention as shown in the attached figure 5. The switching frequency of the frequency converter is 5kHz, and when the rotating speed is lower than the rated synchronous rotating speed, the ratio of the induced potential to the frequency is ensured to be unchanged; and when the synchronous rotating speed exceeds the rated synchronous rotating speed, taking the induction potential as a constant value. It can be seen that the higher the modulation factor of the PWM inverter, the higher the rotation speed of the induction motor, the larger the iron loss equivalent resistance.
Example 3
The method of the invention based on the analytical method, the classical model based on the time-step finite element and the piecewise variable coefficient model are respectively utilized to calculate the iron loss of a variable frequency induction motor with the specification of 5.5kW as shown in the table 1 under the sinusoidal power supply conditions of different power supply voltages. The measurement and simulation pair is shown in fig. 6a and 6b, for example. It can be seen that the calculated value of the piecewise coefficient variable model is very close to the measured value.
Example 4
The method of the invention based on the analytical method, the classical model based on the time-step finite element and the piecewise variable coefficient model are respectively utilized to calculate the iron loss of a variable frequency induction motor with the specification of 30kW as shown in the table 2 under the condition of different power supply voltage sine power supply. The measurement and simulation pair is shown in fig. 7a and 7b, for example. It can be seen that the calculated value of the method and the piecewise variable coefficient model is very close to the measured value.
TABLE 15.5 kW Induction machine gauge parameters
Figure BDA0002111238570000141
Table 230 kW induction machine specification parameters
Figure BDA0002111238570000142
Figure BDA0002111238570000151
In summary, the method for obtaining the iron loss resistance of the down-conversion motor under the condition of considering the PWM harmonic wave disclosed by the present invention represents the iron loss of the induction motor as a function of the induction potential and the rotation speed based on the sectional variable coefficient iron loss model. The method takes the additional iron loss generated by the harmonic waves of the PWM frequency converter into account, and takes the influence of the space harmonic wave component of the motor on the tooth surface of the stator and the rotor and the pulse vibration loss into account. Based on the analytic calculation method of the iron loss of the PWM variable-frequency power supply induction motor, the equivalent circuit model of the variable-frequency induction motor under the condition of considering the iron loss is obtained. Finally, taking a 5.5kW variable frequency induction motor and a 30kW variable frequency induction motor as examples, the equivalent resistance change rules are obtained by solving by using the method provided by the invention. The method, the classical model based on the time-step finite element and the segmented variable coefficient model based on the time-step finite element are respectively utilized to calculate the iron loss of the two variable frequency induction motors at different rotating speeds, and compared with actual measurement, the comparison result shows that the precision of the method is higher.
The above-mentioned contents are only for illustrating the technical idea of the present invention, and the protection scope of the present invention is not limited thereby, and any modification made on the basis of the technical idea of the present invention falls within the protection scope of the claims of the present invention.

Claims (5)

1. A calculation method for calculating the iron loss resistance of a down-conversion motor under a PWM harmonic condition is characterized by comprising the following steps of:
on the basis of the sectional variable coefficient model, replacing the flux density and the frequency variable in the sectional variable coefficient model with an induced potential and a rotating speed variable respectively; calculating the change of the basic iron loss of the motor along with the power frequency in a sectional variable coefficient mode, so as to solve and obtain the basic iron loss of the down-conversion motor under the PWM harmonic condition;
the specific method for establishing the down-conversion motor basic iron loss under the PWM harmonic condition is as follows:
hysteresis loss P expressed in terms of rotation speed and induced potentialH_sinAs shown in formula (8):
Figure FDA0002831060890000011
in the formula (I), the compound is shown in the specification,
Figure FDA0002831060890000012
khand alpha is a classical hysteresis loss term coefficient; k is a radical of1And beta1To add a coefficient of magnetic density term, k, of hysteresis loss1And beta1Changes with magnetic density and frequency; n is a radical of*Each phase of the stator is connected with equivalent turns in series, and S is the cross section area of an equivalent iron core of the motor; em1Is the amplitude of the fundamental induced potential; synchronous speed omega1(ii) a p is the number of pole pairs of the motor;
eddy current loss in terms of rotational speed and induced potentialConsumption PE_sinAs shown in formula (9)
Figure FDA0002831060890000013
In the formula (I), the compound is shown in the specification,
Figure FDA0002831060890000014
keis a classical eddy current loss term coefficient; k is a radical of2And beta2To add the eddy current loss magnetic density term coefficient, k2And beta2Varying with magnetic density and frequency.
2. The method for calculating the iron loss resistance of the down-conversion motor under the condition of considering the PWM harmonic waves as claimed in claim 1, wherein the influence of the harmonic waves of the PWM frequency converter on the basic iron loss of the motor is also calculated by introducing a coefficient related to the output voltage of the PWM frequency converter while replacing the flux density and the frequency variable of the piecewise variable coefficient model with an induced potential and a rotating speed variable.
3. The method of claim 2, wherein hysteresis and eddy current losses due to supply voltage harmonics are compensated by a factor related to the supply voltage of the induction machine, and wherein the compensated hysteresis loss P is calculated by taking into account the PWM harmonic condition down-conversion motor iron loss resistanceH_PWMAnd eddy current loss PE_PWMRespectively as follows:
Figure FDA0002831060890000021
PE_PWM＝χ2PE_sin (11)
Figure FDA0002831060890000022
Figure FDA0002831060890000023
in the formula, EavIs the average value of the induced potential; eav1Is the average value of the fundamental induced potential; ermsIs an effective value of the induced potential; erms1Is an effective value of the fundamental induced potential; e (t) is a function of the induced potential over time; t is the period of the fundamental wave content in the induced potential.
4. The method for calculating the iron loss resistance of the down-conversion motor under the PWM harmonic condition according to claim 3, wherein the influence of motor tooth slots on the iron loss of the down-conversion motor is considered while the flux density and the frequency variable of the piecewise variable coefficient model are replaced by the induction potential and the rotation speed variable.
5. The method for calculating the iron loss resistance of the down-conversion motor under the PWM harmonic condition according to claim 4, wherein the specific solution is as follows:
surface additional iron loss P generated by motor slottingsurfLAs shown in formula (16):
PsurfL＝CsurfLEm1 2Ω1.5 (16)
Figure FDA0002831060890000024
CsurfL＝KL1Csurf0 (18)
in the formula (I), the compound is shown in the specification,
Figure FDA0002831060890000025
Z1the number of stator teeth; alpha's'pTo calculate the pole arc coefficient, the coefficient is related to the core saturation level; lmThe motor shaft length; d2Is the outer diameter of the rotor; lδIs the motor air gap width; omega is the motor rotating speed; csurf0Adding a loss coefficient to the no-load surface; k is a radical of0Is related to the material and processing factor of silicon steel sheetA coefficient of correlation; kσ1Air gap coefficient when the stator is grooved and the surface of the rotor is smooth; beta is a01The specific numerical value of the function of the width of the stator slot can be obtained by looking up a table; kL1The harmonic load coefficient of the stator teeth is determined by the motor load and the size of the stator and rotor tooth spaces; t is t1The pitch of the stator teeth; t is t2Is the rotor pitch; b02Is the rotor slot width;
pulse vibration loss P of induction motor stator caused by motor slottingpsLAnd rotor pulsation loss PprLRespectively as follows:
PpsL＝CpsLEm1 2Ω2 (19)
PprL＝CprLEm1 2Ω2 (20)
wherein the content of the first and second substances,
Figure FDA0002831060890000031
Figure FDA0002831060890000032
in the formula, Z2The number of rotor teeth; gt1And Gt2The weights of the motor stator and rotor teeth are respectively; kL1And KL2Harmonic load coefficients of the stator and rotor teeth are respectively; gamma ray1And gamma2The coefficients related to the stator and rotor slot widths, respectively, are given by:
Figure FDA0002831060890000033
total pulse vibration loss P of stator and rotor of variable frequency motorpLComprises the following steps:
Figure FDA0002831060890000034
the model enables the iron loss of the motor to be equivalent to the loss generated by iron consumption resistance in an equivalent circuit diagram of the induction motor, and the total iron loss of the variable frequency motor is shown as the formula (25):
Figure FDA0002831060890000041
Figure FDA0002831060890000042
iron loss equivalent resistance RFeAs shown in formula (26), formula (26) gives a feedback that the iron loss resistance changes with the change of the induced potential and the rotation speed:
Figure FDA0002831060890000043
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