CN108667381B - TLDMC-PMSM system control method based on dynamic torque hysteresis - Google Patents

TLDMC-PMSM system control method based on dynamic torque hysteresis Download PDF

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CN108667381B
CN108667381B CN201810374959.XA CN201810374959A CN108667381B CN 108667381 B CN108667381 B CN 108667381B CN 201810374959 A CN201810374959 A CN 201810374959A CN 108667381 B CN108667381 B CN 108667381B
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matrix converter
level direct
flux linkage
torque
direct matrix
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CN108667381A (en
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程启明
陈路
孙伟莎
李涛
程尹曼
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Shanghai University of Electric Power
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
    • H02P23/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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Abstract

The invention relates to a TLDMC-PMSM system control method based on dynamic torque hysteresis, which comprises the following steps: 1. collecting three-phase input voltage and three-phase output current of a three-level direct matrix converter and rotor rotating speed of a permanent magnet synchronous motor; 2. calculating to obtain an input current space vector and an output voltage space vector of the three-level direct matrix converter; 3. observing the average value of the electromagnetic torque, the stator flux linkage and the input power factor of the permanent magnet synchronous motor; 4. and 3, obtaining a switching signal of the three-level direct matrix converter by adopting a direct torque control mode according to the observed value obtained in the step 3, and realizing system control. Compared with the prior art, the three-level direct matrix converter can reduce the problem of stator flux linkage imbalance of the permanent magnet synchronous motor in low-speed operation, and reduces the common-mode voltage output by the system while improving the electromagnetic torque and stator flux linkage control performance of the system.

Description

TLDMC-PMSM system control method based on dynamic torque hysteresis
Technical Field
The invention relates to a control method of a permanent magnet synchronous motor, in particular to a TLDMC-PMSM system control method based on dynamic torque hysteresis.
Background
Matrix Converters (MC) are a new type of ac power Converter with many advantages: energy flows in two directions and can run in four quadrants; sine input/output, small harmonic distortion rate; adjustable power factor, etc. Permanent Magnet Synchronous Motors (PMSM) adopt Permanent Magnet materials to realize Motor excitation, and have high operating power factor, high efficiency and high power density. Therefore, the matrix converter-permanent magnet synchronous motor (MC-PMSM) system has wide application prospect. Direct Torque Control (DTC) is a simple and powerful new advanced Control technique: 1) the structure is simple, and complex coordinate transformation is not needed; 2) a current controller is not needed, so that the hardware cost is reduced; 3) the quick and accurate torque control response can be realized; 4) but also in a sensorless mode, etc.
To combine the advantages of the three, some have applied DTC to MC-PMSM systems. However, the high dynamic performance of the DTC makes the MC-PMSM system electromagnetic torque and stator flux ripple still large. To solve this problem, the relevant scholars have done a lot of research work: 1) the improvement of the traditional DTC comprises the expansion and optimization of a switching table, the adoption of a duty ratio control method, the DTC based on space vector modulation and the like; 2) novel control theories are adopted to be combined with DTC, such as fuzzy control, neural networks, predictive torque control methods and the like; 3) it is proposed to have more control vectors, for example using the discrete space vector approach, using new converters, etc. The low speed performance of the above solution is not ideal. When the rotor is at low speed and zero speed, the stator flux linkage is easy to distort due to the long acting time of the zero vector selected for reducing the electromagnetic torque, so that the stator flux linkage cannot be accurately adjusted, the stator winding current is deteriorated, the coupling between the stator and the rotor is reduced, and the operation of the motor is not favorable.
In recent years, a Three-Level Direct Matrix Converter (TLDMC) combines the characteristics of a Matrix Converter and a Three-Level inverter, and can further reduce the output total harmonic distortion, reduce the voltage stress of a switching device, reduce the common-mode voltage and the high power density, so that the Three-Level Direct Matrix Converter is applied to a permanent magnet synchronous motor Direct torque control system, and can improve the control performance of electromagnetic torque and stator flux linkage.
Disclosure of Invention
The invention aims to overcome the defects of the prior art and provide a TLDMC-PMSM system control method based on dynamic torque hysteresis.
The purpose of the invention can be realized by the following technical scheme:
a dynamic torque hysteresis based TLDMC-PMSM system control method, wherein a permanent magnet synchronous motor is controlled by a three-level direct matrix converter, comprises the following steps:
s1, collecting three-phase input voltage and three-phase output current of the three-level direct matrix converter and rotor rotating speed of the permanent magnet synchronous motor;
s2, calculating an input current space vector and an output voltage space vector of the three-level direct matrix converter through the three-phase input voltage, the three-phase output current and the switching function matrix of the three-level direct matrix converter;
s3, observing the electromagnetic torque, the stator flux linkage and the input power factor average value of the permanent magnet synchronous motor through the input current space vector and the output voltage space vector of the three-level direct matrix converter to obtain an observed value;
and S4, obtaining a switching signal of the three-level direct matrix converter by adopting a direct torque control mode according to the observation value obtained in the step S3, and realizing system control.
Preferably, the calculation formula of the output voltage space vector in step S2 is as follows:
Figure BDA0001639360890000021
wherein u isoA、uoB、uoCThe three-phase output voltage of the three-level direct matrix converter is represented by the following calculation formula:
Figure BDA0001639360890000022
wherein S (t) represents the trigger signal S of 12 bidirectional switches included in the circuit topology of the three-level direct matrix converterijThe corresponding switch function matrix, i belongs to { A, B, C }, j belongs to { a, B, C, n }, u }iRepresenting three-phase input voltage u of a three-level direct matrix converteria、uib、uicObtaining a three-phase input voltage space vector through vector transformation;
the input current space vector is:
Figure BDA0001639360890000031
wherein iia、iib、iicThe three-phase input current of the three-level direct matrix converter is represented by the following calculation formula:
Figure BDA0001639360890000032
where T represents the transpose of the matrix, ioRepresenting three-phase output currents i of a three-level direct matrix converteroA、ioB、ioCThree-phase output current space vector, i, obtained by vector transformationnIs the neutral point current.
Preferably, in step S3, the observing the stator flux linkage of the permanent magnet synchronous motor includes: respectively calculating alpha-axis component psi of stator flux linkagesαStator flux linkage beta-axis component psisβAnd stator flux linkage amplitude | ψs|：
ψsα＝∫(usα-Rsisα)dt
ψsβ＝∫(usβ-Rsisβ)dt
Figure BDA0001639360890000033
Wherein R issRepresenting stator winding resistance, usα、usβAnd isα、isβRespectively representing stator terminal voltages usAnd stator winding current isAlpha, beta components, u, in a stationary coordinate systemsOutput voltage space vector, i, equal to three-level direct matrix convertersThe three-phase output current space vector is obtained by vector transformation of three-phase output current equal to the three-level direct matrix converter;
the observing the electromagnetic torque of the permanent magnet synchronous motor comprises: calculating electromagnetic torque Te：
Te＝1.5p(ψsαisβ-ψsβisα)
Wherein p is the number of pole pairs;
the observing the input power factor average value comprises: calculating an input voltage space vector uiPhase difference with input current space vector
Figure BDA0001639360890000034
Obtaining an average value of input power factors
Figure BDA0001639360890000035
Wherein u isiThe three-phase input voltage through the three-level direct matrix converter is obtained through vector conversion.
Preferably, the step S4 includes:
s41, passing the error between the stator flux set value and the observed stator flux amplitude through a first two-stage hysteresis comparator to obtain an output CψObtaining a torque reference value by a PI regulator according to the set reference rotating speed and the actual rotating speed of the rotor of the permanent magnet synchronous motor, and obtaining an output C by a dynamic hysteresis comparator according to the error between the torque reference value and the observed electromagnetic torqueTThe power factor set value 0 and the observed input power factor average value pass through a second two-stage hysteresis comparator to obtain output
Figure BDA0001639360890000036
S42, according to Cψ、CT、
Figure BDA0001639360890000037
And inquiring and selecting a proper voltage vector in a formulated direct torque control switch table to obtain a switching signal of the three-level direct matrix converter, thereby realizing system control.
Preferably, the dynamic hysteresis comparator comprises a plurality of different hysteresis comparators connected in parallel, and an appropriate hysteresis comparator is selected according to the error of the stator flux linkage amplitude and the set lower limit of flux linkage maladjustment.
Preferably, the dynamic hysteresis comparator comprises a five-stage hysteresis comparator and a four-stage hysteresis comparator which are connected in parallel.
Compared with the prior art, the invention has the following advantages:
1. the three-level direct matrix converter has more control vectors than the traditional matrix converter, can effectively reduce the electromagnetic torque and stator flux linkage pulsation in the steady state of a system in a medium-high speed operation range, and can reduce the common-mode voltage output by the converter.
2. A dynamic torque hysteresis method is adopted in direct torque control, and when stator flux linkage misadjustment occurs during low-speed operation of a system, the stator flux linkage misadjustment can be improved by selecting a reverse voltage vector to replace a zero vector, so that the flux linkage adjustment performance of the system is effectively improved.
Drawings
FIG. 1 is a schematic diagram of a three level direct matrix converter-permanent magnet synchronous motor (TLDMC-PMSM) system of the present invention;
FIG. 2 is a process diagram of the control method of the present invention;
FIG. 3 is a schematic diagram of the space vector of the output voltage of the three-level direct matrix converter according to the present invention;
FIG. 4 is a schematic diagram of the space vector of the input current of the three-level direct matrix converter according to the present invention;
FIG. 5 is a schematic view of flux linkage reduction during low speed operation of a conventional matrix converter-PMSM system;
FIG. 6 shows a voltage vector u3A schematic diagram of the variation of the radial component and the tangential component at sector 2;
FIG. 7 is a graph of simulation results of radial and tangential components of a voltage vector affecting flux linkage adjustment;
FIG. 8 is a schematic diagram of discrete waveforms of flux linkage, torque and torque states during hysteresis comparator dynamics;
FIG. 9 is a schematic diagram of a dynamic hysteresis comparator according to the present invention;
FIG. 10 is a waveform diagram of a simulation of a conventional MC-DTC at a low speed and a light load in an embodiment;
FIG. 11 is a waveform diagram of the simulation at low speed with light load of the improved MC-DTC in the embodiment;
FIG. 12 is a simulation waveform diagram of TLDMC-DTC at low speed with light load in the embodiment;
FIG. 13 is a waveform diagram of simulation at a rotation speed of 5rad/s and a light load of 2 N.m in the example;
FIG. 14 is a waveform diagram of a conventional MC-DTC load-rated start-up simulation in an embodiment;
FIG. 15 is a waveform of an embodiment of an improved MC-DTC load-rated start simulation;
FIG. 16 is a waveform of load-rated start simulation of TLDMC-DTC in an embodiment.
Detailed Description
The invention is described in detail below with reference to the figures and specific embodiments. The present embodiment is implemented on the premise of the technical solution of the present invention, and a detailed implementation manner and a specific operation process are given, but the scope of the present invention is not limited to the following embodiments.
The application provides a control method of a TLDMC-PMSM system based on dynamic torque hysteresis, wherein a permanent magnet synchronous motor is controlled in the TLDMC-PMSM system through a three-level direct matrix converter, as shown in figure 1. The three-level direct matrix converter (TLDMC) structure has 3 more bidirectional switches than the traditional matrix converter, and a neutral line n of an input three-phase R-L-C low-pass filter is connected to an output switch of each phase to form a 4 x 3 matrix circuit. When the topology is switched between different switch states, the amplitude of each phase output voltage can be between three-phase input voltage and neutral line voltage unTo switch between. In the figure, uia、uib、uicIs a three-phase input voltage; i.e. iia、iib、iicThree-phase input current; l isf、CfAnd RfThe filter inductor, the filter capacitor and the damping resistor are respectively input into the R-L-C low-pass filter; u. ofoA、uoB、uoCThe voltage of the stator end of the permanent magnet synchronous motor is the three-phase output voltage of the three-level direct matrix converter; i.e. ioA、ioB、ioCThe current of the stator winding of the permanent magnet synchronous motor is the three-phase output current of the three-level direct matrix converter;Sija bi-directional switch having bi-directional turn-off and bi-directional turn-on capabilities is shown.
The on and off states of the bi-directional switch are defined as:
Figure BDA0001639360890000051
according to the safety principle that the operation process of the matrix converter must follow, the limiting conditions of the switching function can be obtained as follows:
Sia+Sib+Sic+Sin＝1 i∈{A,B,C}
the three-level direct matrix converter also adopts a fully-controlled bidirectional switch, so that the control mode is a chopping control mode, and the output quantity and the input quantity can be represented by a switch function. According to the topology of the three-level direct matrix converter shown in fig. 1, the relationships between the output phase voltage and the input phase voltage, and between the input phase current and the output phase current are respectively obtained as follows:
Figure BDA0001639360890000052
Figure BDA0001639360890000061
in the formula: s (t) is a trigger signal S of 12 bidirectional switches contained in a circuit topology of a three-level direct matrix converterijThe corresponding switch function matrix, T represents the transposition of the matrix, uiRepresenting a three-phase input voltage space vector, i, obtained by vector conversion of the three-phase input voltage of a three-level direct matrix converteroRepresenting a three-phase output current space vector, i, obtained by vector conversion of the three-phase output currents of a three-level direct matrix converternIs the midpoint current.
The circuit topology has a total of 64 control vectors, of which 36 are valid vectors, 4 are zero vectors, and 24 rotation vectors that are not temporarily used for DTC control. It can be seen that the three levels are directThe matrix converter has much more active vectors than are available in conventional matrix converters and therefore can further reduce the electromagnetic torque and the stator flux ripple. Furthermore, a new zero vector 0 is addednThe common mode voltage is zero, which can further reduce the common mode voltage.
Of the 36 valid vectors, the first 18 vectors are defined as MC (+ -1 to 9), as shown in Table 1. The last 18 small vectors (± 10 '± 18') exist to make the neutral point current inThe phenomenon is not zero, so it is necessary to adopt a vector synthesis method to combine the 18 different switches so as to satisfy the condition that the average value of the current flowing through the neutral point is zero (± 10 to ± 18), as shown in table 2. In the table, uab、ubc、ucaFor inputting line three-phase voltages, iA、iB、iCThree-phase output current.
TABLE 1 conventional matrix converter switch states
Figure BDA0001639360890000062
Figure BDA0001639360890000071
TABLE 2 TLDMC switch states at zero midpoint current
Figure BDA0001639360890000072
Figure BDA0001639360890000081
The space vector diagram of the three-level direct matrix converter, as shown in fig. 3 and 4, shows that the selectivity of the effective vector is more. Therefore, the DTC method of dividing the magnitude vector is adopted, i.e. the torque hysteresis comparator is in five-stage form (2, 1, 0, -1, -2), and the specific implementation is shown in table 3 and table 4. In the table, L represents a large vector, and S represents a small vector.
TABLE 3 DTC SWITCH METER FOR TORQUE FIVE-STAGE LACKING RING
Figure BDA0001639360890000082
Table 4 DTC switching table using TLDMC
Figure BDA0001639360890000091
The table does not contain the selection mode of the zero vector, and the invention adopts the newly added 0nAs a selection of zero vectors, i.e. switches SAn、SBnAnd SCnAnd conducting, wherein the amplitude of the generated common mode voltage is minimum and is equal to zero.
However, when the direct torque control of the three-level direct matrix converter-permanent magnet synchronous motor system under low-speed operation uses a zero vector, the problem that the stator flux linkage is easy to be maladjusted still exists, and for this reason, a mathematical model of the permanent magnet synchronous motor needs to be analyzed, so that the principle of the dynamic torque hysteresis comparator method adopted by the invention is introduced.
Neglecting the saturation of the iron core of the motor, neglecting the eddy current and hysteresis loss in the motor, and the rotor has no damping winding, the mathematical model of the permanent magnet synchronous motor under the static reference coordinate system is as follows:
Figure BDA0001639360890000092
ψs＝Lsis+ψr
Figure BDA0001639360890000093
Te＝1.5pψs×is＝1.5p(ψαiβ-ψβiα)
in the formula: rs、LsRespectively stator winding resistance, electricityA sub-inductor; u. ofs、is、ψsAnd psirStator terminal voltage, stator winding current, stator flux linkage and rotor flux linkage vectors are respectively; p is the number of pole pairs; t iseIs an electromagnetic torque; thetaeIs the rotor electrical angle.
Therefore, the electromagnetic torque change rate can be obtained as follows:
Figure BDA0001639360890000101
then, during a time interval Δ t during which each voltage space vector acts, the magnitude of the change in electromagnetic torque is:
ΔTe(≈dTe)＝ΔT1+ΔT2+ΔT3
wherein:
Figure BDA0001639360890000102
Figure BDA0001639360890000103
Figure BDA0001639360890000104
as can be seen from the above equation, Δ T1And stator winding resistance R in motor parameterssStator inductance LsRelatively, the amplitude is small and always negative; delta T2With the rotor speed omegarThe motor has great relevance, and has great difference when the motor runs at low speed and high speed, and is always a negative value; delta T3Dependent on the effective vector usIs varied by the rotor flux linkage psirAnd the effective vector usThe included angles are related. At low speed operation, ωrInduced Δ T2Small and thus zero vector participation, resulting in an electromagnetic torque variation Δ TeThe electromagnetic torque ripple reduction device is small, can slowly reduce the electromagnetic torque, and effectively reduces the electromagnetic torque ripple.
On the other hand, the mathematical expression of the stator flux linkage is as follows:
Figure BDA0001639360890000105
during a time interval Δ t during which each voltage space vector acts, the change in the stator flux linkage vector may describe the following relationship:
Δψs1＝(us-Rsis)·Δt
when the input voltage vector is an effective voltage vector, neglecting the influence of the resistance voltage drop of the stator, the above formula can be simplified as follows:
Δψs1＝us·Δt
it can be analytically determined that the electromagnetic torque T is reduced when requiredeWhen, according to the switching table, the selected zero vector will maintain the stator flux linkage psisIs substantially unchanged. However, in practical applications, especially at low PMSM speeds, the zero vector actually acts to reduce stator flux linkage due to the large stator resistive voltage drop. Therefore, when the stator resistance voltage drop is not negligible, the stator flux linkage variation caused by the zero vector is as follows:
Δψs2＝-Rsis·Δt
Figure BDA0001639360890000106
Figure BDA0001639360890000111
in the formula: st +、St -The rate of change of the electromagnetic torque, respectively increasing or decreasing, is mainly influenced by several factors:
1) electromagnetic torque Te；
2) Motor rotor speed (i.e. rotor electrical angular velocity) omegar；
3) StatorSub-terminal voltage vector us。
Wherein T iseAnd ωrCan greatly influence St -FIG. 5 shows the simulation results of the conventional MC-DTC in the low speed range (5 rad/s). The waveforms of the figure from top to bottom are the rotor rotation speed omegarStator flux linkage amplitude | ψsL, electromagnetic torque TeAnd output C of the torque hysteresis comparatorT. As can be seen from the figure, in zero vector selection (C)T0) the flux linkage amplitude is reduced by | Δ ψs2L. In this example, | Δ ψ appearss1|<|Δψs2And | the flux linkage amplitude is integrally reduced, and the stator flux linkage amplitude adjustment is influenced in serious cases. The main reason for this is because the rate of change (S) of the electromagnetic torque is increasedt +) Much greater than the rate of change (S) that reduces the electromagnetic torquet -) The action time of the zero vector is longer than that of the effective voltage vector, which is more common in light-load and low-speed operation.
For the third factor usIt helps to increase St +. Radial component u of stator voltage vectorsαAnd a tangential component usβRespectively linked with stator flux psisAnd electromagnetic torque TeIs concerned with the dynamic behavior of. With phisRotation of usαAnd usβIs also according to phisThe change in position. FIG. 6 shows the stator flux linkage within sector 2, with voltage vector u3The radial component and the tangential component of (a) vary.
In FIG. 5, whensEntering the initial stage of sector 2, the effective vector u3Radial component u of3αVery small, tangential component u3βBut is quite large. In this case, since the duration of the effective voltage vector is short, when u is selected3At this time, the increment of the stator flux linkage is small, so that | Δ ψ appearss1|<|Δψs2L. When the stator flux linkage moves to the middle of sector 2, the voltage vector u3αBecomes large and its tangential component u becomes large3βThe stator flux linkage is increased with a decrease. However, if the motor is rotatingThe speed is very low, and the stator flux linkage still appears | delta psi in the initial stage of each sectors1|<|Δψs2If the current is too large, the flux linkage adjustment is failed, so that the stator current is distorted, and extra current harmonics are generated.
To illustrate this, as shown in FIG. 7, a simulation waveform of a conventional MC-DTC is shown when the PMSM rotor speed is controlled at 5 rad/s. In fig. 7, the stator flux linkage amplitude | ψ is shown in order from top to bottomsI, stator flux sector number hθAnd torque hysteresis comparator output CT. It can be seen from the figure that the phenomenon that the amplitude of the stator flux linkage is reduced and even maladjusted occurs has several reasons: 1) radial component u of the voltage vector3αWeak; 2) due to strong tangential component u of the voltage vector3βResulting in a shorter duration of the effective voltage vector; 3) due to reduced rate of change S of torquet -Smaller, resulting in longer action time of the zero voltage vector.
In order to solve the problem of the failure of stator flux amplitude adjustment during sector switching, an effective method is to control the flux linkage size by selecting a proper voltage vector, that is, selecting a reverse voltage vector, when the electromagnetic torque needs to be reduced. As shown in fig. 8, when the flux linkage error | ψerrAnd if the | is larger, selecting the effective voltage vector to replace the traditional zero vector.
The application provides a method for adopting a dynamic torque hysteresis comparator in direct torque control of a TLDMC-PMSM system, which is mainly reflected in flux linkage error | psierrWhen | is less than or equal to the set lower limit of flux linkage offset, a five-stage hysteresis comparator is selected, and when the flux linkage error | ψ iserrWhen | is greater than the set value, which causes the stator flux linkage adjustment failure, the torque hysteresis comparator is dynamically changed, namely, on the basis of the five-stage hysteresis comparator, the four-stage hysteresis comparator (2, 1, -1, -2) is dynamically changed through the selector, and the principle is as shown in fig. 9.
Based on the above analysis content, a process schematic diagram of the TLDMC-PMSM system control method is shown in fig. 2, and specifically includes:
s1, collecting three-phase input voltage u of three-level direct matrix converteria、uib、uicThree-phase output current ioA、ioB、ioCAnd rotor speed ω of the permanent magnet synchronous motorr；
S2, calculating an input current space vector and an output voltage space vector of the three-level direct matrix converter through the three-phase input voltage, the three-phase output current and the switching function matrix of the three-level direct matrix converter;
s3, observing the electromagnetic torque, the stator flux linkage and the input power factor average value of the permanent magnet synchronous motor through the input current space vector and the output voltage space vector of the three-level direct matrix converter to obtain an observed value;
and S4, obtaining a switching signal of the three-level direct matrix converter by adopting a direct torque control mode according to the observation value obtained in the step S3, and realizing system control.
The calculation formula of the output voltage space vector in step S2 is:
Figure BDA0001639360890000121
wherein u isoA、uoB、uoCSee formula (1).
The input current space vector is:
Figure BDA0001639360890000122
wherein iia、iib、iicSee formula (2).
In step S3, the space vector u is calculated from the obtained output voltage of the three-level direct matrix convertero(i.e. stator terminal voltage u)s) And output current space vector io(i.e. stator winding current i)s) For stator flux linkage psi under stationary alpha beta coordinate systemsObserving, i.e. calculating separately, the α -component ψ of the stator flux linkagesαStator flux linkage beta-axis component psisβAnd stator flux linkage amplitude | ψs|。
Is represented by the formula:
Figure BDA0001639360890000131
to obtain psisThe integral form expresses:
ψs＝∫(us-Rsis)dt+ψs|t＝0
this gives:
Figure BDA0001639360890000132
Figure BDA0001639360890000133
wherein R issRepresenting stator winding resistance, usα、usβAnd isα、isβRespectively representing stator terminal voltages usAnd stator winding current isAlpha and beta components in a stationary coordinate system.
Observing the electromagnetic torque of the permanent magnet synchronous motor comprises: calculating the electromagnetic torque T according to the obtained alpha and beta axis components of the stator flux linkagee：
Te＝1.5p(ψsαisβ-ψsβisα)；
Observing the input power factor average includes: calculating an input voltage space vector uiPhase difference with input current space vector
Figure BDA0001639360890000134
Obtaining an average value of input power factors
Figure BDA0001639360890000135
Step S4 includes:
s41, setting the stator flux linkage value | psisrefI and the observed stator flux linkage amplitudeValue | ψsError of | ψerrI passes through a first two-stage hysteresis comparator to obtain an output CψReference speed ω to be setrefAnd the actual rotor speed omega of the permanent magnet synchronous motorrObtaining a torque reference value T through a PI regulatorerefThen reference the torque to the value TerefWith the observed electromagnetic torque TeThe error of (A) is passed through a dynamic hysteresis comparator to obtain an output CTThe power factor set value 0 is averaged with the observed input power factor
Figure BDA0001639360890000136
The output is obtained through a second two-stage hysteresis comparator
Figure BDA0001639360890000137
S42, according to Cψ、CT、
Figure BDA0001639360890000138
And inquiring and selecting a proper voltage vector in a formulated direct torque control switch table to obtain a switching signal of the three-level direct matrix converter, thereby realizing system control.
In step S42, the voltage vectors in the direct torque control switch table are in one-to-one correspondence with the voltage vectors generated by the combination of the 12 bidirectional switches of the three-level direct matrix converter.
The dynamic hysteresis comparator comprises a five-stage hysteresis comparator and a four-stage hysteresis comparator which are connected in parallel, and the error phi is obtained according to the amplitude of the stator flux linkageerr| and a predetermined lower limit E of flux linkage offsetψSelects a suitable hysteresis comparator.
Examples
In this embodiment, the correctness and superiority of the method of the present invention are verified, and a matrix converter-permanent magnet synchronous motor (conventional MC-DTC) direct torque control method, a matrix converter-permanent magnet synchronous motor (improved MC-DTC) direct torque control method using a dynamic hysteresis loop, and a three-level direct matrix converter-permanent magnet synchronous motor (TLDMC-DTC) direct torque control method based on a dynamic hysteresis loop adopted in the present invention are respectively subjected to simulation comparison on a Matlab/Simulink software platform, and sampling periods are all 50 μ s. The specific simulation parameters are as follows:
parameters of the permanent magnet synchronous motor: rated power is 1.5kW, pole pair number p is 2, permanent magnet flux linkage psif0.42Wb, stator resistance R s1 omega, d-axis inductance LdIs 12mH, q-axis inductance LqAt 12mH, rated speed nN1500r/min, rated voltage UN110V, rated torque TNIs 10 N.m;
controlling parameters: hysteresis loop widths are respectively 1N · m and 0.5N · m, a stator flux linkage hysteresis loop width is 0.002Wb, a power factor hysteresis loop width is 0, a stator flux linkage reference value is 0.513Wb, and a flux linkage error Eψ0.006 Wb;
inputting filter parameters: filter capacitor CfIs 50 muF, and the filter inductance LfIs 2mH, damping resistance RfIs 8 omega.
Simulation comparison 1: and the light load runs at low speed. Setting the initial speed of the motor to be 5rad/s and the initial load torque to be 0 N.m; the load is suddenly added at 1s for 2 N.m; subsequently, the given speed was adjusted to 10rad/s at 2 s. The simulation waveforms are shown in FIGS. 10 to 13.
FIG. 10 is a simulated waveform of a conventional MC-DTC, and it can be seen that, under the low-speed operation of no load or light load, the duration of the zero vector is long, which causes the imbalance of the stator flux linkage, so that the current of the stator winding of the motor is distorted, and additional harmonic waves are generated; on the other hand, the pulsation of the electromagnetic torque is large, so that the rotation speed fluctuation of the motor is large, and the non-ideal performance of the traditional method under the condition of low-speed operation is reflected. FIG. 11 is a simulation waveform of an improved MC-DTC, and it can be seen from the diagram that the method has no flux linkage maladjustment phenomenon, the motor stator current has good sine degree, and the rotation speed fluctuation is small. However, the ripple in the electromagnetic torque and stator flux linkage amplitude is still large. Fig. 12 is a simulation waveform of TLDMC-DTC, and it is obvious from the diagram that the current waveform is smoother, the torque and stator flux ripple are greatly reduced, the dynamic response speed is fast, and the tracking is smooth.
FIG. 13 shows the simulation waveform amplification of the three control methods at a motor speed of 5rad/s and a load of 2 N.m within the simulation time of 1.56s to 1.60 s.
Simulation comparison 2: starting under rated load. The simulation waveforms of the permanent magnet synchronous motor are shown in fig. 14-16 when the speed is increased from zero to the rated speed under the rated load of 10 N.m.
It can be seen from fig. 14-16 that, under the method of the present invention, the permanent magnet synchronous motor can more smoothly rise from zero speed to rated speed, and the motor is accelerated by the maximum torque of 26N · m at the starting stage and stably operates at 10N · m when reaching the rated speed; the amplitude of the stator flux linkage is well controlled to be 0.513Wb, the current sine degree is good, and the harmonic wave is less.
In summary, the above description is only a preferred embodiment of the present invention, and is not intended to limit the scope of the present invention. Any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the protection scope of the present invention.

Claims (3)

1. A dynamic torque hysteresis based TLDMC-PMSM system control method is characterized in that a permanent magnet synchronous motor is controlled in the system through a three-level direct matrix converter, and the control method comprises the following steps:
s1, collecting three-phase input voltage and three-phase output current of the three-level direct matrix converter and the rotor speed of the permanent magnet synchronous motor,
s2, calculating the input current space vector and the output voltage space vector of the three-level direct matrix converter through the three-phase input voltage, the three-phase output current and the switch function matrix of the three-level direct matrix converter,
s3, observing the electromagnetic torque, the stator flux linkage and the input power factor average value of the permanent magnet synchronous motor through the input current space vector and the output voltage space vector of the three-level direct matrix converter to obtain an observed value,
s4, according to the observation value obtained in the step S3, a direct torque control mode is adopted to obtain a switching signal of the three-level direct matrix converter, and system control is achieved;
the calculation formula of the output voltage space vector in step S2 is:
Figure FDA0003036385460000011
wherein u isoA、uoB、uoCThe three-phase output voltage of the three-level direct matrix converter is represented by the following calculation formula:
Figure FDA0003036385460000012
wherein S (t) represents the trigger signal S of 12 bidirectional switches included in the circuit topology of the three-level direct matrix converterijThe corresponding switch function matrix, i belongs to { A, B, C }, j belongs to { a, B, C, n }, u }iRepresenting three-phase input voltage u of a three-level direct matrix converteria、uib、uicThree-phase input voltage space vector obtained by vector transformation,
the input current space vector is:
Figure FDA0003036385460000013
wherein iia、iib、iicThe three-phase input current of the three-level direct matrix converter is represented by the following calculation formula:
Figure FDA0003036385460000014
where T represents the transpose of the matrix, ioRepresenting three-phase output currents i of a three-level direct matrix converteroA、ioB、ioCThree-phase output current space vector, i, obtained by vector transformationnIs a neutral point current;
in step S3, the observing the stator flux linkage of the permanent magnet synchronous motor includes: separately counting statorsAlpha-axis component psi of flux linkagesαStator flux linkage beta-axis component psisβAnd stator flux linkage amplitude | ψs|：
ψsα＝∫(usα-Rsisα)dt
ψsβ＝∫(usβ-Rsisβ)dt
Figure FDA0003036385460000021
Wherein R issRepresenting stator winding resistance, usα、usβAnd isα、isβRespectively representing stator terminal voltages usAnd stator winding current isAlpha, beta components, u, in a stationary coordinate systemsOutput voltage space vector, i, equal to three-level direct matrix convertersEqual to the three-phase output current space vector obtained by vector conversion of the three-phase output current of the three-level direct matrix converter,
the observing the electromagnetic torque of the permanent magnet synchronous motor comprises: calculating electromagnetic torque Te：
Te＝1.5p(ψsαisβ-ψsβisα)
Wherein p is the number of pole pairs;
the observing the input power factor average value comprises: calculating an input voltage space vector uiPhase difference with input current space vector
Figure FDA0003036385460000022
Obtaining an average value of input power factors
Figure FDA0003036385460000023
Wherein u isiThree-phase input voltage of the three-level direct matrix converter is obtained through vector conversion;
the step S4 includes:
s41, setting the stator flux linkage value and the observed valueThe error of the stator flux linkage amplitude is output C through a first two-stage hysteresis comparatorψObtaining a torque reference value by a PI regulator according to the set reference rotating speed and the actual rotating speed of the rotor of the permanent magnet synchronous motor, and obtaining an output C by a dynamic hysteresis comparator according to the error between the torque reference value and the observed electromagnetic torqueTThe power factor set value 0 and the observed input power factor average value pass through a second two-stage hysteresis comparator to obtain output
Figure FDA0003036385460000024
S42, according to Cψ、CT、
Figure FDA0003036385460000025
And inquiring and selecting a proper voltage vector in a formulated direct torque control switch table to obtain a switching signal of the three-level direct matrix converter, thereby realizing system control.
2. The dynamic torque hysteresis based TLDMC-PMSM system control method according to claim 1, wherein the dynamic hysteresis comparator comprises a plurality of different hysteresis comparators connected in parallel, and a suitable hysteresis comparator is selected according to the error of comparing the stator flux linkage amplitude with the set lower limit of flux linkage misalignment.
3. The dynamic torque hysteresis based TLDMC-PMSM system control method of claim 2, wherein the dynamic hysteresis comparator comprises a five stage hysteresis comparator and a four stage hysteresis comparator in parallel.
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Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101863413A (en) * 2010-06-25 2010-10-20 天津城市建设学院 Energy-saving control system for permanent magnet synchronous escalator
CN103828213A (en) * 2011-09-26 2014-05-28 大金工业株式会社 Power converter control method
JP2016220324A (en) * 2015-05-15 2016-12-22 株式会社安川電機 Matrix converter, power generating system, control device, and control method
CN106911277A (en) * 2017-04-07 2017-06-30 哈尔滨理工大学 Control system for permanent-magnet synchronous motor based on matrix converter
CN106953570A (en) * 2017-04-25 2017-07-14 南京福致通电气自动化有限公司 Energy feedback type elevator traction drive system control method based on matrix converter
CN107528478A (en) * 2017-09-26 2017-12-29 上海电力学院 A kind of SVPWAM modulator approaches based on three level direct matrix transform devices
CN107689760A (en) * 2017-11-02 2018-02-13 哈尔滨理工大学 Based on the magneto of matrix converter without position vector control system and method

Patent Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101863413A (en) * 2010-06-25 2010-10-20 天津城市建设学院 Energy-saving control system for permanent magnet synchronous escalator
CN103828213A (en) * 2011-09-26 2014-05-28 大金工业株式会社 Power converter control method
JP2016220324A (en) * 2015-05-15 2016-12-22 株式会社安川電機 Matrix converter, power generating system, control device, and control method
CN106911277A (en) * 2017-04-07 2017-06-30 哈尔滨理工大学 Control system for permanent-magnet synchronous motor based on matrix converter
CN106953570A (en) * 2017-04-25 2017-07-14 南京福致通电气自动化有限公司 Energy feedback type elevator traction drive system control method based on matrix converter
CN107528478A (en) * 2017-09-26 2017-12-29 上海电力学院 A kind of SVPWAM modulator approaches based on three level direct matrix transform devices
CN107689760A (en) * 2017-11-02 2018-02-13 哈尔滨理工大学 Based on the magneto of matrix converter without position vector control system and method

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
"矩阵变换器及其在调速***中的应用研究";赖文焯;《中国优秀硕士学位论文全文数据库·工程科技Ⅱ辑》;20130110;全文 *

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