CN102347695B - High efficient series resonance converter - Google Patents
High efficient series resonance converter Download PDFInfo
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- CN102347695B CN102347695B CN201110030999.0A CN201110030999A CN102347695B CN 102347695 B CN102347695 B CN 102347695B CN 201110030999 A CN201110030999 A CN 201110030999A CN 102347695 B CN102347695 B CN 102347695B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
Abstract
The invention provides an efficient series resonant converter capable of reducing switch loss and improving efficiency by arranging a separate capacitor in a secondary winding of a transformer for increasing resonance current of a primary winding of the transformer rapidly when a switch unit is switched on. The series resonant converter is capable of reducing the switch loss by satisfying the condition of zero voltage and zero current switch when the switching unit is switched on and the condition of zero voltage switch when the switching unit is switched off. Moreover, the efficiency of the series resonant coverter can be improved through rapid increase of the resonance current during charging of the secondary capacitor on the secondary winding.
Description
Technical field
The present invention relates to series resonant converter, relate in particular to the series resonant converter that the efficiency of the switching loss that reduces and increase is provided by improving resonance current waveform.
Background technology
DC-to-DC converter is for the electronic circuit from a voltage level conversion to another voltage level by direct current (DC) source.Conventionally, DC-to-DC converter converts direct voltage to interchange (AC) voltage, is raise or is reduced this alternating voltage, and will convert direct voltage to through the alternating voltage raising or reduce by transformer.
Series resonant converter (SRC) is an example of DC-to-DC converter.
Fig. 1 is the circuit diagram of conventional SRC.This SRC is used the resonance being generated by inductor Lr and capacitor Cr, and presents good conversion efficiency.
With reference to figure 1, this SRC comprises switch element 20, LC resonant circuit 30, transformer TX, bridge rectifier 40 and gate driver 51.Switch element 20 comprises that a plurality of switch S 1-S4 are to change over alternating voltage by alternation direct voltage by the direct voltage from input voltage source 10.This LC resonant circuit 30 is connected to switch element 20, and comprises resonant inductor Lr and the resonant capacitor Cr that is one another in series and connects.The resonance that LC resonant circuit 30 use are generated by resonant inductor Lr and resonant capacitor Cr changes the frequency characteristic from the alternating voltage of switch element 20.Transformer TX is by primary voltage, and the alternating voltage from LC resonant circuit 30 converts secondary voltage to.Bridge rectifier 40 converts secondary alternating voltage to direct voltage.Gate driver 51 control switch unit 20 are with amplitude and the shape of control load electric current.
SRC also comprises to the direct voltage from bridge rectifier 40 is carried out filtering and the direct voltage through filtering is applied to the capacitor C of load 60
0.
SRC is Full-bridge pulse width (PWM) transducer that uses four the semiconductor switch S1-S4 such as insulated gate bipolar transistor (IGBT) or mos field effect transistor (MOSFET) with full bridge structure interconnection to realize.Switch S 1-S4 is connected in parallel to anti-paralleled diode D1-D4, and in parallel with buffer condenser CS1-CS4.
Switch element 20 is by synchronously connecting or turn-off pair of switches S1 and S4 or another pair of switches S2 and S3 by converting direct-current voltage into alternating-current voltage under the control at gate driver 51.Alternating voltage is transferred to the secondary winding TX2 of transformer TX by LC resonant circuit 30.
LC resonant circuit 30 comprises resonant inductor Lr and resonant capacitor Cr, and they are connected in series to the armature winding TX1 of transformer TX between the contact node of switch S 1 and S2 and another contact node of switch S 3 and S4.This LC resonant circuit 30 is stored energy and output energy in resonant inductor Lr and resonant capacitor Cr.
This transformer TX is the energy from LC resonant circuit 30 by secondary winding TX2 output.Induced potential in secondary winding TX2 is pressed the ratio of the quantity of the number of turn in secondary winding TX2 and the quantity of the number of turn in armature winding TX1 and is determined.
The bridge rectifier 40 that comprises four rectifier diode RD1-RD4 will convert direct voltage to from the alternating voltage of inducting of secondary winding TX2 output.This direct voltage is by capacitor C
0carry out filtering, and then output to load 60.
This gate driver 51 turns on and off switch S 1-S4 during the driving of SRC and power conversion process.Gate driver 51 received pulse voltage signals are as input and generate to drive signal, gating signal so that switch S 1-S4 turn on and off.
During the switching manipulation of semiconductor switch S1-S4, with the predetermined delay in each switch and gradient, change voltage and current.Therefore,, when switch S 1-S4 turns on and off, can exist voltage and current can synchronously be applied to the section of switch, the i.e. section of voltage and current part crossover.In this section, can there is the switching loss corresponding to the product V * I of voltage and current.
For example, when IGBT ends, at IGBT two ends, apply tail current after voltage completely and can continue to flow, thereby cause serious switching loss.
Switching loss reduces the efficiency of transducer and causes switch heating.In addition, the switching frequency of switching loss and switch increases pro rata, thus the maximum switching frequency of limit switch.
In order to reduce switching loss, the various switching mechanisms such as zero voltage switch (ZVS), Zero Current Switch (ZCS) and zero-voltage zero-current (ZVZCS) have been proposed.
In order to realize ZVS, ZCS and ZVZCS to reduce switching loss, as shown in Figure 1, provide load resonant transducer, it uses LC resonance by inductor Lr and capacitor Cr being connected to the armature winding TX1 of transformer TX.LC resonance can allow transducer to generate the voltage and current waveform that meets no-voltage and zero current condition.In addition, LC resonant circuit can allow load voltage and load current vibration, realizes thus ZVS, ZVC or ZVZCS.
Fig. 2 is the resonance current i that describes to depend on the switching frequency fs in load resonant transducer
lthe curve chart of characteristic.The switching manipulation of load resonant transducer depends on that switching frequency can be divided into two patterns, i.e. discontinuous conduction mode (DCM) and continuously conduction mode (CCM).In DCM, in the switching frequency fs section lower than the resonance frequency fr of inductor and capacitor, carry out switching manipulation.In CCM, in the switching frequency fs section higher than resonance frequency fr, carry out switching manipulation.
Fig. 3 a and 3b describe exemplary voltages that the LC resonance in conventional load resonance converter causes and the curve chart (for convenience's sake, half-bridge structure being shown) of current waveform.Fig. 3 a illustrates the harmonic wave in DCM, and Fig. 3 b illustrates the harmonic wave in CCM.Current i
lindication inductor current and voltage v
cindication condenser voltage.
With reference to figure 3a, the switching frequency fs in DCM than the low one-period of resonance frequency fr during, when the pair of switches of full bridge structure is connected to allow current flowing, electric current gathers in inductor Lr.The electric energy gathering is passed to capacitor Cr, increases thus condenser voltage v
c.Accumulate in after electric energy in inductor Lr dissipates completely, the polarity upset of capacitor, thus cause current i
lflow in the other direction.Then pair of switches is turn-offed.Therefore, there is wherein not having the discontinuous segment of current flowing.
In this discontinuous segment, when connecting, another pair of switches applies in the reverse direction voltage v
cand current i
l.Correspondingly, because there is the wherein discontinuous mobile discontinuous segment of electric current, so can carry out zero current turn on-switch.
With reference to figure 3b, the switching frequency fs in CCM than the high one-period of resonance frequency fr during, inductor current i when pair of switches is connected
lincrease, and condenser voltage v then
cinductor current i during increase
lreduce.When pair of switches is turn-offed, electric current is no longer mobile.When another pair of switches is connected (, zero current turn on-switch), to apply voltage in the other direction.As a result, when starting to decline, voltage applies in the reverse direction electric current.Like this, completed a switch periods.That is, electric current continuous flow during one-period.
In these two patterns, by pulse frequency modulated (PFM), control output current.In DCM, higher frequency causes the increase of output current.In CCM, lower frequency causes the increase of resonance current, increases thus the load current of exporting after full-wave rectification.
In CCM, as the resonance current i when giving resonance current by the waveform that approaches square wave of turn on-switch under same frequency operating condition
lwhile sharply increasing, this resonance current can have the effective value of increase.As a result, transducer can present the efficiency of improvement.
Therefore, need for increasing fast resonance current i at CCM
lto obtain the technology of the transducer with improved efficiency.
Summary of the invention
The present invention is contemplated to the problem that solves above-described prior art, and an aspect of of the present present invention provides the series resonant converter that can reduce switching loss and improve efficiency by improving the resonance current waveform in switching manipulation in continuous conduction mode.
According to an aspect of the present invention, series resonant converter comprises: comprise the switch element of a plurality of switches, for direct current (DC) voltage transitions being become to exchange (AC) voltage by alternation direct voltage; LC resonant circuit, it comprises resonant inductor and the resonant capacitor being connected in series, and is connected to switch element, and uses the resonance being generated by resonant inductor and resonant capacitor to come converting transmission from the frequency characteristic of the alternating voltage of switch element; The transformer that comprises armature winding and secondary winding, this armature winding is connected to LC resonant circuit, and in the induced potential in secondary winding and secondary winding in the quantity of the number of turn and armature winding the ratio of the quantity of the number of turn proportional; Secondary capacitor, its be connected to transformer secondary winding and and transformers connected in parallel; The full-bridge rectifier that comprises a plurality of rectifier diodes, for converting the alternating voltage of inducting at secondary winding to direct voltage; And gate driver, its detection is connected to the conducting of the anti-paralleled diode of switch, and exports conducting gating signal with turn on-switch when anti-paralleled diode conducting.
Secondary capacitor can have the electric capacity less than resonant capacitor.
Secondary capacitor can have the impedance lower than the circuit being comprised of rectifier diode and load, and the secondary capacitor that is used in the load current charging of inducting in secondary winding can cause the resonance current of LC resonant circuit to increase fast.
Gate driver can comprise: be connected to the first resistor of the input of its input pulse voltage signal; Be parallel-connected to the first resistor of input end and use the capacitor of the conducting pulse voltage charging applying by input; The semiconductor switch that comprises source electrode, grid and drain electrode, source electrode, grid and drain electrode are connected respectively to the output node of input, capacitor and are connected to the gate node of output of the switch of switch element; The 3rd resistor connecting between the drain electrode of semiconductor switch and the gate node of output; And be connected to input to form the 4th resistor of conducting path by the first resistor and capacitor.
This second resistor can have than the large resistance of the 4th resistor.
Accompanying drawing summary
According to the following description of the exemplary embodiment providing in conjunction with appended accompanying drawing, above and other aspect of the present invention, feature and advantage will become apparent, wherein:
Fig. 1 is the circuit diagram of conventional series resonant converter;
Fig. 2 is the curve chart of describing to depend on the resonance current characteristic of the switching frequency in load resonant transducer;
Fig. 3 a and 3b describe exemplary voltages that the LC resonance in conventional load resonance converter causes and the curve chart of current waveform;
Fig. 4 is the circuit diagram of the series resonant converter of one exemplary embodiment according to the present invention;
Fig. 5 is the curve chart of describing the voltage and current waveform of the continuous conduction mode in the series resonant converter of one exemplary embodiment according to the present invention;
Fig. 6 is the comparative curve chart of the voltage and current waveform of the continuous conduction mode in depicted example series resonant converter and conventional series resonant converter;
Fig. 7-14th, illustrates the circuit diagram of the operation of each pattern in the series resonant converter of one exemplary embodiment according to the present invention;
Figure 15 is the circuit diagram of the gate driver of the series resonant converter of one exemplary embodiment according to the present invention;
Figure 16 illustrates as the pulse voltage signal to the input signal of the gate driver shown in Figure 15;
Figure 17 illustrates the gating signal as the output signal from the gate driver shown in Figure 15;
Figure 18 is the resonance current of depicted example series resonant converter and the gating signal curve chart with respect to the time; And
Figure 19-22nd, illustrates the circuit diagram of the operation of each pattern in the gate driver of one exemplary embodiment according to the present invention.
Embodiment
Referring now to accompanying drawing, the exemplary embodiment of invention is specifically described.
The present invention provides and can reduce the series resonant converter that switching loss is improved efficiency simultaneously by improving resonance current waveform.More specifically, the resonance current waveform that the series resonant converter of one exemplary embodiment can improve by being added into the capacitor of the secondary winding of transformer in the switching manipulation of continuous conduction mode according to the present invention reduces switching loss and improves efficiency.
Fig. 4 is the circuit diagram of the series resonant converter (SRC) of one exemplary embodiment according to the present invention.
With reference to figure 4, this SRC comprises switch element 20, LC resonant circuit 30, transformer TX, capacitor C2, bridge rectifier 40 and gate driver 51.Switch element 20 comprises a plurality of switch S 1-S4, and the plurality of switch S 1-S4 is for becoming to exchange (AC) voltage by alternation direct voltage by the direct current from input voltage source 10 (DC) voltage transitions.LC resonant circuit 30 uses LC resonance to convert the frequency characteristic from the alternating voltage of switch element 20.Transformer TX is by primary voltage, and the alternating voltage from LC resonant circuit 30 converts secondary voltage to.Capacitor C2 is connected to the secondary winding TX2 of transformer TX, and in parallel with transformer TX.Bridge rectifier 40 converts the alternating voltage of inducting in the secondary winding TX2 at transformer TX to direct voltage.Gate driver 51 control switch unit 20 are with amplitude and the shape of control load electric current.
The structure that this SRC has is similar to the structure of conventional SRC.That is, switch element 20 comprises four semiconductor switch S1-S4 such as IGBT or MOSFET, and it connects with full bridge structure; Anti-paralleled diode D1-D4 is parallel-connected to switch S 1-S4; And buffer condenser CS1-CS4 is parallel-connected to diode D1-D4.
In addition, to be similar to the structure of conventional SRC structure, realize this SRC.; pair of switches S1 in switch element 20 and S4 or another pair of switches S2 and S3 are lower synchronous the connection or shutoff of driving signal (gating signal) of gate driver 51; with by converting direct-current voltage into alternating-current voltage, and this alternating voltage is transferred to the armature winding TX1 of transformer TX by LC resonant circuit 30; LC resonant circuit 30 comprises resonant inductor Lr and resonant capacitor Cr, and the armature winding TX1 that they are connected in series to transformer TX between the contact node of switch S 1 and S2 and another contact node of switch S 3 and S4 is to reduce switching loss; Transformer TX converts primary voltage to the secondary voltage at the secondary winding TX2 two ends of transformer TX; The bridge rectifier 40 that comprises rectifier diode RD1-RD4 converts the alternating voltage of inducting at secondary winding TX2 two ends to direct voltage; And this direct voltage through rectification is by capacitor C
0carry out filtering, then output to load 60.
But in this embodiment, SRC also comprises another capacitor C2, and this capacitor C2 is parallel-connected to transformer TX on secondary winding TX2.This secondary capacitor C2 allows SRC to produce the resonance current waveform improving, and reduces thus the efficiency that switching loss is improved SRC simultaneously.
For the operation of each pattern of the following stated, secondary capacitor C2 has the electric capacity less than resonant capacitor Cr.For example, secondary capacitor C2 can have 1/20～1/5 electric capacity (for example 0.3 μ F) of the electric capacity (for example 3 μ F) that is resonant capacitor Cr.
The secondary capacitor C2 adding can make SCR have the low impedance of combination than rectifier diode RD1-RD4 and load 60.
Operation in each pattern of the SRC that comprises secondary capacitor C2 will be described with reference to the drawings.
Fig. 5 is voltage Vc and the electric current I of describing the continuous conduction mode (CCM) in the SRC of one exemplary embodiment according to the present invention
lthe curve chart of waveform.Fig. 6 is voltage Vc and the electric current I of the CCM in depicted example SRC and conventional SRC
lthe comparative curve chart of waveform.
With reference to figure 5, I
lrepresent resonant inductor current, V
crepresent resonant capacitor voltage, the moment that the pair of switches S1 in the initial point indication full-bridge switch S1-S4 of pattern 1 and S4 connect, t
2the moment that indication pair of switches S1 and S4 turn-off, t
4the moment of indicating another pair of switches S2 and S3 to connect, and t
7the moment that indication pair of switches S2 and S3 turn-off.
Fig. 7-14th, illustrates the circuit diagram of the operation of SRC in each pattern.
pattern 1: turn on-switch S1 and S4 and to secondary capacitor charging (with reference to the pattern 1 of figure 5
and Fig. 7)
When switch S 1 and S4 connection, from the resonance current of input voltage source 10, flow through armature winding TX1, resonant capacitor Cr and the switch S 4 of switch S 1, resonant inductor Lr, transformer TX.Resonance current is inductor current I
las shown in pattern 1, flow, simultaneously condenser voltage V
craise.Load current the secondary winding TX2 inducting from armature winding TX1 is to having the secondary capacitor C2 charging that is parallel-connected to transformer TX of low electric capacity.The interpolation of capacitor C2 makes SRC have the low impedance of combination than rectifier diode RD1-RD4 and load 60.In addition, to make resonance current be induced current I to secondary capacitor C2
lincrease fast, as shown in Figure 5 and Figure 6.
pattern 2: apply electric current (with reference to pattern 2 and Fig. 8 of figure 5) to load
When secondary capacitor C2 charges completely with the load current on secondary winding TX2 and becomes while equaling the voltage at secondary winding TX2 two ends in pattern 1, load current flows to load 60 by rectifier diode RD1 and RD4.
mode 3: switch S 1 and S4 turn-off and electric current afterflow (with reference to mode 3 and Fig. 9 of figure 5)
When switch S 1 and S4 in pattern 2 when removing gating signal and turn-off, owing to buffer condenser CS1 and CS4 switch S 1 and the S4 voltage rising at two ends separately, and the buffer condenser CS2 having charged in preceding mode and the CS3 electric discharge that are connected in parallel with switch S 2 and S3.Buffer condenser CS1 and CS4 are with passing through the mobile inductor current I of resonant inductor Lr
lcharging, and the inductor current of afterflow simultaneously I
l.In mode 3, buffer condenser CS2 and CS3 electric discharge, and simultaneous buffering capacitor CS1 and CS4 charging.When the voltage at buffer condenser CS1 and CS4 two ends becomes while equaling voltage source V dc, buffer condenser CS2 and CS3 discharge completely, thereby cause switch S 1 and S4 to turn-off.The energy gathering in resonant inductor Lr and resonant capacitor Cr allows load current to continue to flow to load 60.
pattern 4: the conducting of anti-paralleled diode D2 and D3 (with reference to pattern 4 and Figure 10 of figure 5)
Because switch S 1 and S4 turn-off in mode 3, so flow through the freewheel current of resonant inductor Lr, flow through anti-paralleled diode D2 and the D3 being connected with S3 with switch S 2, and therefore by voltage source, sharply reduce.The voltage at switch S 2 and S3 two ends approaches zero.Correspondingly, wherein in the no-voltage section of electric current by anti-paralleled diode D2 and D3 afterflow, wherein anti-paralleled diode D2 and D3 connect or the no-voltage section of positive bias in conducting gating signal apply the zero voltage switch that allows switch S 2 and S3.
pattern 5: the no-voltage of switch S 2 and S3 and the turn on-switch of zero current are (with reference to the mould of figure 5
formula 5 and Figure 11)
The freewheel current that flows through anti-paralleled diode D2 and D3 in mode 3 is sharply decreased to zero by voltage source.Then, voltage source allows electric current to flow in the reverse direction, and by switch S 2 and S3, flows to the resonance current I of resonant inductor Lr
lflow in the reverse direction.The secondary capacitor C2 repid discharge of charging in pattern 1 and pattern 2.By with pattern 1 in the direction of opposite current on flow through load current on the secondary winding TX2 that the primary current of armature winding TX1 inducts with pattern 1 in the direction of opposite current on flow through secondary capacitor C2.Correspondingly, secondary capacitor C2 is charged again by load current, thereby cause resonance current, is inductor current I
lincrease fast, as shown in Figure 5 and Figure 6.That is, when anti-paralleled diode D2 and D3 conducting, (, the voltage of switch ends be zero) applies conducting gating signal, and when the reversing of electric current, with zero current condition switch switch S 2 and S3.Its result is, realizes no-voltage and Zero Current Switch condition, minimizes thus switching loss.
pattern 6: apply electric current (with reference to pattern 6 and Figure 12 of figure 5) to load
Except when switch S 2 and S3 inductor current I while connecting
lflow through in the reverse direction outside resonant inductor Lr and resonant capacitor Cr, pattern 6 is identical with pattern 2.When secondary capacitor C2 equals the secondary voltage on secondary winding TX2 by the load current charging on secondary winding TX2 and voltage completely, load current flows to load 60 by rectifier diode RD2 and RD3.
mode 7: switch S 2 and S3 turn-off and electric current afterflow (with reference to mode 7 and Figure 13 of figure 5)
Except inductor current I
lflow through in the reverse direction outside resonant inductor Lr and resonant capacitor Cr and switch S 2 and S3 shutoff, mode 7 is identical with mode 3.Particularly, in mode 7, the voltage at switch S 2 and S3 two ends slowly increases by buffer condenser CS2 and CS3, and the buffer condenser CS1 having charged in preceding mode and CS4 electric discharge.At this moment, the inductor current that flows through resonant inductor Lr is to buffer condenser CS2 and CS3 charging and by afterflow.When the voltage at buffer condenser CS2 and CS3 two ends becomes while equaling voltage source V dc, buffer condenser CS1 and CS4 discharge completely, and switch S 2 and S3 turn-off.The energy gathering in resonant inductor Lr and resonant capacitor Cr allows load current to continue to flow to load 60.
pattern 8: the conducting of anti-paralleled diode D1 and D4 (with reference to pattern 8 and Figure 14 of figure 5)
Except inductor current I
lflow through in the reverse direction resonant inductor Lr and resonant capacitor Cr and be connected to outside the anti-paralleled diode D1 and D4 conducting of switch S 1 and S4, pattern 8 is identical with pattern 4.
Eight patterns have been described during a switch periods.After pattern 8, inductor current I
lreversing, and repeat that wherein switch S 1 and S4 are connected and the pattern 1 (with reference to figure 7) of secondary capacitor C2 charging.After pattern 1, also repeat pattern 2-pattern 8.
After pattern 8 during recovery pattern 1, when anti-paralleled diode D1 and D4 conducting, (, when the voltage at switch S 1 and S4 two ends is zero) applies conducting gating signal.When the reversing of electric current, with zero current condition diverter switch S1 and S4.Therefore, meet no-voltage and Zero Current Switch condition, minimize thus switching loss.
Thus, because the secondary winding TX2 to transformer TX has added the secondary capacitor C2 less than the electric capacity of the resonant capacitor Cr in LC resonant circuit unit 30, thus in initial level electric current flow through secondary capacitor C2 and unsupported 60 and armature winding TX1 on resonance current I
lincrease fast, thereby produce trapezoidal current waveform, as shown in Figure 5.Correspondingly, as shown in Figure 6, under same frequency, this resonance current I
lcan have than thering is the larger effective value of sine-shaped conventional resonance current.
In other words, as shown in Figure 6, because SRC allows resonance current to increase quickly than conventional SRC, so the effective value of resonance current increases the amount (' A+C-B ') of shaded area under same switch frequency, improve thus efficiency.
It is the electric capacity of 10 times of conventional condenser capacitance that the inductor energy of area increased ' C ' allows buffer condenser CS1-CS4 to have, and realizes thus no-voltage turn-off criterion.
Correspondingly, owing to can obtain larger effective current value under same frequency, therefore likely improve the efficiency of this SRC.In addition, likely by being reduced in the resonance current I under same load current conditions
lmaximum reduce conduction loss.
In addition the inductor current I increasing fast by secondary capacitor C2,
lthe energy that allows to be stored in inductor Lr increases.Therefore, likely enlarge markedly the electric capacity of the buffer condenser CS1-CS4 that is connected to switch S 1-S4 two ends, the switching loss while reducing thus switch S 1-S4 shutoff.
Correspondingly, carry out for when switch S 1-S4 connects by switch S 1-S4 separately the voltage at two ends remain on zero zero voltage switch, reduce thus switching loss.
In conventional SRC because under no-voltage and zero current condition turn on-switch S1-S4, and switch S 1-S4 does not turn-off under no-voltage and zero current condition, so there is switching loss.But in exemplary SRC, switch S 1-S4 turn-offs and connects under zero voltage condition.
On the other hand, when anti-paralleled diode D1-D4 conducting, be not easy exactly turn on-switch S1-S4 so that can meet the zero current of switch S 1-S4 and the condition of no-voltage.
In the application with wide loading range because shutoff separately of switch S 1-S4 and the cycle between diode D1-D4 conducting separately depend on loading condition and change, so be difficult to estimate when to apply conducting gating signal.
In one embodiment, when the voltage approaches zero at switch S 1-S4 two ends, gate driver automatically applies conducting gating signal when diode D1-D4 conducting.
Gate driver allows in wide opereating specification, to carry out no-voltage and zero current turn on-switch, and is also provided for the dead time compensation of stable operation.
Figure 15 is the circuit diagram of the gate driver in the SRC of one exemplary embodiment according to the present invention.
With reference to Figure 15, gate driver 51 comprises the first resistor R11, capacitor C11, semiconductor switch, the second resistor R12, the 3rd resistor R13 and the 4th resistor R14.The first resistor R11 is connected to input 52, and pulse voltage signal inputs to this input 52.Capacitor C11 is parallel-connected to the first resistor R11 at input 52 places, and uses the conducting pulse voltage applying by input 52 and 53 to charge.This semiconductor switch comprises source electrode, grid and drain electrode, the gate node 55 of output that this source electrode, grid and drain electrode are connected respectively to the output node of input 51, capacitor C11 and are connected to the switch S 1-S4 of switch element 20.The second resistor R12 is connected between the first resistor R11 and the collector node 54 of output.The 3rd resistor R13 is connected between the drain electrode of semiconductor switch and the gate node of output 55.The 4th resistor R14 is connected to input 52 to form conducting path by the first resistor R11 and capacitor C11.
Gate driver 51 also comprises the first diode D11, the second diode D12 and the 6th resistor R16.This first diode D11 is connected in series to the second resistor R12 between the second resistor R12 and the collector node 54 of output.The second diode D12 is placed in the subcircuits between input 53 and output 56.The 6th resistor R16 is connected to the 5th resistor R15 and the 3rd resistor R13 to form conducting path between them.
In the circuit of configuration thus, as input signal, to input 52 and 53, apply there is (+) and (-) polarity pulse voltage signal with generation gating signal.Figure 16 illustrates the example of the pulse voltage signal applying by input.
Gate node 55 on the output of gate driver 51, collector node 54 and emitter node 56 are connected respectively to grid, the collector and emitter of switch S 1-S4.
Semiconductor switch can be used mos field effect transistor (MOSFET) to realize.When semiconductor applies positive conducting pulse voltage by the input 52 and 53 to gate driver 51 and when capacitor C11 is charged to connect, this conducting gating signal provides the switch S 1-S4 with turn on-switch unit 20 (being hereinafter called main switch) by the gate node 55 of the 3rd resistor R13 and output.
On the other hand, when the input 52 and 53 to gate driver 51 applies negative conducting pulse voltage, MOSFET turn-offs, and wherein electric current flows through its built-in body diode, thereby main switch S1-S4 is turn-offed.
Gate driver 51 detects the conducting (that is, the zero-voltage state of main switch) of the anti-paralleled diode D1-D4 that is connected to main switch S1-S4, and when anti-paralleled diode D1-D4 conducting, provides conducting gating signal to connect main switch S1-S4.Therefore, realize the no-voltage turn on-switch of main switch S1-S4.
In addition, the second resistor R12 and the 4th resistor R14 are connected in series to the first resistor R11, so that flow through the electric current of the first resistor R11, can flow through in the second resistor R12 and the 4th resistor R14.This second resistor R12 can have the large resistance than the 4th resistor R14.
As described hereinafter, while realizing the zero voltage condition of main switch S1-S4 before the maximum dead time being set by capacitor C11 (, when being connected to the anti-paralleled diode D1-D4 conducting of main switch S1-S4), the electric current that flows through the 4th resistor R14 can flow through resistor R12, thereby connects MOSFET.
To the operation of gate driver in each pattern be described.
With reference to Figure 15, the input of gate driver 15 connects to apply pulse voltage by transformer TX11.When the armature winding to transformer TX11 applies as the pulse voltage signal of input signal when controlling main switch S1-S4, the secondary pulses voltage signal of inducting from armature winding on secondary winding causes gate driver 51 to produce signals so that main switch S1-S4 turns on and off.
In each operator scheme, the input 52 and 53 that applies (+) and (-) voltage will be called pin _ 1 (pin _ 1) and pin _ 2 (pin _ 2) in the situation that not mentioning transformer.
Figure 17 illustrates the gating signal of exporting from the gate driver shown in Figure 15 when the pulse voltage signal shown in Figure 16 is used as to input signal.Figure 18 describes the curve chart of the gating signal of resonance current and SRC with respect to the time.Figure 19-22nd, the circuit diagram of the operation of the gate driver that one exemplary embodiment according to the present invention is shown in each pattern.
(+) gate voltage applies pattern 1: the applying of positive pulse voltage (with reference to Figure 19)
When the input to gate driver 51 applies positive gate voltage, as shown in figure 16, pin _ 1 52 are positive terminals, and pin _ 2 53 are negative pole ends.Therefore, electric current flows through capacitor C11, the first resistor R11 and the 4th resistor R14 (high resistance) from pin _ 1 52, and capacitor C11 is recharged.When capacitor C11 continues to be charged to the conducting voltage of P channel mosfet, MOSFET conducting and electric current flow to the 6th resistor R16 by MOSFET and the 3rd resistor R13, thereby connect main switch S1-S4 (with reference to Figure 21).This operator scheme is called maximum dead time pattern.When gate driver 51 is configured to be connected to the anti-paralleled diode D1-D4 conducting of main switch S1-S4 before the maximum dead time in the following stated (, when realizing the zero voltage condition of main switch), the automatic conducting of MOSFET, connects main switch S1-S4 thus.
(+) gate voltage applies pattern 2: the no-voltage of no-voltage detecting pattern and main switch is connected and opened
close (with reference to Figure 20 and 21)
Anti-paralleled diode D1-D4 conducting before the maximum dead time when capacitor C11 is charged to conducting MOSFET in pattern 1, and during the voltage approaches zero between the collector and emitter of main switch S1-S4, electric current flows through capacitor C11, resistor R11 and resistor R12 (low resistance), as shown in figure 20.At this constantly, be different from maximum dead time pattern, because current flows through resistor R12 and capacitor C11 are by quick charge, connect thus MOSFET.As a result, with reference to Figure 21, along with MOSFET conducting, electric current flows through the 3rd resistor R13 and the 6th resistor R16, thereby connects main switch S1-S4.In other words, when before the maximum dead time that anti-paralleled diode D1-D4 is being scheduled to, the voltage at conducting and main switch S1-S4 two ends equals zero, main switch S1-S4 has nothing to do and automatically connects in predetermined maximum dead time.Correspondingly, because gate driver 51 switch element 20 to transducer when anti-paralleled diode D1-D4 conducting provides conducting gating signal, so realize the no-voltage turn on-switch (with reference to the pattern 5 of transducer) of main switch S1-S4 in switch element 20.The on-delay that the step indication that generates gating signal when pulse is risen shown in Figure 17 is caused by the capacitor C11 through charging.
(-) gate voltage applies pattern 1: shutdown mode
The shutoff of this pattern indication main switch S1-S4.When the negative grid voltage shown in Figure 16 is applied to the input of gate driver 51, pin _ 1 52 and pin _ 2 53 are respectively negative pole end and positive terminal.Correspondingly, from the electric current of pin _ 2 53, in the situation that not having to postpone, flow through the body diode of the 6th resistor R16 and MOSFET, thereby turn-off main switch S1-S4.
Therefore,, when the voltage approaches zero at main switch S1-S4 two ends and diode D1-D4 conducting, gate driver 51 detects the conducting of diode D1-D4 and automatically applies grid Continuity signal.As a result, likely realize the zero voltage switch of the main switch S1-S4 in the SRC shown in Fig. 4.
Like this, because the secondary winding to transformer has added the little secondary capacitor of electric capacity than the resonant capacitor in LC resonant circuit unit, so in initial level, electric current flows through secondary capacitor and resonance current on unsupported and armature winding increases fast, thereby generates trapezoidal current waveform.Therefore,, under same frequency, this resonance current can have than the larger effective value of conventional resonance current with sinusoidal waveforms.
Therefore,, owing to the effective current increasing under same switch frequency, this SRC can present the efficiency higher than conventional SRC.In addition, it is the electric capacity of 10 times of conventional condenser capacitance that the inductor energy of increase allows buffer condenser to have, and realizes thus no-voltage turn-off criterion.
In addition, the effective current value of the increase under same frequency allows SRC to present the efficiency of increase.Under same load current conditions, maximum resonance electric current can be lowered, and reduces thus conduction loss.
In addition, the electric current increasing fast by secondary capacitor increases the energy of storing in inductor.Therefore, likely enlarge markedly the electric capacity of the buffer condenser that is connected to switch, thus the switching loss while reducing switch shutoff.
Correspondingly, when switch turn-offs, likely realize and allow the voltage of each switch ends to remain on zero zero voltage switch.As a result, likely reduce switching loss.
In other words, in conventional SRC, because switch S 1-S4 connects under no-voltage and zero current bar line, and switch S 1-S4 does not turn-off under no-voltage and zero current condition, and therefore switching loss occurs when switch turn-offs.But in exemplary SRC, switch S 1-S4 carries out and turn-offs and connect under zero voltage condition.
In addition, gate driver is simple to operate.That is,, in order to realize zero current and the no-voltage on-condition of switch, when switch ends voltage approaches zero, gate driver detects the conducting of anti-paralleled diode, and when diode current flow, automatically applies grid Continuity signal.
In addition, this gate driver allows in wide opereating specification, to carry out no-voltage and zero current turn on-switch, and is also provided for the dead time compensation of stable operation.
Although described some embodiment in the disclosure, these embodiment provide as just example for those of ordinary skills, and can to make various modifications and change and not deviate from the spirit and scope of the present invention be apparent.Correspondingly, scope of the present invention should only be limited by claims and its equivalents.
Claims (6)
1. a series resonant converter, comprising:
The switch element that comprises a plurality of switches, for converting DC voltage to AC voltage by alternation direct voltage;
Comprise the resonant inductor that is connected in series and the LC resonant circuit of resonant capacitor, described LC resonant circuit is connected to described switch element, and uses the resonance being generated by described resonant inductor and described resonant capacitor to come converting transmission from the frequency characteristic of the described AC voltage of described switch element;
The transformer that comprises armature winding and secondary winding, described armature winding is connected to described LC resonant circuit, and the ratio of the quantity of the number of turn in the induced potential in described secondary winding and described secondary winding and the quantity of the number of turn in described armature winding is proportional;
Be connected to described transformer described secondary winding and with the secondary capacitor of described transformers connected in parallel;
The bridge rectifier that comprises a plurality of rectifier diodes, for converting the alternating voltage of inducting at described secondary winding to direct voltage; And
Gate driver, described gate driver detects the conducting of the anti-paralleled diode that is connected to described switch and when described anti-paralleled diode conducting, exports conducting gating signal to connect described switch,
Wherein, described gate driver comprises:
The first resistor that is connected to input, pulse voltage signal inputs to described input;
Be parallel-connected to described first resistor of described input end and use the capacitor of the conducting pulse voltage charging applying by described input;
The semiconductor switch that comprises source electrode, grid and drain electrode, the gate node of output that described source electrode, grid and drain electrode are connected respectively to the output node of described input, described capacitor and are connected to the described switch of described switch element;
The 3rd resistor connecting between the described drain electrode of described semiconductor switch and the described gate node of described output; And
By described the first resistor and described capacitor, be connected to described input to form the 4th resistor of conducting path.
2. series resonant converter as claimed in claim 1, is characterized in that, described secondary capacitor has the electric capacity less than described resonant capacitor.
3. series resonant converter as claimed in claim 1 or 2, is characterized in that, described secondary capacitor has the electric capacity of the 1/20-1/5 that is described resonant capacitor electric capacity.
4. series resonant converter as claimed in claim 1 or 2, it is characterized in that, described secondary capacitor can have than the low impedance of circuit being comprised of described rectifier diode and load, and the described secondary capacitor that is used in the load current charging of inducting in described secondary winding causes the resonance current of described LC resonant circuit to increase fast.
5. series resonant converter as claimed in claim 1, it is characterized in that, between described the first resistor and the collector node of described output, be further in series with the second resistor and the first diode, this second resistor is one another in series and is connected with the first diode.
6. series resonant converter as claimed in claim 5, is characterized in that, described the second resistor has than the large resistance of described the 4th resistor.
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| KR10-2010-0070290 | 2010-07-21 | ||
| KR1020100070290A KR101000561B1 (en) | 2010-07-21 | 2010-07-21 | Series resonant converter |
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| KR101385979B1 (en) | 2012-11-26 | 2014-04-16 | 한국전기연구원 | Series resonant converter |
| KR101628133B1 (en) * | 2014-07-04 | 2016-06-08 | 현대자동차 주식회사 | A Pulse Width Modulation Resonance Converter and Charger for Vehicle Using the Same |
| KR101721321B1 (en) * | 2015-01-28 | 2017-03-31 | 단국대학교 산학협력단 | Hybride type LED Power Supply |
| KR102009200B1 (en) | 2016-07-04 | 2019-08-09 | 숭실대학교산학협력단 | Bidirectional Converter with Auxiliary LC Resonant Circuit and Operating Method thereof |
| US10992233B2 (en) | 2017-01-31 | 2021-04-27 | Vestas Wind Systems A/S | Sub-resonant mode dual control of DC-DC converter with secondary-side resonant tank |
| FR3064849B1 (en) * | 2017-03-31 | 2019-06-07 | Centum Adetel Transportation | HYBRID POWER CELL |
| CN107528477A (en) * | 2017-08-08 | 2017-12-29 | 西南交通大学 | A kind of quasi-resonance soft switch double-transistor flyback DC/DC converters |
| TWI745800B (en) | 2018-12-21 | 2021-11-11 | 美商米沃奇電子工具公司 | Battery pack chargers, methods of controlling an operation of a battery pack charger, and charging circuits for a battery pack charger |
| KR102393664B1 (en) | 2020-11-17 | 2022-05-02 | 한국전기연구원 | Inverter switch gate driver and inverter |
| KR102674726B1 (en) * | 2022-09-06 | 2024-06-12 | 중앙대학교 산학협력단 | Gate driver and resonant converter |
| KR20240047770A (en) | 2022-10-05 | 2024-04-12 | 한국전기연구원 | Resonant converter and inverter with soft switching function |
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| KR100577472B1 (en) | 2004-12-23 | 2006-05-10 | 재단법인 포항산업과학연구원 | Boost type series resonant converter |
| JP4640783B2 (en) | 2005-03-24 | 2011-03-02 | 新電元工業株式会社 | Series resonant converter |
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| US6580255B1 (en) * | 2002-02-12 | 2003-06-17 | Technical Witts, Inc. | Adaptive gate drivers for zero voltage switching power conversion circuits |
| CN1438759A (en) * | 2002-02-15 | 2003-08-27 | 索尼株式会社 | Switch power-supply circuit |
| CN1832316A (en) * | 2005-03-08 | 2006-09-13 | 索尼株式会社 | Switching power supply circuit |
| CN1925300A (en) * | 2005-09-02 | 2007-03-07 | 索尼株式会社 | Switching power supply circuit |
| CN1941593A (en) * | 2005-09-30 | 2007-04-04 | 索尼株式会社 | Switching power supply circuit |
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| KR101000561B1 (en) | 2010-12-14 |
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