CN101902423A - Alternate binary offset carrier (AltBOC) signal acquisition device - Google Patents
Alternate binary offset carrier (AltBOC) signal acquisition device Download PDFInfo
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Abstract
The invention discloses an alternate binary offset carrier (AltBOC) signal acquisition device. The device comprises a sideband processing module, a correlator module, a decision variable generating module and a threshold decision module, wherein the sideband processing module comprises a sub-carrier signal generator, a low pass filter and a multiplier; the correlator module comprises a spectrum-spreading code generator and an integrator; and the decision variable generating module comprises two component synthesizers, four comparators, an incoherent accumulator and three adders. In the device, the AltBOC signal is divided into two-path like quadrature phase-shift keying (QPSK) signals by frequency shift to eliminate the wrong acquisition caused by the sub-carrier signal of the AltBOC signal; and the upper and lower sideband signals of the AltBOC signal adopt an incoherent accumulation manner so as to reduce the complexity of realization.
Description
Technical field
The invention belongs to GNSS signal processing technology field, be specifically related to a kind of alternate binary offset carrier (AltBOC) signal acquisition device.
Background technology
The gps system of a new generation and the satellite navigation signals of Galileo system will generally adopt binary offset carrier (BOC) modulation technique.Compare with traditional BPSK modulation system, the BOC modulation system has the navigation of raising frequency range utilance, suppresses the signal multipath error, reduces the signal coherence loss, improves pseudo range measurement precision, advantages such as enhancing signal interference free performance.Alternate binary offset carrier (AltBOC) signal is the expansion to the BOC signal, and the spectrum structure with the BOC signal similar is arranged.The AltBOC signal carries out frequency spectrum shift by the subcarrier signal of plural form, realized in the signal frequency range on lower sideband transmission of one line navigation signal respectively.At present, the Galileo system will at first broadcast AltBOC (15,10) signal in the E5 frequency range.The E5 frequency range comprises E5a and E5b two parts, and E5a and E5b all have mutually orthogonal spreading code, and the AltBOC modulation is moved lower sideband with the spreading code of E5a, the spreading code of E5b is moved upper sideband, and realized the permanent envelope modulation of signal.Because AltBOC (15,10) signal takies the frequency range of broad, therefore utilize the whole E5 frequency band signals of double-side band technical finesse can obtain the obvious raising of positioning performance.
The last lower sideband of AltBOC signal has been modulated E5a and E5b two-way navigation signal respectively, each road signal has also comprised data (Data) and pilot tone (Pilot) path, usually data (Data) and pilot tone (Pilot) path of representing E5a with E5a-I and E5a-Q are represented data (Data) and pilot tone (Pilot) path of E5b with E5b-I and E5b-Q.According to the design feature of AltBOC signal, can adopt multiple scheme to realize signal capture.The catching method of AltBOC (15,10) is summarized as follows:
(1) monolateral band acquisition algorithm (SSB)
Only catch E5a pilot channel (E5a-Q) or E5b pilot channel (E5b-Q)
Only catch E5 signal (E5a-I and E5a-Q) or E5b signal (E5b-I and E5b-Q)
(2) double-side band acquisition algorithm (DSB)
Catch E5a-Q and E5b-Q simultaneously
Noncoherent accumulation is in conjunction with E5a and E5b
(3) full frequency band independent acquisition algorithm (FIC)
Catch one of E5a-I, E5a-Q, E5b-I, E5b-Q
Coherent accumulation is in conjunction with E5a-Q and E5b-Q, or E5a-I and E5b-I
Noncoherent accumulation is in conjunction with E5a-I and E5a-Q, or E5b-I and E5b-Q
Noncoherent accumulation is in conjunction with E5a and E5b
(4) direct acquisition algorithm
Class 8-PSK (Phase Shift Keying) handles
Monolateral band acquisition algorithm by band pass filter (bandwidth is generally 20.46MHz), obtains E5a or E5b signal component with AltBOC, carries out relevant treatment with local spreading code then.Monolateral tape handling method does not need the local subcarrier signal, can obtain the correlation function characteristic similar to BPSK (10).The double-side band facture is handled E5a and E5b signal component simultaneously, can obtain the correlation function characteristic similar to BPSK (10) equally, compares signal to noise ratio with monolateral tape handling method and is improved.The local signal of full frequency band independent acquisition algorithm is that spreading code and subcarrier signal multiply each other and obtain, and intersects intermodulation component (for the signal component that realizes that permanent envelope is introduced) but do not utilize.Direct prize law makes full use of each signal component of AltBOC, and local signal is the 8-PSK signal.
The acquisition performance of FIC and direct acquisition algorithm slightly is better than the sideband Processing Algorithm.But because FIC and direct acquisition algorithm can bring the erroneous arrest phenomenon, actual AltBOC signal capture generally adopts sideband Processing Algorithm (SSB or DSB).Traditional sideband Processing Algorithm by the Combined Treatment of noncoherent accumulation realization data and pilot channel signal, has been introduced a square loss, influences acquisition performance.
Summary of the invention
The present invention proposes a kind of new AltBOC signal capture device,, adopt coherent accumulation, reduced a square loss, improved acquisition performance in conjunction with homophase and the quadrature component of (descending) sideband on the AltBOC signal by posteriority data estimator position information.
A kind of alternate binary offset carrier (AltBOC) signal acquisition device of the present invention comprises sideband processing module, correlator block, decision variable generating module and threshold judgement module.
The sideband processing module comprises the complex carrier signal signal generator, low pass filter and multiplier;
The input signal of sideband processing module is s
AltBOC (m, n)(t)+n (t), wherein: s
AltBOC (m, n)(t) be the base band alternate binary offset carrier (AltBOC) signal, n (t) is a complex radical band noise item;
The complex carrier signal signal generator is output as
With
Wherein: f
Sc=mf
0The subcarrier frequency of expression AltBOC signal, f
0=1.023MHz;
Phase error behind the expression frequency spectrum shift;
The complex carrier signal signal and the input signal of the output of complex carrier signal signal generator pass through multiplier, realize AltBOC (m, n) the sideband signals frequency spectrum oppositely moves, obtaining two-way does not have the baseband signal of subcarrier, it is passed through low pass filter, the low pass filter filters out out of band signal, the output s of low pass filter filter
L(t) and s
U(t) be class QPSK signal plus noise, QPSK represents the quarternary phase-shift keying (QPSK) signal, that is:
Wherein,
Be the phase error behind the frequency spectrum shift, n
L(t) and n
U(t) be the result that n (t) obtains after frequency spectrum shift and Filtering Processing; e
L-I(t)=c
L-I(t) d
L-1(t), e
L-Q(t)=c
L-Q(t) d
L-Q(t), e
U-I(t)=c
U-I(t) d
U-I(t) and e
U-Q(t)=c
U-Q(t) d
U-Q(t) be AltBOC (m, n) quadrature component of the in-phase component of the quadrature component of the in-phase component of lower sideband signal, lower sideband signal, upper side band signal and upper side band signal respectively; C wherein
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) be corresponding spread-spectrum code signals; d
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be corresponding data bit information, i.e. navigation message; s
L(t) and s
U(t) comprised required navigator fix signal e in
L-I(t)+je
L-Q(t) and e
U-1(t)+je
U-Q(t);
Correlator block comprises spreading code maker and integrator;
The spreading code maker generates four local spread-spectrum code signals: c
L-I(t+ τ), c
L-Q(t+ τ), c
U-I(t+ τ) and c
U-Q(t+ τ), wherein τ is the delay of local spreading code with respect to the input signal spreading code; The output s of sideband processing module
L(t) with local spread-spectrum code signals c
L-I(t+ τ) and c
L-Q(t+ τ) is by the multiplier processing of multiplying each other, the output s of sideband processing module
U(t) with local spread-spectrum code signals c
U-I(t+ τ) and c
U-Q(t+ τ) by the multiplier processing of multiplying each other, and four output signal results of multiplier are carried out integral processing by four integrators respectively, and the output of four integrators is respectively Z
L-I(t), Z
L-Q(t), Z
U-I(t) and Z
U-Q(t), be specially:
Wherein, T is the time of integration, d among the present invention
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be changeless in the time of integration, value is ± 1; R
L-I(τ), R
L-Q(τ), R
U-I(τ) and R
U-Q(τ) be c respectively
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) normalized autocorrelation functions; I
L-I(τ), I
L-Q(τ), I
U-I(τ) and I
U-Q(τ) real part of representing integrator to export, Q
L-I(τ), Q
L-Q(τ), Q
U-I(τ) and Q
U-Q(τ) imaginary part of representing integrator to export; n
L-I(τ), n
L-Q(τ), n
U-I(τ) and n
U-Q(τ) be corresponding noise item;
Decision variable generating module comprises two component synthesizers, four comparators, incoherent accumulator and three adders;
The Z of correlator block output
L-I(t) and Z
L-Q(t) input to the one-component synthesizer, component synthesizers is according to Z
L-I(t) real part and Z
L-Q(t) the relative syntactics of imaginary part, and Z
L-I(t) imaginary part and Z
L-Q(t) the relative syntactics of real part makes up and is squared to it, obtains two groups of signal components, i.e. { [I
L-I(τ)+Q
L-Q(τ)]
2, [I
L-I(τ)-Q
L-Q(τ)]
2And { [Q
L-I(τ)+I
L-Q(τ)]
2, [Q
L-I(τ)-I
L-Q(τ)]
2; Every group of signal component exports a comparator to, and comparator is selected value the greater in every group, and the output of two comparators is carried out component summation processing through an adder, obtains;
V
LK(τ)＝max{[I
L-I(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-Q
L-Q(τ)]
2}+
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}
Wherein, max{A, B} represent to get A and B intermediate value the maximum;
In like manner, the Z of correlator block output
U-I(t) and Z
U-Q(t) also input to another component synthesizers, component synthesizers is according to Z
U-I(t) real part and Z
U-Q(t) the relative syntactics of imaginary part, and Z
U-I(t) imaginary part and Z
U-Q(t) the relative syntactics of real part makes up, and squared to it, obtains two groups of signal components, i.e. { [I
U-I(τ)+Q
U-Q(τ)]
2, [I
U-I(τ)-Q
U-Q(τ)]
2And { [Q
U-I(τ)+I
U-Q(τ)]
2, [Q
U-I(τ)-I
U-Q(τ)]
2; Every group of signal component exports a comparator to, and comparator is selected value the greater in every group, and the output of two comparators is carried out component summation processing through an adder, obtains;
V
Uk(τ)＝max{[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
With V
Lk(τ) and V
Uk(τ) input to another adder, obtain:
V
k(τ)＝V
LK(τ)+V
Uk(τ)
＝max{[I
L-I(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-Q
L-Q(τ)]
2}+
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}+
max{[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
Incoherent accumulator is to the V of input
k(τ) carry out non-coherent accumulation, obtain:
Wherein, K is the non-coherent accumulation number of times; The output y (τ) of incoherent accumulator is the prize judgment variable;
The judgment variables y (τ) of decision variable generating module output inputs to the threshold judgement module, the threshold judgement module compares y (τ) and default decision threshold, determines whether acquisition success of signal, if the signal capture success, end signal is caught process, the processing stage that entering signal being followed the tracks of; If do not capture signal, repeat signal capture, up to the signal capture success.
The invention has the advantages that:
(1) by frequency spectrum shift, be two-way class QPSK signal with the AltBOC signal decomposition, eliminated the erroneous arrest phenomenon that the subcarrier signal of AltBOC signal brings.
(2) by posteriority data estimator position information, coherent accumulation has improved acquisition performance in conjunction with homophase and the quadrature component of (descending) sideband on the AltBOC signal;
(3) the last lower sideband signal of AltBOC adopts the noncoherent accumulation mode, has reduced implementation complexity.
Description of drawings
Fig. 1 is the structural representation of a kind of alternate binary offset carrier (AltBOC) signal acquisition device of the present invention;
Fig. 2 is the subcarrier schematic diagram of AltBOC signal;
Fig. 3 is a relatively schematic diagram of acquisition performance.
Among the figure:
1-sideband processing module 2-correlator block 3-decision variable generating module
4-threshold judgement module
101-complex carrier signal signal generator 102-low pass filter 103-multiplier
201-spreading code maker 202-integrator
301-component synthesizers 302-comparator 303-incoherent accumulator
The 304-adder
Embodiment
Below in conjunction with drawings and Examples the present invention is elaborated.
The present invention is a kind of alternate binary offset carrier (AltBOC) signal acquisition device, as shown in Figure 1, comprises sideband processing module 1, correlator block 2, decision variable generating module 3 and threshold judgement module 4.
Base band AltBOC (m, n) the universal expression formula of signal is:
In the formula,
M and n are natural number, represent AltBOC (m, n) the subcarrier frequency f of signal respectively
Sc=mf
0With the spreading code frequency f
c=nf
0, f wherein
0=1.023MHz; e
L-I(t)=c
L-I(t) d
L-I(t), e
L-Q(t)=c
L-Q(t) d
L-Q(t), e
U-I(t)=c
U-I(t) d
U-I(t) and e
U-Q(t)=c
U-Q(t) d
U-Q(t) be AltBOC (m, the n) quadrature component of the in-phase component of lower sideband signal, lower sideband signal, the in-phase component of upper side band signal and the quadrature component of upper side band signal respectively; C wherein
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) being corresponding spread-spectrum code signals, is mutually orthogonal between them; d
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be corresponding data bit information (navigation message); In addition,
Sc
As(t) and sc
Ap(t) be subcarrier signal:
Sign () is-symbol function wherein.Sc
As(t) and sc
Ap(t) period T
Sc=1/f
Sc, sc
As(t) and sc
Ap(t) time domain waveform as shown in Figure 2.In the formula (1) last two
With
It is the intermodulation component of introducing for the permanent envelope modulation that realizes signal.Traditionally, (m n) represents alternate binary offset carrier (AltBOC) signal, then adopts s when relating to mathematical operation to adopt AltBOC when not relating to mathematical operation
AItBOC (m, n)(t) represent.
The input signal of sideband processing module 1 is s
AItBOC (m, n)(t)+and n (t), wherein n (t) is a complex radical band noise item.(m, n) sc is passed through in modulation to AltBOC
As(t)-jsc
As(t-T
Sc/ 4) component is with e
L-I(t)+je
L-Q(t) move lower sideband in the frequency range, passed through sc
As(t)+jsc
As(t-T
Sc/ 4) component is with e
U-I(t)+je
U-Q(t) moved upper sideband in the frequency range.In order to extract navigation signal, among the present invention, complex carrier signal signal generator 101 is exported
With
The complex carrier signal signal of complex carrier signal signal generator 101 outputs and input signal are by multiplier 103, realize AltBOC (m, n) the sideband signals frequency spectrum oppositely moves, obtaining two-way does not have the baseband signal of subcarrier, then it is passed through low pass filter 102, low pass filter 102 filtering out of band signals, the bandwidth of low pass filter 102 is by AltBOC (m, n) (descending) the sideband signals frequency spectrum main lobe width of going up of signal determines the output s of low pass filter 102
L(t) and s
U(t) be class QPSK signal (baseband form) plus noise, that is:
Wherein,
Be the phase error behind the frequency spectrum shift, n
L(t) and n
U(t) be the result that n (t) obtains after frequency spectrum shift and Filtering Processing.s
L(t) and s
U(t) comprised required navigator fix signal e in
L-I(t)+je
L-Q(t) and e
U-I(t)+je
U-Q(t).
Spreading code maker 201 generates four local spread-spectrum code signals: c
L-I(t+ τ), c
L-Q(t+ τ), c
U-I(t+ τ) and c
U-Q(t+ τ), wherein τ is the delay of local spreading code with respect to the input signal spreading code.The output s of sideband processing module 1
L(t) with local spread-spectrum code signals c
L-I(t+ τ) and c
L-Q(t+ τ) is by the multiplier processing of multiplying each other, the output s of sideband processing module 1
U(t) with local spread-spectrum code signals c
U-I(t+ τ) and c
U-Q(t+ τ) is by the multiplier processing of multiplying each other.Local spread-spectrum code signals and S
L(t) and s
U(t) multiply each other after, carry out integral processing by four integrators 202 respectively.
Four integrators 202 are output as and are respectively:
In like manner
Wherein, T is the time of integration, thinks data message (d generally speaking
L-I (t), d
L-Q(t), d
U-I(t) and d
U-Q(t)) be changeless in the time of integration, and value is ± 1; R
L-I(τ), R
L-Q(τ), R
U-I(τ) and R
U-Q(τ) be c respectively
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) normalized autocorrelation functions; I
L-I(τ), I
L-Q(τ), I
U-I(τ) and I
U-Q(τ) real part of representing integrator 202 to export, Q
L-I(τ), Q
L-Q(τ), Q
U-I(τ) and Q
U-Q(τ) imaginary part of representing integrator 202 to export; n
L-I(τ), n
L-Q(τ), n
U-I(τ) and n
U-Q(τ) be corresponding noise item.
Decision variable generating module 3 comprises two component synthesizers 301, four comparators 302, incoherent accumulator 303 and three adders 304.Decision variable generating module 3 is utilized the output signal of correlator block 2, generates to detect judgment variables.
Because d
L-I(t), d
L-Q(t), d
U-I(t) and d
U-QTherefore (t) be unknown, can have ambiguity when on carrying out, (descending) in-phase component of sideband and quadrature component coherent accumulation.The value of considering data bit information is ± 1, can enumerate possible value condition earlier, comes the value of data estimator position again by comparator.
The Z of correlator block 2 outputs
L-I(t) and Z
L-Q(t) input to one-component synthesizer 301, component synthesizers 301 is according to Z
L-I(t) real part and Z
L-Q(t) the relative syntactics that imaginary part is possible, and Z
L-I(t) imaginary part and Z
L-Q(t) the possible relative syntactics of real part makes up, and squared to it, obtains two groups of signal components, i.e. { [I
L-I(τ)+Q
L-Q(τ)]
2, [I
L-I(τ)-Q
L-Q(τ)]
2And { [Q
L-I(τ)+I
L-Q(τ)]
2, [Q
L-I(τ)-I
L-Q(τ)]
2.Every group of signal component exports a comparator 302 to, and comparator 302 is selected value the greater in every group, and the output of two comparators 302 is carried out component summation processing through an adder 304, obtains;
V
LK(τ)＝max{[I
L-1(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-Q
L-Q(τ)]
2}+(10)
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}
Wherein, max{A, B} represent to get A and B intermediate value the maximum.Because I
L-I(τ) and Q
L-IBe that in-phase component by lower sideband signal obtains (τ), and I
L-Q(τ) and Q
L-QBe that quadrature component by lower sideband signal obtains (τ), so V
Lk(τ) homophase of AltBOC signal lower sideband and the coherent accumulation of quadrature component have in fact been finished.
In like manner, the Z of correlator block 2 outputs
U-I(t) and Z
U-Q(t) also input to one-component synthesizer 301, component synthesizers 301 is according to Z
U-I(t) real part and Z
U-Q(t) the relative syntactics that imaginary part is possible, and Z
U-I(t) imaginary part and Z
U-Q(t) the possible relative syntactics of real part makes up, and squared to it, obtains two groups of signal components, i.e. { [I
U-I(τ)+Q
U-Q(τ)]
2, [I
U-I(τ)-Q
U-Q(τ)]
2And { [Q
U-I(τ)+I
U-Q(τ)]
2, [Q
U-I(τ)-I
U-Q(τ)]
2.Every group of signal component exports a comparator 302 to, and comparator 302 is selected value the greater in every group, and the output of two comparators 302 is carried out component summation processing through an adder 304, obtains;
V
Uk(τ)＝max[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+(11)
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
Because I
U-I(τ) and Q
U-IBe that in-phase component by upper side band signal obtains (τ), and I
U-Q(τ) and Q
U-QBe that quadrature component by upper side band signal obtains (τ), so V
Uk(τ) AltBOC (m, n) homophase of signal upper sideband and quadrature component coherent accumulation have in fact been finished.
With V
Lk(τ) and V
Uk(τ) input to an adder 304, adder 304 finish AltBOC (m, the n) noncoherent accumulation of lower sideband signal on the signal obtains:
V
k(τ)＝V
Lk(τ)+V
Uk(τ)
＝max{[I
L-I(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-Q
L-Q(τ)]
2}+
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}+ (12)
max{[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
The V of 303 pairs of inputs of incoherent accumulator
k(τ) carry out non-coherent accumulation, obtain:
Wherein, K is the non-coherent accumulation number of times.The output y (τ) of incoherent accumulator 303 is the prize judgment variable.
The judgment variables y (τ) of decision variable generating module 3 output inputs to threshold judgement module 4, and threshold judgement module 4 compares y (τ) with default decision threshold, determines whether acquisition success of signal.If the signal capture success, end signal is caught process, the processing stage that entering signal being followed the tracks of.If do not capture signal, repeat signal capture, up to the signal capture success.
Embodiment:
In the embodiment in front, (m, n) signal is illustrated embodiments of the present invention to utilize general AltBOC.The AltBOC (15,10) that adopts with the Galileo system is an example below, and further the present invention will be described.
The last lower sideband signal of the AltBOC of Galileo system (15,10) signal is respectively E5a and E6a, and every road signal all comprised data and pilot channel, and data and pilot channel are corresponding to general AltBOC (m, n) homophase in the signal and quadrature component.Wherein, the data path modulation has navigation message (data bit information), pilot channel and unmodulated navigation message (data bit information).So, have at present embodiment
e
L-I(t)＝e
E5a-I(t)＝c
E5a-I(t)d
E5a-I(t) (14)
e
L-Q(t)＝e
E5a-Q(t)＝c
E5a-Q(t) (15)
e
U-I(t)＝e
E5b-I(t)＝c
E5b-I(t)d
E5b-I(t) (16)
e
U-Q(t)＝e
E5b-Q(t)＝c
E5b-Q(t) (17)
E wherein
E5a-I(t) the data path signal of expression E5a, e
E5a-Q (t)The pilot channel signal of expression E5a, e
E5b-I(t) the data path signal of expression E5b, e
E5b-Q(t) the pilot channel signal of expression E5b; c
E5a-I(t), c
E5a-Q(t), c
E5b-I(t), c
E5b-Q(t) be the spread-spectrum code signals of respective channels; d
E5a-I(t), d
E5b-I(t) be the textual information of corresponding data path.The spreading code cycle of AltBOC (15,10) is 1ms, and spread-spectrum code rate is f
c=10*1.023MHz, subcarrier data rate f
Sc=15*1.023MHz.
Present embodiment realizes on the hardware platform that with FPGA+DSP is core, and FPGA selects the VC4VSX55 in the VIRTEX-4 series of Xilinx company for use, and DSP is the floating type TMS320C6713 of TI company.Sideband processing module 1, correlator block 2 realizes in FPGA; Decision variable generating module 3 and threshold judgement module 4 realize in DSP.
Complex carrier signal signal generator 101 in the sideband processing module 1 generates the required complex carrier signal signal of sideband signals frequency spectrum shift:
With
Utilize
With
Only need to generate cosine and sine signal
With
Get final product." just/cosine signal generator " IP kernel in FPGA can satisfy the demand that generates carrier signal fully.After last lower sideband spectrum is moved and finished, need carry out low-pass filtering treatment.Generally speaking, only be concerned about and go up lower sideband signal main lobe spectrum information, so the bandwidth of low pass filter generally equals main lobe width, i.e. BW=20.46MHz.So far, the output s of sideband processing module 1
L(t) and s
U(t) be approximately, the data of E5a and pilot signal are through baseband form (comprising noise) and the data of E5b and the baseband form (comprising noise) of pilot signal signal after the QPSK modulation of signal after the QPSK modulation, promptly
Wherein,
Be the phase error behind the frequency spectrum shift, n
E5a(t) and n
E5b(t) be noise item.
The output s of sideband processing module 1
L(t) and s
U(t) enter correlator block 2, carry out relevant treatment with local spreading code.Spreading code maker 201 generates four local spread-spectrum code signals: c
E5a-I(t+ τ), c
E5a-Q(t+ τ), c
E5b-I(t+ τ) and c
E5b-Q(t+ τ), wherein τ is the delay of the relative input signal spreading code of local spreading code.The spreading code cycle of considering AltBOC (15,10) is 1ms, and the T time of integration of integrator 202 also is made as 1ms.The output of correlator block 2 is s
L(t) and s
U(t) carry out result after the relevant treatment with local spreading code, promptly
(21)
(23)
Wherein, R
E5a-I(τ), R
E5a-Q(τ), R
E5b-I(τ) and R
E5b-Q(τ) be c respectively
E5a-I(t), c
E5a-Q(t), c
E5b-I(t) and c
E5b-Q(t) normalized autocorrelation functions; I
E5a-I(τ), I
E5a-Q(τ), I
E5b-I(τ) and I
E5b-Q(τ) real part of representing integrator to export, Q
E5a-I(τ), Q
E5a-Q(τ), Q
E5b-I(τ) and Q
E5b-Q(τ) imaginary part of representing integrator to export; n
E5a-I(τ), n
E5a-Q(τ), n
E5b-I(τ) and n
E5b-Q(τ) be corresponding noise item.
Decision variable generating module 3 is utilized the output signal of correlator block 2, generates to detect judgment variables.Through the signal after component synthesizers 301, comparator 302 and the adder 304 be
V
k(τ)＝max{[I
E5a-I(τ)+Q
E5a-Q(τ)]
2，[I
E5a-I(τ)-Q
E5a-Q(τ)]
2}+
max{[Q
E5a-I(τ)+I
E5a-Q(τ)]
2，[Q
E5a-I(τ)-I
E5a-Q(τ)]
2}+
(24)
max{[I
E5b-I(τ)+Q
E5b-Q(τ)]
2，[I
E5b-I(τ)-Q
E5b-Q(τ)]
2}+
max{[Q
E5b-I(τ)+I
E5b-Q(τ)]
2，[Q
E5b-I(τ)-I
E5b-Q(τ)]
2}
Through the output judgment variables after the incoherent accumulator 303 be
Wherein, K is the non-coherent accumulation number of times.
The judgment variables y (τ) of decision variable generating module 3 outputs and the thresholding η of threshold judgement module 4 compare, and whether catch with judgement.Generally, thresholding η can adopt the Newman Pearson criterion to choose, and this criterion is at constraint false alarm probability P
FaUnder the constant situation, make alarm dismissal probability minimum (or detection probability P
dMaximum).
Fig. 3 has provided the present invention and has adopted under the Newman Pearson criterion condition, detection probability P
dResult of the test with carrier-to-noise ratio C/NO relation.Experimental condition: P
Fa=10
-3, the time of integration T=1ms, non-coherent accumulation number of times K=5 postpones τ=0.As a comparison, provided traditional noncoherent accumulation algorithm among Fig. 3 in conjunction with last (descending) data of sideband and the acquisition performance of pilot signal (being labeled as " Non-coherent " among the figure).Obviously, adopt acquisition equipment of the present invention can bring tangible acquisition performance to promote.
Claims (2)
1. an alternate binary offset carrier (AltBOC) signal acquisition device is characterized in that: comprise sideband processing module, correlator block, decision variable generating module and threshold judgement module;
The sideband processing module comprises the complex carrier signal signal generator, low pass filter and multiplier;
The input signal of sideband processing module is s
AltBOC (m, n)(t)+n (t), wherein: s
AltBOC (m, n)(t) be the base band alternate binary offset carrier (AltBOC) signal, n (t) is a complex radical band noise item;
The complex carrier signal signal generator is output as
With
Wherein: f
Sc=mf
0The subcarrier frequency of expression AltBOC signal, f
0=1.023MHz;
Phase error behind the expression frequency spectrum shift;
The complex carrier signal signal and the input signal of the output of complex carrier signal signal generator pass through multiplier, realize AltBOC (m, n) the sideband signals frequency spectrum oppositely moves, obtaining two-way does not have the baseband signal of subcarrier, it is passed through low pass filter, the low pass filter filters out out of band signal, the output s of low pass filter filter
L(t) and s
U(t) be class QPSK signal plus noise, QPSK represents the quarternary phase-shift keying (QPSK) signal, that is:
Wherein,
Be the phase error behind the frequency spectrum shift, n
L(t) and n
U(t) be the result that n (t) obtains after frequency spectrum shift and Filtering Processing; e
L-I(t)=c
L-I(t) d
L-I(t), e
L-Q(t)=c
L-Q(t) d
L-Q(t), e
U-I(t)=c
U-I(t) d
U-I(t) and e
U-Q(t)=c
U-Q(t) d
U-Q(t) be AltBOC (m, the n) quadrature component of the in-phase component of lower sideband signal, lower sideband signal, the in-phase component of upper side band signal and the quadrature component of upper side band signal respectively; C wherein
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) be corresponding spread-spectrum code signals; d
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be corresponding data bit information, i.e. navigation message; s
L(t) and s
U(t) comprised required navigator fix signal e in
L-I(t)+je
L-Q(t) and e
U-I(t)+je
U-Q(t);
Correlator block comprises spreading code maker and integrator;
The spreading code maker generates four local spread-spectrum code signals: c
L-I(t+ τ), c
L-Q(t+ τ), c
U-I(t+ τ) and c
U-Q(t+ τ), wherein τ is the delay of local spreading code with respect to the input signal spreading code; The output s of sideband processing module
L(t) with local spread-spectrum code signals c
L-I(t+ τ) and c
L-Q(t+ τ) is by the multiplier processing of multiplying each other, the output s of sideband processing module
U(t) with local spread-spectrum code signals c
U-I(t+ τ) and c
U-Q(t+ τ) carries out integral processing by four integrators by the multiplier processing of multiplying each other respectively with four output signals of multiplier, and the output of four integrators is respectively Z
L-I(t), Z
L-Q(t), Z
U-I(t) and Z
U-Q(t), be specially:
Wherein, T is the time of integration, d among the present invention
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be changeless in the time of integration, value is ± 1; R
L-I(τ), R
L-Q(τ), R
U-I(τ) and R
U-Q(τ) be c respectively
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) normalized autocorrelation functions; I
L-I(τ), I
L-Q(τ), I
U-I(τ) and I
U-Q(τ) real part of representing integrator to export, Q
L-I(τ), Q
L-Q(τ), Q
U-I(τ) and Q
U-Q(τ) imaginary part of representing integrator to export; n
L-I(τ), n
L-Q(τ), n
U-I(τ) and n
U-Q(τ) be corresponding noise item;
Decision variable generating module comprises two component synthesizers, four comparators, incoherent accumulator and three adders;
The Z of correlator block output
L-I(t) and Z
L-Q(t) input to the one-component synthesizer, component synthesizers is according to Z
L-I(t) real part and Z
L-Q(t) the relative syntactics of imaginary part, and Z
L-I(t) imaginary part and Z
L-Q(t) the relative syntactics of real part makes up and is squared to it, obtains two groups of signal components, i.e. { [I
L-I(τ)+Q
L-Q(τ)]
2, [I
L-I(τ)-Q
L-Q(τ)]
2And { [Q
L-I(τ)+I
L-Q(τ)]
2, [Q
L-I(τ)-I
L-Q(τ)]
2; Every group of signal component exports a comparator to, and comparator is selected value the greater in every group, and the output of two comparators is carried out component summation processing through an adder, obtains;
V
Lk(τ)＝max{[I
L-I(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-
L-Q(τ)]
2}+
(7)
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}
Wherein, max{A, B} represent to get A and B intermediate value the maximum;
In like manner, the Z of correlator block output
U-I(t) and Z
U-Q(t) also input to another component synthesizers, component synthesizers is according to Z
U-I(t) real part and Z
U-Q(t) the relative syntactics of imaginary part, and Z
U-I(t) imaginary part and Z
U-Q(t) the relative syntactics of real part makes up, and squared to it, obtains two groups of signal components, i.e. { [I
U-I(τ)+Q
U-Q(τ)]
2, [I
U-I(τ)-Q
U-Q(τ)]
2And { [Q
U-I(τ)+I
U-Q(τ)]
2, [Q
U-I(τ)-I
U-Q(τ)]
2; Every group of signal component exports a comparator to, and comparator is selected value the greater in every group, and the output of two comparators is carried out component summation processing through an adder, obtains;
V
Uk(τ)＝max{[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+
(8)
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
With V
Lk(τ) and V
Uk(τ) input to another adder, obtain:
V
k(τ)＝V
Lk(τ)+V
uk(τ)
＝max{[I
L-I(τ)+Q
L-Q(τ)]
2，[I
L-I(τ)-Q
L-Q(τ)]
2}+
max{[Q
L-I(τ)+I
L-Q(τ)]
2，[Q
L-I(τ)-I
L-Q(τ)]
2}+ (9)
max{[I
U-I(τ)+Q
U-Q(τ)]
2，[I
U-I(τ)-Q
U-Q(τ)]
2}+
max{[Q
U-I(τ)+I
U-Q(τ)]
2，[Q
U-I(τ)-I
U-Q(τ)]
2}
Incoherent accumulator is to the V of input
k(τ) carry out non-coherent accumulation, obtain:
Wherein, K is the non-coherent accumulation number of times; The output y (τ) of incoherent accumulator is the prize judgment variable;
The judgment variables y (τ) of decision variable generating module output inputs to the threshold judgement module, the threshold judgement module compares y (τ) and default decision threshold, determines whether acquisition success of signal, if the signal capture success, end signal is caught process, the processing stage that entering signal being followed the tracks of; If do not capture signal, repeat signal capture, up to the signal capture success.
2. a kind of alternate binary offset carrier (AltBOC) signal acquisition device according to claim 1 is characterized in that: the base band AltBOC of sideband processing module input (m, n) the universal expression formula of signal is:
In the formula, m and n are natural number, represent AltBOC (m, n) the subcarrier frequency f of signal respectively
Sc=mf
0With the spreading code frequency f
c=nf
0, f wherein
0=1.023MHz; e
L-I(t)=c
L-I(t) d
L-I(t), e
L-Q(t)=c
L-Q(t) d
L-Q(t), e
U-I(t)=c
U-I(t) d
U-I(t) and e
U-Q(t)=c
U-Q(t) d
U-Q(t) be AltBOC (m, the n) quadrature component of the in-phase component of lower sideband signal, lower sideband signal, the in-phase component of upper side band signal and the quadrature component of upper side band signal respectively; C wherein
L-I(t), c
L-Q(t), c
U-I(t) and c
U-Q(t) being corresponding spread-spectrum code signals, is mutually orthogonal between them; d
L-I(t), d
L-Q(t), d
U-I(t) and d
U-Q(t) be corresponding data bit information (navigation message); In addition,
Sc
As(t) and sc
Ap(t) be subcarrier signal:
Wherein: sign () is-symbol function; Sc
As(t) and sc
Ap(t) period T
Sc=1/f
Sc
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| CN105116425A (en) * | 2015-08-21 | 2015-12-02 | 西安空间无线电技术研究所 | Parallel AltBOC navigation signal intermediate frequency generation method |
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