WO2023009198A1

WO2023009198A1 - Passive class of modulators for mm-wave applications (5g/6g and radar)

Info

Publication number: WO2023009198A1
Application number: PCT/US2022/030622
Authority: WO
Inventors: Rashad RAMZAN; Muhammad Omar; Muhammad USMAN
Original assignee: Wi-LAN Research Inc.
Priority date: 2021-07-28
Filing date: 2022-05-24
Publication date: 2023-02-02

Abstract

A new class of passive modulators is described herein which can be manufactured using a standard CMOS or any other IC design technology. The proposed mm-wave modulator exhibits both phase and amplitude modulation, which can be combined to make a polar modulator using a single passive structure. The proposed modulators described herein have an excellent linearity and have good power handling capability compared to the active class of modulators.

Description

PASSIVE CLASS OF MODULATORS FOR MM-WAVE APPLICATIONS (5G/6G AND RADAR)

CROSS REFERENCE TO RELATED APPLICATIONS

This PCT International Application claims priority from Pakistani Provisional Patent Application No. 557/2021, filed on July 28, 2021, entitled “PASSIVE CLASS OF POLAR MODULATOR FOR MM-WAVE APPLICATIONS (5G/6G AND RADAR).

FIELD OF THE INVENTION

The invention described herein is related to Polar Modulators, sometimes called Quadrature Amplitude (QAM) Modulators, and to the related devices incorporating them for various communication systems including, but not limited to, 5G/6G Systems. Different aspects of the invention herein include Amplitude Modulation, Phase Modulation, Polar Modulation, and related systems for use in 5G/6G Systems and subsequent generations of communication protocols and systems including, but not limited to, mm-Wave and Terahertz (THz) communication systems. The polar modulators described herein support two modulation schemes, amplitude and angular modulation, and are suitable for implementation in bulk CMOS technology. The inventions herein further apply to on-chip magnitude and phase modulators for mm-wave and THz applications. The modulators described herein not only exhibit linear behavior but also have the capability to better manage power, making them power efficient compared with active modulators of the prior art.

BACKGROUND

The fifth and sixth generations of mobile communication technology demand high-speed communication systems at higher frequencies; as a result, designers must address the challenges and complexities of small size components that enable significantly higher data rates in millimeter band communications. The prior art of microwave components is mostly limited to below 6 GHz; therefore, it cannot be applied to the mm-wave bands (28-300 GHz). This significant disparity between mm-wave and microwave systems impacts design requirement of all components, including their fundamental architecture, the Digital Signal Processing (DSP), and signal performance of the baseband and user devices. The persons skilled in the art of communication understand different forms of modulations including amplitude, phase, and frequency modulations. These modulation schemes must use smaller-sized filters, avoid signal mixing, increase the communication range, and permit multiplexing of information signals in both time and frequency domains.

A modulator is a device that modulates an information signal onto a carrier signal in such a way such that it can be fully recovered. A typical modulator architecture 100 is shown in Figure 1, where the stub patches 104, 110, 108, and 112 are used for modulation. As a result, variations in the signal parameters at Port-3 106 cause changes in the signal at Port-1 102, and the modulated signal is received at Port-2 114. In a broader perspective, modulation schemes can be divided into two main categories: Continuous Modulation and Digital Modulation. Continuous modulation can be further divided into Amplitude Modulation (AM) and Angular Modulation. Angular modulation can be further divided into Frequency Modulation (FM) and Phase Modulation (PM). Analogous to Continuous Modulation, Digital Modulation is further divided into Amplitude Shift Keying (ASK), Phase Shift Keying (PSK), and Frequency Shift Keying (FSK).

A highly efficient transmitter is needed in mm-Wave MIMO communication due to large number of transmitters on a single chip. This demand can he met using Polar Modulators, as they use highly efficient non-linear powor amplifiers to amplify non- constant envelop modulation like QAM. Figure 2, shows a classic Polar Modulator 200, where non-linear class E or F switching Pas (power amplifiers) work as combiner 212,

A Polar Modulator separates the envelope in amplitude 204 and phase 208 components of a modulated input signal received at RF input port 202, Amplifiers 206 and 210 are low pow⁷er and low gain amplifiers which also work as isolators or buffers. The combiner 212, which is usually a non-linear switching PA, changes the amplitude of the carrier signal which is already phase-modulated. It is important to note that, the combiner is usually implemented using switch-mode amplifiers like class E or F or S PAs. In contrast, in the disclosed invention set forth herein, phase and amplitude modulation takes place on an unmodulated carrier inside the polar modulator block.

In prior art, the majority of polar modulators for mm-wave are designed using active components like combination of low⁷ pow⁷er linear and high-power non-linear power amplifiers. Linearity and power handling are the bottlenecks that limit their use in fully integrated chip designs, phase array radar, and MΪMΌ systems. The passive class of modulators are linear, and their power handling capability is much higher compared to active counterparts.

Polar modulators have gained more importance over the last few decades since they can do both angular and magnitude modulations. The Polar modulator described in U.S. Patent No. 6,834,08462 (Hietala) creates the amplitude and frequency signals by processing them in the digital domain. The angle modulated signal is used as a PA's input, while the amplitude signal controls its power supply.

The polar modulator described in U.S. Patent No. 8,369,80262 (Nakamura, et ai.) uses a temperature sensing and feedback mechanism to control the power amplification of a signal. This feature is primarily designed for enhanced power control to avoid degradation of performance of a modulator in real world environments in which the temperature varies significantly.

U.S. Patent No. 8,259,82261 (Feher) presents a polar modulation usage in WLAN, WiFi, remote control, and position finding applications. It describes designs for multiuse, multimode operable systems that can integrate various communication technologies and architectures.

A polar modulator with a delay-controlled transmitter is described in U.S. Patent No. 7,072,626 B2 (Hadjichristos). The system is described with one sample control circuit that acts as a negative-feedback closed-loop controller and adjusts the relative delay between the transmitter's envelope and phase to reduce the output signal’s ACPR.

In U.S. Patent No. 8,995,56762 (Rofougaran et ah), a polar modulator architecture is discussed in which power supply adjustments are used to drive a PA. However, the given system is composed of active elements such as mixers, adder circuits, etc., and its application to MMIMO applications is not feasible, especially in systems where chip area is limited.

A Wilkinson combiner with two different PAs to achieve polar modulation is another example, but it is undesirable for MMIMO applications because it needs a large number of PAs. Similarly, the multi -mode modulator described in German Patent No.

DEI 02008028326B4 (Van Waasen) also takes a large area for 5GNew Radio (NR.) MMIMO applications. SUMMARY OF THE INVENTION: in one embodiment, an amplitude modulator for communication applications is provided that includes a microstrip filter section having a rectangular shape, with a first connection port provided at a first end of the rectangular shape and a second connection port provided at a second end of the rectangular shape, the first end being located opposite from the second end, a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape, and a dedicated voltage source structured and configured to apply a modulating voltage signal to the microstrip filter section. The microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at one of a microwave frequency and a millimeter-wave (mm-wave) frequency, the split-ring microstrip filter possessing at least one of a Fano, EIT, and Lorentz resonance.

In another embodiment, a phase modulator for communication applications is provided that includes a microstrip filter section having a rectangular shape, with a first connection port provided at a first end of the rectangular shape and a second connection port provided at. a second end of the rectangular shape, the first end being located opposite from the second end, a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape, and at least one variable capacitor disposed in the open space gap of the ring resonator filter section. The microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at one of a microwave frequency, a millimeter-wave (mm- wave) frequency and a THz frequency, the split-ring microstrip filter possessing at least one of a Fano, EIT, and Lorentz resonance.

In still another embodiment, a polar modulator for communication applications is provided that includes a microstrip filter section having a rectangular shape, with a first connection port provided at a first end of the rectangular shape and a second connection port provided at a second end of the rectangular shape, the first end being located opposite from the second end, a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape, at least one variable capacitor disposed in the open space gap of the ring resonator filter section, and a dedicated voltage source structed and configured to apply a modulating voltage signal to the microstrip filter section. The microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at a predetermined frequency band, the split-ring microstrip filter possessing at. least, one of a Fano, EGG, and Lorentz resonance. BRIEF DESCRIPTION OF THE DRAWINGS

Details of one or more implementations of the inventions’ subject matter are outlined in the accompanying drawings briefly described below and the related description set forth herein. Other objects, features, aspects, and advantages will become apparent from the description, the drawings, and the claims. The reference numbers and designations in various drawings indicate the corresponding elements wherever deemed necessary, fire descriptions and features disclosed herein can be applied to mm-wave applications, THz applications, 5G/6G, and high-frequency Radars. Note that the relative dimensions of the drawings may not be drawn to scale.

Figure 1 shows the basic working principle of a typical modulator. Figure 2 shows the basic working principal of a polar modulator.

Figure 3 is a block diagram of 5G/6G front-end deploying several antennas.

Figure 4 shows an example of a Fano-based Amplitude Modulator according to aspects of the invention.

Figure 5 shows the amplitude response vs. frequency for a Fano-based modulator according to aspects of the invention.

Figure 6 shows an example of a Fano-based Amplitude Modulator in which the mutual inductance can be used to achieve the required attenuation while isolating the two sources according to aspects of the invention. Figure 7 shows an example of a Fano-based Amplitude Modulator in which the mutual inductance is replaced with a coupled microstrip line according to aspects of the invention.

Figure 8 shows the attenuation response of a Fano-based Amplitude Modulator resulting from changing the voltage provided by the voltage's source according to aspects of the invention.

Figure 9 shows the variation in magnitude of the signal when the applied voltage changes from 0V to 2V according to aspects of the invention.

Figure 10 shows the reduction in attenuation achieved by applying negative voltages according to aspects of the invention.

Figure 11 shows the amplitude variation caused by varying the voltage for example distances between the microstrip lines of d^:::0.5mm, lmm, and 1.5mm according to aspects of the invention.

Figure 12 shows an example of a multilayer Fano Amplitude Modulator according Figure 13 shows the magnitude response of a Multilayer Fano Amplitude Modulator according to aspects of the invention.

Figure 14 shows an example of a multilayer Fano amplitude modulator, changing the distance between the modulator and the plane carrying voltage, according to aspects of the invention. Figure 15 shows the response of the transmittance variation caused by the separation of the metal layers carrying the modulator structure and modulating signal according to aspects of the invention.

Figure 16 shows an embodiment of modulator with a variable capacitor placed at the point of maximum electric field intensity according to aspects of the invention. Figure 17 shows an embodiment of phase modulator in which variable capacitors are placed at different positions according to aspects of the invention.

Figure 18 shows a normalized plot of a phase change phase modulator caused by changing the capacitance according to aspects of the invention. Figure 19 shows a phase plot of Fano resonance for capacitances of IfF, 30 fF, and 75 fF according to aspects of the invention.

Figure 20 shows a magnitude plot of Fano resonance for capacitances of IfF, 30 fF^', and 75 fF according to aspects of the invention. Figure 21 show's a phase plot of EIT resonance for capacitances of IfF^", 30 fF^', and

75 fF according to aspects of the invention.

Figure 22 shows a magnitude plot of EIT resonance for capacitances of IfF, 30 fF, and 75 fF according to aspects of the invention.

Figure 23 shows a phase plot of Lorentz resonance for capacitances of IfF, 30 fF, and 75 fF according to aspects of the invention.

Figure 24 show's a magnitude plot of Lorentz resonance for capacitances of IfF, 30£F, and 75fF according to aspects of the invention.

Figure 25 shows the frequency response caused by changing the capacitance and placing capacitors at. different positions according to aspects of the invention. Figure 26A show's the phase and magnitude plots aligned in frequency for a narrowband EIT filter according to aspects of the invention.

Figure 26B shows the transmittance response of the signal caused by varying the frequency in an EIT filter according to aspects of the invention.

Figure 27 show's plots of phase versus frequency at varying capacitances according to aspects of the invention.

Figure 28 shows a multilayer Polar Modulator that simultaneously achieves both phase and amplitude modulations according to aspects of the invention.

Figure 29 show's the frequency response of a Polar Modulator at different values of capacitances according to aspects of the invention. Figure 30 show's an example of the polar modulator suitable for 5G systems and beyond according to aspects of the invention. Figure 31 is a plot of size versus frequency wherein the size of the polar modulator decreases as the operating frequency increases according to aspects of the invention.

Figure 32 shows an example of a Polar Modulator in which the substrate is selectively removed under the conductive layer of the modulator according to aspects of the invention.

Figure 33 shows a structure using an impedance transformer (tapper) to compensate for impedance mismatch caused by a change in relative permittivity of the substrate according to aspects of the invention.

Figure 34 show's the transmittance of the signal for 8i02 and Air by considering their dielectric loss according to aspects of the invention.

Figure 35 shows a guard ring that is used to isolate the modulator from the rest of on chip structures according to aspects of the invention.

DETAILED DESCRIPTION

The prior art is generally not seen to be suitable for use in mm-wave 5G/6G systems and beyond because of their power or area requirements. There is a need for a modulator that is of small size and is constructed using passive components. The expected number of antennas deployed on a single chip in mm-wave and THz MMIMO phased array systems ranges from 32 to 512 (see Figure 3). Inherently, the same number of modulators as antennas must also be populated on a single microchip. If the chip size becomes large, for example more than 30 x 30 mm2 in CMOS technology, the matching and reliability issues become more dominant, resulting in a decreased yield of microchips. Therefore, an innovative modulator design is proposed herein that can be fabricated using a passive on-chip structures and is suitable for fully integrated mm-wave 5G/6G Systems.

The hardware of modern mobile communication systems is miniature, low-power, and highly efficient. MMIMO technology is at the core of fifth and beyond generations of mobile communication systems. A typical MIMO mm-wave FDD RF front end architecture 301 is shown in Figure 3. These systems have several antennas 339. For each antenna they have LNAs (e.g., 309, 315), Switches (e.g., 319, 321, 323, 325, 327, 329, 331, 333), filters (341, 343) and Polar Modulators (e.g., 341, 343) and all of them are implemented on the same chip. The polar modulator has three input ports VA (305, 317), VO (307, 311) and Vc (303, 313). The output signal is both phase and amplitude modulated. The size of all components preferably must not be larger than the antenna's size at the top thick metal layer. Despite rapid innovations in the technology, the need for miniaturize radio components that are immune to high dielectric losses still exists and no satisfactory' solution is seen to be proposed in the prior art.

In pending U.S. Patent Application No. 17/198,712 (Ramzan et al.), it is shown that Fano, Lorentzian, and EIT transparency windows can be constructed in the Lorentzian absorption spectra. Designing a modulator, for on-chip implementation, by taking advantage of a transparency window in the Lorentzian absorption spectra would be beneficial.

AM modulation can be achieved by different methods. An embodiment of a fano AM modulator 400 according to aspects of the current invention is depicted in Figure 4. Fano AM modulator 400 belongs to the passive class of modulators and hence does not suffer from the issues mentioned above with respect to the active modulators. Fano AM modulator 400 comprises three-ports. Port-1 404 receives the input signal and Port-2406 provides the output signal. Fano AM modulator 400 produces attenuation of the input signal with respect, to the modulating voltage applied to Port-3 402. The size of split ring 408 is used to tune the frequency in the required band which can be defined at the time of manufacturing. The current produced by the voltage at Port-3 402 interferes with the signal on the microstrip path 412 resulting in an AM modulated signal as shown in Figure 5. The transmitted signal magnitude plot of fano AM Modulator 400 is shown in Figure 5. At a given frequency, an attenuation response (e.g., 506, 508, 510, 512, and 514) produced as a function of the applied voltage, achieving the AM effect. In this structure all ports are physically connected; therefore, the signal source on port-1 404 and the voltage source on Port-3 402 are not isolated from each other.

In Figure 6, an fano AM Modulator 600 is provided that isolates the voltage source 602 and signal source Port-1 604. The modulated signal is received at Port-2 612. The working principle of Fano AM modulator 600 in Figure 6 is similar to that of the Fano AM modulator 400 shown in Figure 4. The split ring resonator 614 tunes the modulator at the required frequency band. The internal resistance of the source 602 is shown as resistance 616. The mutual inductance 608 of the inductors 606 and 610 is exploited to produce the required attenuation, but unlike in Fano AM modulator 400 the sources of Fano AM modulator 600 are isolated from each other.

An alternate fano AM Modulator 700 is shown in Figure 7. In fano AM Modulator 700, the inductors 606 and 610 of fano AM Modulator 600 in Figure 6 are replaced with the coupled microstrip lines 704 and 708. The internal resistance of the source 702 is resistance 710. The resistance that a signal experiences while traveling from Port-1 706 to Port-2 712 is proportional to the change in the voltage Vss 702 at the microstrip 704. The attenuation of the signal received at Port-1 706 is shown in graph 800 of Figure 8, which show's its magnitude response for five different voltages (806, 808,

810, 812, and 814). As seen in Figure 8, the change is linearly proportional to the applied voltage. Figure 9 is a graph 900 of amplitude 902 versus voltage 904 winch shows that the ratio 906 between the applied voltage and attenuation for fano AM Modulator 700 of Figure 7 is linear, which is a desirable behavior that is currently seen to be impossible to achieve in active class modulators. It is evident from graph 900 of Figure 9 that the attenuation is around 0.5 (3-dB) when the applied voltage is 0 V, and the attenuation only increases with higher positive voltages which indicates a high-power loss as at least half of the signal is attenuated. This attribute is taken advantage of by applying a negative voltage to generate constructive interference. The signal passes through the modulator without any attenuation at a certain voltage level, as shown in graph 1000 of Figure 10. It achieves a transmittance of 1 dB at an applied voltage of -3V resulting in a significant gain. For an AM application, the negative voltages are produced by adding a DC bias to the signal of voltage source 702 of fano AM Modulator 700 of Figure 7 with a required signal headroom strength. An alternate solution is to increase the distance between the microstrip lines 704 and 708 by keeping in mind the direct, trade-off of reduced signal strength because of lower coupling. On the other hand, if the distance between the microstrip lines 704 and 708 is increased, a high potential signal is required to get the same transmittance as shown in the three different distance curves (1106, 1108, 1110) of graph 1100 of Figure 11.

The size and area of an AM modulator structure, such as fano AM Modulator 700 of Figure 7, is a key concern for implementing in IC technology. The size and area of fano AM Modulator 700 can further be reduced using a vertical integration option in the CMOS technology. A multilayered configuration of a Fano AM modulator 1200 according to aspects of the invention is shown in Figure 12. The microstrips 1206 and 1208 are in different layers. 1210 is the source which provides attenuation to the signal at Port-1 1202. The modulated signal is received at Port-2 1204. Such a multilayer structure provides two key advantages over the single-layer designs:

1) It requires a smaller area.

2) It provides for a larger overlap area, which provides larger attenuation with a lower voltage. Graph 1300 of Figure 13 validates that relatively more attenuation 1302 can be achieved with smaller applied voltage 1304.

In the multilayer structure, attenuation is a function of the separation between metal layers. A multilayer Fano AM modulator 1400 comprising three metal layers is shown in Figure 14, The Port-1 1402 and Port-2 1404 are the ports of the modulator. Layers 1406,1408 and 1410 represent different layers of the modulator. Source 1412 is the voltage source at layer 1406, source 1414 is the voltage source at layer 1408 and source 1416 is the voltage source at layer 1410, respectively, impact of the separation between the layers on the attenuation is shown in graph 1500 of Figure 15. The attenuation increases (1506) with an increase in the distance between the layers (1508),

Phase modulation can be achieved by changing the capacitance between the split edges of a ring oscillator. This is achieved by adding a variable capacitor 1610 in the split ring 1608 as shown in phase modulator 1600 of Figure 16, Port-1 1602 and Port-2 1604 are provided, along with microstrip line 1606. The variable capacitance can be implemented using any type of capacitor available in IC fabrication process. The electric field's intensity is maximum at the edges of the split ring 1608. The resonance frequency of the modulator changes with the capacitor in the following manner:

Where L is the loop inductance, and € is the capacitance. The resonance frequency shifts to the lower part of the band with an increase in capacitance. To achieve this, a variable capacitor 1610 is used as shown in Figure! 6. Moreover, a capacitor may possibly be added to split ring 1608 at any one of the three different locations, such as location A1712, location B 1714, and location C 1706 as shown in phase modulator 1700 of Figure 17. Fano, HIT, and Lorentz resonance phenomena can be achieved depending on the location of the capacitor. However, the capacitor must be used at only one location at the time of manufacturing and the other two locations will be filled with the metal so that the split ring resonator becomes a continuous structure. Port-1 1702 and Port-2 1708 are provided, along with microstrip line 1704. Capacitors with different values of 100 £F, 200 fF, and 500 £F, for example, are used in these areas of maximum electric field strength that result in a remarkable shift in the resonance frequency. This provides desired tunability to achieve Phase Modulation as shown in graph 1800 of Figure 18.

The normalized phase-shifted plot for PM Modulator 1600 of Figure 16 is shown in graph 1800 of Figure 18. When the capacitor is placed alternately at each of the three different locations 1706, 1712, and 1714 as shown in Figure 17, the effect of each combination of position at a variety of capacitor sizes is given in graphs 1900 to 2400 of Figures 19 to 24, respectively. Graph 1900 of Figure 19 shows the effect when the capacitor is placed at position 1712 of Figure 17. Fano resonance is achieved with phase change at IFF 1908, 30fF 1910 and 75fF 1912, as seen in graph 1900. A change in phase of 700 is evident for a change in the value of capacitance from 1 to75 fix The magnitude response for placing the capacitor at position 1712 is shown in graph 2000 of Figure 20. Highly selective resonance curves are achieved in the case of fano resonance. EGG resonance is achieved when the variable capacitor is placed at position 1714. Graph 2100 of Figure 2!shows the phase change achieved at IFF 2106, 30fF 2108, and 75fF 2110 when placing the capacitor at position 1714. The magnitude response for placing the capacitor at position 1714 is shown in graph 2200 of Figure 22. The magnitude show's the resonance with larger bandwidth as compared to the fano based resonance. Lorentzian resonance is achieved by placing the capacitor at position 1706. The phase and magnitude plot are shown in graph 2300 of Figure 23 and graph 2400 of Figure 24, respectively. The Lorentzian resonance achieved by placing the capacitor at position 1706 is less sensitive to the capacitor changes as compared to placing the capacitor at positions 1712 and 1714 because the electric field intensity when the capacitor is placed at position 1706 is weak as compared to when it is placed at position 1712 or position 1714.

As seen in graph 2500 of Figure 25, the plots 2508 and 2506 of frequency versus capacitance when placing the capacitor at positions 1712 and 1706, respectively, are relatively more linear than the plot for placing the capacitor at position 1714. If the capacitance is changed in small values, AC, plots of all positions will be linear albeit at the expense of a small change in the phase. This desirable behavior has not been achieved in the active class of on-chip modulators.

The frequency response shown in graphs 2000, 2200 and 2400 of Figure 20,

Figure 22, and Figure 24, respectively, is narrow band in nature and produces unwanted passband attenuation, which causes AM along with PM as shown in graph 2600 of Figure 26A. The passband attenuation 2618 shown is for the BIT filter, and the response shows unwanted AM with PM. Unwanted AM can he avoided by increasing the bandwidth 2605, as shown in graph 2601 of Figure 26B. The phase response 2700 is shown in the graph 2700 of Figure 27. The bandwidth can be increased by different methods. The first method is to change the overlapping area of the ring and the microstrip line, in standard CMOS IC technology, controlling small capacitances still remains a significant challenge. Varactors or MOS transistors can be used to construct the on-Chip variable capacitors. However, capacitors in the specific capacitance ranges can be manufactured with relatively tight tolerances. These problems are solved combining large value capacitors in series to achieve the desired value of capacitance. Another method is to add the capacitor at the area of less sensitivity 1706 (refer to Figure 17).

Polar Modulation is used to simultaneously modulate the phase and amplitude of a signal. Figure 28 shows a novel Polar modulator 2800 according to aspects of the invention. Polar modulator 2800 is created by combining Amplitude Modulator 1200 of Figure 12 and Phase Modulator 1600 of Figure 16. Amplitude modulation is achieved by changing the voltage levels of the microstrip under the transmission line between Port-1 2806 and Port-22812. Phase modulation is achieved by varying the capacitance inserted between the split ring resonator 2808. The voltage source 2802 is connected to the microstrip line 2804. The frequency response is shown in graph 2900 of Figure 29. The results shown in graph 2900 are taken at three different capacitance values (1 fF, 50 fly and 100 fF), with constant potentials of QV and -3V. As shown in graph 2900, the transmittance of the signal is reduced with an increase in the applied voltage. This is the desired behavior for a passive polar modulator in mm -wave frequency bands and makes it suitable for implementation in standard bulk CMOS technology.

An application scenario of the modulator described in graph 2900 is shown in modulator 3000 of Figure 30. N-QAM modulation is achieved by modulating both phase and amplitude of the carrier signal 3012, The polar modulator has a split ring resonator 3018, a microstrip line 3008, and a wider microstrip line 3002 placed below microstrip line 3008. PAs 3004 and 3016 amplify the amplitudes respective phase and amplitude modulating signals 3006 and 3014, respectively. The variable capacitor 3024 provides the phase modulation, and ground 3010 is connected to the microstrip line 3002. By increasing the voltage at source 3014, the phase of the carrier 3012 decreases. Similarly, by increasing the voltage at source 3006, the amplitude of the carrier 3012 decreases. Polar coordinate plot 3026 shows different combinations of phase and amplitude. The unmodulated carrier signal is applied at port 3012, which is matched with the microstrip line 3008. The amplitude and phase modulated signal is received at output 3010, which further passes through the bandpass filter 3020 and antenna 3022, The described modulator 3000 can modulate both amplitude and phase of the carrier signal, simultaneously. One skilled in the art would understand that the carrier signal may be either an intermediate frequency or the final carrier frequency.

Results of a scalability analysis are shown in graph 3100 of Figure 31, which demonstrates that the polar modulator would work at higher frequencies. It is a well- known fact in electromagnetics that any microwave structure with relatively small dimensions resonates at higher frequencies. Therefore, one skilled in the art would understand that even 300 GHz is not an upper limit, and the microstructures described can resonate at higher frequencies, including in the THz region by reducing their dimensions.

Due to low cost, low power, and better performance at high unity gain frequencies, CMOS technology has gained attention as the primary technology for mm- wave applications. The System on Chip (SoC) paradigm has many advantages, such as relaxed impedance matching criteria, small size, and improved SNR, Despite all these advantages, CMOS technology is still considered unsuitable for passive components due to high dielectric losses and low resistivity substrates. While these characteristics are suitable for active components since they reduce latch-up in transistors, the criteria for passive components is opposite. The lower thickness of interconnect layers causes the radiation to accumulate due to the substrate's lower resistivity. These factors increase the losses in passive structures and make them power hungry and inefficient. Many techniques have already been developed to reduce these dielectric losses, including Electromagnetic Bandgap (EBG), Reflective surfaces, Selective Removal of Dielectric,

reflections and can be resolved by making a microstrip tapper 3312 at. that location point 3306, as shown in Figure 33.

In 5G and beyond ICs, such as IC 3502 of Figure 35, the isolation from the adjacent components is also highly critical. The problem doubles in passive resonating structure, where minor intersections significantly affect the performance. The isolation is achieved by inserting the vias 3504 to form a guard ring as depicted in IC 3502 of Figure 35. The length of one side of the square via is l/2, and the distance 3506 between adjacent vias, for instance, at 28 GHz are computed using the following relation:

where X_c is the cut-off wavelength.

Those of skill in the art. will appreciate that the various examples, logical and functional blocks, components, devices, graphs, modules and units described in connection with the aspects disclosed herein can be implemented as hardware blocks inside an application specific integrated chip (ASIC) or in discrete blocks in 3D integrated circuits or in multichip modules or hybrids or reconfigurabie modules of software defined radios (implemented in any known technology). To clearly illustrate this interchangeability of hardware and functionality, various illustrative devices, components, blocks, modules, and/or steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software, or a combination thereof, depends upon the particular constraints imposed on the overall system and devices. Skilled persons can implement the described functionality in varying ways for each particular system, but such implementation decisions should not be interpreted as causing a departure from the scope of the invention as described herein. In addition, the grouping of functions, components or devices within a unit, module, block, or step is for ease of description. Specific functions, components or steps can be moved from one unit, module, or block without departing from the invention.

The above description of the disclosed aspects, and that provided in the accompanying documents, is provided to enable any person skilled in the art to make or use the invention. Various modifications to these aspects will be readily apparent to those skilled in the art, and the generic principles described herein, and in the accompanying documents, can be applied to other aspects without departing from the spirit or scope of the invention. Thus, it is to be understood that the description and drawings presented herein, and presented in the accompanying documents, represent particular aspects of the invention and are therefore representative examples of the subject matter that is broadly contemplated by the present invention. It is further understood that the scope of the present invention fully encompasses other aspects that are, or may become, understood to those skilled in the art based on the descriptions presented herein and that the scope of the present invention is accordingly not limited by the descriptions of aspects presented herein, or by the descriptions of aspects presented in the accompanying documents.

Claims

CLAIMS What we claim is:

1. An amplitude modulator for communication applications, the amplitude m odul ator cornpri si ng : a microstrip filter section having a rectangular shape, with a first connection port, provided at a first end of the rectangular shape and a second connection port provided at a second end of the rectangular shape, the first end being located opposite from the second end; a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape; and a dedicated voltage source structured and configured to apply a modulating voltage signal to the microstrip filter section; wherein the microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at one of a microwave frequency and a millimeter-wave (mm-wave) frequency, the split-ring microstrip filter possessing at least one of a Fano, BIT, and Lorentz resonance.

2. The amplitude modulator of Claim 1 wherein an operational frequency band of the amplitude modulator depends upon a physical size of the ring resonator filter section.

3. The amplitude modulator of Claim 1, further including a third connection port provided at a second side of the rectangular shape between the first end and the second end, third connection port being coupled to the dedicated voltage source, the second side of the rectangular shape being located opposite from the first side of the rectangular shape.

4. The amplitude modulator of Claim 1 wherein the amplitude modulator has a planar shape and is fabricated using one of a standard CMOS technology and a planar IC manufacturing technology.

5. The amplitude modulator of Claim 1, wherein the amplitude modulator has a linear structure.

6. The amplitude modulator of Claim 1, wherein the amplitude modulator consists of passive components and thereby operates in a power efficient manner.

7. The amplitude modulator of Claim 1 , wherein the amplitude modulator provides continuous and discrete amplitude modulation of an input signal.

8. The amplitude modulator of Claim 1, wherein the microstrip filter section, the ring resonator filter section and the dedicated voltage source are part of a single layer structure.

9. The amplitude modulator of Claim 1 , wherein the microstrip filter section, the ring resonator filter section and the dedicated voltage source are part multilayer layer structure, wherein the microstrip filter section and the ring resonator filter section are part of a first layer of the multilayer layer structure, the dedicated voltage source is part of a second layer of the multilayer layer structure that is different than the first layer.

10. The amplitude modulator of Claim 9, wherein the second layer further includes a microstrip line coupled to the dedicated voltage source.

11. The amplitude modulator of Claim 1, wherein the microstrip filter section and the ring resonator filter section are not electrically isolated from the dedicated voltage source.

12. The amplitude modulator of Claim 1, wherein the microstrip filter section and the ring resonator filter section are electrically isolated from the dedicated voltage source.

13. The amplitude modulator of Claim 12, wherein the dedicated voltage source is coupled to the microstrip filter section by mutual inductance of a first inductor coupled to the dedicated voltage source and a second inductor coupled to the microstrip filter section.

14. The amplitude modulator of Claim 12, rvherein the dedicated voltage source is coupled to the microstrip filter section through a microstrip line coupled to the dedicated voltage source.

15. The amplitude modulator of Claim 1 , wherein the split-ring microstrip filter is fabricated using a CMOS fabrication process or an IC fabrication process.

16. The amplitude modulator of Claim 15 wherein the IC fabrication process is one of a GaN IC fabrication process or a GaAs fabrication process.

17. The amplitude modulator of Claim 1, wherein the split-ring microstrip filter is fabricated by a non-pianar technology.

18. The amplitude modulator of Claim 17, wherein the non-planar technology is one of a MEMS technology and a 3D printing technology.

19. A phase modulator for communication applications, the phase modulator comprising: a microstrip filter section having a rectangular shape, with a first connection port provided at a first end of the rectangular shape and a second connection port provided at a second end of the rectangular shape, the first end being located opposite from the second end; a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape; and at least one variable capacitor disposed in the open space gap of the ring resonator filter section; wherein the microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at one of a microwave frequency, a millimeter-wave (mm-wave) frequency and a THz frequency, the split-ring microstrip filter possessing at least one of a Fano, EGT, and Lorentz resonance.

20. The phase modulator of Claim 19, wherein the at least one variable capacitor is also disposed in an open space gap provided in the microstrip filter section.

21. The phase modulator of Claim 19, wherein the variable capacitor provides a required phase shift to an input signal.

22. The phase modulator of Claim 19, wherein the phase modulator provides both continuous and discrete phase modulation to an input signal.

23. The phase modulator of Claim 19, wherein the phase modulator has a linear structure.

24. The phase modulator of Claim 19, wherein the split-ring microstrip filter is fabricated by a non-pfanar technology.

25. The phase modulator of Claim 24, wherein the non-planar technology is one of a

MEMS technology and a 3D printing technology.

26, A polar modulator for communication applications, the polar modulator comprising: a microstrip filter section having a rectangular shape, with a first connection port provided at a first end of the rectangular shape and a second connection port provided at a second end of the rectangular shape, the first end being located opposite from the second end; a ring resonator filter section having a circular shape, an outer edge of the ring resonator filter section being coupled to a first side of the rectangular shape of the microstrip filter section, the ring resonator filter section having an open space gap in the circular shape; at least one variable capacitor disposed in the open space gap of the ring resonator filter section; and a dedicated voltage source structed and configured to apply a modulating voltage signal to the microstrip filter section; wherein the microstrip filter section and the ring resonator filter section operate together in a combined manner as a split-ring microstrip filter that is tuned at a predetermined frequency band, the split-ring microstrip filter possessing at least one of a Fano, EIT, and Lorentz resonance.

27. The polar modulator of Claim 26, wherein an operational frequency band of the polar modulator depends upon a physical size of the ring resonator filter section.

28. The polar modulator of Claim 26, wherein the predetermined frequency band is a 5(3 millimeter wave (mm-wave) band of 28-32 GHz.

29. The polar modulator of Claim 26, wherein the variable capacitor is implemented using one or more varactors.

30. The polar modulator of Claim 26, wherein the variable capacitor is implemented using a CMOS transistor provided by a standard CMOS fabrication process.

31. The polar modulator of Claim 26, wherein the variable capacitor is implemented using a CMOS transistor provided by a GaN or a GaAs fabrication process.

32. The polar modulator of Claim 26, wherein the polar modulator has a planar shape and is manufactured by a standard CMOS technology.

33. The polar modulator of Claim 26, wherein the polar modulator has a linear structure in which the microstrip filter section is comprised of two microstrips placed close to each other.

34. The polar modulator of Claim 26, wherein the at least one variable capacitor provides a required phase shift to an input signal.

35. The polar modulator of Claim 26, wherein the polar modulator is comprised only of passive components and is thereby power efficient.

36. The polar modulator of Claim 26, wherein the polar modulator provides continuous and discrete amplitude modulation to an input signal.

37. The polar modulator of Claim 26, wherein the polar modulator is one of a single layer topology and a multilayer topology.

38. The polar modulator of Claim 26, wherein the split-ring microstrip filter is fabricated by a non-planar technology .

39. The polar modulator of Claim 38 wherein the non-planar technology is one of a MEMS technology and a 3D printing technology.