WO2022261995A1 - 一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法 - Google Patents

一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法 Download PDF

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Publication number
WO2022261995A1
WO2022261995A1 PCT/CN2021/101535 CN2021101535W WO2022261995A1 WO 2022261995 A1 WO2022261995 A1 WO 2022261995A1 CN 2021101535 W CN2021101535 W CN 2021101535W WO 2022261995 A1 WO2022261995 A1 WO 2022261995A1
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phase
fault
tolerant
coordinate system
permanent magnet
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PCT/CN2021/101535
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English (en)
French (fr)
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周华伟
江光耀
张多
陶炜国
陈前
田翔
毛彦欣
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江苏大学
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Priority to GB2219569.7A priority Critical patent/GB2610788B/en
Publication of WO2022261995A1 publication Critical patent/WO2022261995A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K41/00—Propulsion systems in which a rigid body is moved along a path due to dynamo-electric interaction between the body and a magnetic field travelling along the path
    • H02K41/02—Linear motors; Sectional motors
    • H02K41/03—Synchronous motors; Motors moving step by step; Reluctance motors
    • H02K41/031—Synchronous motors; Motors moving step by step; Reluctance motors of the permanent magnet type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K41/00—Propulsion systems in which a rigid body is moved along a path due to dynamo-electric interaction between the body and a magnetic field travelling along the path
    • H02K41/02—Linear motors; Sectional motors
    • H02K41/03—Synchronous motors; Motors moving step by step; Reluctance motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K21/00—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets
    • H02K21/12—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets
    • H02K21/14—Synchronous motors having permanent magnets; Synchronous generators having permanent magnets with stationary armatures and rotating magnets with magnets rotating within the armatures
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/12—Stationary parts of the magnetic circuit
    • H02K1/14—Stator cores with salient poles
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/22—Rotating parts of the magnetic circuit
    • H02K1/27—Rotor cores with permanent magnets
    • H02K1/2706—Inner rotors
    • H02K1/272—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis
    • H02K1/274—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets
    • H02K1/2753—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets the rotor consisting of magnets or groups of magnets arranged with alternating polarity
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/22—Rotating parts of the magnetic circuit
    • H02K1/27—Rotor cores with permanent magnets
    • H02K1/2706—Inner rotors
    • H02K1/272—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis
    • H02K1/274—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets
    • H02K1/2753—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets the rotor consisting of magnets or groups of magnets arranged with alternating polarity
    • H02K1/276—Magnets embedded in the magnetic core, e.g. interior permanent magnets [IPM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K3/00—Details of windings
    • H02K3/04—Windings characterised by the conductor shape, form or construction, e.g. with bar conductors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K3/00—Details of windings
    • H02K3/04—Windings characterised by the conductor shape, form or construction, e.g. with bar conductors
    • H02K3/28—Layout of windings or of connections between windings
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K7/00—Arrangements for handling mechanical energy structurally associated with dynamo-electric machines, e.g. structural association with mechanical driving motors or auxiliary dynamo-electric machines
    • H02K7/14—Structural association with mechanical loads, e.g. with hand-held machine tools or fans
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/04—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for very low speeds
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/141—Flux estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20—Estimation of torque
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/0243—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being a broken phase
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/028—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the motor continuing operation despite the fault condition, e.g. eliminating, compensating for or remedying the fault
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/06—Details of the magnetic circuit characterised by the shape, form or construction
    • H02K1/22—Rotating parts of the magnetic circuit
    • H02K1/27—Rotor cores with permanent magnets
    • H02K1/2706—Inner rotors
    • H02K1/272—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis
    • H02K1/274—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets
    • H02K1/2753—Inner rotors the magnetisation axis of the magnets being perpendicular to the rotor axis the rotor consisting of two or more circumferentially positioned magnets the rotor consisting of magnets or groups of magnets arranged with alternating polarity
    • H02K1/278—Surface mounted magnets; Inset magnets
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K2201/00—Specific aspects not provided for in the other groups of this subclass relating to the magnetic circuits
    • H02K2201/18—Machines moving with multiple degrees of freedom
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K29/00—Motors or generators having non-mechanical commutating devices, e.g. discharge tubes or semiconductor devices
    • H02K29/03—Motors or generators having non-mechanical commutating devices, e.g. discharge tubes or semiconductor devices with a magnetic circuit specially adapted for avoiding torque ripples or self-starting problems
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • H02P2207/055—Surface mounted magnet motors

Definitions

  • the invention relates to a linear rotation two-degree-of-freedom five-phase permanent magnet fault-tolerant actuator and a two-phase open-circuit fault-tolerant direct torque control method, which are suitable for the application occasion of the steering suspension integrated system and belong to the technical field of special motors.
  • two-degree-of-freedom drives have attracted attention in automotive steering and suspension integration systems.
  • the two-degree-of-freedom motor avoids the intermediate transmission structure, which can effectively improve the control accuracy and reduce the system volume, so it can better meet the requirements of the steering suspension drive system.
  • a linear rotary permanent magnet actuator is a two-degree-of-freedom motor with linear, rotary, and helical motion.
  • the literature Analysis of a double stator linear rotary permanent magnet motor with orthogonally arrayed permanent magnets proposes a two-degree-of-freedom permanent magnet motor with radially inner and outer double stator linear rotation.
  • the rotor core is thicker and heavier, and the torque and thrust pulsation is higher.
  • the structure of the magnetically coupled linear rotary permanent magnet motor is relatively compact, as shown in the literature Analytical Magnetic Field Analysis and Prediction of Cogging Force and Torque of a Linear and Rotary Permanent Magnet Actuator (IEEE Transactions on Magnetics,47(10):3004-3007,2011)
  • the structure proposed in but this structure has serious magnetic flux leakage, large torque and thrust pulsation, and no good fault tolerance performance. If a certain phase fails, it cannot operate normally, and cannot meet high reliability occasions.
  • the motor has a two-phase open-circuit fault, although it can still output a certain amount of torque, the torque fluctuates greatly, and the motor running noise and loss after the fault become larger, resulting in a decrease in the performance and service life of the motor, and even Cause permanent damage to the motor drive system.
  • the two-phase open-circuit fault occurs in the motor, the addition of the fault-tolerant algorithm can make the motor basically reach the steady state and dynamic performance before the fault, so that the motor after the fault can output smooth torque.
  • the former has obvious disadvantages such as complicated switch table making, unstable switching frequency, and large motor running noise; the latter has poor torque dynamic response, low torque control accuracy, neglect of cross-coupling items of control targets, and the application of multiple transformation matrices leads to Disadvantages such as complex structure.
  • the purpose of this patent is to design a new type of rotary linear permanent magnet actuator with fault-tolerant performance for steering suspension system and its two-phase open-circuit fault-tolerant DTC strategy.
  • the purpose of the present invention is to provide a linear rotation two-degree-of-freedom permanent magnet fault-tolerant actuator and its fault-tolerant direct torque control strategy in the case of two-phase open-circuit faults for the steering suspension integrated system.
  • the actuator can realize the decoupling of the rotating magnetic field and the traveling wave magnetic field in a single air gap, and then output torque and thrust, and overcome the shortcomings of the existing linear rotating two-degree-of-freedom permanent magnet motor with low fault tolerance and large torque ripple. , effectively reduce the amount of permanent magnets, reduce torque ripple, reduce magnetic flux leakage between poles, improve motor fault tolerance, and improve motor reliability.
  • the actuator can still output stable torque and thrust, and the dynamic performance and Normal is similar.
  • the actuator greatly reduces the volume of the traditional steering suspension integrated system, improves the control precision, and enhances the reliability of the steering suspension system.
  • an integrated five-phase permanent magnet fault-tolerant actuator for steering suspension which is used for linear, rotary and helical motion; includes a stator and a mover;
  • the stator is mainly composed of auxiliary teeth (35), armature teeth (32), fault-tolerant teeth (34), windings (21) of the linear motion part and windings (22) of the rotary motion part;
  • the stator is axially It consists of a pair of auxiliary teeth (35) including several alternating armature teeth (33) and fault-tolerant teeth (34); in the circumferential direction, several armature teeth (31) and fault-tolerant teeth (32) are alternately arranged
  • the end of the armature tooth (31) in the circumferential direction is a split tooth structure;
  • the armature tooth is provided with windings, and the windings are divided into the winding (21) of the linear motion part and the winding (21) of the rotary motion part ( 22);
  • the mover is made of permanent magnets (5),
  • the winding (21) of the linear motion part is wound into a circular shape and embedded in the bottom of the stator slot between the armature teeth (33) and the fault-tolerant teeth (34) in the axial direction, and the coils in two adjacent slots of the armature teeth are connected in series Winding in one phase, and filling the non-magnetically conductive ring (41) in the axial stator slot; the winding (22) of the rotary motion part is respectively wound on the armature teeth in the circumferential direction from one end to the other end.
  • the five-phase winding (21) of the rectilinear motion part and the five-phase winding (22) of the rotary motion part both adopt centralized winding.
  • the armature tooth (31) pole shoe part is a split tooth structure in the circumferential direction, and the split tooth structure can also use multiple teeth to reduce the magnetic flux leakage between the poles of the armature, but it is not a split tooth structure in the axial direction.
  • ; the number of pole pairs P 1 of the permanent magnet along the axial direction, the number of pole pairs of the armature winding P w1 of the linear motion part, and the number of modulation teeth N r1 along the axial direction, the three satisfy the relationship: P w1
  • the fault-tolerant direct torque control method under the condition of two-phase open-circuit fault includes the following steps:
  • Step 1 establish the model of the steering suspension integrated five-phase permanent magnet fault-tolerant actuator
  • Step 2 Two-phase open-circuit faults occur in the rotating part of the steering-suspension integrated five-phase permanent magnet fault-tolerant actuator. Assuming that non-adjacent two-phase open-circuit faults occur on phases B and E, according to the principle of equal magnetomotive force before and after the fault, Based on the principle that the sum of the non-fault phase currents is zero, the fault-tolerant currents i A BE , i B BE , i C BE , i D BE , and i E BE under the condition of B and E phase open-circuit faults are calculated, and the fault-tolerant currents are deduced respectively according to the fault-tolerant currents Fault-tolerant transformation matrix T BE under the condition of B and E phase open-circuit faults;
  • a two-phase open-circuit fault occurs in the rotating part of the steering-suspension integrated five-phase permanent magnet fault-tolerant actuator, assuming that the adjacent two-phase open-circuit fault occurs on C and D phases, according to the principle of equal magnetomotive force before and after the fault, Based on the principle that the sum of the non-fault phase currents is zero, the fault-tolerant currents i A CD , i B CD , i C CD , i D CD , and i E CD under the condition of C and D phase open-circuit faults are obtained; according to the fault-tolerant currents, C and the fault-tolerant transformation matrix T CD under the condition of D-phase open-circuit fault;
  • Step 3 using the traditional Clark transformation matrix to transform the stator flux linkage, voltage and current into the ⁇ coordinate system under the condition of two-phase open-circuit fault, then the stator flux linkage ⁇ ⁇ and ⁇ ⁇ on the ⁇ coordinate system can be expressed as
  • Step 4 Calculate the non-faulted phase voltages of phase B and E in the case of two-phase open-circuit faults according to the state of the inverter switch tube, and combine them with the faulted phase voltages to transform them into the ⁇ coordinate system by using the Clark transformation matrix
  • step 4 calculate the non-fault phase voltages under the condition of C and D phase open-circuit faults according to the state of the inverter switch tube, and combine them with the fault phase voltages, and use the Clark transformation matrix to transform them into the ⁇ coordinate system
  • Step 5 using the improved integrator to design the stator flux observer under the condition of two-phase open circuit fault. Furthermore, the torque under the condition of two-phase open-circuit fault is obtained.
  • Step 6 Based on the principle of stator magnetic field orientation, the voltage, current, and stator flux linkage on the ⁇ coordinate system are transformed to the MT coordinate system by using the T ⁇ -MT transformation matrix. Since the MT coordinate system is based on the orientation of the stator magnetic field, the stator flux observer on the ⁇ coordinate system is transformed into the MT coordinate system to construct a torque observer.
  • Step 7 the stator flux amplitude and torque are estimated by the stator flux observer and torque observer built in step 6, and they are compared with the given stator flux amplitude ⁇ * and given torque T* respectively
  • the given voltage command on the MT coordinate system is obtained through the PI controller with Inversely transform the voltage command to the voltage command on the ⁇ coordinate system through the T MT- ⁇ inverse transformation matrix in step 6 with Then, use the transformation matrix T BE or T CD in step 2 to transform it into the natural coordinate system, and get the phase voltage command If an open-circuit fault occurs on phases B and E, the phase voltage command is
  • phase voltage command is sent to the voltage source inverter, combined with the carrier pulse width modulation technology based on zero-sequence voltage injection to realize the integration of steering and suspension. Fault-tolerant bumpless direct torque control operation in case of C and D phase open circuit faults.
  • the two-phase open-circuit fault when an open-circuit fault occurs in the integrated five-phase permanent magnet fault-tolerant actuator of the steering suspension, it is assumed that a two-phase open-circuit fault occurs in the rotating winding of the actuator, and the two-phase open-circuit fault can be divided into adjacent phases and non-adjacent phases.
  • phase faults There are two types of phase faults. Assuming that non-adjacent two-phase faults occur on phase B and phase E, first derive the fault-tolerant current of the non-faulty phase after the non-adjacent phase B and E are open. According to the fault-tolerant current, the fault-tolerant transformation matrix for transforming the variables on the two-phase stationary coordinate system ( ⁇ coordinate system) to the natural coordinate system is obtained.
  • the stator flux linkage on the natural coordinate system is transformed to the ⁇ coordinate system by using the Clark transformation matrix.
  • stator flux observer and the torque observer are designed using the improved integrator on the ⁇ coordinate system.
  • stator flux linkage and torque observers in the ⁇ coordinate system are transformed to the MT coordinate system.
  • the stator flux amplitude and torque are estimated by the stator flux observer and torque observer on the MT coordinate system, and after making differences with the given stator flux and given torque, the PI controller is obtained
  • the given voltage command on the MT coordinate system is inversely transformed into a voltage command on the ⁇ coordinate system.
  • use the fault-tolerant transformation matrix to transform it to the natural coordinate system, and obtain the voltage command of the non-fault phase.
  • CPWM carrier pulse width modulation technology
  • the permanent magnet fault-tolerant actuator of the present invention realizes high integration of vehicle steering and suspension systems, greatly reduces the complexity of the original integrated system, and improves the accuracy, steady state and dynamic performance of system control.
  • the permanent magnet fault-tolerant actuator of the present invention uses two sets of windings on a stator to realize the linear and rotating armature magnetic fields, and the magnetic flux paths of the linear winding and the rotating winding are perpendicular to each other, which can achieve better linear motion and rotary motion Decoupled drive control.
  • the linear and rotating windings of the permanent magnet fault-tolerant actuator of the present invention are all wound in the adjacent slots of the armature teeth, and no coils are wound on the fault-tolerant teeth, thus physically isolating the phases of the actuator , thermal isolation and magnetic circuit decoupling, so as to achieve better fault tolerance performance and improve the reliability of the actuator.
  • Actuator windings adopt centralized winding, which is convenient for winding and the end winding is short, which can effectively reduce winding resistance and copper loss.
  • the circumferential direction of the permanent magnet fault-tolerant actuator of the present invention adopts a unique split armature tooth structure, which reduces the magnetic flux leakage between the poles of the armature, reduces the torque ripple, and improves the ability to output torque.
  • the circumferential and axial permanent magnets of the mover of the permanent magnet fault-tolerant actuator of the present invention are arranged in alternating poles, which reduces the amount of permanent magnets used, thereby reducing the cost of the actuator.
  • An air gap is added between the axial permanent magnet and the mover teeth to reduce the magnetic flux leakage between the poles of the permanent magnet.
  • the fault-tolerant DTC method of the present invention is different from the traditional fault-tolerant DTC method, and what the traditional fault-tolerant DTC adopted is to select the target voltage vector in the switch table by a hysteresis comparator.
  • There is a voltage discrimination error in the hysteresis comparator which leads to a large torque or thrust pulsation; at the same time, because the switch table query and sector discrimination involve the division of sectors, the calculation of trigonometric functions and irrational functions, the complexity of the program is greatly increased; and
  • the fault-tolerant DTC method of the present invention adopts the CPWM method based on the principle of stator magnetic field orientation and zero-sequence voltage signal injection, and can obtain the same effect as space vector pulse width modulation without distinguishing sectors and calculations, saving controller CPU memory resources, and effectively The calculation time of the CPU is reduced, and the torque ripple is greatly suppressed at the same time, and the torque control precision is improved.
  • the stator flux observer based on the improved integrator is designed, and it is used for the observation of the stator flux under the condition of two-phase open circuit fault, which not only improves the stator flux during the low-speed operation of the permanent magnet fault-tolerant actuator. It not only improves the sine degree of the chain waveform, but also improves the robustness of the stator flux linkage to parameters during high-speed operation, thereby enhancing the parameter robustness and anti-disturbance performance of fault-tolerant DTC operation.
  • the error-tolerant transformation matrix used to convert the variables in the natural coordinate system to the two-phase stationary coordinate system can be multiplied by the Clark transformation matrix to obtain the identity matrix, that is, whether it is under normal conditions or under fault-tolerant conditions, it is
  • the Clark transformation matrix is used to transform the variables sampled in the natural coordinate system to the ⁇ coordinate system
  • the fault-tolerant transformation matrix is used to transform the control command to the natural coordinate system only in the case of fault tolerance. Therefore, this strategy greatly reduces the complexity of the structural reconfiguration of the control system before and after the fault.
  • the fault-tolerant DTC method proposed by the present invention does not need to consider whether the actuator body adopts a fault-tolerant design scheme, and does not need to consider whether there is mutual inductance and coupling between the phase windings of the actuator.
  • the proposed fault-tolerant DTC method extends the application object from specially designed fault-tolerant permanent magnet motors to ordinary permanent magnet motors, which is more general and practical.
  • the fault-tolerant DTC method proposed by the present invention is based on stator magnetic field orientation, and the torque and stator flux linkage are decoupled on the MT coordinate system. Combining it with fault-tolerant DTC improves the torque and stator flux linkage under fault conditions Excellent dynamic performance and control accuracy further reduce torque or thrust ripple and simplify the design difficulty of fault-tolerant controllers. Therefore, compared with the vector control based on the rotor field orientation, it has the advantages of fast torque dynamic response, high torque or thrust control accuracy, and is also conducive to the realization of field weakening control; compared with the traditional DTC strategy, it has small torque ripple , high precision torque or thrust control.
  • the fault-tolerant DTC method of the present invention only needs to replace the transformation matrix from the two-phase static coordinate system to the natural coordinate system from the normal to the fault-tolerant control structure, and does not modify other places.
  • the derived voltage of the fault phase is only related to the resistance, leakage inductance, and permanent magnet amplitude of the non-fault phase where the ⁇ axis coincides.
  • the leakage inductance of the motor is very small, almost negligible, while the amplitude of the resistance and permanent magnet changes slightly, and the voltage change caused by it is compared with the phase voltage of the non-fault phase. is very small and can be ignored, so the changes of these three parameters will not affect the fault-tolerant DTC operation effect of the present invention.
  • the steering suspension integrated five-phase permanent magnet fault-tolerant actuator of the present invention can not only effectively suppress the fault after adopting the fault-tolerant DTC proposed in the present invention
  • the resulting torque and thrust pulsation can make the dynamic performance under fault conditions similar to that under normal conditions, and more importantly, the accuracy of torque and thrust control is improved. Therefore, combining the actuator with fault-tolerant DTC can effectively improve the steady-state and dynamic performance of the steering suspension system, and enhance the reliability of the steering suspension system.
  • Fig. 1 is a cross-sectional view of a steering suspension integrated five-phase permanent magnet fault-tolerant actuator according to an embodiment of the present invention
  • Fig. 2 is a front view of the steering suspension integrated five-phase permanent magnet fault-tolerant actuator according to the embodiment of the present invention
  • Fig. 3 is a side view cutaway view of the integrated five-phase permanent magnet fault-tolerant actuator of the steering suspension according to the embodiment of the present invention
  • Fig. 4 is the structure of split armature teeth, fault-tolerant teeth and axial stator slot non-magnetic material fillers according to the embodiment of the present invention
  • FIG. 5 is a schematic diagram of the fault-tolerant DTC strategy under the open-circuit fault of the rotating part B and E phases of the five-phase permanent magnet fault-tolerant actuator integrated with the steering suspension according to the embodiment of the present invention
  • Fig. 6 is the torque waveform of the rotating part B and E phases of the permanent magnet fault-tolerant actuator according to the embodiment of the present invention from normal to open-circuit fault conditions when there is no fault-tolerant DTC operation;
  • Fig. 7 is the torque waveform of the permanent magnet fault-tolerant actuator rotating part B and E phases when the fault-tolerant DTC runs from normal to open-circuit fault conditions;
  • Fig. 8 is the output torque waveform of the permanent magnet fault-tolerant actuator when the torque command is step-up during the fault-tolerant DTC operation under the open-circuit fault of phase B and E of the rotating part of the permanent magnet fault-tolerant actuator according to the embodiment of the present invention.
  • an embodiment of the present invention is an integrated five-phase permanent magnet fault-tolerant actuator for steering suspension.
  • Groove structure The stator of the actuator is composed of 10 pairs of split armature teeth and fault-tolerant teeth arranged alternately along the circumference; along the axial direction, it is composed of 10 pairs of armature teeth and 9 fault-tolerant teeth and a pair of auxiliary teeth located at their two ends
  • the armature teeth are provided with windings, and the windings are divided into windings for rotating the mover (referred to as rotating windings) and windings for moving the mover linearly (referred to as linear windings).
  • the linear winding is formed into a circular ring and embedded in the bottom of the stator slot between the axial armature teeth and the fault-tolerant teeth, and the coils in two adjacent slots of the armature teeth are connected in series to form a phase winding, and the linear winding is centralized winding;
  • the coils wound on the ten axial armature teeth are A1 phase, C1 phase, E1 phase, B1 phase, D1 phase, A2 phase, C2 phase, E2 phase, B2 phase, D2 phase, and the winding of each coil
  • the direction of the line is the same, and the A1 and A2 phases are connected in series in the forward direction to obtain the A phase, and the other four phases can be obtained in the same way.
  • the axial stator slot is filled with a non-magnetic ring made of a non-magnetic material such as aluminum or epoxy resin.
  • the rotating winding is wound on the armature teeth in the circumferential direction from one end to the other end.
  • the rotating winding is wound in a centralized manner, and the coils wound on the ten armature teeth in the circumferential direction are A1 phases in turn.
  • D1 phase, B1 phase, E1 phase, C1 phase, A2 phase, D2 phase, B2 phase, E2 phase, C2 phase, and the winding direction of each coil is the same, connect the A1 and A2 phases in forward series to get the A phase, other
  • the four phases can be obtained in the same way.
  • the mover is composed of permanent magnets, mover teeth, mover yokes and magnetically non-conductive rings, and the permanent magnets are embedded in mover slots.
  • the two-phase open-circuit fault can be divided into adjacent phase faults and non-adjacent phase faults. types. Assuming that non-adjacent two-phase faults occur on phase B and phase E, first derive the fault-tolerant current of the non-faulty phase after the non-adjacent phase B and E are open.
  • the rotating part of the permanent magnet fault-tolerant actuator adopts the direct torque control strategy shown in Fig. 5, and uses the Clark transformation matrix shown in formula (1) to transform the variable equal amplitude values on the five-phase natural coordinate system to the ⁇ - ⁇ coordinate system
  • phase currents of the five phases A, B, C, D, and E can be expressed as
  • i A , i B , i C , i D , and i E are the phase currents of A, B, C, D, and E phases respectively, and i ⁇ and i ⁇ are the components of the stator current on the ⁇ coordinate system, respectively.
  • the synthetic magnetomotive force MMF of the rotating winding of the permanent magnet fault-tolerant actuator is expressed as
  • N is the effective number of turns of the stator winding of each phase in the rotating part of the actuator.
  • phase B and E When an open-circuit fault occurs in phase B and E, the phase current is zero, so the magnetomotive force of the non-fault phase is zero
  • i A BE , i C BE , and i D BE are the phase fault currents of non-fault phases A, C, and D, respectively.
  • phase currents i A BE , i B BE , i C BE , i D BE , and i E BE after the open-circuit fault tolerance of phase B and E are calculated as
  • stator flux linkage ⁇ A BE , ⁇ B BE , ⁇ C BE , ⁇ D BE , ⁇ E BE of the rotating part of the actuator can be expressed as
  • L A , L B , L C , L D , LE are the inductances of A, B, C, D and E phases; L ls is the leakage inductance;
  • I 5 ⁇ 5 is the fifth order Identity matrix;
  • stator flux linkage and phase voltage on the natural coordinate system are transformed into the stator flux linkage on the ⁇ coordinate system by using the Clark matrix shown in formula (1).
  • Chain ⁇ ⁇ , ⁇ ⁇ and stator voltage u ⁇ , u ⁇ are
  • phase B and E Due to the existence of mutual inductance of non-fault relative to fault and the existence of back electromotive force of fault phase itself, when phase B and E are open circuited, the voltage of phase B and E can be expressed as
  • emf B emf E are the counter potentials of B and E phases.
  • m 1 L m i ⁇ - L ⁇ i ⁇ cos2 ⁇ - L ⁇ i ⁇ sin2 ⁇
  • m 2 L m i ⁇ + L ⁇ i ⁇ cos2 ⁇ - L ⁇ i ⁇ sin2 ⁇ .
  • phase voltage of the non-fault phase is represented by the state of the switch tube of the upper bridge arm of the voltage source inverter, and then combined with formula (12), the phase voltage in the fault state is transformed into the ⁇ coordinate system by using the Clark matrix above, can be expressed as
  • S a , S c , S d are the signals of the switch tubes of the upper bridge arm of the voltage source inverter A, C, and D phases.
  • the corresponding signal is equal to 1 when the switch tube is turned on, and the corresponding signal is 0 when it is turned off.
  • U dc is the DC bus Voltage, are the phase voltages of B and E phase open-circuit fault states, respectively;
  • the transformation matrix shown in equation (16) is used to transform the voltage, current, and stator flux linkage on the ⁇ coordinate system to the MT coordinate system. Transform the voltage model of formula (10) to the voltage u M , u T on the MT coordinate system, as shown in formula (18)
  • ⁇ s is the stator flux angle
  • ⁇ s is the stator flux angular velocity
  • i M , i T , ⁇ M , ⁇ T are the current and stator flux components on the MT coordinate system, respectively.
  • stator flux observer shown in formula (20) is constructed.
  • the direct decoupling control of torque and flux linkage can be realized by using two PI controllers.
  • the stator flux amplitude and torque are estimated by the stator flux observer shown in formula (20) and the torque observer shown in formula (21), and they are compared with the given torque T * , given Stator flux linkage
  • the given voltage command on the MT coordinate system is obtained through the PI controller.
  • the voltage command is reverse-transformed to the voltage command on the ⁇ coordinate system through the inverse transformation matrix T MT- ⁇ shown in formula (17).
  • the fault-tolerant transformation matrix shown in formula (7) is used to transform it into the natural coordinate system, and the voltage command of the non-fault phase is obtained, and the voltage command of the fault phase is 0.
  • the disturbance-free high-performance operation of the rotating part of the permanent magnet fault-tolerant actuator after B and E phase open-circuit faults is realized.
  • This fault-tolerant strategy not only has the characteristics of fast torque dynamic response and simple structure of DTC, but also can effectively suppress the torque ripple caused by the open-circuit fault of phase B and E, and realize the stable operation of DTC. The more important thing is the steady-state performance and vector control quite.
  • the fault-tolerant transformation matrix for transforming the variables on the ⁇ coordinate system to the natural coordinate system can be obtained as
  • the non-faulted phase voltage and the faulted phase voltage based on the switch state signal of the upper bridge arm of the voltage source inverter can be expressed in the ⁇ coordinate system as
  • S a , S b , S e are the signals of the switch tubes of the upper bridge arms of the voltage source inverters A, B, and E; are the phase voltages of C and D phases under open-circuit fault conditions, respectively.
  • stator flux linkage and torque observers shown in equations (20) and (21) on the MT coordinate system can be derived according to the same method as above.
  • the present invention takes the open-circuit faults of rotating parts B and E of the permanent magnet fault-tolerant actuator as an example to simulate and analyze the performance of the proposed fault-tolerant DTC strategy.
  • the block diagram of the control strategy is shown in Figure 5 . Compared with normal operation, the control structure is almost unchanged, so the proposed fault tolerance strategy is concise and efficient.
  • Figure 6 shows the torque waveforms when the rotating part of the actuator operates without fault tolerance when an open-circuit fault occurs in phase B and E at 0.2s and enters an open-circuit fault from normal. It can be seen that the torque fluctuation of the rotating part of the actuator is obvious.
  • FIG. 7 is the torque waveform of the rotating part of the actuator in fault-tolerant operation when the B and E phases are from normal to open-circuit faults.
  • the fault-tolerant DTC strategy of the present invention is started immediately. It can be seen that compared with the fault condition, the output torque ripple of the actuator is suppressed obviously, and there is almost no ripple.
  • Fig. 8 is the output torque waveform of the actuator when the torque command steps down during the operation of the B and E phase open-circuit fault-tolerant DTC, and the response time is about 1ms. It can be seen that the fault-tolerant DTC strategy proposed by the present invention maintains the advantage of rapid torque response of DTC.
  • the output torque has almost no fluctuation, the sine degree of the phase current is better, and the actuator has a dynamic performance similar to that under normal conditions after adopting the fault-tolerant DTC strategy of the present invention.
  • the steering suspension integrated five-phase permanent magnet fault-tolerant actuator and its two-phase open-circuit fault-tolerant DTC method of the present invention not only realize the rotary motion, linear motion, spiral motion, rotary motion and
  • the decoupling of linear motion can ensure that the output torque or thrust of the actuator is consistent with the normal situation in the case of two-phase open-circuit faults, and can obviously suppress the torque or thrust pulsation caused by two-phase open-circuit faults; more importantly, , the dynamic performance during fault-tolerant operation is similar to that of DTC under normal conditions, and the steady-state performance is similar to that of vector control, with strong versatility, no need for complex calculations, and low CPU overhead. Therefore, the present invention has good application prospects in automobiles, aerospace and other systems that require high operating performance.
  • references to the terms “one embodiment,” “some embodiments,” “exemplary embodiments,” “example,” “specific examples,” or “some examples” are intended to mean that the implementation A specific feature, structure, material, or characteristic described by an embodiment or example is included in at least one embodiment or example of the present invention.
  • schematic representations of the above terms do not necessarily refer to the same embodiment or example.
  • the specific features, structures, materials or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.

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Abstract

一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法，包括建立直线和旋转双自由度单气隙作动器模型；首先，在两相开路故障下推导出将两相静止坐标系上的变量变换到αβ坐标系上的容错变换矩阵；在αβ坐标系上基于改进积分器设计定子磁链和转矩观测器，并将其变换到基于定子磁链定向的坐标系上；其次，将定子磁链幅值和转矩指令分别与观测出的定子磁链和转矩作差，并经PI控制器获得电压指令；然后，采用容错变换矩阵该电压指令经变换到自然坐标系并送电压源逆变器控制作动器高性能运行。不但实现了车辆转向和悬架的一体化集成驱动，提高了其动态性能，而且更为关键的是增强了转向悬架***的可靠性。

Description

一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法 技术领域
本发明涉及一种直线旋转两自由度五相永磁容错作动器及其两相开路容错直接转矩控制方法，适用于转向悬架集成***的应用场合，属于特种电机技术领域。
背景技术
随着现代工业驱动***发展，一方面增加了***集成度，另一方面对驱动控制的精度要求不断提高，因此两自由度驱动在汽车转向和悬架集成***中得到关注。两台单自由度电机通过机械传动组合机构实现两维驱动存在诸多问题，如控制精度低、***体积大等。两自由度电机避免了中间的传动结构，能有效提高控制精度和减小***体积，因此能更好的满足转向悬架驱动***的要求。
直线旋转永磁作动器是一种具备直线、旋转和螺旋运动的两自由度电机。经过多年研究发展，直线旋转两自由度电机的研究取得一些阶段性成果。许多新型直线旋转两自由度电机的拓扑结构不断提出，如螺旋式结构、组合式结构和磁耦合式结构等。文献Analysis of a double stator linear rotary permanent magnet motor with orthogonally arrayed permanent magnets(IEEE Transactions on Magnetics，52(7):8203104,2016)提出一种径向内外双定子直线旋转两自由度永磁电机，该电机动子铁芯较厚、重量较重，转矩和推力脉动较高。磁耦合式直线旋转永磁电机结构较紧凑，如文献Analytical Magnetic Field Analysis and Prediction of Cogging Force and Torque of a Linear and Rotary Permanent Magnet Actuator(IEEE Transactions on Magnetics,47(10):3004-3007,2011)中提出的结构，但这种结构漏磁较严重，转矩和推力脉动较大，没有较好的容错性能，某一相发生故障，就不能正常运行，无法满足高可靠性的场合。
当电机发生两相开路故障时，虽然仍然能输出一定大小的转矩，但转矩存在很大的波动，而且故障后的电机运行噪声、损耗都变大，导致电机性能以及使用寿命下降，甚至导致电机驱动***永久损坏。而当电机发生两相开路故障时，容错算法的加入能够使电机基本达到故障前的稳态和动态性能，使得故障后的电机输出平滑转矩。文献中国电机工程学报36(1):231-239,2016“基于虚拟变量的六相永磁同步电机缺任意两相容错型直接转矩控制”针对六相永磁电机两相故障，提出一种基于多个虚拟变量的容错直接转矩控制(DTC)，该方法沿用传统开关表与滞环比较器的组合，这引入了开关表制作复杂、开关频率不稳定、电机运行噪声大等缺点，同时多个虚拟变量的设置，减弱了DTC简洁高效的优势。目前，针对多相电机发生两相开路故障的容错方法主要思想是以DTC和矢量控制为基础。前者存在开关表制作复杂、开关频率不稳定、电机运行噪声大等显著缺点；后者存在转矩动态响应不佳、转矩控制精度低、 忽略控制目标的交叉耦合项、应用多个变换矩阵导致结构复杂等缺点。
本专利的目的就是设计一种用于转向悬架***的具有容错性能的新型的旋转直线永磁作动器及其两相开路容错DTC策略。
发明内容
本发明的目的是提供用于转向悬架集成***的一种直线旋转两自由度永磁容错作动器及其两相开路故障情况下的容错直接转矩控制策略。该作动器在单气隙中能实现旋转磁场和行波磁场的解耦，进而输出转矩和推力，并克服已有直线旋转两自由度永磁电机容错性能低、转矩脉动大的缺点，有效降低永磁体用量，降低转矩脉动，减小极间漏磁、提高电机容错性能，提高电机可靠性。在作动器发生相邻或不相邻两相开路故障的情况下，采用本发明提出的容错DTC策略后，作动器仍然能输出平稳的转矩和推力，且故障情况下的动态性能和正常情况下相似。该作动器极大减小了传统转向悬架集成***的体积，提高了控制精度，增强了转向悬架***的可靠性。
本发明的永磁容错作动器的技术方案是：一种转向悬架一体化五相永磁容错作动器，该作动器用于直线、旋转和螺旋运动；包括定子、动子；所述定子主要由辅助齿(35)、电枢齿(32)、容错齿(34)、直线运动部分的绕组(21)和旋转运动部分的绕组(22)构成；所述定子在沿轴向上是由一对辅助齿(35)包含若干个交替的电枢齿(33)与容错齿(34)排列而成；在圆周向上是由若干个电枢齿(31)与容错齿(32)交替排列而成，在圆周方向上的电枢齿(31)端部为***齿结构；所述电枢齿上设置绕组，所述绕组分为直线运动部分的绕组(21)和旋转运动部分的绕组(22)；所述动子由永磁体(5)、动子齿(6)、动子轭(7)以及不导磁圆环(42)构成；所述永磁体(5)在圆周方向是瓦片状的且径向充磁、内嵌在动子齿(6)之间，沿圆周向分布的角度圆心角为tc＝(0.6～1)*360°/P，P为永磁体沿圆周向的极对数，在圆周向按交替极阵列排列；永磁体(5)沿轴向分布的长度为t l＝(0.25～0.5)*τ，τ为轴向动子极距，在轴向相邻两永磁体的充磁方向相反，按错位交替阵列排列，错位角度为t z＝180°/P；所述永磁体(5)与动子齿(6)在轴向上两两之间设置的气隙长度为t 0＝0.5*τ-t l；所述气隙可采用不导磁圆环(42)填充；所述动子齿和永磁体在圆周面上是等高的；所述辅助齿(35)轴向宽度为t l＝(0.25～0.5)*τ。
进一步，所述直线运动部分的绕组(21)绕制成圆环状嵌入轴向上电枢齿(33)与容错齿(34)之间的定子槽底部，电枢齿相邻两槽线圈串联成一相绕组，并且在轴向的定子槽内填充不导磁圆环(41)；所述旋转运动部分的绕组(22)分别从一边端部到另一边端部绕制在圆周向电枢齿(31)上；直线运动 部分的五相绕组(21)与旋转运动部分的五相绕组(22)均采用集中式绕制。
进一步，所述电枢齿(31)极靴部分在圆周方向是***齿结构，该***齿结构还可采用多齿来降低电枢极间漏磁，然而其在轴向方向不是***齿结构。
进一步，永磁体(5)沿圆周向的极对数P，旋转运动部分的电枢绕组极对数P w，沿圆周向的调制齿极数N r，三者满足关系：P w＝|N r-P|；永磁体沿轴向的极对数P 1，直线运动部分的电枢绕组极对数P w1，沿轴向的调制齿极数N r1，三者满足关系：P w1＝|N r1-P 1|。
本发明的一种由权利要求1所述的转向悬架一体化五相永磁容错作动器的两相开路容错直接转矩控制方法，当作动器旋转部分和直线部分的相数分别为m＝5时，可分别分为A、B、C、D、E五相，两相开路故障情况下的容错直接转矩控制方法包括如下步骤：
步骤1，建立转向悬架一体化五相永磁容错作动器的模型；
步骤2，转向悬架一体化五相永磁容错作动器旋转部分绕组发生两相开路故障，假设不相邻两相开路故障发生在B和E相上，根据故障前后磁动势相等原理、非故障相电流和为零的原则，求出B和E相开路故障情况下的容错电流i A BE、i B BE、i C BE、i D BE、i E BE，根据该容错电流分别推导出B和E相开路故障情况下的容错变换矩阵T BE；
或步骤2，转向悬架一体化五相永磁容错作动器旋转部分绕组发生两相开路故障，假设相邻两相开路故障发生在C和D相上，根据故障前后磁动势相等原理、非故障相电流和为零的原则，求出C和D相开路故障情况下的容错电流i A CD、i B CD、i C CD、i D CD、i E CD；根据该容错电流分别推导C和D相开路故障情况下的容错变换矩阵T CD；
步骤3，采用传统Clark变换矩阵将两相开路故障情况下定子磁链、电压和电流变换到αβ坐标系上，则αβ坐标系上的定子磁链ψ α、ψ β可表示为
Figure PCTCN2021101535-appb-000001
式中：i α、i β分别是定子电流在αβ坐标系上的分；θ为电角度；L m＝0.2(L d+L q)，L θ＝0.2(L q-L d)，L d、L q分别为旋转部分绕组的d轴和q轴电感；L ls为漏感；ψ pm为永磁磁链幅值；
步骤4，根据逆变器开关管状态计算出B相和E两相开路故障情况下的非故障相电压，并结合故障相电压，采用Clark变换矩阵将它们变换到αβ坐标系上
Figure PCTCN2021101535-appb-000002
或步骤4，根据逆变器开关管状态计算出C和D相开路故障情况下的非故障相电压，并结合故障相 电压，采用Clark变换矩阵将它们变换到αβ坐标系上
Figure PCTCN2021101535-appb-000003
步骤5，采用改进型积分器设计两相开路故障情况下的定子磁链观测器。进而求出两相开路故障情况下的转矩。
步骤6，基于定子磁场定向原则，采用T αβ-MT变换矩阵将αβ坐标系上的电压、电流、定子磁链变换到MT坐标系上。由于MT坐标系是基于定子磁场定向的，将αβ坐标系上的定子磁链观测器变换到MT该坐标系上，构建的转矩观测器。
步骤7，通过步骤6构建的定子磁链观测器和转矩观测器估算出定子磁链幅值和转矩，并将其与给定定子磁链幅值ψ*和给定转矩T*分别作差后，经PI控制器得到MT坐标系上的给定电压指令
Figure PCTCN2021101535-appb-000004
和
Figure PCTCN2021101535-appb-000005
通过步骤6中T MT-αβ反变换矩阵将该电压指令反变换至αβ坐标系上的电压指令
Figure PCTCN2021101535-appb-000006
和
Figure PCTCN2021101535-appb-000007
然后，采用步骤2中的变换矩阵T BE或T CD将其变换到自然坐标系上，得到相电压指令
Figure PCTCN2021101535-appb-000008
若B和E相发生开路故障，则相电压指令为
Figure PCTCN2021101535-appb-000009
若C和D相发生开路故障，则相电压指令为
Figure PCTCN2021101535-appb-000010
将该相电压指令送给电压源逆变器，再结合基于零序电压注入的载波脉宽调制技术实现转向悬架一体化五相永磁容错作动器旋转部分在B和E相开路故障或C和D相开路故障情况下的容错无扰直接转矩控制运行。
本发明中：当转向悬架一体化五相永磁容错作动器发生开路故障时，假设该作动器旋转绕组发生两相开路故障，两相开路故障可分为相邻相和不相邻相故障两种类型。假设不相邻两相故障发生在B相和E相上，先推导不相邻的B相和E相开路后非故障相的容错电流。根据容错电流求出将两相静止坐标系 (αβ坐标系)上的变量变换到自然坐标系上的容错变换矩阵。
采用Clark变换矩阵将自然坐标系上的定子磁链变换到αβ坐标系上。
由于非故障相对故障相互感的存在以及故障相自身反电势的存在，当B和E相发生开路后，求出B和E相的电压。在此基础上，采用Clark变换矩阵将自然坐标系上的相电压变换到αβ坐标系上。
在此基础上，根据定子磁链在故障情况下的电压和电流模型，在αβ坐标系上采用改进的积分器设计了定子磁链观测器和转矩观测器。基于定子磁场定向原则，将αβ坐标系上定子磁链和转矩观测器变换到MT坐标系上。
通过MT坐标系上的定子磁链观测器与转矩观测器估算定子磁链幅值和转矩，并将其与给定定子磁链、给定转矩分别作差后，经PI控制器得到MT坐标系上的给定电压指令。将该电压指令反变换至αβ坐标系上的电压指令。再采用容错变换矩阵将其变换到自然坐标系上，得到非故障相的电压指令。结合基于零序电压注入的载波脉宽调制技术(CPWM)实现永磁容错作动器旋转部分在B和E相开路故障后的无扰DTC高性能运行。
本发明的一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法，具有以下有益效果：
1)本发明的永磁容错作动器实现了车辆转向和悬架***的高度集成，大大降低了原有集成***的复杂性，提高了***控制的精度、稳态和动态性能。
2)本发明的永磁容错作动器在一个定子上采用两套绕组实现直线和旋转电枢磁场，且直线绕组和旋转绕组磁通路径相互垂直，能够实现较好的直线运动和旋转运动的解耦驱动控制。
3)本发明的永磁容错作动器的直线和旋转绕组都绕制在电枢齿相邻的槽内，容错齿上不绕线圈，起到了对作动器相与相之间的物理隔离、热隔离和磁路解耦的作用，从而达到较好的容错性能，提高了作动器的可靠性。作动器绕组均采用集中式绕制，绕线方便且端部绕组较短，可以有效减少绕组电阻和铜耗。
4)本发明的永磁容错作动器的圆周向采用独特的***电枢齿结构，降低了电枢极间漏磁，降低了转矩脉动，提高了输出转矩的能力。
5)本发明的永磁容错作动器的动子在圆周向和轴向上的永磁体均采用交替极方式排列，减小了永磁体用量，从而降低了作动器的成本。在轴向永磁体与动子齿两两之间增加气隙，降低了永磁体极间漏磁。
6)本发明的永磁容错作动器的动子与定子之间只有一层气隙，相比传统的双自由度的电机，该类作动器的结构大大简化，降低了安装的难度。
7)本发明的容错DTC方法不同于传统的容错DTC方法，传统的容错DTC采用的是通过滞环比较 器来选择开关表中的目标电压矢量。滞环比较器存在电压判别误差，导致较大的转矩或推力脉动；同时由于开关表查询和扇区判别涉及扇区的划分、三角函数和无理函数的计算，大大增加程序的复杂性；而本发明的容错DTC方法采用基于定子磁场定向原则和基于零序电压信号注入的CPWM方法，无需判别扇区和计算就能获得空间矢量脉宽调制相同的效果，节省了控制器CPU内存资源，有效减小了CPU的计算时间，同时大大抑制了转矩脉动，提高了转矩控制精度。
8)本发明中设计了基于改进积分器的定子磁链观测器，并将其用于两相开路故障情况下的定子磁链观测，不但改善了永磁容错作动器低速运行时的定子磁链波形的正弦度，而且提高了其高速运行时定子磁链对参数的鲁棒性，进而增强了容错DTC运行的参数鲁棒性和抗扰性能。
9)本发明中用于将自然坐标系中变量转换到两相静止坐标系的容错变换矩阵与Clark变换矩阵相乘能得到单位矩阵，也就是无论是在正常情况下还是容错情况下，都是采用Clark变换矩阵将在自然坐标系上采样的变量变换到αβ坐标系上，仅在容错情况下才采用容错变换矩阵将控制指令变换到自然坐标系上。因此，该策略极大的降低了故障前后控制***结构重构的复杂性。
10)本发明所提出的容错DTC方法无需考虑作动器本体是否采用容错设计方案，无需考虑作动器相绕组之间是否存在互感以及耦合情况。相比传统容错DTC方法，所提出的容错DTC方法将应用对象从特殊设计的容错永磁电机推广到普通的永磁电机，更具一般性和实用价值。
11)本发明所提出的容错DTC方法是基于定子磁场定向的，转矩和定子磁链在MT坐标系上是解耦的，将其与容错DTC结合改善了故障情况下转矩和定子磁链的动态性能和控制精度，进一步降低了转矩或推力脉动，简化了容错控制器设计难度。因此，与基于转子磁场定向的矢量控制相比，具有转矩动态响应快、转矩或推力控制精度高等优点，同时也有利于实现弱磁控制；与传统DTC策略相比，具有转矩脉动小、转矩或推力控制精度高等优点。
12)本发明的容错DTC方法从正常到容错的控制结构只需要更换从两相静止坐标系到自然坐标系的变换矩阵，其他地方不做修改。故障相的衍生电压只与α轴重合的非故障相的电阻、漏感、永磁体幅值有关。在恒转矩区域运行时，电机漏感值很小，几乎可以忽略不计，而电阻、永磁体幅值变化幅度较小，其引起的电压变化部分相比于非故障相的相电压，其值很小，可以忽略不计，因此这三个参数的变化不会影响本发明的容错DTC运行效果。
13)本发明的转向悬架一体化五相永磁容错作动器在发生不相邻或相邻两相开路故障的情况下，采用本发明中所提出的容错DTC后，不但能有效抑制故障导致的转矩和推力脉动，而且能使故障情况下的动态性能和正常情况下相似，更为关键的是提高了转矩和推力控制的精度。因此，将该作动器和容错DTC相结合能有效提高转向悬架***的稳态和动态性能，增强转向悬架***的可靠性。
附图说明
图1为本发明实施例转向悬架一体化五相永磁容错作动器剖视图；
图2为本发明实施例转向悬架一体化五相永磁容错作动器正视图；
图3为本发明实施例转向悬架一体化五相永磁容错作动器侧视剖分图；
图4为本发明实施例***电枢齿、容错齿以及轴向定子槽不导磁材料填充物结构；
图5为本发明实施例转向悬架一体化五相永磁容错作动器旋转部分B和E相开路故障下容错DTC策略原理图；
图6为本发明实施例永磁容错作动器旋转部分B和E相从正常到开路故障情况下无容错DTC运行时的转矩波形；
图7为本发明实施例永磁容错作动器旋转部分B和E相从正常到开路故障情况下容错DTC运行时的转矩波形；
图8为本发明实施例永磁容错作动器旋转部分B和E相开路故障下容错DTC运行过程中转矩指令阶跃上升时的作动器输出转矩波形。
图中：1为定子；2为绕组，21为直线运动部分的绕组，22为旋转运动部分的绕组；3为定子齿，31为圆周向电枢齿，32为圆周向容错齿，33为轴向电枢齿，34为轴向容错齿，35为辅助齿；4为不导磁材料填充物，41为轴向定子槽内的不导磁圆环，42为动子齿与永磁体在轴向上两两之间的不导磁圆环；5为永磁体；6为动子齿；7为动子轭。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述。
为了能够更加简单明了地说明本发明的转向悬架一体化永磁容错作动器及其两相开路容错直接转矩控制方法的特点和有益效果，下面结合一个具体的转向悬架一体化永磁容错作动器进行详细表述。
建立转向悬架一体化永磁容错作动器模型；
如图1-4所示为本发明的一个实施例，转向悬架一体化五相永磁容错作动器，该实施例直线运动部分采用18极20槽结构，旋转运动部分采用21对极20槽结构。所述作动器的定子沿圆周向由10对***电枢齿与容错齿交替排列而成；沿轴向由交替的10个电枢齿与9个容错齿以及位于他们两端的一对辅助齿排列而成，所述辅助齿轴向宽度为t 1＝5.3mm；所述轴向定子槽内除放置直线绕组外还用不导磁材料如铝或环氧树脂等材料制成的不导磁圆环填充。
所述电枢齿上设置绕组，所述绕组分为使动子发生旋转运动的绕组(简称旋转绕组)和使动子发 生直线运动的绕组(简称直线绕组)。所述直线绕组绕制成圆环状嵌入轴向的电枢齿与容错齿之间的定子槽底部，电枢齿相邻两槽线圈串联成一相绕组，所述直线绕组为集中式绕制；轴向的十个电枢齿上绕制的线圈依次为A1相，C1相，E1相，B1相，D1相，A2相，C2相，E2相，B2相，D2相，而且各个线圈的绕线方向一致，将A1、A2相正向串联得到A相，其他四相利用同样方式可得。在轴向的定子槽内填充不导磁材料如铝或环氧树脂等不导磁材料制成的不导磁圆环。所述旋转绕组从一边端部到另一边端部绕制在圆周向电枢齿上，所述旋转绕组为集中式绕制，圆周向的十个电枢齿上绕制的线圈依次为A1相，D1相，B1相，E1相，C1相，A2相，D2相，B2相，E2相，C2相，而且各个线圈的绕线方向一致，将A1和A2相正向串联得到A相，其他四相利用同样方式可得。
所述动子由永磁体、动子齿、动子轭以及不导磁环构成，所述永磁体嵌入在动子槽内。沿圆周向的永磁体极对数为21，均为径向且同方向充磁；沿轴向的永磁体极对数为9，在径向上的充磁方向是错位交替排列，错位角度为t z＝8.57°。永磁体沿圆周向分布的角度圆心角t c＝10.84°；沿轴向分布的长度t l＝8mm；所述永磁体与动子齿在轴向上两两之间设置的气隙长度t 0＝2mm，所述气隙采用不导磁材料如铝或环氧树脂等不导磁材料制成的不导磁圆环填充；永磁体均采用径向充磁。
当转向悬架一体化五相永磁容错作动器发生开路故障时，假设该作动器旋转绕组部分发生两相开路故障，两相开路故障可分为相邻相和不相邻相故障两种类型。假设不相邻两相故障发生在B相和E相上，先推导不相邻的B相和E相开路后非故障相的容错电流。
永磁容错作动器旋转部分采用图5所示直接转矩控制策略，采用式(1)所示Clark变换矩阵将五相自然坐标系上的变量等幅值变换到α-β坐标系上
Figure PCTCN2021101535-appb-000011
式中，a＝2π/5。
当作动器在正常情况下稳态运行时，假设其三维空间电流i x和i y已经控制为零，则A、B、C、D、E五相的相电流可表示为
Figure PCTCN2021101535-appb-000012
式中：i A、i B、i C、i D、i E分别为A、B、C、D、E相的相电流，i α、i β分别是定子电流在αβ坐标系上的分量。
永磁容错作动器旋转绕组的合成磁动势MMF表示为
Figure PCTCN2021101535-appb-000013
式中：ε＝e j2π/5，N为作动器旋转部分各相定子绕组的有效匝数。
当B和E相发生开路故障时，其相电流为零，故非故障相合成的磁动势为
Figure PCTCN2021101535-appb-000014
式中：i A BE、i C BE、i D BE分别为非故障相A、C、D的相容错电流。
为了保证作动器旋转部分故障后能继续无扰运行，需要保持故障前后合成磁动势的幅值与速度相等，即令式(3)和(4)相等。由于旋转绕组采用星形连接且中心点与直流母线中点不连接，故非故障相电流和为零
Figure PCTCN2021101535-appb-000015
由以上约束条件，求出B和E相开路容错后的相电流i A BE、i B BE、i C BE、i D BE、i E BE为
Figure PCTCN2021101535-appb-000016
根据式(6)求出将αβ坐标系上的变量变换到自然坐标系上的容错变换矩阵
Figure PCTCN2021101535-appb-000017
当B和E相发生开路故障后，注入式(6)所示的容错电流后，作动器旋转部分的定子磁链ψ A BE、ψ B BE、ψ C BE、ψ D BE、ψ E BE可表示为
Figure PCTCN2021101535-appb-000018
其中：
Figure PCTCN2021101535-appb-000019
Figure PCTCN2021101535-appb-000020
Figure PCTCN2021101535-appb-000021
L(θ)为旋转部分绕组的电感矩阵；θ为电角度；L m＝0.2(L d+L q)，L θ＝0.2(L q-L d)；L d、L q分别为旋转部分绕组的d轴和q轴电感；L A、L B、L C、L D、L E为A、B、C、D和E相的电感；L ls为漏感；I 5×5为五阶单位矩阵；ψ f为永磁体耦合到定子侧的永磁磁链，其可表示为ψ f＝ψ pm[cosθ cos(θ-a) cos(θ-2a) cos(θ-3a) cos(θ-4a)] T；ψ pm为永磁磁链幅值。
由于发生两相开路故障后，作动器旋转部分只剩两个自由度，采用式(1)所示Clark矩阵将自然坐标系上的定子磁链和相电压变换到αβ坐标系上的定子磁链ψ α、ψ β和定子电压u α、u β为
Figure PCTCN2021101535-appb-000022
Figure PCTCN2021101535-appb-000023
由于非故障相对故障相互感的存在以及故障相自身反电势的存在，当B和E相发生开路后，B和E相电压可表示为
Figure PCTCN2021101535-appb-000024
其中：emf B、emf E为B和E相的反电势。
根据式(6)和(11)可得
Figure PCTCN2021101535-appb-000025
其中：m 1＝L mi α-L θi αcos2θ-L θi βsin2θ，m 2＝L mi β+L θi βcos2θ-L θi αsin2θ。
为估算定子磁链，先采用电压源逆变器上桥臂开关管的状态表示非故障相的相电压，再结合式(12)，采用Clark矩阵将故障状态下的相电压变换到αβ坐标系上，可表示为
Figure PCTCN2021101535-appb-000026
式中：
Figure PCTCN2021101535-appb-000027
S a、S c、S d为电压源逆变器A、C、D相上桥臂开关管信号，开关管导通时对应信号等于1，关断时对应信号为0，U dc为直流母线电压，
Figure PCTCN2021101535-appb-000028
Figure PCTCN2021101535-appb-000029
分别是B和E相开路故障状态下的相电压；
当B和E相发生开路故障后，根据式(9)、(10)和(13)，采用改进积分器构建定子磁链观测器
Figure PCTCN2021101535-appb-000030
式中：s为微分算子，ω c为滤波截止频率。由此，故障情况下的转矩可表示为
Figure PCTCN2021101535-appb-000031
在构建好定子磁链与转矩观测器之后，基于定子磁场定向原则，采用式(16)所示的变换矩阵将αβ坐标系上的电压、电流、定子磁链变换到MT坐标系上。将式(10)电压模型变换到MT坐标系上的电压u M、u T，如式(18)所示
Figure PCTCN2021101535-appb-000032
Figure PCTCN2021101535-appb-000033
Figure PCTCN2021101535-appb-000034
式中：θ s为定子磁链角，ω s为定子磁链角速度，i M、i T、ψ M、ψ T分别为MT坐标系上的电流和定子磁链分量。
将式(15)所示的转矩变换到MT坐标系上，可表示为
T e＝2.5p(ψ Mi T-ψ Ti M)      (19)
由于MT坐标系是基于定子磁场定向，将式(14)观测出的定子磁链变换到MT该坐标系上应满足式(20)。由此，构建如式(20)所示的定子磁链观测器。
Figure PCTCN2021101535-appb-000035
将式(20)代入式(19)，作动器旋转部分的转矩观测器可设计为
T e＝2.5pψ Mi T          (21)
将式(20)和(21)代入式(18)，得到
Figure PCTCN2021101535-appb-000036
由此根据式(22)可知，采用两个PI控制器就可以实现转矩、磁链的直接解耦控制。首先，通过如式(20)所示的定子磁链观测器与式(21)所示的转矩观测器估算定子磁链幅值和转矩，并将其与给定转矩T *、给定定子磁链
Figure PCTCN2021101535-appb-000037
分别作差后，经PI控制器得到MT坐标系上的给定电压指令。其次，通过式(17)所示反变换矩阵T MT-αβ将该电压指令反变换至αβ坐标系上的电压指令。然后，采用式(7)所示容错变换矩阵将其变换到自然坐标系上，得到非故障相的电压指令，故障相电压指令为0。最后，结合基于零序电压注入的CPWM实现永磁容错作动器旋转部分在B和E相开路故障后的无扰高性能运行。该容错策略不但具备DTC的转矩动态响应迅速、结构简单的特点，而且能够有效抑制B和E相开路故障导致的转矩脉动，实现DTC平稳运行，更为关键的是稳态性能和矢量控制相当。
当C和D相发生开路故障后，按照上面的推导方法可得将αβ坐标系上的变量变换到自然坐标系上的容错变换矩阵为
Figure PCTCN2021101535-appb-000038
C和D相开路故障情况下，基于电压源逆变器上桥臂开关状态信号的非故障相电压和故障相电压在αβ坐标系上可表示为
Figure PCTCN2021101535-appb-000039
式中：
Figure PCTCN2021101535-appb-000040
S a、S b、S e为电压源逆变器A、B、E上桥臂开关管信号；
Figure PCTCN2021101535-appb-000041
分别是C和D相开路故障状态下的相电压。
当C和D相发生开路故障后，按照上述相同的方法可推导出在MT坐标系上如式(20)和(21)所示的定子磁链和转矩观测器。
根据以上两种开路故障情况的分析可知，容错运行时只需要更换如式(7)或(23)所示的容错变换矩阵，就能实现两相开路故障情况下的无扰DTC运行。
同样，当永磁容错作动器直线部分的B和E相发生开路故障或者C和D相发生开路故障时，仅需修改上述容错方法的转矩观测器，将其改成MT坐标系上的推力观测器
Figure PCTCN2021101535-appb-000042
不失一般性，本发明以该永磁容错作动器旋转部分B和E开路故障为例，对所提出的容错DTC策略的性能进行仿真分析，其控制策略框图如图5所示。与正常运行时相比，控制结构几乎不变，因此提出的容错策略简洁高效。图6为B和E相在0.2s发生开路故障，从正常进入开路故障情况下，作动器旋转部分无容错运行时的转矩波形。可见，作动器旋转部分的转矩波动明显。图7为B和E相从正常到开路故障情况下，作动器旋转部分容错运行时的转矩波形。0.2s时B和E相开路故障发生，立即启动本发明的容错DTC策略。可见，和故障情况下相比，作动器输出转矩脉动得到明显抑制，几乎没有脉动。图8为B和E相开路容错DTC运行过程中转矩指令阶跃下降时的作动器输出转矩波形，响应时间约为 1ms。可见，本发明提出的容错DTC策略保持了DTC转矩响应迅速的优点。因此当电机发生两相开路故障后，采用本发明的容错DTC策略后，输出转矩几乎没有波动，相电流正弦度较好，同时作动器具有和正常情况下相似的动态性能。
当其它两相发生故障时，只需将自然坐标系逆时针旋转0.4kπ(k＝0、1、2、3、4；C和D相故障，或B和E相故障时，k＝0；D和E相故障，或C和A相故障时，k＝1；E和A相故障，或D和B相故障时，k＝2；A和B相故障，或E和C相故障时，k＝3；B和C相故障，或A和D相故障时，k＝4)电角度。
综上所述，本发明的一种转向悬架一体化五相永磁容错作动器及其两相开路容错DTC方法不但实现了作动器的旋转运动和直线运动、螺旋运动、旋转运动和直线运动的解耦，而且能保证两相开路故障情况下作动器输出转矩或推力和正常情况下一致，同时能明显抑制两相开路故障导致的转矩或推力脉动；更为关键的是，在容错运行时的动态性能和DTC正常情况下相似，稳态性能和矢量控制时相似，通用性强、无需复杂计算、CPU开销小。因此，本发明在汽车、航空航天等对运行性能要求高的***中拥有很好的应用前景。
虽然本发明已以较佳实施例公开如上，但实施例并不是用来限定本发明的。在不脱离本发明之精神和范围内，所做的任何等效变化或润饰，均属于本申请所附权利要求所限定的保护范围。
在本说明书的描述中，参考术语“一个实施例”、“一些实施例”、“示意性实施例”、“示例”、“具体示例”、或“一些示例”等的描述意指结合该实施例或示例描述的具体特征、结构、材料或者特点包含于本发明的至少一个实施例或示例中。在本说明书中，对上述术语的示意性表述不一定指的是相同的实施例或示例。而且，描述的具体特征、结构、材料或者特点可以在任何的一个或多个实施例或示例中以合适的方式结合。
尽管已经示出和描述了本发明的实施例，本领域的普通技术人员可以理解：在不脱离本发明的原理和宗旨的情况下可以对这些实施例进行多种变化、修改、替换和变型，本发明的范围由权利要求及其等同物限定。

Claims (9)

  1. 一种转向悬架一体化五相永磁容错作动器，其特征在于：该作动器用于直线、旋转和螺旋运动；包括定子、动子；所述定子主要由辅助齿(35)、电枢齿(32)、容错齿(34)、直线运动部分的绕组(21)和旋转运动部分的绕组(22)构成；所述定子在沿轴向上是由一对辅助齿(35)包含若干个交替的电枢齿(33)与容错齿(34)排列而成；在圆周向上是由若干个电枢齿(31)与容错齿(32)交替排列而成，在圆周方向上的电枢齿(31)端部为***齿结构；所述电枢齿上设置绕组，所述绕组分为直线运动部分的绕组(21)和旋转运动部分的绕组(22)；所述动子由永磁体(5)、动子齿(6)、动子轭(7)以及不导磁圆环(42)构成；所述永磁体(5)在圆周方向是瓦片状的且径向充磁、内嵌在动子齿(6)之间，沿圆周向分布的角度圆心角为t c＝(0.6～1)*360°/P，P为永磁体沿圆周向的极对数，在圆周向按交替极阵列排列；永磁体(5)沿轴向分布的长度为t l＝(0.25～0.5)*τ，τ为轴向动子极距，在轴向相邻两永磁体的充磁方向相反，按错位交替阵列排列，错位角度为t z＝180°/P；所述永磁体(5)与动子齿(6)在轴向上两两之间设置的气隙长度为t 0＝0.5*τ-t l；所述气隙可采用不导磁圆环(42)填充；所述动子齿和永磁体在圆周面上是等高的；所述辅助齿(35)轴向宽度为t l＝(0.25～0.5)*τ。
  2. 根据权利要求1所述的一种转向悬架一体化五相永磁容错作动器，其特征在于：所述直线运动部分的绕组(21)绕制成圆环状嵌入轴向上电枢齿(33)与容错齿(34)之间的定子槽底部，电枢齿相邻两槽线圈串联成一相绕组，并且在轴向的定子槽内填充不导磁圆环(41)；所述旋转运动部分的绕组(22)分别从一边端部到另一边端部绕制在圆周向电枢齿(31)上；直线运动部分的五相绕组(21)与旋转运动部分的五相绕组(22)均采用集中式绕制。
  3. 根据权利要求1所述的一种转向悬架一体化五相永磁容错作动器，其特征在于：所述电枢齿(31)极靴部分在圆周方向是***齿结构，该***齿结构还可采用多齿来降低电枢极间漏磁，然而其在轴向方向不是***齿结构。
  4. 根据权利要求1所述的一种转向悬架一体化五相永磁容错作动器，其特征在于：永磁体(5)沿圆周向的极对数P，旋转运动部分的电枢绕组极对数P w，沿圆周向的调制齿极数N r，三者满足关系：P w＝|N r-P|；永磁体沿轴向的极对数P 1，直线运动部分的电枢绕组极对数P w1，沿轴向的调制齿极数N r1，三者满足关系：P w1＝|N r1-P 1|。
  5. 一种由权利要求1所述的转向悬架一体化五相永磁容错作动器的两相开路容错直接转矩控制方法，其特征在于，当作动器旋转部分和直线部分的相数分别为m＝5时，可分别分为A、B、C、D、E 五相，两相开路故障情况下的容错直接转矩控制方法包括如下步骤：
    步骤1，建立转向悬架一体化五相永磁容错作动器的模型；
    步骤2，转向悬架一体化五相永磁容错作动器旋转部分绕组发生两相开路故障，假设不相邻两相开路故障发生在B和E相上，根据故障前后磁动势相等原理、非故障相电流和为零的原则，求出B和E相开路故障情况下的容错电流i A BE、i B BE、i C BE、i D BE、i E BE，根据该容错电流分别推导出B和E相开路故障情况下的容错变换矩阵T BE
    Figure PCTCN2021101535-appb-100001
    或步骤2，转向悬架一体化五相永磁容错作动器旋转部分绕组发生两相开路故障，假设相邻两相开路故障发生在C和D相上，根据故障前后磁动势相等原理、非故障相电流和为零的原则，求出C和D相开路故障情况下的容错电流i A CD、i B CD、i C CD、i D CD、i E CD；根据该容错电流分别推导C和D相开路故障情况下的容错变换矩阵T CD
    Figure PCTCN2021101535-appb-100002
    步骤3，采用传统Clark变换矩阵将两相开路故障情况下定子磁链、电压和电流变换到αβ坐标系上，则αβ坐标系上的定子磁链ψ α、ψ β可表示为
    Figure PCTCN2021101535-appb-100003
    式中：i α、i β分别是定子电流在αβ坐标系上的分；θ为电角度；L m＝0.2(L d+L q)，L θ＝0.2(L q-L d)，L d、L q分别为旋转部分绕组的d轴和q轴电感；L ls为漏感；ψ pm为永磁磁链幅值；
    步骤4，根据逆变器开关管状态计算出B相和E两相开路故障情况下的非故障相电压，并结合故障相电压，采用Clark变换矩阵将它们变换到αβ坐标系上
    Figure PCTCN2021101535-appb-100004
    式中：
    Figure PCTCN2021101535-appb-100005
    S a、S c、S d分别为A、C、D相的上桥臂功率开关管信号，开关管导通时对应信号等于1，关断时对应信号为0，U dc为直流母线电压，
    Figure PCTCN2021101535-appb-100006
    Figure PCTCN2021101535-appb-100007
    分别是B和E相开路故障状态下的相电压；
    或步骤4，根据逆变器开关管状态计算出C和D相开路故障情况下的非故障相电压，并结合故障相电压，采用Clark变换矩阵将它们变换到αβ坐标系上
    Figure PCTCN2021101535-appb-100008
    式中：
    Figure PCTCN2021101535-appb-100009
    S a、S b、S e为A、B、E相的上桥臂功率开关管开关信号，
    Figure PCTCN2021101535-appb-100010
    分别是C和D相开路故障状态下的相电压；
    步骤5，采用改进型积分器设计两相开路故障情况下的定子磁链观测器
    Figure PCTCN2021101535-appb-100011
    式中：s为微分算子，ω c为滤波截止频率，由此，两相开路故障情况下的转矩可表示为
    Figure PCTCN2021101535-appb-100012
    步骤6，基于定子磁场定向原则，采用T αβ-MT变换矩阵将αβ坐标系上的电压、电流、定子磁链变换到MT坐标系上
    Figure PCTCN2021101535-appb-100013
    Figure PCTCN2021101535-appb-100014
    由于MT坐标系是基于定子磁场定向的，将αβ坐标系上的定子磁链观测器变换到MT该坐标系上，其可表示为
    Figure PCTCN2021101535-appb-100015
    由此，构建的转矩观测器为
    T e＝2.5pψ Mi T；
    式中：i T为MT坐标系上的电流分量。
    步骤7，通过步骤6构建的定子磁链观测器和转矩观测器估算出定子磁链幅值和转矩，并将其与给定定子磁链幅值ψ*和给定转矩T*分别作差后，经PI控制器得到MT坐标系上的给定电压指令
    Figure PCTCN2021101535-appb-100016
    和
    Figure PCTCN2021101535-appb-100017
    通过步骤6中T MT-αβ反变换矩阵将该电压指令反变换至αβ坐标系上的电压指令
    Figure PCTCN2021101535-appb-100018
    和
    Figure PCTCN2021101535-appb-100019
    然后，采用步骤2中的变换矩阵T BE或T CD将其变换到自然坐标系上，得到相电压指令
    Figure PCTCN2021101535-appb-100020
    若B和E相发生开路故障，则相电压指令为
    Figure PCTCN2021101535-appb-100021
    若C和D相发生开路故障，则相电压指令为
    Figure PCTCN2021101535-appb-100022
    将该相电压指令送给电压源逆变器，再结合基于零序电压注入的载波脉宽调制技术实现转向悬架一体化五相永磁容错作动器旋转部分在B和E相开路故障或C和D相开路故障情况下的容错无扰直接转矩控制运行。
  6. 由权利要求5所述的转向悬架一体化五相永磁容错作动器两相开路容错直接转矩控制方法，其特征在于，所述步骤2的具体过程为：
    步骤2.1，根据故障前后磁动势相等原理、非故障相电流和为零的原则求出B和E相开路故障情况下的容错电流i A BE、i B BE、i C BE、i D BE、i E BE
    Figure PCTCN2021101535-appb-100023
    或步骤2.1，根据故障前后磁动势相等原理、非故障相电流和为零的原则求出C和D相开路故障情况下的容错电流i A CD、i B CD、i C CD、i D CD、i E CD
    Figure PCTCN2021101535-appb-100024
    步骤2.2，根据非故障相的容错电流推导出将αβ坐标系上的变量变换到自然坐标系上的变换矩阵。
  7. 由权利要求5所述的转向悬架一体化五相永磁容错作动器两相开路容错直接转矩控制方法，其特征在于，所述步骤3的具体过程为：
    步骤3.1，将自然坐标系上的变量变换到αβ坐标系上的Clark变换矩阵为
    Figure PCTCN2021101535-appb-100025
    步骤3.2，当B和E相发生开路故障时，作动器旋转部分的定子磁链ψ A BE、ψ B BE、ψ C BE、ψ D BE、ψ E BE可表示为
    Figure PCTCN2021101535-appb-100026
    其中：L(θ)为旋转部分绕组的电感矩阵；ψ f为永磁体耦合到定子侧的永磁磁链，表示为
    ψ f＝ψ pm[cosθ cos(θ-a) cos(θ-2a) cos(θ-3a) cos(θ-4a)] T；
    步骤3.3，采用Clark变换矩阵将两相开路故障情况下自然坐标系上的定子磁链变换到αβ坐标系上。
  8. 由权利要求5所述的转向悬架一体化五相永磁容错作动器两相开路容错直接转矩控制方法，其特征在于，所述步骤4的具体过程为：
    步骤4.1，当B和E相发生开路故障后，非故障相对故障相互感以及故障相的反电势仍然存在。因此，故障相的相电压为
    Figure PCTCN2021101535-appb-100027
    其中：emf B、emf E为B和E相的反电势，L AB、L BC、L BD、L AE、L CE、L DE分别为相互感；
    步骤4.2，在B和E相故障情况下，它们的相电压之间的关系为
    Figure PCTCN2021101535-appb-100028
    其中：m 1＝L mi α-L θi αcos2θ-L θi βsin2θ，m 2＝L mi β+L θi βcos2θ-L θi αsin2θ；
    或步骤4.1，当C和D相发生开路故障后，非故障相对故障相互感以及故障相的反电势仍然存在，因此，故障相的相电压为
    Figure PCTCN2021101535-appb-100029
    其中：emf C、emf D为C和D相的反电势，L AC、L BC、L CE、L AD、L BD、L DE分别为相互感；
    或步骤4.2，在C和D相故障情况下，它们的相电压之间的关系为
    Figure PCTCN2021101535-appb-100030
    步骤4.3，根据电压源逆变器开关状态计算出两相开路故障情况下的非故障相电压，并结合计算出的故障相电压，采用Clark变换矩阵将它们变换到αβ坐标系上。
  9. 由权利要求5所述的转向悬架一体化五相永磁容错作动器两相开路容错直接转矩控制方法，其特征在于，当转向悬架一体化五相永磁容错作动器直线部分绕组发生相邻或不相邻两相开路故障时，仅需将推力观测器
    Figure PCTCN2021101535-appb-100031
    代替步骤6中的转矩观测器，本发明的容错直接转矩控制方法就能满足作动器在直线部分两相开路故障情况下的高性能运行。
PCT/CN2021/101535 2021-06-17 2021-06-22 一种转向悬架一体化五相永磁容错作动器及其两相开路容错直接转矩控制方法 WO2022261995A1 (zh)

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