WO2022085050A1 - Motor driving device, electric blower, electric vacuum cleaner, and hand dryer - Google Patents

Motor driving device, electric blower, electric vacuum cleaner, and hand dryer Download PDF

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Publication number
WO2022085050A1
WO2022085050A1 PCT/JP2020/039275 JP2020039275W WO2022085050A1 WO 2022085050 A1 WO2022085050 A1 WO 2022085050A1 JP 2020039275 W JP2020039275 W JP 2020039275W WO 2022085050 A1 WO2022085050 A1 WO 2022085050A1
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WO
WIPO (PCT)
Prior art keywords
motor
voltage
phase motor
rotation speed
phase
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Application number
PCT/JP2020/039275
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French (fr)
Japanese (ja)
Inventor
裕次 ▲高▼山
和徳 畠山
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三菱電機株式会社
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Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to PCT/JP2020/039275 priority Critical patent/WO2022085050A1/en
Priority to JP2022556837A priority patent/JP7366288B2/en
Publication of WO2022085050A1 publication Critical patent/WO2022085050A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/26—Arrangements for controlling single phase motors

Definitions

  • the present disclosure relates to a motor drive device for driving a single-phase motor, an electric blower equipped with a single-phase motor driven by the motor drive device, an electric vacuum cleaner, and a hand dryer.
  • Patent Document 1 in a method of starting a three-phase brushless motor without a position sensor, the initial position of the rotor is set by one energization, and the rotation speed of the motor is increased based on the information of the set initial position. , A method of re-detecting the position of the rotor based on the information of the motor-induced voltage after the rotation speed has increased is disclosed.
  • Patent Document 1 does not consider restarting the motor from the free-run state in which the motor is free-running due to inertia.
  • the motor-induced voltage generated in the motor may be small depending on the rotation speed of the motor, and an overcurrent may flow in the motor. Therefore, it is desired to establish a technique for suppressing an overcurrent that can flow in a motor regardless of the rotation speed of the motor when restarting from a free-run state.
  • Patent Document 1 is a technique related to position sensorless control, it is useful to establish a technique for suppressing overcurrent even in control with a position sensor.
  • the present disclosure has been made in view of the above, and an object of the present invention is to obtain a motor drive device capable of suppressing an overcurrent that may occur when a single-phase motor is restarted from a free-run state.
  • the motor drive device is a motor drive device that drives a single-phase motor in order to solve the above-mentioned problems and achieve the object.
  • the motor drive device includes an inverter and a first detector.
  • the inverter converts the DC voltage into an AC voltage and applies the converted AC voltage to the single-phase motor.
  • the first detector detects a first physical quantity that correlates with the motor-induced voltage induced in the single-phase motor.
  • the absolute value of the average value of the motor applied voltage applied to the single-phase motor by the inverter is the motor applied voltage when the single-phase motor is started from the stopped state. Greater than the absolute value of the average.
  • the motor drive device According to the motor drive device according to the present disclosure, there is an effect that the overcurrent that may occur when the single-phase motor is restarted from the free-run state can be suppressed.
  • Sectional drawing which provides the explanation of the structure of the single-phase motor in Embodiment 1.
  • the figure which shows the torque characteristic of the single-phase motor shown in FIG. Circuit diagram of the inverter shown in FIG.
  • a circuit diagram showing a modified example of the inverter shown in FIG. A block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. 1.
  • PWM pulse width modulation
  • FIG. 7 A block diagram showing another example of the carrier comparison unit shown in FIG. The figure which shows the waveform example of the main part when it operated using the carrier comparison part shown in FIG.
  • Configuration diagram of the vacuum cleaner according to the second embodiment Configuration diagram of the hand dryer according to the second embodiment
  • FIG. 1 is a block diagram showing a configuration of a motor drive system 1 including a motor drive device 2 according to the first embodiment.
  • the motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, and a battery 10.
  • the motor drive device 2 is a drive device that supplies AC power to the single-phase motor 12 to drive the single-phase motor 12.
  • the battery 10 is a DC power source that supplies DC power to the motor drive device 2.
  • the motor drive device 2 includes an inverter 11, an analog-digital converter 30, a control unit 25, and a drive signal generation unit 32.
  • the inverter 11 and the single-phase motor 12 are connected by two connecting lines 18a and 18b.
  • the motor drive system 1 includes voltage detectors 20 and 21 and current detectors 22 and 24.
  • the motor drive system 1 is a so-called position sensorless control drive system that does not use a position sensor signal for detecting the rotation position of the rotor 12a.
  • the voltage detector 20 is a detector that detects the DC voltage Vdc output from the battery 10 to the motor drive device 2.
  • the DC voltage V dc is the output voltage of the battery 10 and is the voltage applied to the inverter 11.
  • the voltage detector 21 is a detector that detects the AC voltage Vac generated between the connection lines 18a and 18b.
  • the AC voltage V ac is a voltage obtained by superimposing the motor applied voltage applied by the inverter 11 to the single-phase motor 12 and the motor-induced voltage induced by the single-phase motor 12.
  • the detected value of the voltage detector 21 is a physical quantity that correlates with the motor-induced voltage. Therefore, in this paper, the detected value of the voltage detector 21 may be described as "a first physical quantity that correlates with the motor-induced voltage".
  • the state in which the inverter 11 has stopped operating and the inverter 11 is not outputting a voltage is referred to as "gate off”. Further, the voltage output by the inverter 11 is appropriately referred to as an "inverter output voltage”.
  • the current detector 22 is a detector that detects the motor current Im .
  • the motor current Im is an alternating current flowing in and out between the inverter 11 and the single-phase motor 12.
  • the motor current Im is equal to the alternating current flowing through the windings (not shown in FIG. 1) wound around the stator 12b of the single-phase motor 12.
  • Examples of the current detector 22 include a current transformer (CT) or a current detector that detects a current using a shunt resistor.
  • the current detector 24 is a detector that detects the power supply current I dc .
  • the power supply current I dc is a direct current flowing between the battery 10 and the inverter 11.
  • the current detector 24 is generally configured to use a shunt resistor as shown in the figure.
  • the detected value of the power supply current I dc flowing through the current detector 24 is converted into a voltage value and input to the analog-digital converter 30.
  • the detection value of the current detector 24 is appropriately referred to as "shunt voltage”.
  • the shunt voltage which is the detected value of the power supply current I dc , has a correlation with the motor current Im .
  • the shunt voltage may be described as "a second physical quantity that correlates with the motor current Im ".
  • the current detector 24 may be referred to as a "second detector”.
  • the single-phase motor 12 is used as a rotary electric machine for rotating an electric blower (not shown). Electric blowers are mounted on devices such as vacuum cleaners and hand dryers.
  • the inverter 11 is a power converter that converts the DC voltage Vdc applied from the battery 10 into an AC voltage.
  • the inverter 11 supplies AC power to the single-phase motor 12 by applying the converted AC voltage to the single-phase motor 12.
  • the analog-to-digital converter 30 is a signal converter that converts analog data into digital data.
  • the analog-digital converter 30 converts the detected value of the DC voltage V dc detected by the voltage detector 20 and the detected value of the AC voltage V ac detected by the voltage detector 21 into digital data, and causes the control unit 25. Output. Further, the analog-digital converter 30 converts the detected value of the motor current Im detected by the current detector 22 and the detected value of the power supply current I dc detected by the current detector 24 into digital data, and the control unit 25. Output to.
  • the control unit 25 is referred to as PWM signals Q1, Q2, Q3, Q4 (hereinafter, appropriately referred to as "Q1 to Q4") based on the digital output value 30a converted by the analog digital converter 30 and the voltage amplitude command V *. ) Is generated.
  • the voltage amplitude command V * will be described later.
  • the drive signal generation unit 32 has drive signals S1, S2, S3, S4 for driving the switching element in the inverter 11 based on the PWM signals Q1 to Q4 output from the control unit 25 (hereinafter, appropriately “S1 to”. S4 ") is generated.
  • the control unit 25 has a processor 31, a carrier generation unit 33, and a memory 34.
  • the processor 31 generates PWM signals Q1 to Q4 for performing PWM control.
  • the processor 31 is a processing unit that performs various operations related to PWM control and advance angle control.
  • a CPU Central Processing Unit
  • a microprocessor a microcomputer
  • a microcomputer a microcomputer
  • a DSP Digital Signal Processor
  • LSI Large Scale Integration
  • the program read by the processor 31 is stored in the memory 34.
  • the memory 34 is also used as a work area when the processor 31 performs arithmetic processing.
  • the memory 34 is generally a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Project ROM), or an EEPROM (registered trademark) (Electrically EPROM). Details of the configuration of the carrier generation unit 33 will be described later.
  • FIG. 2 is a cross-sectional view for explaining the structure of the single-phase motor 12 in the first embodiment.
  • FIG. 2 shows the cross-sectional shapes of the rotor 12a and the stator 12b of the single-phase permanent magnet brushless motor as an example of the single-phase motor 12 used in the embodiment.
  • the rotor 12a is fitted to the shaft 12c and is configured to be rotatable in the direction of the arrow shown in the figure, that is, counterclockwise.
  • Four permanent magnets are arranged in the circumferential direction on the rotor 12a. These four permanent magnets are arranged so that the magnetizing directions are alternately reversed in the circumferential direction to form a magnetic pole in the rotor 12a.
  • the case where the number of magnetic poles of the rotor 12a is 4 poles is illustrated, but the number of magnetic poles of the rotor 12a may be other than 4 poles.
  • a stator 12b is arranged around the rotor 12a.
  • the stator 12b is configured by connecting four divided cores 12d in an annular shape.
  • the split core 12d has an asymmetrically shaped tooth 12e.
  • a winding 12f is wound around the teeth 12e.
  • the teeth 12e has a first tip portion 12e1 and a second tip portion 12e2 protruding toward the rotor 12a.
  • the side ahead of the rotation direction is the first tip portion 12e1, and the side behind the rotation direction is the second tip portion 12e2.
  • the distance between the first tip portion 12e1 and the rotor 12a is referred to as a "first gap” and is represented by G1.
  • the distance between the second tip portion 12e2 and the rotor 12a is called a "second gap” and is represented by G2.
  • G1 ⁇ G2 between the first gap G1 and the second gap G2.
  • the single-phase motor 12 may be a motor having a structure in which a permanent magnet is arranged on the surface of the rotor 12a (Surface Permanent Magnet: SPM), or a magnet-embedded type (Interior) in which the permanent magnet is embedded inside the rotor 12a. It may be a motor having a Permanent Magnet (IPM) structure.
  • SPM Surface Permanent Magnet
  • IPM Permanent Magnet
  • the single-phase motor 12 is a motor having an SPM structure, there is an effect that the torque pulsation due to the reluctance torque can be reduced. Further, when the single-phase motor 12 is a motor having an IPM structure, there is an effect that the structure for holding the permanent magnet becomes easy.
  • FIG. 3 is a diagram showing changes in the rotor position when the single-phase motor 12 shown in FIG. 2 is excited.
  • FIG. 4 is a diagram showing torque characteristics of the single-phase motor 12 shown in FIG.
  • the stop position of the rotor 12a is shown in the upper part of FIG.
  • the magnetic pole center line representing the center of the magnetic pole and the tooth center line representing the structural center of the stator 12b are deviated so that the magnetic pole center line precedes the rotation direction. This occurs because the single-phase motor 12 has a structure having an asymmetrically shaped teeth 12e. With this structure, the torque characteristics as shown in FIG. 4 appear.
  • the curve K1 shown by the solid line represents the motor torque
  • the curve K2 shown by the broken line represents the cogging torque.
  • the motor torque is the torque generated in the rotor 12a by the current flowing through the winding of the stator 12b.
  • the cogging torque is the torque generated in the rotor 12a by the magnetic force of the permanent magnet when no current is flowing in the winding of the stator 12b. Take the counterclockwise direction to the positive torque.
  • the horizontal axis of FIG. 4 represents the machine angle
  • the stop position of the rotor 12a whose magnetic pole center line coincides with the teeth center line is the machine angle 0 °.
  • the cogging torque is positive when the mechanical angle is 0 °. Therefore, the rotor 12a rotates counterclockwise and stops at the position of the mechanical angle ⁇ 1 where the cogging torque becomes zero.
  • the position of the mechanical angle ⁇ 1 is the stop position shown in the upper part of FIG.
  • FIG. 5 is a circuit diagram of the inverter 11 shown in FIG.
  • the inverter 11 has a plurality of switching elements 51, 52, 53, 54 (hereinafter, appropriately referred to as “51 to 54”) to be bridge-connected.
  • the switching elements 51 and 52 constitute the first leg, the leg 5A.
  • the leg 5A is a series circuit in which a switching element 51, which is a first switching element, and a switching element 52, which is a second switching element, are connected in series.
  • the switching elements 53 and 54 constitute the second leg, the leg 5B.
  • the leg 5B is a series circuit in which a switching element 53, which is a third switching element, and a switching element 54, which is a fourth switching element, are connected in series.
  • the legs 5A and 5B are connected between the DC bus 16a on the high potential side and the DC bus 16b on the low potential side so as to be in parallel with each other. As a result, the legs 5A and 5B are connected in parallel to both ends of the battery 10.
  • the switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side.
  • the high potential side is referred to as an "upper arm” and the low potential side is referred to as a "lower arm”. Therefore, the switching elements 51 and 53 may be referred to as “upper arm switching element”, and the switching elements 52 and 54 may be referred to as "lower arm switching element”.
  • connection end 6A between the switching element 51 and the switching element 52 and the connection end 6B between the switching element 53 and the switching element 54 form an AC end in the bridge circuit.
  • a single-phase motor 12 is connected between the connection end 6A and the connection end 6B.
  • MOSFET Metal-Oxide-Semiconductor Field-Effective Transistor
  • FET Field-Effective Transistor
  • the switching element 51 is formed with a body diode 51a connected in parallel between the drain and the source of the switching element 51.
  • the switching element 52 is formed with a body diode 52a connected in parallel between the drain and the source of the switching element 52.
  • the switching element 53 is formed with a body diode 53a connected in parallel between the drain and the source of the switching element 53.
  • the switching element 54 is formed with a body diode 54a connected in parallel between the drain and the source of the switching element 54.
  • Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode. A separate freewheeling diode may be connected. Further, instead of the MOSFET, an insulated gate bipolar transistor (IGBT) may be used.
  • IGBT insulated gate bipolar transistor
  • the switching elements 51 to 54 are not limited to MOSFETs formed of silicon-based materials, and may be MOSFETs formed of wide bandgap (Wide Band Gap: WBG) semiconductors such as silicon carbide, gallium nitride, gallium oxide, or diamond.
  • WBG Wide Band Gap
  • WBG semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the plurality of switching elements 51 to 54, the withstand voltage resistance and the allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element can be miniaturized.
  • WBG semiconductors have high heat resistance. Therefore, it is possible to reduce the size of the heat radiating portion for radiating the heat generated in the semiconductor module. In addition, it is possible to simplify the heat dissipation structure that dissipates heat generated by the semiconductor module.
  • FIG. 6 is a circuit diagram showing a modified example of the inverter 11 shown in FIG.
  • the inverter 11A shown in FIG. 6 has shunt resistors 55a and 55b added to the configuration of the inverter 11 shown in FIG.
  • the shunt resistor 55a is a detector for detecting the current flowing through the leg 5A
  • the shunt resistor 55b is a detector for detecting the current flowing through the leg 5B.
  • the shunt resistor 55a is connected between the terminal on the low potential side of the switching element 52 and the DC bus 16b
  • the shunt resistor 55b is connected to the terminal on the low potential side of the switching element 54 and the DC bus. It is connected to 16b.
  • the current detector 22 shown in FIG. 1 can be omitted.
  • the detected values of the shunt resistors 55a and 55b are sent to the processor 31 via the analog-digital converter 30.
  • the processor 31 implements activation control, which will be described later, based on the detected values of the shunt resistors 55a and 55b.
  • the shunt resistor 55a is not limited to that of FIG. 6 as long as it can detect the current flowing through the leg 5A.
  • the shunt resistor 55a is located between the DC bus 16a and the terminal on the high potential side of the switching element 51, between the terminal on the low potential side of the switching element 51 and the connection end 6A, or between the connection end 6A and the high potential of the switching element 52. It may be arranged between the terminal on the side.
  • the shunt resistor 55b is between the DC bus 16a and the terminal on the high potential side of the switching element 53, between the terminal on the low potential side of the switching element 53 and the connection end 6B, or between the connection end 6B and the switching element 54. It may be arranged between the terminal on the high potential side of the.
  • the on-resistance of the MOFFET may be used to detect the current with the voltage generated across the on-resistance.
  • FIG. 7 is a block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit 25 shown in FIG.
  • the carrier comparison unit 38 is input with the advance angle controlled advance phase ⁇ v and the reference phase ⁇ e used when generating the voltage command V m described later.
  • the reference phase ⁇ e is a phase obtained by converting the rotor mechanical angle, which is the angle of the rotor 12a from the reference position, into an electric angle.
  • the motor drive device 2 according to the first embodiment has a so-called position sensorless control configuration that does not use the position sensor signal from the position sensor. Therefore, the rotor mechanical angle and the reference phase ⁇ e are estimated by calculation.
  • the "advance angle phase” referred to here is a phase representing the "advance angle” which is the "advance angle” of the voltage command Vm .
  • the "advance angle” referred to here is a phase difference between the motor applied voltage applied to the winding 12f of the stator 12b and the motor induced voltage induced in the winding 12f of the stator 12b.
  • the “advance angle” takes a positive value when the voltage applied to the motor is ahead of the voltage induced by the motor.
  • the carrier comparison unit 38 in addition to the advance phase ⁇ v and the reference phase ⁇ e , the carrier generated by the carrier generation unit 33, the DC voltage V dc , and the voltage which is the amplitude value of the voltage command V m . Amplitude command V * is input.
  • the carrier comparison unit 38 generates PWM signals Q1 to Q4 based on the carrier, the advance phase ⁇ v , the reference phase ⁇ e , the DC voltage V dc , and the voltage amplitude command V *.
  • FIG. 8 is a block diagram showing an example of the carrier comparison unit 38 shown in FIG. 7.
  • FIG. 8 shows the detailed configuration of the carrier comparison unit 38A and the carrier generation unit 33.
  • a triangular wave carrier moving up and down between “0” and “1” is shown.
  • the PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the advance phase ⁇ v . On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carrier with the advance phase ⁇ v .
  • the carrier comparison unit 38A has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion unit. It has 38i and an output inversion unit 38j.
  • the absolute value calculation unit 38a calculates the absolute value
  • is divided by the DC voltage V dc detected by the voltage detector 20.
  • the output of the division unit 38b is the modulation factor.
  • the battery voltage which is the output voltage of the battery 10, fluctuates as the current continues to flow.
  • the value of the modulation factor can be adjusted so that the motor applied voltage does not decrease due to the decrease in the battery voltage.
  • the multiplication unit 38c calculates a sine value of “ ⁇ e + ⁇ v ”, which is the reference phase ⁇ e plus the advance phase ⁇ v .
  • the calculated sine value of " ⁇ e + ⁇ v " is multiplied by the modulation factor which is the output of the division unit 38b.
  • "1/2" is multiplied by the voltage command Vm , which is the output of the multiplication unit 38c.
  • Vm which is the output of the multiplication unit 38c.
  • the addition unit 38e "1/2" is added to the output of the multiplication unit 38d.
  • "-1" is multiplied by the output of the addition unit 38e.
  • the output of the addition unit 38e is input to the comparison unit 38g as a positive voltage command Vm1 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54, and is input to the comparison unit 38g of the multiplication unit 38f.
  • the output is input to the comparison unit 38h as a negative voltage command Vm2 for driving the two switching elements 52 and 54 of the lower arm.
  • the comparison unit 38 g the positive voltage command V m1 and the amplitude of the carrier are compared.
  • the output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52.
  • the negative voltage command Vm2 and the amplitude of the carrier are compared.
  • the output of the output inversion unit 38j which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54.
  • the output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time
  • the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
  • FIG. 9 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38A shown in FIG.
  • the waveform of the positive voltage command V m1 output from the addition unit 38e, the waveform of the negative voltage command V m2 output from the multiplication unit 38f, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown.
  • the voltage waveform is shown.
  • the PWM signal Q1 becomes “Low” when the positive voltage command V m1 is larger than the carrier, and becomes “High” when the positive voltage command V m1 is smaller than the carrier.
  • the PWM signal Q2 is an inverted signal of the PWM signal Q1.
  • the PWM signal Q3 becomes “Low” when the negative voltage command V m2 is larger than the carrier, and becomes “High” when the negative voltage command V m2 is smaller than the carrier.
  • the PWM signal Q4 is an inverted signal of the PWM signal Q3.
  • the circuit shown in FIG. 8 is configured with “Low Active", but even if each signal is configured with "High Active” having opposite values. good.
  • the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.
  • Bipolar modulation and unipolar modulation are known as modulation methods used when generating PWM signals Q1 to Q4.
  • Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential every cycle of the voltage command Vm .
  • Unipolar modulation is a modulation method that outputs a voltage pulse that changes at three potentials in each cycle of the voltage command Vm , that is, a voltage pulse that changes between a positive potential, a negative potential, and a zero potential.
  • the waveform shown in FIG. 9 is due to unipolar modulation.
  • any modulation method may be used. In applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation having a lower harmonic content than bipolar modulation.
  • the waveform shown in FIG. 9 shows four switchings of the switching elements 51 and 52 constituting the leg 5A and the switching elements 53 and 54 constituting the leg 5B during the half-cycle T / 2 period of the voltage command V m . It is obtained by a method of switching the element. This method is called “both-sided PWM" because the switching operation is performed by both the positive side voltage command V m1 and the negative side voltage command V m2 . On the other hand, in one half cycle of one cycle T of the voltage command V m , the switching operation of the switching elements 51 and 52 is suspended, and in the other half cycle of the one cycle T of the voltage command V m , the switching operation is suspended. There is also a method of suspending the switching operation of the switching elements 53 and 54.
  • one-sided PWM This method is called “one-sided PWM”.
  • double-sided PWM mode the operation mode operated by double-sided PWM
  • one-sided PWM mode the operation mode operated by one-sided PWM
  • the PWM signal by "two-sided PWM” may be called “two-sided PWM signal”
  • the PWM signal by "one-sided PWM” may be called “one-sided PWM signal”.
  • FIG. 10 is a block diagram showing another example of the carrier comparison unit 38 shown in FIG. 7.
  • FIG. 10 shows an example of a one-sided PWM signal generation circuit, and specifically, a detailed configuration of a carrier comparison unit 38B and a carrier generation unit 33 is shown.
  • the configuration of the carrier generation unit 33 shown in FIG. 10 is the same as or equivalent to that shown in FIG.
  • the configuration of the carrier comparison unit 38B shown in FIG. 10 the same or equivalent components as the carrier comparison unit 38A shown in FIG. 8 are designated by the same reference numerals.
  • the carrier comparison unit 38B has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.
  • the absolute value calculation unit 38a calculates the absolute value
  • is divided by the DC voltage V dc detected by the voltage detector 20. Even in the configuration of FIG. 10, the output of the division unit 38b is the modulation factor.
  • the multiplication unit 38c calculates a sine value of “ ⁇ e + ⁇ v ”, which is the reference phase ⁇ e plus the advance phase ⁇ v .
  • the calculated sine value of " ⁇ e + ⁇ v " is multiplied by the modulation factor which is the output of the division unit 38b.
  • "-1" is multiplied by the voltage command Vm , which is the output of the multiplication unit 38c.
  • Vm which is the output of the multiplication unit 38c.
  • “1” is added to the voltage command Vm which is the output of the multiplication unit 38c.
  • "1" is added to the output of the multiplication unit 38k, that is, the inverted output of the voltage command Vm .
  • the output of the addition unit 38m is input to the comparison unit 38g as a first voltage command Vm3 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54.
  • the output of the addition unit 38n is input to the comparison unit 38h as a second voltage command Vm4 for driving the two switching elements 52 and 54 of the lower arm.
  • the comparison unit 38 g the first voltage command V m3 and the amplitude of the carrier are compared.
  • the output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52.
  • the second voltage command Vm4 and the amplitude of the carrier are compared.
  • the output of the output inversion unit 38j which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54.
  • the output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time
  • the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
  • FIG. 11 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38B shown in FIG.
  • the waveform of the first voltage command V m3 output from the adder 38 m the waveform of the second voltage command V m4 output from the adder 38n, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown.
  • the voltage waveform is shown.
  • the waveform portion of the first voltage command V m3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V m4 whose amplitude value is larger than the peak value of the carrier.
  • the corrugated portion is represented by a flat straight line.
  • the PWM signal Q1 becomes “Low” when the first voltage command V m3 is larger than the carrier, and becomes “High” when the first voltage command V m3 is smaller than the carrier.
  • the PWM signal Q2 is an inverted signal of the PWM signal Q1.
  • the PWM signal Q3 becomes “Low” when the second voltage command V m4 is larger than the carrier, and becomes “High” when the second voltage command V m4 is smaller than the carrier.
  • the PWM signal Q4 is an inverted signal of the PWM signal Q3.
  • the circuit shown in FIG. 10 is configured with “Low Active", but even if each signal is configured with "High Active” having opposite values. good.
  • the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.
  • the switching element 52 is controlled to be always on in one half cycle of one cycle T of the voltage command V m , and the one cycle T of the voltage command V m is controlled.
  • the switching element 54 is controlled to be always on.
  • FIG. 11 is an example, in which the switching element 51 is controlled to be always on in one half cycle, and the switching element 53 is controlled to be always on in the other half cycle. Is also possible. That is, the waveform shown in FIG. 11 is characterized in that at least one of the switching elements 51 to 54 is controlled to be in the ON state in the half cycle of the voltage command V m .
  • the waveform of the inverter output voltage is unipolar modulation that changes at three potentials in each cycle of the voltage command Vm .
  • bipolar modulation may be used instead of unipolar modulation, but in applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation.
  • FIG. 12 is a block diagram showing a functional configuration for calculating the advance angle phase ⁇ v input to the carrier comparison unit 38 shown in FIG. 7.
  • the function of calculating the advance angle phase ⁇ v can be realized by the rotation speed calculation unit 42 and the advance angle phase calculation unit 44.
  • the rotation speed calculation unit 42 calculates the rotation speed ⁇ of the single-phase motor 12 based on the detection value of the motor current Im detected by the current detector 22. Further, the rotation speed calculation unit 42 calculates the reference phase ⁇ e based on the detected value of the motor current Im.
  • the reference phase ⁇ e is a phase obtained by converting the rotor mechanical angle, which is the angle of the rotor 12a from the reference position, into an electric angle.
  • the rotor machine angle is a calculated value calculated inside the rotation speed calculation unit 42.
  • the advance phase phase calculation unit 44 calculates the advance phase phase ⁇ v based on the rotation speed ⁇ , the reference phase ⁇ e , and the motor-induced voltage.
  • the motor induced voltage can be obtained from the detected value of the AC voltage Vac .
  • the detected value of the AC voltage Vac includes the motor applied voltage applied to the single-phase motor 12 by the inverter 11 and the motor-induced voltage induced by the single-phase motor 12. Of these voltages, the motor-induced voltage can be detected during the gate-off period when the inverter 11 does not output the voltage. The details of the calculation method of the advance phase ⁇ v will be described later.
  • FIG. 13 is a first diagram used for explaining the operation of a main part in the acceleration control of the first embodiment.
  • FIG. 14 is a second diagram used for explaining the operation of the main part in the acceleration control of the first embodiment.
  • the above-mentioned FIGS. 2 and 3 are examples in which the single-phase motor 12 having the asymmetrically shaped teeth 12e is the drive target, but the single-phase motor to be driven has the structure of FIGS. 2 and 3. Not limited. That is, this control is not limited to the case where the teeth 12e has an asymmetrical shape, and can be applied even when the teeth 12e has a symmetrical shape.
  • FIG. 13 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is low.
  • FIG. 14 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is high.
  • the low speed or high speed referred to here means a relative relationship between the two, and the first acceleration control shown in FIG. 13 and the second acceleration control shown in FIG. 14 are at preset rotation speeds. Switch. In this paper, this rotation speed is referred to as "rotation speed A".
  • rotation speed A When the rotation speed of the single-phase motor 12 is less than the rotation speed A, the single-phase motor 12 is driven by the first acceleration control shown in FIG.
  • the rotation speed of the single-phase motor 12 is equal to or higher than the rotation speed A, the single-phase motor 12 is driven by the second acceleration control shown in FIG.
  • the waveform of the motor-induced voltage is shown in the upper part of FIG. In the lower part of FIG. 13, the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown.
  • the gate-on period in which the inverter 11 gates on is indicated by a coarse hatch pattern
  • the gate-off period in which the inverter 11 gates off is indicated by a fine hatch pattern.
  • the gate-on period T1 is a period in which the polarity of the motor applied voltage is positive
  • the gate-on period T2 is a period in which the polarity of the motor applied voltage is negative.
  • the gate-on period T1 may be referred to as the "first period” and the gate-on period T2 may be referred to as the "second period”.
  • the gate-off period T3 may be referred to as an "application stop period”.
  • T4 represents a period of 1 ⁇ 2 of the rotation cycle of the single-phase motor 12, that is, a rotation half cycle.
  • one cycle of the electric angle is described as one rotation cycle, but one cycle of the rotor mechanical angle may be one rotation cycle.
  • a voltage having a positive polarity is applied during the gate-on period T1.
  • the voltage of this polarity is called "first voltage”.
  • the gate-on period T1 begins at the zero crossing point where the polarity of the motor-induced voltage switches from negative to positive.
  • a voltage having a negative polarity is applied.
  • the voltage of this polarity is called "second voltage”.
  • the gate-on period T2 begins at the zero crossing point where the polarity of the motor-induced voltage switches from positive to negative.
  • FIG. 13 illustrates a case where the first voltage and the second voltage are voltages of one pulse, but the present invention is not limited to this.
  • the first voltage and the second voltage may be voltages of a plurality of PWM-controlled pulse trains.
  • the polarity switching of the motor applied voltage is performed based on the rotation speed of the single-phase motor 12 and the motor-induced voltage.
  • the gate-off period T3 since the inverter 11 is gate-off, the motor-induced voltage can be detected by the voltage detector 21. Therefore, it is possible to detect the zero crossing point of the motor induced voltage.
  • the zero cross point is a phase obtained by converting the rotor mechanical angle into an electric angle, and it is also possible to use the reference phase ⁇ e obtained by calculation.
  • the zero crossing point of the motor-induced voltage is set as the switching point of the polarity of the motor applied voltage. That is, when the rotation speed is less than the rotation speed A, the threshold value for switching the polarity of the motor applied voltage is set to a zero value. Therefore, the gate-on period T1 or the gate-on period T2 is started at the zero crossing point of the motor induced voltage. Then, by repeating the gate-on periods T1 and T2, rotational torque is applied to the single-phase motor 12, and the single-phase motor 12 accelerates and rotates. In this paper, this threshold is referred to as "threshold A".
  • the length of the gate-on periods T1 and T2 and the amplitude of the motor applied voltage can be determined based on the duty ratio, the modulation factor and the rotation speed.
  • the duty ratio is the ratio of the gate-on periods T1 and T2 to the rotation half cycle T4.
  • the motor-induced voltage may be calculated based on the detection value of the voltage detector 20 or the detection value of the current detector 24.
  • a control means for reducing the output voltage of the battery 10 to zero or a mechanism for disconnecting the electrical connection between the battery 10 and the inverter 11 is required.
  • FIG. 14 shows the waveform of the motor-induced voltage
  • the lower part shows the waveform of the motor applied voltage and the waveform of the motor-induced voltage.
  • the hatching patterns attached to the gate-on period and the gate-off period are the same as those in FIG.
  • the polarity of the motor applied voltage is switched based on the rotation speed of the single-phase motor 12 and the motor-induced voltage.
  • the gate-off period ⁇ 3 since the inverter 11 is gate-off, the motor-induced voltage can be detected by the voltage detector 21.
  • a first voltage having a positive polarity is applied.
  • the gate-on period ⁇ 1 starts when the absolute value of the amplitude of the motor-induced voltage reaches ⁇ V.
  • a second voltage having a negative polarity is applied in the gate-on period ⁇ 2 .
  • the gate-on period ⁇ 2 starts when the absolute value of the amplitude of the motor-induced voltage reaches ⁇ V. That is, in the second acceleration control, the value of ⁇ V to be compared with the absolute value of the amplitude of the motor-induced voltage is set as the threshold value A.
  • the inverter 11 inverts the polarity of the voltage applied to the single-phase motor 12 each time the absolute value of the amplitude of the motor-induced voltage reaches the threshold value A. Further, in the second acceleration control, the threshold value A is set so as to tend to increase with an increase in the rotation speed. Note that FIG. 14 illustrates a case where the first voltage and the second voltage are voltages of one pulse, but the present invention is not limited to this.
  • the first voltage and the second voltage may be voltages of a plurality of PWM-controlled pulse trains.
  • the duty ratios T1 / T4, ⁇ 1 / ⁇ 4, T2 / T4, ⁇ 2 / ⁇ 4 contribute to the motor applied voltage
  • the threshold value A ( ⁇ V) is the phase difference of the motor applied voltage with respect to the motor induced voltage. Contribute to.
  • the reactance component ( ⁇ L) is smaller than at high speeds. Therefore, as for the motor current flowing through the single-phase motor 12, the phase delay of the motor applied voltage with respect to the motor current is smaller at low speed than at high speed.
  • a small phase lag means a large power factor. If the power factor is large, it is possible to apply an effective motor torque to the single-phase motor 12.
  • the reactance component ( ⁇ L) becomes large.
  • the phase delay of the motor applied voltage with respect to the motor current becomes large, but by increasing the advance phase ⁇ v , it is possible to suppress the decrease in the power factor.
  • the motor-induced voltage generated in the single-phase motor 12 increases as the rotation speed increases. When the motor induced voltage is large, the overcurrent can be suppressed even if the inverter output voltage is increased. Therefore, by increasing the inverter output voltage according to the increase in the rotation speed, it is possible to shorten the acceleration time while suppressing the overcurrent.
  • the threshold value A ( ⁇ V) is controlled to increase as the rotation speed increases, and the advance phase ⁇ v is increased. Since a high power factor can be maintained by this control, the acceleration torque applied to the single-phase motor 12 can be efficiently obtained, and the electric power supplied to the single-phase motor 12 can be effectively utilized.
  • FIG. 15 is a first diagram used for explaining the operation of a main part in the restart control of the first embodiment.
  • FIG. 16 is a second diagram used for explaining the operation of the main part in the restart control of the first embodiment.
  • the free run refers to a state in which the single-phase motor 12 is rotating by inertia.
  • FIGS. 2 and 3 are examples in which the single-phase motor 12 having the asymmetrically shaped teeth 12e is the drive target, but the single-phase motor to be driven has the structure of FIGS. 2 and 3. Not limited. That is, this control is not limited to the case where the teeth 12e has an asymmetrical shape, and can be applied even when the teeth 12e has a symmetrical shape.
  • FIG. 15 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is high.
  • FIG. 16 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is low.
  • the high speed or low speed referred to here means a relative relationship between the two, and the first restart control shown in FIG. 15 and the second restart control shown in FIG. 16 are preset rotation speeds. Switch by speed. In this paper, this rotation speed is referred to as "rotation speed B".
  • rotation speed B When the rotation speed of the single-phase motor 12 is equal to or higher than the rotation speed B, the single-phase motor 12 is driven by the first restart control shown in FIG.
  • the rotation speed of the single-phase motor 12 is less than the rotation speed B, the single-phase motor 12 is driven by the second restart control shown in FIG.
  • the rotation speed B does not depend on the above-mentioned rotation speed A. That is, the rotation speed B may be larger or smaller than the rotation speed A.
  • the rotation speed B may be referred to as a "first rotation speed”.
  • the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown in the upper part of FIG. 15.
  • the solid line is the voltage applied to the motor, and the broken line is the voltage induced by the motor. In this free-run state, no voltage is applied to the single-phase motor 12, so it is possible to observe only the motor-induced voltage.
  • the absolute value of the average value of the motor applied voltage is shown in the middle part of FIG.
  • the broken line is the absolute value of the average value of the motor applied voltage when the single-phase motor 12 is started from the stopped state. Further, the waveform of the motor current is shown in the lower part of FIG.
  • the absolute value of the average value of the motor applied voltage applied to the single-phase motor 12 by the inverter 11 is the average value of the motor applied voltage when the single-phase motor 12 is started from the stopped state. It is larger than the absolute value of.
  • the motor current increases in proportion to the difference voltage between the motor applied voltage and the motor induced voltage. Therefore, when the rotation speed of the single-phase motor 12 becomes relatively small, the motor-induced voltage also becomes relatively small. On the contrary, when the rotation speed of the single-phase motor 12 becomes relatively large, the motor-induced voltage also becomes relatively large. Therefore, when starting from a stopped state, it is necessary to control so that the average value of the motor applied voltage gradually increases from a small value. On the other hand, in the restart from the free-run state, the average value of the motor applied voltage can be increased as compared with the case of starting from the stopped state. Even with this control, it is possible to suppress the generation of excessive motor current.
  • the on time and the off time are alternately repeated within the rotation half cycle.
  • the off time is the gate off period and the on time is the gate on period.
  • a reflux current flows through the switching elements 51, 53 or the switching elements 52, 54 during the gate-off period. Therefore, if the restart control described above is performed, the time for the reflux current to flow can be reduced. This makes it possible to suppress the generation of an excessive return current.
  • the rate of increase of the motor applied voltage until the rotation speed of the single-phase motor 12 reaches the rotation speed C larger than the rotation speed B is such that the rotation speed of the single-phase motor 12 is the rotation speed C. It is characterized in that it is smaller than the rate of increase of the motor applied voltage after reaching.
  • the rotation speed C may be referred to as a "second rotation speed".
  • the rotation speed in the free run state is calculated using the information of the zero cross point of the motor induced voltage.
  • the advance angle phase ⁇ v is changed according to the rotation speed, as in the second acceleration control described above. By doing so, it is possible to suppress the deterioration of the power factor at the time of the first restart control.
  • FIG. 16 is a diagram showing restart control when the rotation speed of the single-phase motor 12 is less than the rotation speed B.
  • the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown.
  • the waveform of the motor current is shown in the middle part of FIG.
  • the waveform of the brake signal is shown in the lower part of FIG.
  • the brake signal is a control signal output when a braking force is applied to the single-phase motor 12.
  • a braking force is applied to the single-phase motor 12 by passing a reflux current through the switching element of the lower arm or the upper arm of the inverter 11.
  • the rotation speed B is defined as the determination value of the rotation speed at which the observation accuracy of the motor-induced voltage cannot be obtained.
  • the rotation speed of the single-phase motor 12 is less than the rotation speed B, the waveform of the motor-induced voltage is small, so that accurate motor-induced voltage may not be detected. If the single-phase motor 12 is restarted in this state, the drive torque synchronized with the rotation speed cannot be applied to the single-phase motor 12, and the restart may fail. In addition to failing to restart, there is a risk of causing an overcurrent in the single-phase motor 12 or causing the single-phase motor to rotate in the opposite direction.
  • the brake signal is controlled from off to on, and the braking force is applied to the single-phase motor 12 to apply the braking force to the single-phase motor 12. Controls to reduce the rotation speed of.
  • the restart of the single-phase motor 12 can be performed after the single-phase motor 12 is stopped, or can be performed after detecting that the motor current flowing through the single-phase motor 12 has become zero.
  • the brake is controlled during the specified time t1 and restarted after the specified time t1 has elapsed. That is, it can be said that the specified time t1 is the time required for the motor current to decrease until it is regarded as zero.
  • the specified time t1 can be arbitrarily set based on the characteristics of the single-phase motor 12, the rotation speed B which is the threshold value for switching between the first and second restart controls, and the like.
  • the brake control for the single-phase motor 12 is performed, but the present invention is not limited to this example.
  • the motor-induced voltage generated in the single-phase motor 12 may be directly detected, and the brake control may be performed based on the detected motor-induced voltage.
  • the motor-induced voltage detected during the free run may be compared with the threshold value B, and when the motor-induced voltage is less than the threshold value B, brake control may be performed on the single-phase motor 12. That is, when the motor induced voltage is less than the threshold value B, the single-phase motor 12 is restarted after the braking force is applied.
  • the threshold value B may be referred to as a "first threshold value".
  • the method of applying the braking force to the single-phase motor 12 by passing a recirculation current through the switching element of the lower arm or the upper arm of the inverter 11 has been described, but the braking force is applied to the single-phase motor 12. Is not limited to this method, and may be carried out by any method or means.
  • the single-phase motor when the rotation speed of the single-phase motor is less than the first rotation speed, the single-phase motor is used after applying a braking force to the single-phase motor. restart.
  • the rotation speed of the single-phase motor is equal to or higher than the first rotation speed, the single-phase motor is restarted without applying a braking force to the single-phase motor.
  • Such restart control makes it possible to suppress the overcurrent that may occur when the single-phase motor is restarted from the free-run state. Further, since the generation of overcurrent can be suppressed, demagnetization of the rotor in the single-phase motor can be prevented.
  • the absolute value of the average value of the motor applied voltage applied to the single-phase motor by the inverter is It is controlled so as to be larger than the absolute value of the average value of the motor applied voltage when the single-phase motor is started from the stopped state.
  • the average value of the motor applied voltage gradually increases from a small value, but when restarting from a free-run state, compared to when starting from a stopped state, The average value of the motor applied voltage can be increased. Therefore, according to this control, it is possible to shorten the time required for restarting while suppressing the generation of an excessive motor current.
  • the distance between the permanent magnet provided in the rotor and the substrate provided with the magnetic pole position sensor becomes close.
  • the substrate is arranged at a position that obstructs the flow of the wind generated by the blades, which increases the pressure loss of the air passage. The increase in pressure loss becomes a factor that deteriorates the suction power of the vacuum cleaner and lowers the suction power.
  • the application example is an electric blower
  • the gas sucked by the electric blower contains a large amount of water
  • the amount of water that directly collides with the substrate increases.
  • a voltage is applied to the substrate
  • ionized metal moves between the electrodes to cause a short circuit, which may cause ion migration.
  • dust or dust accumulating on the substrate there is a concern about a short circuit caused by dust or dust accumulating on the substrate.
  • a method of applying a moisture-proof agent to the substrate or a method of isolating the substrate from the air passage is adopted, but both of them lead to an increase in manufacturing cost.
  • the degree of freedom in board placement is increased, so that the board can be placed while avoiding the air passage.
  • the amount of water that directly collides with the substrate is reduced, so that the occurrence of ion migration can be suppressed and the amount of the moisture-proofing agent can be reduced.
  • the degree of freedom in arranging the substrate is increased, the quality of the substrate can be improved by arranging the substrate outside the housing.
  • the position sensor is a magnetic pole position sensor
  • the accuracy of the mounting work for correctly detecting the magnetic pole position is required, and it is necessary to carry out the position adjusting work according to the mounting position. For this reason, it becomes difficult to control the manufacturing, and the manufacturing cost including the installation work increases.
  • the inverter and the single-phase motor can be configured separately. This makes it possible to reduce the restrictions when applying the product. For example, when the application example is a product used in a water place or the like, the inverter can be isolated from the position of the water place or the like and arranged.
  • the configuration is equipped with a current detector.
  • the current detector can detect a motor abnormality such as a shaft lock or a phase loss by detecting the motor current. This makes it possible to safely stop without a position sensor.
  • a second threshold value for determining an overcurrent is set. Then, when the shunt voltage reaches the second threshold value, it is determined that the motor is abnormal. Further, when it is determined that the motor is abnormal, the output of the inverter is cut off. By doing so, it is possible to detect a motor abnormality and safely stop the operation of the product.
  • Embodiment 2 In the second embodiment, an application example of the motor drive device 2 described in the first embodiment will be described.
  • the motor drive device 2 described above can be used, for example, in a vacuum cleaner.
  • a vacuum cleaner In the case of a product such as a vacuum cleaner that is used immediately after the power is turned on, the effect of shortening the start-up time of the motor drive device 2 according to the first embodiment becomes large.
  • FIG. 17 is a configuration diagram of the vacuum cleaner 61 according to the second embodiment.
  • the vacuum cleaner 61 shown in FIG. 17 is a so-called stick-type vacuum cleaner.
  • the vacuum cleaner 61 includes a battery 10 shown in FIG. 1, a motor driving device 2 shown in FIG. 1, an electric blower 64 driven by a single-phase motor 12 shown in FIG. 1, and dust collector.
  • a chamber 65, a sensor 68, a suction port 63, an extension pipe 62, and an operation unit 66 are provided.
  • the user who uses the vacuum cleaner 61 has an operation unit 66 and operates the vacuum cleaner 61.
  • the motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 10 as a power source. By driving the electric blower 64, dust is sucked from the suction port 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.
  • stick-type vacuum cleaner is illustrated in FIG. 17, it is not limited to the stick-type vacuum cleaner.
  • the technique of the present disclosure can be applied to any product as long as it is an electric device equipped with an electric blower.
  • FIG. 17 shows a configuration in which the battery 10 is used as a power source, but the present invention is not limited to this. Instead of the battery 10, an AC power supply supplied from an outlet may be used.
  • the motor drive device 2 described above can be used for, for example, a hand dryer.
  • a hand dryer the shorter the time from inserting the hand to driving the electric blower, the better the user's usability. Therefore, the effect of shortening the acceleration time of the motor drive device 2 according to the first embodiment is greatly exhibited.
  • FIG. 18 is a block diagram of the hand dryer 90 according to the second embodiment.
  • the hand dryer 90 includes the motor drive device 2 shown in FIG. 1, the casing 91, the hand detection sensor 92, the water receiving unit 93, the drain container 94, the cover 96, the sensor 97, and the intake air. It includes a port 98 and an electric blower 95 driven by the single-phase motor 12 shown in FIG.
  • the sensor 97 is either a gyro sensor or a motion sensor.
  • the hand dryer 90 when a hand is inserted into the hand insertion portion 99 at the upper part of the water receiving portion 93, water is blown off by the blown air by the electric blower 95, and the blown water is collected by the water receiving portion 93. After that, it is stored in the drain container 94.
  • the position sensor is a sensitive sensor, high-precision mounting accuracy is required for the installation position of the position sensor.
  • it is necessary to make adjustments according to the mounting position of the position sensor.
  • the position sensorless configuration the position sensor itself becomes unnecessary, and the adjustment step of the position sensor can be eliminated. As a result, the manufacturing cost can be significantly reduced.
  • the quality of the product can be improved because the position sensor is not affected by the secular variation.
  • the inverter and the single-phase motor can be configured separately. This makes it possible to relax restrictions on the product. For example, in the case of a product used in a water place with a large amount of water, the mounting position of the inverter in the product can be arranged at a place far from the water place. As a result, the possibility of failure of the inverter can be reduced, and the reliability of the device can be improved.
  • the motor abnormality such as the shaft lock and the open phase can be detected by detecting the motor current or the inverter current by the current detector arranged instead of the position sensor. Therefore, the product can be safely stopped without the position sensor.
  • the motor drive device 2 can be widely applied to an electric device on which a motor is mounted.
  • Examples of electrical equipment equipped with motors are incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators. , OA equipment, and electric blowers.
  • the electric blower is a blowing means for transporting an object, sucking dust, or for general blowing and exhausting.
  • the configuration shown in the above embodiments is an example, and can be combined with another known technique, or can be combined with each other, and deviates from the gist. It is also possible to omit or change a part of the configuration to the extent that it does not.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

A motor driving device (2) comprises: an inverter (11) that converts a DC voltage into an AC voltage and that applies the converted AC voltage to a single-phase motor (12); and a voltage detector (21) that detects a motor induced voltage induced on the single-phase motor (12). In reactivation control for reactivating the single-phase motor (12) from a free running state, the absolute value of an average value of a motor applied voltage applied to the single-phase motor (12) by the inverter (11) is greater than the absolute value of an average value of a motor applied voltage when the single-phase motor (12) is activated from a stopped state.

Description

モータ駆動装置、電動送風機、電気掃除機及びハンドドライヤMotor drive, electric blower, vacuum cleaner and hand dryer
　本開示は、単相モータを駆動するモータ駆動装置、モータ駆動装置によって駆動される単相モータを搭載した電動送風機、電気掃除機及びハンドドライヤに関する。 The present disclosure relates to a motor drive device for driving a single-phase motor, an electric blower equipped with a single-phase motor driven by the motor drive device, an electric vacuum cleaner, and a hand dryer.
　従来、モータを安価に駆動するための手段として、位置センサを用いずにモータ電流に基づいてモータの回転子位置を推定する位置センサレス制御が広く知られている。下記特許文献１には、三相のブラシレスモータを位置センサレスで起動する方法において、１回の通電でロータの初期位置を設定し、設定した初期位置の情報に基づいてモータの回転速度を上昇させ、回転速度が上昇した後のモータ誘起電圧の情報に基づいて、ロータの位置検出を再度行う方法が開示されている。 Conventionally, as a means for driving a motor at low cost, a position sensorless control that estimates the rotor position of a motor based on a motor current without using a position sensor is widely known. In Patent Document 1 below, in a method of starting a three-phase brushless motor without a position sensor, the initial position of the rotor is set by one energization, and the rotation speed of the motor is increased based on the information of the set initial position. , A method of re-detecting the position of the rotor based on the information of the motor-induced voltage after the rotation speed has increased is disclosed.
特開平１－３０８１９２号公報Japanese Unexamined Patent Publication No. 1-308192
　しかしながら、上記特許文献１には、モータが惰性で回転しているフリーラン状態から再起動することについての考慮がなされていない。フリーラン状態から再起動する場合、モータの回転速度によっては、モータに発生するモータ誘起電圧が小さく、モータに過電流が流れるおそれがある。このため、フリーラン状態から再起動する場合において、モータの回転速度に依らずに、モータに流れ得る過電流を抑制する技術の確立が望まれている。なお、特許文献１は、位置センサレス制御に関する技術であるが、位置センサ付き制御においても、過電流を抑制する技術の確立は有用である。 However, Patent Document 1 does not consider restarting the motor from the free-run state in which the motor is free-running due to inertia. When restarting from the free-run state, the motor-induced voltage generated in the motor may be small depending on the rotation speed of the motor, and an overcurrent may flow in the motor. Therefore, it is desired to establish a technique for suppressing an overcurrent that can flow in a motor regardless of the rotation speed of the motor when restarting from a free-run state. Although Patent Document 1 is a technique related to position sensorless control, it is useful to establish a technique for suppressing overcurrent even in control with a position sensor.
　本開示は、上記に鑑みてなされたものであって、単相モータをフリーラン状態から再起動する際に発生し得る過電流を抑制できるモータ駆動装置を得ることを目的とする。 The present disclosure has been made in view of the above, and an object of the present invention is to obtain a motor drive device capable of suppressing an overcurrent that may occur when a single-phase motor is restarted from a free-run state.
　上述した課題を解決し、目的を達成するため、本開示に係るモータ駆動装置は、単相モータを駆動するモータ駆動装置である。モータ駆動装置は、インバータと、第１の検出器と、を備える。インバータは、直流電圧を交流電圧に変換し、変換した交流電圧を単相モータに印加する。第１の検出器は、単相モータに誘起されるモータ誘起電圧と相関のある第１の物理量を検出する。単相モータをフリーラン状態から再起動する再起動制御において、インバータが単相モータに印加するモータ印加電圧の平均値の絶対値は、単相モータを停止状態から起動する際のモータ印加電圧の平均値の絶対値よりも大きい。 The motor drive device according to the present disclosure is a motor drive device that drives a single-phase motor in order to solve the above-mentioned problems and achieve the object. The motor drive device includes an inverter and a first detector. The inverter converts the DC voltage into an AC voltage and applies the converted AC voltage to the single-phase motor. The first detector detects a first physical quantity that correlates with the motor-induced voltage induced in the single-phase motor. In the restart control that restarts the single-phase motor from the free-run state, the absolute value of the average value of the motor applied voltage applied to the single-phase motor by the inverter is the motor applied voltage when the single-phase motor is started from the stopped state. Greater than the absolute value of the average.
　本開示に係るモータ駆動装置によれば、単相モータをフリーラン状態から再起動する際に発生し得る過電流を抑制できるという効果を奏する。 According to the motor drive device according to the present disclosure, there is an effect that the overcurrent that may occur when the single-phase motor is restarted from the free-run state can be suppressed.
実施の形態１に係るモータ駆動装置を含むモータ駆動システムの構成を示すブロック図A block diagram showing a configuration of a motor drive system including a motor drive device according to the first embodiment. 実施の形態１における単相モータの構造の説明に供する断面図Sectional drawing which provides the explanation of the structure of the single-phase motor in Embodiment 1. 図２に示す単相モータを励磁した際のロータ位置の変化を示す図The figure which shows the change of the rotor position when the single-phase motor shown in FIG. 2 is excited. 図２に示す単相モータのトルク特性を示す図The figure which shows the torque characteristic of the single-phase motor shown in FIG. 図１に示すインバータの回路図Circuit diagram of the inverter shown in FIG. 図５に示すインバータの変形例を示す回路図A circuit diagram showing a modified example of the inverter shown in FIG. 図１に示す制御部の機能部位のうちのパルス幅変調（Ｐｕｌｓｅ　Ｗｉｄｔｈ　Ｍｏｄｕｌａｔｉｏｎ：ＰＷＭ）信号を生成する機能部位を示すブロック図A block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. 1. 図７に示すキャリア比較部の一例を示すブロック図A block diagram showing an example of the carrier comparison unit shown in FIG. 図８に示すキャリア比較部を用いて動作させたときの要部の波形例を示す図The figure which shows the waveform example of the main part when it operated using the carrier comparison part shown in FIG. 図７に示すキャリア比較部の他の例を示すブロック図A block diagram showing another example of the carrier comparison unit shown in FIG. 図１０に示すキャリア比較部を用いて動作させたときの要部の波形例を示す図The figure which shows the waveform example of the main part when it operated using the carrier comparison part shown in FIG. 図７に示されるキャリア比較部へ入力される進角位相を算出するための機能構成を示すブロック図A block diagram showing a functional configuration for calculating the advance phase input to the carrier comparison unit shown in FIG. 7. 実施の形態１の加速制御における要部の動作説明に使用する第１の図The first figure used for explaining the operation of the main part in the acceleration control of Embodiment 1. 実施の形態１の加速制御における要部の動作説明に使用する第２の図The second figure used for explaining the operation of the main part in the acceleration control of Embodiment 1. 実施の形態１の再起動制御における要部の動作説明に使用する第１の図The first figure used for explaining the operation of the main part in the restart control of Embodiment 1. 実施の形態１の再起動制御における要部の動作説明に使用する第２の図The second figure used for explaining the operation of the main part in the restart control of Embodiment 1. 実施の形態２に係る電気掃除機の構成図Configuration diagram of the vacuum cleaner according to the second embodiment 実施の形態２に係るハンドドライヤの構成図Configuration diagram of the hand dryer according to the second embodiment
　以下、本開示の実施の形態に係るモータ駆動装置、電動送風機、電気掃除機及びハンドドライヤを図面に基づいて詳細に説明する。 Hereinafter, the motor drive device, the electric blower, the electric vacuum cleaner, and the hand dryer according to the embodiment of the present disclosure will be described in detail with reference to the drawings.
実施の形態１．
　図１は、実施の形態１に係るモータ駆動装置２を含むモータ駆動システム１の構成を示すブロック図である。図１に示すモータ駆動システム１は、単相モータ１２と、モータ駆動装置２と、バッテリ１０と、を備える。モータ駆動装置２は、単相モータ１２に交流電力を供給して単相モータ１２を駆動する駆動装置である。バッテリ１０は、モータ駆動装置２に直流電力を供給する直流電源である。 Embodiment 1.
FIG. 1 is a block diagram showing a configuration of a motor drive system 1 including a motor drive device 2 according to the first embodiment. The motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, and a battery 10. The motor drive device 2 is a drive device that supplies AC power to the single-phase motor 12 to drive the single-phase motor 12. The battery 10 is a DC power source that supplies DC power to the motor drive device 2.
　モータ駆動装置２は、インバータ１１と、アナログディジタル変換器３０と、制御部２５と、駆動信号生成部３２とを備える。インバータ１１と単相モータ１２は、２本の接続線１８ａ，１８ｂによって接続されている。 The motor drive device 2 includes an inverter 11, an analog-digital converter 30, a control unit 25, and a drive signal generation unit 32. The inverter 11 and the single-phase motor 12 are connected by two connecting lines 18a and 18b.
　モータ駆動システム１は、電圧検出器２０，２１及び電流検出器２２，２４を備えている。モータ駆動システム１は、ロータ１２ａの回転位置を検出するための位置センサ信号を用いない、いわゆる位置センサレス制御の駆動システムである。 The motor drive system 1 includes voltage detectors 20 and 21 and current detectors 22 and 24. The motor drive system 1 is a so-called position sensorless control drive system that does not use a position sensor signal for detecting the rotation position of the rotor 12a.
　電圧検出器２０は、バッテリ１０からモータ駆動装置２に出力される直流電圧Ｖｄｃを検出する検出器である。直流電圧Ｖｄｃは、バッテリ１０の出力電圧であり、インバータ１１への印加電圧である。 The voltage detector 20 is a detector that detects the DC voltage Vdc output from the battery 10 to the motor drive device 2. The DC voltage V dc is the output voltage of the battery 10 and is the voltage applied to the inverter 11.
　電圧検出器２１は、接続線１８ａ，１８ｂ間に生じる交流電圧Ｖａｃを検出する検出器である。交流電圧Ｖａｃは、インバータ１１が単相モータ１２に印加するモータ印加電圧と、単相モータ１２によって誘起されるモータ誘起電圧とが重畳された電圧である。インバータ１１が動作を停止し、単相モータ１２が回転している場合、モータ誘起電圧が観測される。従って、電圧検出器２１の検出値は、モータ誘起電圧と相関のある物理量である。このため、本稿では、電圧検出器２１の検出値を「モータ誘起電圧と相関のある第１の物理量」と記載する場合がある。また、本稿では、インバータ１１が動作を停止し、インバータ１１が電圧を出力していない状態を「ゲートオフ」と呼ぶ。また、インバータ１１が出力する電圧を、適宜「インバータ出力電圧」と呼ぶ。 The voltage detector 21 is a detector that detects the AC voltage Vac generated between the connection lines 18a and 18b. The AC voltage V ac is a voltage obtained by superimposing the motor applied voltage applied by the inverter 11 to the single-phase motor 12 and the motor-induced voltage induced by the single-phase motor 12. When the inverter 11 is stopped and the single-phase motor 12 is rotating, the motor induced voltage is observed. Therefore, the detected value of the voltage detector 21 is a physical quantity that correlates with the motor-induced voltage. Therefore, in this paper, the detected value of the voltage detector 21 may be described as "a first physical quantity that correlates with the motor-induced voltage". Further, in this paper, the state in which the inverter 11 has stopped operating and the inverter 11 is not outputting a voltage is referred to as "gate off". Further, the voltage output by the inverter 11 is appropriately referred to as an "inverter output voltage".
　電流検出器２２は、モータ電流Ｉｍを検出する検出器である。モータ電流Ｉｍは、インバータ１１と単相モータ１２との間で流出入する交流電流である。モータ電流Ｉｍは、単相モータ１２のステータ１２ｂに巻かれている、図１では不図示の巻線に流れる交流電流に等しい。電流検出器２２には、変流器（Ｃｕｒｒｅｎｔ　Ｔｒａｎｓｆｏｒｍｅｒ：ＣＴ）、又はシャント抵抗を用いて電流を検出する電流検出器を例示できる。 The current detector 22 is a detector that detects the motor current Im . The motor current Im is an alternating current flowing in and out between the inverter 11 and the single-phase motor 12. The motor current Im is equal to the alternating current flowing through the windings (not shown in FIG. 1) wound around the stator 12b of the single-phase motor 12. Examples of the current detector 22 include a current transformer (CT) or a current detector that detects a current using a shunt resistor.
　電流検出器２４は、電源電流Ｉｄｃを検出する検出器である。電源電流Ｉｄｃは、バッテリ１０とインバータ１１との間に流れる直流電流である。電流検出器２４としては、図示のようにシャント抵抗を用いる構成が一般的である。電流検出器２４に流れる電源電流Ｉｄｃの検出値は、電圧値に変換されてアナログディジタル変換器３０に入力される。なお、本稿では、電流検出器２４の検出値を、適宜「シャント電圧」と呼ぶ。また、電源電流Ｉｄｃの検出値であるシャント電圧は、モータ電流Ｉｍと相関関係がある。即ち、モータ電流Ｉｍが増加すればシャント電圧も増加し、モータ電流Ｉｍが減少すればシャント電圧も減少する。このため、本稿では、シャント電圧を「モータ電流Ｉｍと相関のある第２の物理量」と記載する場合がある。また、本稿では、電流検出器２４を「第２の検出器」と呼ぶ場合がある。 The current detector 24 is a detector that detects the power supply current I dc . The power supply current I dc is a direct current flowing between the battery 10 and the inverter 11. The current detector 24 is generally configured to use a shunt resistor as shown in the figure. The detected value of the power supply current I dc flowing through the current detector 24 is converted into a voltage value and input to the analog-digital converter 30. In this paper, the detection value of the current detector 24 is appropriately referred to as "shunt voltage". Further, the shunt voltage, which is the detected value of the power supply current I dc , has a correlation with the motor current Im . That is, if the motor current Im increases, the shunt voltage also increases, and if the motor current Im decreases, the shunt voltage also decreases. Therefore, in this paper, the shunt voltage may be described as "a second physical quantity that correlates with the motor current Im ". Further, in this paper, the current detector 24 may be referred to as a "second detector".
　単相モータ１２は、不図示の電動送風機を回転させる回転電機として利用される。電動送風機は、電気掃除機及びハンドドライヤといった装置に搭載される。 The single-phase motor 12 is used as a rotary electric machine for rotating an electric blower (not shown). Electric blowers are mounted on devices such as vacuum cleaners and hand dryers.
　インバータ１１は、バッテリ１０から印加される直流電圧Ｖｄｃを交流電圧に変換する電力変換器である。インバータ１１は、変換した交流電圧を単相モータ１２に印加することで、単相モータ１２に交流電力を供給する。 The inverter 11 is a power converter that converts the DC voltage Vdc applied from the battery 10 into an AC voltage. The inverter 11 supplies AC power to the single-phase motor 12 by applying the converted AC voltage to the single-phase motor 12.
　アナログディジタル変換器３０は、アナログデータをディジタルデータに変換する信号変換器である。アナログディジタル変換器３０は、電圧検出器２０によって検出された直流電圧Ｖｄｃの検出値、及び電圧検出器２１によって検出された交流電圧Ｖａｃの検出値をディジタルデータに変換して制御部２５に出力する。また、アナログディジタル変換器３０は、電流検出器２２によって検出されたモータ電流Ｉｍの検出値、及び電流検出器２４によって検出され電源電流Ｉｄｃの検出値をディジタルデータに変換して制御部２５に出力する。 The analog-to-digital converter 30 is a signal converter that converts analog data into digital data. The analog-digital converter 30 converts the detected value of the DC voltage V dc detected by the voltage detector 20 and the detected value of the AC voltage V ac detected by the voltage detector 21 into digital data, and causes the control unit 25. Output. Further, the analog-digital converter 30 converts the detected value of the motor current Im detected by the current detector 22 and the detected value of the power supply current I dc detected by the current detector 24 into digital data, and the control unit 25. Output to.
　制御部２５は、アナログディジタル変換器３０で変換されたディジタル出力値３０ａと、電圧振幅指令Ｖ＊とに基づいて、ＰＷＭ信号Ｑ１，Ｑ２，Ｑ３，Ｑ４（以下、適宜「Ｑ１～Ｑ４」と表記）を生成する。電圧振幅指令Ｖ＊については、後述する。 The control unit 25 is referred to as PWM signals Q1, Q2, Q3, Q4 (hereinafter, appropriately referred to as "Q1 to Q4") based on the digital output value 30a converted by the analog digital converter 30 and the voltage amplitude command V *. ) Is generated. The voltage amplitude command V * will be described later.
　駆動信号生成部３２は、制御部２５から出力されるＰＷＭ信号Ｑ１～Ｑ４に基づいて、インバータ１１内のスイッチング素子を駆動するための駆動信号Ｓ１，Ｓ２，Ｓ３，Ｓ４（以下、適宜「Ｓ１～Ｓ４」と表記）を生成する。 The drive signal generation unit 32 has drive signals S1, S2, S3, S4 for driving the switching element in the inverter 11 based on the PWM signals Q1 to Q4 output from the control unit 25 (hereinafter, appropriately "S1 to". S4 ") is generated.
　制御部２５は、プロセッサ３１、キャリア生成部３３及びメモリ３４を有する。プロセッサ３１は、ＰＷＭ制御を行うためのＰＷＭ信号Ｑ１～Ｑ４を生成する。プロセッサ３１は、ＰＷＭ制御及び進角制御に関する各種演算を行う処理部である。プロセッサ３１としては、ＣＰＵ（Ｃｅｎｔｒａｌ　Ｐｒｏｃｅｓｓｉｎｇ　Ｕｎｉｔ）、マイクロプロセッサ、マイコン、マイクロコンピュータ、ＤＳＰ（Ｄｉｇｉｔａｌ　Ｓｉｇｎａｌ　Ｐｒｏｃｅｓｓｏｒ）、又はシステムＬＳＩ（Ｌａｒｇｅ　Ｓｃａｌｅ　Ｉｎｔｅｇｒａｔｉｏｎ）を例示できる。 The control unit 25 has a processor 31, a carrier generation unit 33, and a memory 34. The processor 31 generates PWM signals Q1 to Q4 for performing PWM control. The processor 31 is a processing unit that performs various operations related to PWM control and advance angle control. As the processor 31, a CPU (Central Processing Unit), a microprocessor, a microcomputer, a microcomputer, a DSP (Digital Signal Processor), or a system LSI (Large Scale Integration) can be exemplified.
　メモリ３４には、プロセッサ３１によって読みとられるプログラムが保存される。メモリ３４は、プロセッサ３１が演算処理を行う際の作業領域としても使用される。メモリ３４は、ＲＡＭ（Ｒａｎｄｏｍ　Ａｃｃｅｓｓ　Ｍｅｍｏｒｙ）、フラッシュメモリ、ＥＰＲＯＭ（Ｅｒａｓａｂｌｅ　Ｐｒｏｇｒａｍｍａｂｌｅ　ＲＯＭ）、ＥＥＰＲＯＭ（登録商標）（Ｅｌｅｃｔｒｉｃａｌｌｙ　ＥＰＲＯＭ）といった不揮発性又は揮発性の半導体メモリが一般的である。キャリア生成部３３の構成の詳細は後述する。 The program read by the processor 31 is stored in the memory 34. The memory 34 is also used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Project ROM), or an EEPROM (registered trademark) (Electrically EPROM). Details of the configuration of the carrier generation unit 33 will be described later.
　図２は、実施の形態１における単相モータ１２の構造の説明に供する断面図である。図２には、実施の形態で用いる単相モータ１２の一例として、単相の永久磁石ブラシレスモータのロータ１２ａ及びステータ１２ｂの断面形状が示されている。 FIG. 2 is a cross-sectional view for explaining the structure of the single-phase motor 12 in the first embodiment. FIG. 2 shows the cross-sectional shapes of the rotor 12a and the stator 12b of the single-phase permanent magnet brushless motor as an example of the single-phase motor 12 used in the embodiment.
　ロータ１２ａはシャフト１２ｃに嵌合され、図示の矢印方向、即ち反時計回りに回転可能に構成される。ロータ１２ａには、４個の永久磁石が周方向に配列されている。これらの４個の永久磁石は、着磁方向が周方向に交互に反転するように配置され、ロータ１２ａにおける磁極を形成する。なお、実施の形態１では、ロータ１２ａの磁極数が４極の場合を例示するが、ロータ１２ａの磁極数は４極以外でもよい。 The rotor 12a is fitted to the shaft 12c and is configured to be rotatable in the direction of the arrow shown in the figure, that is, counterclockwise. Four permanent magnets are arranged in the circumferential direction on the rotor 12a. These four permanent magnets are arranged so that the magnetizing directions are alternately reversed in the circumferential direction to form a magnetic pole in the rotor 12a. In the first embodiment, the case where the number of magnetic poles of the rotor 12a is 4 poles is illustrated, but the number of magnetic poles of the rotor 12a may be other than 4 poles.
　ロータ１２ａの周囲には、ステータ１２ｂが配置される。ステータ１２ｂは、４つの分割コア１２ｄが環状に連結されて構成されている。 A stator 12b is arranged around the rotor 12a. The stator 12b is configured by connecting four divided cores 12d in an annular shape.
　分割コア１２ｄは、非対称形状のティース１２ｅを有する。ティース１２ｅには、巻線１２ｆが巻回されている。ティース１２ｅは、ロータ１２ａ側に突出する第１先端部１２ｅ１及び第２先端部１２ｅ２を有する。回転方向に対し、回転方向の先にある側が第１先端部１２ｅ１であり、回転方向の後にある側が第２先端部１２ｅ２である。ここで、第１先端部１２ｅ１とロータ１２ａとの距離を「第１ギャップ」と呼び、Ｇ１で表す。また、第２先端部１２ｅ２とロータ１２ａとの距離を「第２ギャップ」と呼び、Ｇ２で表す。第１ギャップＧ１と第２ギャップＧ２との間には、Ｇ１＜Ｇ２の関係がある。 The split core 12d has an asymmetrically shaped tooth 12e. A winding 12f is wound around the teeth 12e. The teeth 12e has a first tip portion 12e1 and a second tip portion 12e2 protruding toward the rotor 12a. The side ahead of the rotation direction is the first tip portion 12e1, and the side behind the rotation direction is the second tip portion 12e2. Here, the distance between the first tip portion 12e1 and the rotor 12a is referred to as a "first gap" and is represented by G1. Further, the distance between the second tip portion 12e2 and the rotor 12a is called a "second gap" and is represented by G2. There is a relationship of G1 <G2 between the first gap G1 and the second gap G2.
　なお、単相モータ１２は、永久磁石をロータ１２ａの表面に配置する（Ｓｕｒｆａｃｅ　Ｐｅｒｍａｎｅｎｔ　Ｍａｇｎｅｔ：ＳＰＭ）構造のモータであってもよいし、永久磁石をロータ１２ａの内部に埋め込む磁石埋込型（Ｉｎｔｅｒｉｏｒ　Ｐｅｒｍａｎｅｎｔ　Ｍａｇｎｅｔ：ＩＰＭ）構造のモータであってもよい。単相モータ１２がＳＰＭ構造のモータである場合、リラクタンストルクによるトルク脈動を小さくできるという効果がある。また、単相モータ１２がＩＰＭ構造のモータである場合、永久磁石を保持する構造が容易になるという効果がある。 The single-phase motor 12 may be a motor having a structure in which a permanent magnet is arranged on the surface of the rotor 12a (Surface Permanent Magnet: SPM), or a magnet-embedded type (Interior) in which the permanent magnet is embedded inside the rotor 12a. It may be a motor having a Permanent Magnet (IPM) structure. When the single-phase motor 12 is a motor having an SPM structure, there is an effect that the torque pulsation due to the reluctance torque can be reduced. Further, when the single-phase motor 12 is a motor having an IPM structure, there is an effect that the structure for holding the permanent magnet becomes easy.
　図３は、図２に示す単相モータ１２を励磁した際のロータ位置の変化を示す図である。図４は、図２に示す単相モータ１２のトルク特性を示す図である。図３の上段部には、ロータ１２ａの停止位置が示されている。ロータ１２ａの停止位置において、磁極の中心を表す磁極中心線と、ステータ１２ｂの構造的な中心を表すティース中心線とは、回転方向に対して磁極中心線が先行するようにずれている。これは、単相モータ１２が非対称形状のティース１２ｅを有する構造であるために生ずる。この構造により、図４に示すようなトルク特性が表れる。 FIG. 3 is a diagram showing changes in the rotor position when the single-phase motor 12 shown in FIG. 2 is excited. FIG. 4 is a diagram showing torque characteristics of the single-phase motor 12 shown in FIG. The stop position of the rotor 12a is shown in the upper part of FIG. At the stop position of the rotor 12a, the magnetic pole center line representing the center of the magnetic pole and the tooth center line representing the structural center of the stator 12b are deviated so that the magnetic pole center line precedes the rotation direction. This occurs because the single-phase motor 12 has a structure having an asymmetrically shaped teeth 12e. With this structure, the torque characteristics as shown in FIG. 4 appear.
　図４において、実線で示す曲線Ｋ１はモータトルク、破線で示す曲線Ｋ２はコギングトルクを表している。モータトルクは、ステータ１２ｂの巻線に流れる電流によってロータ１２ａに発生するトルクである。コギングトルクは、ステータ１２ｂの巻線に電流が流れていないときに永久磁石の磁力によってロータ１２ａに発生するトルクである。反時計方向をトルクの正にとる。また、図４の横軸は機械角を表しており、磁極中心線がティース中心線に一致するロータ１２ａの停止位置が機械角０°である。図４に示されるように、機械角０°のときのコギングトルクは正である。このため、ロータ１２ａは反時計方向に回転し、コギングトルクがゼロとなる機械角θ１の位置で停止する。この機械角θ１の位置が、図３の上段部に示す停止位置である。 In FIG. 4, the curve K1 shown by the solid line represents the motor torque, and the curve K2 shown by the broken line represents the cogging torque. The motor torque is the torque generated in the rotor 12a by the current flowing through the winding of the stator 12b. The cogging torque is the torque generated in the rotor 12a by the magnetic force of the permanent magnet when no current is flowing in the winding of the stator 12b. Take the counterclockwise direction to the positive torque. Further, the horizontal axis of FIG. 4 represents the machine angle, and the stop position of the rotor 12a whose magnetic pole center line coincides with the teeth center line is the machine angle 0 °. As shown in FIG. 4, the cogging torque is positive when the mechanical angle is 0 °. Therefore, the rotor 12a rotates counterclockwise and stops at the position of the mechanical angle θ1 where the cogging torque becomes zero. The position of the mechanical angle θ1 is the stop position shown in the upper part of FIG.
　図２に示す単相モータ１２の場合、ロータ１２ａの停止位置は２箇所ある。停止位置の１つは、上述した図３の上段部に示す停止位置であり、もう１つは図３の下段部に示す停止位置である。巻線１２ｆに直流電圧を印加すると、反時計回りに回転し、図３の中段部に示す励磁中の状態を経て図３の下段部に示す状態で停止する。図３の例の場合、直流電圧の印加によってティース１２ｅに発生する磁力が、対向するロータ１２ａの磁極と同極であるため、回転方向にトルクがかかり、ロータ１２ａは回転する。そして、ある時間が経過し、ティース１２ｅに発生する磁力と、対向するロータ１２ａの磁極とが異極となる図３の下段部の位置で安定的に停止する。 In the case of the single-phase motor 12 shown in FIG. 2, there are two stop positions for the rotor 12a. One of the stop positions is the stop position shown in the upper part of FIG. 3 described above, and the other is the stop position shown in the lower part of FIG. When a DC voltage is applied to the winding 12f, it rotates counterclockwise, passes through the state of excitation shown in the middle part of FIG. 3, and stops in the state shown in the lower part of FIG. In the case of the example of FIG. 3, since the magnetic force generated in the teeth 12e by applying the DC voltage is the same pole as the magnetic poles of the opposing rotors 12a, torque is applied in the rotation direction and the rotor 12a rotates. Then, after a certain period of time elapses, the magnetic force generated in the teeth 12e and the magnetic poles of the rotors 12a facing each other are stably stopped at the position of the lower portion of FIG.
　図５は、図１に示すインバータ１１の回路図である。インバータ１１は、ブリッジ接続される複数のスイッチング素子５１，５２，５３，５４（以下、適宜「５１～５４」と表記）を有する。 FIG. 5 is a circuit diagram of the inverter 11 shown in FIG. The inverter 11 has a plurality of switching elements 51, 52, 53, 54 (hereinafter, appropriately referred to as “51 to 54”) to be bridge-connected.
　スイッチング素子５１，５２は、第１のレグであるレグ５Ａを構成する。レグ５Ａは、第１のスイッチング素子であるスイッチング素子５１と、第２のスイッチング素子であるスイッチング素子５２とが直列に接続された直列回路である。 The switching elements 51 and 52 constitute the first leg, the leg 5A. The leg 5A is a series circuit in which a switching element 51, which is a first switching element, and a switching element 52, which is a second switching element, are connected in series.
　スイッチング素子５３，５４は、第２のレグであるレグ５Ｂを構成する。レグ５Ｂは、第３のスイッチング素子であるスイッチング素子５３と、第４のスイッチング素子であるスイッチング素子５４とが直列に接続された直列回路である。 The switching elements 53 and 54 constitute the second leg, the leg 5B. The leg 5B is a series circuit in which a switching element 53, which is a third switching element, and a switching element 54, which is a fourth switching element, are connected in series.
　レグ５Ａ，５Ｂは、高電位側の直流母線１６ａと低電位側の直流母線１６ｂとの間に、互いに並列になるように接続される。これにより、レグ５Ａ，５Ｂは、バッテリ１０の両端に並列に接続される。 The legs 5A and 5B are connected between the DC bus 16a on the high potential side and the DC bus 16b on the low potential side so as to be in parallel with each other. As a result, the legs 5A and 5B are connected in parallel to both ends of the battery 10.
　スイッチング素子５１，５３は、高電位側に位置し、スイッチング素子５２，５４は、低電位側に位置する。一般的に、インバータ回路では、高電位側は「上アーム」と称され、低電位側は「下アーム」と称される。よって、スイッチング素子５１，５３を「上アームのスイッチング素子」と呼び、スイッチング素子５２，５４を「下アームのスイッチング素子」と呼ぶ場合がある。 The switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side. Generally, in an inverter circuit, the high potential side is referred to as an "upper arm" and the low potential side is referred to as a "lower arm". Therefore, the switching elements 51 and 53 may be referred to as "upper arm switching element", and the switching elements 52 and 54 may be referred to as "lower arm switching element".
　スイッチング素子５１とスイッチング素子５２との接続端６Ａと、スイッチング素子５３とスイッチング素子５４との接続端６Ｂとは、ブリッジ回路における交流端を構成する。接続端６Ａと接続端６Ｂとの間には、単相モータ１２が接続される。 The connection end 6A between the switching element 51 and the switching element 52 and the connection end 6B between the switching element 53 and the switching element 54 form an AC end in the bridge circuit. A single-phase motor 12 is connected between the connection end 6A and the connection end 6B.
　スイッチング素子５１～５４のそれぞれには、金属酸化膜半導体電界効果型トランジスタであるＭＯＳＦＥＴ（Ｍｅｔａｌ－Ｏｘｉｄｅ－Ｓｅｍｉｃｏｎｄｕｃｔｏｒ　Ｆｉｅｌｄ－Ｅｆｆｅｃｔ　Ｔｒａｎｓｉｓｔｏｒ）が使用される。ＭＯＳＦＥＴは、ＦＥＴ（Ｆｉｅｌｄ－Ｅｆｆｅｃｔ　Ｔｒａｎｓｉｓｔｏｒ）の一例である。 For each of the switching elements 51 to 54, a MOSFET (Metal-Oxide-Semiconductor Field-Effective Transistor), which is a metal oxide film semiconductor field effect transistor, is used. MOSFET is an example of FET (Field-Effective Transistor).
　スイッチング素子５１には、スイッチング素子５１のドレインとソースとの間に並列接続されるボディダイオード５１ａが形成される。スイッチング素子５２には、スイッチング素子５２のドレインとソースとの間に並列接続されるボディダイオード５２ａが形成される。スイッチング素子５３には、スイッチング素子５３のドレインとソースとの間に並列接続されるボディダイオード５３ａが形成される。スイッチング素子５４には、スイッチング素子５４のドレインとソースとの間に並列接続されるボディダイオード５４ａが形成される。複数のボディダイオード５１ａ，５２ａ，５３ａ，５４ａのそれぞれは、ＭＯＳＦＥＴの内部に形成される寄生ダイオードであり、還流ダイオードとして使用される。なお、別途の還流ダイオードを接続してもよい。また、ＭＯＳＦＥＴに代えて絶縁ゲートバイポーラトランジスタ（Ｉｎｓｕｌａｔｅｄ　Ｇａｔｅ　Ｂｉｐｏｌａｒ　Ｔｒａｎｓｉｓｔｏｒ：ＩＧＢＴ）を用いてもよい。 The switching element 51 is formed with a body diode 51a connected in parallel between the drain and the source of the switching element 51. The switching element 52 is formed with a body diode 52a connected in parallel between the drain and the source of the switching element 52. The switching element 53 is formed with a body diode 53a connected in parallel between the drain and the source of the switching element 53. The switching element 54 is formed with a body diode 54a connected in parallel between the drain and the source of the switching element 54. Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode. A separate freewheeling diode may be connected. Further, instead of the MOSFET, an insulated gate bipolar transistor (IGBT) may be used.
　スイッチング素子５１～５４は、シリコン系材料により形成されたＭＯＳＦＥＴに限定されず、炭化珪素、窒化ガリウム、酸化ガリウム又はダイヤモンドといったワイドバンドギャップ（Ｗｉｄｅ　Ｂａｎｄ　Ｇａｐ：ＷＢＧ）半導体により形成されたＭＯＳＦＥＴでもよい。 The switching elements 51 to 54 are not limited to MOSFETs formed of silicon-based materials, and may be MOSFETs formed of wide bandgap (Wide Band Gap: WBG) semiconductors such as silicon carbide, gallium nitride, gallium oxide, or diamond.
　一般的にＷＢＧ半導体はシリコン半導体に比べて耐電圧及び耐熱性が高い。そのため、複数のスイッチング素子５１～５４のうちの少なくとも１つにＷＢＧ半導体を用いることにより、スイッチング素子の耐電圧性及び許容電流密度が高くなり、スイッチング素子を組み込んだ半導体モジュールを小型化できる。また、ＷＢＧ半導体は、耐熱性も高い。このため、半導体モジュールで発生した熱を放熱するための放熱部の小型化が可能である。また、半導体モジュールで発生した熱を放熱する放熱構造の簡素化が可能である。 Generally, WBG semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the plurality of switching elements 51 to 54, the withstand voltage resistance and the allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element can be miniaturized. In addition, WBG semiconductors have high heat resistance. Therefore, it is possible to reduce the size of the heat radiating portion for radiating the heat generated in the semiconductor module. In addition, it is possible to simplify the heat dissipation structure that dissipates heat generated by the semiconductor module.
　また、図６は、図５に示すインバータ１１の変形例を示す回路図である。図６に示すインバータ１１Ａは、図５に示すインバータ１１の構成において、更にシャント抵抗５５ａ，５５ｂを追加したものである。シャント抵抗５５ａは、レグ５Ａに流れる電流を検出するための検出器であり、シャント抵抗５５ｂは、レグ５Ｂに流れる電流を検出するための検出器である。図６に示すように、シャント抵抗５５ａは、スイッチング素子５２の低電位側の端子と、直流母線１６ｂとの間に接続され、シャント抵抗５５ｂは、スイッチング素子５４の低電位側の端子と直流母線１６ｂとの間に接続されている。シャント抵抗５５ａ，５５ｂを備えるインバータ１１Ａを用いた場合、図１に示す電流検出器２２は、省略することができる。この構成の場合、シャント抵抗５５ａ，５５ｂの検出値は、アナログディジタル変換器３０を介してプロセッサ３１に送られる。プロセッサ３１は、シャント抵抗５５ａ，５５ｂの検出値に基づいて、後述する起動制御を実施する。 Further, FIG. 6 is a circuit diagram showing a modified example of the inverter 11 shown in FIG. The inverter 11A shown in FIG. 6 has shunt resistors 55a and 55b added to the configuration of the inverter 11 shown in FIG. The shunt resistor 55a is a detector for detecting the current flowing through the leg 5A, and the shunt resistor 55b is a detector for detecting the current flowing through the leg 5B. As shown in FIG. 6, the shunt resistor 55a is connected between the terminal on the low potential side of the switching element 52 and the DC bus 16b, and the shunt resistor 55b is connected to the terminal on the low potential side of the switching element 54 and the DC bus. It is connected to 16b. When the inverter 11A provided with the shunt resistors 55a and 55b is used, the current detector 22 shown in FIG. 1 can be omitted. In this configuration, the detected values of the shunt resistors 55a and 55b are sent to the processor 31 via the analog-digital converter 30. The processor 31 implements activation control, which will be described later, based on the detected values of the shunt resistors 55a and 55b.
　なお、シャント抵抗５５ａは、レグ５Ａに流れる電流を検出できるものであればよく、図６のものに限定されない。シャント抵抗５５ａは、直流母線１６ａとスイッチング素子５１の高電位側の端子との間、スイッチング素子５１の低電位側の端子と接続端６Ａとの間、又は接続端６Ａとスイッチング素子５２の高電位側の端子との間に配置されるものであってもよい。同様に、シャント抵抗５５ｂは、直流母線１６ａとスイッチング素子５３の高電位側の端子との間、スイッチング素子５３の低電位側の端子と接続端６Ｂとの間、又は接続端６Ｂとスイッチング素子５４の高電位側の端子との間に配置されるものであってもよい。また、シャント抵抗５５ａ，５５ｂに代え、ＭＯＦＦＥＴのオン抵抗を利用し、オン抵抗の両端に生じる電圧で電流検出を行う構成としてもよい。 The shunt resistor 55a is not limited to that of FIG. 6 as long as it can detect the current flowing through the leg 5A. The shunt resistor 55a is located between the DC bus 16a and the terminal on the high potential side of the switching element 51, between the terminal on the low potential side of the switching element 51 and the connection end 6A, or between the connection end 6A and the high potential of the switching element 52. It may be arranged between the terminal on the side. Similarly, the shunt resistor 55b is between the DC bus 16a and the terminal on the high potential side of the switching element 53, between the terminal on the low potential side of the switching element 53 and the connection end 6B, or between the connection end 6B and the switching element 54. It may be arranged between the terminal on the high potential side of the. Further, instead of the shunt resistors 55a and 55b, the on-resistance of the MOFFET may be used to detect the current with the voltage generated across the on-resistance.
　図７は、図１に示す制御部２５の機能部位のうちのＰＷＭ信号を生成する機能部位を示すブロック図である。 FIG. 7 is a block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit 25 shown in FIG.
　図７において、キャリア比較部３８には、後述する電圧指令Ｖｍを生成するときに用いる進角制御された進角位相θｖと基準位相θｅとが入力される。基準位相θｅは、ロータ１２ａの基準位置からの角度であるロータ機械角を電気角に換算した位相である。なお、前述したように、実施の形態１に係るモータ駆動装置２は、位置センサからの位置センサ信号を用いない、いわゆる位置センサレス制御の構成である。このため、ロータ機械角及び基準位相θｅは、演算によって推定される。また、ここで言う「進角位相」とは、電圧指令Ｖｍの「進み角」である「進角」を位相で表したものである。更に、ここで言う「進み角」とは、ステータ１２ｂの巻線１２ｆに印加されるモータ印加電圧と、ステータ１２ｂの巻線１２ｆに誘起されるモータ誘起電圧との間の位相差である。なお、モータ印加電圧がモータ誘起電圧よりも進んでいるときに「進み角」は正の値をとる。 In FIG. 7, the carrier comparison unit 38 is input with the advance angle controlled advance phase θ v and the reference phase θ e used when generating the voltage command V m described later. The reference phase θ e is a phase obtained by converting the rotor mechanical angle, which is the angle of the rotor 12a from the reference position, into an electric angle. As described above, the motor drive device 2 according to the first embodiment has a so-called position sensorless control configuration that does not use the position sensor signal from the position sensor. Therefore, the rotor mechanical angle and the reference phase θ e are estimated by calculation. Further, the "advance angle phase" referred to here is a phase representing the "advance angle" which is the "advance angle" of the voltage command Vm . Further, the "advance angle" referred to here is a phase difference between the motor applied voltage applied to the winding 12f of the stator 12b and the motor induced voltage induced in the winding 12f of the stator 12b. The "advance angle" takes a positive value when the voltage applied to the motor is ahead of the voltage induced by the motor.
　また、キャリア比較部３８には、進角位相θｖと基準位相θｅとに加え、キャリア生成部３３で生成されたキャリアと、直流電圧Ｖｄｃと、電圧指令Ｖｍの振幅値である電圧振幅指令Ｖ＊とが入力される。キャリア比較部３８は、キャリア、進角位相θｖ、基準位相θｅ、直流電圧Ｖｄｃ及び電圧振幅指令Ｖ＊に基づいて、ＰＷＭ信号Ｑ１～Ｑ４を生成する。 Further, in the carrier comparison unit 38, in addition to the advance phase θ v and the reference phase θ e , the carrier generated by the carrier generation unit 33, the DC voltage V dc , and the voltage which is the amplitude value of the voltage command V m . Amplitude command V * is input. The carrier comparison unit 38 generates PWM signals Q1 to Q4 based on the carrier, the advance phase θ v , the reference phase θ e , the DC voltage V dc , and the voltage amplitude command V *.
　図８は、図７に示すキャリア比較部３８の一例を示すブロック図である。図８には、キャリア比較部３８Ａ及びキャリア生成部３３の詳細構成が示されている。 FIG. 8 is a block diagram showing an example of the carrier comparison unit 38 shown in FIG. 7. FIG. 8 shows the detailed configuration of the carrier comparison unit 38A and the carrier generation unit 33.
　図８において、キャリア生成部３３には、キャリアの周波数であるキャリア周波数ｆＣ［Ｈｚ］が設定される。キャリア周波数ｆＣの矢印の先には、キャリア波形の一例として、“０”と“１”との間を上下する三角波キャリアが示される。インバータ１１のＰＷＭ制御には、同期ＰＷＭ制御と非同期ＰＷＭ制御とがある。同期ＰＷＭ制御の場合、進角位相θｖにキャリアを同期させる必要がある。一方、非同期ＰＷＭ制御の場合、進角位相θｖにキャリアを同期させる必要はない。 In FIG. 8, the carrier frequency f C [Hz], which is the frequency of the carrier, is set in the carrier generation unit 33. At the tip of the arrow of the carrier frequency f C , as an example of the carrier waveform, a triangular wave carrier moving up and down between “0” and “1” is shown. The PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the advance phase θ v . On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carrier with the advance phase θ v .
　キャリア比較部３８Ａは、図８に示すように、絶対値演算部３８ａ、除算部３８ｂ、乗算部３８ｃ、乗算部３８ｄ、乗算部３８ｆ、加算部３８ｅ、比較部３８ｇ、比較部３８ｈ、出力反転部３８ｉ及び出力反転部３８ｊを有する。 As shown in FIG. 8, the carrier comparison unit 38A has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion unit. It has 38i and an output inversion unit 38j.
　絶対値演算部３８ａでは、電圧振幅指令Ｖ＊の絶対値｜Ｖ＊｜が演算される。除算部３８ｂでは、絶対値｜Ｖ＊｜が、電圧検出器２０で検出された直流電圧Ｖｄｃによって除算される。図８の構成では、除算部３８ｂの出力が変調率となる。バッテリ１０の出力電圧であるバッテリ電圧は、電流を流し続けることにより変動する。一方、絶対値｜Ｖ＊｜を直流電圧Ｖｄｃで除算することにより、変調率の値を調整し、バッテリ電圧の低下によってモータ印加電圧が低下しないようにできる。 The absolute value calculation unit 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the division unit 38b, the absolute value | V * | is divided by the DC voltage V dc detected by the voltage detector 20. In the configuration of FIG. 8, the output of the division unit 38b is the modulation factor. The battery voltage, which is the output voltage of the battery 10, fluctuates as the current continues to flow. On the other hand, by dividing the absolute value | V * | by the DC voltage Vdc , the value of the modulation factor can be adjusted so that the motor applied voltage does not decrease due to the decrease in the battery voltage.
　乗算部３８ｃでは、基準位相θｅに進角位相θｖを加えた“θｅ＋θｖ”の正弦値が演算される。演算された“θｅ＋θｖ”の正弦値は、除算部３８ｂの出力である変調率に乗算される。乗算部３８ｄでは、乗算部３８ｃの出力である電圧指令Ｖｍに“１／２”が乗算される。加算部３８ｅでは、乗算部３８ｄの出力に“１／２”が加算される。乗算部３８ｆでは、加算部３８ｅの出力に“－１”が乗算される。加算部３８ｅの出力は、複数のスイッチング素子５１～５４のうち、上アームの２つのスイッチング素子５１，５３を駆動するための正側電圧指令Ｖｍ１として比較部３８ｇに入力され、乗算部３８ｆの出力は、下アームの２つのスイッチング素子５２，５４を駆動するための負側電圧指令Ｖｍ２として比較部３８ｈに入力される。 The multiplication unit 38c calculates a sine value of “θ e + θ v ”, which is the reference phase θ e plus the advance phase θ v . The calculated sine value of "θ e + θ v " is multiplied by the modulation factor which is the output of the division unit 38b. In the multiplication unit 38d, "1/2" is multiplied by the voltage command Vm , which is the output of the multiplication unit 38c. In the addition unit 38e, "1/2" is added to the output of the multiplication unit 38d. In the multiplication unit 38f, "-1" is multiplied by the output of the addition unit 38e. The output of the addition unit 38e is input to the comparison unit 38g as a positive voltage command Vm1 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54, and is input to the comparison unit 38g of the multiplication unit 38f. The output is input to the comparison unit 38h as a negative voltage command Vm2 for driving the two switching elements 52 and 54 of the lower arm.
　比較部３８ｇでは、正側電圧指令Ｖｍ１と、キャリアの振幅とが比較される。比較部３８ｇの出力を反転した出力反転部３８ｉの出力は、スイッチング素子５１へのＰＷＭ信号Ｑ１となり、比較部３８ｇの出力は、スイッチング素子５２へのＰＷＭ信号Ｑ２となる。同様に、比較部３８ｈでは、負側電圧指令Ｖｍ２と、キャリアの振幅とが比較される。比較部３８ｈの出力を反転した出力反転部３８ｊの出力は、スイッチング素子５３へのＰＷＭ信号Ｑ３となり、比較部３８ｈの出力は、スイッチング素子５４へのＰＷＭ信号Ｑ４となる。出力反転部３８ｉにより、スイッチング素子５１とスイッチング素子５２とが同時にオンされることはなく、出力反転部３８ｊにより、スイッチング素子５３とスイッチング素子５４とが同時にオンされることはない。 In the comparison unit 38 g, the positive voltage command V m1 and the amplitude of the carrier are compared. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, in the comparison unit 38h, the negative voltage command Vm2 and the amplitude of the carrier are compared. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time, and the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
　図９は、図８に示すキャリア比較部３８Ａを用いて動作させたときの要部の波形例を示す図である。図９には、加算部３８ｅから出力される正側電圧指令Ｖｍ１の波形と、乗算部３８ｆから出力される負側電圧指令Ｖｍ２の波形と、ＰＷＭ信号Ｑ１～Ｑ４の波形と、インバータ出力電圧の波形とが示されている。 FIG. 9 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38A shown in FIG. In FIG. 9, the waveform of the positive voltage command V m1 output from the addition unit 38e, the waveform of the negative voltage command V m2 output from the multiplication unit 38f, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown. The voltage waveform is shown.
　ＰＷＭ信号Ｑ１は、正側電圧指令Ｖｍ１がキャリアよりも大きいときに“ロー（Ｌｏｗ）”となり、正側電圧指令Ｖｍ１がキャリアよりも小さいときに“ハイ（Ｈｉｇｈ）”となる。ＰＷＭ信号Ｑ２は、ＰＷＭ信号Ｑ１の反転信号である。ＰＷＭ信号Ｑ３は、負側電圧指令Ｖｍ２がキャリアよりも大きいときに“ロー（Ｌｏｗ）”となり、負側電圧指令Ｖｍ２がキャリアよりも小さいときに“ハイ（Ｈｉｇｈ）”となる。ＰＷＭ信号Ｑ４は、ＰＷＭ信号Ｑ３の反転信号である。このように、図８に示される回路は、“ローアクティブ（Ｌｏｗ　Ａｃｔｉｖｅ）”で構成されているが、それぞれの信号が逆の値となる“ハイアクティブ（Ｈｉｇｈ　Ａｃｔｉｖｅ）”で構成されていてもよい。 The PWM signal Q1 becomes “Low” when the positive voltage command V m1 is larger than the carrier, and becomes “High” when the positive voltage command V m1 is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. The PWM signal Q3 becomes “Low” when the negative voltage command V m2 is larger than the carrier, and becomes “High” when the negative voltage command V m2 is smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 8 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.
　インバータ出力電圧の波形は、図９に示されるように、ＰＷＭ信号Ｑ１とＰＷＭ信号Ｑ４との差電圧による電圧パルスと、ＰＷＭ信号Ｑ３とＰＷＭ信号Ｑ２との差電圧による電圧パルスとが表れる。これらの電圧パルスが、モータ印加電圧として、単相モータ１２に印加される。 As shown in FIG. 9, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.
　ＰＷＭ信号Ｑ１～Ｑ４を生成する際に使用する変調方式としては、バイポーラ変調と、ユニポーラ変調とが知られている。バイポーラ変調は、電圧指令Ｖｍの１周期ごとに正又は負の電位で変化する電圧パルスを出力する変調方式である。ユニポーラ変調は、電圧指令Ｖｍの１周期ごとに３つの電位で変化する電圧パルス、即ち正の電位と負の電位と零の電位とに変化する電圧パルスを出力する変調方式である。図９に示される波形は、ユニポーラ変調によるものである。実施の形態１に係るモータ駆動装置２においては、何れの変調方式を用いてもよい。なお、モータ電流波形をより正弦波に制御する必要がある用途では、バイポーラ変調よりも、高調波含有率が少ないユニポーラ変調を採用することが好ましい。 Bipolar modulation and unipolar modulation are known as modulation methods used when generating PWM signals Q1 to Q4. Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential every cycle of the voltage command Vm . Unipolar modulation is a modulation method that outputs a voltage pulse that changes at three potentials in each cycle of the voltage command Vm , that is, a voltage pulse that changes between a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 9 is due to unipolar modulation. In the motor drive device 2 according to the first embodiment, any modulation method may be used. In applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation having a lower harmonic content than bipolar modulation.
　また、図９に示される波形は、電圧指令Ｖｍの半周期Ｔ／２の期間において、レグ５Ａを構成するスイッチング素子５１，５２と、レグ５Ｂを構成するスイッチング素子５３，５４の４つのスイッチング素子をスイッチング動作させる方式によって得られる。この方式は、正側電圧指令Ｖｍ１と負側電圧指令Ｖｍ２の双方でスイッチング動作させることから、「両側ＰＷＭ」と呼ばれる。これに対し、電圧指令Ｖｍの１周期Ｔのうちの一方の半周期では、スイッチング素子５１，５２のスイッチング動作を休止させ、電圧指令Ｖｍの１周期Ｔのうちの他方の半周期では、スイッチング素子５３，５４のスイッチング動作を休止させる方式もある。この方式は、「片側ＰＷＭ」と呼ばれる。以下、「片側ＰＷＭ」について説明する。なお、以下の説明において、両側ＰＷＭで動作させる動作モードを「両側ＰＷＭモード」と呼び、片側ＰＷＭで動作させる動作モードを「片側ＰＷＭモード」と呼ぶ。また、「両側ＰＷＭ」によるＰＷＭ信号を「両側ＰＷＭ信号」と呼び、「片側ＰＷＭ」によるＰＷＭ信号を「片側ＰＷＭ信号」と呼ぶ場合がある。 Further, the waveform shown in FIG. 9 shows four switchings of the switching elements 51 and 52 constituting the leg 5A and the switching elements 53 and 54 constituting the leg 5B during the half-cycle T / 2 period of the voltage command V m . It is obtained by a method of switching the element. This method is called "both-sided PWM" because the switching operation is performed by both the positive side voltage command V m1 and the negative side voltage command V m2 . On the other hand, in one half cycle of one cycle T of the voltage command V m , the switching operation of the switching elements 51 and 52 is suspended, and in the other half cycle of the one cycle T of the voltage command V m , the switching operation is suspended. There is also a method of suspending the switching operation of the switching elements 53 and 54. This method is called "one-sided PWM". Hereinafter, "one-sided PWM" will be described. In the following description, the operation mode operated by double-sided PWM is referred to as "double-sided PWM mode", and the operation mode operated by one-sided PWM is referred to as "one-sided PWM mode". Further, the PWM signal by "two-sided PWM" may be called "two-sided PWM signal", and the PWM signal by "one-sided PWM" may be called "one-sided PWM signal".
　図１０は、図７に示すキャリア比較部３８の他の例を示すブロック図である。図１０には、片側ＰＷＭ信号の生成回路の一例が示され、具体的には、キャリア比較部３８Ｂ及びキャリア生成部３３の詳細構成が示されている。なお、図１０に示されるキャリア生成部３３の構成は、図８に示されるものと同一又は同等である。また、図１０に示されるキャリア比較部３８Ｂの構成において、図８に示されるキャリア比較部３８Ａと同一又は同等の構成部には同一の符号を付して示している。 FIG. 10 is a block diagram showing another example of the carrier comparison unit 38 shown in FIG. 7. FIG. 10 shows an example of a one-sided PWM signal generation circuit, and specifically, a detailed configuration of a carrier comparison unit 38B and a carrier generation unit 33 is shown. The configuration of the carrier generation unit 33 shown in FIG. 10 is the same as or equivalent to that shown in FIG. Further, in the configuration of the carrier comparison unit 38B shown in FIG. 10, the same or equivalent components as the carrier comparison unit 38A shown in FIG. 8 are designated by the same reference numerals.
　キャリア比較部３８Ｂは、図１０に示されるように、絶対値演算部３８ａ、除算部３８ｂ、乗算部３８ｃ、乗算部３８ｋ、加算部３８ｍ、加算部３８ｎ、比較部３８ｇ、比較部３８ｈ、出力反転部３８ｉ及び出力反転部３８ｊを有する。 As shown in FIG. 10, the carrier comparison unit 38B has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.
　絶対値演算部３８ａでは、電圧振幅指令Ｖ＊の絶対値｜Ｖ＊｜が演算される。除算部３８ｂでは、絶対値｜Ｖ＊｜が、電圧検出器２０で検出された直流電圧Ｖｄｃによって除算される。図１０の構成でも、除算部３８ｂの出力が変調率となる。 The absolute value calculation unit 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the division unit 38b, the absolute value | V * | is divided by the DC voltage V dc detected by the voltage detector 20. Even in the configuration of FIG. 10, the output of the division unit 38b is the modulation factor.
　乗算部３８ｃでは、基準位相θｅに進角位相θｖを加えた“θｅ＋θｖ”の正弦値が演算される。演算された“θｅ＋θｖ”の正弦値は、除算部３８ｂの出力である変調率に乗算される。乗算部３８ｋでは、乗算部３８ｃの出力である電圧指令Ｖｍに“－１”が乗算される。加算部３８ｍでは、乗算部３８ｃの出力である電圧指令Ｖｍに“１”が加算される。加算部３８ｎでは、乗算部３８ｋの出力、即ち電圧指令Ｖｍの反転出力に“１”が加算される。加算部３８ｍの出力は、複数のスイッチング素子５１～５４のうち、上アームの２つのスイッチング素子５１，５３を駆動するための第１電圧指令Ｖｍ３として比較部３８ｇに入力される。加算部３８ｎの出力は、下アームの２つのスイッチング素子５２，５４を駆動するための第２電圧指令Ｖｍ４として比較部３８ｈに入力される。 The multiplication unit 38c calculates a sine value of “θ e + θ v ”, which is the reference phase θ e plus the advance phase θ v . The calculated sine value of "θ e + θ v " is multiplied by the modulation factor which is the output of the division unit 38b. In the multiplication unit 38k, "-1" is multiplied by the voltage command Vm , which is the output of the multiplication unit 38c. In the addition unit 38m, “1” is added to the voltage command Vm which is the output of the multiplication unit 38c. In the addition unit 38n, "1" is added to the output of the multiplication unit 38k, that is, the inverted output of the voltage command Vm . The output of the addition unit 38m is input to the comparison unit 38g as a first voltage command Vm3 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output of the addition unit 38n is input to the comparison unit 38h as a second voltage command Vm4 for driving the two switching elements 52 and 54 of the lower arm.
　比較部３８ｇでは、第１電圧指令Ｖｍ３と、キャリアの振幅とが比較される。比較部３８ｇの出力を反転した出力反転部３８ｉの出力は、スイッチング素子５１へのＰＷＭ信号Ｑ１となり、比較部３８ｇの出力は、スイッチング素子５２へのＰＷＭ信号Ｑ２となる。同様に、比較部３８ｈでは、第２電圧指令Ｖｍ４と、キャリアの振幅とが比較される。比較部３８ｈの出力を反転した出力反転部３８ｊの出力は、スイッチング素子５３へのＰＷＭ信号Ｑ３となり、比較部３８ｈの出力は、スイッチング素子５４へのＰＷＭ信号Ｑ４となる。出力反転部３８ｉにより、スイッチング素子５１とスイッチング素子５２とが同時にオンされることはなく、出力反転部３８ｊにより、スイッチング素子５３とスイッチング素子５４とが同時にオンされることはない。 In the comparison unit 38 g, the first voltage command V m3 and the amplitude of the carrier are compared. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, in the comparison unit 38h, the second voltage command Vm4 and the amplitude of the carrier are compared. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time, and the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
　図１１は、図１０に示すキャリア比較部３８Ｂを用いて動作させたときの要部の波形例を示す図である。図１１には、加算部３８ｍから出力される第１電圧指令Ｖｍ３の波形と、加算部３８ｎから出力される第２電圧指令Ｖｍ４の波形と、ＰＷＭ信号Ｑ１～Ｑ４の波形と、インバータ出力電圧の波形とが示されている。なお、図１１では、便宜的に、キャリアのピーク値よりも振幅値が大きくなる第１電圧指令Ｖｍ３の波形部分と、キャリアのピーク値よりも振幅値が大きくなる第２電圧指令Ｖｍ４の波形部分は、フラットな直線で表されている。 FIG. 11 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38B shown in FIG. In FIG. 11, the waveform of the first voltage command V m3 output from the adder 38 m, the waveform of the second voltage command V m4 output from the adder 38n, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown. The voltage waveform is shown. In FIG. 11, for convenience, the waveform portion of the first voltage command V m3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V m4 whose amplitude value is larger than the peak value of the carrier. The corrugated portion is represented by a flat straight line.
　ＰＷＭ信号Ｑ１は、第１電圧指令Ｖｍ３がキャリアよりも大きいときに“ロー（Ｌｏｗ）”となり、第１電圧指令Ｖｍ３がキャリアよりも小さいときに“ハイ（Ｈｉｇｈ）”となる。ＰＷＭ信号Ｑ２は、ＰＷＭ信号Ｑ１の反転信号である。ＰＷＭ信号Ｑ３は、第２電圧指令Ｖｍ４がキャリアよりも大きいときに“ロー（Ｌｏｗ）”となり、第２電圧指令Ｖｍ４がキャリアよりも小さいときに“ハイ（Ｈｉｇｈ）”となる。ＰＷＭ信号Ｑ４は、ＰＷＭ信号Ｑ３の反転信号である。このように、図１０に示される回路は、“ローアクティブ（Ｌｏｗ　Ａｃｔｉｖｅ）”で構成されているが、それぞれの信号が逆の値となる“ハイアクティブ（Ｈｉｇｈ　Ａｃｔｉｖｅ）”で構成されていてもよい。 The PWM signal Q1 becomes “Low” when the first voltage command V m3 is larger than the carrier, and becomes “High” when the first voltage command V m3 is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. The PWM signal Q3 becomes “Low” when the second voltage command V m4 is larger than the carrier, and becomes “High” when the second voltage command V m4 is smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 10 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.
　インバータ出力電圧の波形は、図１１に示されるように、ＰＷＭ信号Ｑ１とＰＷＭ信号Ｑ４との差電圧による電圧パルスと、ＰＷＭ信号Ｑ３とＰＷＭ信号Ｑ２との差電圧による電圧パルスとが表れる。これらの電圧パルスが、モータ印加電圧として、単相モータ１２に印加される。 As shown in FIG. 11, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.
　図１１に示される波形では、電圧指令Ｖｍの１周期Ｔのうちの一方の半周期では、スイッチング素子５１，５２のスイッチング動作が休止し、電圧指令Ｖｍの１周期Ｔのうちの他方の半周期では、スイッチング素子５３，５４のスイッチング動作が休止している。 In the waveform shown in FIG. 11, in one half cycle of one cycle T of the voltage command V m , the switching operation of the switching elements 51 and 52 is suspended, and the other of the one cycle T of the voltage command V m is stopped. In the half cycle, the switching operation of the switching elements 53 and 54 is suspended.
　また、図１１に示される波形では、電圧指令Ｖｍの１周期Ｔのうちの一方の半周期では、スイッチング素子５２は常時オン状態となるように制御され、電圧指令Ｖｍの１周期Ｔのうちの他方の半周期では、スイッチング素子５４は常時オン状態となるように制御される。なお、図１１は一例であり、一方の半周期では、スイッチング素子５１が常時オン状態となるように制御され、他方の半周期では、スイッチング素子５３が常時オン状態となるように制御される場合も有り得る。即ち、図１１に示される波形には、電圧指令Ｖｍの半周期において、スイッチング素子５１～５４のうちの少なくとも１つがオン状態となるように制御されるという特徴がある。 Further, in the waveform shown in FIG. 11, the switching element 52 is controlled to be always on in one half cycle of one cycle T of the voltage command V m , and the one cycle T of the voltage command V m is controlled. In the other half cycle, the switching element 54 is controlled to be always on. Note that FIG. 11 is an example, in which the switching element 51 is controlled to be always on in one half cycle, and the switching element 53 is controlled to be always on in the other half cycle. Is also possible. That is, the waveform shown in FIG. 11 is characterized in that at least one of the switching elements 51 to 54 is controlled to be in the ON state in the half cycle of the voltage command V m .
　また、図１１において、インバータ出力電圧の波形は、電圧指令Ｖｍの１周期ごとに３つの電位で変化するユニポーラ変調となる。前述の通り、ユニポーラ変調に代えてバイポーラ変調を用いてもよいが、モータ電流波形をより正弦波に制御する必要がある用途では、ユニポーラ変調を採用することが好ましい。 Further, in FIG. 11, the waveform of the inverter output voltage is unipolar modulation that changes at three potentials in each cycle of the voltage command Vm . As described above, bipolar modulation may be used instead of unipolar modulation, but in applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation.
　図１２は、図７に示されるキャリア比較部３８へ入力される進角位相θｖを算出するための機能構成を示すブロック図である。進角位相θｖの算出機能は、図１２に示されるように、回転速度算出部４２と、進角位相算出部４４とによって実現できる。回転速度算出部４２は、電流検出器２２によって検出されたモータ電流Ｉｍの検出値に基づいて単相モータ１２の回転速度ωを算出する。また、回転速度算出部４２は、モータ電流Ｉｍの検出値に基づいて、基準位相θｅを算出する。前述したように、基準位相θｅは、ロータ１２ａの基準位置からの角度であるロータ機械角を電気角に換算した位相である。ロータ機械角は、回転速度算出部４２の内部で演算される演算値である。 FIG. 12 is a block diagram showing a functional configuration for calculating the advance angle phase θ v input to the carrier comparison unit 38 shown in FIG. 7. As shown in FIG. 12, the function of calculating the advance angle phase θ v can be realized by the rotation speed calculation unit 42 and the advance angle phase calculation unit 44. The rotation speed calculation unit 42 calculates the rotation speed ω of the single-phase motor 12 based on the detection value of the motor current Im detected by the current detector 22. Further, the rotation speed calculation unit 42 calculates the reference phase θ e based on the detected value of the motor current Im. As described above, the reference phase θ e is a phase obtained by converting the rotor mechanical angle, which is the angle of the rotor 12a from the reference position, into an electric angle. The rotor machine angle is a calculated value calculated inside the rotation speed calculation unit 42.
　進角位相算出部４４は、回転速度ω、基準位相θｅ及びモータ誘起電圧に基づいて進角位相θｖを算出する。モータ誘起電圧は、交流電圧Ｖａｃの検出値により取得することができる。前述したように、交流電圧Ｖａｃの検出値には、インバータ１１が単相モータ１２に印加するモータ印加電圧と、単相モータ１２によって誘起されるモータ誘起電圧とが含まれている。これらの電圧のうち、モータ誘起電圧は、インバータ１１が電圧を出力していないゲートオフ期間に検出することができる。進角位相θｖの算出手法の詳細は、後述する。 The advance phase phase calculation unit 44 calculates the advance phase phase θ v based on the rotation speed ω, the reference phase θ e , and the motor-induced voltage. The motor induced voltage can be obtained from the detected value of the AC voltage Vac . As described above, the detected value of the AC voltage Vac includes the motor applied voltage applied to the single-phase motor 12 by the inverter 11 and the motor-induced voltage induced by the single-phase motor 12. Of these voltages, the motor-induced voltage can be detected during the gate-off period when the inverter 11 does not output the voltage. The details of the calculation method of the advance phase θ v will be described later.
　次に、実施の形態１に係るモータ駆動装置２の駆動制御の第１の要点である単相モータ１２の加速制御について、図１３及び図１４を参照して説明する。図１３は、実施の形態１の加速制御における要部の動作説明に使用する第１の図である。図１４は、実施の形態１の加速制御における要部の動作説明に使用する第２の図である。なお、上述の図２及び図３は、非対称形状のティース１２ｅを有する単相モータ１２を駆動対象とする例であるが、駆動対象の単相モータは、図２及び図３の構造のものに限定されない。即ち、本制御は、ティース１２ｅが非対称形状である場合に限定されず、ティース１２ｅが対称形状である場合においても適用可能である。 Next, the acceleration control of the single-phase motor 12, which is the first main point of the drive control of the motor drive device 2 according to the first embodiment, will be described with reference to FIGS. 13 and 14. FIG. 13 is a first diagram used for explaining the operation of a main part in the acceleration control of the first embodiment. FIG. 14 is a second diagram used for explaining the operation of the main part in the acceleration control of the first embodiment. The above-mentioned FIGS. 2 and 3 are examples in which the single-phase motor 12 having the asymmetrically shaped teeth 12e is the drive target, but the single-phase motor to be driven has the structure of FIGS. 2 and 3. Not limited. That is, this control is not limited to the case where the teeth 12e has an asymmetrical shape, and can be applied even when the teeth 12e has a symmetrical shape.
　図１３には、単相モータ１２を加速するときの回転速度が低速である場合の動作波形が示されている。また、図１４には、単相モータ１２を加速するときの回転速度が高速である場合の動作波形が示されている。ここで言う低速又は高速は、両者の相対的な関係を意味するものであり、図１３に示す第１の加速制御と図１４に示す第２の加速制御とは、予め設定された回転速度で切り替える。本稿では、この回転速度を「回転速度Ａ」と呼ぶ。単相モータ１２の回転速度が回転速度Ａ未満のときは、図１３に示す第１の加速制御で単相モータ１２を駆動する。一方、単相モータ１２の回転速度が回転速度Ａ以上のときは、図１４に示す第２の加速制御で単相モータ１２を駆動する。 FIG. 13 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is low. Further, FIG. 14 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is high. The low speed or high speed referred to here means a relative relationship between the two, and the first acceleration control shown in FIG. 13 and the second acceleration control shown in FIG. 14 are at preset rotation speeds. Switch. In this paper, this rotation speed is referred to as "rotation speed A". When the rotation speed of the single-phase motor 12 is less than the rotation speed A, the single-phase motor 12 is driven by the first acceleration control shown in FIG. On the other hand, when the rotation speed of the single-phase motor 12 is equal to or higher than the rotation speed A, the single-phase motor 12 is driven by the second acceleration control shown in FIG.
　次に、第１及び第２の加速制御の詳細について、図１３及び図１４を参照して説明する。まず、図１３を参照して第１の加速制御について説明する。 Next, the details of the first and second acceleration controls will be described with reference to FIGS. 13 and 14. First, the first acceleration control will be described with reference to FIG.
　図１３の上段部にはモータ誘起電圧の波形が示されている。図１３の下段部には、モータ印加電圧の波形と、モータ誘起電圧の波形とが示されている。下段部において、インバータ１１がゲートオンするゲートオン期間は粗いハッチングパターンで示され、インバータ１１がゲートオフするゲートオフ期間は細かなハッチングパターンで示されている。ゲートオン期間Ｔ１はモータ印加電圧の極性が正の期間であり、ゲートオン期間Ｔ２はモータ印加電圧の極性が負の期間である。ゲートオン期間Ｔ１とゲートオン期間Ｔ２との間には、ゲートオフ期間Ｔ３が存在する。 The waveform of the motor-induced voltage is shown in the upper part of FIG. In the lower part of FIG. 13, the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown. In the lower part, the gate-on period in which the inverter 11 gates on is indicated by a coarse hatch pattern, and the gate-off period in which the inverter 11 gates off is indicated by a fine hatch pattern. The gate-on period T1 is a period in which the polarity of the motor applied voltage is positive, and the gate-on period T2 is a period in which the polarity of the motor applied voltage is negative. There is a gate-off period T3 between the gate-on period T1 and the gate-on period T2.
　なお、本稿では、ゲートオン期間Ｔ１を「第１の期間」と呼び、ゲートオン期間Ｔ２を「第２の期間」と呼ぶ場合がある。また、ゲートオフ期間Ｔ３を「印加停止期間」と呼ぶ場合がある。また、Ｔ４は、単相モータ１２の回転周期の１／２の期間、即ち回転半周期を表している。回転半周期Ｔ４、ゲートオン期間Ｔ１，Ｔ２及びゲートオフ期間Ｔ３との間には、Ｔ４＝Ｔ１＋Ｔ３及びＴ４＝Ｔ２＋Ｔ３の関係がある。なお、本稿では、電気角の１周期を１回転周期として説明するが、ロータ機械角の１周期を１回転周期としてもよい。 In this paper, the gate-on period T1 may be referred to as the "first period" and the gate-on period T2 may be referred to as the "second period". Further, the gate-off period T3 may be referred to as an "application stop period". Further, T4 represents a period of ½ of the rotation cycle of the single-phase motor 12, that is, a rotation half cycle. There is a relationship of T4 = T1 + T3 and T4 = T2 + T3 between the rotation half cycle T4, the gate-on period T1, T2, and the gate-off period T3. In this paper, one cycle of the electric angle is described as one rotation cycle, but one cycle of the rotor mechanical angle may be one rotation cycle.
　前述したように、ゲートオン期間Ｔ１では、正の極性の電圧が印加される。本稿では、この極性の電圧を「第１電圧」と呼ぶ。ゲートオン期間Ｔ１は、モータ誘起電圧の極性が負から正に切り替わるゼロクロス点から始まる。また、ゲートオン期間Ｔ２では、負の極性の電圧が印加される。本稿では、この極性の電圧を「第２電圧」と呼ぶ。ゲートオン期間Ｔ２は、モータ誘起電圧の極性が正から負に切り替わるゼロクロス点から始まる。なお、図１３では、第１電圧及び第２電圧が１パルスの電圧である場合を例示しているが、これに限定されない。第１電圧及び第２電圧は、ＰＷＭ制御された複数のパルス列の電圧でもよい。 As described above, a voltage having a positive polarity is applied during the gate-on period T1. In this paper, the voltage of this polarity is called "first voltage". The gate-on period T1 begins at the zero crossing point where the polarity of the motor-induced voltage switches from negative to positive. Further, during the gate-on period T2, a voltage having a negative polarity is applied. In this paper, the voltage of this polarity is called "second voltage". The gate-on period T2 begins at the zero crossing point where the polarity of the motor-induced voltage switches from positive to negative. Note that FIG. 13 illustrates a case where the first voltage and the second voltage are voltages of one pulse, but the present invention is not limited to this. The first voltage and the second voltage may be voltages of a plurality of PWM-controlled pulse trains.
　第１の加速制御において、モータ印加電圧の極性の切り替えは、単相モータ１２の回転速度及びモータ誘起電圧に基づいて行われる。ゲートオフ期間Ｔ３では、インバータ１１がゲートオフしているので、電圧検出器２１によってモータ誘起電圧の検出が可能である。従って、モータ誘起電圧のゼロクロス点の検出も可能である。なお、ゼロクロス点は、ロータ機械角を電気角に換算した位相であり、演算によって求められている基準位相θｅを用いることも可能である。 In the first acceleration control, the polarity switching of the motor applied voltage is performed based on the rotation speed of the single-phase motor 12 and the motor-induced voltage. In the gate-off period T3, since the inverter 11 is gate-off, the motor-induced voltage can be detected by the voltage detector 21. Therefore, it is possible to detect the zero crossing point of the motor induced voltage. The zero cross point is a phase obtained by converting the rotor mechanical angle into an electric angle, and it is also possible to use the reference phase θ e obtained by calculation.
　単相モータ１２の回転速度が回転速度Ａ未満である場合、モータ誘起電圧のゼロクロス点をモータ印加電圧の極性の切り替え点とする。即ち、回転速度が回転速度Ａ未満である場合、モータ印加電圧の極性を切り替える閾値は零値に設定される。従って、モータ誘起電圧のゼロクロス点において、ゲートオン期間Ｔ１又はゲートオン期間Ｔ２が開始される。そして、ゲートオン期間Ｔ１，Ｔ２が繰り返されることにより、単相モータ１２には回転トルクが付与され、単相モータ１２は加速して回転する。なお、本稿では、この閾値を「閾値Ａ」と呼ぶ。 When the rotation speed of the single-phase motor 12 is less than the rotation speed A, the zero crossing point of the motor-induced voltage is set as the switching point of the polarity of the motor applied voltage. That is, when the rotation speed is less than the rotation speed A, the threshold value for switching the polarity of the motor applied voltage is set to a zero value. Therefore, the gate-on period T1 or the gate-on period T2 is started at the zero crossing point of the motor induced voltage. Then, by repeating the gate-on periods T1 and T2, rotational torque is applied to the single-phase motor 12, and the single-phase motor 12 accelerates and rotates. In this paper, this threshold is referred to as "threshold A".
　ゲートオン期間Ｔ１，Ｔ２の長さ、及びモータ印加電圧の振幅は、デューティ比、変調率及び回転速度に基づいて決定することができる。デューティ比は回転半周期Ｔ４に対するゲートオン期間Ｔ１，Ｔ２の比である。 The length of the gate-on periods T1 and T2 and the amplitude of the motor applied voltage can be determined based on the duty ratio, the modulation factor and the rotation speed. The duty ratio is the ratio of the gate-on periods T1 and T2 to the rotation half cycle T4.
　なお、モータ誘起電圧を電圧検出器２１にて直接検出する手法に代え、電圧検出器２０の検出値、又は電流検出器２４の検出値に基づいてモータ誘起電圧を算出してもよい。なお、電圧検出器２０の検出値を用いる場合には、バッテリ１０の出力電圧をゼロにする制御手段、又はバッテリ１０とインバータ１１との間の電気的接続を切り離す機構が必要である。 Instead of the method of directly detecting the motor-induced voltage by the voltage detector 21, the motor-induced voltage may be calculated based on the detection value of the voltage detector 20 or the detection value of the current detector 24. When the detection value of the voltage detector 20 is used, a control means for reducing the output voltage of the battery 10 to zero or a mechanism for disconnecting the electrical connection between the battery 10 and the inverter 11 is required.
　次に、図１４を参照して第２の加速制御について説明する。図１３と同様に、図１４の上段部にはモータ誘起電圧の波形が示され、下段部にはモータ印加電圧の波形とモータ誘起電圧の波形とが示されている。ゲートオン期間及びゲートオフ期間に付しているハッチングパターンも図１３と同じである。 Next, the second acceleration control will be described with reference to FIG. Similar to FIG. 13, the upper part of FIG. 14 shows the waveform of the motor-induced voltage, and the lower part shows the waveform of the motor applied voltage and the waveform of the motor-induced voltage. The hatching patterns attached to the gate-on period and the gate-off period are the same as those in FIG.
　第２の加速制御において、モータ印加電圧の極性の切り替えは、単相モータ１２の回転速度及びモータ誘起電圧に基づいて行われる。ゲートオフ期間τ３では、インバータ１１がゲートオフしているので、電圧検出器２１によってモータ誘起電圧の検出が可能である。 In the second acceleration control, the polarity of the motor applied voltage is switched based on the rotation speed of the single-phase motor 12 and the motor-induced voltage. In the gate-off period τ3, since the inverter 11 is gate-off, the motor-induced voltage can be detected by the voltage detector 21.
　ゲートオン期間τ１では、正の極性の第１電圧が印加される。ゲートオン期間τ１は、モータ誘起電圧の振幅の絶対値がΔＶに到達したときから始まる。また、ゲートオン期間τ２では、負の極性の第２電圧が印加される。ゲートオン期間τ２は、モータ誘起電圧の振幅の絶対値がΔＶに到達したときから始まる。即ち、第２の加速制御では、モータ誘起電圧の振幅の絶対値と比較するΔＶの値が閾値Ａとして設定される。インバータ１１は、モータ誘起電圧の振幅の絶対値が閾値Ａに到達する都度、単相モータ１２に印加する電圧の極性を反転する。また、第２の加速制御において、閾値Ａは、回転速度の増加に対して増加傾向となるように設定される。なお、図１４では、第１電圧及び第２電圧が１パルスの電圧である場合を例示しているが、これに限定されない。第１電圧及び第２電圧は、ＰＷＭ制御された複数のパルス列の電圧でもよい。 During the gate-on period τ1, a first voltage having a positive polarity is applied. The gate-on period τ1 starts when the absolute value of the amplitude of the motor-induced voltage reaches ΔV. Further, in the gate-on period τ2, a second voltage having a negative polarity is applied. The gate-on period τ2 starts when the absolute value of the amplitude of the motor-induced voltage reaches ΔV. That is, in the second acceleration control, the value of ΔV to be compared with the absolute value of the amplitude of the motor-induced voltage is set as the threshold value A. The inverter 11 inverts the polarity of the voltage applied to the single-phase motor 12 each time the absolute value of the amplitude of the motor-induced voltage reaches the threshold value A. Further, in the second acceleration control, the threshold value A is set so as to tend to increase with an increase in the rotation speed. Note that FIG. 14 illustrates a case where the first voltage and the second voltage are voltages of one pulse, but the present invention is not limited to this. The first voltage and the second voltage may be voltages of a plurality of PWM-controlled pulse trains.
　なお、図１４においても図１３と同様に、ゲートオン期間τ１を「第１の期間」と呼び、ゲートオン期間τ２を「第２の期間」と呼ぶ場合がある。また、ゲートオフ期間τ３を「印加停止期間」と呼ぶ場合がある。また、回転半周期τ４、ゲートオン期間τ１，τ２及びゲートオフ期間τ３との間には、τ４＝τ１＋τ３及びτ４＝τ２＋τ３の関係がある。また、ゲートオン期間τ１，τ２の長さ、及びモータ印加電圧の振幅は、デューティ比、変調率及び回転速度に基づいて決定することができる。 In FIG. 14, as in FIG. 13, the gate-on period τ1 may be referred to as a “first period” and the gate-on period τ2 may be referred to as a “second period”. Further, the gate-off period τ3 may be referred to as an “application stop period”. Further, there is a relationship of τ4 = τ1 + τ3 and τ4 = τ2 + τ3 between the rotation half cycle τ4, the gate-on period τ1, τ2, and the gate-off period τ3. Further, the length of the gate-on period τ1 and τ2 and the amplitude of the motor applied voltage can be determined based on the duty ratio, the modulation factor and the rotation speed.
　次に、実施の形態１の第１の加速制御におけるデューティ比Ｔ１／Ｔ４，Ｔ２／Ｔ４，第２の加速制御におけるデューティ比τ１／τ４，τ２／τ４、及び第２の加速制御における閾値Ａ、即ちΔＶの意義について説明する。 Next, the duty ratio T1 / T4, T2 / T4 in the first acceleration control of the first embodiment, the duty ratio τ1 / τ4, τ2 / τ4 in the second acceleration control, and the threshold value A in the second acceleration control, That is, the significance of ΔV will be described.
　まず、デューティ比Ｔ１／Ｔ４，τ１／τ４，Ｔ２／Ｔ４，τ２／τ４はモータ印加電圧に寄与し、閾値Ａ（ΔＶ）はモータ誘起電圧に対するモータ印加電圧の位相差である進角位相θｖに寄与する。加速時において、回転速度が低い低速時においては、高速時と比較して、リアクタンス成分（ωＬ）が小さい。このため、単相モータ１２に流れるモータ電流は、高速時と比較すると、低速時の方が、モータ電流に対するモータ印加電圧の位相遅れが小さい。位相遅れが小さいことは、力率が大きいことを意味する。力率が大きければ、単相モータ１２に対して、有効なモータトルクを付与することが可能となる。 First, the duty ratios T1 / T4, τ1 / τ4, T2 / T4, τ2 / τ4 contribute to the motor applied voltage, and the threshold value A (ΔV) is the phase difference of the motor applied voltage with respect to the motor induced voltage. Contribute to. At low speeds when the rotation speed is low during acceleration, the reactance component (ωL) is smaller than at high speeds. Therefore, as for the motor current flowing through the single-phase motor 12, the phase delay of the motor applied voltage with respect to the motor current is smaller at low speed than at high speed. A small phase lag means a large power factor. If the power factor is large, it is possible to apply an effective motor torque to the single-phase motor 12.
　また、高速時においては、リアクタンス成分（ωＬ）が大きくなる。このとき、モータ電流に対するモータ印加電圧の位相遅れが大きくなるが、進角位相θｖを大きくすることで、力率が小さくなるのを抑制することができる。更に、単相モータ１２に発生するモータ誘起電圧は、回転速度の増加に応じて増大する。モータ誘起電圧が大きい場合、インバータ出力電圧を大きくしても過電流を抑制することができる。このため、回転速度の増加に応じてインバータ出力電圧を大きくすることで、過電流を抑制しつつ、加速時間の短縮化を図ることが可能となる。 Further, at high speed, the reactance component (ωL) becomes large. At this time, the phase delay of the motor applied voltage with respect to the motor current becomes large, but by increasing the advance phase θ v , it is possible to suppress the decrease in the power factor. Further, the motor-induced voltage generated in the single-phase motor 12 increases as the rotation speed increases. When the motor induced voltage is large, the overcurrent can be suppressed even if the inverter output voltage is increased. Therefore, by increasing the inverter output voltage according to the increase in the rotation speed, it is possible to shorten the acceleration time while suppressing the overcurrent.
　上記の制御動作を実現するため、実施の形態１の制御では、回転速度の増加に応じて閾値Ａ（ΔＶ）が大きくなるように制御して、進角位相θｖを増加させる。この制御により、高い力率を維持できるので、単相モータ１２に付与する加速トルクを効率良く得ることができ、単相モータ１２に供給する電力を有効に活用することが可能となる。 In order to realize the above control operation, in the control of the first embodiment, the threshold value A (ΔV) is controlled to increase as the rotation speed increases, and the advance phase θ v is increased. Since a high power factor can be maintained by this control, the acceleration torque applied to the single-phase motor 12 can be efficiently obtained, and the electric power supplied to the single-phase motor 12 can be effectively utilized.
　次に、実施の形態１に係るモータ駆動装置２の駆動制御の第２の要点である単相モータ１２におけるフリーラン後の再起動制御について、図１５及び図１６を参照して説明する。図１５は、実施の形態１の再起動制御における要部の動作説明に使用する第１の図である。図１６は、実施の形態１の再起動制御における要部の動作説明に使用する第２の図である。前述したように、フリーランは、単相モータ１２が惰性で回転している状態を指している。なお、上述の図２及び図３は、非対称形状のティース１２ｅを有する単相モータ１２を駆動対象とする例であるが、駆動対象の単相モータは、図２及び図３の構造のものに限定されない。即ち、本制御は、ティース１２ｅが非対称形状である場合に限定されず、ティース１２ｅが対称形状である場合においても適用可能である。 Next, the restart control after the free run in the single-phase motor 12, which is the second main point of the drive control of the motor drive device 2 according to the first embodiment, will be described with reference to FIGS. 15 and 16. FIG. 15 is a first diagram used for explaining the operation of a main part in the restart control of the first embodiment. FIG. 16 is a second diagram used for explaining the operation of the main part in the restart control of the first embodiment. As described above, the free run refers to a state in which the single-phase motor 12 is rotating by inertia. The above-mentioned FIGS. 2 and 3 are examples in which the single-phase motor 12 having the asymmetrically shaped teeth 12e is the drive target, but the single-phase motor to be driven has the structure of FIGS. 2 and 3. Not limited. That is, this control is not limited to the case where the teeth 12e has an asymmetrical shape, and can be applied even when the teeth 12e has a symmetrical shape.
　図１５には、単相モータ１２を加速するときの回転速度が高速である場合の動作波形が示されている。また、図１６には、単相モータ１２を加速するときの回転速度が低速である場合の動作波形が示されている。ここで言う高速又は低速は、両者の相対的な関係を意味するものであり、図１５に示す第１の再起動制御と図１６に示す第２の再起動制御とは、予め設定された回転速度で切り替える。本稿では、この回転速度を「回転速度Ｂ」と呼ぶ。単相モータ１２の回転速度が回転速度Ｂ以上のときは、図１５に示す第１の再起動制御で単相モータ１２を駆動する。一方、単相モータ１２の回転速度が回転速度Ｂ未満のときは、図１６に示す第２の再起動制御で単相モータ１２を駆動する。なお、回転速度Ｂは、前述した回転速度Ａに依存しない。即ち、回転速度Ｂは、回転速度Ａと比較して、大きくても小さくてもよい。 FIG. 15 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is high. Further, FIG. 16 shows an operation waveform when the rotation speed when accelerating the single-phase motor 12 is low. The high speed or low speed referred to here means a relative relationship between the two, and the first restart control shown in FIG. 15 and the second restart control shown in FIG. 16 are preset rotation speeds. Switch by speed. In this paper, this rotation speed is referred to as "rotation speed B". When the rotation speed of the single-phase motor 12 is equal to or higher than the rotation speed B, the single-phase motor 12 is driven by the first restart control shown in FIG. On the other hand, when the rotation speed of the single-phase motor 12 is less than the rotation speed B, the single-phase motor 12 is driven by the second restart control shown in FIG. The rotation speed B does not depend on the above-mentioned rotation speed A. That is, the rotation speed B may be larger or smaller than the rotation speed A.
　次に、第１及び第２の再起動制御の詳細について、図１５及び図１６を参照して説明する。まず、図１５を参照して第１の再起動制御について説明する。なお、本稿では、回転速度Ｂを「第１の回転速度」と呼ぶこともある。 Next, the details of the first and second restart controls will be described with reference to FIGS. 15 and 16. First, the first restart control will be described with reference to FIG. In this paper, the rotation speed B may be referred to as a "first rotation speed".
　図１５の上段部には、モータ印加電圧の波形と、モータ誘起電圧の波形とが示されている。実線はモータ印加電圧であり、破線はモータ誘起電圧である。このフリーラン状態では、単相モータ１２に電圧が印加されないので、モータ誘起電圧のみを観測することが可能である。 The waveform of the motor applied voltage and the waveform of the motor induced voltage are shown in the upper part of FIG. 15. The solid line is the voltage applied to the motor, and the broken line is the voltage induced by the motor. In this free-run state, no voltage is applied to the single-phase motor 12, so it is possible to observe only the motor-induced voltage.
　図１５の中段部には、モータ印加電圧の平均値の絶対値が示されている。破線は、単相モータ１２を停止状態から起動する際のモータ印加電圧の平均値の絶対値である。また、図１５の下段部には、モータ電流の波形が示されている。 The absolute value of the average value of the motor applied voltage is shown in the middle part of FIG. The broken line is the absolute value of the average value of the motor applied voltage when the single-phase motor 12 is started from the stopped state. Further, the waveform of the motor current is shown in the lower part of FIG.
　図１５の中段部に示されるように、インバータ１１が単相モータ１２に印加するモータ印加電圧の平均値の絶対値は、単相モータ１２を停止状態から起動する際のモータ印加電圧の平均値の絶対値よりも大きくなっている。モータ電流は、モータ印加電圧とモータ誘起電圧との差分電圧に比例して大きくなる。従って、単相モータ１２の回転速度が相対的に小さくなると、モータ誘起電圧も相対的に小さくなる。逆に、単相モータ１２の回転速度が相対的に大きくなると、モータ誘起電圧も相対的に大きくなる。このため、停止状態からの起動では、モータ印加電圧の平均値が小さな値から徐々に大きくなるように制御する必要がある。これに対して、フリーラン状態からの再起動では、停止状態から起動する場合に比して、モータ印加電圧の平均値を大きくすることができる。このように制御しても、過大なモータ電流の発生を抑制することができる。 As shown in the middle part of FIG. 15, the absolute value of the average value of the motor applied voltage applied to the single-phase motor 12 by the inverter 11 is the average value of the motor applied voltage when the single-phase motor 12 is started from the stopped state. It is larger than the absolute value of. The motor current increases in proportion to the difference voltage between the motor applied voltage and the motor induced voltage. Therefore, when the rotation speed of the single-phase motor 12 becomes relatively small, the motor-induced voltage also becomes relatively small. On the contrary, when the rotation speed of the single-phase motor 12 becomes relatively large, the motor-induced voltage also becomes relatively large. Therefore, when starting from a stopped state, it is necessary to control so that the average value of the motor applied voltage gradually increases from a small value. On the other hand, in the restart from the free-run state, the average value of the motor applied voltage can be increased as compared with the case of starting from the stopped state. Even with this control, it is possible to suppress the generation of excessive motor current.
　また、モータ印加電圧の波形が図１５に示されるようなＰＷＭ波形である場合、オン時間とオフ時間とが回転半周期内で交互に繰り返される。モータ印加電圧の平均値が大きい場合、オフ時間に対するオン時間の比率が大きくなる。オフ時間はゲートオフ期間であり、オン時間はゲートオン期間である。図５のインバータ１１において、ゲートオフ期間では、スイッチング素子５１，５３又はスイッチング素子５２，５４を経由した還流電流が流れる。従って、上述した再起動制御を行えば、還流電流が流れる時間を小さくできる。これにより、過大な還流電流の発生を抑制することが可能となる。 Further, when the waveform of the motor applied voltage is a PWM waveform as shown in FIG. 15, the on time and the off time are alternately repeated within the rotation half cycle. When the average value of the motor applied voltage is large, the ratio of the on time to the off time becomes large. The off time is the gate off period and the on time is the gate on period. In the inverter 11 of FIG. 5, a reflux current flows through the switching elements 51, 53 or the switching elements 52, 54 during the gate-off period. Therefore, if the restart control described above is performed, the time for the reflux current to flow can be reduced. This makes it possible to suppress the generation of an excessive return current.
　なお、上記の第１の再起動制御について、幾つかの点を補足する。まず、フリーラン後の再起動制御では、単相モータ１２の回転速度が回転速度Ｂ以上である場合において、回転速度の増加に応じてモータ印加電圧の平均値を大きくする。一方、ある回転速度以上の運転域では、モータ印加電圧の平均値を線形に上昇させることができない。このため、ある回転速度以上では、モータ印加電圧の増加量を低減させる制御を行う。本稿では、この回転速度を「回転速度Ｃ」と呼ぶ。従って、第１の再起動制御では、単相モータ１２の回転速度が回転速度Ｂよりも大きな回転速度Ｃに達するまでのモータ印加電圧の増加率は、単相モータ１２の回転速度が回転速度Ｃに達した後のモータ印加電圧の増加率よりも小さいという特徴がある。なお、本稿では、回転速度Ｃを「第２の回転速度」と呼ぶこともある。 Note that some points are supplemented regarding the above-mentioned first restart control. First, in the restart control after the free run, when the rotation speed of the single-phase motor 12 is the rotation speed B or more, the average value of the motor applied voltage is increased according to the increase in the rotation speed. On the other hand, in the operating range above a certain rotation speed, the average value of the motor applied voltage cannot be increased linearly. Therefore, at a certain rotation speed or higher, control is performed to reduce the amount of increase in the motor applied voltage. In this paper, this rotation speed is referred to as "rotation speed C". Therefore, in the first restart control, the rate of increase of the motor applied voltage until the rotation speed of the single-phase motor 12 reaches the rotation speed C larger than the rotation speed B is such that the rotation speed of the single-phase motor 12 is the rotation speed C. It is characterized in that it is smaller than the rate of increase of the motor applied voltage after reaching. In this paper, the rotation speed C may be referred to as a "second rotation speed".
　また、第１の再起動制御において、フリーラン状態時における回転速度は、モータ誘起電圧のゼロクロス点の情報を用いて算出する。このようにすれば、回転速度の算出精度が高くなるので、より正確な磁極位置の検出が可能となる。 Further, in the first restart control, the rotation speed in the free run state is calculated using the information of the zero cross point of the motor induced voltage. By doing so, the calculation accuracy of the rotation speed becomes high, and more accurate detection of the magnetic pole position becomes possible.
　また、フリーラン後の再起動制御においては、上述した第２の加速制御と同様に、回転速度に応じて進角位相θｖを変化させるようにする。このようにすれば、第１の再起動制御時の力率悪化を抑制することが可能となる。 Further, in the restart control after the free run, the advance angle phase θ v is changed according to the rotation speed, as in the second acceleration control described above. By doing so, it is possible to suppress the deterioration of the power factor at the time of the first restart control.
　次に、図１６を参照して第２の再起動制御について説明する。前述したように図１６は、単相モータ１２の回転速度が回転速度Ｂ未満のときの再起動制御を示す図である。図１６の上段部には、モータ印加電圧の波形とモータ誘起電圧の波形とが示されている。図１６の中段部には、モータ電流の波形が示されている。図１６の下段部には、ブレーキ信号の波形が示されている。ブレーキ信号は、単相モータ１２にブレーキ力を付与するときに出力される制御信号である。実施の形態１では、インバータ１１の下アーム又は上アームのスイッチング素子に還流電流を通流させることで単相モータ１２にブレーキ力を付与することを想定している。 Next, the second restart control will be described with reference to FIG. As described above, FIG. 16 is a diagram showing restart control when the rotation speed of the single-phase motor 12 is less than the rotation speed B. In the upper part of FIG. 16, the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown. The waveform of the motor current is shown in the middle part of FIG. The waveform of the brake signal is shown in the lower part of FIG. The brake signal is a control signal output when a braking force is applied to the single-phase motor 12. In the first embodiment, it is assumed that a braking force is applied to the single-phase motor 12 by passing a reflux current through the switching element of the lower arm or the upper arm of the inverter 11.
　インバータ１１の下アーム又は上アームのスイッチング素子に還流電流を通流させることでブレーキ力を得る手法は、回転速度が低速になるほどブレーキトルクが大きくなることが知られている。従って、回転速度が回転速度Ｂ未満のときにブレーキ制御を行う本手法は、単相モータ１２を、確実且つ安定的に再起動する手法として適している。 It is known that the method of obtaining the braking force by passing a reflux current through the switching element of the lower arm or the upper arm of the inverter 11 increases the braking torque as the rotation speed becomes lower. Therefore, this method of performing brake control when the rotation speed is less than the rotation speed B is suitable as a method of reliably and stably restarting the single-phase motor 12.
　単相モータ１２の回転速度が小さくなると、観測されるモータ誘起電圧は小さくなる。実施の形態１では、モータ誘起電圧の観測精度が得られなくなる回転速度の判定値を回転速度Ｂとしている。単相モータ１２の回転速度が回転速度Ｂ未満のとき、モータ誘起電圧の波形は小さいので、正確なモータ誘起電圧を検出できないおそれがある。この状態で単相モータ１２を再起動すると、回転速度に同期した駆動トルクを単相モータ１２に付与することができず、再起動に失敗するおそれがある。また、再起動に失敗するだけでなく、単相モータ１２に過電流を生じさせ、或いは単相モータを逆方向へ回転させてしまうおそれがある。そこで、第２の再起動制御では、単相モータ１２の回転速度が回転速度Ｂ未満のとき、ブレーキ信号をオフからオンに制御し、単相モータ１２にブレーキ力を付与して単相モータ１２の回転速度を低下させる制御を行う。 As the rotation speed of the single-phase motor 12 decreases, the observed motor induced voltage decreases. In the first embodiment, the rotation speed B is defined as the determination value of the rotation speed at which the observation accuracy of the motor-induced voltage cannot be obtained. When the rotation speed of the single-phase motor 12 is less than the rotation speed B, the waveform of the motor-induced voltage is small, so that accurate motor-induced voltage may not be detected. If the single-phase motor 12 is restarted in this state, the drive torque synchronized with the rotation speed cannot be applied to the single-phase motor 12, and the restart may fail. In addition to failing to restart, there is a risk of causing an overcurrent in the single-phase motor 12 or causing the single-phase motor to rotate in the opposite direction. Therefore, in the second restart control, when the rotation speed of the single-phase motor 12 is less than the rotation speed B, the brake signal is controlled from off to on, and the braking force is applied to the single-phase motor 12 to apply the braking force to the single-phase motor 12. Controls to reduce the rotation speed of.
　単相モータ１２の再起動は、単相モータ１２が停止してから行うこともできるし、単相モータ１２に流れるモータ電流が零になったことを検出してから行うこともできる。しかしながら、これらの手法では、再起動に要する時間がかかってしまう。そこで、図１６の例では、規定時間ｔ１の間ブレーキ制御を行い、規定時間ｔ１の経過後に再起動を行うようにしている。即ち、規定時間ｔ１は、モータ電流が零と見なされるまでに低下するのに要する時間であると言える。規定時間ｔ１は、単相モータ１２の特性、第１及び第２の再起動制御の切り替えの閾値である回転速度Ｂなどに基づいて任意に設定することができる。 The restart of the single-phase motor 12 can be performed after the single-phase motor 12 is stopped, or can be performed after detecting that the motor current flowing through the single-phase motor 12 has become zero. However, with these methods, it takes time to restart. Therefore, in the example of FIG. 16, the brake is controlled during the specified time t1 and restarted after the specified time t1 has elapsed. That is, it can be said that the specified time t1 is the time required for the motor current to decrease until it is regarded as zero. The specified time t1 can be arbitrarily set based on the characteristics of the single-phase motor 12, the rotation speed B which is the threshold value for switching between the first and second restart controls, and the like.
　なお、上記の説明では、単相モータ１２の回転速度が回転速度Ｂ未満のときに、単相モータ１２に対するブレーキ制御を行うこととしているが、この例に限定されない。単相モータ１２に発生するモータ誘起電圧を直接検出し、検出したモータ誘起電圧に基づいてブレーキ制御を行ってもよい。具体的には、フリーラン時に検出されたモータ誘起電圧を閾値Ｂと比較し、モータ誘起電圧が閾値Ｂ未満であるときに、単相モータ１２に対してブレーキ制御を行うこととしてもよい。即ち、モータ誘起電圧が閾値Ｂ未満である場合、単相モータ１２は、ブレーキ力が付与された後に再起動される。一方、モータ誘起電圧が閾値Ｂ以上である場合、単相モータ１２は、ブレーキ力が付与されずに再起動される。なお、本稿では、閾値Ｂを「第１の閾値」と呼ぶ場合がある。 In the above description, when the rotation speed of the single-phase motor 12 is less than the rotation speed B, the brake control for the single-phase motor 12 is performed, but the present invention is not limited to this example. The motor-induced voltage generated in the single-phase motor 12 may be directly detected, and the brake control may be performed based on the detected motor-induced voltage. Specifically, the motor-induced voltage detected during the free run may be compared with the threshold value B, and when the motor-induced voltage is less than the threshold value B, brake control may be performed on the single-phase motor 12. That is, when the motor induced voltage is less than the threshold value B, the single-phase motor 12 is restarted after the braking force is applied. On the other hand, when the motor induced voltage is equal to or higher than the threshold value B, the single-phase motor 12 is restarted without applying the braking force. In this paper, the threshold value B may be referred to as a "first threshold value".
　なお、上記では、インバータ１１の下アーム又は上アームのスイッチング素子に還流電流を通流させることで単相モータ１２にブレーキ力を付与する手法について説明したが、単相モータ１２に対するブレーキ力の付与は、この手法に限定されず、どのような手法又は手段を用いて実施してもよい。 In the above, the method of applying the braking force to the single-phase motor 12 by passing a recirculation current through the switching element of the lower arm or the upper arm of the inverter 11 has been described, but the braking force is applied to the single-phase motor 12. Is not limited to this method, and may be carried out by any method or means.
　以上説明したように、実施の形態１に係るモータ駆動装置によれば、単相モータの回転速度が第１の回転速度未満である場合、単相モータにブレーキ力を付与した後に単相モータを再起動する。また、単相モータの回転速度が第１の回転速度以上である場合、単相モータにブレーキ力を付与せずに単相モータを再起動する。このような再起動制御により、単相モータをフリーラン状態から再起動する際に発生し得る過電流を抑制することが可能となる。また、過電流の発生を抑制できるので、単相モータにおけるロータの減磁を防止することが可能となる。 As described above, according to the motor drive device according to the first embodiment, when the rotation speed of the single-phase motor is less than the first rotation speed, the single-phase motor is used after applying a braking force to the single-phase motor. restart. When the rotation speed of the single-phase motor is equal to or higher than the first rotation speed, the single-phase motor is restarted without applying a braking force to the single-phase motor. Such restart control makes it possible to suppress the overcurrent that may occur when the single-phase motor is restarted from the free-run state. Further, since the generation of overcurrent can be suppressed, demagnetization of the rotor in the single-phase motor can be prevented.
　また、実施の形態１に係るモータ駆動装置によれば、単相モータをフリーラン状態から再起動する再起動制御において、インバータが単相モータに印加するモータ印加電圧の平均値の絶対値は、単相モータを停止状態から起動する際のモータ印加電圧の平均値の絶対値よりも大きくなるように制御される。停止状態からの起動では、モータ印加電圧の平均値が小さな値から徐々に大きくなるように制御する必要があるが、フリーラン状態からの再起動では、停止状態から起動する場合に比して、モータ印加電圧の平均値を大きくすることができる。従って、本制御によれば、過大なモータ電流の発生を抑制しつつ、再起動に要する時間の短縮化を図ることが可能となる。 Further, according to the motor drive device according to the first embodiment, in the restart control for restarting the single-phase motor from the free-run state, the absolute value of the average value of the motor applied voltage applied to the single-phase motor by the inverter is It is controlled so as to be larger than the absolute value of the average value of the motor applied voltage when the single-phase motor is started from the stopped state. When starting from a stopped state, it is necessary to control so that the average value of the motor applied voltage gradually increases from a small value, but when restarting from a free-run state, compared to when starting from a stopped state, The average value of the motor applied voltage can be increased. Therefore, according to this control, it is possible to shorten the time required for restarting while suppressing the generation of an excessive motor current.
　次に、実施の形態１に係るモータ駆動装置２において、単相モータ１２を位置センサレスで駆動することによる効果について説明する。 Next, in the motor drive device 2 according to the first embodiment, the effect of driving the single-phase motor 12 without a position sensor will be described.
　まず、適用例が電気掃除機であり、位置センサとして磁極位置センサが用いられる場合、ロータに具備される永久磁石と、磁極位置センサを備えた基板との距離が近くなる。この場合、羽根で発生させた風の流れを妨げる位置に基板を配置することとなり、風路の圧力損失を増大させてしまう。圧力損失の増大は、電気掃除機の吸い込み仕事率を悪化させ、吸引力を低下させてしまう要因となる。 First, when an application example is a vacuum cleaner and a magnetic pole position sensor is used as a position sensor, the distance between the permanent magnet provided in the rotor and the substrate provided with the magnetic pole position sensor becomes close. In this case, the substrate is arranged at a position that obstructs the flow of the wind generated by the blades, which increases the pressure loss of the air passage. The increase in pressure loss becomes a factor that deteriorates the suction power of the vacuum cleaner and lowers the suction power.
　これに対し、位置センサレスでは、位置センサを備えないことから基板配置の自由度が増えるので、基板を風路に対し平行に配置することができる。これにより、基板が風路を遮断しないので、風路の圧力損失を抑制し、吸引力を向上させることができる。その結果、電気掃除機の吸い込み仕事率を向上させることが可能となる。 On the other hand, in the case of no position sensor, since the position sensor is not provided, the degree of freedom in board placement is increased, so that the board can be placed parallel to the air passage. As a result, since the substrate does not block the air passage, the pressure loss of the air passage can be suppressed and the suction force can be improved. As a result, it becomes possible to improve the suction work rate of the vacuum cleaner.
　また、適用例が電動送風機である場合において、電動送風機が吸引した気体に水分が多く含まれている場合、基板に直接衝突する水分量が多くなる。この場合、基板に電圧を印加した際に、電極間をイオン化した金属が移動して短絡が生じるという、イオンマイグレーションの発生が懸念される。更に、塵又は埃が基板に堆積することに起因して発生する短絡が懸念される。この対策として、防湿剤を基板に塗布すること、又は基板を風路から隔離する方法が採られるが、何れも製造コストの増大を招く。 Further, in the case where the application example is an electric blower, if the gas sucked by the electric blower contains a large amount of water, the amount of water that directly collides with the substrate increases. In this case, when a voltage is applied to the substrate, ionized metal moves between the electrodes to cause a short circuit, which may cause ion migration. Further, there is a concern about a short circuit caused by dust or dust accumulating on the substrate. As a countermeasure, a method of applying a moisture-proof agent to the substrate or a method of isolating the substrate from the air passage is adopted, but both of them lead to an increase in manufacturing cost.
　これに対し、位置センサレスでは、位置センサを備えないことから基板配置の自由度が増えるので、風路を避けて基板を配置することができる。これにより、基板に直接衝突する水分量が減少するので、イオンマイグレーションの発生を抑制し、防湿剤の量を低減することができる。また、基板配置の自由度が増加しているので、基板を筺体の外部に配置することで、基板の品質を向上させることができる。 On the other hand, in the case of no position sensor, since the position sensor is not provided, the degree of freedom in board placement is increased, so that the board can be placed while avoiding the air passage. As a result, the amount of water that directly collides with the substrate is reduced, so that the occurrence of ion migration can be suppressed and the amount of the moisture-proofing agent can be reduced. Further, since the degree of freedom in arranging the substrate is increased, the quality of the substrate can be improved by arranging the substrate outside the housing.
　また、位置センサが磁極位置センサである場合、磁極位置を正しく検出するための取り付け作業の精度が要求されると共に、取り付け位置に応じた位置調整作業を実施する必要がある。このため、製造上の管理が難しくなり、設置作業を含めた製造コストの増大を招く。 Further, when the position sensor is a magnetic pole position sensor, the accuracy of the mounting work for correctly detecting the magnetic pole position is required, and it is necessary to carry out the position adjusting work according to the mounting position. For this reason, it becomes difficult to control the manufacturing, and the manufacturing cost including the installation work increases.
　これに対し、位置センサレスでは、位置センサを取り付ける設置工程、及び取り付け後の調整工程が不要であるので、製造コストの大幅な削減が可能となる。また、位置センサの経年変化による影響が発生しないので、製品の品質を向上させることができる。 On the other hand, in the case of no position sensor, the installation process for installing the position sensor and the adjustment process after installation are not required, so that the manufacturing cost can be significantly reduced. Moreover, since the influence of the secular variation of the position sensor does not occur, the quality of the product can be improved.
　更に、位置センサレスでは、位置センサを必要としないため、インバータと単相モータとを分離して構成することができる。これにより、製品適用時の制約を小さくできる。例えば、適用例が水場等で使用する製品である場合、水場等の位置からインバータを隔離して配置することができる。 Furthermore, since the position sensorless type does not require a position sensor, the inverter and the single-phase motor can be configured separately. This makes it possible to reduce the restrictions when applying the product. For example, when the application example is a product used in a water place or the like, the inverter can be isolated from the position of the water place or the like and arranged.
　また、位置センサレスの場合、電流検出器を備えた構成となる。電流検出器は、モータ電流を検出することで、軸ロック又は欠相といったモータ異常を検知可能である。これにより、位置センサが無くても、安全に停止させることができる。 Also, in the case of no position sensor, the configuration is equipped with a current detector. The current detector can detect a motor abnormality such as a shaft lock or a phase loss by detecting the motor current. This makes it possible to safely stop without a position sensor.
　なお、モータ異常を検出するには、例えば過電流を判定するための第２の閾値を設定する。そして、シャント電圧が第２の閾値に到達した場合には、モータ異常と判定する。更に、モータ異常と判定した場合には、インバータの出力を遮断する。このようにすれば、モータ異常を検出して、製品の動作を安全に停止することができる。 In order to detect a motor abnormality, for example, a second threshold value for determining an overcurrent is set. Then, when the shunt voltage reaches the second threshold value, it is determined that the motor is abnormal. Further, when it is determined that the motor is abnormal, the output of the inverter is cut off. By doing so, it is possible to detect a motor abnormality and safely stop the operation of the product.
実施の形態２．
　実施の形態２では、実施の形態１で説明したモータ駆動装置２の応用例について説明する。上述したモータ駆動装置２は、例えば電気掃除機に用いることができる。電気掃除機のように、電源の投入直後から直ぐに使用する製品の場合、実施の形態１に係るモータ駆動装置２が有する起動時間短縮による効果が大きくなる。 Embodiment 2.
In the second embodiment, an application example of the motor drive device 2 described in the first embodiment will be described. The motor drive device 2 described above can be used, for example, in a vacuum cleaner. In the case of a product such as a vacuum cleaner that is used immediately after the power is turned on, the effect of shortening the start-up time of the motor drive device 2 according to the first embodiment becomes large.
　図１７は、実施の形態２に係る電気掃除機６１の構成図である。図１７に示す電気掃除機６１は、いわゆるスティック型の電気掃除機である。図１７において、電気掃除機６１は、図１に示されるバッテリ１０と、図１に示されるモータ駆動装置２と、図１に示される単相モータ１２により駆動される電動送風機６４と、集塵室６５と、センサ６８と、吸込口体６３と、延長管６２と、操作部６６とを備える。 FIG. 17 is a configuration diagram of the vacuum cleaner 61 according to the second embodiment. The vacuum cleaner 61 shown in FIG. 17 is a so-called stick-type vacuum cleaner. In FIG. 17, the vacuum cleaner 61 includes a battery 10 shown in FIG. 1, a motor driving device 2 shown in FIG. 1, an electric blower 64 driven by a single-phase motor 12 shown in FIG. 1, and dust collector. A chamber 65, a sensor 68, a suction port 63, an extension pipe 62, and an operation unit 66 are provided.
　電気掃除機６１を使用するユーザは、操作部６６を持ち、電気掃除機６１を操作する。電気掃除機６１のモータ駆動装置２は、バッテリ１０を電源として電動送風機６４を駆動する。電動送風機６４が駆動されることにより、吸込口体６３からごみの吸込みが行われる。吸込まれたごみは、延長管６２を介して集塵室６５へ集められる。 The user who uses the vacuum cleaner 61 has an operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 10 as a power source. By driving the electric blower 64, dust is sucked from the suction port 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.
　なお、図１７では、スティック型の電気掃除機を例示したが、スティック型の電気掃除機に限定されるものではない。電動送風機を搭載した電気機器であれば、任意の製品に本開示の技術を適用できる。 Although the stick-type vacuum cleaner is illustrated in FIG. 17, it is not limited to the stick-type vacuum cleaner. The technique of the present disclosure can be applied to any product as long as it is an electric device equipped with an electric blower.
　また、図１７は、バッテリ１０を電源として用いる構成であるが、これに限定されない。バッテリ１０に代えて、コンセントから供給する交流電源を用いる構成でもよい。 Further, FIG. 17 shows a configuration in which the battery 10 is used as a power source, but the present invention is not limited to this. Instead of the battery 10, an AC power supply supplied from an outlet may be used.
　また、上述したモータ駆動装置２は、例えばハンドドライヤに用いることができる。ハンドドライヤの場合、手を挿入してから電動送風機を駆動するまでの時間が短い程、ユーザの使用感は向上する。このため、実施の形態１に係るモータ駆動装置２が有する加速時間短縮の効果が大いに発揮される。 Further, the motor drive device 2 described above can be used for, for example, a hand dryer. In the case of a hand dryer, the shorter the time from inserting the hand to driving the electric blower, the better the user's usability. Therefore, the effect of shortening the acceleration time of the motor drive device 2 according to the first embodiment is greatly exhibited.
　図１８は、実施の形態２に係るハンドドライヤ９０の構成図である。図１８において、ハンドドライヤ９０は、図１に示されるモータ駆動装置２と、ケーシング９１と、手検知センサ９２と、水受け部９３と、ドレン容器９４と、カバー９６と、センサ９７と、吸気口９８と、図１に示される単相モータ１２により駆動される電動送風機９５とを備える。ここで、センサ９７は、ジャイロセンサ及び人感センサの何れかである。ハンドドライヤ９０では、水受け部９３の上部にある手挿入部９９に手が挿入されることにより、電動送風機９５による送風で水が吹き飛ばされ、吹き飛ばされた水は、水受け部９３で集められた後、ドレン容器９４に溜められる。 FIG. 18 is a block diagram of the hand dryer 90 according to the second embodiment. In FIG. 18, the hand dryer 90 includes the motor drive device 2 shown in FIG. 1, the casing 91, the hand detection sensor 92, the water receiving unit 93, the drain container 94, the cover 96, the sensor 97, and the intake air. It includes a port 98 and an electric blower 95 driven by the single-phase motor 12 shown in FIG. Here, the sensor 97 is either a gyro sensor or a motion sensor. In the hand dryer 90, when a hand is inserted into the hand insertion portion 99 at the upper part of the water receiving portion 93, water is blown off by the blown air by the electric blower 95, and the blown water is collected by the water receiving portion 93. After that, it is stored in the drain container 94.
　上述した電気掃除機６１及びハンドドライヤ９０は、何れも実施の形態１に係るモータ駆動装置２を備えた位置センサレスの製品であるため、以下に示す効果が得られる。 Since the above-mentioned electric vacuum cleaner 61 and the hand dryer 90 are both position sensorless products equipped with the motor drive device 2 according to the first embodiment, the following effects can be obtained.
　まず、位置センサレスの構成の場合、位置センサが無くても起動することができるため、位置センサの材料費、加工費等のコストを削減することができる。また、位置センサが無いため、位置センサの取り付けずれによる性能影響を無くすことができる。これにより、安定した性能を確保することができる。 First, in the case of a position sensorless configuration, since it can be started without a position sensor, it is possible to reduce costs such as material cost and processing cost of the position sensor. Further, since there is no position sensor, it is possible to eliminate the performance effect due to the misalignment of the position sensor. As a result, stable performance can be ensured.
　また、位置センサはセンシティブなセンサであるため、位置センサの設置位置に関して、高精度な取り付け精度が要求される。また、取り付け後に位置センサの取り付け位置に応じた調整が必要になる。これに対し、位置センサレスの構成の場合、位置センサそのものが不要となり、位置センサの調整工程も排除することができる。これにより、製造コストを大幅に削減することができる。また、位置センサの経年変化による影響が発生しないため、製品の品質を向上させることができる。 Also, since the position sensor is a sensitive sensor, high-precision mounting accuracy is required for the installation position of the position sensor. In addition, after mounting, it is necessary to make adjustments according to the mounting position of the position sensor. On the other hand, in the case of the position sensorless configuration, the position sensor itself becomes unnecessary, and the adjustment step of the position sensor can be eliminated. As a result, the manufacturing cost can be significantly reduced. In addition, the quality of the product can be improved because the position sensor is not affected by the secular variation.
　また、位置センサレスの構成の場合、位置センサが不要であるため、インバータと単相モータとを分離して構成することができる。これにより、製品に対する制約を緩和することが可能となる。例えば、水分の多い水場で使用する製品の場合、製品におけるインバータの搭載位置を水場から遠い箇所に配置することができる。これにより、インバータが故障する可能性を小さくできるので、装置の信頼性を高めることができる。 Also, in the case of a position sensorless configuration, since a position sensor is not required, the inverter and the single-phase motor can be configured separately. This makes it possible to relax restrictions on the product. For example, in the case of a product used in a water place with a large amount of water, the mounting position of the inverter in the product can be arranged at a place far from the water place. As a result, the possibility of failure of the inverter can be reduced, and the reliability of the device can be improved.
　また、位置センサレスの構成の場合、位置センサに代えて配置した電流検出器により、モータ電流又はインバータ電流を検出することで、軸ロック及び欠相と言ったモータの異常を検知することができる。このため、位置センサが無くても、製品を安全に停止させることができる。 Further, in the case of the position sensorless configuration, the motor abnormality such as the shaft lock and the open phase can be detected by detecting the motor current or the inverter current by the current detector arranged instead of the position sensor. Therefore, the product can be safely stopped without the position sensor.
　以上の通り、実施の形態１，２に係るモータ駆動装置２を電気掃除機及びハンドドライヤに適用した構成例を説明したが、これらの例に限定されない。モータ駆動装置２は、モータが搭載された電気機器に広く適用することができる。モータが搭載された電気機器の例は、焼却炉、粉砕機、乾燥機、集塵機、印刷機械、クリーニング機械、製菓機械、製茶機械、木工機械、プラスチック押出機、ダンボール機械、包装機械、熱風発生機、ＯＡ機器、及び電動送風機である。電動送風機は、物体輸送用、吸塵用、又は一般送排風用の送風手段である。 As described above, a configuration example in which the motor drive device 2 according to the first and second embodiments is applied to a vacuum cleaner and a hand dryer has been described, but the present invention is not limited to these examples. The motor drive device 2 can be widely applied to an electric device on which a motor is mounted. Examples of electrical equipment equipped with motors are incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators. , OA equipment, and electric blowers. The electric blower is a blowing means for transporting an object, sucking dust, or for general blowing and exhausting.
　なお、以上の実施の形態に示した構成は、一例を示すものであり、別の公知の技術と組み合わせることも可能であるし、実施の形態同士を組み合わせることも可能であるし、要旨を逸脱しない範囲で、構成の一部を省略、変更することも可能である。 The configuration shown in the above embodiments is an example, and can be combined with another known technique, or can be combined with each other, and deviates from the gist. It is also possible to omit or change a part of the configuration to the extent that it does not.
　１　モータ駆動システム、２　モータ駆動装置、５Ａ，５Ｂ　レグ、６Ａ，６Ｂ　接続端、１０　バッテリ、１１，１１Ａ　インバータ、１２　単相モータ、１２ａ　ロータ、１２ｂ　ステータ、１２ｃ　シャフト、１２ｄ　分割コア、１２ｅ　ティース、１２ｅ１　第１先端部、１２ｅ２　第２先端部、１２ｆ　巻線、１６ａ，１６ｂ　直流母線、１８ａ，１８ｂ　接続線、２０，２１　電圧検出器、２２，２４　電流検出器、２５　制御部、３０　アナログディジタル変換器、３０ａ　ディジタル出力値、３１　プロセッサ、３２　駆動信号生成部、３３　キャリア生成部、３４　メモリ、３８，３８Ａ，３８Ｂ　キャリア比較部、３８ａ　絶対値演算部、３８ｂ　除算部、３８ｃ，３８ｄ，３８ｆ，３８ｋ　乗算部、３８ｅ，３８ｍ，３８ｎ　加算部、３８ｇ，３８ｈ　比較部、３８ｉ，３８ｊ　出力反転部、４２　回転速度算出部、４４　進角位相算出部、５１，５２，５３，５４　スイッチング素子、５１ａ，５２ａ，５３ａ，５４ａ　ボディダイオード、５５ａ，５５ｂ　シャント抵抗、６１　電気掃除機、６２　延長管、６３　吸込口体、６４，９５　電動送風機、６５　集塵室、６６　操作部、６８，９７　センサ、９０　ハンドドライヤ、９１　ケーシング、９２　手検知センサ、９３　水受け部、９４　ドレン容器、９６　カバー、９８　吸気口、９９　手挿入部。 1 motor drive system, 2 motor drive device, 5A, 5B leg, 6A, 6B connection end, 10 battery, 11, 11A inverter, 12 single-phase motor, 12a rotor, 12b stator, 12c shaft, 12d split core, 12e teeth, 12e1 1st tip, 12e2 2nd tip, 12f winding, 16a, 16b DC bus, 18a, 18b connection line, 20,21 voltage detector, 22,24 current detector, 25 control unit, 30 analog digital conversion Instrument, 30a digital output value, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 34 memory, 38, 38A, 38B carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c, 38d, 38f, 38k Multiplying unit, 38e, 38m, 38n addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 42 rotation speed calculation unit, 44 advance phase calculation unit, 51, 52, 53, 54 switching element, 51a, 52a , 53a, 54a body diode, 55a, 55b shunt resistance, 61 electric vacuum cleaner, 62 extension tube, 63 suction port, 64,95 electric blower, 65 dust collection chamber, 66 operation unit, 68,97 sensor, 90 hand dryer , 91 casing, 92 hand detection sensor, 93 water receiving part, 94 drain container, 96 cover, 98 intake port, 99 hand insertion part.

Claims (13)

  1. 　単相モータを駆動するモータ駆動装置であって、
    　直流電圧を交流電圧に変換し、変換した前記交流電圧を前記単相モータに印加するインバータと、
    　前記単相モータに誘起されるモータ誘起電圧と相関のある第１の物理量を検出する第１の検出器と、
    　を備え、
    　前記単相モータをフリーラン状態から再起動する再起動制御において、前記インバータが前記単相モータに印加するモータ印加電圧の平均値の絶対値は、前記単相モータを停止状態から起動する際の前記モータ印加電圧の平均値の絶対値よりも大きい
    　モータ駆動装置。     A motor drive that drives a single-phase motor,
    An inverter that converts a DC voltage into an AC voltage and applies the converted AC voltage to the single-phase motor.
    A first detector that detects a first physical quantity that correlates with the motor-induced voltage induced in the single-phase motor.
    Equipped with
    In the restart control for restarting the single-phase motor from the free-run state, the absolute value of the average value of the motor applied voltage applied to the single-phase motor by the inverter is when the single-phase motor is started from the stopped state. A motor drive device that is larger than the absolute value of the average value of the motor applied voltage.
  2. 　前記単相モータの回転速度が第１の回転速度未満である場合、前記単相モータにブレーキ力を付与した後に前記単相モータを再起動し、
    　前記単相モータの回転速度が前記第１の回転速度以上である場合、前記単相モータにブレーキ力を付与せずに前記単相モータを再起動する
    　請求項１に記載のモータ駆動装置。     When the rotation speed of the single-phase motor is less than the first rotation speed, the single-phase motor is restarted after applying a braking force to the single-phase motor.
    The motor drive device according to claim 1, wherein when the rotation speed of the single-phase motor is equal to or higher than the first rotation speed, the single-phase motor is restarted without applying a braking force to the single-phase motor.
  3. 　前記フリーラン状態からの再起動制御において、前記単相モータの回転速度が前記第１の回転速度以上である場合、前記モータ印加電圧は増加傾向にあり、且つ、前記単相モータの回転速度が前記第１の回転速度よりも大きな第２の回転速度に達するまでの前記モータ印加電圧の増加率は、前記単相モータの回転速度が前記第２の回転速度に達した後の前記モータ印加電圧の増加率よりも小さい
    　請求項２に記載のモータ駆動装置。     In the restart control from the free-run state, when the rotation speed of the single-phase motor is equal to or higher than the first rotation speed, the motor applied voltage tends to increase and the rotation speed of the single-phase motor is high. The rate of increase of the motor applied voltage until the second rotation speed higher than the first rotation speed is reached is the motor applied voltage after the rotation speed of the single-phase motor reaches the second rotation speed. The motor drive device according to claim 2, which is smaller than the rate of increase of.
  4. 　前記第１の物理量が第１の閾値未満である場合、前記単相モータにブレーキ力を付与した後に前記単相モータを再起動し、
    　前記第１の物理量が前記第１の閾値以上である場合、前記単相モータにブレーキ力を付与せずに前記単相モータを再起動する
    　請求項１に記載のモータ駆動装置。     When the first physical quantity is less than the first threshold value, the single-phase motor is restarted after applying a braking force to the single-phase motor.
    The motor drive device according to claim 1, wherein when the first physical quantity is equal to or greater than the first threshold value, the single-phase motor is restarted without applying a braking force to the single-phase motor.
  5. 　前記単相モータに対するブレーキ力は、前記インバータの下アームのスイッチング素子に還流電流を通流させ、又は前記インバータの上アームのスイッチング素子に還流電流を通流させることで得られる
    　請求項２から４の何れか１項に記載のモータ駆動装置。     The braking force for the single-phase motor is obtained by passing a recirculation current through the switching element of the lower arm of the inverter or passing a recirculation current through the switching element of the upper arm of the inverter. The motor drive device according to any one of the above items.
  6. 　前記単相モータにブレーキ力を付与した後は、規定時間の経過後に前記単相モータを再起動する
    　請求項２から５の何れか１項に記載のモータ駆動装置。     The motor drive device according to any one of claims 2 to 5, wherein after applying a braking force to the single-phase motor, the single-phase motor is restarted after a lapse of a predetermined time.
  7. 　前記単相モータは位置センサレスで駆動される
    　請求項１から６の何れか１項に記載のモータ駆動装置。     The motor driving device according to any one of claims 1 to 6, wherein the single-phase motor is driven without a position sensor.
  8. 　前記単相モータに流れる電流と相関のある第２の物理量を検出する第２の検出器を備え、
    　前記第２の物理量が第２の閾値に到達した際には、前記インバータは動作を停止する
    　請求項１から７の何れか１項に記載のモータ駆動装置。     A second detector for detecting a second physical quantity that correlates with the current flowing through the single-phase motor is provided.
    The motor drive device according to any one of claims 1 to 7, wherein the inverter stops operating when the second physical quantity reaches the second threshold value.
  9. 　前記インバータのスイッチング素子のうちの少なくとも１つはワイドバンドギャップ半導体で形成されている
    　請求項１から８の何れか１項に記載のモータ駆動装置。     The motor drive device according to any one of claims 1 to 8, wherein at least one of the switching elements of the inverter is formed of a wide bandgap semiconductor.
  10. 　前記ワイドバンドギャップ半導体は、炭化珪素、窒化ガリウム、酸化ガリウム又はダイヤモンドである
    　請求項９に記載のモータ駆動装置。     The motor drive device according to claim 9, wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond.
  11. 　請求項１から１０の何れか１項に記載のモータ駆動装置を備えた電動送風機。 An electric blower provided with the motor drive device according to any one of claims 1 to 10.
  12. 　請求項１１に記載の電動送風機を備えた電気掃除機。 A vacuum cleaner provided with the electric blower according to claim 11.
  13. 　請求項１１に記載の電動送風機を備えたハンドドライヤ。 A hand dryer equipped with the electric blower according to claim 11.
PCT/JP2020/039275 2020-10-19 2020-10-19 Motor driving device, electric blower, electric vacuum cleaner, and hand dryer WO2022085050A1 (en)

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