WO2021081968A1 - 基于mtpa无参数无位置传感的永磁同步电机控制方法 - Google Patents

基于mtpa无参数无位置传感的永磁同步电机控制方法 Download PDF

Info

Publication number
WO2021081968A1
WO2021081968A1 PCT/CN2019/114919 CN2019114919W WO2021081968A1 WO 2021081968 A1 WO2021081968 A1 WO 2021081968A1 CN 2019114919 W CN2019114919 W CN 2019114919W WO 2021081968 A1 WO2021081968 A1 WO 2021081968A1
Authority
WO
WIPO (PCT)
Prior art keywords
mtpa
angle
current
permanent magnet
idq
Prior art date
Application number
PCT/CN2019/114919
Other languages
English (en)
French (fr)
Inventor
刘宁
郭伟
杨妍
Original Assignee
中山大洋电机股份有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 中山大洋电机股份有限公司 filed Critical 中山大洋电机股份有限公司
Priority to CA3154822A priority Critical patent/CA3154822A1/en
Priority to MX2022005162A priority patent/MX2022005162A/es
Publication of WO2021081968A1 publication Critical patent/WO2021081968A1/zh
Priority to US17/686,391 priority patent/US11689132B2/en

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/26—Rotor flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
    • H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
    • H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/34—Modelling or simulation for control purposes
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Definitions

  • the invention relates to a permanent magnet synchronous motor control method based on MTPA without parameters and position sensing.
  • US patent US7525269 discloses a position sensorless 3-phase synchronous motor vector controller, which only discloses the current torque control mode for constant torque control.
  • Chinese patent CN103929109(A) also discloses a constant speed control method based on a position sensorless vector control permanent magnet synchronous motor.
  • the most conventional FOC theory of the position sensorless vector control permanent magnet synchronous motor control method is mostly based on the rotor frame.
  • the position sensorless algorithm will be used to estimate the rotor position, so the FOC theory can continue to be used.
  • Estimating the rotor position adopts theoretical and motor-related parameters.
  • the mathematical model is complex, the calculation is time-consuming and complicated, and it takes up a lot of control chip (microprocessor MCU) resources. It requires high microprocessor MCU and high cost.
  • the FOC theory has a high degree of dependence on motor parameters: motor resistance Rs, inductance Lq, Lq and magnetic flux ⁇ m, resulting in a narrower application range.
  • the purpose of the present invention is to provide a permanent magnet synchronous motor control method based on MTPA without parameters and position sensing, which solves the problem of conventional FOC theory in the prior art for estimating rotor position.
  • the theory is closely related to the relevant parameters of the motor, the mathematical model is complicated, and the calculation is time-consuming. , Occupies a lot of control chip resources, high requirements for the microprocessor MCU, and high cost technical problems.
  • the permanent magnet synchronous motor control method based on MTPA without parameters and position sensing is characterized in that it contains a current control mode, which includes the following steps:
  • Step 1 Receive user input for given current Idq* and ⁇ angle, ⁇ angle is the angle between current vector Idq* and q axis, and calculate Id_r* and Iq_r*, said Id_r* and Iq_r* are current vector Idq* The current value of the d-axis and the current value of the q-axis are projected on the rotor rotating coordinate system dq.
  • the given current Idq* and ⁇ angle are data conforming to the MTPA mode;
  • Step 2 Find the corresponding ⁇ angle in the MTPA data table according to Id_r* and Iq_r*.
  • the ⁇ angle is the angle between the rotor coordinate system dq and the voltage coordinate system VdVq.
  • the MTPA data table refers to the maximum torque Data obtained in the mode per ampere;
  • Step 3 Use ⁇ angle, Id_r* and Iq_r* to calculate Id_Ref and Iq_Ref.
  • Id_Ref and Iq_Ref are the projections of the current vector Idq* in the voltage coordinate system VdVq;
  • Step 4 Input the current Iq_Ref and the real-time feedback current Iq of the motor into the PLL phase-locked loop to obtain ⁇ v, ⁇ v is the angle between the voltage vector and the static coordinate system ⁇ , using the current Id_Ref and the real-time motor running feedback current Id.
  • Id_ref Id_r* ⁇ cos( ⁇ )+Iq_r* ⁇ sin( ⁇ )
  • Iq_ref -Id_r* ⁇ sin( ⁇ )+Iq_r* ⁇ cos( ⁇ ).
  • the above-mentioned MTPA data table is data obtained through experiments or data obtained through theoretical calculations or data obtained through computer finite element analysis software.
  • the permanent magnet synchronous motor control method based on MTPA without parameters and position sensing is characterized in that it contains a speed control mode, which includes the following steps:
  • Step 1 Receive the user input given speed spd command and ⁇ angle, make the vector angle of the voltage vector Vdq rotate at the given speed spd to obtain ⁇ v, ⁇ v is the angle between the voltage vector and the static coordinate system ⁇ , and the ⁇ angle is the current vector The angle between Idq and q axis;
  • Step 2 Calculate Id_r* and Iq_r* according to the feedback current Id and ⁇ angle of the motor running in real time.
  • the Id_r* and Iq_r* are the current value of the d-axis and the q-axis of the current current vector Idq in the rotor rotation coordinate system dq Id_r* and Iq_r* are data conforming to the MTPA mode.
  • Use Id_r* and Iq_r* to find the corresponding ⁇ angle in the MTPA data table.
  • ⁇ is the angle between the voltage vector Vdq and the current vector Idq in the MTPA mode, so that The current vector Idq coincides with the Vd axis to obtain the angle ⁇ i;
  • the above-mentioned MTPA data table is data obtained through experiments or data obtained through theoretical calculations or data obtained through computer finite element analysis software.
  • the above-mentioned step 2 is to use the feedback current Iq of the real-time operation of the motor as one input of the PLL phase-locked loop, and set the other input Iq* of the PLL phase-locked loop to 0.
  • Iq* is the current vector Idq in the VdVq coordinate system Vq
  • spd is the speed value
  • Pole_pair is the number of pole pairs of the motor
  • ⁇ t is the time variable
  • the PI regulator In the speed control mode, when the voltage value Vq is greater than or equal to the set threshold value Vmax, the PI regulator enters the saturation state, its output is limited to Vmax, and automatically switches to the field weakening control mode.
  • the present invention has the following beneficial effects:
  • the optimization of the motor is to make its current run along the calibratable MTPA track.
  • the motor has a full-load start function, and its operating range includes from no BEMF to field weakening control, with complete functions.
  • the present invention is based on the MTPA parameterless and position sensing permanent magnet synchronous motor control method.
  • the PLSL-MTPA mathematical model is no longer based on a single rotor coordinate system.
  • the algorithm projects the motor current vector on the current and voltage coordinates at the same time. In the system, the positionless control is completed by analyzing the vector angle.
  • the mathematical model is simple, the algorithm is simple, and the operation is simple. It does not occupy a lot of control chip resources. It does not require high microprocessor MCUs, which is beneficial to reduce costs.
  • Figure 1 is a traditional FOC control block diagram of a position sensorless vector control permanent magnet synchronous motor.
  • Figure 2 is a perspective view of the permanent magnet synchronous motor of the present invention
  • Figure 3 is a perspective view of the motor controller of the permanent magnet synchronous motor of the present invention.
  • Figure 4 is a cross-sectional view of the permanent magnet synchronous motor of the present invention.
  • Figure 5 is a schematic block diagram of the motor controller of the permanent magnet synchronous motor of the present invention.
  • Fig. 6 is a circuit diagram corresponding to Fig. 5;
  • Fig. 7 is a schematic diagram of the static coordinate system ABC of a three-phase permanent magnet synchronous motor
  • Fig. 8 is a schematic diagram of a stationary orthogonal coordinate system ⁇ of a three-phase permanent magnet synchronous motor
  • Figure 9 is a diagram of the relationship between the coordinate systems of the vector control of the three-phase permanent magnet synchronous motor
  • Fig. 11 is a diagram of three time-domain variables at the same frequency of ⁇ v, ⁇ i, and ⁇ r of the present invention.
  • FIG. 12 is a diagram showing the angle relationship between the marked voltage vector and the current vector of the present invention.
  • FIG. 13 is a schematic diagram of the principle of the MTPA mode of the present invention.
  • Figure 14 is a schematic diagram of the principle of the current control mode of the present invention.
  • Figure 15 is a schematic block diagram of the current control mode of the present invention.
  • Figure 16 is a schematic block diagram of the speed control mode of the present invention.
  • Figure 17 is an analysis data diagram corresponding to the angle ⁇ and Id_r* and Iq_r* of the present invention.
  • Fig. 18 is a graph of analysis data corresponding to the angle ⁇ of the present invention and Id_r* and Iq_r*.
  • FIG. 2 Figure 3, Figure 4, for example: suppose the present invention is a three-phase permanent magnet synchronous motor, consisting of a motor controller 2 and a motor unit 1, and the motor unit 1 includes a stator assembly 12, The rotor assembly 13 and the housing assembly 11, the stator assembly 13 is installed on the housing assembly 11, the rotor assembly 13 is sleeved on the inside or outside of the stator assembly 12, and the motor controller 2 includes a control box 22 and a control box 22 installed inside.
  • the control circuit board 21 generally includes a power supply circuit, a microprocessor, a bus voltage detection circuit, and an inverter.
  • the power supply circuit supplies power to each part of the circuit.
  • the bus voltage detection circuit inputs the DC bus voltage Uabc to the microprocessor.
  • the microprocessor controls the inverter, and the inverter controls the on and off of each phase coil winding of the stator assembly 12.
  • the phase line current detection circuit of a 3-phase brushless DC permanent magnet synchronous motor inputs the currents Ia, Ib, and Ic of each phase to the microprocessor.
  • the DC bus voltage Vdc bus is output at one end of the capacitor C1.
  • the DC bus voltage Vdc bus is related to the input AC voltage.
  • the processor inputs the PWM signal to the inverter.
  • the inverter is composed of electronic switch tubes Q1, Q2, Q3, Q4, Q5, and Q6.
  • the control ends of the electronic switch tubes Q1, Q2, Q3, Q4, Q5, and Q6 are respectively composed of Controlled by 6 PWM signals (P1, P2, P3, P4, P5, P6) output by the microprocessor.
  • the three-phase motor currents are Ia, Ib, Ic.
  • There is a phase angle of 120 degrees in the time domain which is generally called a stationary coordinate system.
  • These three time-domain current quantities can be simplified into two orthogonal current quantities I ⁇ and I ⁇ , as shown in Fig. 8. It is described by a triangular vector diagram, as shown in Figure 9. The mathematical relationship is:
  • I d I ⁇ *cos( ⁇ )+I ⁇ *sin( ⁇ )
  • ⁇ is the azimuth of our observation of I ⁇ and I ⁇ .
  • the projections on different rotating platforms that is, coordinate systems
  • the Park Transformation changes the forward rotations I ⁇ , I ⁇ into direct current Id, Iq.
  • the north pole of the rotor magnetic field is usually positioned at 0 degrees.
  • One rotation of the rotor is 360 degrees, and the relationship between the position of the rotor and the number of pole pairs is as follows:
  • ⁇ r Pole_pair ⁇ 0, where ⁇ 0 is the mechanical angle of the rotor, and Pole_pair is the number of pole pairs of the motor.
  • the present invention is based on a permanent magnet synchronous motor control method with no parameter and no position sensor based on MTPA [PLSL-MTPA algorithm (parameterless sensorless--MTPA)].
  • PLSL-MTPA we use another method to analyze I ⁇ and I ⁇ . That is to say, use different ⁇ angles to build a rotating platform to analyze I ⁇ , I ⁇ .
  • the working principle of the phase-locked loop is just the opposite of the Park transformation. The latter is to change the positive spinner into a direct current by using an angle.
  • the phase-locked loop is to find out the ⁇ angle to establish the rotating platform by locking one of the direct currents.
  • the present invention introduces ⁇ v and ⁇ i respectively representing the angle between the voltage vector and the current vector and the stationary coordinate system ABC, as shown in FIG. 12.
  • ⁇ v is derived from open loop speed integration. It is used to generate V ⁇ , V ⁇ , and then generate Va, Vb, Vc.
  • Vabc is the combined voltage vector of A-phase, B-phase, and C-phase windings
  • Iabc is the combined current vector of A-phase, B-phase, and C-phase windings
  • w is the angular velocity
  • t is time
  • ⁇ is the voltage vector Va(t) of A-phase. The angle with the current vector Ia(t).
  • ⁇ v, ⁇ i, ⁇ r are the relationship between voltage and current in the time domain as shown in Figure 11.
  • the projection of the current vector Idq of I ⁇ and I ⁇ on the q axis is related to the current
  • the projection of the vector Idq on the Vq axis of the VdVq coordinate system can be processed by the PLL phase-locked loop to obtain ⁇ v.
  • the projection of the current vector Idq of I ⁇ and I ⁇ on the d axis is the same as the Vd axis of the current vector Idq on the VdVq coordinate system.
  • Pole_pair is the number of pole pairs of the motor
  • ⁇ t is the time variable
  • the voltage vector Vdq PI( ⁇ iv- ⁇ ), since the ⁇ v and Vdq are obtained, the positive spinner can be turned into a direct current to obtain V ⁇ , V ⁇ .
  • the current PI device in the above two control modes uses angle or current error control to obtain Vdq, among which: speed mode:
  • Vdq PI( ⁇ ), and Vabc is generated by ⁇ v and Vdq.
  • the PLSL-MTPA static full-load start of the present invention start with the maximum current Idq_Max.
  • Idq_Max the current PI controller tends to be saturated with the maximum current, and Idq_Max is used to drive the motor .
  • the back electromotive force also increases. This also causes the current-voltage difference angle ⁇ to gradually become non-zero. So that the current PI controller enters the normal working range, the difference between Vdq and Idq varies with the actual load, and the motor current responds accordingly.
  • the starting ability depends on the target speed, the speed increasing slope, the current PI gain, and the maximum current limit.
  • the PLSL-MTPA control method of the present invention complies with 4 control laws:
  • the adjustable angle ⁇ is maintained between the current vector and the voltage vector, which can realize the comprehensive control of the synchronous motor.
  • the ⁇ angle is also commonly referred to as the power factor angle.
  • the rotor position of the synchronous motor can be determined by the voltage vector plus a controllable angle ⁇ .
  • the ⁇ angle may lead or lag the voltage vector. Based on the given ⁇ angle, implement a voltage vector to the synchronous motor, and its synchronization condition can be sustained, that is, the synchronous motor is controllable.
  • the maximum torque per unit current MTPA control can be completed by converting current commands into ⁇ and ⁇ angles.
  • the generation of the ⁇ and ⁇ angles follows the MTPA principle, and is used by the control law 1 and the control law 2 to control the synchronous motor.
  • the rotor position estimated by the control law 2 is the actual motor rotor position.
  • the torque generated by the motor is the MTPA torque.
  • the ratio of voltage to speed can be approximated as BEMF.
  • the operating state of the synchronous motor can be determined by comparing this value with a threshold.
  • the speed mode of PLSL-MTPA is actually an open-loop speed control that can carry a load and also enter and exit the field weakening area. In addition to being simple and optimizable, its speed and position only change with commands. This feature has broad application prospects in drag control.
  • the present invention is based on the MTPA parameterless and position-sensing permanent magnet synchronous motor control method, which is characterized in that it contains a current control mode, which includes the following steps:
  • Step 1 Receive user input for given current Idq* and ⁇ angle, ⁇ angle is the angle between current vector Idq* and q axis, calculate Id_r* and Iq_r*, said Id_r* and Iq_r* are current vector Idq* In the rotor rotating coordinate system dq, the current value of the d-axis and the current value of the q-axis, the given current Idq* and ⁇ angle are data that conform to the MTPA mode;
  • Step 2 Find the corresponding ⁇ angle in the MTPA data table according to Id_r* and Iq_r*.
  • the ⁇ angle is the angle between the rotor coordinate system dq and the voltage coordinate system VdVq.
  • the MTPA data table refers to the maximum torque Data obtained in the mode per ampere;
  • Step 3 Use ⁇ angle, Id_r* and Iq_r* to calculate Id_Ref and Iq_Ref.
  • Id_Ref and Iq_Ref are the projections of the current vector Idq* on the Vd and Vq axes in the voltage coordinate system VdVq;
  • Step 4 Input the current Iq_Ref and the real-time feedback current Iq of the motor into the PLL phase-locked loop to obtain ⁇ v, ⁇ v is the angle between the voltage vector and the static coordinate system ⁇ , using the current Id_Ref and the real-time motor running feedback current Id.
  • Id_ref Id_r* ⁇ cos( ⁇ )+Iq_r* ⁇ sin( ⁇ )
  • Iq_ref -Id_r* ⁇ sin( ⁇ )+Iq_r* ⁇ cos( ⁇ ).
  • the above-mentioned MTPA data table is data obtained through experiments or data obtained through theoretical calculations or data obtained through computer finite element analysis software.
  • the permanent magnet synchronous motor control method based on MTPA without parameters and position sensing is characterized in that it contains a speed control mode, which includes the following steps:
  • Step 1 Receive the user input given speed spd command and ⁇ angle, and make the vector angle of voltage vector Vdq rotate at the given speed spd to obtain ⁇ v, ⁇ v is the angle between the voltage vector and the static coordinate system ⁇ , and the ⁇ angle is the current vector The angle between Idq and q axis;
  • Step 2 Calculate Id_r* and Iq_r* according to the feedback current Id and ⁇ angle of the motor running in real time.
  • the Id_r* and Iq_r* are the current value of the d-axis and the q-axis of the current current vector Idq in the rotor rotation coordinate system dq Id_r* and Iq_r* are data conforming to the MTPA mode.
  • Use Id_r* and Iq_r* to find the corresponding ⁇ angle in the MTPA data table.
  • ⁇ is the angle between the voltage vector Vdq and the current vector Idq in the MTPA mode, so that The current vector Idq coincides with the Vd axis to obtain the angle ⁇ i;
  • the above-mentioned MTPA data table is data obtained through experiments or data obtained through theoretical calculations or data obtained through computer finite element analysis software.
  • the above step 2 is to use the feedback current Iq of the motor real-time operation as one input of the PLL phase-locked loop, and set the other input Iq* of the PLL phase-locked loop to 0.
  • Iq* is the Vq axis of the current vector Idq in the VdVq coordinate system.
  • spd is the speed value
  • Pole_pair is the number of pole pairs of the motor
  • ⁇ t is the time variable
  • the PI regulator when the voltage value Vq is greater than or equal to the set threshold value Vmax, the PI regulator enters a saturated state, its output is limited to Vmax, and it automatically switches to the field weakening control mode.
  • MTPA_Angle_Lookup the main purpose is to obtain the angle ⁇ and angle ⁇ in Figure 13, see the relationship as follows:
  • the MTPA data sheet can be obtained through experiments.
  • the measured motor is 1/3HP. It is measured by a dynamometer in the motor laboratory.
  • the data of the maximum output torque is regarded as a data of the MTPA data sheet, as shown in Table 1.
  • the data of the maximum output torque includes ⁇ angle, Id_r*, Iq_r*, ⁇ angle, ⁇ angle and other data. In the same way, we can also measure the speed at
  • a set of data of the maximum output torque corresponding to different speeds such as 1400rpm, 1350rpm, 1300rpm...etc., and write them into the MTPA data table for easy searching.
  • Table 1 is for the speed control mode.
  • the test method is roughly the same.
  • the MTPA data sheet can be obtained through experiments.
  • the measured motor is 1/3HP.
  • the MTPA data table can also be obtained through the finite element analysis software of the computer, as shown in Figure 17 and Figure 18.
  • the relevant data is obtained through the computer finite element analysis, and the value of the coordinate Id_r* and the value of the coordinate Iq_r* are used in the figure. Get the ⁇ angle or ⁇ angle.
  • the data in the MTPA data sheet can also be obtained through theoretical calculations.
  • the permanent magnet synchronous motor control method based on MTPA without parameter and position sensing of the present invention no longer uses a magnetic flux observer to analyze the rotor position, thereby greatly reducing the calculation time of the CPU, and the positionless motor control becomes more simple and intuitive.
  • the current and speed control modes of the motor are simultaneously completed by two decoupling PI regulators, and the stability and dynamic response of the control are better than the multi-stage sleeve control loop; in the current and speed control mode of the present invention, the motor is optimized It is to make the current run along the calibratable MTPA track.
  • the motor has a full-load start function, and its operating range includes from no BEMF to field weakening control, with perfect functions; the present invention is based on the MTPA parameter-free and position-sensing permanent magnet synchronous motor control method, and its PLSL-MTPA mathematical model is no longer a single
  • the algorithm projects the motor current vector on the current and voltage coordinate system at the same time, and completes the positionless control by analyzing the angle of the vector.
  • the mathematical model is simple, the algorithm is simple, and the operation is simple, and does not take up a lot of control chips.
  • the PLSL-MTPA mathematical model is a position sensorless optimization without motor parameters Motor control technology, this technology solves the bottleneck problem that is highly dependent on motor resistance Rs, inductance Lq, Lq and magnetic flux ⁇ m in the implementation of positionless and optimized control of the motor.
  • the field weakening control of the traditional FOC control theory is: the control of the synchronous permanent magnet motor is mostly carried out in two intervals: the MTPA interval and the field weakening interval. As the speed increases, there will be a maximum torque or maximum current range outside the field weakening range, but it is rarely used in practice.
  • the FOC vector control of the motor is the control of the current Id and Iq. This is a control method with two degrees of freedom. How to control Id and Iq separately to make the motor work in the optimal state, so there is the MTPA theory. When entering the field weakening zone, the degree of freedom in the Id direction is locked, leaving only Iq proportional to the torque output. Motor control no longer needs to be optimized.
  • Id and field weakening It should be that the permanent magnet is embedded on the rotor, and the rotor magnetic field will induce a back electromotive force BEMF that offsets the stator voltage when the motor rotates, and the back electromotive force BEMF is proportional to the speed . When the speed is high to a certain degree, BEMF will be greater than the stator voltage, causing the motor to fail to work in an electric state.
  • the so-called field weakening control means that when the back electromotive force BEMF is high to a certain degree, let Id continue to increase in the negative direction, so as to generate a magnetic field specifically to weaken the rotor magnetic field.
  • the size of Id depends on the speed and motor load, but the ultimate goal is to make the back electromotive force BEMF less than the maximum stator voltage.
  • Iq can be obtained by the above formula.
  • Id and Iq are also limited by the voltage ellipse when running at high speed, and can run along the MTPA track at low speed.
  • the solution of Id is not given by the formula, but usually by the control logic (such as PI regulator, look-up table method, etc.).
  • the core of field weakening is to find Id so that the voltage does not overshoot, that is, to calculate Iq to meet the FOC theory and complete current control.
  • a permanent magnet synchronous motor control method based on MTPA without parameters and no position sensing that is, the PLPS-MTPA control method is very different in field weakening control.
  • This strategy fully satisfies the field weakening theory, but the realization method is greatly simplified, and the control is more stable.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

一种基于MTPA无参数无位置传感的永磁同步电机控制方法，其含有一种电流控制模式，包括步骤1：接收用户输入给定电流Idq*和γ角，计算出Id_r*和Iq_r*；步骤2：根据Id_r*和Iq_r*在MTPA数据表查找对应的α角；步骤3：利用α角、Id_r*和Iq_r*计算出Id_Ref和Iq_Ref；步骤4：通过对电流Iq_Ref和电机实时运行的反馈电流Iq输入到PLL锁相环中获取θv，利用对电流Id_Ref和电机实时运行的反馈电流Id的PI处理获取Vq，因为在MTPA模式下，Vd＝0，Vdq＝Vq，利用θv和Vdq可以获取Vα和Vβ，从而实现对电流的控制，该方法算法精简，运算简单，减少芯片资源占用，降低成本，解决了在对电机电阻Rs、电感Lq、Lq和磁通λm有高度依赖性的的瓶颈问题。

Description

基于MTPA无参数无位置传感的永磁同步电机控制方法 技术领域：
本发明涉及基于MTPA无参数无位置传感的永磁同步电机控制方法。
背景技术：
目前，无位置传感器矢量控制永磁同步电机的控制方法，一般有恒力矩控制模式、或恒转速控制模式，或恒风量控制模式三种。
例如美国专利US7525269公开了无位置传感器的3相同步电机电机矢量控制器，只公开了电流力矩控制模式，进行恒力矩控制。
中国专利CN103929109(A)也公开，基于无位置传感器矢量控制永磁同步电机的恒转速控制方法。
如图1所示，一般恒力矩控制的方框图如图1所示，由于力矩T只与q轴电流有关，根据力矩计算公式T＝K*iq0，给出力矩设定值T就换算成q轴的设定电流iq0，就可以利用q轴PI电流环进行闭环控制实现恒力矩的控制。
无位置传感器矢量控制永磁同步电机的控制方法最常规的FOC理论，大多都是建立在转子坐标系(rotor frame)上的。当转子位置未知时，无位置传感器算法就会被用来估算转子位置，从而FOC理论可以继续沿用。估算转子位置采用理论与电机相关参数，数学模型复杂，运算耗时复杂，占用大量的控制芯片(微处理器MCU)资源，对微处理器MCU要求较高，成本也高。另外，FOC理论对电机参数:电机电阻Rs、电感Lq、Lq和磁通λm有高度依赖性，造成方案适应范围较窄。
发明内容：
本发明的目的是提供基于MTPA无参数无位置传感的永磁同步电机控制方法，解决现有技术中常规FOC理论估算转子位置采用理论与电机相关参数密相关，数学模型复杂，运算耗时复杂，占用大量的控制芯片资源，对微处理器MCU要求较高，成本也高的技术问题。
本发明的目的是通过以下的技术方案予以实现的。
基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种电流控制模式，包括如下步骤：
步骤1：接收用户输入给定电流Idq*和γ角，γ角是电流矢量Idq*与q轴的夹角，计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是电流矢量Idq*投影在转子旋转坐标系dq中d轴的电流值和q轴的电流值，给定电流Idq*和γ角是符合MTPA模式的数据；
步骤2：根据Id_r*和Iq_r*在MTPA数据表查找对应的α角，所述的α角是转子坐标系dq与电压坐标系VdVq之间的夹角，所述的MTPA数据表是指最大力矩每安培的模式下获得的数据；
步骤3：利用α角、Id_r*和Iq_r*计算出Id_Ref和Iq_Ref,Id_Ref和Iq_Ref是电流矢量Idq*在电压坐标系VdVq中的投影；
步骤4：通过对电流Iq_Ref和电机实时运行的反馈电流Iq输入到PLL锁相环中获取θv，θv是电压矢量与静止坐标系αβ的夹角，利用对电流Id_Ref和电机实时运行的反馈电流Id的PI处理获取Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对电流的控制。
上述所述的Id_Ref和Iq_Ref是这样获得的：
Id_ref＝Id_r*×cos(α)+Iq_r*×sin(α)
Iq_ref＝-Id_r*×sin(α)+Iq_r*×cos(α)。
上述所述的步骤2的Id_r*和Iq_r*是这样计算得到：
Id_r*＝-Idq*×sin(γ)
Iq_r*＝Idq*×cos(γ)。
上述所述的MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
上述所述的电流控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其电压输出被限制在Vmax，Id不再受控，此状态即为弱磁控制方式。
基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种速度控制模式，包括如下步骤：
步骤1：接收用户输入给定速度spd指令和γ角，使电压矢量Vdq的矢量角按给定速度spd旋转，获得θv，θv是电压矢量与静止坐标系αβ的夹角，γ角是电流矢量Idq与q轴的夹角；
步骤2：根据电机实时运行的反馈电流Id和γ角计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是当前电流矢量Idq在转子旋转坐标系dq中d轴的电流值和q轴的电流值，Id_r*和Iq_r*是符合MTPA模式的数据，利用Id_r*和Iq_r*在MTPA数据表查找对应的β角，β是在MTPA模式下电压矢量Vdq与电流矢量Idq的夹角，使电流矢量Idq与Vd轴重合获取角度θi；
步骤3：获得θiv＝θv-θi,利用对角度β和θiv的PI处理获取电压 Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对转速的控制。
上述所述的MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
上述所述的步骤2是将电机实时运行的反馈电流Iq作为PLL锁相环一个输入，将PLL锁相环的另一个输入Iq*设置为0，Iq*是电流矢量Idq在VdVq坐标系下Vq轴的投影，PLL锁相环输出角度θi，θi使Iq*＝0是PLL锁相环解析Iα,Iβ所产生的角度。
上述所述的θv是这样获得：
θν＝∫spd×(pole_pair×360×Δt÷60)·dt
其中：spd是速度值，Pole_pair为电机极对数，Δt是时间变量。
在速度控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其输出被限制在Vmax，自动转换入弱磁控制方式。
本发明的与现有技术相比具有的有益效果是：
1)本发明的基于MTPA无参数无位置传感的永磁同步电机控制方法不再通过用磁通观测器来解析转子位置，从而大大降低了CPU的计算时间，无位置电机控制变得更加简捷直观，电机的电流和速度控制模式同时由两路解耦的PI调节器完成，控制的稳定性和动态响应均优于多级坎套式控制回路；
2)本发明在电流和速度控制模式中，电机的优化是使其电流沿可标定的MTPA轨迹运行。电机具有满载启动功能，其运行区间包括从无BEMF至弱磁控制，功能完善。
3)本发明基于MTPA无参数无位置传感的永磁同步电机控制方法，其 PLSL-MTPA数学模型不再以单一的转子坐标系为前提，该算法将电机电流矢量同时投影在电流和电压坐标系上，以解析矢量夹角的方式完成无位置控制，数学模型简单，算法简单，运算简单，不占用大量的控制芯片资源，对微处理器MCU要求不高，有利于降低成本。
4)基于MTPA无参数无位置传感的永磁同步电机控制方法，其PLSL-MTPA数学模型是一种无需电机参数的无位置传感器的优化电机控制技术，该技术解决了在对电机实行对无位置和优化控制中，对电机电阻Rs、电感Lq、Lq和磁通λm有高度依赖性的的瓶颈问题。
附图说明：
图1是传统无位置传感器矢量控制永磁同步电机FOC控制方框图。
图2是本发明永磁同步电机的立体图；
图3是本发明永磁同步电机的电机控制器的立体图；
图4是本发明永磁同步电机的剖视图；
图5是本发明永磁同步电机的电机控制器的原理方框图；
图6是图5对应的电路图；
图7是三相永磁同步电机的静止坐标系ABC的示意图；
图8是三相永磁同步电机的静止的正交坐标系αβ的示意图；
图9是三相永磁同步电机矢量控制的各坐标系关系图；
图10是本发明的的正交坐标系αβ与转子坐标系dq的关系图；
图11是本发明的θv、θi，θr是三个同频率的时域变量图；
图12是本发明的标示电压矢量和电流矢量的角度关系图；
图13是本发明的MTPA模式下的原理示意图；
图14是本发明电流控制模式下的原理示意图；
图15是本发明的电流控制模式下的原理方框图；
图16是本发明的速度控制模式下的原理方框图；
图17是本发明的α角与Id_r*、Iq_r*对应的分析数据图；
图18是本发明的β角与Id_r*、Iq_r*对应的分析数据图。
具体实施方式：
下面通过具体实施例并结合附图对本发明作进一步详细的描述。
如图2、图3、图4所示，举例：假设本发明是一种三相永磁同步电机,由电机控制器2和电机单体1,所述的电机单体1包括定子组件12、转子组件13和机壳组件11，定子组件13安装在机壳组件11上，转子组件13套装在定子组件12的内侧或者外侧组成，电机控制器2包括控制盒22和安装在控制盒22里面的控制线路板21，控制线路板21一般包括电源电路、微处理器、母线电压检测电路、逆变器，电源电路为各部分电路供电，母线电压检测电路将直流母线电压Uabc输入到微处理器，微处理器控制逆变器，逆变器控制定子组件12的各相线圈绕组的通断电。
如图5、图6所示，假设3相无刷直流永磁同步电机的相线电流检测电路将各相的电流Ia、Ib、Ic输入到微处理器。交流输入(AC INPUT)经过由二级管D7、D8、D9、D10组成的全波整流电路后，在电容C1的一端输出直流母线电压Vdc bus,直流母线电压Vdc bus与输入交流电压有关，微处理器输入到逆变器的PWM信号，逆变器由电子开关管Q1、Q2、Q3、Q4、Q5、Q6组成，电子开关管Q1、Q2、Q3、Q4、Q5、Q6的控制端分别由微处理器输出的6路PWM信号(P1、P2、P3、P4、P5、P6)控制。
如图7所示，三相电机电流为Ia,Ib,Ic.在时域中有120度的相位角， 一般称为静止坐标系。这三个时域电流量可以简化为两个正交的电流量Iα、Iβ，如图8所示。由三角矢量图来描述，见图9所示。其数学关系式为：
Figure PCTCN2019114919-appb-000001
Figure PCTCN2019114919-appb-000002
这就是常用的克拉克变换(Clarke Transformation)，Iα,Iβ是随时间变化的正旋电流。如果我们站在一个同频率的旋转平台去观测Iα,Iβ，引入转子旋转坐标系dq，其正旋特性就会被销除，只由相位特性得到保留。建立这个旋转平台就是常用的PARK变换(Park Transformation:)。
I d＝I α*cos(θ)+I β*sin(θ)
I q＝I β*cos(θ)+I α*sin(θ)
                                     -------(公式2)
如图10所示，在PARK变换中我们引入了变量θ。θ即为我们观测Iα,Iβ的观测方位角。当然从不同的观测方位去解析Iα,Iβ，其投影在不同的旋转平台(也就是坐标系)上的投影是不同的。一但选好了一个观测角，Park变换就将正旋Iα,Iβ变成直流Id,Iq。在无位置传感器的永磁同步电机中，该如何选择θ角成为最重要的问题，在电机控制中通常把转子磁场的北极定位0度。转子转一圈为360度,转子位置与极对数的关系如下：
θr＝Pole_pair×θ0，其中θ0为转子机械角度，Pole_pair为电机极对数。
在绝大多数无位置传感器的永磁同步电机控制中，其转子机械角度当然是未知的。算法的中心任务就是估算θr。一但估算出θr，电机的矢量控制算法就立即可实现。但传统的FOC理论的估算θr的算法极为复杂，运算时间长，数学模型复杂，高度依赖电机的参数。
本发明基于MTPA无参数无位置传感的永磁同步电机控制方法【简称PLSL-MTPA算法(parameterless sensorless--MTPA)】，在PLSL-MTPA中我们用另外的方法来对Iα,Iβ进行解析。也就是说用不同的θ角来建立旋转平台来分析Iα,Iβ。在此，我们引入VdVq电压坐标系并用了锁相环逻辑，锁相环的工作原理正好与Park变换相反。后者是通过用角度将正旋量变成直流量。锁相环则是通过锁定直流量其中的一个，找出建立旋转平台的θ角来。
本发明引入θv，θi分别代表电压矢量和电流矢量与静止坐标系ABC的夹角，见图12所示。
在速度控制模式下θv是由开环速度积分得来的。它用来产生Vα,Vβ,继而生成Va,Vb,Vc.。θi锁相环解析Iα,Iβ使Iq*＝0所产生的角度。
根据上述公式2和(Id,0)＝Park转换(Iα,Iβ)byθi满足：
Id＝Iα×Cos(θi)+Iβ×sin(θi)
0＝Iβ×Cos(θi)－Iα×sin(θi)
在同步电机的理论中，θv、θi，θr是三个同频率的时域变量。以A相为例,当Va(t)超前Ia(t)β时，见图12所示：
Va(t)＝Vabc×cos(θv)＝Vabc×cos(wt+β)
Ia(t)＝Iabc×cos(θi)＝Iabc×cos(wt)
其中Vabc是A相、B相、C相绕组合成的电压矢量，IabcA相、B相、C相绕组合成的电流矢量，w是角速度，t是时间，β是A相的电压矢量Va(t)与电流矢量Ia(t)的夹角。
θv,θi，θr三者间的关系为：
β＝θv-θi-------公式3
θr＝θv-α-----公式4
θv,θi，θr是电压电流在时域上的关系见图11所示，PLSL-MTPA算法的核心就是依照上述公式3永远让θv和θi保持一个给定的β角——形成速度模式，或依照上述公式4将Iα,Iβ在电压坐标系(即VdVq坐标系)上的投影满足α角——形成电流模式。因为PLSL-MTPA始终让Vd＝0,Vdq＝Vq，形成电压，电流和转子间的矢量图见图13所示，图13是简化的PLSL-MTPA矢量图。
在图13中，可以得到角度α＝β+γ，其中，角度α是在MTPA模式下转子坐标系dq与电压坐标系VdVq之间的夹角，角度β是在MTPA模式下电压矢量Vdq与电流矢量Idq的夹角，γ是电流矢量dq与q轴的夹角。
在图14、图15中，在电压坐标系-VdVq坐标系中，始终让Vd＝0,Vdq＝Vq，在电流控制模式中，将Iα,Iβ的电流矢量Idq在q轴上的投影与电流矢量Idq在VdVq坐标系上的Vq轴的投影进行PLL锁相环处理就能获得θv，将Iα,Iβ的电流矢量Idq在d轴上的投影与电流矢量Idq在VdVq坐标系上的Vd轴的投影进行PI处理获得Vq，由于Vd＝0,Vdq＝Vq，那么利用θv 和Vdq＝Vq就可将直流量转成正旋量Vα,Vβ。
如图14、图16所示，是本发明的PLSL-MTPA的另一种控制模式，即速度控制模式，其原理是：使电压矢量Vdq角度按给定速度指令spd旋转,即开环控制运行，
θν＝∫spd×(pole_pair×360×Δt÷60)·dt
其中spd是速度值，Pole_pair为电机极对数，Δt是时间变量；
PLL锁相器使Idq与Vd轴重合(即用Iq＝0)生成θi，得到角度差θiv＝θv-θi
电压矢量Vdq＝PI(θiv-β),由于获得θv和Vdq就可将正旋量变成直流量获得Vα,Vβ。
上述两种控制模式的电流PI器利用角度或电流误差控制获得Vdq,其中：速度模式:
Δ＝θiv-β
电流模式：
Δ＝I d_ref-I d
而电压矢量Vdq＝PI(Δ)，Vabc由θv和Vdq生成。
这两种模式的θv和Vdq的计算方式是不同的。
本发明的PLSL-MTPA静态满载起动:用最大的电流Idq_Max启动,在低速时由于电机呈阻性状态,电流和电压同相位，这时电流PI控制器趋于最大电流饱和，Idq_Max用于驱动电机，当转速开始上升，反电动势也随之增强。这也导致电流电压差角β渐不为0。从而使电流PI控制器进入正常工作范围，Vdq和Idq差角随实际负载变化，电机电流也随之响应。起动能 力取决于目标转速，速度递增斜率，电流PI增益，及最大电流限制。
常规的FOC理论大多都是建立在转子坐标系(rotor frame)上的。当转子位置未知时，无位置传感器算法就会被用来估算转子位置，从而FOC理论可以继续沿用,本发明的PLSL-MTPA控制方法与常规FOC的不同点就在于前者从一开始就脱离了对转子坐标系的依赖性，转而采用了与电机参数无关的角度转换方案并对电流电压相位角进行调节，以实现对电机的同步控制。该设计理念大大简化了无位置传感器电机控制过程。
本发明的PLSL-MTPA控制方法符合4种控制定律：
控制定律1:
在同步运行条件下，在电流矢量和电压矢量之间维持可调控夹角β，可以实现对同步电机的全面控制。该β角也就是通常所称的功率因数角。
控制定律2:
在同步运行条件下，同步电机的转子位置可以由电压矢量加一个可控夹角α来确定。该α角可能超前或滞后电压矢量。基于已给定的α角，对同步电机实施一个电压矢量，其同步条件可以持续，即同步电机可控。
控制定律3:
单位电流最大力矩MTPA控制可以将电流指令转换成α,β角度来完成。其α,β角度的生成均遵循MTPA原理，并被控制定律1和控制定律2用于对同步电机的控制。
控制定律4:
仅当电流控制指令沿MTPA轨迹,且延用MTPA准则将其转换成α,β角，由控制定律2估算的转子位置为实际电机转子位置。此时电机产生的力矩 为MTPA力矩.
控制定律5:
当电机在小电流运行条件下，电压与转速的比值可近似为BEMF。同步电机运行状态可以通过将该值与阀值的比较来确定。
PLSL-MTPA的速度模式实际上是一个能带负载也能进出弱磁区的速度开环控制。除了简单可优化外，其转速和位置仅随指令变化。这种特性在拖动控制中有着广泛的应用前景。
具体实施例一：
如图14、图15所示，本发明基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种电流控制模式，包括如下步骤：
步骤1：接收用户输入给定电流Idq*和γ角，γ角是电流矢量Idq*与q轴的夹角，计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是电流矢量Idq*在转子旋转坐标系dq中d轴的电流值和q轴的电流值，给定电流Idq*和γ角是符合MTPA模式的数据；
步骤2：根据Id_r*和Iq_r*在MTPA数据表查找对应的α角，所述的α角是转子坐标系dq与电压坐标系VdVq之间的夹角，所述的MTPA数据表是指最大力矩每安培的模式下获得的数据；
步骤3：利用α角、Id_r*和Iq_r*计算出Id_Ref和Iq_Ref,Id_Ref和Iq_Ref是电流矢量Idq*在电压坐标系VdVq中Vd轴和Vq轴的投影；
步骤4：通过对电流Iq_Ref和电机实时运行的反馈电流Iq输入到PLL 锁相环中获取θv，θv是电压矢量与静止坐标系αβ的夹角，利用对电流Id_Ref和电机实时运行的反馈电流Id的PI处理获取Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对电流的控制。
上述的Id_Ref和Iq_Ref是这样获得的：
Id_ref＝Id_r*×cos(α)+Iq_r*×sin(α)
Iq_ref＝-Id_r*×sin(α)+Iq_r*×cos(α)。
上述的步骤2的Id_r*和Iq_r*是这样计算得到：
Id_r*＝-Idq*×sin(γ)
Iq_r*＝Idq*×cos(γ)。
上述的电流控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其电压输出被限制在Vmax，Id不再受控，此状态即为弱磁控制方式。
上述的MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
实施例二：
如图14、图16所示，基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种速度控制模式，包括如下步骤：
步骤1：接收用户输入给定速度spd指令和γ角，使电压矢量Vdq的矢量角按给定速度spd旋转，获得θv，θv是电压矢量与静止坐标系αβ的 夹角，γ角是电流矢量Idq与q轴的夹角；
步骤2：根据电机实时运行的反馈电流Id和γ角计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是当前电流矢量Idq在转子旋转坐标系dq中d轴的电流值和q轴的电流值，Id_r*和Iq_r*是符合MTPA模式的数据，利用Id_r*和Iq_r*在MTPA数据表查找对应的β角，β是在MTPA模式下电压矢量Vdq与电流矢量Idq的夹角，使电流矢量Idq与Vd轴重合获取角度θi；
步骤3：获得θiv＝θv-θi,利用对角度β和θiv的PI处理获取电压Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对转速的控制。
上述的MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
上述的步骤2是将电机实时运行的反馈电流Iq作为PLL锁相环一个输入，将PLL锁相环的另一个输入Iq*设置为0，Iq*是电流矢量Idq在VdVq坐标系下Vq轴的投影，PLL锁相环输出角度θi，θi使Iq*＝0是PLL锁相环解析Iα,Iβ所产生的角度。
上述的θv是这样获得：
θν＝∫spd×(pole_pair×360×Δt÷60)·dt
其中：spd是速度值，Pole_pair为电机极对数，Δt是时间变量。
上述在速度控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其输出被限制在Vmax，自动转换入弱磁控制方式。
MTPA数据表或者称为MTPA查找表(MTPA_Angle_Lookup),主要目的是 获得图13中的α角和β角，见关系式如下：
Figure PCTCN2019114919-appb-000003
即通过Id_r*和Iq_r*利用MTPA数据表查表获取α角和β角。
MTPA数据表可以通过实验来获得，实测电机为1/3HP，在电机试验室利用测功机进行测量，电流夹角γ测试步骤，设定电机转速＝1450rpm，设定测功机力矩间隔:10,15,20,26,31oz-in搜索最高***效率或最大MTPA_Index，对每个测功机力矩挡位，记录Idq和γ角值:验证测试结果，从五组测试数据中针对电机转速＝1450rpm，最大输出力矩的一组数据作为MTPA数据表一项数据，见表1所示，在最大输出力矩的一组数据中有γ角、Id_r*、Iq_r*、α角、β角等数据。同理，我们也可以测量转速在
转速 力矩 γ角 Id_r* Iq_r* α角 β角
1450rpm 10.46 1728 500 1652 3391 13356
1450rpm 15.98 2640 1240 2344 6592 11308
1450rpm 20.94 3500 1900 2964 8640 10456
1450rpm 26.46 4483 2608 3672 10177 9988
1450rpm 31.64 5498 3356 4388 11328 9604
1400rpm 。。。 。。。 。。。 。。。 。。。 。。。
1400rpm 。。。 。。。 。。。 。。。 。。。 。。。
1400rpm 。。。 。。。 。。。 。。。 。。。 。。。
1400rpm 。。。 。。。 。。。 。。。 。。。 。。。
1400rpm 。。。 。。。 。。。 。。。 。。。 。。。
             
表1
1400rpm、1350rpm、1300rpm....等不同转速对应的最大输出力矩的一组数据，并写入到MTPA数据表中，以便查找。表1是针对转速控制模式。
对于电流控制模式，其测试的方法是大致相同，MTPA数据表可以通过实验来获得，实测电机为1/3HP，在电机试验室利用测功机进行测量，电流夹角γ测试步骤，设定电流Idq＝3.2A，设定测功机力矩间隔:12,18,24,28,33oz-in搜索最高***效率或最大MTPA_Index，对每个测功机力矩挡位，记录Idq和γ角值:验证测试结果，从五组测试数据中针对Idq＝3.2A，最大输出力矩的一组数据作为MTPA数据表一项数据，见表2所示。
电流 力矩 γ角 Id_r* Iq_r* α角 β角
3.2A 12 1455 650 2253 2351 13450
3.2A 18 2228 1440 2698 4323 12011
3.2A 24 3400 1800 3298 6543 10244
3.2A 28 4578 2708 3789 8678 951
3.2A 33 5677 3676 4565 10233 8867
。。。 。。。 。。。 。。。 。。。 。。。 。。。
。。。 。。。 。。。 。。。 。。。 。。。 。。。
。。。 。。。 。。。 。。。 。。。 。。。 。。。
。。。 。。。 。。。 。。。 。。。 。。。 。。。
             
表2
另外，对于MTPA数据表也可以通过计算机的有限元分析软件获取，见图17和图18所示，通过计算机有限元分析获取相关的数据，图中通过坐标Id_r*的值和坐标Iq_r*的值获取α角或β角。
当然，对于MTPA数据表的数据也可以通过理论计算获取数据。
本发明的基于MTPA无参数无位置传感的永磁同步电机控制方法不再通过用磁通观测器来解析转子位置，从而大大降低了CPU的计算时间，无位置电机控制变得更加简捷直观，电机的电流和速度控制模式同时由两路解耦的PI调节器完成，控制的稳定性和动态响应均优于多级坎套式控制回路；本发明在电流和速度控制模式中，电机的优化是使其电流沿可标定的MTPA轨迹运行。电机具有满载启动功能，其运行区间包括从无BEMF至弱磁控制，功能完善；本发明基于MTPA无参数无位置传感的永磁同步电机控制方法，其PLSL-MTPA数学模型不再以单一的转子坐标系为前提，该算法将电机电流矢量同时投影在电流和电压坐标系上，以解析矢量夹角的方式完成无位置控制，数学模型简单，算法简单，运算简单，不占用大量的控制芯片资源，对微处理器MCU要求不高，有利于降低成本；基于MTPA无参数无位置传感的永磁同步电机控制方法，其PLSL-MTPA数学模型是一种无需电机参数的无位置传感器的优化电机控制技术，该技术解决了在对电机实行对无位置和优化控制中，对电机电阻Rs、电感Lq、Lq和磁通λm有高度依赖性的的瓶颈问题。
关于本发明基于MTPA无参数无位置传感的永磁同步电机控制方法中的弱磁控制与传统的FOC控制理论的弱磁控制的对比：
传统的FOC控制理论的弱磁控制是：对同步永磁电机的控制，多数是在两个区间内进行的：MTPA区间和弱磁区间。随着转速的增高在弱磁区间之 外还会有最大力矩或最大电流区间，但在实际中很少用到。电机的FOC矢量控制就是对电流Id,Iq的控制。这是一个具有两维自由度的控制手段。如何分别控制Id和Iq从而让电机工作在最优状态，于是就有了MTPA理论。当进入弱磁区后，Id方向的自由度被锁定，仅剩下Iq正比于力矩输出。电机控制再无需优化。Id和弱磁的关系:应为永磁体是嵌在转子上的，转子磁场在电机旋转时会感应出一个与定子电压相抵消的反向电动势BEMF，而反向电动势BEMF是与转速成正比的。当转速高到一定成度时，BEMF会大过定子电压，造成电机无法工作在电动状态。所谓弱磁控制就是当反向电动势BEMF高到一定成度时，让Id继续往负方向增加,以此产生一个磁场专门削弱转子磁场。Id的大小取决于转速和电机负载，但最终目标是让反向电动势BEMF小于最大定子电压。
经典的弱磁控制理论：电机电磁方程：
v ds＝r si ds-ω rλ qs＝r si ds-ω rL qi qs
v qs＝r si qs+ω rλ ds＝r si qs+ω r(L di ds+λ PM)
由于受到电压限制，Id,Iq满足椭圆关系：
Figure PCTCN2019114919-appb-000004
在弱磁条件下
Figure PCTCN2019114919-appb-000005
Iq的解是唯一的：
Figure PCTCN2019114919-appb-000006
一但有了Id,Iq就可由上式求出。
从实际测试模拟图象上看，Id,Iq在高速运行时还受到电压椭圆的限制，低速时可沿MTPA轨迹运行。Id的求解不是由公式给出的，通常是由控制逻辑(如PI调节器，查表法等)。总之，弱磁的核心就是找到Id使电压不超调，即而算出Iq,满足FOC理论，完成电流控制。
关于本发明基于MTPA无参数无位置传感的永磁同步电机控制方法,即PLPS-MTPA控制方法是在弱磁控制上有很大不同。弱磁控制不再是以找Id为核心，而是直接控制电压，只要把Vdq(Vdq＝Vq,Vd＝0)限制在直流母线电压Vdc_Bus以内，由此而产生的Id完全由逆变器吸收，继续控制电流电压相位角产生所需力矩。该策略完全满足弱磁理论，但实现手段大大简化，而且控制更平稳。
以上实施例为本发明的较佳实施方式，但本发明的实施方式不限于此，其他任何未背离本发明的精神实质与原理下所作的改变、修饰、替代、组合、简化，均为等效的置换方式，都包含在本发明的保护范围之内。

Claims (10)

  1. 基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种电流控制模式，包括如下步骤：
    步骤1：接收用户输入给定电流Idq*和γ角，γ角是电流矢量Idq*与q轴的夹角，计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是电流矢量Idq*在转子旋转坐标系dq中d轴的电流值和q轴的电流值，给定电流Idq*和γ角是符合MTPA模式的数据；
    步骤2：根据Id_r*和Iq_r*在MTPA数据表查找对应的α角，所述的α角是转子坐标系dq与电压坐标系VdVq之间的夹角，所述的MTPA数据表是指最大力矩每安培的模式下获得的数据；
    步骤3：利用α角、Id_r*和Iq_r*计算出Id_Ref和Iq_Ref,Id_Ref和Iq_Ref是电流矢量Idq*在电压坐标系VdVq中Vd轴和Vq轴的投影；
    步骤4：通过对电流Iq_Ref和电机实时运行的反馈电流Iq输入到PLL锁相环中获取θv，θv是电压矢量与静止坐标系αβ的夹角；利用对电流Id_Ref和电机实时运行的反馈电流Id的PI处理获取Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对电流的控制。
  2. 根据权利要求1所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：Id_Ref和Iq_Ref是这样获得的：
    Id_ref＝Id_r*×cos(α)+Iq_r*×sin(α)
    Iq_ref＝-Id_r*×sin(α)+Iq_r*×cos(α)。
  3. 根据权利要求1或2所述的基于MTPA无参数无位置传感的永磁同步 电机控制方法，其特征在于：步骤1的Id_r*和Iq_r*是这样计算得到：
    Id_r*＝-Idq*×sin(γ)
    Iq_r*＝Idq*×cos(γ)。
  4. 根据权利要求1或2或3所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
  5. 根据权利要求4所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：电流控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其电压输出被限制在Vmax，Id不再受控，此状态即为弱磁控制方式。
  6. 基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：它含有一种速度控制模式，包括如下步骤：
    步骤1：接收用户输入给定速度spd指令和γ角，使电压矢量Vdq的矢量角按给定速度spd旋转，获得θv，θv是电压矢量与静止坐标系αβ的夹角，γ角是电流矢量Idq与q轴的夹角；
    步骤2：根据电机实时运行的反馈电流Id和γ角计算出Id_r*和Iq_r*，所述的Id_r*和Iq_r*是当前电流矢量Idq在转子旋转坐标系dq中d轴的电流值和q轴的电流值，Id_r*和Iq_r*是符合MTPA模式的数据，利用Id_r*和Iq_r*在MTPA数据表查找对应的β角，β是在MTPA模式下电压矢量Vdq与电流矢量Idq的夹角；使电流矢量Idq与Vd轴重合获取角度θi；
    步骤3：获得θiv＝θv-θi,利用对角度β和θiv的PI处理获取电压Vq,因为在MTPA模式下，Vd＝0,Vdq＝Vq,利用θv和Vdq可以获取Vα和Vβ，从而实现对转速的控制。
  7. 根据权利要求6所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：MTPA数据表是通过实验获取的数据或者通过理论计算获取的数据或者是通过计算机的有限元分析软件获取的数据。
  8. 根据权利要求7所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：步骤2是将电机实时运行的反馈电流Iq作为PLL锁相环一个输入，将PLL锁相环的另一个输入Iq*设置为0，Iq*是电流矢量Idq在VdVq坐标系下Vq轴的投影，PLL锁相环输出角度θi，θi使Iq*＝0是PLL锁相环解析Iα,Iβ所产生的角度。
  9. 根据权利要求8所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：θv是这样获得：
    θν＝∫spd×(pole_pair×360×△t÷60)·dt
    其中：spd是速度值，Pole_pair为电机极对数，Δt是时间变量。
  10. 根据权利要求9所述的基于MTPA无参数无位置传感的永磁同步电机控制方法，其特征在于：在速度控制模式中，当电压值Vq大于或等于设定的阀值Vmax时，PI调节器进入饱和状态，其输出被限制在Vmax，自动转换入弱磁控制方式。
PCT/CN2019/114919 2019-10-28 2019-11-01 基于mtpa无参数无位置传感的永磁同步电机控制方法 WO2021081968A1 (zh)

Priority Applications (3)

Application Number Priority Date Filing Date Title
CA3154822A CA3154822A1 (en) 2019-10-28 2019-11-01 Parameterless position-sensorless permanent magnet synchronous motor control method based on mtpa
MX2022005162A MX2022005162A (es) 2019-10-28 2019-11-01 Metodo de control de motor sincrono de imanes permanentes sin sensor de posicion sin parametros basado en mtpa.
US17/686,391 US11689132B2 (en) 2019-10-28 2022-03-03 MTPA based method for parameterless and position-sensorless control of a permanent magnet synchronous motor

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN201911028408.9A CN110768601B (zh) 2019-10-28 2019-10-28 基于mtpa无参数无位置传感的永磁同步电机控制方法
CN201911028408.9 2019-10-28

Related Child Applications (1)

Application Number Title Priority Date Filing Date
US17/686,391 Continuation-In-Part US11689132B2 (en) 2019-10-28 2022-03-03 MTPA based method for parameterless and position-sensorless control of a permanent magnet synchronous motor

Publications (1)

Publication Number Publication Date
WO2021081968A1 true WO2021081968A1 (zh) 2021-05-06

Family

ID=69334118

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2019/114919 WO2021081968A1 (zh) 2019-10-28 2019-11-01 基于mtpa无参数无位置传感的永磁同步电机控制方法

Country Status (5)

Country Link
US (1) US11689132B2 (zh)
CN (1) CN110768601B (zh)
CA (1) CA3154822A1 (zh)
MX (1) MX2022005162A (zh)
WO (1) WO2021081968A1 (zh)

Families Citing this family (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111538254B (zh) * 2020-05-11 2023-04-07 成都华川电装有限责任公司 一种获取永磁同步电机性能map表的仿真电路
CN112865636A (zh) * 2021-01-15 2021-05-28 珠海格力电器股份有限公司 一种最大转矩电流比控制方法及装置
US11936313B2 (en) * 2021-08-31 2024-03-19 Kinetic Technologies International Holdings Lp Method of aligning a rotor of a synchronous motor at a specified rotor angle and a controller therefor

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102594250A (zh) * 2012-02-17 2012-07-18 哈尔滨工业大学 无位置传感器内置式永磁同步电机最大转矩电流比矢量控制***及控制方法
EP2544362A2 (en) * 2011-07-05 2013-01-09 LSIS Co., Ltd. Apparatus for operating an interior permanent magnet synchronous motor
CN103607155A (zh) * 2013-10-28 2014-02-26 浙江大学 基于旋转电流矢量的永磁同步电机无位置传感器控制方法
CN103929109A (zh) * 2014-05-12 2014-07-16 哈尔滨工业大学 一种大功率内置式永磁同步电机无位置传感器控制***及控制方法
CN109889117A (zh) * 2019-04-04 2019-06-14 合肥工业大学 基于旋转高频注入法的ipmsm位置观测方法、***及驱动***
CN110071674A (zh) * 2019-05-14 2019-07-30 天津大学 一种无位置传感器永磁同步电机最大转矩电流比控制方法

Family Cites Families (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2003081110A (ja) * 2001-09-12 2003-03-19 Hitachi Unisia Automotive Ltd 電動パワーステアリング装置の制御装置
CN101110559B (zh) * 2006-07-20 2011-10-12 比亚迪股份有限公司 一种混合动力车用永磁同步发电机的控制***
US8339094B2 (en) * 2010-03-11 2012-12-25 GM Global Technology Operations LLC Methods, systems and apparatus for overmodulation of a five-phase machine
TWI439041B (zh) * 2011-12-19 2014-05-21 Ind Tech Res Inst 永磁同步馬達驅動方法與裝置
US8829827B2 (en) * 2011-12-28 2014-09-09 Gm Global Technology Operations, Llc Methods, systems and apparatus for controlling operation of an electric machine in an overmodulation region
US9966889B2 (en) * 2013-05-12 2018-05-08 Infineon Technologies Ag Optimized control for synchronous motors
KR101535036B1 (ko) * 2014-08-25 2015-07-24 현대자동차주식회사 구동모터의 전류지령에 대한 토크 보상장치 및 방법

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP2544362A2 (en) * 2011-07-05 2013-01-09 LSIS Co., Ltd. Apparatus for operating an interior permanent magnet synchronous motor
CN102594250A (zh) * 2012-02-17 2012-07-18 哈尔滨工业大学 无位置传感器内置式永磁同步电机最大转矩电流比矢量控制***及控制方法
CN103607155A (zh) * 2013-10-28 2014-02-26 浙江大学 基于旋转电流矢量的永磁同步电机无位置传感器控制方法
CN103929109A (zh) * 2014-05-12 2014-07-16 哈尔滨工业大学 一种大功率内置式永磁同步电机无位置传感器控制***及控制方法
CN109889117A (zh) * 2019-04-04 2019-06-14 合肥工业大学 基于旋转高频注入法的ipmsm位置观测方法、***及驱动***
CN110071674A (zh) * 2019-05-14 2019-07-30 天津大学 一种无位置传感器永磁同步电机最大转矩电流比控制方法

Also Published As

Publication number Publication date
MX2022005162A (es) 2022-06-08
CA3154822A1 (en) 2021-05-06
CN110768601B (zh) 2021-04-06
US11689132B2 (en) 2023-06-27
US20220190762A1 (en) 2022-06-16
CN110768601A (zh) 2020-02-07

Similar Documents

Publication Publication Date Title
Bierhoff A general PLL-type algorithm for speed sensorless control of electrical drives
CN108599651B (zh) 基于虚拟电压注入的感应电机无速度传感器驱动控制方法
WO2021081968A1 (zh) 基于mtpa无参数无位置传感的永磁同步电机控制方法
JP4531751B2 (ja) 同期機制御装置
Wang et al. An EMF observer for PMSM sensorless drives adaptive to stator resistance and rotor flux linkage
Tang et al. IPMSMs sensorless MTPA control based on virtual q-axis inductance by using virtual high-frequency signal injection
US7944163B2 (en) Sensorless controlling apparatus of brushless motor
CN111786607A (zh) 一种可靠平顺基于无位置传感器永磁同步电动机起动方法
JP2008043030A (ja) 電動機の制御装置
JP2002095300A (ja) 永久磁石同期電動機の制御方法
Lee et al. Reactive-power-based robust MTPA control for v/f scalar-controlled induction motor drives
Xing et al. Research on startup process for sensorless control of PMSMs based on IF method combined with an adaptive compensator
Yu et al. A Stator Current Oriented Closed-Loop $ I\text {--} f $ Control of Sensorless SPMSM With Fully Unknown Parameters for Reverse Rotation Prevention
JP2004297966A (ja) 交流電動機の制御装置
JP5648310B2 (ja) 同期モータの制御装置、及び同期モータの制御方法
CN112701969B (zh) 一种同步磁阻电机最大转矩电流比在线寻优方法
CN112039392B (zh) 电机控制方法、装置、设备及计算机可读介质
WO2021215083A1 (ja) 電動工具システム、制御方法、及びプログラム
CN114337426A (zh) 一种d-q轴静止坐标系下永磁同步电机偏差解耦控制方法
CN111431455A (zh) 基于最大转矩电流比控制的永磁同步电主轴矢量控制方法
Ding et al. Starting strategy and self-tuning technology of permanent magnet linear synchronous motor (PMLSM) based on Hall sensor
Yao et al. An Improved Adaptive Full-Order Sliding-Mode Observer for Sensorless Control of High-Speed Permanent-Magnet Synchronous Motor
WO2023134379A1 (zh) 用于永磁同步电机的控制方法和控制模块
TWI760228B (zh) 馬達控制方法
Zhang et al. An adaptive sliding mode observer for sensorless vector control of surface-mounted PMSM

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 19950623

Country of ref document: EP

Kind code of ref document: A1

ENP Entry into the national phase

Ref document number: 3154822

Country of ref document: CA

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 19950623

Country of ref document: EP

Kind code of ref document: A1